LT1228_12_技术文档

LT1228

100MHz Current Feedback

Amplifier with DC Gain Control

FEATURES

DESCRIPTION

The LT^®1228 makes it easy to electronically control the

gain of signals from DC to video frequencies. The LT1228

implements gain control with a transconductance ampli-

fier (voltage to current) whose gain is proportional to an

externally controlled current. A resistor is typically used

to convert the output current to a voltage, which is then

amplified with a current feedback amplifier. The LT1228

combinesbothamplifiersintoan8-pinpackage, andoper-

ates on any supply voltage from 4V ( 2V) to 30V ( 1ꢀV).

A complete differential input, gain controlled amplifier can

be implemented with the LT1228 and just a few resistors.

n

Very Fast Transconductance Amplifier

Bandwidth: 7ꢀMHz

g = 10 × I

m

SET

Low THD: 0.2% at 30mV

Input

RMS

Wide I Range: 1µA to 1mA

SET

n

Very Fast Current Feedback Amplifier

Bandwidth: 100MHz

Slew Rate: 1000V/µs

Output Drive Current: 30mA

Differential Gain: 0.04%

Differential Phase: 0.1°

High Input Impedance: 2ꢀMΩ, 6pF

Wide Supply Range: 2V to 1ꢀV

Inputs Common Mode to Within 1.ꢀV of Supplies

Outputs Swing Within 0.8V of Supplies

Supply Current: 7mA

TheLT1228transconductanceamplifierhasahighimped-

ancedifferentialinputandacurrentsourceoutputwithwide

n

output voltage compliance. The transconductance, g , is

m

setbythecurrentthatflowsintoPinꢀ,I .Thesmallsignal

SET

g isequaltotentimesthevalueofI andthisrelationship

m

SET

Available in 8-Lead PDIP and SO Packages

holds over several decades of set current. The voltage at

Pin ꢀ is two diode drops above the negative supply, Pin 4.

APPLICATIONS

TheLT1228currentfeedbackamplifierhasveryhighinput

impedanceandthereforeitisanexcellentbufferfortheout-

putofthetransconductanceamplifier.Thecurrentfeedback

amplifiermaintainsitswidebandwidthoverawiderangeof

voltage gains making it easy to interface the transconduc-

tance amplifier output to other circuitry. The current feed-

back amplifier is designed to drive low impedance loads,

suchascables,withexcellentlinearityathighfrequencies.

n

Video DC Restore (Clamp) Circuits

n

Video Differential Input Amplifiers

n

Video Keyer/Fader Amplifiers

n

AGC Amplifiers

n

Tunable Filters

n

Oscillators

L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks of Linear

Technology Corporation. All other trademarks are the property of their respective owners.

TYPICAL APPLICATION

Frequency Response

6

Differential Input Variable Gain Amp

V

=

15V

S

L

R

= 100Ω

3

0

15V

4.7µF

I

= 1mA

SET

+

R3A

–3

10k

7

4

3

2

+

–

+

–6

1

8

g

R2A

10k

V

m

+

–

IN

–9

6

I

= 300µA

SET

V

–

5

CFA

OUT

–12

–15

–18

I

SET

R

F

–15V

R2

470Ω

R4

1.24k

R5

R3

100Ω

= 100µA

1M

4.7µF

10k

SET

+

100Ω

R1

270Ω

R

G

–21

–24

HIGH INPUT RESISTANCE

EVEN WHEN POWER IS OFF

–18dB < GAIN < 2dB

10Ω

R6

6.19k

100k

10M

100M

V

≤ 3V

IN

RMS

LT1228 • TA02

LT1228 • TA01

FREQUENCY (Hz)

1228fd

1

LT1228

ABSOLUTE MAXIMUM RATINGS

PIN CONFIGURATION

(Note 1)

Supply Voltage....................................................... ± 18V

Input Current, Pins 1, 2, 3, ꢀ, 8 (Note 8) ..............±1ꢀmA

Output Short Circuit Duration (Note 2) .........Continuous

Operating Temperature Range

LT1228C...................................................0°C to 70°C

LT1228I................................................–40°C to 8ꢀ°C

LT1228M (OBSOLETE) ...................... –ꢀꢀ°C to 12ꢀ°C

Storage Temperature Range ..................–6ꢀ°C to 1ꢀ0°C

Junction Temperature

TOP VIEW

I

1

2

3

4

8

7

6

5

GAIN

OUT

+

–

+

V

–IN

g

m

+IN

V

OUT

–

V

I

SET

N8 PACKAGE

8-LEAD PDIP

S8 PACKAGE

8-LEAD PLASTIC SO

T

= 1ꢀ0°C, θ = 100°C/W (N)

JA

JMAX

= 1ꢀ0°C, θ = 1ꢀ0°C/W (N)

JA

Plastic Package................................................. 1ꢀ0°C

Ceramic Package (OBSOLETE)......................... 17ꢀ°C

Lead Temperature (Soldering, 10 sec)...................300°C

J8 PACKAGE

8-LEAD CERDIP

T

= 17ꢀ°C, θ = 100°C/W (J)

JMAX

JA

OBSOLETE PACKAGE

ORDER INFORMATION

LEAD FREE FINISH

LT1228CN8#PBF

LT1228IN8#PBF

LT1228CS8#PBF

LT1228IS8#PBF

TAPE AND REEL

PART MARKING

LT1228CN8

LT1228IN8

1228

PACKAGE DESCRIPTION

8-Lead Plastic DIP

8-Lead Plastic SO

TEMPERATURE RANGE

0°C to 70°C

LT1228CN8#TRPBF

LT1228IN8#TRPBF

LT1228CS8#TRPBF

LT1228IS8#TRPBF

–40°C to 8ꢀ°C

0°C to 70°C

1228I

–40°C to 8ꢀ°C

OBSOLETE PACKAGE

LT1228MJ8

LT1228CJ8

LT1228MJ8#TRPBF

LT1228CJ8#TRPBF

LT1228MJ8

LT1228CJ8

8-Lead CERDIP

–ꢀꢀ°C to 12ꢀ°C

0°C to 70°C

Consult LTC Marketing for parts specified with wider operating temperature ranges.

Consult LTC Marketing for information on nonstandard lead based finish parts.

For more information on lead free part marking, go to: http://www.linear.com/leadfree/

For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/

ELECTRICAL CHARACTERISTICS ^Thel denotes the specifications which apply over the full operating

temperature range, otherwise specifications are at T_A= 25°C. Current Feedback Amplifier, Pins 1, 6, 8. 5V ≤ V_S≤ 15V, I_SET= 0µA,

V_CM= 0V unless otherwise noted.

SYMBOL

PARAMETER

CONDITIONS

T = 2ꢀ°C

MIN

TYP

MAX

UNITS

V

Input Offset Voltage

3

10

1ꢀ

mV

OS

A

l

Input Offset Voltage Drift

Noninverting Input Current

10

µV/°C

+

I

T = 2ꢀ°C

A

0.3

3

10

µA

IN

l

–

Inverting Input Current

T = 2ꢀ°C

A

10

6ꢀ

100

µA

IN

e

Input Noise Voltage Density

Input Noise Current Density

f = 1kHz, R = 1k, R = 10Ω, R = 0Ω

6

nV/√Hz

pV/√Hz

n

F

G

S

i

f = 1kHz, R = 1k, R = 10Ω, R = 10k

1.4

n

F

G

S

1228fd

2

LT1228

ELECTRICAL CHARACTERISTICS ^Thel denotes the specifications which apply over the full operating

temperature range, otherwise specifications are at T_A= 25°C. Current Feedback Amplifier, Pins 1, 6, 8. 5V ≤ V_S≤ 15V, I_SET= 0µA,

V_CM= 0V unless otherwise noted.

SYMBOL

PARAMETER

CONDITIONS

MIN

TYP

MAX

UNITS

l

R

IN

Input Resistance

V

IN

V

IN

=

13V, V = 1ꢀV

2

2ꢀ

MΩ

S

3V, V = ꢀV

S

C

Input Capacitance (Note 3)

Input Voltage Range

V = ꢀV

6

pF

IN

S

V = 1ꢀV, T = 2ꢀ°C

S

13

12

13.ꢀ

V

A

l

V = ꢀV, T = 2ꢀ°C

S

3

2

3.ꢀ

V

A

CMRR

PSRR

Common Mode Rejection Ratio

V = 1ꢀV, V

S

=

13V, T = 2ꢀ°C

ꢀꢀ

69

dB

S

CM

A

l

V = 1ꢀV, V

12V

V = ꢀV, V

S

=

3V, T = 2ꢀ°C

S

CM

A

V = ꢀV, V

=

2V

Inverting Input Current Common

Mode Rejection

V = 1ꢀV, V

S

=

13V, T = 2ꢀ°C

2.ꢀ

10

µA/V

S

CM

A

l

V = 1ꢀV, V

12V

V = ꢀV, V

S

=

3V, T = 2ꢀ°C

S

CM

A

V = ꢀV, V

2V

Power Supply Rejection Ratio

V = 2V to 1ꢀV, T = 2ꢀ°C

60

80

10

dB

S

A

l

V = 3V to 1ꢀV

S

Noninverting Input Current Power

Supply Rejection

V = 2V to 1ꢀV, T = 2ꢀ°C

ꢀ0

nA/V

S

A

V = 3V to 1ꢀV

S

Inverting Input Current Power Supply V = 2V to 1ꢀV, T = 2ꢀ°C

Rejection

0.1

ꢀ

µA/V

S

A

V = 3V to 1ꢀV

S

l

A

Large-Signal Voltage Gain

V = 1ꢀV, V

=

10V, R = 1k

LOAD

ꢀꢀ

6ꢀ

dB

V

S

OUT

V = ꢀV, V

=

2V, R

= 1ꢀ0Ω

S

OUT

LOAD

–

l

R

Transresistance, ∆V /∆I

V = 1ꢀV, V

=

10V, R = 1k

LOAD

100

200

kΩ

kΩ

OL

OUT IN

S

OUT

V = ꢀV, V

=

2V, R

= 1ꢀ0Ω

S

OUT

LOAD

V

Maximum Output Voltage Swing

Maximum Output Current

V = 1ꢀV, R

= 400Ω, T = 2ꢀ°C

12

10

13.ꢀ

3.7

6ꢀ

V

OUT

S

LOAD

A

l

V = ꢀV, R

= 1ꢀ0Ω, T = 2ꢀ°C

3

2.ꢀ

V

S

LOAD

A

I

R

LOAD

= 0Ω, T = 2ꢀ°C

30

2ꢀ

12ꢀ

mA

OUT

S

A

l

Supply Current

V

OUT

= 0V, I = 0V

6

11

mA

V/µs

ns

SET

SR

Slew Rate (Notes 4 and 6)

Slew Rate

T = 2ꢀ°C

A

300

ꢀ00

3ꢀ00

10

V = 1ꢀV, R = 7ꢀ0Ω, R = 7ꢀ0Ω, R = 400Ω

S

F

G

L

t

Rise Time (Notes ꢀ and 6)

Small-Signal Bandwidth

Small-Signal Rise Time

Propagation Delay

T = 2ꢀ°C

A

20

r

BW

V = 1ꢀV, R = 7ꢀ0Ω, R = 7ꢀ0Ω, R = 100Ω

100

3.ꢀ

MHz

ns

S

F

G

L

t

V = 1ꢀV, R = 7ꢀ0Ω, R = 7ꢀ0Ω, R = 100Ω

S F G L

r

V = 1ꢀV, R = 7ꢀ0Ω, R = 7ꢀ0Ω, R = 100Ω

3.ꢀ

ns

S

F

G

L

Small-Signal Overshoot

Settling Time

V = 1ꢀV, R = 7ꢀ0Ω, R = 7ꢀ0Ω, R = 100Ω

1ꢀ

%

S

F

G

L

t

0.1%, V

= 10V, R =1k, R = 1k, R =1k

4ꢀ

ns

S

OUT

F

G

L

Differential Gain (Note 7)

Differential Phase (Note 7)

Differential Gain (Note 7)

Differential Phase (Note 7)

V = 1ꢀV, R = 7ꢀ0Ω, R = 7ꢀ0Ω, R = 1k

0.01

0.04

0.1

%

S

F

G

L

V = 1ꢀV, R = 7ꢀ0Ω, R = 7ꢀ0Ω, R = 1k

DEG

%

S

F

G

L

V = 1ꢀV, R = 7ꢀ0Ω, R = 7ꢀ0Ω, R = 1ꢀ0Ω

S

F

G

L

V = 1ꢀV, R = 7ꢀ0Ω, R = 7ꢀ0Ω, R = 1ꢀ0Ω

DEG

S

F

G

L

1228fd

3

LT1228

ELECTRICAL CHARACTERISTICS ^Thel denotes the specifications which apply over the full operating

temperature range, otherwise specifications are at T_A= 25°C. Transconductance Amplifier, Pins 1, 2, 3, 5. 5V ≤ V_S≤ 15V,

I_SET= 100µA, V_CM= 0V unless otherwise noted.

SYMBOL

PARAMETER

CONDITIONS

= 1mA, T = 2ꢀ°C

MIN

TYP

MAX

UNITS

V

Input Offset Voltage

I

0.ꢀ

ꢀ

10

mV

OS

SET

A

l

Input Offset Voltage Drift

Input Offset Current

10

40

µV/°C

I

T = 2ꢀ°C

A

200

ꢀ00

nA

OS

l

Input Bias Current

T = 2ꢀ°C

A

0.4

1

ꢀ

µA

B

e

Input Noise Voltage Density

f = 1kHz

20

nV/√Hz

n

l

R

IN

Input Resistance-Differential Mode

Input Resistance-Common Mode

V

≈

30mV

30

200

kΩ

IN

l

V = 1ꢀV, V

=

CM

12V

2V

ꢀ0

1000

MΩ

MΩ

S

V = ꢀV, V

=

S

C

IN

Input Capacitance

3

pF

Input Voltage Range

V = 1ꢀV, T = 2ꢀ°C

S

13

12

14

V

S

A

l

V = 1ꢀV

V = ꢀV, T = 2ꢀ°C

3

2

4

V

S

A

V = ꢀV

CMRR

PSRR

Common Mode Rejection Ratio

Power Supply Rejection Ratio

V = 1ꢀV, V

S

=

13V, T = 2ꢀ°C

12V

60

100

dB

S

CM

A

V = 1ꢀV, V

V = ꢀV, V

S

=

3V, T = 2ꢀ°C

2V

60

dB

S

CM

A

V = ꢀV, V

V = 2V to 1ꢀV, T = 2ꢀ°C

60

dB

S

A

V = 3V to 1ꢀV

g

Transconductance

I

= 100µA, I

=

30µA, T = 2ꢀ°C

0.7ꢀ

1.00

–0.33

100

1.2ꢀ

130

µA/mV

%/°C

µA

m

SET

OUT

A

l

Transconductance Drift

Maximum Output Current

Output Leakage Current

I

= 100µA

70

OUT

SET

= 0µA (+I of CFA), T = 2ꢀ°C

0.3

3

10

µA

OL

SET

IN

A

l

V

Maximum Output Voltage Swing

Output Resistance

V = 1ꢀV , R1 = ∞

S

13

3

14

4

V

OUT

S

V = ꢀV , R1 = ∞

l

R

V = 1ꢀV, V =

OUT

13V

3V

2

8

MΩ

O

S

V = ꢀV, V

=

S

OUT

Output Capacitance (Note 3)

Supply Current, Both Amps

Total Harmonic Distortion

Small-Signal Bandwidth

Small-Signal Rise Time

Propagation Delay

V = ꢀV

6

9

pF

mA

%

S

l

I

= 1mA

1ꢀ

S

SET

THD

BW

V

= 30mV

at 1kHz, R1 = 100k

0.2

80

ꢀ

IN

RMS

R1 = ꢀ0Ω, I = ꢀ00µA

MHz

ns

SET

t

r

R1 = ꢀ0Ω, I = ꢀ00µA, 10% to 90%

SET

R1 = ꢀ0Ω, I = ꢀ00µA, ꢀ0% to ꢀ0%

ꢀ

ns

SET

Note 1: Stresses beyond those listed under Absolute Maximum Ratings

may cause permanent damage to the device. Exposure to any Absolute

Maximum Rating condition for extended periods may affect device

reliability and lifetime.

Note 5: Rise time is measured from 10% to 90% on a ꢀ00mV output signal

while operating on 1ꢀV supplies with R = 1k, R = 110Ω and R = 100Ω.

This condition is not the fastest possible, however, it does guarantee the

internal capacitances are correct and it makes automatic testing practical.

F

G

L

Note 2: A heat sink may be required depending on the power supply voltage.

Note 6: AC parameters are 100% tested on the ceramic and plastic DIP

packaged parts (J and N suffix) and are sample tested on every lot of the SO

packaged parts (S suffix).

Note 3: This is the total capacitance at Pin 1. It includes the input capacitance

of the current feedback amplifier and the output capacitance of the

transconductance amplifier.

Note 7: NTSC composite video with an output level of 2V.

Note 4: Slew rate is measured at ꢀV on a 10V output signal while operating

Note 8: Back to back 6V Zener diodes are connected between Pins 2 and 3

for ESD protection.

on 1ꢀV supplies with R = 1k, R = 110Ω and R = 400Ω. The slew rate is

F

G

L

much higher when the input is overdriven, see the Applications Information

section.

1228fd

4

LT1228

TYPICAL PERFORMANCE CHARACTERISTICS ^{Transconductance Amplifier, Pins 1, 2, 3, 5}

Small-Signal Bandwidth

vs Set Current

Small-Signal Transconductance

and Set Current vs Bias Voltage

Small-Signal Transconductance

vs DC Input Voltage

100

10

1

2.0

1.8

1.6

1.4

1.2

1.0

0.8

0.6

0.4

100

10

10000

1000

100

10

R1 = 100Ω

V

=

2V TO 15V

V

= 1ꢀV

S

A

S

V

SET

=

2V TO 15V

= 100µA

S

T

= 25°C

I

R1 = 1k

–55°C

1

R1 = 10k

25°C

0.1

125°C

0.01

0.001

1.0

R1 = 100k

0.2

0

0.1

1.5

0.1

10

100

SET CURRENT (µA)

1000

–200 –150 –100 –50

0

50 100 150 200

0.9

1.0

1.1

1.2

1.3

1.4

INPUT VOLTAGE (mVDC)

LT1228 • TPC03

BIAS VOLTAGE, PIN 5 TO 4, (V)

LT1228 • TPC01

LT1228 • TPC02

Total Harmonic Distortion

vs Input Voltage

Spot Output Noise Current

vs Frequency

Input Common Mode Limit

vs Temperature

+

10

1

V

1000

100

10

V

= 1ꢀV

V

= ±2V TO ±15V

= 25°C

+

S

A

V

= 2V TO 15V

–0.5

–1.0

–1.5

–2.0

T

I

= 1mA

SET

I

= 100µA

SET

2.0

1.5

1.0

0.5

0.1

0.01

–

V

= –2V TO –15V

= 100µA

10k

SET

I

= 1mA

10

SET

–

V

1

100

1000

–50 –25

0

25

50

75 100 125

10

100

1k

FREQUENCY (Hz)

100k

INPUT VOLTAGE (mV

)

LT1228 • TPC04

TEMPERATURE (°C)

P–P

LT1228 • TPC06

LT1228 • TPC05

Small-Signal Control Path

Bandwidth vs Set Current

Small-Signal Control Path

Gain vs Input Voltage

Output Saturation Voltage

vs Temperature

+

100

10

1

1.0

0.9

0.8

0.7

0.6

0.5

0.4

0.3

0.2

0.1

0

V

= ±±V Tꢀ ±1ꢁV

= ±00mV

S

IN

–0.5

–1.0

(PIN 2 TO 3)

2V ꢀ V

ꢀ 15V

R1 = ∞^S

∆I

ꢀUT

OUT

∆I

SET

+1.0

+0.5

–

V

10

100

SET CURRENT (µA)

1000

0

40

80

120

160

200

–50 –25

0

25

50

75 100 125

INPUT VOLTAGE, PIN 2 TO 3, (mVDC)

TEMPERATURE (°C)

LT1228 • TPC07

LT1228 • TPC08

LT1228 • TPC09

1228fd

5

LT1228

Transconductance Amplifier, Pins 1, 6, 8

–3dB Bandwidth vs Supply –3dB Bandwidth vs Supply

TYPICAL PERFORMANCE CHARACTERISTICS

Voltage Gain and Phase

vs Frequency, Gain = 6dB

Voltage, Gain = 2, R_L= 100Ω

Voltage, Gain = 2, R_L= 1k

180

160

140

180

160

140

8

7

6

0

PHASE

PEAKING ≤ 0.5dB

PEAKING ≤ 5dB

45

90

R = 500Ω

F

GAIN

R = 500Ω

R = 750Ω

F

135

180

225

5

4

120

100

80

60

40

20

0

120

100

80

60

40

20

0

R = 750Ω

F

3

2

1

R = 1k

F

PEAKING ≤ 0.5dB

PEAKING ≤ 5dB

V

= 15V

S

L

F

R

= 100Ω

0

–1

–2

R = 2k

F

R = 1k

F

R = 2k

F

R = 750Ω

0

2

4

6

8

10 12 14 16 18

0

2

4

6

8

10 12 14 16 18

0.1

1

10

100

SUPPLY VOLTAGE ( Vꢀ

FREQUENCY (MHz)

LT1228 • TPC10

LT1228 • TPC11

LT1228 • TPC12

Voltage Gain and Phase

vs Frequency, Gain = 20dB

–3dB Bandwidth vs Supply

Voltage, Gain = 10, R_L= 100Ω

–3dB Bandwidth vs Supply

Voltage, Gain = 10, R_L= 1k

22

21

20

0

180

160

140

180

160

140

PHASE

PEAKING ≤ 0.5dB

PEAKING ≤ 5dB

PEAKING ≤ 0.5dB

PEAKING ≤ 5dB

45

90

GAIN

135

180

225

19

18

120

100

80

60

40

20

0

120

100

80

60

40

20

0

R = 250Ω

R = 500Ω

F

R = 250Ω

F

17

16

15

R = 500Ω

F

R = 750Ω

F

R = 750Ω

F

R = 1k

F

V

= 15V

S

L

R = 1k

F

R

= 100Ω

14

13

12

R = 750Ω

R = 2k

F

R = 2k

F

0

2

4

6

8

10 12 14 16 18

0.1

1

10

100

2

4

6

8

10 12 14 16 18

SUPPLY VOLTAGE ( Vꢀ

FREQUENCY (MHz)

SUPPLY VOLTAGE ( Vꢀ

LT1228 • TPC13

LT1228 • TPC14

LT1228 • TPC15

Voltage Gain and Phase

vs Frequency, Gain = 40dB

–3dB Bandwidth vs Supply

Voltage, Gain = 100, R_L= 100Ω

–3dB Bandwidth vs Supply

Voltage, Gain = 100, R_L= 1k

42

41

40

0

18

16

14

18

16

14

PHASE

45

90

R

= 500Ω

GAIN

F

135

180

225

39

38

12

10

8

12

10

8

R

= 500Ω

F

R

= 1k

F

37

36

35

R

= 1k

= 2k

F

R

= 2k

F

6

R

F

V

= 15V

S

L

R

= 100Ω

4

34

33

32

R = 750Ω

F

2

0

0.1

1

10

100

2

4

6

8

10 12 14 16 18

0

2

4

6

8

10 12 14 16 18

FREQUENCY (MHz)

SUPPLY VOLTAGE ( Vꢀ

LT1228 • TPC16

LT1228 • TPC18

LT1228 • TPC17

1228fd

6

LT1228

Transconductance Amplifier, Pins 1, 6, 8

TYPICAL PERFORMANCE CHARACTERISTICS

Maximum Capacitive Load

vs Feedback Resistor

Total Harmonic Distortion

vs Frequency

2nd and 3rd Harmonic

Distortion vs Frequency

0.10

–20

–30

10k

1k

V

=

15V

P–P

S

O

L

V

=

1ꢀV

S

L

= 2V

R

= 400Ω

R

= 100Ω

V

= ±±V

R = R = 7ꢀ0Ω

S

F

G

R = 750Ω

A

F

= 10dB

2nd

V

–40

–50

–60

–70

V

= ±1±V

S

100

10

1

0.01

3rd

V

= 7V

= 1V

O

RMS

R

= 1k

L

PEAKING ≤ ±dB

GAIN = 2

0.001

10

100

1k

FREQUENCY (Hz)

10k

100k

1

10

FREQUENCY (MHz)

100

0

1

2

3

FEEDBACK RESISTOR (kΩ)

LT1228 • TPC20

LT1228 • TPC19

LT1228 • TPC21

Input Common Mode Limit

vs Temperature

Output Saturation Voltage

vs Temperature

Output Short-Circuit Current

vs Temperature

+

70

60

50

40

30

V

–0.5

–1.0

–1.5

–2.0

–0.5

–1.0

+

V

= 2V TO 15V

R

= ∞

L

2V ꢀ V ꢀ 15V

S

2.0

1.5

1.0

0.5

–

= –2V TO –15V

1.0

0.5

–

V

–50 –25

0

25 50 75 100 125 150 175

TEMPERATURE (°C)

–50 –25

0

25

50

75 100 125

–50 –25

0

25

50

75 100 125

TEMPERATURE (°C)

LT1228 • TPC22

LT1228 • TPC23

LT1228 • TPC24

Spot Noise Voltage and Current

vs Frequency

Power Supply Rejection

vs Frequency

Output Impedance vs Frequency

80

100

10

V

= 1ꢀV

S

V

R

=

1ꢀV

S

L

F

= 100Ω

= R = 7ꢀ0Ω

G

60

POSITIVE

1.0

–i

n

R

= R = 2k

G

F

40

20

0

10

R

= R = 750Ω

G

F

e

0.1

0.01

n

NEGATIVE

+i

n

1

0.001

10k

100k

1M

10M

100M

10

100

1k

FREQUENCY (Hz)

10k

100k

10k

100k

1M

FREQUENCY (Hz)

10M

100M

FREQUENCY (Hz)

LT1228 • TPC26

LT1228 • TPC25

LT1228 • TPC27

1228fd

7

LT1228

Current Feedback Amplifier, Pins 1, 6, 8

TYPICAL PERFORMANCE CHARACTERISTICS

Setting Time to 10mV

vs Output Step

Setting Time to 1mV

vs Output Step

Supply Current vs Supply Voltage

10

8

10

8

10

9

NONINVERTING

INVERTING

NONINVERTING

6

8

–55°C

25°C

INVERTING

4

7

2

6

V

= 1ꢀV

G

S

F

V

=

1ꢀV

G

S

F

0

5

125°C

175°C

R = R = 1k

–2

–4

–2

–4

4

3

2

1

0

INVERTING

–6

NONINVERTING

–8

NONINVERTING

20

INVERTING

–10

0

40

60

80

100

0

4

8

12

16

20

0

2

4

6

8

10 12 14 16 18

SETTLING TIME (ns)

SETTLING TIME (µs)

SUPPLY VOLTAGE ( Vꢀ

LT1228 • TPC28

LT1228 • TPC29

LT1228 • TPC30

SIMPLIFIED SCHEMATIC

+

V

7

6

4

BIAS

+IN

3

–IN

I

OUT

2

8

GAIN

OUT

1

I

SET

5

–

LT1228 • TA03

1228fd

8

LT1228

APPLICATIONS INFORMATION

The LT1228 contains two amplifiers, a transconductance

amplifier(voltage-to-current)andacurrentfeedbackampli-

fier(voltage-to-voltage).Thegainofthetransconductance

amplifier is proportional to the current that is externally

programmed into Pin ꢀ. Both amplifiers are designed to

operate on almost any available supply voltage from 4V

( 2V) to 30V ( 1ꢀV). The output of the transconductance

amplifier is connected to the noninverting input of the

current feedback amplifier so that both fit into an eight

pin package.

Resistance Controlled Gain

If the set current is to be set or varied with a resistor or

potentiometeritispossibletousethenegativetemperature

coefficient at Pin ꢀ (with respect to Pin 4) to compensate

for the negative temperature coefficient of the transcon-

ductance. The easiest way is to use an LT1004-2.ꢀ, a 2.ꢀV

reference diode, as shown below:

Temperature Compensation of g_mwith a 2.5V Reference

R

I

SET

TRANSCONDUCTANCE AMPLIFIER

g

m

I

V

be

TheLT1228transconductanceamplifierhasahighimped-

ance differential input (Pins 2 and 3) and a current source

output (Pin 1) with wide output voltage compliance. The

4

2.5V

2E

g

5

R

SET

be

voltage to current gain or transconductance (g ) is set

m

LT1004-2.5

by the current that flows into Pin ꢀ, I . The voltage at

SET

–

V

LT1228 • TA04

Pin ꢀ is two forward biased diode drops above the nega-

tive supply, Pin 4. Therefore the voltage at Pin ꢀ (with

The current flowing into Pin ꢀ has a positive temperature

coefficient that cancels the negative coefficient of the

transconductance. The following derivation shows why a

2.ꢀVreferenceresultsinzerogainchangewithtemperature:

–

respect to V ) is about 1.2V and changes with the log of

the set current (120mV/decade), see the characteristic

curves.Thetemperaturecoefficientofthisvoltageisabout

–4mV/°C (–3300ppm/°C) and the temperature coefficient

oftheloggingcharacteristicis3300ppm/°C.Itisimportant

that the current into Pin ꢀ be limited to less than 1ꢀmA.

THE LT1228 WILL BE DESTROYED IF PIN ꢀ IS SHORTED

TO GROUND OR TO THE POSITIVE SUPPLY. A limiting

resistor (2k or so) should be used to prevent more than

1ꢀmA from flowing into Pin ꢀ.

q

I

SET

Sinceg =

×

= 10 •I

SET

m

kT 3.87

akT

n





cT

andV = E –

wherea = In

be

g





q

Ic





≈ 19.4 at 27°C c = 0.001, n = 3, Ic = 100µA

(

)

The small-signal transconductance (g ) is given as

m

E is about 1.2ꢀV so the 2.ꢀV reference is 2E . Solving

the loop for the set current gives:

g

g = 10 • I , with g in (A/V) and I in (A).This rela-

m

SET

m

SET

tionship holds over many decades of set current (see the

characteristic curves). The transconductance is inversely

proportionaltoabsolutetemperature(–3300ppm/°C).The

input stage of the transconductance amplifier has been

designed to operate with much larger signals than is pos-

sible with an ordinary diff-amp. The transconductance of

the input stage varies much less than 1% for differential

input signals over a 30 mV range (see the characteristic

curveSmall-SignalTransconductancevsDCInputVoltage).



akT

q

2E_g–2 E_g–









2akT

Rq

I_SET

=

or I_SET=

R

1228fd

9

LT1228

APPLICATIONS INFORMATION

Substitutingintothe equationfortransconductancegives:

diode drops above the negative supply, a single resistor

from the control voltage source to Pin ꢀ will suffice in

many applications. The control voltage is referenced to

the negative supply and has an offset of about 900mV.

The conversion will be monotonic, but the linearity is

determined by the change in the voltage at Pin ꢀ (120mV

per decade of current). The characteristic is very repeat-

able since the voltage at Pin ꢀ will vary less than ꢀ%

from part to part. The voltage at Pin ꢀ also has a negative

temperature coefficient as described in the previous sec-

tion. When the gain of several LT1228s are to be varied

together, the current can be split equally by using equal

value resistors to each Pin ꢀ.

a

10

R

g_m=

=

1.94R

The temperature variation in the term “a” can be ignored

since it is much less than that of the term “T” in the equa-

tion for V . Using a 2.ꢀV source this way will maintain the

be

gain constant within 1% over the full temperature range of

–ꢀꢀ°C to 12ꢀ°C. If the 2.ꢀV source is off by 10%, the gain

willvaryonlyabout 6%overthesametemperaturerange.

Wecanalsotemperaturecompensatethetransconductance

withoutusinga2.ꢀVreferenceifthenegativepowersupply

is regulated. A Thevenin equivalent of 2.ꢀV is generated

from two resistors to replace the reference. The two resis-

tors also determine the maximum set current, approxi-

Formoreaccurate(andlinear)control,avoltage-to-current

converter circuit using one op amp can be used. The fol-

lowing circuit has several advantages. The input no longer

has to be referenced to the negative supply and the input

can be either polarity (or differential). This circuit works

on both single and split supplies since the input voltage

and the Pin ꢀ voltage are independent of each other. The

temperature coefficient of the output current is set by Rꢀ.

mately 1.1V/R . By rearranging the Thevenin equations

TH

to solve for R4 and R6 we get the following equations in

terms of R and the negative supply, V .

TH

EE

R_TH

R_THV_EE

2.5V

R4=

and R6=













2.5V

V_EE

1–

R3

1M

Temperature Compensation of g_mwith a Thevenin Voltage

R1

1M

1.03k

R'

R5

1k

V1

V2

+

–

I

SET

I

OUT

R2

1M

LT1006

50pF

TO PIN 5

g

OF LT1228

m

I

V

be

4

R4

1M

R6

V

= 2.5V

TH

6.19kΩ

5

R'

SET

R1 = R2

R3 = R4

R4

1.24kΩ

(V1 – V2) R3

–15V

I

=

•

= 1mA/V

OUT

LT1228 • TA05

R5

R1

LT1228 • TA19

Voltage Controlled Gain

Digital control of the transconductance amplifier gain is

done by converting the output of a DAC to a current flow-

ing into Pin ꢀ. Unfortunately most current output DACs

sink rather than source current and do not have output

To useavoltagetocontrolthegainofthetransconductance

amplifier requires converting the voltage into a current

that flows into Pin ꢀ. Because the voltage at Pin ꢀ is two

1228fd

10

LT1228

APPLICATIONS INFORMATION

Transconductance Amp Small-Signal Response

compliance compatible with Pin ꢀ of the LT1228. There-

fore, the easiest way to digitally control the set current

is to use a voltage output DAC and a voltage-to-current

circuit. Thepreviousvoltage-to-currentconverterwilltake

the output of any voltage output DAC and drive Pin ꢀ with

a proportional current. The R, 2R CMOS multiplying DACs

operatinginthevoltageswitchingmode workwellonboth

single and split supplies with the above circuit.

I_SET= 500µA, R1 = 50Ω

Logarithmic control is often easier to use than linear

control. A simple circuit that doubles the set current

for each additional volt of input is shown in the voltage

controlled state variable filter application near the end of

this data sheet.

Transconductance Amplifier Frequency Response

CURRENT FEEDBACK AMPLIFIER

The bandwidth of the transconductance amplifier is a

function of the set current as shown in the characteristic

curves. At set currents below 100µA, the bandwidth is

approximately:

The LT1228 current feedback amplifier has very high

noninvertinginputimpedanceandisthereforeanexcellent

buffer for the output of the transconductance amplifier.

The noninverting input is at Pin 1, the inverting input at

Pin 8 and the output at Pin 6. The current feedback ampli-

fier maintains its wide bandwidth for almost all voltage

gains making it easy to interface the output levels of the

transconductance amplifier to other circuitry. The cur-

rent feedback amplifier is designed to drive low imped-

ance loads such as cables with excellent linearity at high

frequencies.

11

–3dB bandwidth = 3 • 10

I

SET

The peak bandwidth is about 80MHz at ꢀ00µA. When a

resistor is used to convert the output current to a volt-

age, the capacitance at the output forms a pole with the

resistor. The best case output capacitance is about ꢀpF

with 1ꢀV supplies and 6pF with ꢀV supplies. You must

add any PC board or socket capacitance to these values to

get the total output capacitance. When using a 1k resistor

at the output of the transconductance amp, the output

capacitance limits the bandwidth to about 2ꢀMHz.

Feedback Resistor Selection

Thesmall-signalbandwidthoftheLT1228currentfeedback

amplifier is set by the external feedback resistors and the

internal junction capacitors. As a result, the bandwidth is

a function of the supply voltage, the value of the feedback

resistor, the closed-loop gain and load resistor. The char-

acteristic curves of bandwidth versus supply voltage are

done with a heavy load (100Ω) and a light load (1k) to

The output slew rate of the transconductance amplifier is

the set current divided by the output capacitance, which

is 6pF plus board and socket capacitance. For example

with the set current at 1mA, the slew rate would be over

100V/µs.

1228fd

11

LT1228

APPLICATIONS INFORMATION

Capacitance on the Inverting Input

show the effect of loading. These graphs also show the

family of curves that result from various values of the

feedback resistor. These curves use a solid line when the

response has less than 0.ꢀdB of peaking and a dashed line

for the response with 0.ꢀdB to ꢀdB of peaking. The curves

stop where the response has more than ꢀdB of peaking.

Current feedback amplifiers want resistive feedback from

the output to the inverting input for stable operation. Take

care to minimize the stray capacitance between the output

and the inverting input. Capacitance on the inverting input

to ground will cause peaking in the frequency response

(and overshoot in the transient response), but it does

not degrade the stability of the amplifier. The amount of

capacitance that is necessary to cause peaking is a func-

tion of the closed-loop gain taken. The higher the gain,

the more capacitance is required to cause peaking. For

example, in a gain of 100 application, the bandwidth can

be increased from 10MHz to 17MHz by adding a 2200pF

Current Feedback Amp Small-Signal Response

V_S= 15V, R_F= R_G= 750Ω, R_L= 100Ω

capacitor, as shown below. C must have very low series

G

resistance, such as silver mica.

1

+

V

IN

6

CFA

V

OUT

8

–

R

F

510Ω

At a gain of two, on 1ꢀV supplies with a 7ꢀ0Ω feedback

resistor, the bandwidth into a light load is over 160MHz

without peaking, but into a heavy load the bandwidth re-

ducesto100MHz.Theloadinghassomucheffectbecause

thereisamildresonanceintheoutputstagethatenhances

the bandwidth at light loads but has its Q reduced by the

heavy load. This enhancement is only useful at low gain

settings, at a gain of ten it does not boost the bandwidth.

At unity gain, the enhancement is so effective the value of

thefeedbackresistorhasverylittleeffectonthebandwidth.

At very high closed-loop gains, the bandwidth is limited

by the gain-bandwidth product of about 1GHz. The curves

show that the bandwidth at a closed-loop gain of 100 is

10MHz, only one tenth what it is at a gain of two.

R

G

C

G

5.1Ω

LT1228 • TA08

Boosting Bandwidth of High Gain Amplifier

with Capacitance On Inverting Input

49

46

C

= 4700pF

G

43

40

37

34

31

28

25

22

19

C

= 2200pF

G

C

= 0

G

1

10

FREQUENCY (MHz)

100

LT1228 • TA09

1228fd

12

LT1228

APPLICATIONS INFORMATION

Capacitive Loads

The output slew rate is set by the value of the feedback

resistors and the internal capacitance. At a gain of ten with

a 1k feedback resistor and 1ꢀV supplies, the output slew

rate is typically ꢀ00V/µs and –8ꢀ0V/µs. There is no input

stage enhancement because of the high gain. Larger feed-

backresistorswillreducetheslewrateaswilllowersupply

voltages, similar to the way the bandwidth is reduced.

TheLT1228currentfeedbackamplifiercandrivecapacitive

loadsdirectlywhenthepropervalueoffeedbackresistoris

used.ThegraphofMaximumCapacitiveLoadvsFeedback

Resistor should be used to select the appropriate value.

The value shown is for ꢀdB peaking when driving a 1k

load, at a gain of 2. This is a worst case condition, the

amplifierismorestableathighergains,anddrivingheavier

loads. Alternatively, a small resistor (10Ω to 20Ω) can be

put in series with the output to isolate the capacitive load

from the amplifier output. This has the advantage that the

amplifier bandwidth is only reduced when the capacitive

load is present and the disadvantage that the gain is a

function of the load resistance.

Current Feedback Amp Large-Signal Response

V_S= 15V, R_F= 1k, R_G= 110Ω, R_L= 400Ω

Slew Rate

The slew rate of the current feedback amplifier is not inde-

pendent of the amplifier gain configuration the way it is in

a traditional op amp. This is because the input stage and

the output stage both have slew rate limitations. The input

stage of the LT1228 current feedback amplifier slews at

about 100V/µs before it becomes nonlinear. Faster input

signalswillturnonthenormallyreversebiasedemitterson

theinputtransistorsandenhancetheslewratesignificantly.

This enhanced slew rate can be as much as 3ꢀ00V/µs!

Settling Time

The characteristic curves show that the LT1228 current

feedback amplifier settles to within 10mV of final value

in 40ns to ꢀꢀns for any output step less than 10V. The

curve of settling to 1mV of final value shows that there

is a slower thermal contribution up to 20µs. The thermal

settling component comes from the output and the input

stage. The output contributes just under 1mV/V of output

changeandtheinputcontributes300µV/Vofinputchange.

Fortunately the input thermal tends to cancel the output

thermal. For this reason the noninverting gain of two

configuration settles faster than the inverting gain of one.

Current Feedback Amp Large-Signal Response

V_S= 15V, R_F= R_G= 750Ω Slew Rate Enhanced

1228fd

13

LT1228

APPLICATIONS INFORMATION

Power Supplies

For example, let’s calculate the worst case power dis-

sipation in a variable gain video cable driver operating on

12V supplies that delivers a maximum of 2V into 1ꢀ0Ω.

The maximum set current is 1mA.

The LT1228 amplifiers will operate from single or split

supplies from 2V (4V total) to 18V (36V total). It is

not necessary to use equal value split supplies, however

the offset voltage and inverting input bias current of the

current feedback amplifier will degrade. The offset voltage

changesabout3ꢀ0µV/Vofsupplymismatch, theinverting

bias current changes about 2.ꢀµA/V of supply mismatch.

V

OMAX

P = 2V I

+ 3.5I

+ V – V

) (

(

)

D

S

SMAX

SET

S

OMAX

R

L

2V

P = 2 •12V • 7mA + 3.5 •1mA + 12V – 2V





(

)

(

)

D





150Ω

Power Dissipation

= 0.252+ 0.133 = 0.385W

The worst case amplifier power dissipation is the total of

the quiescent current times the total power supply voltage

plus the power in the IC due to the load. The quiescent

supplycurrentoftheLT1228transconductanceamplifieris

equal to 3.ꢀ times the set current at all temperatures. The

quiescent supply current of the LT1228 current feedback

amplifier has a strong negative temperature coefficient

and at 1ꢀ0°C is less than 7mA, typically only 4.ꢀmA. The

power in the IC due to the load is a function of the output

voltage, the supply voltage and load resistance. The worst

case occurs when the output voltage is at half supply, if

it can go that far, or its maximum value if it cannot reach

half supply.

The total power dissipation times the thermal resistance

of the package gives the temperature rise of the die above

ambient. The above example in SO-8 surface mount pack-

age (thermal resistance is 1ꢀ0°C/W) gives:

Temperature Rise = P

θ

= 0.38ꢀW • 150°C/W

D JA

= ꢀ7.7ꢀ°C

Therefore the maximum junction temperature is 70°C

+ꢀ7.7ꢀ°C or 127.7ꢀ°C, well under the absolute maximum

junction temperature for plastic packages of 1ꢀ0°C.

TYPICAL APPLICATIONS

Basic Gain Control

Lastly the signal is buffered and amplified by the current

feedback amplifier (CFA). The voltage gain of the current

feedback amplifier is:

The basic gain controlled amplifier is shown on the front

page of the data sheet. The gain is directly proportional

to the set current. The signal passes through three stages

from the input to the output.

R_F

1+

R_G

First the input signal is attenuated to match the dynamic

range of the transconductance amplifier. The attenuator

should reduce the signal down to less than 100mV peak.

The characteristic curves can be used to estimate how

much distortion there will be at maximum input signal.

For single ended inputs eliminate R2A or R3A.

The overall gain of the gain controlled amplifier is the

product of all three stages:







R3

R_F





A =

•10•I_SET•R1• 1+









V





R3+ R3A

R

_G

More than one output can be summed into R1 because

the output of the transconductance amplifier is a current.

This is the simplest way to make a video mixer.

The signal is then amplified by the transconductance

amplifier (g ) and referred to ground. The voltage gain

m

of the transconductance amplifier is:

g • R1 = 10 • I • R1

m

SET

1228fd

14

LT1228

TYPICAL APPLICATIONS

Video Fader

Video DC Restore (Clamp) Circuit

NOT NECESSARY IF THE SOURCE RESISTANCE IS LESS THAN 50Ω

1k

3

V

+

IN1

1

200Ω

+

g

m

+

–

V

1000pF

2

LT1223

CFA

–

V

OUT

3

7

+

5

1

8

g

+

–

m

1k

100Ω

2

6

V

0.01µF

–

CFA

5

OUT

4

10k

5.1k

10k

R

F

–

V

=

5V

10k

5.1k

V

–5V

1k

S

R

G

5V

1k

3k

3

2

5

VIDEO

INPUT

V

+

IN2

1

g

m

100Ω

LOGIC

INPUT

–

2N3906

LT1228 • TA12

RESTORE

LT1228 • TA13

The video fader uses the transconductance amplifiers

from two LT1228s in the feedback loop of another cur-

rent feedback amplifier, the LT1223. The amount of signal

from each input at the output is set by the ratio of the

set currents of the two LT1228s, not by their absolute

value. The bandwidth of the current feedback amplifier

is inversely proportional to the set current in this

configuration. Therefore, the set currents remain high

over most of the pot’s range, keeping the bandwidth over

1ꢀMHz even when the signal is attenuated 20dB. The pot

is set up to completely turn off one LT1228 at each end

of the rotation.

The video restore (clamp) circuit restores the black level

of the composite video to zero volts at the beginning of

every line. This is necessary because AC coupled video

changes DC level as a function of the average brightness

of the picture. DC restoration also rejects low frequency

noise such as hum.

The circuit has two inputs: composite video and a logic

signal. The logic signal is high except during the back

porch time right after the horizontal sync pulse. While

the logic is high, the PNP is off and I is zero. With I

SET

equal to zero the feedback to Pin 2 has no affect. The

video input drives the noninverting input of the current

feedback amplifier whose gain is set by R and R . When

F

G

the logic signal is low, the PNP turns on and I goes to

SET

about 1mA. Then the transconductance amplifier charges

the capacitor to force the output to match the voltage at

Pin 3, in this case zero volts.

ThiscircuitcanbemodifiedsothatthevideoisDCcoupled

by operating the amplifier in an inverting configuration.

Just ground the video input shown and connect R to the

G

video input instead of to ground.

1228fd

15

LT1228

TYPICAL APPLICATIONS

Single Supply Wien Bridge Oscillator

3 at resonance; therefore the attenuation of the 1.8k resis-

tor and the transconductance amplifier must be about 11,

resulting in a set current of about 600µA at oscillation. At

start-upthereisnosetcurrentandthereforenoattenuation

for a net gain of about 11 around the loop. As the output

oscillation builds up it turns on the PNP transistor which

generates the set current to regulate the output voltage.

100Ω

2N3906

+

V

6V TO 30V

+

V

+

470Ω

10µF

1

10kΩ

7

3

+

–

5

12MHz Negative Resistance LC Oscillator

g

+

m

0.1µF

51Ω

2

6

CFA

+

V

8

4

–

9.1k

3

2

7

R

680Ω

+

–

F

1

8

V

g

–

O

+

–

m

1k

V

51Ω

O

6

R

5

G

CFA

1.8k

20Ω

4

+

160Ω

1000pF

10µF

1000pF

V

750Ω

+

1k

50Ω

10µF

160Ω

50Ω

4.7µH

30pF

4.3k

330Ω

10k

2N3906

f = 1MHz

= 6dBm (450mV

V

)

RMS

O

2N3904

2nd HARMONIC = –38dBc

3rd HARMONIC = –54dBc

FOR 5V OPERATION SHORT OUT 100Ω RESISTOR

0.1µF

LT1228 • TA14

In this application the LT1228 is biased for operation from

a single supply. An artificial signal ground at half supply

voltage is generated with two 10k resistors and bypassed

–

V

= 10dB

O

AT V = ±5V ALL HARMONICS 40dB DOWN

AT V = ±12V ALL HARMONICS 50dB DOWN

S

LT1228 • TA15

with a capacitor. A capacitor is used in series with R to

G

This oscillator uses the transconductance amplifier as a

negative resistor to cause oscillation. A negative resistor

results when the positive input of the transconductance

amplifier is driven and the output is returned to it. In

this example a voltage divider is used to lower the signal

level at the positive input for less distortion. The negative

resistor will not DC bias correctly unless the output of the

transconductance amplifier drives a very low resistance.

Here it sees an inductor to ground so the gain at DC is

zero. The oscillator needs negative resistance to start

and that is provided by the 4.3k resistor to Pin ꢀ. As the

output level rises it turns on the PNP transistor and in turn

the NPN which steals current from the transconductance

amplifier bias input.

set the DC gain of the current feedback amplifier to unity.

The transconductance amplifier is used as a variable

resistor to control gain. A variable resistor is formed by

driving the inverting input and connecting the output back

to it. The equivalent resistor value is the inverse of the

gm. This works with the 1.8k resistor to make a variable

attenuator. The 1MHz oscillation frequency is set by the

Wien bridge network made up of two 1000pF capacitors

and two 160Ω resistors.

For clean sine wave oscillation, the circuit needs a net gain

ofonearoundtheloop. Thecurrentfeedbackamplifierhas

a gain of 34 to keep the voltage at the transconductance

amplifier input low. The Wien bridge has an attenuation of

1228fd

16

LT1228

TYPICAL APPLICATIONS

Filters

Single Pole Low/High/Allpass Filter

R3A

V

1k

IN

3

LOWPASS

+

INPUT

1

g

+

m

R3

120Ω

2

C

6

V

–

CFA

OUT

330pF

5

8

–

I

SET

R

G

F

V

1k

IN

HIGHPASS

INPUT

R2A

1k

10

2π

9

I

R + 1

R2

R2 + R2A

R2

120Ω

SET

C

F

f

=

×

C

R

G

LT1228 • TA16

f

= 10 I FOR THE VALUES SHOWN

C

SET

Allpass Filter Phase Response

0

–45

1mA SET CURRENT

–90

–135

–180

100µA SET CURRENT

10k

100k

1M

10M

FREQUENCY (Hz)

LT1228 • TA17

Using the variable transconductance of the LT1228 to

make variable filters is easy and predictable. The most

straight forward way is to make an integrator by putting a

capacitor at the output of the transconductance amp and

buffering it with the current feedback amplifier. Because

the input bias current of the current feedback amplifier

must be supplied by the transconductance amplifier, the

setcurrentshouldnotbeoperatedbelow10µA. Thislimits

the filters to about a 100:1 tuning range.

values shown give a 100kHz corner frequency for 100µA

set current. The circuit has two inputs, a lowpass filter

input and a highpass filter input. To make a lowpass filter,

ground the highpass input and drive the lowpass input.

Conversely for a highpass filter, ground the lowpass input

and drive the highpass input. If both inputs are driven, the

result is an allpass filter or phase shifter. The allpass has

flat amplitude response and 0° phase shift at low frequen-

cies, going to –180° at high frequencies. The allpass filter

has –90° phase shift at the corner frequency.

The Single Pole circuit realizes a single pole filter with a

corner frequency (f ) proportional to the set current. The

C

1228fd

17

LT1228

TYPICAL APPLICATIONS

Voltage Controlled State Variable Filter

+

–

1k

LT1006

10k

2N3906

V

C

100pF

180Ω

51k

5V

3k

–5V

3k

7

3.3k

100Ω

3

2

+

V

5

IN

1

8

g

+

m

6

BANDPASS

OUTPUT

–

CFA

4

18pF

–

–5V

1k

3.3k

5

100Ω

5V

7

3

2

+

–

1

8

g

100Ω

+

m

LOWPASS

OUTPUT

6

CFA

4

18pF

3.3k

–

–5V

1k

f

= 100kHz AT V = 0V

C

O

= 200kHz AT V = 1V

C

= 400kHz AT V = 2V

C

= 800kHz AT V = 3V

C

= 1.6MHz AT V = 4V

LT1228 • TA18

C

The state variable filter has both lowpass and bandpass

outputs.EachLT1228isconfiguredasavariableintegrator

whose frequency is set by the attenuators, the capacitors

and the set current. Because the integrators have both

positive and negative inputs, the additional op amp nor-

mally required is not needed. The input attenuators set

for best accuracy. If discrete transistors are used, the

51k resistor should be trimmed to give proper frequency

response with V equal zero. The circuit generates 100µA

C

for V equal zero volts and doubles the current for every

C

additional volt. The two 3k resistors divide the current

between the two LT1228s. Therefore the set current of

each amplifier goes from 50µA to 800µA for a control

the circuit up to handle 3V

signals.

P–P

voltage of 0V to 4V. The resulting filter is at 100kHz for V

C

The set current is generated with a simple circuit that

gives logarithmic voltage to current control. The two PNP

transistors should be a matched pair in the same package

equal zero, and changes it one octave/V of control input.

1228fd

18

LT1228

PACKAGE DESCRIPTION

Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.

J8 Package

3-Lead CERDIP (Narrow .300 Inch, Hermetic)

(Reference LTC DWG # 05-08-1110)

.405

(10.287)

MAX

CORNER LEADS OPTION

(4 PLCS)

.005

(0.127)

MIN

6

5

4

8

7

2

.023 – .045

(0.584 – 1.143)

HALF LEAD

OPTION

.025

(0.635)

RAD TYP

.220 – .310

(5.588 – 7.874)

.045 – .068

(1.143 – 1.650)

FULL LEAD

OPTION

1

3

.200

.300 BSC

(5.080)

MAX

(7.62 BSC)

.015 – .060

(0.381 – 1.524)

.008 – .018

(0.203 – 0.457)

0° – 15°

.045 – .065

(1.143 – 1.651)

.125

3.175

MIN

NOTE: LEAD DIMENSIONS APPLY TO SOLDER DIP/PLATE

OR TIN PLATE LEADS

.014 – .026

(0.360 – 0.660)

.100

(2.54)

BSC

J8 0801

OBSOLETE PACKAGE

1228fd

19

LT1228

PACKAGE DESCRIPTION

Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.

N Package

8-Lead PDIP (Narrow .300 Inch)

(Reference LTC DWG # 05-08-1510 Rev I)

.400*

(10.160)

MAX

.130 ±.005

.300 – .325

.045 – .065

(3.302 ±0.127)

(1.143 – 1.651)

(7.620 – 8.255)

8

1

7

6

5

4

.065

(1.651)

TYP

.255 ±.015*

(6.477 ±0.381)

.008 – .015

(0.203 – 0.381)

.120

.020

(0.508)

MIN

(3.048)

MIN

+.035

–.015

2

3

.325

.018 ±.003

(0.457 ±0.076)

.100

(2.54)

BSC

+0.889

8.255

N8 REV I 0711

(

)

–0.381

NOTE:

INCHES

1. DIMENSIONS ARE

MILLIMETERS

*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.

MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED .010 INCH (0.254mm)

S8 Package

8-Lead Plastic Small Outline (Narrow .150 Inch)

(Reference LTC DWG # 05-08-1610 Rev G)

.189 – .197

(4.801 – 5.004)

.045 ±.005

NOTE 3

.050 BSC

7

5

8

6

.245

MIN

.160 ±.005

.150 – .157

(3.810 – 3.988)

NOTE 3

.228 – .244

(5.791 – 6.197)

.030 ±.005

TYP

1

3

4

2

RECOMMENDED SOLDER PAD LAYOUT

.010 – .020

(0.254 – 0.508)

× 45°

.053 – .069

(1.346 – 1.752)

.004 – .010

(0.101 – 0.254)

.008 – .010

(0.203 – 0.254)

0°– 8° TYP

.016 – .050

(0.406 – 1.270)

.050

(1.270)

BSC

.014 – .019

(0.355 – 0.483)

TYP

NOTE:

INCHES

1. DIMENSIONS IN

(MILLIMETERS)

2. DRAWING NOT TO SCALE

3. THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.

MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED .006" (0.15mm)

4. PIN 1 CAN BE BEVEL EDGE OR A DIMPLE

SO8 REV G 0212

1228fd

20

LT1228

REVISION HISTORY ^{(Revision history begins at Rev D)}

REV

DATE

06/12 Updated Order Information table to new format

Clarified units used in g = 10 • I relationship

DESCRIPTION

PAGE NUMBER

D

2

9

m

SET

1228fd

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-

tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.

21

LT1228

TYPICAL APPLICATIONS

RF AGC Amplifier (Leveling Loop)

15V

7

10k

RF INPUT

to 1.3V

3

2

+

–

0.6V

RMS

25MHz

1

g

+

100Ω

m

OUTPUT

P–P

300Ω

CFA

2V

5

8

–

4

470Ω

0.01µF

10k

–15V

10k

4pF

10

Ω

0.01µF

15V

10k

100k

4.7k

–

–15V

A3

LT1006

AMPLITUDE

ADJUST

1N4148’s

COUPLE THERMALLY

LT1004

1.2V

+

LT1228 • TA20

–15V

Inverting Amplifier with DC Output Less Than 5mV

Amplitude Modulator

5V

4.7µF

+

V

2

3

7

–

+

3

2

7

1

8

+

–

g

+

–

m

1

+

6

g

+

5

m

CFA

V

O

V

OUT

100µF

6

5

CFA

4

0dBm(230mV) AT

R5

CARRIER

INPUT

30mV

MODULATION = 0V

8

4

–

10k

1k

V

–

R

1k

F

R

750Ω

F

V

=

5Vꢀ R5 = 3ꢁ6k

15Vꢀ R5 = 13ꢁ6k

MUST BE LESS THAN

4.7µF

S

+

R

G

R

1k

–5V

G

OUT

200mV

750Ω

FOR LOW OUTPUT OFFSET

P–P

BW = 30Hz TO 20MHz

MODULATION

INPUT ≤ 8V

V

LT1228 • TA22

IN

INCLUDES DC

LT1228 • TA21

P–P

RELATED PARTS

PART NUMBER

DESCRIPTION

COMMENTS

LT1227

140MHz Current Feedback Amplifier

40MHz Video Fader

1100V/µs Slew Rate, 0.01% Differential Gain, 0.03% Differential Phase

Accurate Linear Gain Control: 1% Typ, 3% Max

800V/µs Slew Rate, 80mA Output Current

LT1251/LT1256

LT1399

400MHz Current Feedback Amplifier

1228fd

LT 0612 REV D • PRINTED IN USA

LinearTechnology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

22

●

 LINEAR TECHNOLOGY CORPORATION 2012

(408) 432-1900 FAX: (408) 434-0507 www.linear.com


型号：	LT1228_12
厂家：	Linear
描述：	100MHz Current Feedback 100MHz的电流反馈
文件：	总22页 (文件大小：660K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

LT1228_12 [Linear]

相关型号：

LT1229

LT1229C

LT1229CJ8

LT1229CN8

LT1229CN8#PBF

LT1229CS8

LT1229CS8#PBF

LT1229CS8#TRPBF

LT1229M

LT1229MJ8

LT1229MJ8/883B

LT1229MS8