LT1507CS8-3.3#TR_技术文档

LT1507

500kHz Monolithic

Buck Mode Switching Regulator

FEATURES

■

with all the necessary oscillator, control and logic cir-

cuitry. High switching frequency allows a considerable

reductioninthesizeofexternalcomponents.Thetopology

is current mode for fast transient response and good loop

stability. Both fixed output voltage (3.3V) and adjustable

parts are available.

Constant 500kHz Switching Frequency

■

Uses All Surface Mount Components

■

Operates with Inputs as Low as 4V

Saturated Switch Design (0.3Ω)

Cycle-by-Cycle Current Limiting

Easily Synchronizable

Inductor Size as Low as 2µH

Shutdown Current: 20µA

■

A special high speed bipolar process and new design

techniques allow this regulator to achieve high efficiency

at a high switching frequency. Efficiency is maintained

over a wide output current range by keeping quiescent

supply current to 4mA and by utilizing a supply boost

capacitor to allow the NPN power switch to saturate. A

shutdown signal will reduce supply current to 20µA. The

LT1507 can be externally synchronized from 570kHz to

1MHz with logic level inputs.

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APPLICATIONS

■

Portable Computers

Battery-Powered Systems

Battery Charger

Distributed Power

■

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The LT1507 fits into standard 8-pin SO and PDIP pack-

ages. Temperature rise is kept to a minimum by the high

efficiency design. Full cycle-by-cycle short-circuit protec-

tionandthermalshutdownareprovided.Standardsurface

mount external parts are used including the inductor and

capacitors.

DESCRIPTION

TheLT^®1507isa500kHzmonolithicbuckmodeswitching

regulator, functionally identical to the LT1375 but opti-

mized for lower input voltage applications. It will operate

over a 4V to 15V input range, compared with 5.5V to 25V

for the LT1375. A 1.5A switch is included on the die along

, LTC and LT are registered trademarks of Linear Technology Corporation.

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TYPICAL APPLICATION

5V to 3.3V Volt Down Converter

5V to 3.3V Efficiency

100

D2^†

1N914

V

IN

V

OUT

= 5V

= 3.3V

90

80

70

60

50

C2

0.1µF

BOOST

OUTPUT

3.3V

5V

V

IN

V

SW

L1***

5µH

1.25A

LT1507-3.3

C3*

47µF

16V

+

DEFAULT

(OPEN)

= ON

SHDN

GND

SENSE

TANTALUM

V

C

C1**

100µF

10V

D1

1N5818

+

C

3.3nF

TANTALUM

AVX TPSD477M016R0150 OR SPRAGUE 593D EQUIVALENT.

RIPPLE CURRENT RATING ≥ 0.6A

*

0

0.25

0.50

0.75

1.00

1.25

AVX TPSD108M010R0100 OR SPRAGUE 593D EQUIVALENT

COILTRONICS CTX5-1. SUBSTITUTION UNITS SHOULD BE RATED

AT ≥ 1.25A, USING LOW LOSS CORE MATERIAL

**

***

LOAD CURRENT (A)

LT1507 • TA02

†

SEE BOOST PIN CONSIDERATIONS IN APPLICATIONS INFORMATION

SECTION FOR ALTERNATIVE D2 CONNECTION

1

LT1507

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ABSOLUTE MAXIMUM RATINGS

PACKAGE/ORDER INFORMATION

Input Voltage ........................................................... 16V

Boost Pin Voltage .................................................... 25V

Shutdown Pin Voltage ............................................... 7V

FB Pin Voltage (Adjustable Part)............................. 3.5V

FB Pin Current (Adjustable Part)............................. 1mA

Sense Voltage (Fixed 3.3V Part) ................................ 5V

Sync Pin Voltage ....................................................... 7V

Operating Ambient Temperature Range

LT1507C.................................................. 0°C to 70°C

LT1507I .............................................. –40°C to 85°C

Max Operating Junction Temperature................... 125°C

Storage Temperature Range ................. –65°C to 150°C

Lead Temperature (Soldering, 10 sec).................. 300°C

ORDER PART

NUMBER

TOP VIEW

LT1507CN8

BOOST

1

2

3

4

8

7

6

5

V

C

LT1507CN8-3.3

LT1507CS8

V

FB/SENSE

GND

IN

V

SW

LT1507CS8-3.3

LT1507IN8

SHDN

SYNC

LT1507IN8-3.3

LT1507IS8

N8 PACKAGE

8-LEAD PDIP

S8 PACKAGE

8-LEAD PLASTIC SO

LT1507IS8-3.3

T_JMAX= 125°C, θ_JA= 80°C/W TO 120°C/ W (N)

T_JMAX= 125°C, θ_JA= 120°C/W TO 170°C/ W (S)

DEPENDING ON PC BOARD LAYOUT

S8 PART MARKING

1507

1507I

15073 1507I3

Consult factory for Military grade parts.

ELECTRICAL CHARACTERISTICS

T = 25°C, V = 5V, V = 1.5V, boost open, switch open unless otherwise specified.

J

IN

C

PARAMETER

CONDITIONS

All Conditions

MIN

TYP

MAX

UNITS

Reference Voltage (Adjustable)

2.39

2.36

3.25

3.23

2.42

2.45

2.48

3.35

3.37

9.5

0.03

2

V

kΩ

%/V

µA

●

Sense Voltage (3.3V)

3.3

●

Sense Pin Resistance

Reference Voltage Line Regulation

FB Input Bias Current

4.0

6.6

0.01

0.5

4.3V ≤ V ≤ 15V

IN

Error Amplifier Voltage Gain (Note 8)

Error Amplifier Transconductance (Note 8) ∆I(V ) = ±10µA

(Note 1)

150

1500

1100

400

2000

2700

3000

µmho

C

●

V Pin to Switch Current

C

Transconductance

Error Amplifier Source Current

Error Amplifier Sink Current

2

225

2

A/V

µA

mA

V

A

V

= 2.1V or V

= 2.7V or V

= 2.9V

= 3.7V

150

320

FB

SENSE

V Pin Switching Threshold

Duty Cycle = 0

= 2.1V or V

0.9

2.1

2

C

V Pin High Clamp

V

= 2.9V

C

FB

SENSE

Switch Current Limit

V Open, V = 2.1V or V

= 2.9V DC ≤ 50%

●

1.50

1.35

3

C

FB

SENSE

V

≥ 5V, V

= V + 5V

DC = 80%

IN

BOOST

IN

Switch On Resistance (Note 6)

Maximum Switch Duty Cycle

I

= 1.5A, V

= V + 5V

0.3

0.4

0.5

Ω

SW

BOOST

IN

●

V

= 2.1V or V

= 2.9V

90

86

93

%

FB

SENSE

2

LT1507

ELECTRICAL CHARACTERISTICS

T = 25°C, V = 5V, V = 1.5V, boost open, switch open unless otherwise specified.

J

IN

C

PARAMETER

Switch Frequency

CONDITIONS

V Set to Give 50% Duty Cycle

MIN

TYP

500

MAX

UNITS

460

440

540

560

570

kHz

C

–25°C ≤ T ≤ 125°C

J

T ≤ –25°C

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Switch Frequency Line Regulation

Frequency Shifting Threshold on F Pin

Minimum Input Voltage (Note 2)

Minimum Boost Voltage (Note 3)

Boost Current (Note 4)

4.3V ≤ V ≤ 15V

∆f = 10kHz

●

0.05

1.0

4

3

12

0.15

1.3

4.3

3.5

22

25

35

40

%/V

V

IN

0.8

B

V

mA

I

≤ 1.5A

SW

V

= V + 5V

I

= 500mA, –25°C ≤ T ≤ 125°C

BOOST

IN

SW

J

T ≤ –25°C

J

= 1.5A, –25°C ≤ T ≤ 125°C

25

J

T ≤ –25°C

J

Input Supply Current (Note 5)

Shutdown Supply Current

●

3.8

15

5.4

50

75

mA

µA

V

= 0V, V ≤ 12V

SHDN

IN

= 0V, V Open

●

SW

C

Lockout Threshold

V Open

C

2.3

2.38

2.46

V

Shutdown Threshold

V Open

C

Device Shutting Down

Device Starting Up

●

0.15

0.25

0.37

0.45

0.70

V

Minimum Synchronizing Amplitude

●

1.5

2.2

V

Synchronizing Frequency Range (Note 7)

580

1000

kHz

The

range.

●

denotes specifications which apply over the operating temperature

Note 5: Input supply current is the bias current drawn by the V pin when

the SHDN pin is held at 1V (switching disabled).

IN

Note 1: Gain is measured with a V swing equal to 200mV above the low

clamp level to 200mV below the upper clamp level.

Note 6: Switch ON resistance is calculated by dividing V to V voltage

IN SW

by the forced current (1.5A). See Typical Performance Characteristics for

the graph of switch voltage at other currents.

Note 7: For synchronizing frequency above 700kHz, with duty cycles

above 50%, external slope compensation may be needed. See Applications

Information.

Note 8: Transconductance and voltage gain refer to the internal amplifier

exclusive of the voltage divider. To calculate gain and transconductance

refer to SENSE pin on fixed voltage parts. Divide values shown by the ratio

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Note 2: Minimum input voltage is not measured directly, but is guaranteed

by other tests. It is defined as the voltage where internal bias lines are still

regulated, so that the reference voltage and oscillator frequency remain

constant. Actual minimum input voltage to maintain a regulated output will

depend on output voltage and load current. See Applications Information.

Note 3: This is the minimum voltage across the boost capacitor needed to

guarantee full saturation of the internal power switch.

V

/2.42.

OUT

Note 4: Boost current is the current flowing into the BOOST pin with the

pin held 5V above input voltage. It flows only during switch ON time.

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TYPICAL PERFORMANCE CHARACTERISTICS

V_CPin Shutdown Threshold

Switch Peak Current Limit

Feedback Pin Voltage and Current

2.44

2.43

2.42

2.41

2.40

2.0

1.5

1.0

0.5

0

100

90

80

70

60

50

1.4

1.2

1.0

0.8

0.6

0.4

V

IN

V

OUT

= 5V

= 3.3V

VOLTAGE

CURRENT

–25

0

25

50

75

125

–50

100

0

0.25

0.50

0.75

1.00

1.25

–25

0

25

50

75

125

–50

100

JUNCTION TEMPERATURE (°C)

LOAD CURRENT (A)

JUNCTION TEMPERATURE (°C)

LT1507 • TPC03

LT1507 • TA02

LT1507 • TPC01

3

LT1507

TYPICAL PERFORMANCE CHARACTERISTICS

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Shutdown Pin Bias Current

Standby and Shutdown Thresholds

Shutdown Supply Current

30

25

2.40

2.36

2.32

0.8

0.4

0

500

400

300

200

8

V

SHDN

= 0V

CURRENT REQUIRED TO FORCE SHUTDOWN

(FLOWS OUT OF PIN). AFTER SHUTDOWN,

CURRENT DROPS TO A FEW µA

STANDBY

20

15

10

5

STARTUP

AT 2.38V STANDBY THRESHOLD

(CURRENT FLOWS OUT OF PIN)

4

SHUTDOWN

0

50

JUNCTION TEMPERATURE (°C)

100 125

0

3

6

9

12

15

50

TEMPERATURE (°C)

100 125

–50 –25

0

25

75

–50 –25

0

25

75

INPUT VOLTAGE (V)

LT1507 • TPC06

LT1507 • TPC05

L11507 • TPC04

Shutdown Supply Current

Error Amplifier Transconductance

2500

2000

1500

1000

500

3000

2500

2000

1500

1000

500

200

150

100

50

150

125

100

75

PHASE

GAIN

V

C

OUT

12pF

R

OUT

200k

–3

V

FB

× 2e

V

= 10V

50

IN

ERROR AMPLIFIER EQUIVALENT CIRCUIT

0

25

R

LOAD

= 50Ω

0

–50

0

–50

0

25

50

75 100 125

100

1k

10k

100k

1M

10M

–25

0

0.1

0.2

0.3

0.4

0.5

JUNCTION TEMPERATURE (°C)

FREQUENCY (Hz)

SHUTDOWN VOLTAGE (V)

LT1507 • TPC09

LT1507 • TPC08

LT1507 • TPC07

Minimum Input Voltage

with 3.3V Output

Frequency Foldback

Switching Frequency

600

550

500

450

400

6.5

6.0

5.5

5.0

4.5

4.0

3.5

3.0

500

400

300

200

100

0

MINIMUM VOLTAGE

TO START WITH

STANDARD CIRCUIT

SWITCHING

FREQUENCY

MINIMUM VOLTAGE

TO RUN WITH

FEEDBACK PIN

CURRENT

STANDARD CIRCUIT

–25

0

25

50

75

125

–50

100

1

10

100

1000

0

0.5

1.0

1.5

2.0

2.5

JUNCTION TEMPERATURE (°C)

LOAD CURRENT (mA)

FEEDBACK PIN VOLTAGE (V)

LT1507 • TPC12

LT1507 • TPC11

LT1507 • TPC10

MINIMUM INPUT VOLTAGE CAN BE REDUCED

BY ADDING A SMALL EXTERNAL PNP. SEE

APPLICATIONS INFORMATION

4

LT1507

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TYPICAL PERFORMANCE CHARACTERISTICS

Maximum Load Current

at V_OUT= 5V

at V_OUT= 3.3V

Current Limit Foldback

2.5

2.0

1.5

1.0

0.5

0

1.50

1.25

1.50

1.25

V

OUT

= 3.3V

L = 20µH

L = 10µH

FOLDBACK

*POSSIBLE

UNDESIRED

CHARACTERISTICS

L = 10µH

L = 5µH

STABLE POINT

FOR CURRENT

SOURCE LOAD

CURRENT

SOURCE LOAD

1.00

0.75

1.00

0.75

L = 5µH

L = 3µH

0.50

0.25

0

0.50

0.25

0

L = 2µH

MOS LOAD

RESISTOR LOAD

0

20

40

60

80

100

4

6

8

10

12

14

0

3

6

9

12

15

INPUT VOLTAGE (V)

OUTPUT VOLTAGE (%)

LT1507 • TPC13

LT1507 • TPC14

LT1507 • TPC15

*SEE "MORE THAN JUST VOLTAGE FEEDBACK"

IN APPLICATIONS INFORMATION SECTION

Inductor Core Loss for 3.3V Output

Boost Pin Current

Switch Voltage Drop

0.8

0.6

0.4

0.2

0

12

10

8

1.0

0.1

V

OUT

= 3.3V

OUT

IN

T

= 25°C

T

= 25°C

J

= 5V

I

= 1A

TYPE 52 POWDERED IRON

Kool Mµ^®

6

0.01

4

PERMALLOY

Metglas^®

µ = 125

2

0

0.001

0

0.25 0.50 0.75 1.00 1.25 1.50

SWITCH CURRENT (A)

0

0.25

0.50

0.75

1.00

1.25

1

2

4

6

8

10

SWITCH CURRENT (A)

INDUCTANCE (µH)

LT1507 • TPC17

LT1507 • TPC18

LT1507 • TPC16

CORE LOSS IS INDEPENDENT OF LOAD CURRENT

UNTIL LOAD CURRENT FALLS LOW ENOUGH

FOR CIRCUIT TO GO INTO DISCONTINUOUS MODE

Kool Mµ is a registered trademark of Magnetics, Incorporated.

Metglas is a registered trademark of AlliedSignal Incorporated.

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PIN FUNCTIONS

BOOST (Pin 1): The BOOST pin is used to provide a drive

voltage, higher than the input voltage, to the internal

bipolar NPN power switch. Without this added voltage the

typical switch voltage loss would be about 1.5V. The

additional boost voltage allows the switch to saturate and

voltage loss approximates that of a 0.3Ω FET structure,

butwithamuchsmallerdiearea.Efficiencyimprovesfrom

70% forconventionalbipolardesignstogreater than 85%

for these new parts.

V_IN(Pin 2): Input Pin. The LT1507 is designed to operate

withaninputvoltagebetween4.5Vand15V.Undercertain

conditions, input voltage may be reduced down to 4V.

Actual minimum operating voltage will always be higher

than the output voltage. It may be limited by switch

5

LT1507

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PIN FUNCTIONS

saturation voltage and maximum duty cycle. A typical

value for minimum input voltage is 1V above output

voltage. Start-up conditions may require more voltage at

light loads. See Minimum Input Voltage for details.

SYNC (Pin 5): The SYNC pin is used to synchronize the

internal oscillator to an external signal. It is directly logic

compatible and can be driven with any signal between

10% and 90% duty cycle. The synchronizing range is

equal to initial operating frequency up to 1MHz. See

Sychronizing section for details.

V

_SW(Pin3):Theswitchpinisdrivenuptotheinputvoltage

in the ON state and is an open circuit in the OFF state. At

higher load currents, pin voltage during the off condition

will be one diode drop below ground as set by the external

catch diode. At lighter loads the pin will assume an

intermediate state equal to output voltage during part of

the switch OFF time. Maximum negative voltage on the

switch pin is 1V with respect to the GND pin, so it must

always be clamped with a catch diode to the GND pin.

FB/SENSE (Pin 7): The feedback pin is used to set output

voltage using an external voltage divider that generates

2.42V at the pin with the desired output voltage. The fixed

voltage(–3.3V)partshavethedividerincludedonthechip

and the feedback pin is used as a sense pin connected

directly to the 5V output. Two additional functions are

performed by the feedback pin. When the pin voltage

drops below 1.7V, switch current limit is reduced. Below

1V, switching frequency is also reduced. See More Than

Just Voltage Feedback.

SHDN (Pin 4): The shutdown pin is used to turn off the

regulator and to reduce input drain current to a few

microamperes. Actually this pin has two separate thresh-

olds, one at 2.38V to disable switching and a second at

0.4V to force complete micropower shutdown. The 2.38V

threshold functions as an accurate undervoltage lockout

(UVLO). This is sometimes used to prevent the regulator

from delivering power until the input voltage has reached

a predetermined level.

V_C(Pin 8): The V_Cpin is the output of the error amplifier

and the input of the peak switch current comparator. It is

normally used for frequency compensation but can do

double duty as a current clamp or control loop override.

This pin sets at about 1V for very light loads and 2V at

maximum load. It can be driven to ground to shut off the

regulator, but if driven high, current must be limited to 4mA.

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BLOCK DIAGRAM

The LT1507 is a constant frequency, current mode buck

converter. This means that there is an internal clock and

twofeedbackloopsthatcontrolthedutycycleofthepower

switch. In addition to the normal error amplifier, there is

a current sense amplifier that monitors switch current on

a cycle-by-cycle basis. A switch cycle starts with an

oscillator pulse which sets the RS flip-flop to turn the

switch on. When switch current reaches a level set by the

inverting input of the comparator, the flip-flop is reset and

the switch turns off. Output voltage control is obtained by

using the output of the error amplifier to set the switch

current trip point. This technique means that the error

amplifier commands current to be delivered to the output

rather than voltage. A voltage fed system will have low

phase shift up to the resonant frequency of the inductor

and output capacitor, then an abrupt 180° shift will occur.

The current fed system will have 90° phase shift at a much

lower frequency, but will not have the additional 90° shift

until well beyond the LC resonant frequency. This makes

itmucheasiertofrequencycompensatethefeedbackloop

and also gives much quicker transient response.

High switch efficiency is attained by using the BOOST pin

to provide a voltage to the switch driver which is higher

than the input voltage, allowing the switch to be saturated.

This boosted voltage is generated with an external capaci-

tor and diode.

Two comparators are connected to the shutdown pin. One

has a 2.38V threshold for undervoltage lockout and the

second has a 0.4V threshold for complete shutdown.

6

LT1507

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BLOCK DIAGRAM

0.1Ω

V

2

IN

+

–

CURRENT

SENSE

2.9V BIAS

REGULATOR

INTERNAL

CC

BIAS

AMPLIFIER

VOLTAGE GAIN = 5

V

BOOST

1

SLOPE COMP

Σ

0.9V

500kHz

OSCILLATOR

5

S

R

SYNC

Q1

POWER

SWITCH

R

DRIVER

CIRCUITRY

S

FLIP-FLOP

+

–

SHUTDOWN

COMPARATOR

V

3

SW

+

–

CURRENT

COMPARATOR

0.4V

FREQUENCY

SHIFT CIRCUIT

SHDN

4

3.5µA

FOLDBACK

CURRENT

LIMIT

Q2

+

–

CLAMP

–

7

6

FB/SENSE

LOCKOUT

COMPARATOR

+

ERROR

AMPLIFIER

= 2000µmho

2.38V

g

m

2.42V

8

V

C

GND

LT1507 • BD

Figure 1. Block Diagram

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APPLICATIONS INFORMATION

Note: This application section is adapted from the more FEEDBACK PIN FUNCTIONS

complete version found in the LT1375/LT1376 data sheet.

The feedback pin (FB or SENSE) on the LT1507 is used to

set output voltage and also to provide several overload

protection features. The first part of this section deals with

selecting resistors to set output voltage and the remaining

part talks about foldback frequency and current limiting

created by the FB pin. Please read both parts before

If more details are desired consult the LT1375/LT1376

Applications Information section, but please acquaint

yourself thoroughly with this LT1507 information first so

that differences between the LT1375 and the LT1507 do

not cause confusion.

7

LT1507

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APPLICATIONS INFORMATION

committing to a final design. The fixed 3.3V LT1507-3.3

has internal divider resistors and the FB pin is renamed

SENSE, connected directly to the output.

pin voltage drops below 1V (see Frequency Foldback

graph). This does not affect operation with normal load

conditions; one simply sees a gear shift in switching

frequency during start-up as the output voltage rises.

ThesuggestedvaluefortheoutputdividerresistorfromFB

to ground (R2) is 5k or less and the formula for R1 is

shownbelow. Theoutputvoltageerrorcausedbyignoring

the input bias current on the FB pin is less than 0.25% with

R2 = 5k. Please read below if R2 is increased above the

suggested value.

In addition to lower switching frequency, the LT1507 also

operates at lower switch current limit when the feedback

pin voltage drops below 1.5V. This foldback current limit

greatly reduces power dissipation in the IC, diode and

inductor during short-circuit conditions. Again, it is nearly

transparent to the user under normal load conditions. The

only loads which may be affected are current source loads

which maintain full-load current with output voltage less

than 50% of final value. In these rare situations, the

feedback pin can be clamped above 1.5V with an external

diode to defeat foldback current limit. Caution: clamping

thefeedbackpinmeansthatfrequencyshiftingwillalsobe

defeated, so a combination of high input voltage and dead

shorted output may cause the LT1507 to lose control of

current limit.

R2(V

– 2.42)

2.42

OUT

R1 =

More Than Just Voltage Feedback

The feedback pin is used for more than just output voltage

sensing. It also reduces switching frequency and current

limit when output voltage is very low (see graph in Typical

Performance Characteristics). This is done to control

power dissipation in both the IC and in the external diode

and inductor during short-circuit conditions. A shorted

output requires the switching regulator to operate at very

low duty cycles and the average current through the diode

andinductorisequaltotheshort-circuitcurrentlimitofthe

switch (typically 2A of the LT1507, folding back to less

than 1A). Minimum switch ON time limitations would

prevent the switcher from attaining a sufficiently low duty

cycle if switching frequency were maintained at 500kHz,

so frequency is reduced by about 5:1 when the feedback

The internal circuitry which forces reduced switching

frequency also causes current to flow out of the feedback

pin when output voltage is low. If the FB pin falls below 1V,

currentbeginstoflowoutofthepinandreducesfrequency

at the rate of approximately 5kHz/µA. To ensure adequate

frequency foldback (under worst-case short-circuit con-

ditions) the external divider Thevinin resistance must be

lowenoughtopull150µAoutoftheFBpinwith0.6Vonthe

D2

1N914

C2

0.1µF

BOOST

OUTPUT

V

IN

SW

FB

5V

L1***

R1

LT1507

C3*

33µF

20V

5µH

+

5.36k

DEFAULT

(OPEN)

= ON

SHDN

GND

TANTALUM

V

C

C1**

100µF

10V

+

D1

1N5818

R2

4.99k

C

3.3nF

C

TANTALUM

AVX TPSD337M020R0200 OR SPRAGUE 593 EQUIVALENT.

RIPPLE CURRENT RATING ≥ 0.6A

*

LT1507 • F01

AVX TPSD108M010R0100 OR SPRAGUE 593 EQUIVALENT

COILTRONICS CTX5-1. SUBSTITUTION UNITS SHOULD BE RATED

AT ≥ 1.25A, USING LOW LOSS CORE MATERIAL. LOAD CURRENTS

ABOVE 0.85A MAY NEED A 10µH OR 20µH INDUCTOR

**

***

Figure 2. Typical Schematic for LT1507 Adjustable Application

8

LT1507

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APPLICATIONS INFORMATION

pin (R_DIV= R1/R2 ≤ 4k). The net result is that reductions

in frequency and current limit are affected by output

voltage divider impedance. Although divider impedance is

not critical, caution should be used if resistors are

increased beyond the suggested values and short-circuit

conditions will occur with high input voltage. High

frequency pickup will also increase and the protection

accordedbyfrequencyandcurrentfoldbackwilldecrease.

may not survive a continuous 1.5A overload condition.

Deadshorts(V_OUT≤1V)willactuallybemoregentleon

the inductor because the LT1507 has foldback current

limiting (see graph in Typical Performance Character-

istics).

2. Calculate peak inductor current at full load current to

ensure that the inductor will not saturate. Peak current

can be significantly higher than output current, espe-

cially with smaller inductors and lighter loads, so don’t

omit this step. Powdered iron cores are forgiving

because they saturate softly, whereas ferrite cores

saturate abruptly. Other core materials fall in between

somewhere. The following formula assumes a con-

tinuous mode of operation, but it errs only slightly on

the high side for discontinuous mode, so it can be used

for all conditions.

CHOOSING THE INDUCTOR AND OUTPUT CAPACITOR

For most applications the value of the inductor will fall in

the range of 2µH to 10µH. Lower values are chosen to

reduce physical size of the inductor. Higher values allow

more output current because they reduce peak current

seen by the LT1507 switch, which has a 1.5A limit. Higher

values also reduce output ripple voltage and reduce core

loss. Graphs in the Typical Performance Characteristics

section show maximum output load current versus induc-

torsizeandinputvoltage. Asecondgraphshowscoreloss

versus inductor size for various core materials.

V

(V – V

)

OUT IN

OUT

I

= I

+

OUT

PEAK

2(f)(L)(V )

IN

V_IN= Maximum input voltage

f = Switching frequency = 500kHz

When choosing an inductor you might have to consider

maximum load current, core and copper losses, allowable

component height, output voltage ripple, EMI, fault cur-

rent in the inductor, saturation and, of course, cost. The

following procedure is suggested as a way of handling

thesesomewhatcomplicatedandconflictingrequirements.

3. Decide if the design can tolerate an “open” core geom-

etry like ferrite rods or barrels, which have high mag-

netic field radiation or whether it needs a closed core

like a toroid to prevent EMI problems. One would not

wantanopencorenexttoamagneticstoragemediafor

instance! This is a tough decision because the rods or

barrelsaretemptinglycheapandsmallandthereareno

helpful guidelines to calculate when the magnetic field

radiationwillbeaproblem. Thefollowingisanexample

of just how subtle the “B” field problems can be with

open geometry cores.

1. Choose a value in microhenries from the graphs of

Maximum Load Current and Inductor Core Loss for

3.3V Output. If you want to double check that the

chosen inductor value will allow sufficient load current,

go to the next section, Maximum Output Load Current.

Choosing a small inductor with lighter loads may result

in discontinuous mode of operation, but the LT1507 is

designed to work well in either mode. Keep in mind that

lower core loss means higher cost, at least for closed-

core geometries like toroids. Type 52 powdered iron,

Kool Mµ and Molypermalloy are old standbys for tor-

oids in ascending order of price. A newcomer, Metglas,

gives very low core loss with high saturation current.

We had selected an open drum shaped ferrite core for

the LTC1376 demonstration board because the induc-

tor was extremely small and inexpensive. It met all the

requirements for current and the ferrite core gave low

core loss. When the boards came back from assembly,

many of them had somewhat higher than expected

output ripple voltage. We removed the inductors and

output capacitors and found them to be no different

than the good boards. After much head scratching and

hours of delicate low level ripple measurements on the

good and bad boards, I realized that the problem must

Assume that the average inductor current is equal to

load current and decide whether or not the inductor

must withstand continuous fault conditions. If maxi-

mum load current is 0.5A, for instance, a 0.5A inductor

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Table 1. Representative Surface Mount Units

be due to a radiated magnetic field coupling into PC

board traces. But why were some boards bad and

others good? In a moment of desperation (or divine

inspiration) I unsoldered a “bad” inductor, rotated it

180° and resoldered it. Problem fixed!!

VALUE

MANUFACTURER (µH)

DC

(A)

CORE SERIES

HEIGHT

(mm)

TYPE

(Ω)

CORE

Coiltronics

CTX5-1

CTX10-1

CTX5-1P

CTX10-1P

5

10

5

2.3

1.9

1.8

1.6

Tor

0.027 KMµ

0.039 KMµ

0.021

4.2

52

It turns out that the inductor was symmetrical in all

regards except that the polarity of the magnetic field

reversed when the unit was rotated 180° because

current flowed in the opposite direction in the coil. In

one direction, the magnetically induced ripple in the

boardtracesadded tooutputripple.Rotatingtheinduc-

tor caused the induced field to reduce output ripple.

Unfortunately the inductor had no physical package

assymmetry to indicate rotation, including part mark-

ing, so we had to visually examine the winding in each

unit before soldering it to the boards. This little horror

story should not preclude the use of open core induc-

tors, but it emphasizes the need to carefully check the

effect these seductively small, low cost inductors may

have on regulator or system performances.

10

0.030

Sumida

CDRH64

CDRH73

CD73

10

1.7

1.4

2.4

SC

0.084

0.055

Fer

4.5

3.4

3.5

4.0

Open 0.062

Open 0.041

CD104

Gowanda

SM20-102K

10

1.3

Open 0.038

Fer

7

Dale

IHSM-4825

IHSM-5832

10

3.1

4.3

Open 0.071

Open 0.053

Fer

5.6

7.1

SC = Semi-closed geometry

Fer = Ferrite core material

52 = Type 52 powdered iron core material

KMµ = Kool Mµ

OUTPUT CAPACITOR

The output capacitor is normally chosen by its effective

series resistance (ESR), because that is what determines

output ripple voltage. At 500kHz any polarized capacitor is

essentially resistive. To get low ESR takes volume; physi-

cally larger capacitors have lower ESR. The ESR range

needed for typical LT1507 applications is 0.05Ω to 0.5Ω.

A typical output capacitor is an AVX type TPS, 100µF at

10V, with a guaranteed ESR less than 0.1Ω. This is a “D”

size surface mount solid tantalum capacitor. TPS capaci-

tors are specially constructed and tested for low ESR so

they give the lowest ESR for a given volume. The value in

microfarads is not particularly critical and values from

22µF to greater than 500µF work well, but you cannot

cheat mother nature on ESR. If you find a tiny 22µF solid

tantalum capacitor, it will have high ESR and output ripple

voltage will be terrible. The chart in Table 2 shows some

typical solid tantalum surface mount capacitors.

4. Look for an inductor (see Table 1) which meets the

requirements of core shape, peak current (to avoid

saturation), average current (to limit heat) and fault

current (if the inductor gets too hot, wire insulation will

melt and cause turn-to-turn shorts). Keep in mind that

all good things like high efficiency, surface mounting,

lowprofileandhightemperatureoperationwillincrease

cost, sometimes dramatically.

5. Aftermakinganinitialchoice,considersecondarythings

like output voltage ripple, second sourcing, etc. Use the

experts in the Linear Technology Applications Depart-

ment if you feel uncertain about the final choice. They

haveexperiencewithawiderangeofinductortypesand

cantellyouaboutthelatestdevelopmentsinlowprofile,

surface mounting, etc.

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Table 2. Surface Mount Solid Tantalum Capacitor ESR

and Ripple Current

appropriate for input bypassing because of their high

ripple current ratings and tolerance of turn-on surges.

E CASE SIZE

AVX TPS, Sprague 593D

AVX TAJ

ESR (MAX Ω)

0.1 to 0.3

RIPPLE CURRENT (A)

0.7 to 1.1

0.4

OUTPUT RIPPLE VOLTAGE

0.7 to 0.9

D CASE SIZE

AVX TPS, Sprague 593D

AVX TAJ

Ripplevoltageisdeterminedbythehighfrequencyimped-

anceoftheoutputcapacitorandripplecurrentthroughthe

inductor. Ripple current is triangular (continuous mode)

with a peak-to-peak value of:

0.1 to 0.3

0.9 to 2.0

0.7 to 1.1

0.36 to 0.24

C CASE SIZE

AVX TPS

0.2 (Typ)

1.8 to 3.0

0.5 (Typ)

(V )(V − V )

OUT

AVX TAJ

0.22 to 0.17

OUT IN

I

=

P-P

(V )(L)(f)

IN

Many engineers have heard that solid tantalum capacitors

are prone to failure if they undergo high surge currents.

This is historically true, and type TPS capacitors are

specially tested for surge capability, but surge rugged-

ness is not a critical issue with the output capacitor. Solid

tantalum capacitors fail during very high turn-on surges

which do not occur at the output of regulators. High

discharge surges, such as when the regulator output is

dead shorted, do not harm the capacitors.

Output ripple voltage is also triangular with peak-to-peak

amplitude of:

V_RIPPLE= (I_P–P)(ESR) (peak-to-peak)

Example:withV_IN=5V, V_OUT=3.3V, L=5µH, ESR=0.1Ω;

(3.3)(5− 3.3)

I

=

= 0.45

P-P

−6

3

5 5 10

500 10

Unlike the input capacitor, RMS ripple current in the

output capacitor is normally low enough that ripple cur-

rent rating is not an issue. The current waveform is

triangularwithatypicalvalueof200mARMS.Theformula

to calculate this is:

V

= (0.45A)(0.1Ω) = 45mV

RIPPLE

P-P

MAXIMUM OUTPUT LOAD CURRENT

Maximum load current will be less than the 1.5A rating of

the LT1507, especially with lower inductor values. Induc-

tor ripple current must be taken into account as well as

reduced switch current at high duty cycles. Maximum

switch current rating (I_P) of the LT1507 is 1.5A up to 50%

duty cycle (DC), decreasing to 1.35A at 80% duty cycle,

shown graphically in Typical Performance Characteristics

and as a formula below. Current rating decreases with

duty cycle because the LT1507 has internal slope com-

pensation to prevent current mode subharmonic switch-

ing. For more details on subharmonic oscillation read

Application Note 19. Peak guaranteed switch current (I_P)

is found from:

Output Capacitor Ripple Current (RMS)

0.29(V )(V – V

)

OUT IN

OUT

I

(RMS) =

RIPPLE

(L)(f)(V )

IN

Ceramic Capacitors

Higher value, lower cost ceramic capacitors are now

becomingavailableinsmallercasesizes.Thesearetempt-

ing for switching regulator use because of their very low

ESR. Unfortunately, the ESR is so low that it can cause

loop stability problems when ceramic is used for the

output capacitor. Solid tantalum capacitor ESR generates

a loop “zero” at 5kHz to 50kHz that is instrumental in

giving acceptable loop phase margin. Ceramic capacitors

remain capacitive to beyond 300kHz and usually resonate

with their ESL before ESR becomes effective. They are

V

OUT

I = 1.5A for

≤ 0.5

P

IN

V

0.5(V

)

OUT

I = 1.75A−

for

≥ 0.5

P

V

IN

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Example: with V_OUT= 3.3V, V_IN= 5V;

V_OUT/V_IN= 3.3/5 = 0.67

Discontinuous mode:

2

(I ) (f)(L)(V )

P

IN

I

=

OUT(MAX)

2(V )(V – V

)

I_P= 1.75 – (0.5)(0.66) = 1.42A

OUT IN

OUT

Maximum load current would be equal to maximum

switch current for an infinitely large inductor, but with

finite inductor size, maximum load current is reduced by

one half peak-to-peak inductor current. The following

formulaassumescontinuousmodeoperation;thetermon

the right must be less than one half of I_P.

Example: with L = 2µH, V_OUT= 5V and V_IN(MAX)= 15V;

2

3

−6

(1.5) 500 10

2 10

15

(

)

(

)

I

=

OUT(MAX)

2(5)(15 – 5)

= 338m

A

Continuous mode:

The main reason for using such a tiny inductor is that it is

physically very small, but keep in mind that peak-to-peak

inductorcurrentwillbeveryhigh. Thiswillincreaseoutput

ripplevoltage.Iftheoutputcapacitorhastobemadelarger

to reduce ripple voltage, the overall circuit could actually

be larger.

(V )(V – V

)

OUT IN

OUT

I

= I –

P

OUT(MAX)

2(L)(f)(V )

IN

For the conditions above, with L = 5µH and f = 500kHz;

(3.3)(5 – 3.3)

I

= 1.42 –

OUT(MAX)

−6

3

2 5 10

500 10

5

(

)

(

)

CATCH DIODE

= 1.42 – 0.22 = 1.2A

The suggested catch diode (D1) is a 1N5818 Schottky or

its Motorola equivalent, MBR130. It is rated at 1A average

forward current and 30V reverse voltage. Typical forward

voltage is 0.42V at 1A. The diode conducts current only

during switch OFF time. Peak reverse voltage is equal to

regulatorinputvoltage.Averageforwardcurrentinnormal

operation can be calculated from:

At V_IN= 8V, V_OUT/V_IN= 0.41, so I_Pis equal to 1.5A and

I_OUT(MAX)is equal to;

(3.3)(8 – 3.3)

1.5 –

−6

3

2 5 10

500 10

8

(

)

(

)

= 1.5 – 0.39 = 1.11A

I

(V – V

)

OUT IN

OUT

I

=

D(AVG)

V

Note that there is less load current available at the higher

input voltage because inductor ripple current increases.

This is not always the case. Certain combinations of

inductor value and input voltage range may yield lower

available load current at the lowest input voltage due to

reduced peak switch current at high duty cycles. If load

current is close to the maximum available, please check

maximum available current at both input voltage

extremes. To calculate actual peak switch current with a

given set of conditions, use:

IN

This formula will not yield values higher than 1A with

maximumloadcurrentof1.25Aunlesstheratioofinputto

output voltage exceeds 5:1. The only reason to consider a

larger diode is the worst-case condition of a high input

voltageand overloaded(notshorted)output. Undershort-

circuit conditions, foldback current limit will reduce diode

current to less than 1A, but if the output is overloaded and

does not fall to less than 1/3 of nominal output voltage,

foldback will not take effect. With the overloaded condi-

tion, output current will increase to a typical value of 1.8A,

determined by peak switch current limit of 2A. With V_IN=

10V, V_OUT= 2V (3.3V overloaded) and I_OUT= 1.8A:

V

(V – V

2(L)(f)(V )

)

OUT IN

OUT

I

= I

+

OUT

SWITCH(PEAK)

IN

Forlighterloadswherediscontinuousmodeoperationcan

be used, maximum load current is equal to:

1.8(10 – 2)

I

=

= 1.44A

D(AVG)

10

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The anode of the diode can be connected to the regulated

output voltage or the unregulated input voltage. The

“boost voltage” generated across the boost capacitor is

then nearly identical to the anode voltage. The input

connection minimizes start-up problems and gives plenty

of boost voltage, but efficiency is slightly lower, especially

with input voltages above 10V. For 5V to 3.3V operation,

or any output voltage less than 3.3V, the diode should be

connected to the input. With input voltage more than 3V

above the output and an output voltage of at least 3.3V the

output connection will give better efficiency. Use the

BAT85 Schottky diode for 3.3V applications where the

anode is connected to the output.

This is safe for short periods of time, but it would be

prudent to check with the diode manufacturer if continu-

ous operation under these conditions must be tolerated.

BOOST PIN CONSIDERATIONS

Formostapplications, theboostcomponentsarea0.22µF

capacitorandanMBR0520orBAT85Schottkydiode. This

capacitor value is twice that suggested for the LT1376

because the lower voltages commonly found in LT1507

applications may require lower ripple voltage across the

capacitor to ensure adequate boost voltage under worst-

case conditions. Efficiency is not affected by the capacitor

value, but the capacitor should have an ESR of less than

2Ωtoensurethatitcanberechargedfullyundertheworst-

casecondition of minimuminputvoltage. Almost any type

of film or ceramic capacitor will work fine.

LAYOUT CONSIDERATIONS

Suggested layout for the LT1507 is shown in Figure 3. The

main concern for layout is to minimize the length of the

INPUT

C

F

MINIMIZE AREA OF

D2

C2

CONNECTIONS TO THE

SWITCH NODE AND

BOOST NODE, BUT OBSERVE

CURRENT DENSITY LIMITATIONS

C

AND R ARE OPTIONAL.

C

F

V

BOOST

IN

C

SEE FREQUENCY

COMPENSATION

IN PATH TO L1

C3

D1

R

C

FB

GND

KEEP INPUT CAPACITOR

AND CATCH DIODE CLOSE

TO REGULATOR AND

TERMINATE THEM

SW

R2

SHDN

SYNC

TO SAME POINT

R1

L1

SYNC

C1

OUTPUT

TERMINATE GND PIN

DIRECTLY TO GROUND

PLANE WITH VIA TO

MINIMIZE EMI. (MINIMIZE

DISTANCE TO INPUT

CAPACITOR C3). CONNECT

FEEDBACK RESISTORS AND

COMPENSATION

GROUND RING NEED

NOT BE AS SHOWN.

(NORMALLY EXISTS AS

INTERNAL PLANE)

COMPONENTS DIRECTLY

TO GROUND PLANE OR TO

SWITCHER GND PIN.

CONNECT OUTPUT CAPACITOR

DIRECTLY TO HEAVY GROUND

TAKE OUTPUT DIRECTLY FROM END OF OUTPUT

CAPACITOR TO AVOID PARASITIC RESISTANCE

AND INDUCTANCE (KELVIN CONNECTION)

LT1507 • F03

Figure 3. Suggested Layout

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The term inside the radical has a maximum value of 0.5

when input voltage is twice output and stays near 0.5 for

a relatively wide range of input voltages. It is common

practice, therefore, tosimplyusetheworst-casevalueand

assumethatRMSripplecurrentisonehalfofloadcurrent.

At maximum output current of 1.5A for the LT1507, the

input bypass capacitor should be rated at 0.75A ripple

current. Note however, that there are many secondary

considerations in choosing the final ripple current rating.

These include ambient temperature, average versus peak

load current, equipment operating schedule and required

productlifetime.FormoredetailsseeApplicationNotes19

and 46.

high speed circulating current path shown in Figure 4 and

to make connections to the output capacitor in a manner

that minimizes output ripple and noise. For more details,

see Applications Information section in the LT1376 data

sheet.

SWITCH NODE

L1

5V

HIGH

FREQUENCY

CIRCULATING

PATH

C

3

C

1

V

IN

LOAD

LT1507 • F04

Figure 4. High Speed Switching Path

Input Capacitor Type

Some caution must be used when selecting the type of

capacitor used at the input of regulators. Aluminum

electrolytics are lowest cost, but are physically large to

achieve adequate ripple current rating, and size con-

straints (especially height) may preclude their use.

Ceramic capacitors are now available in larger values and

their high ripple current and voltage rating make them

ideal for input bypassing. Cost is slightly higher and

footprint may also be somewhat larger. Solid tantalum

capacitors are a good choice except that they have a

history of occasional spectacular failures when they are

subjected to very large current surges during power-up.

The capacitors can short and then burn with a brilliant

white light and lots of nasty smoke. This phenomenon

occurs in only a small percentage of units, but it has led

some OEM companies to forbid their use in high surge

applications. The input bypass capacitor of regulators can

see such high surges when a battery or high capacitance

source is connected.

INPUT BYPASSING AND VOLTAGE RANGE

Input Bypass Capacitor

Stepdown converters draw current from the input supply

in pulses. The average height of these pulses is equal to

load current and the duty cycle is equal to V_OUT/V_IN. Rise

and fall time of the current is very fast. A local bypass

capacitor across the input supply is necessary to ensure

proper operation of the regulator and minimize the ripple

current fed back into the input supply. The capacitor also

forces switching current to flow in a tight local loop,

minimizing EMI.

Do not cheat on the ripple current rating of the input

bypass capacitor, but also don’t get hung up on the value

in microfarads. The input capacitor is intended to absorb

all the switching current ripple, which can have an RMS

value as high as one half of load current. Ripple current

ratings on the capacitor must be observed to ensure

reliable operation. The actual value of the capacitor in

microfarads is not particularly important because at

500kHz, any value above 5µF is essential resistive. Ripple

current rating is the critical parameter. RMS ripple current

can be calculated from:

Several manufacturers have developed a line of solid

tantalum capacitors specially tested for surge capability

(AVX TPS series for instance, see Table 2). Even these

units may fail if the input current surge exceeds a value

equal to the voltage rating of the capacitor divided by 1Ω

(10A for a 10V capacitor). For this reason, AVX recom-

mends using the highest voltage rating possible for the

input capacitor. For equal case size, this means that lower

values of capacitance must be used. As stated above, this

V

(V – V

)

OUT IN

OUT

I

(RMS) = I

OUT

RIPPLE

2

V

IN

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is not a problem, but it should be noted that for equal case

size, the ripple current rating and ESR of higher voltage

capacitors will be somewhat worse. The lower input

operating voltages of the LT1507 allow considerable

derating of capacitor voltage. If solid tantalum units are

used, it would be wise to use units rated at 25V or more,

as long as ripple current requirements are met. Design

Note 122 discusses the problem of showing typical input

capacitorsurgesthatoccurwhenbatteriesoradaptersare

hot plugged to typical regulator systems.

to 1.5V higher than the standard running voltage, espe-

cially at light loads. An approximate formula to calculate

minimum running voltage at load currents above 100mA

is:

V

+ (I

0.85

)(0.3Ω)

OUT

V

=

(I

≥ 100mA)

IN(MIN)

OUT

With V_OUT= 3.3V and I_OUT= 0.1A, this formula yields

V_IN(MIN)= 3.9V. Increasing load current to 1A raises

minimum input to 4.2V. For start-up and operation at light

loads, see the next section.

A new capacitor type known as OS-CON uses a “semicon-

ductor” dielectric to achieve extremely low ESR and high

ripple current rating. These are ideal for input bypassing

because they are not surge sensitive. They are not sug-

gested for output capacitors because the very low ESR

maypresentloopstabilityproblems.Priceandsize(height)

are issues to be considered. The original manufacturer is

Sanyo but there are now additional sources.

Minimum Start-Up Voltage and Operation

at Light Loads

The boost capacitor supplies current to the BOOST pin

during switch ON time. This capacitor is recharged only

during switch OFF time. Under certain conditions of light

load and low input voltage, the capacitor may not be fully

rechargedduringtherelativelyshortOFFtime.Thiscauses

the boost voltage to collapse and minimum input voltage

is increased. Start-up voltage at light loads is higher than

normal running voltage for the same reasons. Figure 5

shows minimum input voltage for a 3.3V output, both for

start-up and for normal operation. This graph indicates

that a 5V to 3.3V converter with 4.7V minimum input

voltage, will not start correctly below a 40mA load current

and will not run correctly below a 4mA load current. If

minimumloadcurrentislessthan50mA, apreloadshould

be added or the circuit in Figure 6 can be used.

Larger capacitors may be necessary when the input volt-

age is very close to the minimum specified on the data

sheet. A 5µF ceramic input capacitor for instance, moves

at about 0.1V/µs during switch ON time when load current

is 1A, creating a ripple voltage due to reactance. This is in

addition to the ripple caused by capacitor ESR. Physically

larger input capacitors will have more capacitance (less

reactance) and lower ESR. Small voltage dips during

switchONtimearenotnormallyaproblem, butatverylow

inputvoltagetheymaycauseerraticoperationbecausethe

input voltage drops below the minimum specification.

Problems can also occur if the input to output voltage

differential is near minimum.

6.5

VALID ONLY FOR V

= 3.3V

OUT

6.0

5.5

5.0

4.5

4.0

3.5

3.0

MINIMUM VOLTAGE

TO START WITH

PNP ADDED

Minimum Input Voltage (After Start-Up)

MINIMUM VOLTAGE

TO START WITH

STANDARD CIRCUITS

Minimum input voltage to make the LT1507 “run” cor-

rectly is typically 3.6V, but to regulate the output, a buck

converter input voltage must always be higher than the

output voltage. To calculate minimum operating input

voltage, switch voltage loss and maximum duty cycle

must be taken into account. With the LT1507 there is the

additional consideration of proper operation of the boost

circuit. The boost circuit allows the power switch to

saturate for high efficiency, but it also sometimes results

in a start-up or low current operating voltage that is 0.5V

MINIMUM VOLTAGE

TO RUN WITH

MINIMUM VOLTAGE

TO RUN WITH

STANDARD CIRCUIT

PNP ADDED

1

10

100

1000

LOAD CURRENT (mA)

LT1400 • GXX

MINIMUM VOLTAGE

TO RUN WITH

Figure 5. Minimum Input Voltage for V_OUT= 3.3V

PNP ADDED

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The circuit in Figure 6 will allow operation at light loads

with low input voltages. It uses a small PNP to charge the

boost capacitor (C2) and an extra diode (D3) to complete

the power path from V_SWto the boost capacitor. Note that

the diodes have been changed to Schottky BAT85s to

optimize low voltage operation. Figure 5 shows that with

the added PNP, minimum load current can be reduced to

6mA and still guarantee proper start-up with 4.7V input.

problems. For low input voltage, high sync frequency

applications, the circuit shown in Figure 7 can be used to

generate an external slope compensation ramp that elimi-

nates subharmonic oscillation. See Frequency Compen-

sation section for a discussion of an entirely different

cause of subharmonic switching before assuming that the

causeisinsufficientslopecompensation.ApplicationNote

19 has more details on the theory of slope compensation.

D2

BAT85

V

SW

OUT

+

C2

0.22µF

LT1507

GND

D3

BAT85

SYNC

V

C

L1

C

BOOST

S

R

C

1000pF

V

= 3.3V

INPUT

V

SW

470Ω

OUT

IN

Q1

2N3906

LT1507-3.3

+

C

R

S

5.2k

SENSE

V

C

2000pF

GND

LT1507 • F07

+

D1

1N5818

C1

C

Figure 7. Adding External Slope Compensation for High

Sync Frequencies

LT1507 • F06

External Slope Compensation Ramp

Figure 6. Adding a Small PNP to Reduce Minimum

Start-Up Voltage

The LT1507 is a current mode switching regulator and

therefore, it requires something called “slope compensa-

tion” when operated above 50% duty cycle in continuous

mode. This condition occurs when input voltage is less

than twice output voltage. Slope compensation adds a

ramp to the switch current sense signal generated on the

chip during switch ON time. Typically the ramp is gener-

ated from a portion of the internal oscillator waveform. In

the LT1507, the ramp is arranged to be zero until the

oscillator waveform reaches about 40% of its final value.

This minimizes the total amount of ramp added to switch

current. The reason for doing it this way is that the ramp

subtracts from switch current limit, so that switch current

limit would be considerably lower at high duty cycle

compared to low duty cycle if the ramp existed at all duty

cycles. By starting the ramp at the 40% point, changes in

current limit are minimized. No ramp is needed when

operating below 50% duty cycle.

SYNCHRONIZING

The LT1507 SYNC pin is used to synchronize the internal

oscillator to an external signal. It is directly logic compat-

ible and can be driven with any signal between 10% and

90% duty cycle. The synchronizing range is equal toinitial

operating frequency up to 1MHz (above 700kHz external

slope compensation may be needed). This means that

minimum practical sync frequency is equal to the worst-

case high self-oscillating frequency (560kHz) not the

typical operating frequency of 500kHz. Caution should be

usedwhensynchronizingabove700kHzbecauseathigher

sync frequencies, the amplitude of the internal slope

compensation used to prevent subharmonic switching is

reduced. This type of subharmonic switching only occurs

at input voltages less than twice the output voltage and

shows up as alternating pulse widths at the switch node.

It does not cause the regulator to lose regulation, but

switch frequency content down to 100kHz may be objec-

tionable. Higher inductor values will tend to eliminate

Problems can occur with this technique if the regulator is

used with a combination of high external sync frequency

and more than 50% duty cycle. The basic sync function

16

LT1507

U

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APPLICATIONS INFORMATION

works by prematurely tripping the oscillator before it

reaches its normal peak value. For instance, if the oscilla-

tor is synchronized at twice its nominal frequency, oscil-

lator amplitude will drop by half. A ramp which previously

started at the 40% point now starts at the 80% point! This

effectively blocks slope compensation and the regulator

may respond with fluctuating pulse widths, a “phase

oscillation” if you will. The regulator output stays in

regulation but subharmonic frequencies are generated at

the switch node.

ForV_IN=4.7, V_OUT=3.3V, f=1MHz, L=5µHandDC_S=25%:

(6.6 − 4.7)(1− 0.25)

V

≥

= 71mV

P-P

6

−6

2 1 10

5 10

1.8

ToavoidsmallvaluesofR_S, thecompensationcapacitor(C_C)

should be made as small as possible. 2000pF will work in

most situations. If we increase V_PPto 90mV for a little

cushion, R_Swill be:

The solution to this problem is to generate an external

ramp that replaces the missing internal ramp. As it turns

out, this is not difficult if the sync signal can be arranged

tohaveafairlylowdutycycle(<35%). Therampiscreated

by AC coupling a resistor from the sync signal to the

compensation capacitor as shown in Figure 7. This gener-

ates a negative ramp on the V_Cpin during switch ON time

that emulates the missing internally generated ramp.

Amplitude of the ramp should be about 100mV to 200mV

peak-to-peak. The formulas for calculating the values of

R_Sand C_Sare shown below. Note that the C_Svalue is

unimportant as long as it exceeds the value given. The

formula assures that the impedance of C_Swill be small

compared to R_S.

(5)(0.25)(0.75)

R =

= 5.2k

S

−9

6

0.09 2 10

1 10

(

)

(

)

20

C ≥

= 612pF

6

2π 1 10

5200

(

)

(

)

THERMAL CALCULATIONS

Power dissipation in the LT1507 chip comes from four

sources: switch DC loss, switch AC loss, boost circuit

current and input quiescent current. The formulas below

show how to calculate each of these losses. These formu-

las assume continuous mode operation, so they should

notbeusedforcalculatingefficiencyatlightloadcurrents.

V

(DC )(1− DC )

S S

SYNC

R =

S

Switch loss:

V

(C )(f)

C

P-P

2

20

2π(f)(R )

R

(I

) (V

V

IN

)

SW OUT

OUT

C >

S

P

=

+16ns(I )(V )(f)

OUT IN

SW

S

V_SYNC= Peak-to-peak value of sync signal

DC_S= Duty cycle of incoming sync signal

V_P-P= Desired amplitude of ramp

f = Sync frequency

Boost current loss:

2

V

I

OUT

75

OUT

P

=

0.008 +

BOOST

V

IN

Theoretical minimum amplitude for the ramp, assuming

no internal ramp, is:

Quiescent current loss:

P = V (0.003)+ V (0.005)

Q

IN

OUT

(2V

− V )(1−DC )

IN S

OUT

V

≥

P-P

R_SW= Switch resistance (≈ 0.4Ω)

16ns = Equivalent switch current/voltage overlap time

f = Switching frequency

2(f)(L)(g

)

mP

g_mP= Transconductance from V_Cpin to switch current

(1.8A/V for the LT1507).

17

LT1507

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APPLICATIONS INFORMATION

2.0

1.5

1.0

0.5

0

80

Example: with V_IN= 5V, V_OUT= 3.3V, I_OUT= 1A;

GAIN (A/V)

2

40

(0.4)(1) (3.3)

5

−9

3

P

=

+ 16 10

1 5 500 10

( )( )

SW

(

)

(

)

0

= 0.26 + 0.04 = 0.3W

PHASE

2

1

(3.3)

P

=

0.008

0.046W

+

=

BOOST

V

= 3.3V

–40

–80

OUT

IN

75

5

I

= 250mA

V

= 5V

L = 10µH

P = 5(0.003)+ 3.3(0.005) = 0.032W

Q

10

100

1k

10k

100k

FREQUENCY (Hz)

Total power dissipation is 0.3 + 0.046 + 0.032 = 0.38W.

LT1507 • F08

Thermal resistance for the LT1507 packages is influenced

by the presence of internal or backside planes. With a full

plane under the SO package, thermal resistance will be

about120°C/W. Noplanewillincreaseresistancetoabout

150°C/W. To calculate die temperature, use the proper

thermal resistance number for the desired package and

add in worst-case ambient temperature;

Figure 8. Phase and Gain from V_CPin Voltage

to Inductor Current

parallel with 12pF. In all practical applications, the com-

pensation network from V_Cpin to ground has a much

lower impedance than the output impedance of the ampli-

fier at frequencies above 500Hz. This means that the error

amplifier characteristics themselves do not contribute

excess phase shift to the loop and the phase/gain charac-

teristics of the error amplifier section are completely

controlled by the external compensation network.

T_J= T_A+ θ_JA(P_TOT

)

With the S8 package (θ_JA= 120°C/W) at an ambient

temperature of 70°C;

T_J= 70 + 120(0.38) = 116°C

The complete small-signal model is shown in Figure 9. R1

andR2arethedividerusedtosetoutputvoltage.Theseare

internal on the fixed voltage LT1507-3.3 with R1 = 1.8k

and R2 = 5k. R_C, C_Cand C_Fare external compensation

FREQUENCY COMPENSATION

The LT1507 uses a “current mode” architecture to help

alleviate phase shift created by the inductor. The basic

connectionsareshowninFigure9.Gainofthepowerstage

can be modeled as 1.8A/V transconductance from the V_C

pin voltage to current delivered to the output. This is

shown in Figure 8 where the transconductance from V_C

pin to inductor current is essentially flat from 50Hz to

50kHz and phase shift is minimal in the important loop

unity-gain band of 1kHz to 50kHz. Inductor variation from

3µH to 20µH will have very little effect on these curves.

V

SW

L1

LT1507

POWER STAGE

= 1.8A/V

OUTPUT

g

m

ERROR AMPLIFIER

= 2000µho

R1

R2

12pF

200k

g

m

F

B

–

ESR

+

2.42V

C1

GND

V

C

Overall gain from the V_Cpin to output is then modeled as

the product of 1.8A/V transconductance multiplied by the

complex impedance of the load in parallel with the output

capacitor model.

R

C

F

C

1507 • F09

The error amplifier can be modeled as a transconductance

of 2000µmho, with an output impedance of 200kΩ in

Figure 9. Small-Signal Model for Loop Stability Analysis

18

LT1507

U

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APPLICATIONS INFORMATION

move around, but at the same time phase moves with it so

that adequate phase margin is maintained over a very wide

range of ESR (≥ 5:1)

components. In many cases only C_Cis needed. Adding R_C

willimprovephasemargin,butthismaynecessitatetheneed

for C_Fto limit switching frequency ripple at the V_Cpin.

80

200

150

100

50

In Figure 10, full loop phase/gain characteristics are shown

with a compensation capacitor (C_C) of 0.0033µF, giving the

error amplifier a pole at 240Hz, with phase rolling off to 90°

and staying there. The overall loop has a gain of 77dB at low

frequency rolling off to unity gain at 20kHz. Phase shows a

2-pole characteristic until the ESR of the output capacitor

brings it back above 10kHz. Phase margin is about 60° at

unity-gain.

V

OUT

= 10V

IN

V

OUT

= 5V, I

= 500mA

OUT

C

= 100µF, 10V, AVX TPS

60

C

= 3.3nF, R = 0

C

L = 10µH

GAIN

40

PHASE

20

0

Analog experts will note that around 1kHz, phase dips to

within 20° of the zero phase margin line. This is typical of

switching regulators because of the 2-pole rolloff generated

by the output capacitor and the compensation network. This

region of low phase is not a problem as long as it does not

occur near unity-gain. In practice, the variability of output

capacitorESRtendstodominateallothereffectswithrespect

to loop response. Variations in ESR will cause unity-gain to

–20

–50

0.01

0.1

1

10

100

FREQUENCY (kHz)

LT1511 • F10

Figure 10. Overall Loop Phase and Gain

Undervoltage Lockout

See Application Information in LT1376 data sheet.

U

PACKAGE DESCRIPTION ^{Dimensions in inches (millimeters) unless otherwise noted.}

N8 Package

8-Lead PDIP (Narrow 0.300)

(LTC DWG # 05-08-1510)

0.400*

(10.160)

MAX

0.130 ± 0.005

(3.302 ± 0.127)

0.300 – 0.325

(7.620 – 8.255)

0.045 – 0.065

(1.143 – 1.651)

8

1

7

6

5

0.065

(1.651)

TYP

0.009 – 0.015

(0.229 – 0.381)

0.255 ± 0.015*

(6.477 ± 0.381)

0.125

(3.175)

MIN

0.005

0.015

(0.380)

MIN

(0.127)

MIN

+0.025

–0.015

0.325

2

4

+0.635

8.255

3

(

)

–0.381

0.100 ± 0.010

(2.540 ± 0.254)

0.018 ± 0.003

(0.457 ± 0.076)

N8 0695

*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.

MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm)

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-

tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.

19

LT1507

U

PACKAGE DESCRIPTION

Dimensions in inches (millimeters) unless otherwise noted.

S8 Package

8-Lead Plastic Small Outline (Narrow 0.150)

(LTC DWG # 05-08-1610)

0.189 – 0.197*

(4.801 – 5.004)

7

5

8

6

0.150 – 0.157**

(3.810 – 3.988)

0.228 – 0.244

(5.791 – 6.197)

1

3

4

2

0.010 – 0.020

(0.254 – 0.508)

× 45°

0.053 – 0.069

(1.346 – 1.752)

0.004 – 0.010

(0.101 – 0.254)

0.008 – 0.010

(0.203 – 0.254)

0°– 8° TYP

0.016 – 0.050

0.406 – 1.270

0.050

(1.270)

BSC

0.014 – 0.019

(0.355 – 0.483)

*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH

SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE

SO8 0695

**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD

FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE

RELATED PARTS

PART NUMBER

DESCRIPTION

COMMENTS

LT1371

3A 500kHz Step-Up Switching Regulator

1.5A 500kHz Step-Up Switching Regulator

1.5A 500kHz Step-Down Switching Regulator

1.5A 1MHz Step-Up Switching Regulator

High Current DC/DC Conversion Uses Small Power Components

Includes Positive and Negative Output Voltage Regulation

Includes Synchronization Capability

LT1372

LT1375

LT1376

Output Biasing Yields 90% Efficiency

LT1377

Highest Frequency Monolithic Switching Regulator

1507f LT/TP 0697 4K • PRINTED IN USA

LINEAR TECHNOLOGY CORPORATION 1996

Linear Technology Corporation

●

1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408)432-1900

20

●

FAX: (408) 434-0507 TELEX: 499-3977 www.linear-tech.com


型号：	LT1507CS8-3.3#TR
厂家：	Linear
描述：	LT1507 - 500kHz Monolithic Buck Mode Switching Regulator; Package: SO; Pins: 8; Temperature Range: 0°C to 70°C 开关光电二极管
文件：	总20页 (文件大小：287K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

LT1507CS8-3.3#TR [Linear]

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