LT1576IS8_技术文档

LT1576/LT1576-5

1.5A, 200kHz Step-Down

Switching Regulator

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FEATURES

DESCRIPTIO

TheLT^®1576isa200kHzmonolithicbuckmodeswitching

regulator. A 1.5A switch is included on the die along with

allthenecessaryoscillator, controlandlogiccircuitry. The

topology is current mode for fast transient response and

goodloopstability.TheLT1576isamodifiedversionofthe

industry standard LT1376 optimized for noise sensitive

applications.

■

Constant 200kHz Switching Frequency

■

1.21V Reference Voltage

■

Fixed 5V Output Option

Easily Synchronizable

Uses All Surface Mount Components

Inductor Size Reduced to 15µH

■

Saturating Switch Design: 0.2Ω

Effective Supply Current: 1.16mA

Shutdown Current: 20µA

Cycle-by-Cycle Current Limiting

Fused Lead SO-8 Package

In addition, the reference voltage has been lowered to

allow the device to produce output voltages down to 1.2V.

Quiescent current has been reduced by a factor of two.

Switchonresistancehasbeenreducedby30%.Switchtran-

sition times have been slowed to reduce EMI generation.

The oscillator frequency has been reduced to 200kHz to

maintain high efficiency over a wide output current range.

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APPLICATIO S

■

Portable Computers

The pinout has been changed to improve PC layout by

allowing the high current high frequency switching cir-

cuitrytobeeasilyisolatedfromlowcurrentnoisesensitive

control circuitry. The new SO-8 package includes a fused

ground lead which significantly reduces the thermal resis-

tance of the device to extend the ambient operating tem-

perature range. There is an optional function of shutdown

orsynchronization.Standardsurfacemountexternalparts

can be used including the inductor and capacitors.

, LTC and LT are registered trademarks of Linear Technology Corporation.

■

Battery-Powered Systems

■

Battery Charger

Distributed Power

■

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TYPICAL APPLICATION

Efficiency vs Load Current

100

5V Buck Converter

V

= 5V

OUT

IN

INPUT

= 10V

95

90

85

80

75

70

6V TO 25V

C2

+

C3*

L = 33µH

D2

0.33µF

10µF TO

50µF

1N914

L1**

15µH

V

BOOST

IN

OUTPUT**

5V, 1.25A

V

SW

LT1576 BIAS

SHDN

GND

FB

OPEN = ON

V

C

R1

15.8k

D1

1N5818

C1

* RIPPLE CURRENT RATING ≥ I /2

OUT

+

R2

4.99k

C

100µF, 10V

SOLID

** INCREASE L1 TO 30µH FOR LOAD

CURRENTS ABOVE 0.6A AND TO

60µH ABOVE 1A

100pF

TANTALUM

SEE APPLICATIONS INFORMATION

0

0.50 0.75 1.00

LOAD CURRENT (A)

1.25 1.50

0.25

1576 TA01

1576 TA02

1

LT1576/LT1576-5

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ABSOLUTE MAXIMUM RATINGS

(Note 1)

PACKAGE/ORDER INFORMATION

ORDER PART NUMBER

Input Voltage .......................................................... 25V

BOOST Pin Above Input Voltage ............................. 10V

SHDN Pin Voltage..................................................... 7V

BIAS Pin Voltage ...................................................... 7V

FB Pin Voltage (Adjustable Part)............................ 3.5V

FB Pin Current (Adjustable Part)............................ 1mA

SYNC Pin Voltage ..................................................... 7V

Operating Junction Temperature Range

LT1576C............................................... 0°C to 125° C

LT1576I ........................................... –40°C to 125°C

Storage Temperature Range ................ –65°C to 150°C

Lead Temperature (Soldering, 10 sec)................. 300°C

LT1576CS8

LT1576CS8-SYNC

TOP VIEW

LT1576IS8

SHDN OR

1

2

3

4

8

7

6

5

V

SW

LT1576IS8-SYNC

LT1576CS8-5

SYNC*

FB OR SENSE*

V

IN

V

C

BOOST

GND

LT1576CS8-5 SYNC

LT1576IS8-5

LT1576IS8-5 SYNC

BIAS

S8 PACKAGE

8-LEAD PLASTIC SO

θ_JA= 80°C/ W WITH FUSED GROUND PIN

CONNECTED TO GROUND PLANE OR

LARGE LANDS

S8 PART MARKING

1576

1576SN 5765SN

1576I 1576I5

576ISN 76I5SN

15765

*Default is the adjustable output voltage device with FB pin and shutdown

function. Option -5 replaces FB with SENSE pin for fixed 5V output

applications. -SYNC replaces SHDN with SYNC pin for applications

requiring synchronization. Consult factory for Military grade parts.

The ● denotes specifications which apply over the full operating temperature

ELECTRICAL CHARACTERISTICS

range, otherwise specifications are T_A, T_J= 25°C, V_IN= 15V, V_C= 1.5V, Boost = V_IN+ 5V, switch open, unless otherwise noted.

PARAMETER

CONDITIONS

All Conditions

MIN

TYP

MAX

UNITS

Feedback Voltage

1.195 1.21

1.18

1.225

1.24

V

●

Sense Voltage (Fixed 5V)

4.94

4.90

5.0

5.06

5.10

V

SENSE Pin Resistance

13

18.5

0.01

0.5

26

0.03

2

kΩ

%/V

µA

Reference Voltage Line Regulation

Feedback Input Bias Current

Error Amplifier Voltage Gain

Error Amplifier Transconductance

5V ≤ V ≤ 25V

IN

●

(Notes 2, 8)

200

400

∆I (V ) = ±10µA (Note 8)

800

400

1050

1300

1700

µMho

C

V Pin to Switch Current Transconductance

1.5

110

130

0.8

2.1

2

A/V

µA

V

C

Error Amplifier Source Current

Error Amplifier Sink Current

V

= 1.1V

= 1.4V

●

40

50

190

200

FB

V Pin Switching Threshold

C

Duty Cycle = 0

V Pin High Clamp

C

V

Switch Current Limit

V Open, V = 1.1V, DC ≤ 50%

C

●

1.5

3.50

A

FB

Slope Compensation (Note 9)

Switch On Resistance (Note 7)

DC = 80%

0.3

0.2

A

I

= 1.5A

0.35

0.45

Ω

SW

●

Maximum Switch Duty Cycle

V

FB

= 1.1V

90

86

94

%

2

LT1576/LT1576-5

The ● denotes specifications which apply over the full operating temperature

ELECTRICAL CHARACTERISTICS

range, otherwise specifications are T_A, T_J= 25°C, V_IN= 15V, V_C= 1.5V, Boost = V_IN+ 5V, switch open, unless otherwise noted.

PARAMETER

CONDITIONS

V Set to Give 50% Duty Cycle

MIN

TYP

8

MAX

UNITS

Minimum Switch Duty Cycle (Note 10)

Switch Frequency

%

180

160

200

220

240

kHz

C

●

Switch Frequency Line Regulation

Frequency Shifting Threshold on FB Pin

Minimum Input Voltage (Note 3)

Minimum Boost Voltage (Note 4)

Boost Current (Note 5)

5V ≤ V ≤ 25V

0

0.15

1.0

5.5

3.0

%/V

V

IN

∆f = 10kHz

0.4

0.74

5.0

2.3

V

I

≤ 1.5A

V

SW

I

= 0.5A

= 1.5A

●

9

27

18

50

mA

SW

V

Supply Current (Note 6)

V

= 5V

●

0.55

1.6

20

0.8

2.2

mA

IN

BIAS

SHDN

BIAS Supply Current (Note 6)

Shutdown Supply Current

= 0V, V ≤ 25V, V = 0V, V Open

50

75

µA

IN

SW

C

●

Lockout Threshold

V Open

C

2.34

2.42

2.50

V

Shutdown Thresholds

V Open Device Shutting Down

Device Starting Up

●

0.13

0.25

0.37

0.45

0.60

0.7

V

C

Synchronization Threshold

Synchronizing Range

1.5

2.2

V

kHz

kΩ

250

400

SYNC Pin Input Resistance

40

Note 1: Absolute Maximum Ratings are those values beyond which the life

Note 7: Switch on resistance is calculated by dividing V to V voltage

IN SW

by the forced current (1.5A). See Typical Performance Characteristics for

the graph of switch voltage at other currents.

of a device may be impaired.

Note 2: Gain is measured with a V swing equal to 200mV above the

C

switching threshold level to 200mV below the upper clamp level.

Note 8: Transconductance and voltage gain refer to the internal amplifier

exclusive of the voltage divider. To calculate gain and transconductance,

refer to the SENSE pin on the fixed voltage parts. Divide values shown by

Note 3: Minimum input voltage is not measured directly, but is guaranteed

by other tests. It is defined as the voltage where internal bias lines are still

regulated so that the reference voltage and oscillator frequency remain

constant. Actual minimum input voltage to maintain a regulated output will

depend on output voltage and load current. See Applications Information.

Note 4: This is the minimum voltage across the boost capacitor needed to

guarantee full saturation of the internal power switch.

the ratio V /1.21.

OUT

Note 9: Slope compensation is the current subtracted from the switch

current limit at 80% duty cycle. See Maximum Output Load Current in the

Applications Information section for further details.

Note 10: Minimum on-time is 400ns typical. For a 200kHz operating

frequency this means the minimum duty cycle is 8%. In frequency

foldback mode, the effective duty cycle will be less than 8%.

Note 5: Boost current is the current flowing into the boost pin with the pin

held 5V above input voltage. It flows only during switch on time.

Note 6: V supply current is the current drawn when the BIAS pin is held

IN

at 5V and switching is disabled. Total input referred supply current is

calculated by summing input supply current (I ) with a fraction of BIAS

SI

supply current (I

)

SB

I

= I + (I )(V

/V )(1.15)

BIAS IN

TOT

SI

SB

with V = 15V, V

= 5V, I = 0.55mA, I = 1.6mA and I

= 1.16mA.

IN

BIAS

SI

SB

TOT

If the BIAS pin is unavailable or open circuit, the sum of V and BIAS

IN

supply currents will be drawn by the V pin.

IN

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LT1576/LT1576-5

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TYPICAL PERFORMANCE CHARACTERISTICS

Switch Drop

Switch Peak Current Limit

Feedback Pin Voltage

0.5

0.4

0.3

0.2

0.1

0

1.23

1.22

1.21

1.20

1.19

2.5

2.0

1.5

1.0

0.5

0

TYPICAL

125°C

25°C

MINIMUM

–20°C

0

20

40

60

80

100

0

0.50 0.75 1.00

1.25 1.50

0.25

–25

0

25

50

75

125

–50

100

SWITCH CURRENT (A)

DUTY CYCLE (%)

JUNCTION TEMPERATURE (°C)

1576 G02

1576 G01

1576 G03

Shutdown Pin Bias Current

Shutdown Thresholds

4

3

2

1

0

180

160

140

120

100

80

0.8

0.7

0.6

0.5

0.4

0.3

0.2

0.1

0

AT 2.44V STANDBY THRESHOLD

(CURRENT FLOWS OUT OF PIN)

START-UP

SHUTDOWN

60

CURRENT REQUIRED TO FORCE

40

SHUTDOWN (FLOWS OUT OF PIN).

AFTER SHUTDOWN, CURRENT

DROPS TO A FEW µA

20

0

–25

0

25

50

75

125

–25

0

25

50

75

125

–25

0

25

50

75

125

–50

100

–50

100

–50

100

JUNCTION TEMPERATURE (°C)

1576 G04

1576 G05

1576 G06

Shutdown Supply Current

Standby Thresholds

Error Amplifier Transconductance

2000

1500

1000

500

200

150

100

50

25

20

15

10

5

2.46

2.45

2.44

2.43

2.42

2.41

2.40

V

= 0V

SHDN

PHASE

GAIN

ON

V

C

STANDBY

C

OUT

2.4pF

R

OUT

570k

V

1 × 10^–3

(

)

FB

0

ERROR AMPLIFIER EQUIVALENT CIRCUIT

= 50Ω

0

R

LOAD

–500

0

–50

0

5

10

15

20

25

10

100

1k

10k

100k

1M

50

100 125

–50 –25

0

25

75

INPUT VOLTAGE (V)

FREQUENCY (Hz)

JUNCTION TEMPERATURE (°C)

1576 G09

1576 G08

1576 G07

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LT1576/LT1576-5

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TYPICAL PERFORMANCE CHARACTERISTICS

Shutdown Supply Current

Error Amplifier Transconductance

Frequency Foldback

100

75

50

25

0

1600

1400

1200

1000

800

600

400

200

0

250

200

150

100

50

SWITCHING FREQUENCY

V

IN

= 25V

V

IN

= 10V

FEEDBACK PIN CURRENT

0

0.2

0.3

0

0.4

–25

0

25

50

75

100

125

0.1

–50

0

0.5

1.0

FEEDBACK VOLTAGE (V)

1.5

2.0

SHUTDOWN VOLTAGE (V)

JUNCTION TEMPERATURE (°C)

1576 G10

1576 G11

1576 G12

Minimum Input Voltage

at V_OUT= 5V

Maximum Load Current

at V_OUT= 10V

Switching Frequency

1.50

1.25

1.00

0.75

0.50

0.25

0

7

6

5

240

220

200

180

160

V

= 5V

OUT

V

OUT

= 10V

L = 60µH

L = 30µH

MINIMUM

STARTING VOLTAGE

L = 15µH

MINIMUM

RUNNING VOLTAGE

–25

0

25

50

75

125

0

5

10

15

20

25

–50

100

1

10

100

1000

INPUT VOLTAGE (V)

LOAD CURRENT (mA)

JUNCTION TEMPERATURE (°C)

1576 G13

1576 G15

1576 G14

Maximum Load Current

at V_OUT= 5V

Maximum Load Current

at V_OUT= 3.3V

Inductor Core Loss

1.50

1.25

1.00

0.75

0.50

0.25

0

1.50

1.25

1.00

0.75

0.50

0.25

0

1.0

0.1

20

12

8

V

= 5V, V = 10V, I

= 1A

OUT

IN

OUT

L = 60µH

L = 30µH

L = 15µH

L = 30µH

L = 15µH

4

2

1.2

0.8

TYPE 52

POWDERED IRON

Kool Mµ^®

0.4

PERMALLOY

µ = 125

0.2

0.01

0.001

CORE LOSS IS

INDEPENDENT OF LOAD

CURRENT UNTIL LOAD CURRENT FALLS

LOW ENOUGH FOR CIRCUIT TO GO INTO

DISCONTINUOUS MODE

0.12

0.08

0.04

0.02

V

OUT

= 5V

5

V

OUT

= 3.3V

5

0

10

15

20

25

0

10

15

20

25

0

5

10

15

20

25

INPUT VOLTAGE (V)

INDUCTANCE (µH)

1576 G16

1576 G17

1576 G18

Kool Mµ is a registered trademark of Magnetics, Inc.

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LT1576/LT1576-5

TYPICAL PERFORMANCE CHARACTERISTICS

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BOOST Pin Current

V_CPin Shutdown Threshold

30

25

20

15

10

5

1.0

0.8

0.6

0.4

0.2

0

–25

0

25

50

75

125

–50

100

0

0.50 0.75 1.00

1.25 1.50

0.25

SWITCH CURRENT (A)

JUNCTION TEMPERATURE (°C)

1576 G20

1576 G19

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PIN FUNCTIONS

V_SW(Pin 1): The switch pin is the emitter of the on-chip

power NPN switch. This pin is driven up to the input pin

voltage during switch on time. Inductor current drives the

switch pin negative during switch off time. Negative volt-

age is clamped with the external catch diode. Maximum

negative switch voltage allowed is –0.8V.

GND pin of the IC. This condition will occur when load

current or other currents flow through metal paths be-

tween the GND pin and the load ground point. Keep the

ground path short between the GND pin and the load and

use a ground plane when possible. The second consider-

ation is EMI caused by GND pin current spikes. Internal

capacitance between the V_SWpin and the GND pin creates

very narrow (<10ns) current spikes in the GND pin. If the

GND pin is connected to system ground with a long metal

trace, this trace may radiate excess EMI. Keep the path

between the input bypass and the GND pin short.

V_IN(Pin 2): This is the collector of the on-chip power NPN

switch. This pin powers the internal circuitry and internal

regulator when the BIAS pin is not present. At NPN switch

on and off, high dI/dt edges occur on this pin. Keep the

external bypass and catch diode close to this pin. All trace

inductanceonthispathwillcreateavoltagespikeatswitch

off, adding to the V_CEvoltage across the internal NPN.

BIAS (Pin 5): The BIAS pin is used to improve efficiency

when operating at higher input voltages and light load

current. Connecting this pin to the regulated output volt-

age forces most of the internal circuitry to draw its

operating current from the output voltage rather than the

input supply. This is a much more efficient way of doing

business if the input voltage is much higher than the

output. Minimum output voltage setting for this mode of

operation is 3.3V. Efficiency improvement at V_IN= 20V,

V_OUT= 5V, and I_OUT= 25mA is over 10%.

BOOST (Pin 3): The BOOST pin is used to provide a drive

voltage, higher than the input voltage, to the internal

bipolarNPNpowerswitch. Withoutthisaddedvoltage, the

typical switch voltage loss would be about 1.5V. The

additional boost voltage allows the switch to saturate and

voltage loss approximates that of a 0.2Ω FET structure.

Efficiency improves from 75% for conventional bipolar

designs to > 88% for these new parts.

V_C(Pin 6): The V_Cpin is the output of the error amplifier

and the input of the peak switch current comparator. It is

normally used for frequency compensation, but can do

double duty as a current clamp or control loop override.

GND(Pin4):TheGNDpinconnectionneedsconsideration

for two reasons. First, it acts as the reference for the

regulated output, so load regulation will suffer if the

“ground” end of the load is not at the same voltage as the

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LT1576/LT1576-5

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PIN FUNCTIONS

This pin sits at about 1V for very light loads and 2V at

maximum load. It can be driven to ground to shut off the

regulator, but if driven high, current must be limited to

4mA.

SYNC (Pin 8): The SYNC pin is used to synchronize the

internal oscillator to an external signal. It is directly logic

compatible and can be driven with any signal between

10% and 90% duty cycle. The synchronizing range is

equal to initial operating frequency, up to 400kHz. This pin

replacesSHDNon-SYNCoptionparts. SeeSynchronizing

section in Applications Information for details.

FB/SENSE (Pin 7): The feedback pin is the input to the

error amplifier which is referenced to an internal 1.21V

source. An external resistive divider is used to set the

output voltage. Three additional functions are performed

by the FB pin. The fixed voltage (-5) parts have the divider

resistors included on-chip and the FB pin is used as a

SENSE pin, connected directly to the 5V output. When the

pin voltage drops below 0.7V, the switch current limit and

theswitchingfrequencyarereducedand theexternalsync

function is disabled. See Feedback Pin Function section in

Applications Information for details.

SHDN (Pin 8): The shutdown pin is used to turn off the

regulator and to reduce input drain current to a few

microamperes. Actually, this pin has two separate thresh-

olds, one at 2.44V to disable switching, and a second at

0.4V to force complete micropower shutdown. The 2.44V

threshold functions as an accurate undervoltage lockout

(UVLO). This can be used to prevent the regulator from

operating until the input voltage has reached a predeter-

mined level.

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BLOCK DIAGRAM

The LT1576 is a constant frequency, current mode buck

converter. This means that there is an internal clock and

twofeedbackloopsthatcontrolthedutycycleofthepower

switch. In addition to the normal error amplifier, there is a

current sense amplifier that monitors switch current on a

cycle-by-cycle basis. A switch cycle starts with an oscilla-

tor pulse which sets the R_Sflip-flop to turn the switch on.

When switch current reaches a level set by the inverting

input of the comparator, the flip-flop is reset and the

switch turns off. Output voltage control is obtained by

using the output of the error amplifier to set the switch

current trip point. This technique means that the error

amplifier commands current to be delivered to the output

rather than voltage. A voltage fed system will have low

phase shift up to the resonant frequency of the inductor

and output capacitor, then an abrupt 180° shift will occur.

The current fed system will have 90° phase shift at a much

lower frequency, but will not have the additional 90° shift

until well beyond the LC resonant frequency. This makes

itmucheasiertofrequencycompensatethefeedbackloop

and also gives much quicker transient response.

Most of the circuitry of the LT1576 operates from an

internal 2.9V bias line. The bias regulator normally draws

power from the regulator input pin, but if the BIAS pin is

connected to an external voltage higher than 3V, bias

powerwillbedrawnfromtheexternalsource(typicallythe

regulated output voltage). This will improve efficiency if

the BIAS pin voltage is lower than regulator input voltage.

High switch efficiency is attained by using the BOOST pin

to provide a voltage to the switch driver which is higher

thantheinputvoltage,allowingtheswitchtosaturate.This

boosted voltage is generated with an external capacitor

and diode. Two comparators are connected to the shut-

down pin. One has a 2.44V threshold for undervoltage

lockout and the second has a 0.4V threshold for complete

shutdown.

7

LT1576/LT1576-5

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BLOCK DIAGRAM

0.025Ω

INPUT

+

–

CURRENT

SENSE

2.9V BIAS

REGULATOR

INTERNAL

CC

BIAS

AMPLIFIER

VOLTAGE GAIN = 35

V

SLOPE COMP

BOOST

Σ

0.8V

200kHz

OSCILLATOR

S

R

SYNC

Q1

POWER

SWITCH

R

DRIVER

CIRCUITRY

CURRENT

COMPARATOR

S

FLIP-FLOP

+

–

SHUTDOWN

COMPARATOR

–

+

V

SW

0.4V

FREQUENCY

SHDN

SHIFT CIRCUIT

3.5µA

FOLDBACK

CURRENT

LIMIT

Q2

+

–

CLAMP

–

FB

LOCKOUT

COMPARATOR

+

ERROR

V

C

AMPLIFIER

2.44V

1.21V

g

= 1000µMho

m

GND

1576 BD

Figure 1. Block Diagram

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APPLICATIONS INFORMATION

FEEDBACK PIN FUNCTIONS

The suggested value for the output divider resistor (see

Figure 2) from FB to ground (R2) is 5k or less, and a

formula for R1 is shown below. The output voltage error

caused by ignoring the input bias current on the FB pin is

less than 0.25% with R2 = 5k. A table of standard 1%

values is shown in Table 1 for common output voltages.

Please read the following if divider resistors are increased

above the suggested values.

The feedback (FB) pin on the LT1576 is used to set output

voltage and provide several overload protection features.

The first part of this section deals with selecting resistors

to set output voltage and the remaining part talks about

foldback frequency and current limiting created by the FB

pin. Please read both parts before committing to a final

design. The fixed 5V LT1576-5 has internal divider resis-

tors and the FB pin is renamed SENSE, connected directly

to the output.

R2 V_OUT−1.21

(

)

R1=

1.21

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LT1576/LT1576-5

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APPLICATIONS INFORMATION

sufficiently low duty cycle if switching frequency were

maintained at 200kHz, so frequency is reduced by about

5:1 when the feedback pin voltage drops below 0.7V (see

FrequencyFoldbackgraph). Thisdoesnotaffectoperation

with normal load conditions; one simply sees a gear shift

in switching frequency during start-up as the output

voltage rises.

Table 1

OUTPUT

VOLTAGE

(V)

R1

% ERROR AT OUTPUT

R2

(NEAREST 1%) DUE TO DISCREET 1%

(kΩ

)

(k

Ω

)

RESISTOR STEPS

–0.50

3

3.3

5

4.99

7.32

8.66

15.8

19.6

28.0

36.5

44.2

56.2

+0.30

+0.83

6

–0.62

In addition to lower switching frequency, the LT1576 also

operates at lower switch current limit when the feedback

pin voltage drops below 0.7V. Q2 in Figure 2 performs this

function by clamping the V_Cpin to a voltage less than its

normal 2.1V upper clamp level. This foldback current limit

greatly reduces power dissipation in the IC, diode and

inductorduringshort-circuitconditions.Externalsynchro-

nization is also disabled to prevent interference with

foldback operation. Again, it is nearly transparent to the

userundernormalloadconditions.Theonlyloadsthatmay

be affected are current source loads which maintain full

loadcurrentwithoutputvoltagelessthan50%offinalvalue.

In these rare situations the feedback pin can be clamped

above0.7Vtodefeatfoldbackcurrentlimit.Caution:clamp-

ingthefeedbackpinmeansthatfrequencyshiftingwillalso

be defeated, so a combination of high input voltage and

deadshortedoutputmaycausetheLT1576tolosecontrol

of current limit.

8

–0.01

10

12

15

+0.61

–0.60

– 1.08

More Than Just Voltage Feedback

The feedback pin is used for more than just output voltage

sensing. It also reduces switching frequency and current

limit when output voltage is very low (see the Frequency

Foldback graph in Typical Performance Characteristics).

ThisisdonetocontrolpowerdissipationinboththeICand

the external diode and inductor during short-circuit con-

ditions. A shorted output requires the switching regulator

to operate at very low duty cycles, and the average current

throughthediodeandinductorisequaltotheshort-circuit

current limit of the switch (typically 2A for the LT1576,

folding back to less than 0.77A). Minimum switch on time

limitations would prevent the switcher from attaining a

LT1576

V

SW

TO FREQUENCY

OUTPUT

5V

SHIFTING

1.4V

Q1

ERROR

AMPLIFIER

R1

1.21V

+

–

R3

1k

R4

1k

FB

+

R5

5k

Q2

R2

5k

TO SYNC CIRCUIT

V

C

GND

1576 F02

Figure 2. Frequency and Current Limit Foldback

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finite inductor size, maximum load current is reduced by

one-half peak-to-peak inductor current. The following

formula assumes continuous mode operation, implying

that the term on the right is less than one-half of I_P.

The internal circuitry which forces reduced switching

frequency also causes current to flow out of the feedback

pin when output voltage is low. The equivalent circuitry is

shown in Figure 2. Q1 is completely off during normal

operation. If the FB pin falls below 0.7V, Q1 begins to

conduct current and reduces frequency at the rate of

approximately 1kHz/µA. To ensure adequate frequency

foldback (under worst-case short-circuit conditions), the

external divider Thevinin resistance must be low enough

V_OUTV − V_OUT

(

)( ^IN

)

I_OUT(MAX)

=

I_P−

Continuous Mode

2 L f V

( )( )( ^IN

)

to pull 35µA out of the FB pin with 0.5V on the pin (R_DIV

≤

For the conditions above and L = 15µH,

14.3k). The net result is that reductions in frequency and

current limit are affected by output voltage divider imped-

ance. Although divider impedance is not critical, caution

should be used if resistors are increased beyond the

suggested values and short-circuit conditions will occur

with high input voltage. High frequency pickup will

increase and the protection accorded by frequency and

current foldback will decrease.

5 8 − 5

( )(

)

I_{OUT MAX}₎= 1.43 −

(

2 15 •10⁻⁶200•10³

8

( )

(

)(

)

=1.43 − 0.31= 1.12A

AtV_IN=15V, dutycycleis33%, soI_Pisjustequaltoafixed

1.5A, and I_OUT(MAX)is equal to:

MAXIMUM OUTPUT LOAD CURRENT

5 15 − 5

( )(

)

Maximum load current for a buck converter is limited by

the maximum switch current rating (I_P) of the LT1576.

This current rating is 1.5A up to 50% duty cycle (DC),

decreasing to 1.3A at 80% duty cycle. This is shown

graphically in Typical Performance Characteristics and as

shown in the formula below:

1.5 −

2 15 •10⁻⁶200•10³15

( )

(

)(

)

= 1.5 − 0.56 = 0.94A

Note that there is less load current available at the higher

input voltage because inductor ripple current increases.

This is not always the case. Certain combinations of

inductor value and input voltage range may yield lower

available load current at the lowest input voltage due to

reduced peak switch current at high duty cycles. If load

current is close to the maximum available, please check

maximum available current at both input voltage

extremes. To calculate actual peak switch current with a

given set of conditions, use:

I_P= 1.5A for DC ≤ 50%

I_P= 1.67 – 0.18 (DC) – 0.32(DC)²for 50% < DC < 90%

DC = Duty cycle = V_OUT/V_IN

Example: with V_OUT= 5V, V_IN= 8V; DC = 5/8 = 0.625, and;

I_SW(MAX)= 1.67 – 0.18 (0.625) – 0.32(0.625)²= 1.43A

Current rating decreases with duty cycle because the

LT1576 has internal slope compensation to prevent cur-

rent mode subharmonic switching. For more details, read

Application Note 19. The LT1576 is a little unusual in this

regardbecauseithasnonlinearslopecompensationwhich

gives better compensation with less reduction in current

limit.

V_OUTV − V_OUT

(

IN

)

I_{SW PEAK}=I_OUT

+

(

)

2 L f V

( )( )( ^IN

)

For lighter loads where discontinuous operation can be

used, maximum load current is equal to:

Maximum load current would be equal to maximum

switch current for an infinitely large inductor, but with

10

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2

Assume that the average inductor current is equal to

load current and decide whether or not the inductor

must withstand continuous fault conditions. If maxi-

mum load current is 0.5A, for instance, a 0.5A inductor

may not survive a continuous 1.5A overload condition.

Dead shorts will actually be more gentle on the induc-

tor because the LT1576 has foldback current limiting.

I_OUT(MAX)

=

I

f L V

IN

( ) ( )( )(

)

P

Discontinuous mode

2 V

V − V

(

)(

)

OUT

IN

OUT

Example: with L = 5µH, V_OUT= 5V, and V_IN(MAX) = 15V,

2

3

−6

1.5 200• 10 5• 10

15

( )

(

)

I

=

= 0.34A

OUT MAX

(

)

2 5 15 − 5

( )(

2. Calculate peak inductor current at full load current to

ensure that the inductor will not saturate. Peak current

can be significantly higher than output current, espe-

cially with smaller inductors and lighter loads, so don’t

omit this step. Powdered iron cores are forgiving

because they saturate softly, whereas ferrite cores

saturate abruptly. Other core materials fall somewhere

in between. The following formula assumes continu-

ous mode of operation, but it errs only slightly on the

high side for discontinuous mode, so it can be used for

all conditions.

)

The main reason for using such a tiny inductor is that it is

physically very small, but keep in mind that peak-to-peak

inductorcurrentwillbeveryhigh. Thiswillincreaseoutput

ripplevoltage.Iftheoutputcapacitorhastobemadelarger

to reduce ripple voltage, the overall circuit could actually

wind up larger.

CHOOSING THE INDUCTOR AND OUTPUT CAPACITOR

For most applications the output inductor will fall in the

rangeof15µHto60µH. Lowervaluesarechosentoreduce

physical size of the inductor. Higher values allow more

output current because they reduce peak current seen by

the LT1576 switch, which has a 1.5A limit. Higher values

also reduce output ripple voltage, and reduce core loss.

GraphsintheTypicalPerformanceCharacteristicssection

show maximum output load current versus inductor size

andinputvoltage. Asecondgraphshowscorelossversus

inductor size for various core materials.

V_OUTV − V

(

)

IN

OUT

I_PEAK=I_OUT+

2 f L V

( )( )(

)

IN

V_IN= Maximum input voltage

f = Switching frequency, 200kHz

3. Decide if the design can tolerate an “open” core geom-

etry like a rod or barrel, with high magnetic field

radiation, orwhetheritneedsaclosedcorelikeatoroid

to prevent EMI problems. One would not want an open

core next to a magnetic storage media, for instance!

Thisisatoughdecisionbecausetherodsorbarrelsare

temptingly cheap and small and there are no helpful

guidelines to calculate when the magnetic field radia-

tion will be a problem.

When choosing an inductor you might have to consider

maximum load current, core and copper losses, allowable

component height, output voltage ripple, EMI, fault cur-

rent in the inductor, saturation, and of course, cost. The

following procedure is suggested as a way of handling

thesesomewhatcomplicatedandconflictingrequirements.

4. Start shopping for an inductor (see representative

surfacemountunitsinTable2)whichmeetstherequire-

mentsofcoreshape,peakcurrent(toavoidsaturation),

average current (to limit heating), and fault current (if

the inductor gets too hot, wire insulation will melt and

cause turn-to-turn shorts). Keep in mind that all good

thingslikehighefficiency,lowprofile,andhightempera-

ture operation will increase cost, sometimes dramati-

cally. Get a quote on the cheapest unit first to calibrate

yourself on price, then ask for what you really want.

1. Choose a value in microhenries from the graphs of

maximumloadcurrentandcoreloss.Choosingasmall

inductor may result in discontinuous mode operation

at lighter loads, but the LT1576 is designed to work

well in either mode. Keep in mind that lower core loss

means higher cost, at least for closed core geometries

like toroids. The core loss graphs show both absolute

lossandpercentlossfora5Woutput,soactualpercent

losses must be calculated for each situation.

11

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Output Capacitor

5. After making an initial choice, consider the secondary

things like output voltage ripple, second sourcing, etc.

Use the experts in the Linear Technology’s applica-

tions department if you feel uncertain about the final

choice. They have experience with a wide range of

inductor types and can tell you about the latest devel-

opments in low profile, surface mounting, etc.

The output capacitor is normally chosen by its Effective

Series Resistance (ESR), because this is what determines

output ripple voltage. To get low ESR takes volume, so

physically smaller capacitors have high ESR. The ESR

range for typical LT1576 applications is 0.05Ω to 0.2Ω. A

typical output capacitor is an AVX type TPS, 100µF at 10V,

with a guaranteed ESR less than 0.1Ω. This is a “D” size

surface mount solid tantalum capacitor. TPS capacitors

are specially constructed and tested for low ESR, so they

give the lowest ESR for a given volume. The value in

microfarads is not particularly critical, and values from

22µF to greater than 500µF work well, but you cannot

cheat mother nature on ESR. If you find a tiny 22µF solid

tantalumcapacitor, itwillhavehighESR, andoutputripple

voltage will be terrible. Table 3 shows some typical solid

tantalum surface mount capacitors.

Table 2

SERIES

CORE

VENDOR/

PART NO.

VALUE

DC

CORE RESIS- MATER- HEIGHT

(µ

H) (Amps) TYPE TANCE(

Ω

)

IAL

(mm)

Coiltronics

CTX15-2

15

33

68

15

33

68

1.7

1.4

1.2

1.4

1.3

1.1

Tor

0.059

KMµ

52

6.0

6.4

4.2

6.0

6.4

CTX33-2

0.106

0.158

0.087

0.126

0.238

CTX68-4

CTX15-1P

CTX33-2P

52

Table 3. Surface Mount Solid Tantalum Capacitor ESR

and Ripple Current

E Case Size

CTX68-4P

52

Sumida

ESR (Max.,

Ω

)

Ripple Current (A)

0.7 to 1.1

0.4

CDRH74-150

CDH115-330

CDRH125-680

CDH74-330

Coilcraft

15

33

68

33

1.47

1.68

1.5

SC

0.081

0.082

0.12

Fer

4.5

5.2

6

AVX TPS, Sprague 593D

AVX TAJ

0.1 to 0.3

0.7 to 0.9

D Case Size

1.45

0.17

5.2

AVX TPS, Sprague 593D

C Case Size

0.1 to 0.3

0.2 (typ)

0.7 to 1.1

0.5 (typ)

DO3308P-153

DO3316P-333

DO3316P-683

Pulse

15

33

68

2

SC

0.12

0.1

Fer

3

AVX TPS

5.21

1.4

0.18

Many engineers have heard that solid tantalum capacitors

are prone to failure if they undergo high surge currents.

This is historically true, and type TPS capacitors are

speciallytestedforsurgecapability,butsurgeruggedness

is not a critical issue with the output capacitor. Solid

tantalum capacitors fail during very high turn-on surges,

which do not occur at the output of regulators. High

discharge surges, such as when the regulator output is

dead shorted, do not harm the capacitors.

PE-53602

35

73

22

40

1.4

1.3

2.7

Tor

0.166

0.290

0.063

0.085

Fer

9.1

10

PE-53604

PE-53632

PE-53633

Gowanda

SMP3316-152K

SMP3316-332K

SMP3316-682K

Tor = Toroid

15

33

68

3.5

2.3

1.7

SC

0.041

0.092

0.178

Fer

6

Unlike the input capacitor, RMS ripple current in the

output capacitor is normally low enough that ripple cur-

rent rating is not an issue. The current waveform is

triangular with a typical value of 200mA_RMS. The formula

to calculate this is:

SC = Semi-closed geometry

Fer = Ferrite core material

52 = Type 52 powdered iron core material

KMµ = Kool Mµ

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Output Capacitor Ripple Current (RMS):

dI

dt

V_RIPPLE= I

ESR + ESL Σ

(

_P-P)(

) (

)

0.29 V

V − V

IN OUT

(

_OUT)(

)

Example: withV_IN=10V,V_OUT=5V,L=30µH,ESR=0.1Ω,

I_{RIPPLE RMS}

=

(

)

L f V

ESL = 10nH:

( )( )( )

IN

5 10 − 5

Ceramic Capacitors

( )(

)

I

=

= 0.42A

P-P

−6

3

Higher value, lower cost ceramic capacitors are now

becomingavailableinsmallercasesizes.Thesearetempt-

ing for switching regulator use because of their very low

ESR. Unfortunately, the ESR is so low that it can cause

loop stability problems. Solid tantalum capacitor’s ESR

generatesaloop“zero”at5kHzto50kHzthatisinstrumen-

tal in giving acceptable loop phase margin. Ceramic

capacitors remain capacitive to beyond 300kHz and usu-

allyresonatewiththeirESLbeforeESRbecomeseffective.

They are appropriate for input bypassing because of their

highripplecurrentratingsandtoleranceofturn-onsurges.

10 30• 10

200• 10

( )

dI

dt

10

6

Σ

=

= 0.33• 10

−6

30• 10

−9

6

V

= 0.42A 0.1 + 10• 10

0.33• 10

(

)( )

RIPPLE

= 0.042 + 0.003 = 45mV

P-P

20mV/DIV

V_OUTAT

_OUT= 1A

I

INDUCTOR

CURRENT

AT I_OUT= 1A

200mA/DIV

OUTPUT RIPPLE VOLTAGE

Figure 3 shows a typical output ripple voltage waveform

for the LT1576. Ripple voltage is determined by the high

frequency impedance of the output capacitor, and ripple

current through the inductor. Peak-to-peak ripple current

through the inductor into the output capacitor is:

20mV/DIV

V_OUTAT

I_OUT= 50mA

INDUCTOR

200mA/DIV

CURRENT

AT I_OUT= 50mA

2µs/DIV

1576 F03

Figure 3. LT1576 Ripple Voltage Waveform

V

V − V

IN OUT

(

_OUT)(

)

I_P-P

=

V

L f

_IN)( )( )

(

CATCH DIODE

For high frequency switchers, the sum of ripple current

slew rates may also be relevant and can be calculated

from:

The suggested catch diode (D1) is a 1N5818 Schottky, or

its Motorola equivalent, MBR130. It is rated at 1A average

forward current and 30V reverse voltage. Typical forward

voltage is 0.42V at 1A. The diode conducts current only

during switch off time. Peak reverse voltage is equal to

regulatorinputvoltage.Averageforwardcurrentinnormal

operation can be calculated from:

dI

dt

V

IN

L

Σ

=

Peak-to-peak output ripple voltage is the sum of a triwave

created by peak-to-peak ripple current times ESR, and a

square wave created by parasitic inductance (ESL) and

ripple current slew rate. Capacitive reactance is assumed

to be small compared to ESR or ESL.

I_OUTV − V

(

)

IN

OUT

I_D₍_AVG

=

)

V

IN

13

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This formula will not yield values higher than 1A with

maximumloadcurrentof1.25Aunlesstheratioofinputto

output voltage exceeds 5:1. The only reason to consider a

larger diode is the worst-case condition of a high input

voltageandoverloaded(notshorted)output. Undershort-

circuit conditions, foldback current limit will reduce diode

current to less than 1A, but if the output is overloaded and

does not fall to less than 1/3 of nominal output voltage,

foldback will not take effect. With the overloaded condi-

tion, output current will increase to a typical value of 1.8A,

determined by peak switch current limit of 2A. With

V_IN= 15V, V_OUT= 4V (5V overloaded) and I_OUT= 1.8A:

For nearly all applications, a 0.33µF boost capacitor works

just fine, but for the curious, more details are provided

here. The size of the boost capacitor is determined by

switch drive current requirements. During switch on time,

draincurrentonthecapacitorisapproximatelyI_OUT/50.At

peakloadcurrentof1.25A,thisgivesatotaldrainof25mA.

Capacitor ripple voltage is equal to the product of on time

and drain current divided by capacitor value;

∆V = (t_ON)(25mA/C). To keep capacitor ripple voltage to

less than 0.5V (a slightly arbitrary number) at the worst-

case condition of t_ON= 4.7µs, the capacitor needs to be

0.24µF. Boost capacitor ripple voltage is not a critical

parameter, but if the minimum voltage across the capaci-

tor drops to less than 3V, the power switch may not

saturate fully and efficiency will drop. An approximate

formula for absolute minimum capacitor value is:

1.8 15 − 4

(

)

I

=

= 1.32A

D AVG

(

)

15

This is safe for short periods of time, but it would be

prudent to check with the diode manufacturer if continu-

ous operation under these conditions must be tolerated.

I

(

/50 V_OUT/ V

)(

)

OUT

IN

C_MIN

=

f V −3V

( )(

)

OUT

f = Switching frequency

V_OUT= Regulated output voltage

V_IN= Minimum input voltage

BOOST PIN CONSIDERATIONS

Formostapplications, theboostcomponentsarea0.33µF

capacitor and a 1N914 or 1N4148 diode. The anode is

connected to the regulated output voltage and this gener-

ates a voltage across the boost capacitor nearly identical

to the regulated output. In certain applications, the anode

may instead be connected to the unregulated input volt-

age. This could be necessary if the regulated output

voltage is very low (< 3V) or if the input voltage is less than

6V. Efficiencyisnotaffectedbythecapacitorvalue, butthe

capacitor should have an ESR of less than 1Ω to ensure

that it can be recharged fully under the worst-case condi-

tion of minimum input voltage. Almost any type of film or

ceramic capacitor will work fine.

This formula can yield capacitor values substantially less

than 0.24µF, but it should be used with caution since it

does not take into account secondary factors such as

capacitor series resistance, capacitance shift with tem-

perature and output overload.

SHUTDOWN FUNCTION AND

UNDERVOLTAGE LOCKOUT

Figure 4 shows how to add undervoltage lockout (UVLO)

to the LT1576. Typically, UVLO is used in situations where

the input supply is current limited, or has a relatively high

source resistance. A switching regulator draws constant

power from the source, so source current increases as

source voltage drops. This looks like a negative resistance

loadtothesourceandcancausethesourcetocurrentlimit

or latch low under low source voltage conditions. UVLO

prevents the regulator from operating at source voltages

where these problems might occur.

WARNING! Peak voltage on the BOOST pin is the sum of

unregulated input voltage plus the voltage across the

boost capacitor. This normally means that peak BOOST

pin voltage is equal to input voltage plus output voltage,

but when the boost diode is connected to the regulator

input, peak BOOST pin voltage is equal to twice the input

voltage. Be sure that BOOST pin voltage does not exceed

its maximum rating.

14

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R

FB

LT1576

OUTPUT

V

SW

IN

INPUT

2.44V

–

+

STANDBY

R

HI

3.5µA

+

SHDN

+

–

TOTAL

SHUTDOWN

R

C1

0.4V

LO

GND

1576 F04

Figure 4. Undervoltage Lockout

Threshold voltage for lockout is about 2.44V. A 3.5µA bias

current flows out of the pin at threshold. This internally

generated current is used to force a default high state on

the shutdown pin if the pin is left open. When low shut-

down current is not an issue, the error due to this current

can be minimized by making R_LO10k or less. If shutdown

currentisanissue, R_LOcanberaisedto100k, buttheerror

due to initial bias current and changes with temperature

should be considered.

R

V − 2. ∆V/V

+1 + ∆V

44

(

)

LO IN

OUT

[

]

R =

HI

R

2.44 −

3.5µA

(

LO

)

R = R

V

/

∆V

(

)(

)

FB

HI OUT

25k suggested for R_LO

V_IN= Input voltage at which switching stops as input

voltage descends to trip level

∆V = Hysteresis in input voltage level

Example: output voltage is 5V, switching is to stop if input

voltage drops below 12V and should not restart unless

input rises back to 13.5V. ∆V is therefore 1.5V and

V_IN= 12V. Let R_LO= 25k.

R

= 10k to 100k 25k suggested

(

)

LO

R

V − 2.44V

(

)

LO IN

R =

HI

2.44V −R 3.5µA

(

LO

)

V_IN= Minimum input voltage

25k 12 − 2. 1.5/5 +1 + 1.5

44

(

)

[

]

R =

HI

Keep the connections from the resistors to the shutdown

pin short and make sure that interplane or surface capaci-

tance to the switching nodes are minimized. If high resis-

tor values are used, the shutdown pin should be bypassed

with a 1000pF capacitor to prevent coupling problems

from the switch node. If hysteresis is desired in the

undervoltage lockout point, a resistor R_FBcan be added to

the output node. Resistor values can be calculated from:

2.44 − 25k 3.5µA

(

)

25k10.33

(

)

=

=110k

2.35

R = 110k 5/1.5 = 366k

(

)

FB

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SWITCH NODE CONSIDERATIONS

under the switcher circuitry to prevent interplane cou-

pling. A suggested layout for the critical components is

shown in Figure 5. Note that the feedback resistors and

compensation components are kept as far as possible

from the switch node. Also note that the high current

groundpathofthecatchdiodeandinputcapacitorarekept

very short and separate from the analog ground line.

For maximum efficiency, switch rise and fall times are

made as short as possible. To prevent radiation and high

frequency resonance problems, proper layout of the com-

ponents connected to the switch node is essential. B field

(magnetic) radiation is minimized by keeping catch diode,

switch pin, and input bypass capacitor leads as short as

possible. E field radiation is kept low by minimizing the

length and area of all traces connected to the switch pin

and BOOST pin. A ground plane should always be used

Thehighspeedswitchingcurrentpathisshownschemati-

cally in Figure 6. Minimum lead length in this path is

essential to ensure clean switching and low EMI. The path

TAKE OUTPUT DIRECTLY FROM END

CONNECT OUTPUT

CAPACITOR DIRECTLY

TO HEAVY GROUND

OF OUTPUT CAPACITOR TO AVOID

PARASITIC RESISTANCE AND

INDUCTANCE (KELVIN CONNECTION)

C1

V

OUT

MINIMUM SIZE

OF FEEDBACK PIN

CONNECTIONS

MINIMIZE AREA

OF CONNECTIONS

TO SWITCH NODE

AND BOOST NODE

L1

D2

TO AVOID PICKUP

SHDN/SYNC

C2

SW

IN

KEEP INPUT

CAPACITOR

AND CATCH

R2

D1

C3

TERMINATE

V

FB

DIODE CLOSE

TO REGULATOR

AND TERMINATE

THEM TO THE

SAME POINT

FEEDBACK

RESISTORS AND

COMPENSATION

COMPONENTS

DIRECTLY TO

SWITCHER

C

BOOST

V

C

R1

GND

R

GROUND PIN

C

GND

GROUND RING NEED NOT BE AS SHOWN

(NORMALLY EXISTS AS INTERNAL PLANE)

1576 F05

Figure 5. Suggested Layout for LT1576

SWITCH NODE

L1

5V

HIGH

FREQUENCY

CIRCULATING

PATH

V

IN

LOAD

1576 F06

Figure 6. High Speed Switching Path

16

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including the switch, catch diode, and input capacitor is

the only one containing nanosecond rise and fall times. If

you follow this path on the PC layout, you will see that it is

irreducibly short. If you move the diode or input capacitor

away from the LT1576, get your resumé in order. The

other paths contain only some combination of DC and

200kHz triwave, so are much less critical.

higher with a poor layout, potentially exceeding the abso-

lute max switch voltage. The path around switch, catch

diode and input capacitor must be kept as short as

possibletoensurereliableoperation.Whenlookingatthis,

a >100MHz oscilloscope must be used, and waveforms

should be observed on the leads of the package. This

switch off spike will also cause the SW node to go below

ground. The LT1576 has special circuitry inside which

mitigates this problem, but negative voltages over 1V

lasting longer than 10ns should be avoided. Note that

100MHz oscilloscopes are barely fast enough to see the

details of the falling edge overshoot in Figure 7.

PARASITIC RESONANCE

Resonance or “ringing” may sometimes be seen on the

switch node (see Figure 7). Very high frequency ringing

following switch rise time is caused by switch/diode/input

capacitor lead inductance and diode capacitance. Schot-

tky diodes have very high “Q” junction capacitance that

can ring for many cycles when excited at high frequency.

Iftotalleadlengthfortheinputcapacitor, diodeandswitch

path is 1 inch, the inductance will be approximately 25nH.

At switch off, this will produce a spike across the NPN

output device in addition to the input voltage. At higher

currents this spike can be in the order of 10V to 20V or

A second, much lower frequency ringing is seen during

switch off time if load current is low enough to allow the

inductor current to fall to zero during part of the switch off

time (see Figure 8). Switch and diode capacitance reso-

nate with the inductor to form damped ringing at 1MHz to

10 MHz. This ringing is not harmful to the regulator and it

hasnotbeenshowntocontributesignificantlytoEMI. Any

attempt to damp it with a resistive snubber will degrade

efficiency.

INPUT BYPASSING AND VOLTAGE RANGE

Input Bypass Capacitor

RISE AND FALL

WAVEFORMS ARE

SUPERIMPOSED

(PULSE WIDTH IS

NOT 350ns)

5V/DIV

Step-down converters draw current from the input supply

in pulses. The average height of these pulses is equal to

load current, and the duty cycle is equal to V_OUT/V_IN. Rise

and fall time of the current is very fast. A local bypass

capacitor across the input supply is necessary to ensure

proper operation of the regulator and minimize the ripple

current fed back into the input supply. The capacitor also

forces switching current to flow in a tight local loop,

minimizing EMI.

50ns/DIV

1374 F07

Figure 7. Switch Node Response

5V/DIV

Do not cheat on the ripple current rating of the Input

bypass capacitor, but also don’t get hung up on the value

in microfarads. The input capacitor is intended to absorb

all the switching current ripple, which can have an RMS

value as high as one half of load current. Ripple current

ratings on the capacitor must be observed to ensure

reliable operation. In many cases it is necessary to parallel

two capacitors to obtain the required ripple rating. Both

capacitors must be of the same value and manufacturer to

SWITCH NODE

VOLTAGE

50mA/DIV

INDUCTOR

CURRENT

1µs/DIV

1374 F08

Figure 8. Discontinuous Mode Ringing

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guaranteepowersharing. Theactualvalueofthecapacitor

in microfarads is not particularly important because at

200kHz, any value above 15µF is essentially resistive.

RMS ripple current rating is the critical parameter. Actual

RMS current can be calculated from:

(AVX TPS series for instance, see Table 3), but even these

units may fail if the input voltage surge approaches the

maximum voltage rating of the capacitor. AVX recom-

mends derating capacitor voltage by 2:1 for high surge

applications. The highest voltage rating is 50V, so 25V

may be a practical upper limit when using solid tantalum

capacitors for input bypassing.

2

I

=I

V

V − V

/V

IN

(

)

RIPPLE RMS

OUT OUT IN

OUT

(

)

Larger capacitors may be necessary when the input volt-

age is very close to the minimum specified on the data

sheet. Small voltage dips during switch on time are not

normallyaproblem, butatverylowinputvoltagetheymay

cause erratic operation because the input voltage drops

below the minimum specification. Problems can also

occur if the input-to-output voltage differential is near

minimum. The amplitude of these dips is normally a

function of capacitor ESR and ESL because the capacitive

reactance is small compared to these terms. ESR tends to

be the dominate term and is inversely related to physical

capacitor size within a given capacitor type.

The term inside the radical has a maximum value of 0.5

when input voltage is twice output, and stays near 0.5 for

a relatively wide range of input voltages. It is common

practice therefore to simply use the worst-case value and

assumethatRMSripplecurrentisonehalfofloadcurrent.

At maximum output current of 1.5A for the LT1576, the

input bypass capacitor should be rated at 0.75A ripple

current. Note however, that there are many secondary

considerations in choosing the final ripple current rating.

These include ambient temperature, average versus peak

load current, equipment operating schedule, and required

product lifetime. For more details, see Application Notes

19 and 46, and Design Note 95.

SYNCHRONIZING (Available as -SYNC Option)

The LT1576-SYNC has the SHDN pin replaced with a

SYNC pin, which is used to synchronize the internal

oscillator to an external signal. The SYNC input must pass

from a logic level low, through the maximum synchroni-

zation threshold with a duty cycle between 10% and 90%.

The input can be driven directly from a logic level output.

The synchronizing range is equal to initial operating fre-

quency up to 400kHz. This means that minimum practical

sync frequency is equal to the worst-case high self-

oscillating frequency (250kHz), not the typical operating

frequency of 200kHz. Caution should be used when syn-

chronizing above 280kHz because at higher sync frequen-

cies the amplitude of the internal slope compensation

used to prevent subharmonic switching is reduced. This

type of subharmonic switching only occurs at input volt-

ages less than twice output voltage. Higher inductor

values will tend to eliminate this problem. See Frequency

Compensation section for a discussion of an entirely

different cause of subharmonic switching before assum-

ing that the cause is insufficient slope compensation.

ApplicationNote19hasmoredetailsonthetheoryofslope

compensation.

Input Capacitor Type

Some caution must be used when selecting the type of

capacitor used at the input to regulators. Aluminum

electrolytics are lowest cost, but are physically large to

achieve adequate ripple current rating, and size con-

straints (especially height), may preclude their use.

Ceramic capacitors are now available in larger values, and

their high ripple current and voltage rating make them

ideal for input bypassing. Cost is fairly high and footprint

may also be somewhat large. Solid tantalum capacitors

would be a good choice, except that they have a history of

occasionalspectacularfailureswhentheyaresubjectedto

large current surges during power-up. The capacitors can

short and then burn with a brilliant white light and lots of

nasty smoke. This phenomenon occurs in only a small

percentage of units, but it has led some OEM companies

to forbid their use in high surge applications. The input

bypass capacitor of regulators can see these high surges

when a battery or high capacitance source is connected.

Several manufacturers have developed a line of solid

tantalum capacitors specially tested for surge capability

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At power-up, when V_Cis being clamped by the FB pin (see

Figure2,Q2),thesyncfunctionisdisabled.Thisallowsthe

frequency foldback to operate in the shorted output con-

dition. During normal operation, switching frequency is

controlledbytheinternaloscillatoruntiltheFBpinreaches

0.7V, after which the SYNC pin becomes operational. If no

synchronization is required, this pin should be connected

to ground.

2

0.2 1 5

(

)( ) ( )

−9

3

P

=

+ 60• 10

1 10 200• 10

( )( )

SW

10

= 0.1 + 0.12 = 0.22W

2

5 1/50

( ) (

)

P

=

= 0.05W

BOOST

10

2

5 0.004

( ) (

)

−3

P =10 0.55• 10

+5 1.6• 10

+

Q

THERMAL CALCULATIONS

10

= 0.02W

Power dissipation in the LT1576 chip comes from four

sources: switch DC loss, switch AC loss, boost circuit

current,andinputquiescentcurrent.Thefollowingformu-

las show how to calculate each of these losses. These

formulas assume continuous mode operation, so they

should not be used for calculating efficiency at light load

currents.

Total power dissipation is 0.22 + 0.05 + 0.02 = 0.29W.

Thermal resistance for LT1576 package is influenced by

the presence of internal or backside planes. With a full

plane under the SO package, thermal resistance will be

about 80°C/W. No plane will increase resistance to about

120°C/W. To calculate die temperature, add in worst-case

ambient temperature:

Switch loss:

2

R

I

V

OUT

(

) (

)

SW OUT

T_J= T_A+ θ_JA(P_TOT

)

P

=

+ 60ns I

V

f

(

)( )( )

SW

OUT IN

V

IN

With the SO-8 package (θ_JA= 80°C/W), at an ambient

temperature of 50°C,

Boost current loss:

T_J= 50 + 80 (0.29) = 73.2°C

2

V

I

/50

(

)

OUT OUT

Die temperature is highest at low input voltage, so use

lowest continuous input operating voltage for thermal

calculations.

P

=

BOOST

V

IN

Quiescent current loss:

FREQUENCY COMPENSATION

−3

P = V 0.55• 10

+ V

1.6• 10

Q

IN

OUT

Loop frequency compensation of switching regulators

can be a rather complicated problem because the reactive

components used to achieve high efficiency also intro-

duce multiple poles into the feedback loop. The inductor

and output capacitor on a conventional step-down con-

verter actually form a resonant tank circuit that can exhibit

peaking and a rapid 180° phase shift at the resonant

frequency. Bycontrast, theLT1576usesa“currentmode”

architecture to help alleviate phase shift created by the

inductor. The basic connections are shown in Figure 9.

Figure 10 shows a Bode plot of the phase and gain of the

power section of the LT1576, measured from the V_Cpin to

2

V

0.004

(

)

OUT

+

V

IN

R_SW= Switch resistance (≈0.2Ω)

60ns = Equivalent switch current/voltage overlap time

f = Switch frequency

Example: with V_IN= 10V, V_OUT= 5V and I_OUT= 1A:

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the output. Gain is set by the 1.5A/V transconductance of

the LT1576 power section and the effective complex

impedance from output to ground. Gain rolls off smoothly

above the 160Hz pole frequency set by the 100µF output

capacitor. Phase drop is limited to about 85°. Phase

recoversandgainlevelsoffatthezerofrequency(≈16kHz)

set by capacitor ESR (0.1Ω).

This means that the error amplifier characteristics them-

selvesdonotcontributeexcessphaseshifttotheloop,and

the phase/gain characteristics of the error amplifier sec-

tion are completely controlled by the external compensa-

tion network.

In Figure 12, full loop phase/gain characteristics are

shownwithacompensationcapacitorof100pF, givingthe

error amplifier a pole at 2.8kHz, with phase rolling off to

90° and staying there. The overall loop has a gain of 66dB

at low frequency, rolling off to unity-gain at 58kHz. Phase

showsatwo-polecharacteristicuntiltheESRoftheoutput

capacitor brings it back above 16kHz. Phase margin is

about 77° at unity-gain.

Erroramplifiertransconductancephaseandgainareshown

in Figure 11. The error amplifier can be modeled as a

transconductance of 1000µMho, with an output imped-

ance of 570kΩ in parallel with 2.4pF. In all practical

applications, the compensation network from V_Cpin to

ground has a much lower impedance than the output

impedance of the amplifier at frequencies above 200Hz.

2000

1500

1000

500

200

150

100

50

LT1576

CURRENT MODE

POWER STAGE

V

SW

FB

PHASE

GAIN

OUTPUT

ERROR

g

= 1.5A/V

m

AMPLIFIER

R1

R2

–

V

C

ESR

C1

+

1.21V

C

R

OUT

2.4pF

–3

OUT

570k

V

1 × 10

(

)

+

FB

V

C

GND

0

ERROR AMPLIFIER EQUIVALENT CIRCUIT

= 50Ω

0

R

C

R

LOAD

C

F

–500

–50

C

10

100

1k

10k

100k

1M

C

FREQUENCY (Hz)

1576 F11

1576 F09

Figure 9. Model for Loop Response

Figure 11. Error Amplifier Gain and Phase

40

20

0

40

80

60

180

135

90

V

= 10V

IN

= 5V

OUT

I

= 500mA

0

GAIN

PHASE

GAIN

40

PHASE

–40

–80

–120

V

= 10V

IN

20

45

= 5V

OUT

I

= 500mA

= 100µF

–20

–40

C

0

10V, AVX TPS

C

= 100pF

C

L = 30µH

–20

–45

1M

10

100

1k

FREQUENCY (Hz)

10k

100k

10

100

1k

10k

100k

FREQUENCY (Hz)

1576 F12

1576 F07

Figure 10. Response from V_CPin to Output

Figure 12. Overall Loop Characteristics

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Analog experts will note that around 7kHz, phase dips

close to the zero phase margin line. This is typical of

switching regulators, especially those that operate over a

wide range of loads. This region of low phase is not a

problem as long as it does not occur near unity-gain. In

practice, the variability of output capacitor ESR tends to

dominate all other effects with respect to loop response.

Variations in ESR will cause unity-gain to move around,

but at the same time phase moves with it so that adequate

phase margin is maintained over a very wide range of ESR

(≥ ±3:1).

subharmonic switching occurs, as evidenced by alternat-

ing pulse widths seen at the switch node. In more severe

cases,theregulatorsquealsorhissesaudiblyeventhough

the output voltage is still roughly correct. None of this will

show on a theoretical Bode plot because Bode is an

amplitude insensitive analysis. Tests have shown that if

ripple voltage on the V_Cis held to less than 100mV_P-P, the

LT1576 will be well behaved. The formula below will give

an estimate of V_Cripple voltage when R_Cis added to the

loop, assuming that R_Cis large compared to the reactance

of C_Cat 200kHz.

What About a Resistor in the Compensation Network?

R G

V − V

ESR 1.21

( )(

)(

)

C

MA IN

OUT

V

=

C RIPPLE

It is common practice in switching regulator design to add

a “zero” to the error amplifier compensation to increase

loop phase margin. This zero is created in the external

network in the form of a resistor (R_C) in series with the

compensation capacitor. Increasing the size of this resis-

tor generally creates better and better loop stability, but

there are two limitations on its value. First, the combina-

tion of output capacitor ESR and a large value for R_Cmay

cause loop gain to stop rolling off altogether, creating a

gain margin problem. An approximate formula for R_C

where gain margin falls to zero is:

(

)

V

L f

(

)( )( )

IN

G_MA= Error amplifier transconductance (1000µMho)

If a computer simulation of the LT1576 showed that a

seriescompensationresistorof15kgavebestoverallloop

response, with adequate gain margin, the resulting V_Cpin

ripple voltage with V_IN= 10V, V_OUT= 5V, ESR = 0.1Ω,

L = 30µH, would be:

15k 1•10⁻³10 − 5 0.1 1.21

(

)

(

)( )(

)

(

)

V_C₍_RIPPLE

=

= 0.151V

)

10 30•10⁻⁶200•10³

( )

(

)(

)

V

OUT

R Loop Gain = 1 =

(

)

C

This ripple voltage is high enough to possibly create

subharmonic switching. In most situations a compromise

value (<10k in this case) for the resistor gives acceptable

phase margin and no subharmonic problems. In other

cases, the resistor may have to be larger to get acceptable

phaseresponse, andsomemeansmustbeusedtocontrol

ripple voltage at the V_Cpin. The suggested way to do this

istoaddacapacitor(C_F)inparallelwiththeR_C/C_Cnetwork

on the V_Cpin. Pole frequency for this capacitor is typically

set at one-fifth of switching frequency so that it provides

significant attenuation of switching ripple, but does not

addunacceptablephaseshiftatloopunity-gainfrequency.

With R_C= 15k,

G

(

G

ESR 1.21

)(

)

MP MA

G_MP= Transconductance of power stage = 1.5A/V

G_MA= Error amplifier transconductance = 1(10^–3)

ESR = Output capacitor ESR

1.21 = Reference voltage

With V_OUT= 5V and ESR = 0.1Ω, a value of 27.5k for R_C

would yield zero gain margin, so this represents an upper

limit. There is a second limitation however which has

nothing to do with theoretical small signal dynamics. This

resistor sets high frequency gain of the error amplifier,

including the gain at the switching frequency. If switching

frequency gain is high enough, output ripple voltage will

appear at the V_Cpin with enough amplitude to muck up

proper operation of the regulator. In the marginal case,

5

)( )(

5

C_F=

=

= 265pF

2π 200•10³15k

2π f R

(

)

C

(

)

(

)

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How Do I Test Loop Stability?

I check switching regulator loop stability by pulse loading

the regulator output while observing transient response at

the output, using the circuit shown in Figure 13. The

regulator loop is “hit” with a small transient AC load

current at a relatively low frequency, 50Hz to 1kHz. This

causes the output to jump a few millivolts, then settle back

totheoriginalvalue,asshowninFigure14. Awellbehaved

loop will settle back cleanly, whereas a loop with poor

phase or gain margin will “ring” as it settles. The number

ofringsindicatesthedegreeofstability, andthefrequency

of the ringing shows the approximate unity-gain fre-

quency of the loop. Amplitude of the signal is not particu-

larlyimportant, aslongastheamplitudeisnotsohighthat

the loop behaves nonlinearly.

The “standard” compensation for LT1576 is a 100pF

capacitor for C_C, with R_C= 0Ω. While this compensation

will work for most applications, the “optimum” value for

loop compensation components depends, to various ex-

tent, on parameters which are not well controlled. These

include inductor value (±30% due to production toler-

ance, load current and ripple current variations), output

capacitance (±20%to±50%duetoproductiontolerance,

temperature, aging and changes at the load), output

capacitor ESR (±200% due to production tolerance, tem-

perature and aging), and finally, DC input voltage and

output load current. This makes it important for the

designer to check out the final design to ensure that it is

“robust” and tolerant of all these variations.

RIPPLE FILTER

TO X1

OSCILLOSCOPE

PROBE

470Ω

4.7k

SWITCHING

REGULATOR

+

100µF TO

1000µF

3300pF

330pF

50Ω

ADJUSTABLE

INPUT SUPPLY

ADJUSTABLE

DC LOAD

TO

OSCILLOSCOPE

SYNC

100Hz TO 1kHz

100mV TO 1V

P-P

1576 F13

Figure 13. Loop Stability Test Circuit

V

_OUTAT

I_OUT= 500mA

BEFORE FILTER

V

_OUTAT

I_OUT= 500mA

AFTER FILTER

V_OUTAT

I_OUT= 50mA

AFTER FILTER

LOAD PULSE

THROUGH 50Ω

f ≈ 780Hz

10mV/DIV

5A/DIV

0.2ms/DIV

1576 F14

Figure 14. Loop Stability Check

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The output of the regulator contains both the desired low

frequency transient information and a reasonable amount

of high frequency (200kHz) ripple. The ripple makes it

difficult to observe the small transient, so a two-pole,

100kHz filter has been added. This filter is not particularly

critical; even if it attenuated the transient signal slightly,

this wouldn’t matter because amplitude is not critical.

probably not be a problem in production. Note that fre-

quency of the light load ringing may vary with component

tolerance but phase margin generally hangs in there.

POSITIVE-TO-NEGATIVE CONVERTER

The circuit in Figure 15 is a classic positive-to-negative

topology using a grounded inductor. It differs from the

standard approach in the way the IC chip derives its

feedback signal, however, because the LT1576 accepts

onlypositivefeedbacksignals,thegroundpinmustbetied

to the regulated negative output. A resistor divider to

ground or, in this case, the sense pin, then provides the

proper feedback voltage for the chip.

After verifying that the setup is working correctly, I start

varying load current and input voltage to see if I can find

any combination that makes the transient response look

suspiciously “ringy.” This procedure may lead to an ad-

justment for best loop stability or faster loop transient

response. Nearly always you will find that loop response

looks better if you add in several kΩ for R_C. Do this only

if necessary, because as explained before, R_Cabove 1k

may require the addition of C_Fto control V_Cpin ripple.

If everything looks OK, I use a heat gun and cold spray on

the circuit (especially the output capacitor) to bring out

any temperature-dependent characteristics.

D1

1N4148

C2

0.33µF

L1*

15µH

INPUT

5.5V TO

20V

BOOST

LT576

V

IN

SW

R1

15.8k

Keep in mind that this procedure does not take initial

component tolerance into account. You should see fairly

cleanresponseunderallloadandlineconditionstoensure

that component variations will not cause problems. One

note here: according to Murphy, the component most

likely to be changed in production is the output capacitor,

because that is the component most likely to have manu-

facturer variations (in ESR) large enough to cause prob-

lems. It would be a wise move to lock down the sources of

the output capacitor in production.

FB

+

C3

10µF TO

50µF

GND

V

C

C1

+

R2

100µF

10V TANT

×2

4.99k

C

D2

1N5818

R

C

OUTPUT**

–5V, 0.5A

* INCREASE L1 TO 30µH OR 60µH FOR HIGHER CURRENT APPLICATIONS.

SEE APPLICATIONS INFORMATION

** MAXIMUM LOAD CURRENT DEPENDS ON MINIMUM INPUT VOLTAGE

AND INDUCTOR SIZE. SEE APPLICATIONS INFORMATION

1576 F15

Figure 15. Positive-to-Negative Converter

A possible exception to the “clean response” rule is at very

light loads, as evidenced in Figure 14 with I_LOAD= 50mA.

Switching regulators tend to have dramatic shifts in loop

response at very light loads, mostly because the inductor

currentbecomesdiscontinuous.Onecommonresultisvery

slow but stable characteristics. A second possibility is low

phase margin, as evidenced by ringing at the output with

transients. The good news is that the low phase margin at

lightloadsisnotparticularlysensitivetocomponentvaria-

tion, so if it looks reasonable under a transient test, it will

Inverting regulators differ from buck regulators in the

basicswitchingnetwork. Currentisdeliveredtotheoutput

as square waves with a peak-to-peak amplitude much

greater than load current. This means that maximum load

current will be significantly less than the LT1576’s 1.5A

maximumswitchcurrent, evenwithlargeinductorvalues.

The buck converter in comparison, delivers current to the

output as a triangular wave superimposed on a DC level

equal to load current, and load current can approach 1.5A

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withlargeinductors.Outputripplevoltageforthepositive-

to-negative converter will be much higher than a buck

converter. Ripple current in the output capacitor will also

be much higher. The following equations can be used to

calculateoperatingconditionsforthepositive-to-negative

converter.

This duty cycle is close enough to 50% that I_Pcan be

assumed to be 1.5A.

OUTPUT DIVIDER

If the adjustable part is used, the resistor connected to

V_OUT(R2) should be set to approximately 5k. R1 is

calculated from:

Maximum load current:

V

(

)(

)

IN OUT

R2 V

− 1.21

(

)

OUT

I −

V

V −

0.35

(

)(

)

P

OUT IN

R1=

2 V

+ V f L

(

)( )( )

OUT

IN

1.21

I

=

MAX

V

+V − 0.35 V

+ V

F

(

)(

)

OUT

IN

OUT

INDUCTOR VALUE

I_P= Maximum rated switch current

V_IN= Minimum input voltage

V_OUT= Output voltage

V_F= Catch diode forward voltage

0.35 = Switch voltage drop at 1.5A

Unlike buck converters, positive-to-negative converters

cannot use large inductor values to reduce output ripple

voltage. At 200kHz, values larger than 75µH make almost

no change in output ripple. The graph in Figure 16 shows

peak-to-peak output ripple voltage for a 5V to –5V con-

verter versus inductor value. The criteria for choosing the

Example: with V_IN(MIN)= 5.5V, V_OUT= 5V, L = 30µH,

V_F= 0.5V, I_P= 1.5A: I_MAX= 0.6A. Note that this equation

does not take into account that maximum rated switch

current (I_P) on the LT1576 is reduced slightly for duty

cyclesabove50%. Ifdutycycleisexpectedtoexceed50%

(input voltage less than output voltage), use the actual I_P

value from the Electrical Characteristics table.

150

5V TO –5V CONVERTER

OUTPUT CAPACITOR’S

ESR = 0.1Ω

120

DISCONTINUOUS

I

= 0.1A

90

60

30

0

LOAD

DISCONTINUOUS

= 0.25A

I

Operating duty cycle:

LOAD

V

_OUT+ V_F

V − 0.3 + V_OUT+ V_F

DC =

CONTINUOUS

IN

I

> 0.38A

LOAD

(This formula uses an average value for switch loss, so it

may be several percent in error.)

0

15

30

45

60

75

INDUCTOR SIZE (µH)

1576 F16

With the conditions above:

Figure 16. Ripple Voltage on Positive-to-Negative Converter

5 + 0.5

5.5 − 0.3 + 5 + 0.5

DC =

= 51%

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inductor is therefore typically based on ensuring that peak

switch current rating is not exceeded. This gives the

lowest value of inductance that can be used, but in some

cases (lower output load currents) it may give a value that

creates unnecessarily high output ripple voltage. A com-

promise value is often chosen that reduces output ripple.

As you can see from the graph, large inductors will not

give arbitrarily low ripple, but small inductors can give

high ripple.

2

)

5.5 1.5

(

) (

I

=

= 0.38A

CONT

4 5.5 + 5 5.5 +5 +0.5

(

)(

)

This says that discontinuous mode can be used and the

minimum inductor needed is found from:

2 5 0.25

( )(

)

L

=

= 5.6µH

MIN

The difficulty in calculating the minimum inductor size

needed is that you must first know whether the switcher

will be in continuous or discontinuous mode at the critical

point where switch current is 1.5A. The first step is to use

the following formula to calculate the load current where

the switcher must use continuous mode. If your load

current is less than this, use the discontinuous mode

formula to calculate minimum inductor needed. If load

current is higher, use the continuous mode formula.

2

)

3

200• 10 1.5

(

Inpractice,theinductorshouldbeincreasedbyabout30%

over the calculated minimum to handle losses and varia-

tionsinvalue. Thissuggestsaminimuminductorof7.3µH

for this application, but looking at the ripple voltage chart

showsthatoutputripplevoltagecouldbereducedbyafac-

toroftwobyusinga30µHinductor.Thereisnoruleofthumb

heretomakeafinaldecision.Ifmodestrippleisneededand

the larger inductor does the trick, go for it. If ripple is non-

critical use the smaller inductor. If ripple is extremely criti-

cal, a second filter may have to be added in any case, and

the lower value of inductance can be used. Keep in mind

thattheoutputcapacitoristheothercriticalfactorindeter-

mining output ripple voltage. Ripple shown on the graph

(Figure 16) is with a capacitor’s ESR of 0.1Ω. This is rea-

sonableforAVXtypeTPS“D”or“E”sizesurfacemountsolid

tantalumcapacitors,butthefinalcapacitorchosenmustbe

looked at carefully for ESR characteristics.

Output current where continuous mode is needed:

2

V

²I

(

_IN) ( _P)

I_CONT

=

4 V + V

V + V + V

IN OUT F

(

_OUT)(

)

IN

Minimum inductor discontinuous mode:

2 V

I

(

_OUT)( _OUT

f I

)

L_MIN

=

2

( )( _P)

Minimum inductor continuous mode:

V

OUT

( )(

)

IN

L_MIN

=

V

+ V_F

(

)

OUT

2 f V + V

I −I

1+

( )(

)

IN

OUT

P

OUT

V

IN

For the example above, with maximum load current of

0.25A:

25

LT1576/LT1576-5

U

W U U

APPLICATIONS INFORMATION

Ripple Current in the Input and Output Capacitors

Diode Current

Positive-to-negativeconvertershavehighripplecurrentin

both the input and output capacitors. For long capacitor

lifetime, the RMS value of this current must be less than

the high frequency ripple current rating of the capacitor.

The following formula will give an approximate value for

RMS ripple current. This formula assumes continuous

mode and large inductor value. Small inductors will give

somewhat higher ripple current, especially in discontinu-

ous mode. The exact formulas are very complex and

appear in Application Note 44, pages 30 and 31. For our

purposes here I have simply added a fudge factor (ff). The

value for ff is about 1.2 for higher load currents and

L ≥10µH. It increases to about 2.0 for smaller inductors at

lower load currents.

Average diodecurrentisequaltoloadcurrent. Peak diode

current will be considerably higher.

Peak diode current:

Continuous Mode =

V + V

V

(

)

(

)(

)

IN

OUT

IN OUT

I

+

OUT

V

2 L f V + V

( )( )(

IN

)

IN

OUT

2 I

(

V

OUT

)(

)

OUT

Discontinuous Mode =

L f

( )( )

Keep in mind that during start-up and output overloads,

average diode current may be much higher than with

normalloads.Careshouldbeusedifdiodesratedlessthan

1A are used, especially if continuous overload conditions

must be tolerated.

V_OUT

Capacitor I_RMS= ff I

( )( _OUT

)

V

IN

ff = Fudge factor (1.2 to 2.0)

26

LT1576/LT1576-5

U

PACKAGE DESCRIPTION ^{Dimensions in inches (millimeters) unless otherwise noted.}

S8 Package

8-Lead Plastic Small Outline (Narrow 0.150)

(LTC DWG # 05-08-1610)

0.189 – 0.197*

(4.801 – 5.004)

7

5

8

6

0.150 – 0.157**

(3.810 – 3.988)

0.228 – 0.244

(5.791 – 6.197)

1

3

4

2

0.010 – 0.020

(0.254 – 0.508)

× 45°

0.053 – 0.069

(1.346 – 1.752)

0.004 – 0.010

(0.101 – 0.254)

0.008 – 0.010

(0.203 – 0.254)

0°– 8° TYP

0.016 – 0.050

(0.406 – 1.270)

0.050

(1.270)

BSC

0.014 – 0.019

(0.355 – 0.483)

TYP

*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH

SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE

**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD

FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE

SO8 1298

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-

tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.

27

LT1576/LT1576-5

U

TYPICAL APPLICATION

Dual Output SEPIC Converter

losses. C4 provides a low impedance path to maintain an

equal voltage swing in L1B, improving regulation. In a

flybackconverter,duringswitchontime,alltheconverter’s

energyisstoredinL1Aonly, sincenocurrentflowsinL1B.

At switch off, energy is transferred by magnetic coupling

into L1B, powering the –5V rail. C4 pulls L1B positive

duringswitchontime, causingcurrenttoflow, andenergy

to build in L1B and C4. At switch off, the energy stored in

both L1B and C4 supply the –5V rail. This reduces the

current in L1A and changes L1B current waveform from

square to triangular. For details on this circuit see Design

Note 100.

The circuit in Figure 17 generates both positive and

negative 5V outputs with a single piece of magnetics. L1

is a 33µH surface mount inductor from Coiltronics. It is

manufactured with two identical windings that can be

connected in series or parallel. The topology for the 5V

output is a standard buck converter. The –5V topology

would be a simple flyback winding coupled to the buck

converter if C4 were not present. C4 creates the SEPIC

(Single-Ended Primary Inductance Converter) topology

which improves regulation and reduces ripple current in

L1. Without C4, the voltage swing on L1B compared to

L1A would vary due to relative loading and coupling

C2

0.33µF

D2

1N914

L1A*

33µH

BOOST

INPUT

OUTPUT

5V

V

IN

V

SW

6V TO 25V

LT1576

BIAS

FB

R1

15.8k

+

R2

4.99k

SHDN

GND

V

C

C1**

100µF

D1

1N5818

10V TANT

+

C3

C

22µF

100pF

35V TANT

GND

+

C5**

C4**

100µF

* L1 IS A SINGLE CORE WITH TWO WINDINGS

COILTRONICS CTX33-2

100µF

L1B*

D3

1N5818

10V TANT

** AVX TSPD107M010

†

OUTPUT

IF LOAD CAN GO TO ZERO, AN OPTIONAL

†

–5V

PRELOAD OF 1k TO 5k MAY BE USED TO

IMPROVE LOAD REGULATION

1576 F17

Figure 17. Dual Output SEPIC Converter

RELATED PARTS

PART NUMBER

LT1074/LT1076

LTC^®1148

DESCRIPTION

COMMENTS

Step-Down Switching Regulators

40V Input, 100kHz, 5A and 2A

External FET Switches

High Efficiency Synchronous Step-Down Switching Regulator

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LTC1149

External FET Switches

LTC1174

0.5A, 150kHz Burst Mode^TMOperation

42V, 6A, 500kHz Switch

35V, 3A, 500kHz Switch

Boost Topology

LT1370

LT1371

High Efficiency DC/DC Converter

LT1372/LT1377

LT1374

500kHz and 1MHz High Efficiency 1.5A Switching Regulators

4.5A, 500kHz Step-Down Switching Regulator

1.5A, 500kHz Step-Down Switching Regulator

High Efficiency Step-Down Converter

LT1376

LT1435/LT1436

LT1676/LT1776

LT1777

External Switches, Low Noise

High Efficiency Step-Down Switching Regulators

Low Noise Step-Down Switching Regulator

7.4V to 60V Input, 100kHz/200kHz

48V Input, Internally Limited dv/dt

Burst Mode is a trademark of Linear Technology Corporation.

1576f LT/TP 0999 4K • PRINTED IN USA

LinearTechnology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

28

●

(408)432-1900 FAX:(408)434-0507 www.linear-tech.com

LINEAR TECHNOLOGY CORPORATION 1999


型号：	LT1576IS8
厂家：	Linear
描述：	1.5A, 200kHz Step-Down Switching Regulator 1.5A ， 200kHz的降压型开关稳压器稳压器开关
文件：	总28页 (文件大小：293K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

LT1576IS8 [Linear]

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