LT3436EFE#TR_技术文档

LT3436

3A, 800kHz Step-Up

Switching Regulator

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DESCRIPTIO

FEATURES

The LT^®3436 is an 800kHz monolithic boost switching

regulator. A high efficiency 3A, 0.1Ω switch is included on

the die together with all the control circuitry required to

complete a high frequency, current-mode switching regu-

lator. Current-mode control provides fast transient re-

sponse and excellent loop stability.

■

Constant 800kHz Switching Frequency

■

Wide Operating Voltage Range: 3V to 25V

■

High Efficiency 0.1Ω/3A Switch

■

1.2V Feedback Reference Voltage

■

±2% Overall Output Voltage Tolerance

■

Uses Low Profile Surface Mount External

Components

New design techniques achieve high efficiency at high

switching frequencies over a wide operating range. A low

dropout internal regulator maintains consistent perfor-

manceoverawiderangeofinputsfrom24VsystemstoLi-

Ion batteries. An operating supply current of 1mA main-

tains high efficiency, especially at lower output currents.

Shutdown reduces quiescent current to 11µA. Maximum

switch current remains constant at all duty cycles. Syn-

chronizationcapabilityallowsanexternallogiclevelsignal

to increase the internal oscillator from 1MHz to 1.4MHz.

■

Low Shutdown Current: 11µA

■

Synchronizable from 1MHz to 1.4MHz

■

Current-Mode Control

■

Constant Maximum Switch Current Rating

at All Duty Cycles*

■

Available in a Small Thermally Enhanced

TSSOP-16 Package

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APPLICATIO S

■

DSL Modems

Full cycle-by-cycle switch current limit protection and ther-

mal shutdown are provided. High frequency operation al-

lows the reduction of input and output filtering components

and permits the use of tiny chip inductors. The LT3436 is

available in an exposed pad, 16-pin TSSOP package.

■

Portable Computers

■

Battery-Powered Systems

Distributed Power

■

, LTC and LT are registered trademarks of Linear Technology Corporation.

All other trademarks are the property of their respective owners.

*Protectd by U.S. Patents including 6535042, 6611131, 6498466

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TYPICAL APPLICATIO

Efficiency vs Load Current

5V to 12V Boost Converter

90

V

= 5V

IN

OUT

3.9µH

= 12V

85

80

75

70

65

60

B220A

OUTPUT

12V

INPUT

5V

4.7µF

CERAMIC

V

IN

SW

†

0.9A

OPEN

OR

HIGH

= ON

LT3436

90.9k

SHDN

SYNC GND

FB

V

C

22µF

CERAMIC

10k

1%

10nF

4.7k

MAXIMUM OUTPUT CURRENT IS SUBJECT TO THERMAL DERATING.

470pF

†

0

0.4

0.6 0.7

0.1 0.2 0.3

0.5

0.8

3436 TA01

LOAD CURRENT (A)

3436 TA01b

3436fa

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LT3436

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ABSOLUTE MAXIMUM RATINGS

PACKAGE/ORDER INFORMATION

(Note 1)

TOP VIEW

ORDER PART NUMBER

Input Voltage .......................................................... 25V

Switch Voltage......................................................... 35V

SHDN Pin ............................................................... 25V

FB Pin Current ....................................................... 1mA

SYNC Pin Current .................................................. 1mA

Operating Junction Temperature Range (Note 2)

LT3436E .......................................... –40°C to 125°C

Storage Temperature Range ................ –65°C to 150°C

Lead Temperature (Soldering, 10 sec)................. 300°C

GND

1

2

3

4

5

6

7

8

16 GND

15 NC

V

IN

LT3436EFE

SW

14 SYNC

13

V

C

17

GND

NC

12 FB

11 SHDN

10 NC

FE PART MARKING

3436EFE

GND

9

GND

FE PACKAGE

16-LEAD PLASTIC TSSOP

EXPOSED PAD IS GND (PIN 17),

MUST BE SOLDERED TO PCB

T_JMAX= 125°C, θ_JA= 45°C/W,

θ

_JC(PAD)= 10°C/W

Consult LTC Marketing for parts specified with wider operating temperature ranges.

ELECTRICAL CHARACTERISTICS ^The●denotesthespecificationswhichapplyoverthefulloperatingtemperature

range, otherwise specifications are at T_A= 25°C. V_IN= 15V, V_C= 0.8V, SHDN, SYNC and switch open unless otherwise noted.

PARAMETER

CONDITION

MIN

3

TYP

MAX

25

UNITS

V

Recommended Operating Voltage

Maximum Switch Current Limit

Oscillator Frequency

●

3

4

6

A

3.3V < V < 25V

640

800

330

2.6

1

960

550

2.73

1.3

kHz

mV

V

IN

Switch On Voltage Drop

I

= 3A

SW

V

Undervoltage Lockout

Supply Current

(Note 3)

2.47

IN

I

= 0A

= 3A

mA

mA/A

SW

Supply Current/I

15

SW

Shutdown Supply Current

V

= 0V, V = 25V, V = 25V

11

25

45

µA

SHDN

IN

SW

●

Feedback Voltage

3V < V < 25V, 0.4V < V < 0.9V

1.182

1.176

1.2

1.218

1.224

V

IN

C

●

FB Input Current

0

–0.2

350

850

–120

110

4.8

–0.4

µA

FB to V Voltage Gain

0.4V < V < 0.9V

150

500

–85

70

C

FB to V Transconductance

∆I = ±10µA

VC

●

1300

–165

165

µMho

µA

A/V

V

C

V Pin Source Current

C

V

= 1V

FB

V Pin Sink Current

C

= 1.4V

V Pin to Switch Current Transconductance

C

V Pin Minimum Switching Threshold

C

Duty Cycle = 0%

0.3

V Pin 3A I Threshold

0.9

V

C

SW

Maximum Switch Duty Cycle

V = 1.2V, I = 350mA

85

80

90

%

C

SW

V = 1.2V, I = 1A

●

C

SW

SHDN Threshold Voltage

1.28

–7

4

1.35

–10

7

1.42

–13

10

V

µA

SHDN Input Current (Shutting Down)

SHDN Threshold Current Hysteresis

SYNC Threshold Voltage

SHDN = 60mV Above Threshold

SHDN = 100mV Below Threshold

µA

1.5

2.2

1.4

V

SYNC Input Frequency

1

MHz

SYNC Pin Resistance

I

= 1mA

20

kΩ

SYNC

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LT3436

ELECTRICAL CHARACTERISTICS

Note 1: Absolute Maximum Ratings are those values beyond which the life

Note 3: Minimum input voltage is defined as the voltage where the internal

regulator enters lockout. Actual minimum input voltage to maintain a

regulated output will depend on output voltage and load current. See

Applications Information.

of a device may be impaired.

Note 2: The LT3436E is guaranteed to meet performance specifications

from 0°C to 125°C. Specifications over the –40°C to 125°C operating

junction temperature range are assured by design, characterization and

correlation with statistical process controls.

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TYPICAL PERFORMANCE CHARACTERISTICS

FB Voltage

Switch On Voltage Drop

Oscillator Frequency

500

450

400

350

300

250

200

150

100

50

1.220

1.215

1.210

1.205

1.200

1.195

1.190

1.185

1.180

920

890

860

830

800

770

740

710

680

T

= 125°C

A

T

A

= 25°C

T

A

= –40°C

0

–25

0

25

50

75

125

0

0.5

1.5

2.0

2.5

3.0

–25

0

25

50

75

125

–50

100

–50

100

1.0

SWITCH CURRENT (A)

TEMPERATURE (°C)

3436 G01

3436 G02

3436 G03

SHDN Threshold

SHDN Supply Current

SHDN Input Current

–12

–10

–8

–6

–4

–2

0

1.40

1.38

1.36

1.34

1.32

1.30

14

12

10

8

T

= 25°C

A

SHDN = 0V

SHUTTING DOWN

6

STARTING UP

4

2

0

50

TEMPERATURE (°C)

100 125

–50 –25

0

25

75

–25

0

25

50

75

125

–50

100

0

10

15

20

25

30

5

INPUT VOLTAGE (V)

TEMPERATURE (°C)

3436 G06

3436 G05

3436 G04

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LT3436

TYPICAL PERFOR A CE CHARACTERISTICS

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Current Limit Foldback

SHDN Supply Current

Input Supply Current

300

250

200

150

100

50

1200

1000

800

600

400

200

0

4.0

3.5

3.0

2.5

2.0

1.5

1.0

0.5

0

40

30

20

10

0

T

= 25°C

IN

T = 25°C

A

T

A

= 25°C

V

= 15V

SWITCH CURRENT

MINIMUM

INPUT

VOLTAGE

0

0.8

1.2 1.4

0

10

5

15

20

25

30

0

0.2

0.4 0.6

1.0

1.2

1.0

0

0.2

0.4

0.6

0.8

INPUT VOLTAGE (V)

SHDN VOLTAGE (V)

FEEDBACK VOLTAGE (V)

3436 G07

3436 G08

3436 G09

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PIN FUNCTIONS

GND (Pins 1, 5, 6, 8, 9, 16, 17): Short GND pins 1, 5, 6,8,

9, 16 and the exposed pad (pin 17) on the PCB. The GND

isthereferencefortheregulatedoutput,soloadregulation

will suffer if the “ground” end of the load is not at the same

voltage as the GND of the IC. This condition occurs when

the load current flows through the metal path between the

GNDpinsandtheloadgroundpoint. Keepthegroundpath

shortbetweentheGNDpinsandtheloadanduseaground

plane when possible. Keep the path between the input

bypass and the GND pins short. The exposed pad should

be attached to a large copper area to improve thermal

performance.

a predetermined level. Float or pull high to put the regula-

tor in the operating mode.

FB (Pin 12): The feedback pin is used to set output voltage

usinganexternalvoltagedividerthatgenerates1.2Vatthe

pinwiththedesiredoutputvoltage. Ifrequired, thecurrent

limit can be reduced during start up when the FB pin is

below 0.5V (see the Current Limit Foldback graph in the

Typical Performance Characteristics section). An imped-

ance of less than 5kΩ at the FB pin is needed for this

feature to operate.

V_C(Pin 13): The V_Cpin is the output of the error amplifier

and the input of the peak switch current comparator. It is

normally used for frequency compensation, but can do

double duty as a current clamp or control loop override.

This pin sits at about 0.3V for very light loads and 0.9V at

maximum load.

V_IN(Pin 2): This pin powers the internal circuitry and

internal regulator. Keep the external bypass capacitor

close to this pin.

SW (Pins 3, 4): The switch pin is the collector of the on-

chip power NPN switch and has large currents flowing

throughit.Keepthetracestotheswitchingcomponentsas

shortaspossibletominimizeradiationandvoltagespikes.

SYNC (Pin 14): The sync pin is used to synchronize the

internal oscillator to an external signal. It is directly logic

compatible and can be driven with any signal between

20% and 80% duty cycle. The synchronizing range is

equal to initial operating frequency, up to 1.4MHz. See

Synchronization section in Applications Information for

details. When not in use, this pin should be grounded.

SHDN (Pin 11): The shutdown pin is used to turn off the

regulator and to reduce input drain current to a few

microamperes. The 1.35V threshold can function as an

accurate undervoltage lockout (UVLO), preventing the

regulatorfromoperatinguntiltheinputvoltagehasreached

3436fa

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LT3436

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BLOCK DIAGRAM

The LT3436 is a constant frequency, current-mode boost

converter. This means that there is an internal clock and

twofeedbackloopsthatcontrolthedutycycleofthepower

switch. In addition to the normal error amplifier, there is a

current sense amplifier that monitors switch current on a

cycle-by-cycle basis. A switch cycle starts with an oscilla-

tor pulse which sets the R_Sflip-flop to turn the switch on.

When switch current reaches a level set by the inverting

input of the comparator, the flip-flop is reset and the

switch turns off. Output voltage control is obtained by

using the output of the error amplifier to set the switch

current trip point. This technique means that the error

amplifier commands current to be delivered to the output

rather than voltage. A voltage fed system will have low

phase shift up to the resonant frequency of the inductor

and output capacitor, then an abrupt 180° shift will occur.

The current fed system will have 90° phase shift at a much

lower frequency, but will not have the additional 90° shift

until well beyond the LC resonant frequency. This makes

itmucheasiertofrequencycompensatethefeedbackloop

and also gives much quicker transient response.

A comparator connected to the shutdown pin disables the

internal regulator, reducing supply current.

INPUT

INTERNAL

CC

2.5V BIAS

REGULATOR

V

SLOPE COMP

Σ

0.3V

SW

800kHz

S

SYNC

Q1

POWER

SWITCH

OSCILLATOR

R

DRIVER

CURRENT

COMPARATOR

S

FLIP-FLOP

CIRCUITRY

R

+

–

CURRENT SENSE

SHUTDOWN

COMPARATOR

AMPLIFIER VOLTAGE

GAIN = 40

7µA

–

+

–

0.005Ω

1.35V

–

SHDN

FB

3µA

+

ERROR

V

C

AMPLIFIER

1.2V

g

= 850µMho

m

GND

3436 F01

Figure 1. Block Diagram

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LT3436

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APPLICATIONS INFORMATION

FB RESISTOR NETWORK

to 22µF range. Since the absolute value of capacitance

defines the pole frequency of the output stage, an X7R or

X5R type ceramic, which have good temperature stability,

is recommended.

The suggested resistance (R2) from FB to ground is 10k

1%. This reduces the contribution of FB input bias current

to output voltage to less than 0.2%. The formula for the

resistor (R1) from V_OUTto FB is:

Tantalum capacitors are usually chosen for their bulk

capacitance properties, useful in high transient load appli-

cations. ESR rather than absolute value defines output

ripple at 800kHz. Values in the 22µF to 100µF range are

generally needed to minimize ESR and meet ripple current

ratings. Care should be taken to ensure the ripple ratings

are not exceeded.

R2 V

1.2 − R2(0.2µA)

−1.2

(

)

OUT

R1=

V

LT3436

SW

Table 1. Surface Mount Solid Tantalum Capacitor ESR and

Ripple Current

OUTPUT

ERROR

AMPLIFIER

E Case Size

ESR (Max,

Ω

)

Ripple Current (A)

1.2V

+

–

AVX TPS, Sprague 593D

D Case Size

0.1 to 0.3

0.2 (typ)

0.7 to 1.1

R1

+

FB

AVX TPS, Sprague 593D

C Case Size

0.7 to 1.1

0.5 (typ)

R2

10k

3436 F02

AVX TPS

V

C

GND

Figure 2. Feedback Network

INPUT CAPACITOR

Unlike the output capacitor, RMS ripple current in the

input capacitor is normally low enough that ripple current

rating is not an issue. The current waveform is triangular,

with an RMS value given by:

OUTPUT CAPACITOR

Step-up regulators supply current to the output in pulses.

The rise and fall times of these pulses are very fast. The

output capacitor is required to reduce the voltage ripple

this causes. The RMS ripple current can be calculated

from:

0.29 V

_IN)(

=

V

− V

(

)

OUT IN

I_{RIPPLE RMS}

(

)

L f V

( )( )(

)

OUT

I_{RIPPLE RMS}= I_OUT

V

OUT

− V

/

V

IN

(

)

IN

Athigherswitchingfrequency,theenergystoragerequire-

ment of the input capacitor is reduced so values in the

range of 2.2µF to 10µF are suitable for most applications.

Y5V or similar type ceramics can be used since the

absolute value of capacitance is less important and has no

significant effect on loop stability. If operation is required

close to the minimum input voltage required by either the

output or the LT3436, a larger value may be necessary.

This is to prevent excessive ripple causing dips below the

minimum operating voltage resulting in erratic operation.

(

)

The LT3436 will operate with both ceramic and tantalum

output capacitors. Ceramic capacitors are generally cho-

sen for their small size, very low ESR (effective series

resistance), and good high frequency operation, reducing

outputripplevoltage. TheirlowESRremovesausefulzero

in the loop frequency response, common to tantalum

capacitors. To compensate for this, the V_Cloop compen-

sationpolefrequencymusttypicallybereducedbyafactor

of 10. Typical ceramic output capacitors are in the 4.7µF

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LT3436

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APPLICATIONS INFORMATION

The recommended minimum inductance is:

(V )²(V_OUT– V )

INDUCTOR CHOICE AND MAXIMUM OUTPUT

CURRENT

IN

L_MIN

=

When choosing an inductor, there are 2 conditions that

limit the minimum inductance; required output current,

and avoidance of subharmonic oscillation. The maximum

output current for the LT3436 in a standard boost con-

verter configuration with an infinitely large inductor is:

0.4(V_OUT)²(I_OUT)(f)

The inductor value may need further adjustment for other

factors such as output voltage ripple and filtering require-

ments. Remember also, inductance can drop significantly

with DC current and manufacturing tolerance.

V • η

V_OUT

IN

I_OUT(MAX)= 3A

The inductor must have a rating greater than its peak

operating current to prevent saturation resulting in effi-

ciency loss. Peak inductor current is given by:

Where η = converter efficiency (typically 0.87 at high

current).

(V_OUT)(I_OUT) V (V_OUT− V )

IN

I_LPEAK

=

+

As the value of inductance is reduced, ripple current

increases and I_OUT(MAX)is reduced. The minimum induc-

tance for a required output current is given by:

V • η

IN

2V_OUT(L)(f)

Also, consideration should be given to the DC resistance

of the inductor. Inductor resistance contributes directly to

the efficiency losses in the overall converter.

V (V_OUT– V )

IN

L_MIN

=

⎛

⎞

SuitableinductorsareavailablefromCoilcraft,Coiltronics,

Dale, Sumida, Toko, Murata, Panasonic and other manu-

factures.

(V_OUT)(I_OUT

)

2V_OUT(f) 3 –

⎜

⎟

V • η

IN

⎝

⎠

The second condition, avoidance of subharmonic oscilla-

tion, mustbemetiftheoperatingdutycycleisgreaterthan

50%. The slope compensation circuit within the LT3436

prevents subharmonic oscillation for inductor ripple cur-

rents of up to 1.4A_P-P, defining the minimum inductor

value to be:

Table 2

PART

NUMBER

VALUE

(µH)

I

DCR

(Ω)

HEIGHT

(mm)

SAT(DC)

(Amps)

Coilcraft

DO1608C-222

Sumida

2.2

2.4

0.07

2.9

CDRH3D16-1R5

CDRH4D18-1R0

CDC5D23-2R2

CR43-1R4

1.5

1.0

2.2

1.4

2.6

3.3

3.0

1.6

1.7

2.2

2.5

2.6

3.5

3.0

0.043

0.035

0.03

1.8

2.0

2.5

3.5

3.0

4.0

3.0

V (V_OUT– V )

1.4V_OUT(f)

IN

L_MIN

=

0.056

0.013

0.02

These conditions define the absolute minimum induc-

tance. However, it is generally recommended that to

prevent excessive output noise, and difficulty in obtaining

stability, the ripple current is no more than 40% of the

average inductor current. Since inductor ripple is:

CDRH5D28-2R6

CDRH6D38-3R3

CDRH6D28-3R0

Toko

0.024

(D62F)847FY-2R4M

(D73LF)817FY-2R2M

2.4

2.2

2.5

2.7

0.037

0.03

2.7

3.0

V (V_OUT– V )

IN

I_{P−P RIPPLE}

=

V_OUT(L)(f)

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LT3436

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APPLICATIONS INFORMATION

CATCH DIODE

shutdown pin can be used. The threshold voltage of the

shutdown pin comparator is 1.35V. A 3µA internal current

sourcedefaultstheopenpinconditiontobeoperating(see

TypicalPerformanceGraphs). Currenthysteresisisadded

above the SHDN threshold. This can be used to set voltage

hysteresis of the UVLO using the following:

The suggested catch diode (D1) is a B220A Schottky. It is

rated at 2A average forward current and 20V reverse

voltage. Typical forward voltage is 0.5V at 2A. The diode

conductscurrentonlyduringswitchofftime.Peakreverse

voltage is equal to regulator output voltage. Average

forward current in normal operation is equal to output

current.

V_H− V_L

R1=

7µA

1.35V

SHUTDOWN AND UNDERVOLTAGE LOCKOUT

R2 =

V − 1.35V

(

)

+ 3µA

H

Figure 4 shows how to add undervoltage lockout (UVLO)

to the LT3436. Typically, UVLO is used in situations where

the input supply is current limited, or has a relatively high

source resistance. A switching regulator draws constant

power from the source, so source current increases as

source voltage drops. This looks like a negative resistance

loadtothesourceandcancausethesourcetocurrentlimit

or latch low under low source voltage conditions. UVLO

prevents the regulator from operating at source voltages

where these problems might occur.

R1

V_H– Turn-on threshold

V_L– Turn-off threshold

Example: switching should not start until the input is

above 4.75V and is to stop if the input falls below 3.75V.

V_H= 4.75V

V_L= 3.75V

4.75V − 3.75V

LT3436

R1=

= 143k

7µA

1.35V

7µA

IN

INPUT

1.35V

R1

R2

3µA

R2 =

= 50.4k

V

CC

4.75V − 1.35V

(

)

+ 3µA

SHDN

143k

C1

GND

3436 F04

Keep the connections from the resistors to the SHDN pin

short and make sure that the interplane or surface capaci-

tance to the switching nodes are minimized. If high resis-

torvaluesareused,theSHDNpinshouldbebypassedwith

a 1nF capacitor to prevent coupling problems from the

switch node.

Figure 4. Undervoltage Lockout

An internal comparator will force the part into shutdown

below the minimum V_INof 2.6V. This feature can be used

to prevent excessive discharge of battery-operated sys-

tems. If an adjustable UVLO threshold is required, the

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LT3436

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APPLICATIONS INFORMATION

SYNCHRONIZATION

high speed switching current path, shown in Figure 5,

must be kept as short as possible. This is implemented in

the suggested layout of Figure 6. Shortening this path will

also reduce the parasitic trace inductance of approxi-

mately 25nH/inch. At switch off, this parasitic inductance

produces a flyback spike across the LT3436 switch. When

operating at higher currents and output voltages, with

poor layout, this spike can generate voltages across the

LT3436 that may exceed its absolute maximum rating. A

ground plane should always be used under the switcher

circuitry to prevent interplane coupling and overall noise.

The SYNC pin, is used to synchronize the internal oscilla-

tor to an external signal. The SYNC input must pass from

a logic level low, through the maximum synchronization

threshold with a duty cycle between 20% and 80%. The

input can be driven directly from a logic level output. The

synchronizing range is equal to initial operating frequency

up to 1.4MHz. This means that minimum practical sync

frequency is equal to the worst-case high self-oscillating

frequency(960kHz),notthetypicaloperatingfrequencyof

800kHz. Caution should be used when synchronizing

above 1.1MHz because at higher sync frequencies the

amplitude of the internal slope compensation used to

prevent subharmonic switching is reduced. Higher induc-

tor values will tend to eliminate this problem. See Fre-

quency Compensation section for a discussion of an

entirely different cause of subharmonic switching before

assuming that the cause is insufficient slope compensa-

tion. Application Note 19 has more details on the theory

of slope compensation.

The V_Cand FB components should be kept as far away as

possible from the switch node. The LT3436 pinout has

been designed to aid in this. The ground for these compo-

nents should be separated from the switch current path.

Failure to do so will result in poor stability or subharmonic

like oscillation.

Board layout also has a significant effect on thermal

resistance. The exposed pad is the copper plate that runs

undertheLT3436die.Thisisthebestthermalpathforheat

out of the package. Soldering the pad onto the board will

reduce die temperature and increase the power capability

of the LT3436. Provide as much copper area as possible

aroundthispad.Addingmultiplesolderfilledfeedthroughs

under and around the pad to the ground plane will also

help. Similar treatment to the catch diode and inductor

terminations will reduce any additional heating effects.

LAYOUT CONSIDERATIONS

As with all high frequency switchers, when considering

layout, care must be taken to achieve optimal electrical,

thermal and noise performance. For maximum efficiency,

switch rise and fall times are typically in the nanosecond

range. To prevent noise both radiated and conducted, the

L1

D1

C3

V

OUT

SW

LT3436

HIGH

FREQUENCY

SWITCHING

PATH

V

IN

C1 LOAD

GND

3436 F05

Figure 5. High Speed Switching Path

3436fa

9

LT3436

U

W U U

APPLICATIONS INFORMATION

L1

3.9µH

D1

B220A

OUTPUT

12V

INPUT

V

IN

SW

FB

5V

†

C3

0.8A

OPEN

OR

HIGH

= ON

LT3436

R1

90.9k

4.7µF

CERAMIC

SHDN

SYNC GND

V

C

C1

R2

10k

1%

C2

10nF

R3

4.7k

22µF

C4

470pF

CERAMIC

†

MAXIMUM OUTPUT CURRENT IS SUBJECT TO THERMAL DERATING.

INPUT

L1

GND

R3

C3

C2

C4

KEEP FB AND V

COMPONENTS

AWAY FROM

C

HIGH FREQUENCY,

HIGH INPUT

COMPONENTS

D1

U1

R2

R1

MINIMIZE

LT3436,

C1, D1 LOOP

GND

C1

V

OUT

PLACE FEEDTHROUGHS

AROUND GROUND PIN FOR

GOOD THERMAL CONDUCTIVITY

SOLDER EXPOSED

GROUND PAD

TO BOARD

KELVIN SENSE

V

OUT

Figure 6. Typical Application and Suggested Layout (Topside Only Shown)

3436fa

10

LT3436

U

W U U

APPLICATIONS INFORMATION

The inductor must have a rating greater than its peak

operating current to prevent saturation resulting in effi-

ciency loss. Peak inductor current is given by:

thermalresistancenumberandaddinworst-caseambient

temperature:

T_J= T_A+ θ_JA(P_TOT

)

If a true die temperature is required, a measurement of

the SYNC to GND pin resistance can be used. The SYNC

pin resistance across temperature must first be cali-

brated, with no device power, in an oven. The same

measurementcanthenbeusedinoperationtoindicatethe

die temperature.

(V_OUT)(I_OUT) V (V_OUT− V )

IN

I_LPEAK

=

+

V • η

IN

2V_OUT(L)(f)

Also, consideration should be given to the DC resistance

of the inductor. Inductor resistance contributes directly to

the efficiency losses in the overall converter.

FREQUENCY COMPENSATION

THERMAL CALCULATIONS

Loop frequency compensation is performed on the output

of the error amplifier (V_Cpin) with a series RC network.

The main pole is formed by the series capacitor and the

output impedance (≈500kΩ) of the error amplifier. The

pole falls in the range of 2Hz to 20Hz. The series resistor

creates a “zero” at 1kHz to 5kHz, which improves loop

stability and transient response. A second capacitor, typi-

cally one-tenth the size of the main compensation capaci-

tor, is sometimes used to reduce the switching frequency

ripple on the V_Cpin. V_Cpin ripple is caused by output

voltage ripple attenuated by the output divider and multi-

plied by the error amplifier. Without the second capacitor,

V_Cpin ripple is:

Power dissipation in the LT3436 chip comes from four

sources:switchDCloss,switchACloss,drivecurrent,and

inputquiescentcurrent.Thefollowingformulasshowhow

to calculate each of these losses. These formulas assume

continuous mode operation, so they should not be used

for calculating efficiency at light load currents.

(V_OUT− V )

IN

DC, duty cycle =

V_OUT

(V_OUT)(I_OUT

)

I_SW

=

V

IN

Switch loss:

P

= (DC)(I_SW)²(R_SW)+ 17n I

V

OUT

f

( )

(

)

(

)

SW

1.2(V

)(g )(R )

m C

RIPPLE

(V

V Pin Ripple =

C

V_INloss:

)

OUT

(V )(I_SW)(DC)

V

m

= Output ripple (V

)

IN

RIPPLE

P–P

P_VIN

=

+ 1mA(V )

IN

g = Error amplifier transconductance

50

(≈850µmho)

R_SW= Switch resistance (≈0.16Ω hot)

R = Series resistor on V pin

C

V

OUT

= DC output voltage

Example: V_IN= 5V, V_OUT= 12V and I_OUT= 0.8A

Total power dissipation = 0.34 + 0.31 + 0.11 + 0.005 =

0.77W

To prevent irregular switching, V_Cpin ripple should be

kept below 50mV_P–P. Worst-case V_Cpin ripple occurs at

maximum output load current and will also be increased if

poor quality (high ESR) output capacitors are used. The

addition of a 150pF capacitor on the V_Cpin reduces

switching frequency ripple to only a few millivolts. A low

value for R_Cwill also reduce V_Cpin ripple, but loop phase

margin may be inadequate.

Thermal resistance for LT3436 package is influenced by

the presence of internal or backside planes. With a full

plane under the package, thermal resistance will be about

40°C/W. Tocalculatedietemperature, usetheappropriate

3436fa

11

LT3436

TYPICAL APPLICATIO S

U

Load Disconnects in Shutdown

D3

1N4148

C6

0.1µF

L1

3.9µH

D2

1N4148

C5

0.1µF

D1

B220A

R4

1M

V

OUT

V

IN

5V

V

IN

V

SW

12V

Q1

Si2306DS

C3

4.7µF

0.8A

C7

22µF

C1

4.7µF

LT3436

R1

90.9k

SHDN

SYNC GND

FB

OFF ON

V

C

R2

10k

1%

C2

10nF

R3

4.7k

C4

470pF

LT3436 • TA02

3V to 20V_IN5V_OUTSEPIC with Either Two Inductors or a Transformer

D1

B220A

L1

CDRH6D28-100

V

OUT

IN

3V TO 20V

5V

+

C1

OPT

C6

C7

C5

R1

OPT

1µF, X5R, 25V

CERAMIC

OPT

31.6K

1%

V

SW

FB

IN

SHDN

SYNC

SHDN

LT3436

C3

10nF

C2

L2

SYNC

V

C

22µF

X5R

10V

CDRH6D28-100

GND GND

R2

10K

1%

CERAMIC

C1

C4

470pF

R3

2.2k

4.7µF

X5R

25V

CERAMIC

GND

OPTION: REPLACE L1, L2 WITH TRANSFORMER CTX5-1A, CTX8-1A, CTX10-2A

3436 TA02b

Maximum Load Current

Increases with Input Voltage

Efficiency

2.0

1.8

1.6

1.4

1.2

1.0

0.8

0.6

0.4

0.2

100

90

80

70

60

50

40

30

20

10

0

12V

IN

5V

IN

3.3V

IN

0

2

4

6

8

10 12 14 16 18 20

(V)

0

500

1.0k

1.5k

2.0k

V

LOAD CURRENT (mA)

IN

3436 TA02c

3436 TA02d

3436fa

12

LT3436

U

TYPICAL APPLICATIO S

4V-9V_INto 5V_OUTSEPIC Converter**

**

4V TO 9V

V

IN

L1A*

15µH

D1

B220A

V

IN

†

•

V

OUT

ON

V

SHDN

GND

SW

OFF

C1

5V

R2

31.6k

1%

C2

LT3436

4.7µF

+

FB

C

•

4.7µF

20V

+

C3

L1B*

15µH

47µF

10V

R3

10k

1%

R1

2.2k

C4

C5

470pF

15nF

LT3436 • TA03

†

MAX I

OUT

*

**

COILTRONICS CTX15-4

INPUT VOLTAGE MAY BE GREATER OR

LESS THAN OUTPUT VOLTAGE

I

V

IN

OUT

0.84A 4V

1.03A 5V

1.18A 6V

1.29A 7V

1.50A 9V

Boost Converter Drives Luxeon III 1A 3.6V White LED with 70% Efficiency

0.05Ω

1%

1A CONSTANT CURRENT

LXHL-PW09 EMITTER

V

IN

3.3V TO 4.2V

V

OUT

= V + V

IN

LED

49.9Ω

1%

UPS120

L1

V

IN

+

V

SW

FB

IN

LT1783

SHDN

LED ON

LT3436

–

Q2

SYNC

V

C

V

OUT

GND GND

4.7µF

X5R

Q1

78.7k

6.3V

22µF

X5R

10V

0.1µF

CERAMIC

1.21k

1%

CERAMIC

8.2k

4.99k

GND

3436 TA03b

Q1: MMBT2222A

Q2: FMMT3906

L1: CDRH6D28-3R0

3436fa

13

LT3436

U

TYPICAL APPLICATIO S

Single Li-Ion Cell to 5V

D1

B220A

L1

4.7µH

V

OUT

5V

R1

31.6k

1%

V

IN

ON

V

SHDN

GND

SW

OFF

LT3436

SINGLE

Li-Ion

CELL

+

C4

FB

C

C1

10µF

47µF

V

10V

R2

10k

1%

C2

3.3nF

C3

470pF

R3

1.5k

LT3436 • TA04

I

V

IN

OUT

1.5A 2.7V

1.86A 3.3V

2.0A 3.6V

SEPIC Converter Drives 5W LumiLEDs Luxeon V White LEDs at 70% Efficiency

D1

B130A

V

OUT

C

COUP

2.2µF, X5R, 25V

D2

L1

L2

CERAMIC

V

IN

3.6V TO 17V

V

IN

LED ON

700mA

+

V

SW

FB

IN

LT1783

SHDN

LT3436

–

R5

23.7k

SYNC

V

C

C1

4.7µF

GND GND

X5R

R7

124k

Q1

25V

C4

0.1µF

C2

CERAMIC

22µF

X5R

V

OUT

R4

1k

1%

R2

0.068Ω

1%

R6

4.99k

16V

CERAMIC

8.2k

GND

3436 TA04b

Q1: DIODES, INC. MMBT2222A

L1: CDRH6D28 10µH 1.7A

L2: CDRH4D28 10µH 1A

D2: LUMILEDS LXHL-PW03 EMITTER OR LXHL-LW6C STAR

3436fa

14

LT3436

U

PACKAGE DESCRIPTION

FE Package

16-Lead Plastic TSSOP (4.4mm)

(Reference LTC DWG # 05-08-1663)

Exposed Pad Variation BB

4.90 – 5.10*

(.193 – .201)

3.58

(.141)

3.58

(.141)

16 1514 13 12 1110

9

6.60 ±0.10

4.50 ±0.10

2.94

(.116)

6.40

(.252)

BSC

SEE NOTE 4

2.94

(.116)

0.45 ±0.05

1.05 ±0.10

0.65 BSC

5

7

8

1

2

3

4

6

RECOMMENDED SOLDER PAD LAYOUT

1.10

(.0433)

MAX

4.30 – 4.50*

(.169 – .177)

0.25

REF

0° – 8°

0.65

(.0256)

BSC

0.09 – 0.20

(.0035 – .0079)

0.50 – 0.75

(.020 – .030)

0.05 – 0.15

(.002 – .006)

0.195 – 0.30

FE16 (BB) TSSOP 0204

(.0077 – .0118)

TYP

NOTE:

1. CONTROLLING DIMENSION: MILLIMETERS 4. RECOMMENDED MINIMUM PCB METAL SIZE

FOR EXPOSED PAD ATTACHMENT

*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH

SHALL NOT EXCEED 0.150mm (.006") PER SIDE

MILLIMETERS

(INCHES)

2. DIMENSIONS ARE IN

3. DRAWING NOT TO SCALE

3436fa

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-

tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.

15

LT3436

U

TYPICAL APPLICATIO

High Voltage Laser Power Supply

0.01µF

5kV

1800pF

10kV

47k

5W

1800pF

10kV

8

11

HV DIODES

L1

3

1

5

4

LASER

2

+

2.2µF

0.47µF

Q1

Q2

150Ω

L2

10µH

MUR405

V

10k

SW

LT3436

GND

10k

V

IN

12V TO 25V

V

FB

IN

V

1N4002

(ALL)

+

V

L1 =

Q1, Q2 =

TBD

0.1µF

IN

190Ω

1%

2.2µF

ZETEX ZTX849

0.47µF =

WIMA 3X 0.15µF TYPE MKP-20

SEMTECH-FM-50

HUGHES 3121H-P

C

HV DIODES =

LASER =

+

10µF

LT3436 • TA05

COILTRONICS (407) 241-7876

RELATED PARTS

PART NUMBER

DESCRIPTION

COMMENTS

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V

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SD

LT1618

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Q

SW

IN

OUT(MAX)

I

= <1µA, MS Package

SD

LT1946/LT1946A

LT1961

1.5A (I ), 1.2MHz/2.7MHz, High Efficiency

Step-Up DC/DC Converter

V

= 2.45V to 16V, V

= <1µA, MS8 Package

= 34V, I = 3.2mA,

SW

IN

OUT(MAX) Q

I

SD

1.5A (I ), 1.25MHz, High Efficiency

90% Efficiency, V = 3V to 25V, V

= 35V, I = 0.9mA,

OUT(MAX) Q

SW

IN

Step-Up DC/DC Converter

I

= 6µA, MS8E Package

SD

LTC3400/LTC3400B

LTC3401

600mA (I ), 1.2MHz, Synchronous

Step-Up DC/DC Converter

92% Efficiency, V = 0.85V to 5V, V

I = 19µA/300µA, I = <1µA, ThinSOT Package

Q SD

= 5V,

SW

IN

OUT(MAX)

1A (I ), 3MHz, Synchronous

97% Efficiency, V = 0.5V to 5V, V

= 6V, I = 38µA,

Q

SW

IN

OUT(MAX)

Step-Up DC/DC Converter

I

= <1µA, MS Package

SD

LTC3402

2A (I ), 3MHz, Synchronous

Step-Up DC/DC Converter

97% Efficiency, V = 0.5V to 5V, V

= 6V, I = 38µA,

Q

SW

IN

OUT(MAX)

I

= <1µA, MS Package

SD

3436fa

LT/LWI/LT 0505 REV A • PRINTED IN USA

LinearTechnology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

16

●

(408) 432-1900 FAX: (408) 434-0507 www.linear.com


型号：	LT3436EFE#TR
厂家：	Linear
描述：	LT3436 - 3A, 800kHz Step-Up Switching Regulator; Package: TSSOP; Pins: 16; Temperature Range: -40°C to 85°C 开关光电二极管
文件：	总16页 (文件大小：173K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

LT3436EFE#TR [Linear]

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