LT3756-2 [Linear]
60V 4-Switch Synchronous; 60V 4开关同步
| 型号： | LT3756-2 |
| 厂家： | 
Linear |
| 描述： | 60V 4-Switch Synchronous |
| 文件： | 总28页 (文件大小：346K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LT3791
60V 4-Switch Synchronous
Buck-Boost LED Driver
Controller
FEATURES
DESCRIPTION
The LT®3791 is a synchronous 4-switch buck-boost LED
driver and voltage regulator controller. The controller
operates from input voltages above, below, or equal to
the output voltage. The LT3791 has a wide 4.7V to 60V
input and 0V to 60V output range along with seamless
transitions between operating modes. A ground reference
voltage FB pin serves as the input for several LED protec-
tion features and also makes it possible for the converter
to operate as a constant-voltage source. The LT3791 is
ideal for a wide variety of applications.
n
4-Switch Single Inductor Architecture Allows V
IN
Above, Below or Equal to V
OUT
n
n
n
n
n
n
n
Wide V Range: 4.7V to 60V
IN
Wide V
Range: 0V to 60V (55V LED)
OUT
±±% Output Voltage Accuracy
Synchronous Switching: Up to 98.5% Efficiency
6ꢀ LED Current Accuracy: 0V ≤ V < 60V
OUT
V
Disconnected from V During Shutdown
OUT
IN
Accurate Rail-to-Rail LED Current Sense with
Monitor Output
n
n
n
n
Input Current Sense with Monitor Output
PWM and Analog Dimming
Capable of 100W or greater per IC
38-Lead TSSOP with Exposed Pad
The LT3791 runs in forced continuous mode, which is
ideal for systems with stringent EMI requirements. Fault
protection is provided to survive and report an open or
shorted LED condition. A timer allows the LT3791 to
continue to run, latch off or restart when a fault occurs.
APPLICATIONS
L, LT, LTC, LTM, Linear Technology and the Linear logo are registered and True Color PWM
is a trademark of Linear Technology Corporation. All other trademarks are the property of their
respective owners.
n
Automotive Headlamps/Running Lamps
General Purpose Lighting
n
TYPICAL APPLICATION
98.5% Efficient 100W (33.3V 3A) Buck-Boost LED Driver
V
IN
15V TO 58V
2.2µF
100V
×5
0.003Ω
V
INTV
CC
IN
4.7µF
1µF
BST2
BST1
TG1
50Ω
Efficiency vs VIN
IVINN
IVINP
0.1µF
0.1µF
4.7µF
100
470nF
50V
1M
×5
499k
SWI
BG1
EN/UVLO
OVLO
98
96
94
92
90
10µH
BOOST
34.2k
BUCK
15.8k
28k
INTV
CC
BUCK-BOOST
LT3791
SNSP
200k 200k
3A, 100W
LED POWER
0.004Ω
0.033Ω
SHORTLED
OPENLED
SNSN
PGND
PWM
BG2
IVINMON
SW2
TG2
FB
ISMON
CLKOUT
10
20
30
40
50
60
V
REF
INPUT VOLTAGE (V)
ISP
ISN
CTRL
3791 TA01b
0.1µF
10nF
PWMOUT
SGND
SS SYNC VC
RT
86.6k
300kHz
2.2k
10nF
3791 TA01a
3791f
1
LT3791
ABSOLUTE MAXIMUM RATINGS
PIN CONFIGURATION
(Note 1)
TOP VIEW
Supply Voltages
1
2
OVLO
FB
38
37
36
35
34
33
32
31
30
29
28
27
26
CTRL
SS
Input Supply (V ).....................................................60V
IN
SW1, SW2......................................................–1V to 60V
OPENLED, SHORTLED ...............................................15V
EN/UVLO, IVINP, IVINN, ISP, ISN ..............................60V
3
VC
PWM
4
RT
OPENLED
SHORTLED
5
SYNC
CLKOUT
TEST2
PWMOUT
SGND
TEST1
SNSN
SNSP
ISN
INTV , (BST1-SW1), (BST2-SW2).............................6V
CC
6
V
REF
TEST2, SYNC, RT, CTRL, OVLO, PWM .......................6V
7
ISMON
IVINMON
EN/UVLO
IVINP
IVINMON, ISMON, FB, SS, VC, V ...........................6V
REF
8
IVINP-IVINN, ISP-ISN, SNSP-SNSN....................... 0.5V
SNSP, SNSN........................................................... 0.3V
Operating Junction Temperature (Notes 2, 3)
9
39
SGND
10
11
12
13
14
15
16
17
18
19
IVINN
LT3791E/LT3791I............................... –40°C to 125°C
LT3791H ............................................ –40°C to 150°C
LT3791MP ......................................... –55°C to 150°C
Storage Temperature Range .................. –65°C to 150°C
Lead Temperature (Soldering, 10 sec)...................300°C
V
IN
INTV
CC
25 ISP
TG1
BST1
SW1
PGND
BG1
TG2
24
23
22
21
20
NC
BST2
SW2
PGND
BG2
FE PACKAGE
38-LEAD PLASTIC TSSOP
T
= 150°C, θ = 28°C/W
JA
JMAX
EXPOSED PAD (PIN 39) IS SGND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH
LT3791EFE#PBF
LT3791IFE#PBF
LT3791HFE#PBF
LT3791MPFE#PBF
TAPE AND REEL
PART MARKING*
LT3791FE
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3791EFE#TRPBF
LT3791IFE#TRPBF
LT3791HFE#TRPBF
LT3791MPFE#TRPBF
38-Lead Plastic TSSOP
38-Lead Plastic TSSOP
38-Lead Plastic TSSOP
38-Lead Plastic TSSOP
–40°C to 125°C
–40°C to 125°C
–40°C to 150°C
–55°C to 150°C
LT3791FE
LT3791FE
LT3791FE
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are at TA = ±5°C (Note ±). VIN = 1±V, VEN/UVLO = 1±V unless otherwise noted.
PARAMETER
Input
CONDITIONS
MIN
TYP
MAX
UNITS
V
IN
V
IN
V
IN
Operating Voltage
4.7
60
1
V
µA
Shutdown I
V
= 0V
EN/UVLO
0.1
3.0
Q
Operating I (Not Switching)
FB = 1.3V, R = 59.0k
4
mA
Q
T
3791f
2
LT3791
ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are at TA = ±5°C (Note ±). VIN = 1±V, VEN/UVLO = 1±V unless otherwise noted.
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Logic Inputs
l
EN/UVLO Falling Threshold
EN/UVLO Rising Hysteresis
EN/UVLO Input Low Voltage
EN/UVLO Pin Bias Current Low
EN/UVLO Pin Bias Current High
CTRL Input Bias Current
CTRL Latch-Off Threshold
OVLO Rising Shutdown Voltage
OVLO Falling Hysteresis
Regulation
1.16
1.2
15
1.24
V
mV
V
I
Drops Below 1µA
0.3
4
VIN
V
V
V
= 1V
2
3
10
20
175
3
µA
nA
nA
mV
V
EN/UVLO
EN/UVLO
= 1.6V
100
50
= 1V
CTRL
l
l
2.85
1.96
3.15
75
mV
V
V
V
Voltage
2.00
2.04
0.04
V
REF
Line Regulation
4.7V < V < 60V
0.002
ꢀ/V
REF
IN
Threshold
V
V
V
V
= 2V
97.5
94
100
100
102.5
106
mV
mV
(ISP-ISN)
CTRL
CTRL
CTRL
CTRL
l
l
l
l
= 1100mV
= 700mV
= 300mV
87
84
90
90
93
96
mV
mV
47.5
46
50
50
52.5
54
mV
mV
6.5
5
10
10
13.5
15
mV
mV
ISP Bias Current
110
20
µA
µA
V
ISN Bias Current
LED Current Sense Common Mode Range
0
60
LED Current Sense Amplifier g
ISMON Monitor Voltage
890
1
µS
V
m
l
l
V
= 100mV
0.96
46.5
1.04
54
(ISP-ISN)
Input Current Sense Threshold V
IVINP Bias Current
3V ≤ V
≤ 60V
50
90
20
mV
µA
µA
V
(IVINP-IVINN)
IVINP
IVINN Bias Current
Input Current Sense Common Mode Range
3
60
Input Current Sense Amplifier g
IVINMON Monitor Voltage
FB Regulation Voltage
2.12
1
mS
V
m
l
l
V
= 50mV
0.96
1.04
(IVINP-IVINN)
1.194
1.176
1.2
1.2
1.206
1.220
V
V
FB Line Regulation
4.7V < V < 60V
0.002
565
0.025
ꢀ/V
µS
IN
FB Amplifier g
m
FB Pin Input Bias Current
FB in Regulation
PWM = 0V
100
150
20
nA
VC Standby Input Bias Current
–20
nA
l
l
V
(V
)
Boost
Buck
42
–56
51
–47.5
60
–39
mV
mV
SENSE(MAX) SNSP-SNSN
Fault
SS Pull-Up Current
SS Discharge Current
V
= 0V
14
µA
µA
V
SS
1.4
FB Overvoltage Rising Threshold
Open LED Rising Threshold (V
1.22
1.25
1.15
l
)
FB
V = 0V
(ISP-ISN)
1.127
1.173
V
3791f
3
LT3791
ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are at TA = ±5°C (Note ±). VIN = 1±V, VEN/UVLO = 1±V unless otherwise noted.
PARAMETER
CONDITIONS
MIN
1.078
5
TYP
1.1
10
MAX
1.122
15
UNITS
V
l
Open LED Falling Threshold (V
Open LED Falling Threshold (V
Short LED Falling Threshold (V
)
FB
)
V
FB
= 1.2V
mV
mV
kΩ
kΩ
V
(ISP-ISN)
)
FB
380
400
1.1
1.1
1.75
0.2
450
2.0
OPENLED Pin Output Impedance
SHORTLED Pin Output Impedance
SS Latch-Off Threshold
SS Reset Threshold
Oscillator
2.0
V
Switching Frequency
R = 147k
190
380
665
200
400
700
210
420
735
kHz
kHz
kHz
T
R = 59.0k
T
R = 29.1k
T
SYNC Frequency
200
0.3
4.8
3.1
700
kHz
kΩ
V
SYNC Pin Resistance to GND
SYNC Threshold Voltage
90
1.5
Internal V Regulator
CC
INTV Regulation Voltage
5
5.2
350
3.9
V
mV
V
CC
Dropout (V – INTV
)
I
= –10mA, V = 5V
240
3.5
67
IN
CC
INTVCC
IN
INTV Undervoltage Lockout
CC
INTV Current Limit
V
= 4V
mA
CC
INTVCC
PWM
PWM Threshold Voltage
PWM Pin Resistance to GND
PWMOUT Pull-Up Resistance
PWMOUT Pull-Down Resistance
NMOS Drivers
0.3
1.5
V
kΩ
Ω
90
10
5
20
10
Ω
TG1, TG2 Gate Driver On-Resistance
Gate Pull-Up
Gate Pull-Down
V – V = 5V
BST SW
2.6
1.7
Ω
Ω
BG1, BG2 Gate Driver On-Resistance
Gate Pull-Up
Gate Pull-Down
V
= 5V
INTVCC
3
1.2
Ω
Ω
TG Off to BG On Delay
BG Off to TG On Delay
C = 3300pF
60
60
ns
ns
ns
L
C = 3300pF
L
TG1, TG2, t
R = 59.0k
T
220
260
OFF(MIN)
operating junction temperature range. The LT3791MP is guaranteed to
meet performance specifications over the –55°C to 150°C operating
junction temperature range. High junction temperatures degrade operating
lifetimes. Operating lifetime is derated for junction temperatures greater
than 125°C.
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note ±: The LT3791E is guaranteed to meet performance from 0°C
to 125°C junction temperature. Specification over the -40°C to
125°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls.
The LT3791I is guaranteed to meet performance specifications over the
–40°C to 125°C operating junction temperature range. The LT3791H is
guaranteed to meet performance specifications over the –40°C to 150°C
Note 3: The LT3791 includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed the maximum operating junction temperature
when overtemperature protection is active. Continuous operation above
the specified absolute maximum operating junction temperature may
impair device reliability.
3791f
4
LT3791
TYPICAL PERFORMANCE CHARACTERISTICS TA = ±5°C, unless otherwise noted.
INTVCC Dropout Voltage
vs Current, Temperature
INTVCC Current Limit
vs Temperature
INTVCC Voltage vs Temperature
90
80
70
60
50
40
30
20
10
0
2.5
2.0
1.5
1.0
0.5
0
5.20
5.15
5.10
5.05
5.00
4.95
4.90
4.85
4.80
T
T
T
= 150°C
= 25°C
= –50°C
A
A
A
V
V
= 60V
= 12V
IN
IN
–50
50
100 125
150
10
20
30
–50 –25
0
25
50 75
100 125 150
–25
0
25
75
0
40
TEMPERATURE (°C)
LDO CURRENT (mA)
TEMPERATURE (°C)
3791 G03
3791 G01
3791 G02
INTVCC Load Regulation
VREF Voltage vs Temperature
VREF Load Regulation
6.00
2.04
2.03
2.02
2.01
2.00
1.99
1.98
1.97
1.96
2.20
2.15
2.10
2.05
2.00
1.95
1.90
1.85
1.80
5.75
5.50
5.25
5.00
4.75
4.50
4.25
V
V
V
= 60V
= 12V
= 4.7V
IN
IN
IN
4.00
50 75
TEMPERATURE (°C)
10
20
40
50
60
70
–50 –25
0
25
100 125 150
0
50
200 250
0
30
100 150
300 350
400
I
(mA)
I
(µA)
REF
LOAD
3791 G05
3791 G04
3791 G06
V(ISP-ISN) Threshold
vs Temperature
V(ISP-ISN) Threshold vs VCTRL
V(ISP-ISN) Threshold vs VISP
110
100
90
80
70
60
50
40
30
20
10
0
108
106
104
102
100
98
108
106
104
102
V
= 12V
IN
100
98
96
96
94
92
V
ISP
V
ISP
V
ISP
= 60V
= 12V
= 0V
94
92
50 75
25
TEMPERATURE (°C)
–50 –25
0
100 125 150
0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0
0
10
20
40
50
60
0
0.2
30
(V)
V
CTRL
(V)
V
ISP
3791 G09
3791 G07
3791 G08
3791f
5
LT3791
TYPICAL PERFORMANCE CHARACTERISTICS TA = ±5°C, unless otherwise noted.
V(ISP-ISN) Threshold vs VFB
ISMON Voltage vs Temperature
ISMON Voltage vs V(ISP-ISN)
120
100
80
60
40
20
0
1.0
0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0
1.04
1.03
1.02
1.01
1.00
0.99
0.98
0.97
0.96
V = 12V
IN
V
(ISP-ISN)
= 100mV
1.17
1.19 1.20 1.21
(V)
1.22 1.23
1.18
50 75
TEMPERATURE (°C)
0
–50 –25
0
25
100 125 150
30 40 50
10 20 60 70 80 90 100
V
FB
V
(mV)
(ISP-ISN)
3791 G10
3791 G11
3791 G12
V(IVINP-IVINN) Threshold
vs Temperature
V(IVINP-IVINN) Threshold
vs VIVINP
IVINMON Voltage vs Temperature
56
54
52.0
51.5
51.0
50.5
1.04
1.03
1.02
1.01
1.00
0.99
0.98
0.97
0.96
V = 12V
IVINP
V
(IVINP-VINN)
= 50mV
52
V
= 60V
= 3V
IVINP
50
48
46
44
V
IVINP
50.0
49.5
49.0
48.5
48.0
42
10
20
40
50 75
0
50
60
–50 –25
–25
0
150
30
0
25
100 125
150
–50
25 50 75 100 125
TEMPERATURE (°C)
V
(V)
TEMPERATURE (°C)
IVINP
3791 G14
3791 G15
3791 G13
FB Regulation Voltage
vs Temperature
SHORTLED Threshold
V
(IVINP-IVINN) Threshold vs VFB
vs Temperature
60
50
40
30
20
10
0
0.500
0.475
0.450
0.425
0.400
0.375
0.350
0.325
0.300
1.24
1.23
1.22
1.21
1.20
1.19
1.18
1.17
1.16
RISING
FALLING
V
V
V
= 60V
= 12V
= 4.7V
IN
IN
IN
50 75
25
TEMPERATURE (°C)
–50 –25
0
100 125 150
1.17
1.19 1.20 1.21
(V)
1.22 1.23
50 75
TEMPERATURE (°C)
1.18
–50 –25
0
25
100 125 150
V
FB
3791 G18
3791 G16
3791 G17
3791f
6
LT3791
TYPICAL PERFORMANCE CHARACTERISTICS TA = ±5°C, unless otherwise noted.
OPENLED Threshold
vs Temperature
OVLO Threshold vs Temperature
Soft-Start Current vs Temperature
3.3
3.2
3.1
3.0
2.9
2.8
2.7
2.6
2.5
16
14
12
10
8
1.200
1.175
1.150
1.125
1.100
1.075
1.050
1.025
1.000
CHARGING
RISING
RISING
FALLING
FALLING
6
4
DISCHARGING
2
0
50 75
TEMPERATURE (°C)
50 75
25
TEMPERATURE (°C)
–50 –25
0
25
100 125 150
–50 –25
0
100 125 150
50 75
TEMPERATURE (°C)
–50 –25
0
25
100 125 150
3791 G20
3791 G21
3791 G19
Supply Current vs Input Voltage
EN/UVLO Pin Current
EN/UVLO Threshold Voltage
1.30
1.28
1.26
1.24
1.22
1.20
1.18
1.16
1.14
1.12
1.10
4.0
3.5
3.0
2.5
8
7
6
5
4
3
2
1
0
V
= 1V
EN/ULO
RISING
2.0
1.5
FALLING
1.0
0.5
0
T
T
T
= 150°C
= 25°C
= –50°C
A
A
A
10
20
40
–50 –25
50 75
25
TEMPERATURE (°C)
0
50
60
0
100 125
150
30
(V)
–50
125
150
–25
0
25 50 75 100
TEMPERATURE (°C)
V
IN
3791 G22
3791 G23
3791 G24
Oscillator Frequency
vs Temperature
TG1, TG± Minimum On-Time
vs Temperature
TG1, TG± Minimum Off-Time
vs Temperature
350
300
100
90
80
70
60
50
40
30
20
800
700
600
500
400
300
200
100
0
TG2
f
f
= 200kHz
= 400kHz
SW
SW
R
R
= 29.1k
T
TG1
250
200
150
100
50
= 59.0k
= 147k
T
f
= 700kHz
SW
R
T
0
–25
0
25 50 75
150
100 125
–50
50 75
TEMPERATURE (°C)
50 75
25
TEMPERATURE (°C)
–50 –25
0
25
100 125 150
–50 –25
0
100 125 150
TEMPERATURE (˚C)
3791 G27
3791 G26
3791 G25
3791f
7
LT3791
TYPICAL PERFORMANCE CHARACTERISTICS TA = ±5°C, unless otherwise noted.
V(BST1-SW1), V(BST±-SW±) UVLO
vs Temperature
BG1, BG± Driver On-Resistance
vs Temperature
TG1, TG± Driver On-Resistance
vs Temperature
3.9
3.8
3.7
3.6
3.5
3.4
3.3
3.2
3.1
4.5
4.0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0
4.0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0
PULL-UP
RISING
PULL-UP
PULL-DOWN
PULL-DOWN
FALLING
50 75
TEMPERATURE (°C)
50 75
TEMPERATURE (°C)
–50 –25
0
25
100 125 150
–50 –25
0
25
100 125 150
50 75
TEMPERATURE (°C)
–50 –25
0
25
100 125 150
3791 G28
3791 G29
3791 G30
PWMOUT On-Resistance
vs Temperature
V(SNSP-SNSN) Buck Threshold
vs VC
VC Voltage vs Duty Cycle
14
12
60
40
1.6
1.4
1.2
1.0
V
= 0V
(SNSP-SNSN)
BG2
BG1
10
PULL-UP
20
8
6
4
2
0.8
0.6
0
PULL-DOWN
–20
–40
–60
0.4
0.2
0
0
–25
0
25 50 75
150
20
40
DUTY CYCLE (%)
80
–50
100 125
0
100
0.6
1.0
1.2
1.4
1.6
1.8
60
0.8
VC (V)
TEMPERATURE (˚C)
3791 G31
3791 G32
3791 G33
V(SNSP-SNSN) Buck Threshold
vs Temperature
V(SNSP-SNSN) Boost Threshold
vs VC
V(SNSP-SNSN) Boost Threshold
vs Temperature
60
40
60
40
60
40
V
V
C(MIN)
C(MAX)
20
20
20
0
0
–20
–40
–60
0
–20
–40
–60
–80
–20
–40
–60
V
C(MAX)
V
C(MIN)
–80
75 100
–50 –25
0
25 50
125 150
–25
0
25 50 75
150
100 125
1.4
1.8
–50
0.6
0.8
1.0
1.2
1.6
TEMPERATURE (°C)
TEMPERATURE (˚C)
VC (V)
3791 G34
3791 G36
3791 G35
3791f
8
LT3791
PIN FUNCTIONS
CTRL(Pin1):CurrentSenseThresholdAdjustmentPinfor
AnalogDimming.RegulatingthresholdV
EN/UVLO (Pin 9): Enable Control Pin. Forcing an accurate
1.2V falling threshold with an externally programmable
hysteresis is generated by the external resistor divider
and a 3µA pull-down current. Above the 1.2V (typical)
threshold (but below 6V), EN/UVLO input bias current is
sub-µA. Below the falling threshold, a 3µA pull-down cur-
rent is enabled so the user can define the hysteresis with
the external resistor selection. An undervoltage condition
resets soft-start. Tie to 0.3V, or less, to disable the device
is1/10th
(ISP-ISN)
of (V
– 200mV). CTRL linear range is from 200mV
CTRL
to 1.1V. For V
> 1.3V, the current sense threshold is
CTRL
constant at the full-scale value of 100mV. For 1.1V < V
CTRL
< 1.3V, the dependence of the current sense threshold
upon V transitions from a linear function to a con-
CTRL
stant value, reaching 98ꢀ of full scale by V
= 1.2V.
CTRL
Connect CTRL to V
for the 100mV default threshold.
REF
Force less than 175mV (typical) to stop switching. Do not
and reduce V quiescent current below 1µA.
IN
leave this pin open.
IVINP (Pin 10): Positive Input for the Input Current Limit
andMonitor.Inputbiascurrentforthispinistypically90µA.
SS (Pin ±): Soft-start reduces the input power sources
surge current by gradually increasing the controller’s cur-
rent limit. A minimum value of 10nF is recommended on
this pin. SS is used as a timer when an open or shorted
LED condition occurs. A 500k resistor placed from SS to
IVINN (Pin 11): Negative Input for the Input Current Limit
and Monitor. The input bias current for this pin is typically
20µA.
V (Pin 1±): Main Input Supply. Bypass this pin to PGND
V
will latch the part off in the event of a fault. A 100k
IN
REF
resistor to V
with a capacitor.
will allow the part to keep running in a
REF
fault. If left open, a 1.4µA current source pulls down on
SS and the part restarts in a fault.
INTV (Pin 13): Internal 5V Regulator Output. The driver
CC
andcontrolcircuitsarepoweredfromthisvoltage. Bypass
this pin to PGND with a minimum 4.7µF ceramic capacitor.
PWM (Pin3):Asignallowturnsoffswitches, idlesswitch-
inganddisconnectstheVCpinfromallexternalloads. The
PWMOUT pin follows the PWM pin. PWM has an internal
TG1 (Pin 14): Top Gate Drive. Drives the top N-channel
MOSFET with a voltage equal to INTV superimposed on
CC
100k pull-down resistor. If not used, connect to INTV .
CC
the switch node voltage SW1.
OPENLED (Pin 4): An open-drain pull-down on OPENLED
BST1 (Pin 15): Bootstrapped Driver Supply. The BST1 pin
asserts if FB is greater than 1.15V (typical) and V
(ISP-ISN)
swings from a diode voltage below INTV up to a diode
CC
is less than 10mV (typical). To function, the pin requires
an external pull-up resistor.
voltage below V + INTV .
IN
CC
SW1 (Pin 16): Switch Node. SW1 pin swings from a diode
SHORTLED (Pin 5): An open-drain pull-down on
voltage drop below ground up to V .
IN
SHORTLED asserts if FB is less than 400mV (typical)
.
PGND (Pins 17, ±0): Power Ground. Connect these pins
closely to the source of the bottom N-channel MOSFET.
To function, the pin requires an external pull-up resistor.
V
(Pin 6): Voltage Reference Output Pin, Typically 2V.
REF
BG1 (Pin 18): Bottom Gate Drive. Drives the gate of the
This pin drives a resistor divider for the CTRL pin, either
for analog dimming or for temperature limit/compensa-
tion of the LED load. Can supply up to 200µA of current.
bottom N-channel MOSFET between ground and INTV .
CC
BG± (Pin 19): Bottom Gate Drive. Drives the gate of the
bottom N-channel MOSFET between ground and INTV .
CC
ISMON (Pin 7): Monitor pin that produces a voltage that
is ten times the voltage V
. ISMON will equal 1V
(ISP-ISN)
SW± (Pin ±1): Switch Node. SW2 pin swings from a diode
when V
= 100mV.
voltage drop below ground up to V
.
(ISP-ISN)
OUT
IVINMON (Pin 8): Monitor pin that produces a voltage
that is twenty times the voltage V . IVINMON
BST± (Pin ±±): Bootstrapped Driver Supply. The BST2 pin
swings from a diode voltage below INTV up to a diode
(IVINP-IVINN)
= 50mV.
CC
will equal 1V when V
voltage below V
+ INTV .
(IVINP-IVINN)
OUT
CC
3791f
9
LT3791
PIN FUNCTIONS
NC (Pin ±3): No Connect Pin. Leave this pin floating.
TEST± (Pin 3±): This pin is used for testing purposes only
and must be connected to INTV (Pin 13) for the part to
CC
TG± (Pin ±4): Top Gate Drive. Drives the top N-channel
operate properly.
MOSFET with a voltage equal to INTV superimposed on
CC
CLKOUT (Pin 33): Clock Output Pin. An in-phase clock is
provided at the oscillator frequency to allow for synchro-
nizing two devices for extending output power capability.
the switch node voltage SW2.
ISP (Pin ±5): Connection Point for the Positive Terminal
of the Output Current Feedback Resistor.
SYNC (Pin 34): External Synchronization Input Pin. This
pin is internally terminated to GND with a 90k resistor.
The rising edge will be synchronized with the rising edge
of the SYNC signal.
ISN (Pin ±6): Connection Point for the Negative Terminal
of the Output Current Feedback Resistor.
SNSP (Pin ±7): The Positive Input to the Current Sense
Comparator. The VC pin voltage and controlled offsets
between the SNSP and SNSN pins, in conjunction with a
resistor, set the current trip threshold.
RT (Pin 35): Frequency Set Pin. Place a resistor to GND
to set the internal frequency. The range of oscillation is
200kHz to 700kHz.
SNSN (Pin ±8): The Negative Input to the Current Sense
Comparator.
VC(Pin36):CurrentControlThresholdandErrorAmplifier
Compensation Point. The current comparator threshold
increases with this control voltage. The voltage ranges
from 0.7V to 1.9V.
TEST1 (Pin ±9): This pin is used for testing purposes only
and must be connected to SGND for the part to operate
properly.
FB (Pin 37): Voltage Loop Feedback Pin. FB is intended
for constant-voltage regulation or for LED protection of
an open or shorted LED. The internal transconductance
amplifier with output VC will regulate FB to 1.2V (typical)
through the DC/DC converter. If the FB input is regulating
SGND (Pin 30, Exposed Pad Pin 39): Signal Ground.
All small-signal components and compensation should
connect to this ground, which should be connected to
PGND at a single point. Solder the exposed pad directly
to the ground plane.
the loop and V
< 10mV, the OPENLED pull-down is
(ISP-ISN)
asserted. If the FB pin is less than 400mV, the SHORTLED
PWMOUT (Pin 31): Buffered Version of PWM Signal for
pull-down is asserted.
Driving LED Load Disconnect N-Channel MOSFET. The
PWMOUT pin is driven from INTV . Use of a MOSFET
OVLO (Pin 38): Overvoltage Input Pin. This pin is used for
OVLO, if OVLO > 3V then SS is pulled low, the part stops
switching and resets. Do not leave this pin open.
CC
with a gate cutoff voltage higher than 1V is recommended.
3791f
10
LT3791
BLOCK DIAGRAM
25
26
10
IVINP
11
IVINN
12
6
13
INTV
CC
ISP
ISN
V
V
REF
IN
–
–
+
+
A2
A1
REGS
TSD
SHDN_INT
A = 10
A = 10 A = 20
A = 24
ISMON
7
BST1
15
ISMON_INT
IVINMON_INT
+
A13
IVINMON
EN/UVLO
TG1
A3
8
9
14
16
SW1
–
–
A4
BUCK
LOGIC
SHDN_INT
3µA
SHDN_INT
SS_RESET
SS LATCH
PWM
+
1.2V
INTV
CC
A14
A15
BG1
PGND
BG2
18
17
19
OSC
SLOPE_COMP_BOOST
INTV
CC
RT
35
34
33
BOOST
LOGIC
SYNC
CLKOUT
SW2
TG2
21
24
22
+
–
A16
A7
SLOPE_COMP_BUCK
BST2
SNSP
+
A10
27
SHORTLED
5
SNSN
0.4V
+
–
28
–
A5
A6
–
–
+
IVINMON_INT
FB
A11
A12
37
FB
V
REF
0.2V
+
–
+
–
A8
A9
OPENLED
1.2V
14µA
4
+
+
–
1.15V
CTRL
1
SS RESET
SS LATCH
+
–
R
ISMON_INT
Q
S
1.75V
INTV
3V
CC
+
–
A18
PWM
A17
3
OVLO
1.4µA
38
SGND
30, 39
PWMOUT
SS
VC
31
2
36
3791 BD
3791f
11
LT3791
OPERATION
The LT3791 is a current mode controller that provides an
output voltage above, equal to or below the input voltage.
TheLTC proprietarytopologyandcontrolarchitectureuses
a current sensing resistor in buck or boost operation. The
sensed inductor current is controlled by the voltage on
the VC pin, which is the output of the feedback amplifiers
A11 and A12. The VC pin is controlled by three inputs,
one input from the output current loop, one input from the
input current loop, and the third input from the feedback
loop.Whicheverfeedbackinputishighertakesprecedence,
forcing the converter into either a constant-current or a
constant-voltage mode.
slowlychargedduringstart-up. This“soft-start”clamping
prevents abrupt current from being drawn from the input
power supply. The SS can also be used as a fault timer
whenever an open or shorted LED is detected.
The top MOSFET drivers are biased from floating boot-
strap capacitors C1 and C2, which are normally recharged
through an external diode when the top MOSFET is turned
off. Schottky diodes across the synchronous switch M4
and synchronous switch M2 are not required, but they do
provide a lower drop during the dead time. The addition
of the Schottky diode typically improves peak efficiency
by 1ꢀ to 2ꢀ at 500kHz.
The LT3791 is designed to transition cleanly between
the two modes of operation. Current sense amplifier A1
senses the voltage between the IVINP and IVINN pins and
provides a pre-gain to amplifier A11. When the voltage
between IVINP and IVINN reaches 50mV, the output of A1
provides IVINMON_INT to the inverting input of A11 and
the converter is in constant-current mode. If the current
sense voltage exceeds 50mV, the output of A1 increases
causing the output of A11 to decrease, thus reducing the
amount of current delivered to the output. In this manner
the current sense voltage is regulated to 50mV.
Power Switch Control
Figure 1 shows a simplified diagram of how the four
power switches are connected to the inductor, V , V
IN OUT
and GND. Figure 2 shows the regions of operation for the
LT3791 as a function of duty cycle D. The power switches
are properly controlled so the transfer between regions is
continuous. When V approaches V , the buck-boost
IN
OUT
region is reached.
V
IN
V
OUT
The output current amplifier works similar to the input
current amplifier but with a 100mV voltage instead of
50mV. The output current sense level is also adjustable
by the CTRL pin. Forcing CTRL to less than 1.2V forces
ISMON_INT to the same level as CTRL, thus providing
current-levelcontrol.Theoutputcurrentamplifierprovides
rail-to-rail operation. Similarly if the FB pin goes above
1.2V the output of A11 decreases to reduce the current
level and regulate the output (constant-voltage mode).
TG1
BG1
M1
M4
TG2
BG2
L1
SW1
SW2
M2
M3
R
SENSE
3791 F01
Figure 1. Simplified Diagram of the Output Switches
D
TheLT3791providesmonitoringpinsIVINMONandISMON
that are proportional to the voltage across the input and
output current amplifiers respectively.
MAX
BOOST
M1 ON, M2 OFF
(BG2)
BOOST REGION
PWM M3, M4 SWITCHES
D
MIN
BOOST
BUCK-BOOST REGION
BUCK REGION
4-SWITCH PWM
D
BUCK
(TG1)
The main control loop is shut down by pulling the EN/
UVLO pin low. When the EN/UVLO pin is higher than 1.2V,
an internal 14µA current source charges soft-start capaci-
MAX
M4 ON, M3 OFF
PWM M2, M1 SWITCHES
D
BUCK
MIN
3791 F02
tor C at the SS pin. The VC voltage is then clamped a
SS
Figure ±. Operating Regions vs Duty Cycle
diode voltage higher than the SS voltage while the C is
SS
3791f
12
LT3791
OPERATION
Buck Region (V > V
)
where D is the duty cycle of the buck-boost
(BUCK-BOOST)
IN
OUT
switch range:
SwitchM4isalwaysonandswitchM3isalwaysoffduring
this mode. At the start of every cycle, synchronous switch
M2 is turned on first. Inductor current is sensed when
synchronous switch M2 is turned on. After the sensed
inductor current falls below the reference voltage, which
isproportionaltoVC, synchronousswitchM2isturnedoff
and switch M1 is turned on for the remainder of the cycle.
Switches M1 and M2 will alternate, behaving like a typical
synchronous buck regulator. The duty cycle of switch M1
increases until the maximum duty cycle of the converter
D
= 8ꢀ
(BUCK-BOOST)
Figure 3 shows typical buck operation waveforms. If V
IN
approaches V , the buck-boost region is reached.
OUT
Buck-Boost (V ~ V
)
IN
OUT
When V is close to V , the controller is in buck-boost
IN
OUT
operation. Figure 4 and Figure 5 show typical waveforms
in this mode. Every cycle the controller turns on switches
M2 and M4, then M1 and M4 are turned on until 180° later
when switches M1 and M3 turn on, and then switches
M1 and M4 are turned on for the remainder of the cycle.
in buck operation reaches D
, given by:
MAX(BUCK, TG1)
D
= 100ꢀ – D
(BUCK-BOOST)
MAX(BUCK,TG1)
M2 + M4
M2 + M4
M2 + M4
M1 + M4
M1 + M4
M1 + M4
3791 F03
Figure 3. Buck Operation (VIN > VOUT
)
M1 + M4
M1 + M4
M1 + M4
M2 + M4
M1 + M4
M2 + M4
M2 + M4
M1+ M3
M1+ M3
M1+ M3
M1 + M4
M1 + M4
3791 F04
Figure 4. Buck-Boost Operation (VIN ≤ VOUT
)
M1 + M4
M1 + M3
M1 + M4
M1 + M3
M1 + M4
M1 + M3
M2 + M4
M2 + M4
M2 + M4
M1 + M4
M1 + M4
M1 + M4
3791 F05
Figure 5. Buck-Boost Operation (VIN ≥ VOUT
)
3791f
13
LT3791
OPERATION
Low Current Operation
Boost Region (V < V
)
OUT
IN
The LT3791 runs in forced continuous mode. In this mode
the controller behaves as a continuous, PWM current
mode synchronous switching regulator. In boost opera-
tion, switch M1 is always on, switch M3 and synchronous
switch M4 are alternately turned on to maintain the output
voltage independent of the direction of inductor current.
In buck operation, synchronous switch M4 is always on,
switch M1 and synchronous switch M2 are alternately
turned on to maintain the output voltage independent of
the direction of inductor current. In this mode, the output
can source or sink current.
Switch M1 is always on and synchronous switch M2 is
always off in boost operation. Every cycle switch M3 is
turned on first. Inductor current is sensed when synchro-
nous switch M3 is turned on. After the sensed inductor
currentexceedsthereferencevoltagewhichisproportional
to VC, switch M3 turns off and synchronous switch M4
is turned on for the remainder of the cycle. Switches M3
and M4 alternate, behaving like a typical synchronous
boost regulator.
The duty cycle of switch M3 decreases until the minimum
duty cycle of the converter in boost operation reaches
DMIN(BOOST,BG2), given by:
D
= D
(BUCK-BOOST)
MIN(BOOST,BG2)
where D
is the duty cycle of the buck-boost
(BUCK-BOOST)
switch range:
D
= 8ꢀ
(BUCK-BOOST)
Figure 6 shows typical boost operation waveforms. If V
IN
approaches V , the buck-boost region is reached.
OUT
M1 + M3
M1 + M3
M1 + M3
M1 + M4
M1 + M4
M1 + M4
3791 F06
Figure 6. Boost Operation (VIN < VOUT
)
3791f
14
LT3791
APPLICATIONS INFORMATION
TheTypicalApplicationonthefrontpageisabasicLT3791
application circuit. External component selection is driven
by the load requirement, and begins with the selection of
TherisingedgeofCLK_OUTcorrespondstotherisingedge
of SYNC thus allowing paralleling converters. The falling
edge of CLK_OUT turns on switch M3 and the rising edge
of CLK_OUT turns on switch M2.
R
and the inductor value. Next, the power MOSFETs
SENSE
areselected. Finally, C andC
areselected. Thiscircuit
IN
OUT
Inductor Selection
can operate up to an input voltage of 60V.
The operating frequency and inductor selection are inter-
related in that higher operating frequencies allow the use
of smaller inductor and capacitor values. The inductor
value has a direct effect on ripple current. The maximum
Programming The Switching Frequency
TheRT frequencyadjustpinallowstheusertoprogramthe
switching frequency from 200kHz to 700kHz to optimize
efficiency/performanceorexternalcomponentsize.Higher
frequency operation yields smaller component size but
increases switching losses and gate driving current, and
maynotallowsufficientlyhighorlowdutycycleoperation.
Lowerfrequencyoperationgivesbetterperformanceatthe
cost of larger external component size. For an appropriate
inductor current ripple ΔI can be seen in Figure 7. This
L
is the maximum ripple that will prevent subharmonic
oscillation and also regulate with zero load. The ripple
should be less than this to allow proper operation over
all load currents. For a given ripple the inductance terms
in continuous mode are as follows:
R resistor value see Table 1. An external resistor from
T
VOUT • VIN(MAX) – VOUT •100
(
)
the RT pin to GND is required; do not leave this pin open.
LBUCK
>
f •ILED •%Ripple• V
IN(MAX)
Table 1. Switching Frequency vs RT Value
f
(kHz)
R (kΩ)
OSC
T
V
2 • VOUT – V
•100
(
)
IN(MIN)
IN(MIN)
200
147
84.5
59.0
45.3
35.7
29.4
LBOOST
>
2
f •ILED •%Ripple• VOUT
300
400
500
600
700
where:
f is operating frequency
ꢀ ripple is allowable inductor current ripple
V
V
V
is minimum input voltage
is maximum input voltage
is output voltage
Frequency Synchronization
IN(MIN)
IN(MAX)
The LT3791 switching frequency can be synchronized
to an external clock using the SYNC pin. Driving SYNC
with a 50ꢀ duty cycle waveform is always a good choice,
otherwise maintain the duty cycle between 10ꢀ and 90ꢀ.
OUT
I
is current through the LEDs
LED
3791f
15
LT3791
APPLICATIONS INFORMATION
200
180
160
140
where ΔI is peak-to-peak inductor ripple current. In buck
L
operation, the maximum average load current is:
47.5mV ∆IL
IOUT(MAX _BUCK)
=
+
BOOST ∆I /
SENSE(MAX)
L
120
100
80
60
40
20
0
R
2
I
LIMIT
SENSE
The maximum current sensing R
operation is:
value for the boost
SENSE
BUCK ∆I /
SENSE(MAX)
L
I
LIMIT
2•51mV•V
IN(MIN)
RSENSE(MAX)
=
2•ILED •VOUT + ∆IL(BOOST) •VIN(MIN)
50 55 60 65 70 75 80 85 90 95 100
BG1, BG2 DUTY CYCLE (%)
3791 F07
The maximum current sensing R
operation is:
value for the buck
SENSE
Figure 7. Maximum Peak-to-Peak Ripple vs Duty Cycle
2•47.5mV
2•ILED – ∆IL(BUCK)
RSENSE(MAX)
=
For high efficiency, choose an inductor with low core
loss. Also, the inductor should have low DC resistance to
2
The final R
SENSE(MAX)
to 30ꢀ margin is usually recommended.
value should be lower than the calculated
reduce the I R losses, and must be able to handle the peak
SENSE
R
in both the boost and buck operation. A 20ꢀ
inductor current without saturating. To minimize radiated
noise, use a shielded inductor.
C and C Selection
R
Selection and Maximum Output Current
IN
OUT
SENSE
In boost operation, input current is continuous. In buck
operation, input current is discontinuous. In buck opera-
R
ischosenbasedontherequiredoutputcurrent.The
SENSE
current comparator threshold sets the peak of the induc-
tor current in boost operation and the maximum inductor
valley current in buck operation. In boost operation, the
tion, the selection of input capacitor, C , is driven by the
IN
need to filter the input square wave current. Use a low ESR
capacitor sized to handle the maximum RMS current. For
buck operation, the input RMS current is given by:
maximum average load current at V
is:
IN(MIN)
VIN(MIN)
51mV ∆IL
IOUT(MAX _BOOST)
=
–
•
2
∆IL
R
2
VOUT
2
SENSE
IRMS = ILED •D+
•D
12
3791f
16
LT3791
APPLICATIONS INFORMATION
Programming V UVLO and OVLO
The formula has a maximum at V = 2V . Note that
IN
IN
OUT
ripple current ratings from capacitor manufacturers are
often based on only 2000 hours of life which makes it
advisable to derate the capacitor.
ThefallingUVLOvaluecanbeaccuratelysetbytheresistor
divider R1 and R2. A small 3µA pull-down current is active
when the EN/UVLO is below the threshold. The purpose
of this current is to allow the user to program the rising
hysteresis. The following equations should be used to
determine the resistor values:
In boost operation, the discontinuous current shifts
from the input to the output, so C
must be capable
OUT
of reducing the output voltage ripple. The effects of ESR
(equivalent series resistance) and the bulk capacitance
must be considered when choosing the right capacitor
for a given output ripple voltage. The steady ripple due to
charginganddischargingthebulkcapacitanceisgivenby:
R1+ R2
R2
V
IN(UVLO ) = 1.2•
–
R1+ R2
V
+ = 3µA •R1+ 1.215•
IN(UVLO )
R2
I
• V
– V
IN(MIN)
OUT
(
)
LED
∆V
=
RIPPLE BOOST _CAP
(
)
The rising OVLO value can be accurately set by the resis-
tor divider R3 and R4. The following equations should be
used to determine the resistor values:
C
• V
• f
OUT
OUT
∆I
8 • f •C
L
∆V
≈
RIPPLE BUCK _CAP
(
)
OUT
R3+ R4
+
V
IN(OVLO ) = 3•
where C
is the output filter capacitor.
R4
OUT
The steady ripple due to the voltage drop across the ESR
is given by:
R3+ R4
V
IN(OVLO ) = 2.925•
–
R4
ΔV
ΔV
= I
• ESR
BOOST(ESR)
LED
V
IN
= I
• ESR
LED
BUCK(ESR)
Multiple capacitors placed in parallel may be needed to
meet the ESR and RMS current handling requirements.
OutputcapacitorsarealsousedforstabilityfortheLT3791.
A good starting point for output capacitors is seen in the
Typical Applications circuits. Ceramic capacitors have
excellent low ESR characteristics but can have a high
voltage coefficient and are recommended for applications
less than 100W. Capacitors available with low ESR and
high ripple current ratings, such as OS-CON and POSCAP
may be needed for applications greater than 100W.
R3
R1
LT3791
OVLO
EN/UVLO
R2
R4
3791 F08
Figure 8. Resistor Connection to Set VIN UVLO and
OVLO Thresholds
3791f
17
LT3791
APPLICATIONS INFORMATION
Programming LED Current
The CTRL pin should not be left open (tie to V
if not
REF
used). The CTRL pin can also be used in conjunction with
a thermistor to provide overtemperature protection for
The LED current is programmed by placing an appropriate
value current sense resistor, R , in series with the LED
LED
the LED load, or with a resistor divider to V to reduce
IN
string. The voltage drop across R
is (Kelvin) sensed
LED
output power and switching current when V is low.
IN
by the ISP and ISN pins. The CTRL pin should be tied to
a voltage higher than 1.2V to get the full-scale 100mV
(typical) threshold across the sense resistor. The CTRL
pin can also be used to dim the LED current, although
relative accuracy decreases with the decreasing sense
threshold. When the CTRL pin voltage is less than 1V,
the LED current is:
The presence of a time varying differential voltage signal
(ripple) across ISP and ISN at the switching frequency
is expected. The amplitude of this signal is increased by
high LED load current, low switching frequency and/or a
smaller value output filter capacitor. Some level of ripple
signal is acceptable: the compensation capacitor on the
VC pin filters the signal so the average difference between
ISP and ISN is regulated to the user-programmed value.
Ripple voltage amplitude (peak-to-peak) in excess of
20mV should not cause mis-operation, but may lead to
noticeable offset between the average value and the user-
programmed value.
VCTRL –200mV
ILED
=
RLED •10
When the CTRL pin voltage is between 1.1V and 1.3V
the LED current varies with V , but departs from the
CTRL
equation above by an increasing amount as V
voltage
CTRL
ISMON
increases. Ultimately, when V
> 1.3V the LED current
CTRL
no longer varies. The typical V
threshold vs V
(ISP-ISN)
CTRL
The ISMON pin provides a linear indication of the cur-
rent flowing through the LEDs. The equation for V
is listed in Table 2.
ISMON
is V
• 10. This pin is suitable for driving an ADC
(ISP–ISN)
Table ±. V(ISP-ISN) Threshold vs CTRL
input,however,theoutputimpedanceofthispinis12.5kΩ
V
(V)
V
(mV)
CTRL
(ISP-ISN)
so care must be taken not to load this pin.
1.1
90
1.15
1.2
94.5
98
Programming Input Current Limit
The LT3791 has a standalone current sense amplifier. It
can be used to limit the input current. The input current
limit is calculated by the following equation:
1.25
1.3
99.5
100
When V
is higher than 1.3V, the LED current is
CTRL
regulated to:
50mV
IIN =
RIN
100mV
RLED
ILED
=
3791f
18
LT3791
APPLICATIONS INFORMATION
For loop stability a lowpass RC filter is needed. For
most applications, a 50Ω resistor and 470nF capacitor
is sufficient.
V
OUT
R5
LT3791
FB
R6
Table 3
R
IN
(mΩ)
20
15
12
10
6
I
(A)
3791 F09
LIMIT
2.5
Figure 9. Resistor Connection for Open LED Threshold
and Constant Output Voltage Regulation
3.3
4.2
5.0
8.3
Dimming Control
5
10.0
12.5
16.7
25
There are two methods to control the current source for
dimming using the LT3791. One method uses the CTRL
pin to adjust the current regulated in the LEDs. A second
method uses the PWM pin to modulate the current source
between zero and full current to achieve a precisely pro-
grammed average current. To make PWM dimming more
accurate, the switch demand current is stored on the VC
node during the quiescent phase when PWM is low. This
featureminimizesrecoverytimewhenthePWM signalgoes
high. To further improve the recovery time a disconnect
switch may be used in the LED current path to prevent the
ISP node from discharging during the PWM signal low
phase. The minimum PWM on- or off-time is affected by
choice of operating frequency and external component
selection. The best overall combination of PWM and
analog dimming capabilities is available if the minimum
PWM pulse is at least six switching cycles and the PWM
pulse is synchronized to the SYNC signal.
4
3
2
IVINMON
TheIVINMONpinprovidesalinearindicationofthecurrent
flowing through the input. The equation for V is
IVINMON
V
• 20. This pin is suitable for driving an ADC
(IVINP-IVINN)
input,however,theoutputimpedanceofthispinis12.5kΩ
so care must be taken not to load this pin.
Programming Output Voltage (Constant Voltage
Regulation) or Open LED/Overvoltage Threshold
For a voltage regulator, the output voltage can be set by
selecting the values of R5 and R6 (see Figure 9) according
to the following equation:
R5+ R6
VOUT = 1.2•
R6
SHORTLED Pin
For an LED driver application, set the resistor from the
The LT3791 provides an open-drain status pin,
SHORTLED, which pulls low when the FB pin is below
400mV. The only time the FB pin will be below 400mV
is during start-up or if the LEDs are shorted. During
output to the FB pin such that the expected V during
FB
normaloperationdoesnotexceed1.1V.OnceV ishigher
FB
thanitsovervoltagethreshold,1.25V(typical),theLT3791
stops switching.
3791f
19
LT3791
APPLICATIONS INFORMATION
start-up the LT3791 ignores the voltage on the FB pin
until the soft-start capacitor reaches 1.75V. To prevent
false tripping after startup, a large enough soft-start
capacitor must be used to allow the output to get up to
approximately 40ꢀ to 50ꢀ of the final value.
The SS pin is also used as a fault timer. Once an open
LED or a shorted LED fault is detected, a 1.4µA pull-down
current source is activated. With a 100k pull-up resistor
to V on the SS pin, the LT3791 will continue to switch
REF
normally. With a 500k pull-up resistor to V on the SS
REF
pin, the LT3791 will latch off until the EN/UVLO pin is
OPENLED Pin
toggled. Without any resistor to V
the SS pin enters
REF
a hiccup mode operation. The 1.4µA pulls SS down until
0.2V is reached, at which point the 14µA pull-up current
sourceturnson. Ifthefaultconditionhasn’tbeenremoved
when SS reaches 1.75V, then the 1.4µA pull-down cur-
rent source turns on again initiating a new cycle. This will
continue until the fault is removed.
The LT3791 provides an open-drain status pin, OPENLED,
which pulls low when the FB pin is above 1.15V and the
voltage across V
is less than 10mV. If the open
(ISP-ISN)
LED clamp voltage is programmed correctly using the FB
pin, then the FB pin should never exceed 1.1V when the
LEDs are connected. Therefore, the only way for the FB
pin to exceed 1.15V is for an open LED event to occur.
Loop Compensation
Soft-Start, Fault Function
TheLT3791usesaninternaltransconductanceerrorampli-
fier whose VC output compensates the control loop. The
external inductor, output capacitor and the compensation
resistor and capacitor determine the loop stability.
Soft-startreducestheinputpowersources’surgecurrents
by gradually increasing the controller’s current limit (pro-
portional to an internally buffered clamped equivalent of
VC).Thesoft-startintervalissetbythesoft-startcapacitor
selection according to the following equation
The inductor and output capacitor are chosen based on
performance, size and cost. The compensation resistor
and capacitor at VC are set to optimize control loop re-
sponse and stability. For typical LED applications, a 10nF
compensation capacitor at VC is adequate, and a series
resistor should always be used to increase the slew rate
on the VC pin to maintain tighter regulation of LED current
during fast transients on the input supply of the converter.
1.2V
14µA
tSS
=
•CSS
Make sure C is large enough when there is loading
SS
during start-up.
3791f
20
LT3791
APPLICATIONS INFORMATION
Power MOSFET Selections and Efficiency
Considerations
Switch M2 operates in buck operation as the synchronous
rectifier. Its power dissipation at maximum output current
is given by:
TheLT3791requiresfourexternalN-channelpowerMOS-
FETs,twoforthetopswitches(switchM1andM4,shownin
Figure 1) and two for the bottom switches (switch M2 and
M3showninFigure1).Importantparametersforthepower
V – V
IN
OUT
PM2(BUCK)
=
•ILED2 •ρT •RDS(ON)
V
IN
Switch M3 operates in boost operation as the control
switch. Its power dissipation at maximum current is
given by:
MOSFETs are the breakdown voltage, V
, threshold
BR(DSS)
, reverse transfer
voltage, V
, on-resistance, R
GS(TH)
DS(ON)
capacitance, C , and maximum current, I
.
RSS
DS(MAX)
V
OUT – V •V
IN
OUT
(
)
The drive voltage is set by the 5V INTV supply. Con-
CC
PM3(BOOST)
=
•ILED2 •ρT •RDS(ON)
•CROSS •f
2
sequently, logic-level threshold MOSFETs must be used
in LT3791 applications. If the input voltage is expected
to drop below the 5V, then sub-logic threshold MOSFETs
should be considered.
V
IN
ILED
3
+ k •VOUT
•
V
IN
where C
is usually specified by the MOSFET manufac-
RSS
In order to select the power MOSFETs, the power dis-
sipated by the device must be known. For switch M1, the
maximum power dissipation happens in boost operation,
when it remains on all the time. Its maximum power dis-
sipation at maximum output current is given by:
turers. The constant k, which accounts for the loss caused
by reverse-recovery current, is inversely proportional to
the gate drive current and has an empirical value of 1.7.
For switch M4, the maximum power dissipation happens
in boost operation, when its duty cycle is higher than
50ꢀ. Its maximum power dissipation at maximum output
current is given by:
2
ILED •VOUT
PM1(BOOST)
=
•ρT •RDS(ON)
V
IN
2
V
VOUT
ILED •VOUT
where ρ is a normalization factor (unity at 25°C)
T
IN
PM4(BOOST)
=
•
•ρT •RDS(ON)
accounting for the significant variation in on-resistance
withtemperature,typically0.4ꢀ/°CasshowninFigure10.
For a maximum junction temperature of 125°C, using a
V
IN
For the same output voltage and current, switch M1 has
the highest power dissipation and switch M2 has the low-
est power dissipation unless a short occurs at the output.
value of ρ = 1.5 is reasonable.
T
3791f
21
LT3791
APPLICATIONS INFORMATION
From a known power dissipated in the power MOSFET, its
junction temperature can be obtained using the following
formula:
switch M3 turn-on, which improves converter efficiency
and reduces switch M3 voltage stress. In order for the
diode to be effective, the inductance between it and the
synchronousswitchmustbeassmallaspossible,mandat-
ing that these components be placed adjacently.
T = T + P • R
J
A
TH(JA)
The R
to be used in the equation normally includes
TH(JA)
the R
for the device plus the thermal resistance from
INTV Regulator
TH(JC)
CC
the case to the ambient temperature (R
). This value
TH(JC)
An internal P-channel low dropout regulator produces 5V
of T can then be compared to the original, assumed value
J
at the INTV pin from the V supply pin. INTV powers
CC
IN
CC
used in the iterative calculation process.
the drivers and internal circuitry within the LT3791. The
INTV pin regulator can supply a peak current of 67mA
CC
2.0
1.5
1.0
0.5
0
and must be bypassed to ground with a minimum of 4.7µF
ceramic capacitor or low ESR electrolytic capacitor. An
additional0.1µFceramiccapacitorplaceddirectlyadjacent
to the INTV and PGND IC pins is highly recommended.
CC
Good bypassing is necessary to supply the high transient
current required by MOSFET gate drivers.
Higher input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the maxi-
mum junction temperature rating for the LT3791 to be
exceeded.Thesystemsupplycurrentisnormallydominated
by the gate charge current. Additional external loading of
50
100
–50
150
0
JUNCTION TEMPERATURE (°C)
3791 F10
the INTV also needs to be taken into account for the
CC
power dissipation calculations. Power dissipation for the
Figure 10. Normalized RDS(ON) vs Temperature
IC in this case is V • I
, and overall efficiency is
IN
INTVCC
lowered. The junction temperature can be estimated by
Optional Schottky Diode (D3, D4) Selection
using the equations given
The Schottky diodes D3 and D4 shown in the Typical Ap-
plications section conduct during the dead time between
the conduction of the power MOSFET switches. They
are intended to prevent the body diode of synchronous
switches M2 and M4 from turning on and storing charge
duringthedeadtime.Inparticular,D4significantlyreduces
reverse-recovery current between switch M4 turn-off and
T = T + (P • θ )
J
A
D
JA
where θ (in °C/W) is the package thermal impedance.
JA
For example, a typical application operating in continuous
current operation might draw 24mA from a 24V supply:
T = 70°C + 24mA • 24V • 28°C/W = 86°C
J
3791f
22
LT3791
APPLICATIONS INFORMATION
To prevent maximum junction temperature from being
2. Transition loss. This loss arises from the brief amount
of time switch M1 or switch M3 spends in the saturated
region during switch node transitions. It depends upon
the input voltage, load current, driver strength and
MOSFET capacitance, among other factors. The loss
is significant at input voltages above 20V and can be
estimated from:
exceeded, the input supply current must be checked
operating in continuous mode at maximum V .
IN
Top Gate (TG) MOSFET Driver Supply (C1, D1, C±, D±)
The external bootstrap capacitors C1 and C2 connected
to the BST1 and BST2 pins supply the gate drive voltage
for the topside MOSFET switches M1 and M4. When the
top MOSFET switch M1 turns on, the switch node SW1
2
Transition Loss ≈ 2.7 • V • I
• C
• f
IN
OUT
RSS
where C
is the reverse-transfer capacitance.
RSS
rises to V and the BST1 pin rises to approximately V +
IN
IN
INTV .WhenthebottomMOSFETswitchM2turnson,the
3. INTV current. This is the sum of the MOSFET driver
CC
CC
switch node SW1 drops low and the bootstrap capacitor
and control currents.
C1 is charged through D1 from INTV . When the bottom
CC
4. C and C
loss. The input capacitor has the difficult
IN
OUT
MOSFET switch M3 turns on, the switch node SW2 drops
job of filtering the large RMS input current to the regu-
lator in buck operation. The output capacitor has the
difficult job of filtering the large RMS output current
lowandthebootstrapcapacitorC2, ischargedthroughD2
from INTV . The bootstrap capacitors C1 and C2 need to
CC
store about 100 times the gate charge required by the top
MOSFET switch M1 and M4. In most applications a 0.1µF
to 0.47µF, X5R or X7R ceramic capacitor is adequate.
in boost operation. Both C and C
are required to
IN
OUT
2
have low ESR to minimize the AC I R loss and sufficient
capacitance to prevent the RMS current from causing
additional upstream losses in fuses or batteries.
Efficiency Considerations
5. Other losses. Schottky diode D3 and D4 are respon-
sible for conduction losses during dead time and light
load conduction periods. Inductor core loss occurs
predominately at light loads. Switch M3 causes reverse
recovery current loss in boost operation.
The power efficiency of a switching regulator is equal to
the output power divided by the input power times 100ꢀ.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Although all dissipative
elements in circuits produce losses, four main sources
account for most of the losses in LT3791 circuits:
Whenmakingadjustmentstoimproveefficiency,theinput
current is the best indicator of changes in efficiency. If you
make a change and the input current decreases, then the
efficiency has increased. If there is no change in the input
current, then there is no change in efficiency.
2
1. DC I R losses. These arise from the resistances of the
MOSFETs, sensing resistor, inductor and PC board
traces and cause the efficiency to drop at high output
currents.
3791f
23
LT3791
APPLICATIONS INFORMATION
PC Board Layout Checklist
n
The path formed by switch M1, switch M2, D1 and the
capacitor should have short leads and PC trace
C
IN
The basic PC board layout requires a dedicated ground
plane layer. Also, for high current, a multilayer board
provides heat sinking for power components.
lengths. The path formed by switch M3, switch M4, D2
and the C
capacitor also should have short leads
OUT
and PC trace lengths.
n
ThePGNDgroundplanelayershouldnothaveanytraces
n
n
Theoutputcapacitor(–)terminalsshouldbeconnected
as close as possible to the (–) terminals of the input
capacitor.
and it should be as close as possible to the layer with
power MOSFETs.
n
PlaceC , switchM1, switchM2andD1inonecompact
IN
Connect the top driver bootstrap capacitor, C1, closely
to the BST1 and SW1 pins. Connect the top driver
bootstrap capacitor, C2, closely to the BST2 and SW2
pins.
area. Place C , switch M3, switch M4 and D2 in one
OUT
compact area.
n
Useimmediateviastoconnectthecomponents(includ-
ing the LT3791’s SGND and PGND pins) to the ground
plane.Useseverallargeviasforeachpowercomponent.
n
n
n
n
Connecttheinputcapacitors,CIN,andoutputcapacitors,
COUT, closely to the power MOSFETs. These capaci-
tors carry the MOSFET AC current in boost and buck
operation.
n
n
Use planes for V and V
to maintain good voltage
OUT
IN
filtering and to keep power losses low.
Floodallunusedareasonalllayerswithcopper.Flooding
with copper will reduce the temperature rise of power
components. Connect the copper areas to any DC net
Route SNSN and SNSP leads together with minimum
PC trace spacing. Avoid sense lines pass through noisy
areas, such as switch nodes. Ensure accurate current
sensing with Kelvin connections at the SENSE resistor.
(V or PGND).
IN
n
Separatethesignalandpowergrounds.Allsmall-signal
componentsshouldreturntotheSGNDpinatonepoint,
which is then tied to the PGND pin close to the sources
of switch M2 and switch M3.
Connect the VC pin compensation network close to the
IC, between VC and the signal ground pins. The capaci-
tor helps to filter the effects of PCB noise and output
voltage ripple voltage from the compensation loop.
n
n
Place switch M2 and switch M3 as close to the control-
ler as possible, keeping the PGND, BG and SW traces
short.
ConnecttheINTV bypasscapacitor,C ,closetothe
CC
VCC
IC,betweentheINTV andthepowergroundpins.This
CC
capacitorcarriestheMOSFETdrivers’currentpeaks.An
additional 0.1µF ceramic capacitor placed immediately
Keep the high dV/dT SW1, SW2, BST1, BST2, TG1 and
TG2 nodes away from sensitive small-signal nodes.
next to the INTV and PGND pins can help improve
CC
noise performance substantially.
3791f
24
LT3791
TYPICAL APPLICATIONS
98% Efficient 50W (±5V ±A) Buck-Boost LED Driver
V
C
IN
IN
4.7V TO 58V
2.2µF
100V
×4
R
IN
0.003Ω
V
INTV
CC
IN
C
VCC
C3
D1 D2
4.7µF
TEST2
BST2
1µF
R7
50Ω
C2
0.1µF
C1
C
IVINN
IVINP
OUT
BST1
TG1
SWI
BG1
C7
470nF
4.7µF
R5
1M
50V
×4
M1
M2
M4
M3
R1
0.1µF
332k
EN/UVLO
OVLO
L1 10µH
R6
44.2k
R2
121k
R3
INTV
R9
CC
LT3791
1M
SNSP
R10
200k
R
R
LED
SENSE
200k
R4
54.9k
0.05Ω
0.004Ω
SHORTLED
OPENLED
SNSN
PGND
25V LED
2A
PWM
BG2
IVINMON
SW2
TG2
FB
ISMON
CLKOUT
V
REF
C8
0.1µF
ISP
ISN
R11
1M
CTRL
TEST1
SS
PWMOUT
RT SGND
R12
SYNC VC
237k
D1, D2: NXP BAT46WJ
M5
R8
R
L1: COOPER HC9-100-R 10µH
C
86.6k
2.2k
M1, M2: RENESAS RJK0651DPB 60V
M3, M4: RENESAS RJK0451DPB 40V
M5: VISHAY Si2318CDS 40V
DS
DS
3791 TA02a
300kHz
C
10nF
C
C
SS
10nF
DS
Efficiency vs VIN
100Hz 50:1 PWM Dimming (VIN = 1±V)
100
98
96
94
92
90
88
86
84
82
80
PWM
5V/DIV
BOOST
BUCK
BUCK-BOOST
I
L1
2A/DIV
I
LED
2A/DIV
3791 TA02c
50µs/DIV
0
10
30
40
50
60
20
INPUT VOLTAGE (V)
3791 TA02b
3791f
25
LT3791
TYPICAL APPLICATIONS
98% Efficient 60W (1±V 5A) Voltage Regulator Runs Down to 3V VIN
V
IN
C
IN
3V TO 55V
4.7µF
100V
×4
R10
200k
INTV
CC
+
C
C
OUT2
VCC
TEST2
BST2
BST1
D1 D2
4.7µF
100µF
25V
SHORTLED
OPENLED
R
OUT
V
0.015Ω
OUT
12V
5A
D5
D6
C2
0.1µF
C1
0.1µF
V
IN
C
OUT
C3
10µF
25V
×3
TG1
SWI
BG1
M1
M2
M4
1µF
D4
L1 6.8µH
D3
IVINN
M3
IVINP
TEST1
SNSP
LT3791
R
SENSE
0.004Ω
IVINMON
ISMON
CLKOUT
EN/UVLO
OVLO
R5
SNSN
PGND
R1
866k
R3
1M
732k
BG2
R6
80.6k
SW2
TG2
FB
R2
576k
R4
57.6k
PWM
V
REF
ISP
ISN
C8
0.1µF
CTRL
3791 TA03a
R
FAULT
SS SYNC VC
RT
SGND PWMOUT
D1, D2: NXP BAT46WJ
D3: IRF 10BQ060
100k
R
C
R8
86.6k
300kHz
D4: IRF 10BQ040
2.2k
D5, D6: DIODES INC. BAT46W
C
SS
C
C
L1: WURTH ELEKTRONIK WE-HCI 7443556680
10nF
22nF
M1, M2: RENASAS RJK0651DPB 60V
DS
M3, M4: VISHAY SiR424DP 40V
DS
Efficiency vs VIN
Maximum Output Current vs VIN
6
5
4
3
2
1
0
100
I
= 5A
BOOST
OUT
98
96
94
92
90
88
86
84
82
80
BUCK
BUCK-BOOST
3
4
7
8
9
10 20 30 40 50 60
0
10
30
40
50
60
5
6
20
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
3791 TA03c
3791 TA03b
3791f
26
LT3791
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
FE Package
38-Lead Plastic TSSOP (4.4mm)
(Reference LTC DWG # 05-08-1772 Rev C)
Exposed Pad Variation AA
4.75 REF
9.60 – 9.80*
(.378 – .386)
4.75
(.187)
REF
38
20
6.60 0.10
4.50 REF
2.74 REF
SEE NOTE 4
6.40
REF (.252)
BSC
2.74
(.108)
0.315 0.05
1.05 0.10
0.50 BSC
RECOMMENDED SOLDER PAD LAYOUT
1
19
1.20
(.047)
MAX
4.30 – 4.50*
(.169 – .177)
0.25
REF
0° – 8°
0.50
(.0196)
BSC
0.09 – 0.20
(.0035 – .0079)
0.50 – 0.75
(.020 – .030)
0.05 – 0.15
(.002 – .006)
0.17 – 0.27
FE38 (AA) TSSOP REV C 0910
(.0067 – .0106)
TYP
NOTE:
1. CONTROLLING DIMENSION: MILLIMETERS 4. RECOMMENDED MINIMUM PCB METAL SIZE
2. DIMENSIONS ARE IN
FOR EXPOSED PAD ATTACHMENT
MILLIMETERS
(INCHES)
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.150mm (.006") PER SIDE
3. DRAWING NOT TO SCALE
3791f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-
tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.
27
LT3791
TYPICAL APPLICATION
1±0W (±4V 5A) Buck-Boost Voltage Regulator
V
IN
12V TO 58V
+
C
2.2µF
100V
C
47µF
100V
IN
IN2
R10
200k
INTV
CC
C
VCC
TEST2
BST2
BST1
D1 D2
4.7µF
SHORTLED
OPENLED
R
OUT
0.015Ω
V
24V
5A
OUT
C2
0.1µF
C1
0.1µF
V
C
IN
OUT
C3
1µF
4.7µF
50V
×6
TG1
SWI
BG1
M1
M2
M4
M3
L1
10µH
IVINN
IVINP
TEST1
SNSP
LT3791
R
SENSE
0.004Ω
IVINMON
ISMON
CLKOUT
R4
SNSN
PGND
R1
R3
732k
499k
499k
EN/UVLO
OVLO
BG2
R5
18.7k
SW2
TG2
FB
R2
56.2k
R4
27.4k
PWM
V
REF
ISP
ISN
C8
0.1µF
CTRL
3791 TA04
R
FAULT
SS SYNC VC
RT
SGND PWMOUT
D1, D2: NXP BAT46WJ
L1: WURTH ELEKTRONICS 74435571100 10µH
100k
R
C
R8
147k
200kHz
M1, M2: RENESAS RJK0651DPB 60V
M3, M4: RENESAS RJK0451DPB 40V
DS
DS
1.1k
C
C
SS
C
10nF
22nF
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC®3780
High Efficiency, Synchronous, 4-Switch Buck-Boost V : 4V to 36V, V
Range: 0.8V to 30V, I < 55µA, SSOP-24, QFN-32
SD
IN
OUT
Controller
Packages
LTC3789
High Efficiency, Synchronous, 4-Switch Buck-Boost
Controller
V : 4V to 38V, V
Range: 0.8V to 38V, I < 40µA, 4mm × 5mm QFN-28,
IN
OUT
SD
SSOP-28 Packages
LT3755/LT3755-1 High Side 60V, 1MHz LED Controller with True Color V : 4.5V to 40V, V
Range: 5V to 60V, 3000:1 True Color PWM™, Analog,
OUT
IN
LT3755-2
3000:1 PWM Dimming
I
< 1µA, 3mm × 3mm QFN-16, MSOP-16E Packages
SD
LT3756/LT3756-1 High Side 100V, 1MHz LED Controller with True Color V : 6V to 100V, V
Range: 5V to 100V, 3000:1 True Color PWM, Analog,
OUT
IN
LT3756-2
3000:1 PWM Dimming
I
< 1µA, 3mm × 3mm QFN-16, MSOP-16E Packages
SD
LT3596
60V, 300mA Step-Down LED Driver
V : 6V to 60V, V
SD
Range: 5V to 55V, 10000:1 True Color PWM, Analog,
OUT
IN
I
< 1µA, 5mm × 8mm QFN-52 Package
LT3743
Synchronous Step-Down 20A LED Driver with
Thee-State LED Current Control
V : 5.5V to 36V, V
SD
Range: 5.5V to 35V, 3000:1 True Color PWM, Analog,
OUT
IN
I
< 1µA, 4mm × 5mm QFN-28, TSSOP-28E Packages
3791f
LT 0312 • PRINTED IN USA
28 LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
●
●
LINEAR TECHNOLOGY CORPORATION 2012
(408) 432-1900 FAX: (408) 434-0507 www.linear.com
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