LT6109-1_15_技术文档

LT6109-1/LT6109-2

High Side Current Sense

Ampliﬁer with Reference

and Comparators

FEATURES

DESCRIPTION

The LT^®6109 is a complete high side current sense device

that incorporates a precision current sense ampliﬁer, an

integrated voltage reference and two comparators. Two

versions of the LT6109 are available. The LT6109-1 has

the comparators connected in opposing polarity and the

LT6109-2hasthecomparatorsconnectedinthesamepolar-

ity.Inaddition,thecurrentsenseampliﬁerandcomparator

inputs and outputs are directly accessible. The ampliﬁer

gainandcomparatortrippointsareconﬁguredbyexternal

resistors. The open-drain comparator outputs allows for

easy interface to other system components.

n

Current Sense Ampliﬁer

– Fast Step Response: 500ns

– Low Offset Voltage: 125µV Maximum

– Low Gain Error: 0.2% Maximum

n

Internal 400mV Precision Reference

n

Internal Latching Comparators with Reset

– Fast Response Time: 500ns

– Total Threshold Error: 1.25% Maximum

– Two Comparator Polarity Options

n

Wide Supply Range: 2.7V to 60V

n

Supply Current: 550µA

Low Shutdown Current: 5µA Maximum

Speciﬁed for –40°C to 125°C Temperature Range

Available in 10-Lead MSOP Package

n

The overall propagation delay of the LT6109 is typically

only 1.4µs, allowing for quick reaction to overcurrent

and undercurrent conditions. The 1MHz bandwidth al-

lows the LT6109 to be used for error detection in critical

applications such as motor control. The high threshold

accuracy of the comparators, combined with the ability to

latch both comparators, ensures the LT6109 can capture

high speed events.

n

APPLICATIONS

n

Overcurrent, Undercurrent and Fault Detection

n

Current Shunt Measurement

n

Battery Monitoring

Motor Control

Automotive Monitoring and Control

Remote Sensing

Industrial Control

The LT6109 is fully speciﬁed for operation from –40°C to

125°C,makingitsuitableforindustrialandautomotiveap-

plications.TheLT6109isavailableinasmall10-leadMSOP.

L, LT, LTC, LTM, TimerBlox, Linear Technology and the Linear logo are registered trademarks

of Linear Technology Corporation. All other trademarks are the property of their respective

owners.

n

TYPICAL APPLICATION

Circuit Fault Protection with Latching Load Disconnect and Early Warning Indication

Response to Overcurrent Event

0.1Ω

IRF9640

12V

TO LOAD

V

LOAD

0.1µF

6.2V*

10V/DIV

0V

100Ω

1k

3.3V

SENSEHI SENSELO

+

V

OUTA

OUT

I

10k 1.62k 100k

LOAD

LT6109-2

6.04k

2.37k

200mA/DIV

0mA

RESET

EN/RST

INC2

V

OUTC1

5V/DIV

1k

100mA WARNING

250mA DISCONNECT

100mA WARNING

OUTC2

OUTC1

0V

250mA DISCONNECT

V

OUTC2

5V/DIV

2N2700

INC1

–

V

1.6k

610912 TA01a

610912 TA01b

5µs/DIV

*CMH25234B

610912fa

1

LT6109-1/LT6109-2

ABSOLUTE MAXIMUM RATINGS

PIN CONFIGURATION

(Note 1)

+

–

TOP VIEW

Total Supply Voltage (V to V ).................................60V

Maximum Voltage

SENSELO

EN/RST

OUTC2

1

2

3

4

5

10 SENSEHI

+

9

8

7

6

V

+

OUTA

INC2

INC1

(SENSELO, SENSEHI, OUTA)............................... V + 1V

OUTC1

+

–

Maximum V – (SENSELO or SENSEHI)....................33V

V

MS PACKAGE

10-LEAD PLASTIC MSOP

Maximum EN/RST Voltage........................................60V

Maximum Comparator Input Voltage........................60V

Maximum Comparator Output Voltage......................60V

Input Current (Note 2)..........................................–10mA

SENSEHI, SENSELO Input Current ....................... 10mA

DifferentialSENSEHIorSENSELOInputCurrent... 2.5mA

θ

JA

= 160°C/W, θ = 45°C/W

JC

–

AmpliﬁerOutputShort-CircuitDuration(toV ).. Indeﬁnite

Operating Temperature Range (Note 3)

LT6109I................................................–40°C to 85°C

LT6109H ............................................ –40°C to 125°C

Speciﬁed Temperature Range (Note 3)

LT6109I................................................–40°C to 85°C

LT6109H ............................................ –40°C to 125°C

Maximum Junction Temperature .......................... 150°C

Storage Temperature Range .................. –65°C to 150°C

Lead Temperature (Soldering, 10 sec)...................300°C

ORDER INFORMATION

LEAD FREE FINISH

LT6109AIMS-1#PBF

LT6109IMS-1#PBF

LT6109AHMS-1#PBF

LT6109HMS-1#PBF

LT6109AIMS-2#PBF

LT6109IMS-2#PBF

LT6109AHMS-2#PBF

LT6109HMS-2#PBF

TAPE AND REEL

PART MARKING*

LTFNJ

PACKAGE DESCRIPTION

10-Lead Plastic MSOP

SPECIFIED TEMPERATURE RANGE

–40°C to 85°C

LT6109AIMS-1#TRPBF

LT6109IMS-1#TRPBF

LT6109AHMS-1#TRPBF

LT6109HMS-1#TRPBF

LT6109AIMS-2#TRPBF

LT6109IMS-2#TRPBF

LT6109AHMS-2#TRPBF

LT6109HMS-2#TRPBF

LTFNJ

–40°C to 85°C

LTFNJ

–40°C to 125°C

LTFNJ

–40°C to 125°C

LTFWY

–40°C to 85°C

LTFWY

–40°C to 85°C

LTFWY

–40°C to 125°C

LTFWY

–40°C to 125°C

Consult LTC Marketing for parts speciﬁed with wider operating temperature ranges. *The temperature grade is identiﬁed by a label on the shipping container.

Consult LTC Marketing for information on non-standard lead based ﬁnish parts.

For more information on lead free part marking, go to: http://www.linear.com/leadfree/

For more information on tape and reel speciﬁcations, go to: http://www.linear.com/tapeandreel/

610912fa

2

LT6109-1/LT6109-2

ELECTRICAL CHARACTERISTICS ^Thel denotes the speciﬁcations which apply over the full operating

temperature range, otherwise speciﬁcations are at T_A= 25°C. V⁺= 12V, V_PULLUP= V⁺, V_EN/RST= 2.7V, R_IN= 100Ω,

R_OUT= R1 + R2 + R3 = 10k, gain = 100, R_C= 25.5k, C_L= C_LC= 2pF, unless otherwise noted. (See Figure 3)

SYMBOL

PARAMETER

CONDITIONS

MIN

TYP

MAX

UNITS

V

+

l

V

Supply Voltage Range

Supply Current (Note 4)

2.7

60

+

I

V = 2.7V, R = 1k, V

= 5mV

475

600

µA

S

IN

SENSE

+

V = 60V, R = 1k, V

700

1000

µA

IN

SENSE

l

+

Supply Current in Shutdown

V = 2.7V, V

= 0V, R = 1k, V

= 0.5V

3

7

5

7

µA

EN/RST

IN

SENSE

+

V = 60V, V

= 0V, R = 1k, V

11

13

µA

EN/RST

IN

SENSE

+

EN/RST Pin Current

EN/RST Pin Input High

EN/RST Pin Input Low

V

= 0V, V = 60V

–200

nA

V

EN/RST

+

l

V

V = 2.7V to 60V

1.9

IH

IL

+

V = 2.7V to 60V

0.8

V

Current Sense Ampliﬁer

V

Input Offset Voltage

V

SENSE

V

SENSE

V

SENSE

V

SENSE

= 5mV, LT6109A

= 5mV, LT6109

= 5mV, LT6109A

= 5mV, LT6109

–125

–350

–250

–450

125

350

250

450

µV

OS

l

Input Offset Voltage Drift

V

= 5mV

0.8

60

µV/°C

∆V /∆T

SENSE

+

OS

I

Input Bias Current

(SENSELO, SENSEHI)

V = 2.7V to 60V

300

350

nA

B

l

+

I

Input Offset Current

V = 2.7V to 60V

5

nA

OS

l

Output Current (Note 5)

1

mA

OUTA

+

PSRR

Power Supply Rejection Ratio

(Note 6)

V = 2.7V to 60V

120

114

127

dB

+

CMRR

Common Mode Rejection Ratio

V = 36V, V

= 5mV, V

= 2.7V to 36V

= 27V to 60V

125

dB

SENSE

ICM

+

V = 60V, V

110

103

dB

l

V

Full-Scale Input Sense Voltage

(Note 5)

R

= 500Ω

IN

500

mV

SENSE(MAX)

+

Gain Error (Note 7)

V = 2.7V to 12V

–0.08

%

l

V = 12V to 60V, V

= 5mV to 100mV

–0.2

0

SENSE

+

l

SENSELO Voltage (Note 8)

V = 2.7V, V

= 100mV, R

= 100mV

= 2k

OUT

2.5

27

V

SENSE

V = 60V, V

+

l

Output Swing High (V to V

)

V = 2.7V, V

= 27mV

0.2

0.5

V

OUTA

SENSE

+

V = 12V, V

= 120mV

BW

Signal Bandwidth

I

= 1mA

1

MHz

kHz

OUT

+

= 100µA

140

t

Input Step Response (to 50% of

Final Output Voltage)

V = 2.7V, V

V = 12V to 60V, V

= 24mV Step, Output Rising Edge

SENSE

500

ns

r

+

= 100mV Step, Output Rising Edge

SENSE

Settling Time to 1%

V

= 10mV to 100mV, R = 2k

OUT

2

µs

SETTLE

SENSE

610912fa

3

LT6109-1/LT6109-2

ELECTRICAL CHARACTERISTICS ^Thel denotes the speciﬁcations which apply over the full operating

temperature range, otherwise speciﬁcations are at T_A= 25°C. V⁺= 12V, V_PULLUP= V⁺, V_EN/RST= 2.7V, R_IN= 100Ω,

R_OUT= R1 + R2 + R3 = 10k, gain = 100, R_C= 25.5k, C_L= C_LC= 2pF, unless otherwise noted. (See Figure 3)

SYMBOL

PARAMETER

CONDITIONS

MIN

TYP

MAX

UNITS

Reference and Comparator

+

l

V

Rising Input Threshold Voltage

(LT6109-1 Comparator 1

LT6109-2 Both Comparators)

V = 2.7V to 60V, LT6109A

395

392

400

405

408

mV

TH(R)

(Note 9)

V = 2.7V to 60V, LT6109

+

l

V

Falling Input Threshold Voltage

(LT6109-1 Comparator 2)

V = 2.7V to 60V, LT6109A

395

392

400

405

408

mV

TH(F)

(Note 9)

V = 2.7V to 60V, LT6109

+

V

= V

– V

TH(F)

V = 2.7V to 60V

3

10

15

mV

nA

HYS

OL

HYS

TH(R)

+

l

Comparator Input Bias Current

Output Low Voltage

V

= 0V, V = 60V

–50

INC1,2

+

V

I

= 500µA, V = 2.7V

60

150

220

mV

OUTC1,C2

High to Low Propagation Delay

5mV Overdrive

100mV Overdrive

3

0.5

µs

Output Fall Time

Reset Time

0.08

0.5

µs

t

RESET

RPW

l

Valid RST Pulse Width

2

15

Note 1: Stresses beyond those listed under Absolute Maximum Ratings

may cause permanent damage to the device. Exposure to any Absolute

Maximum Rating condition for extended periods may affect device

reliability and lifetime.

Note 5: The full-scale input sense voltage and the maximum output

current must be considered to achieve the speciﬁed performance.

Note 6: Supply voltage and input common mode voltage are varied while

ampliﬁer input offset voltage is monitored.

Note 2: Input and output pins have ESD diodes connected to ground. The

SENSEHI and SENSELO pins have additional current handling capability

speciﬁed as SENSEHI, SENSELO input current.

Note 7: Speciﬁed gain error does not include the effects of external

resistors R and R . Although gain error is only guaranteed between

IN

OUT

+

12V and 60V, similar performance is expected for V < 12V, as well.

Note 3: The LT6109I is guaranteed to meet speciﬁed performance from

–40°C to 85°C. LT6109H is guaranteed to meet speciﬁed performance

from –40°C to 125°C.

Note 4: Supply current is speciﬁed with the comparator outputs high.

When the comparator outputs go low the supply current will increase by

75µA typically per comparator.

Note 8: Refer to SENSELO, SENSEHI Range in the Applications

Information section for more information.

Note 9: The input threshold voltage which causes the output voltage of the

comparator to transition from high to low is speciﬁed. The input voltage

which causes the comparator output to transition from low to high is

the magnitude of the difference between the speciﬁed threshold and the

hysteresis.

610912fa

4

LT6109-1/LT6109-2

Performance characteristics taken at T = 25°C,

TYPICAL PERFORMANCE CHARACTERISTICS

A

V⁺= 12V, V_PULLUP= V⁺, V_EN/RST= 2.7V, R_IN= 100Ω, R_OUT= R1 + R2 + R3 = 10k, gain = 100, R_C= 25.5k, C_L= C_LC= 2pF, unless

otherwise noted. (See Figure 3)

Supply Current vs Supply Voltage

Start-Up Supply Current

Enable/Disable Response

700

600

500

400

300

200

100

0

+

V

5V/DIV

V

EN/RST

2V/DIV

0V

I

S

I

S

500µA/DIV

0µA

40

SUPPLY VOLTAGE (V)

60

0

10

20

30

50

10µs/DIV

100µs/DIV

610912 G02

610912 G01

610912 G03

Input Offset Voltage

vs Temperature

Ampliﬁer Offset Voltage

vs Supply Voltage

Offset Voltage Drift Distribution

300

200

100

0

12

10

8

100

80

5 TYPICAL UNITS

60

40

20

6

0

–20

–40

–60

–80

–100

–200

–300

4

2

0

–2 –1.5 –1 –0.5

OFFSET VOLTAGE DRIFT (µV/°C)

0

0.5

1

1.5

2

0

10

30

40

50

60

–40 –25 –10

5

20 35 50 65 80 95 110 125

TEMPERATURE (°C)

610912 G04

20

SUPPLY VOLTAGE (V)

610912 G06

610912 G05

Ampliﬁer Output Swing

vs Temperature

Ampliﬁer Gain Error

vs Temperature

Ampliﬁer Gain Error Distribution

0.50

0.45

0.40

0.35

0.30

0.25

0.20

0.15

0.10

0.05

0

0.05

0

25

20

V

= 5mV TO 100mV

V

= 5mV TO 100mV

SENSE

+

V

= 12V

R

IN

= 1k

V

= 120mV

SENSE

–0.05

–0.10

–0.15

–0.20

15

R

IN

= 100Ω

10

5

+

V

= 2.7V

V

= 27mV

SENSE

0

50

75 100 125

–50 –25

0

25

50

75 100 125

–0.048 –0.052

–0.064 –0.68

–50

–25

0

25

–0.056 –0.060

TEMPERATURE (°C)

GAIN ERROR (%)

610912 G18

610912 G07

610912 G08

610912fa

5

LT6109-1/LT6109-2

Performance characteristics taken at T = 25°C,

TYPICAL PERFORMANCE CHARACTERISTICS

A

V⁺= 12V, V_PULLUP= V⁺, V_EN/RST= 2.7V, R_IN= 100Ω, R_OUT= R1 + R2 + R3 = 10k, gain = 100, R_C= 25.5k, C_L= C_LC= 2pF, unless

otherwise noted. (See Figure 3)

Common Mode Rejection Ratio

vs Frequency

Power Supply Rejection Ratio

vs Frequency

Ampliﬁer Gain vs Frequency

46

40

34

28

22

16

140

120

100

80

160

140

120

100

80

G = 100

G = 50, R

G = 20, R

I

= 5k

= 2k

OUT

60

OUT

60

40

20

= 1mA

= 100µA

20

OUTA

I

OUTA

0

1k

10k

100k

FREQUENCY (Hz)

1M

10M

1

10 100 1k 10k 100k 1M 10M

FREQUENCY (Hz)

1

10 100 1k 10k 100k 1M 10M

FREQUENCY (Hz)

610912 G11

610912 G10

610912 G09

Ampliﬁer Input Bias Current

vs Temperature

Ampliﬁer Step Response

(V_SENSE= 0mV to 100mV)

System Step Response

100

90

80

70

60

50

40

30

20

10

0

V

SENSE

R

= 100Ω

IN

100mV/DIV

G = 100V/V

0V

V

OUTA

1V/DIV

0V

V

OUTA

SENSEHI

2V/DIV

SENSELO

V

OUTC1

0V

2V/DIV

0V

V

SENSE

V

EN/RST

50mV/DIV

5V/DIV

0V

R

OUT

= 2k,100mV INC1 OVERDRIVE

2µs/DIV

–40 –25 –10

5

20 35 50 65 80 95 110 125

TEMPERATURE (°C)

610912 G13

2µs/DIV

610912 G12

610912 G14

Ampliﬁer Step Response

(V_SENSE= 10mV to 100mV)

Ampliﬁer Step Response

(V_SENSE= 10mV to 100mV)

Ampliﬁer Step Response

(V_SENSE= 0mV to 100mV)

R

= 100Ω

R

= 1k

R

= 1k

IN

OUT

G = 100V/V

R

= 20k

OUT

G = 20V/V

V

OUTA

V

OUTA

1V/DIV

2V/DIV

0V

V

SENSE

V

SENSE

100mV/DIV

0V

100mV/DIV

0V

50mV/DIV

0V

2µs/DIV

610912 G16

610912 G15

610912 G17

610912fa

6

LT6109-1/LT6109-2

Performance characteristics taken at T = 25°C,

TYPICAL PERFORMANCE CHARACTERISTICS

A

V⁺= 12V, V_PULLUP= V⁺, V_EN/RST= 2.7V, R_IN= 100Ω, R_OUT= R1 + R2 + R3 = 10k, gain = 100, R_C= 25.5k, C_L= C_LC= 2pF, unless

otherwise noted. (See Figure 3)

Comparator Threshold

vs Temperature

Comparator Threshold

Distribution

Hysteresis Distribution

30

25

20

15

10

5

408

406

404

402

25

20

5 TYPICAL PARTS

–40°C

25°C

125°C

15

400

398

10

5

396

394

392

0

–10

5

20 35

80 95

3.0

6.2 7.7 9.3 10.9 12.5 14.1 15.7 17.3

4.6

COMPARATOR HYSTERESIS (mV)

–40 –25

110 125

50 65

396

397.6 399.2 400.8 402.8

COMPARATOR THRESHOLD (mV)

404

TEMPERATURE (°C)

610912 G20

610912 G21

610912 G19

EN/RST Current vs Voltage

Hysteresis vs Temperature

Hysteresis vs Supply Voltage

20

18

16

14

12

10

8

50

0

14

12

10

8

5 TYPICAL PARTS

–50

–100

–150

–200

–250

6

4

2

0

–40 –25 –10

5

20 35 50 65 80 95 110 125

TEMPERATURE (°C)

610912 G22

0

20

30

40

50

60

40

60

10

0

10

20

30

50

+

EN/RST VOLTAGE (V)

V

(V)

610912 G24

610912 G23

Comparator Input Bias Current

vs Input Voltage

Comparator Input Bias Current

vs Input Voltage

Comparator Output Low Voltage

vs Output Sink Current

10

5

10

5

1.00

0.75

0.50

0.25

0

125°C

25°C

–40°C

0

–5

–10

–15

–20

–10

–15

–20

125°C

25°C

–40°C

0

20

40

60

0

0.2

0.4

0.6

0.8

1.0

0

1

2

3

I

(mA)

COMPARATOR INPUT VOLTAGE (V)

OUTC

610912 G25

610912 G26

610912 G27

610912fa

7

LT6109-1/LT6109-2

Performance characteristics taken at T = 25°C,

TYPICAL PERFORMANCE CHARACTERISTICS

A

V⁺= 12V, V_PULLUP= V⁺, V_EN/RST= 2.7V, R_IN= 100Ω, R_OUT= R1 + R2 + R3 = 10k, gain = 100, R_C= 25.5k, C_L= C_LC= 2pF, unless

otherwise noted. (See Figure 3)

Comparator Propagation Delay

vs Input Overdrive

Comparator Rise/Fall Time

vs Pull-Up Resistor

Comparator Output Leakage

Current vs Pull-Up Voltage

23

18

13

8

5.0

4.5

4.0

3.5

3.0

2.5

2.0

1.5

1.0

0.5

0

10000

1000

100

V

= 0.9 • V

= 0.1 • V

OH

OL

PULLUP

100mV INC1 OVERDRIVE

C

= 2pF

L

RISE TIME

125°C

H TO L

L TO H

FALL TIME

3

–40°C AND 25°C

–2

10

0

20

30

40

50

60

10

0

40

80

120

160

200

1

10

100

1000

COMPARATOR OUTPUT PULL-UP VOLTAGE (V)

R

PULL-UP RESISTOR (kΩ)

COMPARATOR INPUT OVERDRIVE (mV)

C

610912 G28

610912 G30

610912 G29

Comparator Step Response

(5mV INC1 Overdrive)

Comparator Step Response

(100mV INC1 Overdrive)

Comparator Reset Response

V

INC

0.5V/DIV

0V

0.5V/DIV

0V

V

OUTC

5V/DIV

0V

V

OUTC

V

OUTC

2V/DIV

0V

V

EN/RST

2V/DIV

V

EN/RST

5V/DIV

0V

5µs/DIV

610912 G32

610912 G33

610912 G31

PIN FUNCTIONS

SENSELO (Pin 1): Sense Ampliﬁer Input. This pin must

be tied to the load end of the sense resistor.

OUTC2 (Pin 3): Open-Drain Comparator 2 Output. Off-

–

state voltage may be as high as 60V above V , regardless

+

of V used.

EN/RST (Pin 2): Enable and Latch Reset Input. When the

EN/RSTpinispulledhightheLT6109isenabled. Whenthe

EN/RST pin is pulled low for longer than typically 40µs,

the LT6109 will enter the shutdown mode. Pulsing this pin

low for between 2µs and 15µs will reset the comparators

of the LT6109.

OUTC1 (Pin 4): Open-Drain Comparator 1 Output. Off-

–

state voltage may be as high as 60V above V , regardless

+

of V used.

–

V (Pin 5): Negative Supply Pin. This pin is normally con-

nected to ground.

610912fa

8

LT6109-1/LT6109-2

PIN FUNCTIONS

INC1 (Pin 6): This is the inverting input of comparator 1.

Thesecondinputofthiscomparatorisinternallyconnected

to the 400mV reference.

+

V (Pin 9): Positive Supply Pin. The V pin can be con-

nected directly to either side of the sense resistor, R

.

SENSE

+

When V is tied to the load end of the sense resistor, the

+

SENSEHI pin can go up to 0.2V above V . Supply current

INC2 (Pin 7): This is the input of comparator 2. For the

LT6109-1 this is the noninverting input of comparator 2.

For the LT6109-2 this is the inverting input of compara-

tor 2. The second input of each of these comparators is

internally connected to the 400mV reference.

is drawn through this pin.

SENSEHI (Pin 10): Sense Ampliﬁer Input. The internal

sense ampliﬁer will drive SENSEHI to the same potential

as SENSELO. A resistor (typically R ) tied from supply

IN

to SENSEHI sets the output current, I

where V

= V

/R ,

SENSE

OUT

SENSE IN

OUTA (Pin 8): Current Output of the Sense Ampliﬁer. This

is the voltage developed across R

.

SENSE

pin will source a current that is equal to the sense voltage

divided by the external gain setting resistor, R .

IN

BLOCK DIAGRAMS

9

+

LT6109-1

V

100Ω

34V

6V

3k

SENSEHI

10

–

+

SENSELO

3k

OUTA

8

1

–

V

–

V

+

–

V

200nA

EN/RST

ENABLE AND

RESET TIMING

2

RESET

+

V

INC2

7

+

–

UNDERCURRENT FLAG

OUTC2

3

–

+

V

400mV

REFERENCE

+

–

OVERCURRENT FLAG

OUTC1

4

INC1

6

–

V

5

610912 F01

Figure 1. LT6109-1 Block Diagram (Comparators with Opposing Polarity)

610912fa

9

LT6109-1/LT6109-2

BLOCK DIAGRAMS

LT6109-2

9

+

V

100Ω

34V

6V

–

3k

SENSEHI

10

–

+

SENSELO

1

OUTA

8

7

6

–

V

+

–

V

200nA

EN/RST

ENABLE AND

RESET TIMING

2

RESET

+

V

–

+

INC2

OVERCURRENT FLAG

OUTC2

3

–

+

V

400mV

REFERENCE

+

–

OVERCURRENT FLAG

OUTC1

4

INC1

–

V

5

610912 F02

Figure 2. LT6109-2 Block Diagram (Comparators with the Same Polarity)

APPLICATIONS INFORMATION

The LT6109 high side current sense ampliﬁer provides

accuratemonitoringofcurrentsthroughanexternalsense

resistor. The input sense voltage is level-shifted from the

sensed power supply to a ground referenced output and

is ampliﬁed by a user-selected gain to the output. The

output voltage is directly proportional to the current ﬂow-

ing through the sense resistor.

Ampliﬁer Theory of Operation

An internal sense ampliﬁer loop forces SENSEHI to have

the same potential as SENSELO as shown in Figure 3.

Connecting an external resistor, R , between SENSEHI

IN

and V

forces a potential, V

, across R . A

SUPPLY

corresponding current, I

SENSE IN

, equal to V

/R , will

SENSE IN

OUTA

ﬂow through R . The high impedance inputs of the sense

IN

TheLT6109comparatorshaveathresholdsetwithabuilt-in

400mV precision reference and have 10mV of hysteresis.

The open-drain outputs can be easily used to level shift

to digital supplies.

ampliﬁer do not load this current, so it will ﬂow through

an internal MOSFET to the output pin, OUTA.

610912fa

10

LT6109-1/LT6109-2

APPLICATIONS INFORMATION

The output current can be transformed back into a voltage

voltage be 100mV. If this application is expected to draw

–

by adding a resistor from OUTA to V (typically ground).

2A at peak load, R

should be set to 50mΩ.

SENSE

The output voltage is then:

Once the maximum R

value is determined, the mini-

SENSE

–

V

OUT

= V + I

• R

OUT

mum sense resistor value will be set by the resolution or

dynamic range required. The minimum signal that can be

accuratelyrepresentedbythissenseampliﬁerislimitedby

theinputoffset.Asanexample,theLT6109hasamaximum

input offset of 125µV. If the minimum current is 20mA, a

OUTA

where R

= R1 + R2 + R3 as shown in Figure 3.

OUT

Table 1. Example Gain Conﬁgurations

GAIN

20

R

V

FOR V

= 5V

I

AT V

= 5V

OUT

IN

OUT

SENSE

OUT

OUTA

sense resistor of 6.25mΩ will set V

to 125µV. This is

499Ω

200Ω

100Ω

10k

250mV

100mV

50mV

500µA

SENSE

the same value as the input offset. A larger sense resistor

will reduce the error due to offset by increasing the sense

50

100

voltage for a given load current. Choosing a 50mΩ R

SENSE

Useful Equations

Input Voltage: V_SENSE= I_SENSE•R_SENSE

will maximize the dynamic range and provide a system

that has 100mV across the sense resistor at peak load

(2A), while input offset causes an error equivalent to only

2.5mA of load current.

V_OUT

V_SENSER_IN

R_OUT

Voltage Gain:

Current Gain:

=

In the previous example, the peak dissipation in R

SENSE

I_OUTAR_SENSE

I_SENSE

is 200mW. If a 5mΩ sense resistor is employed, then

the effective current error is 25mA, while the peak sense

voltage is reduced to 10mV at 2A, dissipating only 20mW.

=

R_IN

Note that V

can be exceeded without damag-

SENSE(MAX)

The low offset and corresponding large dynamic range of

theLT6109makeitmoreﬂexiblethanothersolutionsinthis

respect.The125µVmaximumoffsetgives72dBofdynamic

range for a sense voltage that is limited to 500mV max.

ing the ampliﬁer, however, output accuracy will degrade

as V exceeds V , resulting in increased

SENSE

SENSE(MAX)

output current, I

.

OUTA

Selection of External Current Sense Resistor

Theexternalsenseresistor,R ,hasasigniﬁcanteffect

Sense Resistor Connection

SENSE

Kelvin connection of the SENSEHI and SENSELO inputs

to the sense resistor should be used in all but the lowest

power applications. Solder connections and PC board

interconnections that carry high currents can cause sig-

niﬁcant error in measurement due to their relatively large

resistances.One10mm× 10mmsquaretraceof1ozcopper

is approximately 0.5mΩ. A 1mV error can be caused by as

little as 2A ﬂowing through this small interconnect. This

on the function of a current sensing system and must be

chosen with care.

First, the power dissipation in the resistor should be

considered. The measured load current will cause power

dissipation as well as a voltage drop in R

. As a

SENSE

result, the sense resistor should be as small as possible

while still providing the input dynamic range required by

the measurement. Note that the input dynamic range is

the difference between the maximum input signal and the

minimum accurately reproduced signal, and is limited

primarily by input DC offset of the internal sense ampli-

ﬁer of the LT6109. To ensure the speciﬁed performance,

will cause a 1% error for a full-scale V

of 100mV.

SENSE

A 10A load current in the same interconnect will cause

a 5% error for the same 100mV signal. By isolating the

sense traces from the high current paths, this error can

be reduced by orders of magnitude. A sense resistor with

integratedKelvinsenseterminalswillgivethebestresults.

Figure3illustratestherecommendedmethodforconnect-

ing the SENSEHI and SENSELO pins to the sense resistor.

R

should be small enough that V

SENSE(MAX)

example, an application may require the maximum sense

does not

SENSE

exceed V

SENSE

under peak load conditions. As an

610912fa

11

LT6109-1/LT6109-2

APPLICATIONS INFORMATION

V

SUPPLY

+

R

IN

R

SENSE

V

SENSE

LT6109-1

–

SENSEHI 10

1

SENSELO

LOAD

+

–

V

SENSE

R

SENSE

C1

I

=

–

+

SENSE

V

9

+

V

2

3

EN/RST

OUTA 8

INC2 7

V

RESET

V

OUT

V

+

PULLUP

V

I

OUTA

R3*

R2*

C

L

+

–

R

C

OUTC2

UNDERCURRENT

FLAG

C

LC

–

+

V

400mV

REFERENCE

R

C

V

+

–

4

5

OUTC1

OVERCURRENT

FLAG

INC1 6

C

LC

–

V

R1*

–

V

610912 F03

*R

OUT

= R1 + R2 + R3

Figure 3. LT6109-1 Typical Connection

+

Selection of External Input Gain Resistor, R

V

IN

R

should be chosen to allow the required speed and

IN

R

D

SENSE

resolution while limiting the output current to 1mA. The

maximum value for R is 1k to maintain good loop sta-

610912 F04

IN

SENSE

LOAD

bility. For a given V

, larger values of R will lower

IN

power dissipation in the LT6109 due to the reduction

Figure 4. Shunt Diode Limits Maximum Input Voltage to Allow

Better Low Input Resolution Without Overranging

in I

while smaller values of R will result in faster

OUT

IN

response time due to the increase in I . If low sense

OUT

currents must be resolved accurately in a system that has

Care should be taken when designing the board layout for

R , especially for small R values. All trace and inter-

connect resistances will increase the effective R value,

causing a gain error.

a very wide dynamic range, a smaller R may be used

IN

if the maximum I

such as with a Schottky diode across R

current is limited in another way,

OUTA

IN

(Figure 4).

SENSE

This will reduce the high current measurement accuracy

by limiting the result, while increasing the low current

measurement resolution.

The power dissipated in the sense resistor can create a

thermal gradient across a printed circuit board and con-

sequently a gain error if R and R

are placed such

IN

OUT

This approach can be helpful in cases where occasional

bursts of high currents can be ignored.

that they operate at different temperatures. If signiﬁcant

power is being dissipated in the sense resistor then care

610912fa

12

LT6109-1/LT6109-2

APPLICATIONS INFORMATION

should be taken to place R and R

such that the gain

In this case, the only error is due to external resistor

mismatch, which provides an error in gain only. However,

offset voltage, input bias current and ﬁnite gain in the

ampliﬁer can cause additional errors:

IN

OUT

error due to the thermal gradient is minimized.

Selection of External Output Gain Resistor, R

OUT

The output resistor, R , determines how the output cur-

OUT

Output Voltage Error, ∆V

, Due to the Ampliﬁer

OUT(VOS)

rent is converted to voltage. V

is simply I

• R

.

OUT

OUTA

OUT

DC Offset Voltage, V

OS

Typically, R

is a combination of resistors conﬁgured

as a resistor divider which has voltage taps going to the

OUT

R_OUT

R_IN

∆V_OUT(VOS)= V_OS•

comparator inputs to set the comparator thresholds.

In choosing an output resistor, the maximum output volt-

age must ﬁrst be considered. If the subsequent circuit is a

The DC offset voltage of the ampliﬁer adds directly to the

valueofthesensevoltage, V . AsV isincreased,

SENSE

buffer or ADC with limited input range, then R

must be

OUT

accuracyimproves.Thisisthedominanterrorofthesystem

and it limits the available dynamic range.

chosen so that I

• R

is less than the allowed

OUTA(MAX)

OUT

maximum input range of this circuit.

In addition, the output impedance is determined by R

.

Output Voltage Error, ∆V

, Due to the Bias

OUT

OUT(IBIAS)

+

–

If another circuit is being driven, then the input impedance

ofthatcircuitmustbeconsidered.Ifthesubsequentcircuit

has high enough input impedance, then almost any use-

ful output impedance will be acceptable. However, if the

subsequent circuit has relatively low input impedance, or

draws spikes of current such as an ADC load, then a lower

outputimpedancemayberequiredtopreservetheaccuracy

oftheoutput. MoreinformationcanbefoundintheOutput

Filtering section. As an example, if the input impedance of

Currents I and I

B

+

The ampliﬁer bias current I ﬂows into the SENSELO pin

B

–

while I ﬂows into the SENSEHI pin. The error due to I

B

is the following:





R_SENSE

R_IN

∆V_OUT(IBIAS)= R_OUTI_B⁺•

–I_B

–









+

–

Since I ≈ I = I

, if R

<< R then,

SENSE IN

B

BIAS

the driven circuit, R

, is 100 times R , then the

IN(DRIVEN)

OUT

∆V

= –R

(I

)

OUT(IBIAS)

OUT BIAS

accuracy of V

will be reduced by 1% since:

R_OUT•R_IN(DRIVEN)

R_OUT+ R_IN(DRIVEN)

100

OUT

It is useful to refer the error to the input:

∆V = –R (I

V_OUT= I_OUTA

•

)

IN BIAS

VIN(IBIAS)

For instance, if I

is 100nA and R is 1k, the input re-

IN

BIAS

= I_OUTA•R_OUT

•

= 0.99•I_OUTA•R_OUT

ferred error is 100µV. This error becomes less signiﬁcant

101

as the value of R decreases. The bias current error can

IN

+

be reduced if an external resistor, R , is connected as

IN

Ampliﬁer Error Sources

shown in Figure 5, the error is then reduced to:

The current sense system uses an ampliﬁer and resistors

to apply gain and level-shift the result. Consequently, the

output is dependent on the characteristics of the ampliﬁer,

suchasgainerrorandinputoffset, aswellasthematching

of the external resistors.

+

–

V

= R

• I ; I = I – I

OUT(IBIAS)

OUT OS OS B B

Minimizing low current errors will maximize the dynamic

range of the circuit.

Ideally, the circuit output is:

R

R_IN

V_OUT= V_SENSE

•

^OUT; V_SENSE= R_SENSE•I_SENSE

610912fa

13

LT6109-1/LT6109-2

APPLICATIONS INFORMATION

+

V

There is also power dissipated due to the quiescent power

supply current:

9

+

V

LT6109

V

BATT

+

P = I • V

S

R

IN

–

+

10 SENSEHI

The comparator output current ﬂows into the comparator

R

SENSE

–

output pin and out of the V pin. The power dissipated in

OUTA

8

1

SENSELO

V

OUT

+

the LT6109 due to each comparator is often insigniﬁcant

and can be calculated as follows:

R

IN

R

OUT

–

V

5

I

SENSE

610912 F05

–

P

= (V

– V ) • I

OUTC1,C2

Figure 5. R_IN⁺Reduces Error Due to I_B

The total power dissipated is the sum of these

dissipations:

Output Voltage Error, ∆V , Due to

OUT(GAIN ERROR)

P

TOTAL

= P

+ P

+ P + P

OUTC2 S

External Resistors

OUTA

OUTC1

At maximum supply and maximum output currents, the

total power dissipation can exceed 100mW. This will

cause signiﬁcant heating of the LT6109 die. In order to

prevent damage to the LT6109, the maximum expected

dissipation in each application should be calculated. This

The LT6109 exhibits a very low gain error. As a result,

the gain error is only signiﬁcant when low tolerance

resistors are used to set the gain. Note the gain error is

systematically negative. For instance, if 0.1% resistors

are used for R and R

then the resulting worst-case

IN

OUT

number can be multiplied by the θ value, 160°C/W, to

gain error is –0.4% with R = 100Ω. Figure 6 is a graph

JA

IN

ﬁnd the maximum expected die temperature. Proper heat

sinking and thermal relief should be used to ensure that

the die temperature does not exceed the maximum rating.

of the maximum gain error which can be expected versus

the external resistor tolerance.

10

1

Output Filtering

The AC output voltage, V , is simply I

• Z . This

OUT

OUTA

R

IN

= 100Ω

makes ﬁltering straightforward. Any circuit may be used

which generates the required Z to get the desired ﬁlter

OUT

R

= 1k

IN

response. For example, a capacitor in parallel with R

OUT

0.1

will give a lowpass response. This will reduce noise at the

output, and may also be useful as a charge reservoir to

keep the output steady while driving a switching circuit

such as a MUX or ADC. This output capacitor in parallel

0.01

0.1

1

10

RESISTOR TOLERANCE (%)

with R

will create an output pole at:

OUT

610912 F06

Figure 6. Gain Error vs Resistor Tolerance

1

f_–3dB

=

2•π •R_OUT•C_L

Output Current Limitations Due to Power Dissipation

The LT6109 can deliver a continuous current of 1mA to the

SENSELO, SENSEHI Range

The difference between V

OUTA pin. This current ﬂows through R and enters the

+

IN

(see Figure 7) and V , as

SENSE

BATT

current sense ampliﬁer via the SENSEHI pin. The power

dissipated in the LT6109 due to the output signal is:

well as the maximum value of V

, must be considered

to ensure that the SENSELO pin doesn’t exceed the range

listed in the Electrical Characteristics table. The SENSELO

and SENSEHI pins of the LT6109 can function from 0.2V

P

= (V

– V

) • I

OUT

SENSEHI

OUTA

+

Since V

≈ V , P

≈ (V – V

) • I

OUTA OUTA

SENSEHI

OUTA

610912fa

14

LT6109-1/LT6109-2

APPLICATIONS INFORMATION

above the positive supply to 33V below it. These operat-

ing voltages are limited by internal diode clamps shown

in Figures 1 and 2. On supplies less than 35.5V, the lower

60

50

40

–

range is limited by V + 2.5V. This allows the monitored

supply, V

, to be separate from the LT6109 positive

BATT

40.2V

supply as shown in Figure 7. Figure 8 shows the range of

operating voltages for the SENSELO and SENSEHI inputs,

VALID SENSELO/

SENSEHI RANGE

+

30

27

for different supply voltage inputs (V ). The SENSELO and

SENSEHI range has been designed to allow the LT6109 to

monitor its own supply current (in addition to the load),

20.2V

20

as long as V

Figure 9.

is less than 200mV. This is shown in

SENSE

10

2.8V

2.5V

Minimum Output Voltage

2.7

10

20

30 35.5 40

(V)

50

60

610912 F08

+

V

The output of the LT6109 current sense ampliﬁer can

produceanon-zerooutputvoltagewhenthesensevoltage

Figure 8. Allowable SENSELO, SENSEHI Voltage Range

is zero. This is a result of the sense ampliﬁer V being

OS

forced across R as discussed in the Output Voltage Er-

IN

ror, ∆V

section. Figure 10 shows the effect of the

OUT(VOS)

9

+

input offset voltage on the transfer function for parts at

V

LT6109

the V limits. With a negative offset voltage, zero input

OS

V

BATT

sense voltage produces an output voltage. With a positive

offset voltage, the output voltage is zero until the input

sense voltage exceeds the input offset voltage. Neglect-

R

IN

–

+

10 SENSEHI

R

SENSE

OUTA

8

1

SENSELO

V

OUT

ing V , the output circuit is not limited by saturation of

OS

R

OUT

pull-down circuitry and can reach 0V.

–

V

5

I

SENSE

610912 F09

Response Time

The LT6109 ampliﬁer is designed to exhibit fast response

to inputs for the purpose of circuit protection or current

monitoring. This response time will be affected by the

external components in two ways, delay and speed.

Figure 9. LT6109 Supply Current Monitored with Load

120

G = 100

100

80

+

V

9

+

V

= –125µV

OS

V

LT6109

V

60

BATT

R

IN

–

+

10 SENSEHI

40

20

0

V

= 125µV

OS

R

SENSE

OUTA

8

1

SENSELO

V

OUT

R

I

OUT

SENSE

–

V

5

0

100 200 300 400 500 600 700 800 900 1000

INPUT SENSE VOLTAGE (µV)

610912 F07

610912 F10

Figure 7. V⁺Powered Separately from Load Supply (V_BATT

)

Figure 10. Ampliﬁer Output Voltage vs Input Sense Voltage

610912fa

15

LT6109-1/LT6109-2

APPLICATIONS INFORMATION

If the output current is very low and an input transient

occurs, there may be an increased delay before the

outputvoltagebeginstochange. TheTypicalPerformance

Characteristics show that this delay is short and it can

be improved by increasing the minimum output current,

require a separate system or user to reset the outputs. In

applications where the comparator output can intervene

and disconnect loads from the supply, latched outputs are

required to avoid oscillation. Latching outputs are also

useful for detecting problems that are intermittent. The

comparator outputs on the LT6109 are always latching

and there is no way to disable this feature.

either by increasing R

or decreasing R . Note that

SENSE

IN

the Typical Performance Characteristics are labeled with

respect to the initial sense voltage.

Eachofthecomparatorshasoneinputavailableexternally,

with the two versions of the part differing by the polarity

of those available inputs. The other comparator inputs are

connected internally to the 400mV precision reference.

The input threshold (the voltage which causes the output

to transition from high to low) is designed to be equal to

that of the reference. The reference voltage is established

The speed is also affected by the external components.

Using a larger R

OUT OUTA OUT

will decrease the response time, since

OUT

V

= I

• Z

where Z

is the parallel combination

OUT

of R

and any parasitic and/or load capacitance. Note

OUT

that reducing R or increasing R

will both have the

IN

OUT

effect of increasing the voltage gain of the circuit. If the

–

output capacitance is limiting the speed of the system, R

with respect to the device V connection.

IN

and R

can be decreased together in order to maintain

OUT

Comparator Inputs

the desired gain and provide more current to charge the

–

output capacitance.

ThecomparatorinputscanswingfromV to60Vregardless

of the supply voltage used. The input current for inputs

well above the threshold is just a few pAs. With decreas-

ing input voltage, a small bias current begins to be drawn

out of the input near the threshold, reaching 50nA max

when at ground potential. Note that this change in input

bias current can cause a small nonlinearity in the OUTA

transfer function if the comparator inputs are coupled to

the ampliﬁer output with a voltage divider. For example, if

the maximum comparator input current is 50nA, and the

resistance seen looking out of the comparator input is 1k,

then a change in output voltage of 50µV will be seen on the

analog output when the comparator input voltage passes

through its threshold. If both comparator inputs are con-

nected to the output then they must both be considered.

The response time of the comparators is the sum of the

propagation delay and the fall time. The propagation

delay is a function of the overdrive voltage on the input

of the comparators. A larger overdrive will result in a

lower propagation delay. This helps achieve a fast system

response time to fault events. The fall time is affected by

the load on the output of the comparator as well as the

pull-up voltage.

The LT6109 ampliﬁer has a typical response time of 500ns

andthecomparatorshaveatypicalresponsetimeof500ns.

When conﬁgured as a system, the ampliﬁer output drives

the comparator input causing a total system response

time which is typically greater than that implied by the

individually speciﬁed response times. This is due to the

overdrive on the comparator input being determined by

the speed of the ampliﬁer output.

Setting Comparator Thresholds

The comparators have an internal precision 400mV refer-

ence. In order to set the trip points of the LT6109-1 com-

Internal Reference and Comparators

parators, the output currents, I

and I

, as well

OVER

UNDER

, must be calculated:

The integrated precision reference and comparators com-

bined with the high precision current sense allow for rapid

andeasydetectionofabnormalloadcurrents. Thisisoften

critical in systems that require high levels of safety and

reliability. The LT6109 comparators are optimized for fault

detection and are designed with latching outputs. Latch-

ing outputs prevent faults from clearing themselves and

as the maximum output current, I

MAX

V

V_SENSE(UNDER)

I_OVER

=

^SENSE(OVER), I_UNDER

=

,

R_IN

V_SENSE(MAX)

R_IN

I_MAX

=

R_IN

610912fa

16

LT6109-1/LT6109-2

APPLICATIONS INFORMATION

where I

and I

are the over and under currents

OVER

UNDER

400mV

R1=

R2 =

R3 =

through the sense resistor which cause the comparators

I

OVER

to trip. I

resistor.

is the maximum current through the sense

MAX

400mV –I

R1

(

)

UNDER

I

UNDER

Depending on the desired maximum ampliﬁer output volt-

age (V ) the three output resistors, R1, R2 and R3, can

V

–I

R1+ R2

(

)

MAX

MAX MAX

be conﬁgured in two ways. If:

I

MAX

If:













400mV –I_UNDERR1

400mV

I_OVER

( )

V_MAX

>

+

I

MAX

I_UNDER













400mV –I_UNDERR1

400mV

I_OVER

( )

V_MAX

<

+

I

MAX

I_UNDER

then use the conﬁguration shown in Figure 3. The desired

trip points and full-scale analog output voltage for the

circuit in Figure 3 can then be achieved using the follow-

ing equations:

then use the conﬁguration shown in Figure 11.

V

SUPPLY

+

R

IN

R

SENSE

V

SENSE

LT6109-1

–

SENSEHI 10

1

SENSELO

LOAD

+

–

V

SENSE

R

SENSE

C1

I

=

–

+

SENSE

V

9

+

V

2

3

EN/RST

8

7

OUTA

INC2

V

RESET

V

+

PULLUP

V

I

C

OUTA

L

+

–

R

C

OUTC2

UNDERCURRENT

FLAG

C

LC

R3

–

+

V

400mV

REFERENCE

V

R

C

OUT

R2

+

–

4

5

OUTC1

OVERCURRENT

FLAG

INC1

6

C

LC

–

V

R1

–

V

610912 F11

Figure 11. Typical Conﬁguration with Alternative R_OUTConﬁguration

610912fa

17

LT6109-1/LT6109-2

APPLICATIONS INFORMATION

Thedesiredtrippointsandfull-scaleanalogoutputvoltage

for the circuit in Figure 13 can be achieved as follows:

OUTC1

(LT6109-1/LT6109-2)

OUTC2

(LT6109-2)

400mV

OUTC2

(LT6109-1)

INCREASING

R1=

V

INC1,2

I

OVER

610912 F12

V

HYS HYS

V

–I

R1

(

)

MAX MAX

V

TH

R2 =

R3 =

I

MAX

Figure 12. Comparator Output Transfer Characteristics

400mV –I

I

R1+ R2

(

)

UNDER

ing input thresholds, V (the actual internal threshold

UNDER

TH

remains unaffected).

Trip points for the LT6109-2 can be set by replacing I

with a second overcurrent, I

UNDER

.

OVER2

Figure 13 shows how to add additional hysteresis to a

noninverting comparator.

Hysteresis

R6canbecalculatedfromtheextrahysteresisbeingadded,

Each comparator has a typical built-in hysteresis of 10mV

to simplify design, ensure stable operation in the pres-

ence of noise at the inputs, and to reject supply noise that

might be induced by state change load transients. The

hysteresis is designed such that the threshold voltage is

altered when the output is transitioning from low to high

as is shown in Figure 12.

V

and the ampliﬁer output current which you

HYS(EXTRA)

want to cause the comparator output to trip, I

HYS(EXTRA)

to the typical 10mV of built-in hysteresis.

. Note

UNDER

, isinaddition

thatthehysteresisbeingadded, V

400mV – V_HYS(EXTRA)

R6=

I_UNDER

External positive feedback circuitry can be employed

to increase the effective hysteresis if desired, but such

circuitry will have an effect on both the rising and fall-

R1 should be chosen such that R1 >> R6 so that V

OUTA

does not change signiﬁcantly when the comparator trips.

+

V

9

+

V

LT6109-1

+

V

R

IN

–

+

10 SENSEHI

R

SENSE

1

SENSELO

OUTA

INC2

8

7

I

LOAD

+

–

V

+

V

R5

R6

V

R1

+

–

VTH

R3

3

OUTC2

400mV

REFERENCE

–

V

5

R2

610912 F13

Figure 13. Noninverting Comparator with Added Hysteresis

610912fa

18

LT6109-1/LT6109-2

APPLICATIONS INFORMATION

R3 should be chosen to allow sufﬁcient V and compara-

In the previous example, this is an error of 4.3mV at the

output of the ampliﬁer or 43µV at the input of the ampliﬁer

assuming a gain of 100.

OL

tor output rise time due to capacitive loading.

R2 can be calculated:

When using the comparators with their inputs decoupled

fromtheoutputoftheampliﬁer,theymaybedrivendirectly

by a voltage source. It is useful to know the threshold

voltageequationswiththeadditionalhysteresis. Theinput

fallingedgethresholdwhichcausestheoutputtotransition

from high to low is:

R1• V⁺– 400mV – V

•R3

(

)

(

)

HYS(EXTRA)

R2 =

V_HYS(EXTRA)

For very large values of R2 PCB related leakage may

become an issue. A tee network can be implemented to

reduce the required resistor values.

+





1

V •R1

R2+ R3





V_TH(F)= 400mV •R1•

+

–

The approximate total hysteresis will be:













R1 R2+ R3





+





V – 400mV

V_HYS= 10mV + R1•

The input rising edge threshold which causes the output

to transition from low to high is:





R2+ R3





For example, to achieve I

= 100µA with 50mV of

UNDER

1





V_TH(R)= 410mV •R1•

+

total hysteresis, R6 = 3.57k. Choosing R1 = 35.7k, R3 =









R1 R2

+

10k and V = 5V results in R2 = 4.12M.

Figure 14 shows how to add additional hysteresis to an

inverting comparator.

The analog output voltage will also be affected when the

comparator trips due to the current injected into R6 by

the positive feedback. Because of this, it is desirable to

R7canbecalculatedfromtheampliﬁeroutputcurrentwhich

have (R1 + R2 + R3) >> R6. The maximum V

caused by this can be calculated as:

error

OUTA

is required to cause the comparator output to trip, I

.

OVER

400mV

I_OVER

R7=

, Assuming R1+R2 >> R7

(

)

R6





∆V_OUTA= V⁺•









R1+ R2+ R3+ R6

+

V

9

+

V

LT6109-1

SENSEHI

+

V

R

IN

–

+

10

1

R

SENSE

SENSELO

OUTA

8

6

I

LOAD

+

–

V

+

R6

R7

V

R1

–

INC1

VTH

R3

4

OUTC1

+

400mV

REFERENCE

–

V

5

V

DD

R2

610912 F14

Figure 14. Inverting Comparator with Added Hysteresis

610912fa

19

LT6109-1/LT6109-2

APPLICATIONS INFORMATION

To ensure (R1 + R2) >> R7, R1 should be chosen such

The input falling edge threshold which causes the output

to transition from low to high is:

that R1 >> R7 so that V

does not change signiﬁcantly

OUTA

when the comparator trips.

R1

R2

R1

R2









V_TH(F)= 390mV • 1+

–V

DD















R3 should be chosen to allow sufﬁcient V and compara-



OL

tor output rise time due to capacitive loading.

R2 can be calculated:

Comparator Outputs

The comparator outputs can maintain a logic low level of

150mV while sinking 500µA. The outputs can sink higher





V_DD–390mV

V_HYS(EXTRA)

R2 = R1•









currents at elevated V levels as shown in the Typical

OL

PerformanceCharacteristics.Loadcurrentsareconducted

Note that the hysteresis being added, V

, is in

HYS(EXTRA)

–

to the V pin. The output off-state voltage may range

additiontothetypical10mVofbuilt-inhysteresis. Forvery

large values of R2 PCB related leakage may become an

issue. A tee network can be implemented to reduce the

required resistor values.

–

between 0V and 60V with respect to V , regardless of the

supply voltage used. As with any open-drain device, the

outputs may be tied together to implement wire-OR logic

functions. The LT6109-1 can be used as a single-output

window comparator in this way.

The approximate total hysteresis is:

V –390mV





DD

EN/RST Pin

V_HYS= 10mV+ R1•









R2

The EN/RST pin performs the two functions of resetting

the latch on the comparators as well as shutting down the

LT6109. After powering on the LT6109, the comparators

must be reset in order to guarantee a valid state at their

outputs.

For example, to achieve I

= 900µA with 50mV of total

OVER

hysteresis, R7 = 442Ω. Choosing R1 = 4.42k, R3 = 10k

and V = 5V results in R2 = 513k.

DD

The analog output voltage will also be affected when the

comparator trips due to the current injected into R7 by

the positive feedback. Because of this, it is desirable to

have (R1 + R2) >> R7. The maximum V

by this can be calculated as:

Applying a pulse to the EN/RST pin will reset the compara-

tors from their tripped state as long as the input on the

comparator is below the threshold and hysteresis for an

invertingcomparatororabovethethresholdandhysteresis

error caused

OUTA

for a noninverting comparator. For example, if V

is

INC1

R7





pulled higher than 400mV and latches the comparator, a

reset pulse will not reset that comparator unless its input

is held below the threshold by a voltage greater than the

10mVtypicalhysteresis.Thecomparatoroutputstypically

unlatch in 0.5µs with 2pF of capacitive load. Increased

capacitive loading will cause increased unlatch time.

∆V_OUTA= V •









DD

R1+ R2+ R7

In the previous example, this is an error of 4.3mV at the

output of the ampliﬁer or 43µV at the input of the ampliﬁer

assuming a gain of 100.

When using the comparators with their inputs decoupled

fromtheoutputoftheampliﬁertheymaybedrivendirectly

by a voltage source. It is useful to know the threshold

voltage equations with additional hysteresis. The input

rising edge threshold which causes the output to transi-

tion from high to low is:

Figure 15 shows the reset functionality of the EN/RST

pin. The width of the pulse applied to reset the compara-

tors must be greater than t

RPW(MAX)

(2µs) but less than

RPW(MIN)

t

(15µs). Applying a pulse that is longer than

40µs typically (or tying the pin low) will cause the part

to enter shutdown. Once the part has entered shutdown,

the supply current will be reduced to 3µA typically and the

ampliﬁer,comparatorsandreferencewillceasetofunction

610912fa

R1

R2





V_TH(R)= 400mV • 1+









20

LT6109-1/LT6109-2

APPLICATIONS INFORMATION

until the EN/RST pin is transitioned high. When the part

is disabled, both the ampliﬁer and comparator outputs

are high impedance.

on V

. Circuitry connected to OUTA can be protected

OUTA

from these transients by using an external diode to clamp

V

or a capacitor to ﬁlter V

.

OUTA

When the EN/RST pin is transitioned from low to high

to enable the part, the ampliﬁer output PMOS can turn

on momentarily causing typically 1mA of current to ﬂow

into the SENSEHI pin and out of the OUTA pin. Once the

ampliﬁer is fully on, the output will go to the correct cur-

rent. Figure 16 shows this behavior and the impact it has

Power Up

After powering on the LT6109, the comparators must be

reset in order to guarantee a valid state at their outputs.

Fast supply ramps may cause a supply current transient

during start-up as shown in the Typical Performance

Characteristics. This current can be lowered by reducing

the edge speed of the supply.

RESET PULSE WIDTH LIMITS

COMPARATOR

EN/RST

RESET

t

RPW(MIN)

2µs

Reverse-Supply Protection

t

RPW(MAX)

15µs

The LT6109 is not protected internally from external rever-

sal of supply polarity. To prevent damage that may occur

during this condition, a Schottky diode should be added

610912 F15

OUTC1

OUTC2

—

in series with V (Figure 17). This will limit the reverse

t

RESET

0.5µs (TYPICAL)

current through the LT6109. Note that this diode will limit

the low voltage operation of the LT6109 by effectively

Figure 15. Comparator Reset Functionality

reducing the supply voltage to the part by V .

D

+

V

R

= 60V

= 100Ω

Alsonotethatthecomparatorreference,comparatoroutput

IN

OUT

= 10k

–

and EN/RST input are referenced to the V pin. In order to

V

EN/RST

preservetheprecisionofthereferenceandtoavoiddriving

2V/DIV

–

the comparator inputs below V , R2 must connect to the

0V

–

V pin. This will shift the ampliﬁer output voltage up by

V .V

canbeaccuratelymeasureddifferentiallyacross

V

OUTA

D

OUTA

2V/DIV

R1 and R2. The comparator output low voltage will also be

shifted up by V . The EN/RST pin threshold is referenced

D

0V

–

to the V pin. In order to provide valid input levels to the

–

50µs/DIV

LT6109 and avoid driving EN/RST below V the negative

610912 F14

–

supply of the driving circuit should be tied to V .

Figure 16. Ampliﬁer Enable Response

610912fa

21

LT6109-1/LT6109-2

APPLICATIONS INFORMATION

+

V

9

+

V

LT6109-1

+

–

V

R

IN

–

+

10

1

SENSEHI

SENSELO

R

SENSE

OUTA

INC

8

6

V

DD

I

+

LOAD

+

V

R1

R2

DD

R3

–

+

4

2

OUTC

V

OUTA

–

400mV

REFERENCE

V

DD

EN/RST

–

V

5

610912 F17

+

V

D

–

Figure 17. Schottky Prevents Damage During Supply Reversal

TYPICAL APPLICATIONS

Overcurrent and Undervoltage Battery Fault Protection

12 LITHIUM

40V CELL STACK

IRF9640

0.1Ω

TO

LOAD

+

10µF

1M

INC2

100k

6.2V*

R10

100Ω

10

9

1

8

SENSEHI SENSELO

+

13.3k

0.1µF

V

OUTA

V

OUT

0.8A

OVERCURRENT

DETECTION

5V

LT6109-1

+

9.53k

475Ω

6

7

2

4

3

10k

RESET

EN/RST

OUTC1

OUTC2

INC1

INC2

100k

2N7000

30V

–

V

UNDERVOLTAGE

DETECTION

5

6109 TA02

*CMH25234B

The comparators monitor for overcurrent and undervolt-

age conditions. If either fault condition is detected the

battery will immediately be disconnected from the load.

The latching comparator outputs ensure the battery stays

disconnected from the load until an outside source resets

the LT6109 comparator outputs.

610912fa

22

LT6109-1/LT6109-2

TYPICAL APPLICATIONS

MCU Interfacing with Hardware Interupts

0.1Ω

+

V

TO LOAD

Example:

5V

100Ω

OUTC2 GOES LOW

0V

10

1

8

SENSEHI SENSELO

+

9

V

OUT

V

OUTA

ADC IN

AtMega1280

5V

MCU INTERUPT

LT6109-1

5

2k

PB0

7

6

2

3

4

RESET

6

7

2

3

1

10k

EN/RST

OUTC2

OUTC1

INC2

INC1

PB1

PCINT2

PCINT3

ADC2

6.65k

UNDERCURRENT ROUTINE

5V

–

V

/ADC IN

1.33k

OUT

5

PB5

RESET COMPARATORS

6109 TA03

610912 TA03b

The comparators are set to have a 50mA undercurrent

threshold and a 300mA overcurrent threshold. The MCU

willreceivethecomparatoroutputsashardwareinterrupts

and immediately run an appropriate fault routine.

Simpliﬁed DC Motor Torque Control

V

MOTOR

100µF

1k

0.1Ω

SENSEHI SENSELO

+

CURRENT SET POINT (0V TO 5V)

BRUSHED

DC MOTOR

(0A TO 5A)

MABUCHI

RS-540SH

V

OUTA

V

OUT

1µF

0.47µF

1N5818

100k

LT6109

5.62k

3.4k

1k

5V

RESET

EN/RST

OUTC2

OUTC1

INC2

INC1

5

–

2

4

7

+

V

1

3

6

MOD OUT

IRF640

³+

–

LTC6246

V

LTC6992-1

100k

4

78.7k

SET DIV

GND

5V

280k

1M

2

610912 TA04

The ﬁgure shows a simpliﬁed DC motor control circuit.

The circuit controls motor current, which is proportional

to motor torque; the LT6109 is used to provide current

feedback to a difference ampliﬁer that controls the current

in the motor. The LTC^®6992 is used to convert the output

of the difference amp to the motors PWM control signal.

610912fa

23

LT6109-1/LT6109-2

TYPICAL APPLICATIONS

Power-On Reset or Disconnect Using a TimerBlox^®Circuit

5V

9

+

V

LT6109-1

+

R

V

IN

100Ω

10 SENSEHI

–

+

R

SENSE

1

SENSELO

OUTA

INC2

8

7

I

–

+

LOAD

V

R1

8.06k

R5

10k

+

–

3

OUTC2

–

+

R4

10k

V

400mV

REFERENCE

R2

1.5k

5V

+

–

4

2

OUTC1

R8

CREATES A DELAYED

10µs RESET PULSE

ON START-UP

30k

C1

INC1

6

Q1

0.1µF

2N2222

EN/RST

R3

499Ω

TRIG

GND

SET

OUT

OPTIONAL:

LTC6993-3

–

R7

1M

V

DISCHARGES C1

WHEN SUPPLY

+

V

610912 TA06

5

IS DISCONNECTED

DIV

R6

487k

TheLTC6993-1providesa10µSresetpulsetotheLT6109-1.

TheresetpulseisdelayedbyR7andC1whosetimeconstant

must be greater than 10ms and longer than the supply

turn-on time. Optional components R8 and Q1 discharge

capacitor C1 when the supply and/or ground are discon-

nected. This ensures that when the power supply and/or

ground are restored, capacitor C1 can fully recharge and

triggertheLTC6993-3toproduceanothercomparatorreset

pulse. These optional components are particularly useful

if the power and/or ground connections are intermittent,

as can occur when PCB are plugged into a connector.

610912fa

24

LT6109-1/LT6109-2

TYPICAL APPLICATIONS

Precision Power-On Reset Using a TimerBlox^®Circuit

5V

9

+

V

LT6109-1

+

R

V

IN

100Ω

10 SENSEHI

–

+

R

SENSE

1

SENSELO

OUTA

INC2

8

7

I

–

+

LOAD

V

R1

8.06k

R5

10k

+

–

3

OUTC2

–

+

R4

10k

V

400mV

REFERENCE

R2

1.5k

R8

+

–

100k

4

2

OUTC1

1 SECOND DELAY

ON START-UP

10µs RESET PULSE

GENERATOR

INC1

6

EN/RST

R3

TRIG

OUT

TRIG

OUT

499Ω

C1

0.1µF

LTC6994-1

LTC6993-1

+

GND

V

GND

V

–

V

C2

0.1µF

R6

1M

610912 TA07

5

SET

DIV

SET

DIV

R7

191k

R5

681k

R4

487k

610912fa

25

LT6109-1/LT6109-2

PACKAGE DESCRIPTION

MS Package

10-Lead Plastic MSOP

(Reference LTC DWG # 05-08-1661 Rev E)

0.889 ± 0.127

(.035 ± .005)

5.23

(.206)

MIN

3.20 – 3.45

(.126 – .136)

3.00 ± 0.102

(.118 ± .004)

(NOTE 3)

0.497 ± 0.076

(.0196 ± .003)

REF

0.50

0.305 ± 0.038

(.0120 ± .0015)

TYP

(.0197)

10 9

8

7 6

BSC

RECOMMENDED SOLDER PAD LAYOUT

3.00 ± 0.102

(.118 ± .004)

(NOTE 4)

4.90 ± 0.152

(.193 ± .006)

DETAIL “A”

0.254

(.010)

0° – 6° TYP

GAUGE PLANE

1

2

3

4 5

0.53 ± 0.152

(.021 ± .006)

0.86

(.034)

REF

1.10

(.043)

MAX

DETAIL “A”

0.18

(.007)

SEATING

PLANE

0.17 – 0.27

(.007 – .011)

TYP

0.1016 ± 0.0508

(.004 ± .002)

0.50

(.0197)

BSC

MSOP (MS) 0307 REV E

NOTE:

1. DIMENSIONS IN MILLIMETER/(INCH)

2. DRAWING NOT TO SCALE

3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.

MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE

4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.

INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE

5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX

610912fa

26

LT6109-1/LT6109-2

REVISION HISTORY

REV

DATE

DESCRIPTION

PAGE NUMBER

A

12/12 Addition of A-grade Performance and Electrical Characteristics

Correction to Typical Application diagram

1, 3, 4, 11, 13, 15 (Fig10), 28

1

Addition of A-grade Order Information

2

Clariﬁcation to Absolute Maximum Short Circuit Duration

Edits to Electrical Characteristics conditions and notes

Clariﬁcation to nomenclature used in Typical Performance Characteristics

Clariﬁcation to Description of Pin Functions

2

3, 4

5-8

8, 9

9, 10, 12, 17, 18, 19, 25, 26

10-16, 18, 20-25

28

Internal Reference Block redrawn for consistency

Edits to Applications Information

Addition of LT6108 to Related Parts

610912fa

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-

tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.

27

LT6109-1/LT6109-2

TYPICAL APPLICATION

ADC Driving Application

SENSE

HIGH

SENSE

LOW

0.1Ω

0.1µF

IN

OUT

V

REF

CC

100Ω

10

COMP

1

8

SENSEHI SENSELO

9

+

V

OUTA

IN

LTC2470

TO

MCU

0.1µF

V

CC

LT6109-1

2k

V

CC

7

6

2

3

4

RESET

EN/RST

OUTC2

OUTC1

INC2

INC1

10k

6.65k

1.33k

–

V

5

OVERCURRENT

6109 TA05

UNDERCURRENT

The low sampling current of the LTC2470 16-bit delta

sigma ADC is ideal for the LT6109.

RELATED PARTS

PART NUMBER DESCRIPTION

COMMENTS

LT1787

LTC4150

LT6100

LTC6101

LTC6102

LTC6103

LTC6104

LT6105

LT6106

LT6107

LT6108

Bidirectional High Side Current Sense Ampliﬁer

Coulomb Counter/Battery Gas Gauge

2.7V to 60V, 75µV Offset, 60µA Quiescent, 8V/V Gain

Indicates Charge Quantity and Polarity

Gain-Selectable High Side Current Sense Ampliﬁer

High Voltage High Side Current Sense Ampliﬁer

Zero Drift High Side Current Sense Ampliﬁer

Dual High Side Current Sense Ampliﬁer

4.1V to 48V, Gain Settings: 10, 12.5, 20, 25, 40, 50V/V

Up to 100V, Resistor Set Gain, 300µV Offset, SOT-23

Up to 100V, Resistor Set Gain, 10µV Offset, MSOP8/DFN

4V to 60V, Resistor Set Gain, 2 Independent Amps, MSOP8

4V to 60V, Separate Gain Control for Each Direction, MSOP8

–0.3V to 44V Input Range, 300µV Offset, 1% Gain Error

2.7V to 36V, 250µV Offset, Resistor Set Gain, SOT-23

2.7V to 36V, –55°C to 150°C, Fully Tested: –55°C, 25°C, 150°C

Bidirectional High Side Current Sense Ampliﬁer

Precision Rail-to-Rail Input Current Sense Amplifer

Low Cost High Side Current Sense Ampliﬁer

High Temperature High Side Current Sense Ampliﬁer

High Side Current Sense Ampliﬁer with Reference and

Comparator

2.7V to 60V, 125µV Offset, Resistor Set Gain, 1.25% Threshold

Error

LT6700

Dual Comparator with 400mV Reference

1.4V to 18V, 6.5µA Supply Current

610912fa

LT 1212 REV A • PRINTED IN USA

28 ^{LinearTechnology Corporation}

1630 McCarthy Blvd., Milpitas, CA 95035-7417

●

 LINEAR TECHNOLOGY CORPORATION 2011

(408) 432-1900 FAX: (408) 434-0507 www.linear.com


型号：	LT6109-1_15
厂家：	Linear
描述：	High Side Current Sense Amplifier with Reference and Comparators
文件：	总28页 (文件大小：323K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

LT6109-1_15 [Linear]

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