LTC1435IG#TR [Linear]
LTC1435 - High Efficiency Low Noise Synchronous Step-Down Switching Regulator; Package: SSOP; Pins: 16; Temperature Range: -40°C to 85°C;
| 型号： | LTC1435IG#TR |
| 厂家： | 
Linear |
| 描述： | LTC1435 - High Efficiency Low Noise Synchronous Step-Down Switching Regulator; Package: SSOP; Pins: 16; Temperature Range: -40°C to 85°C 光电二极管 |
| 文件： | 总20页 (文件大小：333K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LTC1435
High Efficiency Low Noise
Synchronous Step-Down
Switching Regulator
U
FEATURES
DESCRIPTION
The LTC®1435 is a synchronous step-down switching
regulator controller that drives external N-channel power
MOSFETs using a fixed frequency architecture. Burst
ModeTM operation provides high efficiency at low load
currents.Amaximumdutycyclelimitof99%provideslow
dropout operation which extends operating time in bat-
tery-operated systems.
■
Dual N-Channel MOSFET Synchronous Drive
■
Programmable Fixed Frequency
■
Wide VIN Range: 3.5V to 36V Operation
■
Ultrahigh Efficiency
■
Very Low Dropout Operation: 99% Duty Cycle
■
Low Standby Current
■
Secondary Feedback Control
■
Programmable Soft Start
The operating frequency is set by an external capacitor
allowing maximum flexibility in optimizing efficiency. A
secondary winding feedback control pin, SFB, guarantees
regulation regardless of load on the main output by
forcing continuous operation. Burst Mode operation is
inhibited when the SFB pin is pulled low which reduces
noise and RF interference.
■
Remote Output Voltage Sense
■
Logic Controlled Micropower Shutdown: IQ < 25µA
■
Foldback Current Limiting (Optional)
■
Current Mode Operation for Excellent Line and Load
Transient Response
Output Voltages from 1.19V to 9V
■
■
Available in 16-LUead Narrow SO and SSOP Packages
Soft start is provided by an external capacitor which can
be used to properly sequence supplies. The operating
currentlevelisuser-programmableviaanexternalcurrent
sense resistor. Wide input supply range allows operation
from 3.5V to 30V (36V maximum).
APPLICATIONS
■
Notebook and Palmtop Computers, PDAs
Cellular Telephones and Wireless Modems
Portable Instruments
Battery-Operated Devices
■
■
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode is a trademark of Linear Technology Corporation.
■
■
DC Power Distribution Systems
U
TYPICAL APPLICATION
V
IN
4.5V TO 28V
C
V
IN
OSC
C
IN
+
C
OSC
22µF
35V
× 2
M1
Si4412DY
RUN/SS
TG
68pF
C
R
SS
0.1µF
SENSE
0.033Ω
V
OUT
I
SW
TH
2.9V/3.5A
L1
C
D
C
B
LTC1435
INTV
10µH
330pF
CMDSH-3
R1
C
C
R
OUT
B
C
32.4k
CC
+
100µF
10V
0.1µF
10k
SGND
BOOST
R2
22.1k
× 2
D1
MBRS140T3
+
100pF
4.7µF
M2
Si4412DY
V
BG
OSENSE
PGND
SENSE
–
+
SENSE
1000pF
1435 F01
Figure 1. High Efficiency Step-Down Converter
1
LTC1435
W W U W
U
W U
ABSOLUTE MAXIMUM RATINGS
PACKAGE/ORDER INFORMATION
Input Supply Voltage (VIN).........................36V to –0.3V
Topside Driver Supply Voltage (Boost)......42V to –0.3V
Switch Voltage (SW)............................. VIN + 5V to –5V
EXTVCC Voltage ........................................ 10V to –0.3V
Sense+, Sense– Voltages ......... INTVCC + 0.3V to –0.3V
ITH, VOSENSE Voltages .............................. 2.7V to –0.3V
SFB, Run/SS Voltages .............................. 10V to –0.3V
Peak Driver Output Current < 10µs (TG, BG) ............. 2A
INTVCC Output Current ........................................ 50mA
Operating Ambient Temperature Range
LTC1435C............................................... 0°C to 70°C
LTC1435I............................................ –40°C to 85°C
Junction Temperature (Note 1)............................. 125°C
Storage Temperature Range ................. –65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
ORDER PART
TOP VIEW
NUMBER
C
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
TG
OSC
RUN/SS
BOOST
SW
LTC1435CG
LTC1435CS
LTC1435IG
LTC1435IS
I
TH
SFB
V
IN
SGND
INTV
BG
CC
V
OSENSE
–
SENSE
PGND
EXTV
+
SENSE
CC
G PACKAGE
S PACKAGE
16-LEAD PLASTIC SSOP 16-LEAD PLASTIC SO
TJMAX = 125°C, θJA = 130°C/ W (G)
TJMAX = 125°C, θJA = 110°C/ W (S)
Consult factory for Military grade parts.
TA = 25°C, VIN = 15V, VRUN/SS = 5V unless otherwise noted.
ELECTRICAL CHARACTERISTICS
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Main Control Loop
I
V
∆V
∆V
V
Feedback Current
Feedback Voltage
Reference Voltage Line Regulation
Output Voltage Load Regulation
(Note 2)
(Note 2)
10
1.19
0.002
0.5
–0.5
50
1.202
0.01
0.8
–0.8
nA
V
%/V
%
%
IN OSENSE
●
1.178
OSENSE
V
= 3.6V to 20V (Note 2)
LINEREG
IN
I
I
Sinking 5µA (Note 2)
Sourcing 5µA
●
●
LOADREG
TH
TH
V
Secondary Feedback Threshold
Secondary Feedback Current
Output Overvoltage Lockout
V
V
Ramping Negative
= 1.5V
●
1.16
1.24
1.19
–1
1.28
1.22
–2
1.32
V
µA
V
SFB
SFB
SFB
I
SFB
V
OVL
I
Input DC Supply Current
Normal Mode
Shutdown
Run Pin Threshold
Soft Start Current Source
Maximum Current Sense Threshold
EXTV = 5V (Note 3)
CC
Q
3.6V < V < 30V
260
16
1.3
3
µA
µA
V
µA
IN
V
= 0V, 3.6V < V < 15V
25
2
4.5
180
RUN/SS
IN
V
I
∆V
●
0.8
1.5
130
RUN/SS
V
V
= 0V
= 0V, 5V
RUN/SS
RUN/SS
OSENSE
150
mV
SENSE(MAX)
TG Transition Time
Rise Time
TG t
TG t
C
C
= 3000pF
= 3000pF
50
50
150
150
ns
ns
r
f
LOAD
LOAD
Fall Time
BG Transition Time
Rise Time
BG t
BG t
C
C
= 3000pF
= 3000pF
50
40
150
150
ns
ns
r
f
LOAD
LOAD
Fall Time
Internal V Regulator
CC
V
V
V
V
Internal V Voltage
6V < V < 30V, V = 4V
EXTVCC
●
●
4.8
5.0
–0.2
130
4.7
5.2
–1
230
V
%
mV
V
INTVCC
CC
IN
INT
EXT
INTV Load Regulation
I
I
I
= 15mA, V
= 15mA, V
= 15mA, V
= 4V
= 5V
Ramping Positive
LDO
LDO
CC
INTVCC
INTVCC
INTVCC
EXTVCC
EXTVCC
EXTVCC
EXTV Voltage Drop
CC
EXTV Switchover Voltage
4.5
EXTVCC
CC
Oscillator
f
Oscillator Frequency
C
= 100pF (Note 4)
OSC
112
125
138
kHz
OSC
2
LTC1435
TA = 25°C, VIN = 15V, VRUN/SS = 5V unless otherwise noted.
ELECTRICAL CHARACTERISTICS
The
temperature range.
LTC1435CG/LTC1435CS: 0°C ≤ T ≤ 70°C
●
denotes specifications which apply over the full operating
Note 2: The LTC1435 is tested in a feedback loop which servos V
OSENSE
to the balance point for the error amplifier (V = 1.19V).
ITH
A
Note 3: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications Information.
LTC1435IG/LTC1435IS: –40°C ≤ T ≤ 85°C
A
Note 1: T is calculated from the ambient temperature T and power
J
A
Note 4: Oscillator frequency is tested by measuring the C
discharge currents and applying the formula:
charge and
OSC
dissipation P according to the following formula:
D
LTC1435CG/LTC1435IG: T = T + (P )(130°C/W)
8.4(108)
J
A
D
–1
1
1
+
f (kHz) =
OSC
LTC1435CS/LTC1435IS: T = T + (P )(110°C/W)
J
A
D
(
C
) (
I
)
(pF) + 11
I
OSC
CHG DIS
W
U
TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency vs Input Voltage
VOUT = 3.3V
Efficiency vs Input Voltage
VOUT = 5V
Efficiency vs Load Current
100
95
90
85
80
75
70
65
60
55
50
100
95
90
85
80
75
70
100
95
90
85
80
75
70
V
V
= 10V
IN
V
OUT
= 3.3V
V
= 5V
OUT
= 5V
OUT
R
= 0.033Ω
SENSE
I
= 1A
LOAD
I
= 1A
LOAD
I
= 100mA
LOAD
CONTINUOUS
MODE
I
= 100mA
Burst Mode
OPERATION
LOAD
0
10
15
20
25
30
0.001
0.01
0.1
1
10
0
10
15
20
25
30
5
5
INPUT VOLTAGE (V)
LOAD CURRENT (A)
INPUT VOLTAGE (V)
1435 G02
1435 G03
1435 G01
VIN – VOUT Dropout Voltage
vs Load Current
Load Regulation
VITH Pin Voltage vs Output Current
0
–0.25
–0.50
–0.75
–1.00
–1.25
–1.50
3.0
2.5
0.5
0.4
0.3
0.2
0.1
R
= 0.033Ω
R
OUT
= 0.033Ω
SENSE
SENSE
V
DROP OF 5%
2.0
1.5
1.0
0.5
0
Burst Mode
OPERATION
CONTINUOUS
MODE
0
0
1.0
1.5
2.0
2.5
3.0
0.5
0
10 20 30 40 50 60 70 80 90 100
OUTPUT CURRENT (%)
0
0.5
1.0
1.5
2.0
2.5
3.0
LOAD CURRENT (A)
LOAD CURRENT (A)
1435 G05
1435 G06
1435 G04
3
LTC1435
TYPICAL PERFORMANCE CHARACTERISTICS
W
U
EXTVCC Switch Drop
vs INTVCC Load Current
Input Supply and Shutdown
Current vs Input Voltage
INTVCC Regulation
vs INTVCC Load Current
2.5
2.0
1.5
1.0
100
80
200
180
160
140
120
100
80
0.5
0.3
V
= 0V
EXTVCC
70°C
V
= 5V
OUT
25°C
60
70°C
25°C
EXTV = V
CC
OUT
0
–55°C
V
OUT
= 3.3V
40
EXTV = OPEN
CC
60
–0.3
–0.5
40
0.5
0
20
0
20
SHUTDOWN
10
INPUT VOLTAGE (V)
0
0
15
20
25
30
0
2
4
6
12 14 16 18 20
5
10
INTV LOAD CURRENT (mA)
8
10
0
15
20
5
INTV LOAD CURRENT (mA)
CC
CC
1435 G07
1435 G09
1435 G08
Normalized Oscillator Frequency
vs Temperature
RUN/SS Pin Current
vs Temperature
SFB Pin Current vs Temperature
10
5
0
–0.25
–0.50
–0.75
4
3
2
1
f
O
–1.00
–1.25
–1.50
–5
–10
0
60
TEMPERATURE (°C)
110 135
–40 –15 10
35
60
85 110 135
60
TEMPERATURE (°C)
110 135
–40 –15
10
35
85
–40 –15
10
35
85
TEMPERATURE (°C)
1435 G10
1435 G11
1435 G12
Maximum Current Sense
Threshold Voltage vs Temperature
Transient Response
Transient Response
154
152
150
148
VOUT
50mV/DIV
VOUT
50mV/DIV
ILOAD = 1A to 3A
1435 G15
I
LOAD = 50mA to 1A
1435 G14
146
–40 –15 10
35
60
85 110 135
TEMPERATURE (°C)
1435 G13
4
LTC1435
W
U
TYPICAL PERFORMANCE CHARACTERISTICS
Soft Start: Load Current vs Time
Burst Mode Operation
VOUT
20mV/DIV
RUN/SS
5V/DIV
INDUCTOR
CURRENT
1A/DIV
VITH
200mV/DIV
1435 G17
ILOAD = 50mA
1435 G16
U
U
U
PIN FUNCTIONS
ever EXTVCC is higher than 4.7V. See EXTVCC connection
in Applications Information section. Do notexceed10V on
this pin. Connect to VOUT if VOUT ≥ 5V.
COSC (Pin 1): External capacitor COSC from this pin to
ground sets the operating frequency.
RUN/SS (Pin 2): Combination of Soft Start and Run
Control Inputs. A capacitor to ground at this pin sets the
ramp timeto fullcurrentoutput. The timeis approximately
0.5s/µF. Forcing this pin below 1.3V causes the device to
be shut down. In shutdown all functions are disabled.
PGND (Pin 10): Driver Power Ground. Connects to source
of bottom N-channel MOSFET and the (–) terminal of CIN.
BG (Pin 11): High Current Gate Drive for Bottom
N-Channel MOSFET. Voltage swing at this pin is from
ground to INTVCC.
ITH (Pin 3): Error Amplifier Compensation Point. The
current comparator threshold increases with this control
voltage. Nominal voltage range for this pin is 0V to 2.5V.
INTVCC (Pin 12): Output of the Internal 5V Regulator and
EXTVCC Switch. The driver and control circuits are pow-
ered from this voltage. Must be closely decoupled to power
ground with a minimum of 2.2µF tantalum or electrolytic
capacitor.
SFB (Pin 4): Secondary Winding Feedback Input. Nor-
mally connected to a feedback resistive divider from the
secondary winding. This pin should be tied to: ground to
force continuous operation; INTVCC in applications that
don’tuseasecondarywinding;andaresistivedividerfrom
the output in applications using a secondary winding.
VIN (Pin 13): Main Supply Pin. Must be closely decoupled
to the IC’s signal ground pin.
SW (Pin 14): Switch Node Connection to Inductor. Volt-
age swing at this pin is from a Schottky diode (external)
voltage drop below ground to VIN.
SGND (Pin 5): Small-Signal Ground. Must be routed
separately from other grounds to the (–) terminal of COUT
.
VOSENSE (Pin 6): Receives the feedback voltage from an
BOOST (Pin 15): Supply to Topside Floating Driver. The
bootstrap capacitor is returned to this pin. Voltage swing
at this pin is from INTVCC to VIN + INTVCC.
external resistive divider across the output.
SENSE– (Pin 7): The (–) Input to the Current Comparator.
SENSE+ (Pin 8): The (+) Input to the Current Comparator.
Built-in offsets between SENSE– and SENSE+ pins in
conjunction with RSENSE set the current trip thresholds.
TG (Pin 16): High Current Gate Drive for Top N-Channel
MOSFET. This is the output of a floating driver with a
voltage swing equal to INTVCC superimposed on the
switch node voltage SW.
EXTVCC (Pin 9): Input to the Internal Switch Connected to
INTVCC. This switch closes and supplies VCC power when-
5
LTC1435
U
U W
FUNCTIONAL DIAGRA
V
IN
C
OSC
+
C
IN
1
C
OSC
SFB
13
V
IN
SGND 5
4
INTV
CC
1.19V
REF
D
B
1µA
BOOST
15
–
C
B
1.19V
+
TG
16
SHUTDOWN
OSC
DROP
OUT
DET
OV
+
S
R
Q
SWITCH
LOGIC
1.28V
–
0.6V
+
–
SW
14
V
SEC
V
OSENSE
I2
V
FB
6
–
+
–
+
–
+
D1
I1
EA
R2
4k
1.19V
Ω
g
= 1m
180k
m
+
V
IN
INTV
CC
C
SEC
+
–
INTV
CC
12
+
SHUTDOWN
5V
LDO
REG
3µA
R1
RUN
SOFT
START
4.8V
+
–
BG
11
30k
8k
V
OUT
6V
+
C
OUT
PGND
10
R
C
+
–
2
8
7
9
EXTV
RUN/SS
3
SENSE
SENSE
I
TH
CC
C
SS
C
C
D
FB
*
R
SENSE
1435 • FD
* FOLDBACK CURRENT LIMITING OPTION
6
LTC1435
U
(Refer to Functional Diagram)
OPERATION
Main Control Loop
Low Current Operation
The LTC1435 uses a constant frequency, current mode
step-down architecture. During normal operation, the top
MOSFET is turned on each cycle when the oscillator sets
the RS latch, and turned off when the main current
comparator I1 resets the RS latch. The peak inductor
current at which I1 resets the RS latch is controlled by the
voltageontheITHpin,whichistheoutputoferroramplifier
EA. The VOSENSE pin, described in the Pin Functions
section, allows EA to receive an output feedback voltage
VFB from an external resistive divider. When the load
current increases, it causes a slight decrease in VFB
relativetothe1.19Vreference,whichinturncausestheITH
voltage to increase until the average inductor current
matches the new load current. While the top MOSFET is
off, the bottom MOSFET is turned on until either the
inductor current starts to reverse, as indicated by current
comparator I2, or the beginning of the next cycle.
The LTC1435 is capable of Burst Mode operation in which
the external MOSFETs operate intermittently based on
load demand. The transition to low current operation
begins when comparator I2 detects current reversal and
turnsoffthebottomMOSFET. IfthevoltageacrossRSENSE
doesnotexceedthehysteresisofI2(approximately20mV)
for one full cycle, then on following cycles the top and
bottom drives are disabled. This continues until an induc-
tor current peak exceeds 20mV/RSENSE or the ITH voltage
exceeds 0.6V, either of which causes drive to be returned
to the TG pin on the next cycle.
Twoconditionscanforcecontinuoussynchronousopera-
tion, even when the load current would otherwise dictate
low current operation. One is when the common mode
voltage of the SENSE+ and SENSE– pins is below 1.4V and
the other is when the SFB pin is below 1.19V. The latter
conditionisusedtoassistinsecondarywindingregulation
as described in the Applications Information section.
The top MOSFET driver is biased from floating bootstrap
capacitor CB, which normally is recharged during each off
cycle. However, when VIN decreases to a voltage close to
VOUT, the loop may enter dropout and attempt to turn on
thetopMOSFETcontinuously.Thedropoutdetectorcounts
the number of oscillator cycles that the top MOSFET
remains on and periodically forces a brief off period to
allow CB to recharge.
INTVCC/EXTVCC Power
Power for the top and bottom MOSFET drivers and most
oftheotherLTC1435circuitryisderivedfromtheINTVCC
pin. The bottom MOSFET driver supply pin is internally
connected to INTVCC in the LTC1435. When the EXTVCC
pin is left open, an internal 5V low dropout regulator
supplies INTVCC power. If EXTVCC is taken above 4.8V,
the 5V regulator is turned off and an internal switch is
turned on to connect EXTVCC to INTVCC. This allows the
INTVCC power to be derived from a high efficiency
external source such as the output of the regulator itself
or a secondary winding, as described in the Applications
Information section.
The main control loop is shut down by pulling the RUN/SS
pin low. Releasing RUN/SS allows an internal 3µA current
source to charge soft start capacitor CSS. When CSS
reaches 1.3V, the main control loop is enabled with the ITH
voltage clamped at approximately 30% of its maximum
value. As CSS continues to charge, ITH is gradually re-
leased allowing normal operation to resume.
Comparator OV guards against transient overshoots
> 7.5% by turning off the top MOSFET and keeping it off
until the fault is removed.
7
LTC1435
U
W U U
APPLICATIONS INFORMATION
300
250
200
150
100
50
The basic LTC1435 application circuit is shown in Figure
1, HighEfficiencyStep-DownConverter. Externalcompo-
nent selection is driven by the load requirement and
begins with the selection of RSENSE. Once RSENSE is
known, COSC and L can be chosen. Next, the power
MOSFETs and D1 are selected. Finally, CIN and COUT are
selected. The circuit shown in Figure 1 can be configured
for operation up to an input voltage of 28V (limited by the
external MOSFETs).
0
0
100
200
300
400
500
RSENSE Selection for Output Current
OPERATING FREQUENCY (kHz)
LTC1435 • F02
RSENSE is chosen based on the required output current.
TheLTC1435currentcomparatorhasamaximumthresh-
old of 150mV/RSENSE and an input common mode range
of SGND to INTVCC. The current comparator threshold
sets the peak of the inductor current, yielding a maximum
average output current IMAX equal to the peak value less
half the peak-to-peak ripple current ∆IL.
Figure 2. Timing Capacitor Value
losses will be higher, reducing efficiency (see Efficiency
Considerations). The maximum recommended switching
frequency is 400kHz.
Inductor Value Calculation
Allowing a margin for variations in the LTC1435 and
external component values yields:
The operating frequency and inductor selection are inter-
related in that higher operating frequencies allow the use
of smaller inductor and capacitor values. So why would
anyone ever choose to operate at lower frequencies with
larger components? The answer is efficiency. A higher
frequency generally results in lower efficiency because of
MOSFET gate charge losses. In addition to this basic
trade-off, the effect of inductor value on ripple current and
low current operation must also be considered.
100mV
R
=
SENSE
I
MAX
The LTC1435 works well with values of RSENSE from
0.005Ω to 0.2Ω.
COSC Selection for Operating Frequency
TheLTC1435 usesa constantfrequencyarchitecture with
thefrequencydeterminedbyanexternaloscillatorcapaci-
tor COSC. Each time the topside MOSFET turns on, the
voltage COSC is reset to ground. During the on-time, COSC
is charged by a fixed current. When the voltage on the
capacitor reaches 1.19V, COSC is reset to ground. The
process then repeats.
Theinductorvaluehasadirecteffectonripplecurrent.The
inductor ripple current ∆IL decreases with higher induc-
tance or frequency and increases with higher VIN or VOUT
:
1
V
OUT
∆I =
V
1–
L
OUT
f L
( )( )
V
IN
Accepting larger values of ∆IL allows the use of low
inductances, but results in higher output voltage ripple
and greater core losses. A reasonable starting point for
setting ripple current is ∆IL = 0.4(IMAX). Remember, the
maximum ∆IL occurs at the maximum input voltage.
The value of COSC is calculated from the desired operating
frequency:
4
1.37(10 )
Frequency (kHz)
C
(pF) =
– 11
OSC
The inductor value also has an effect on low current
operation. The transition to low current operation begins
when the inductor current reaches zero while the bottom
A graph for selecting COSC vs frequency is given in Figure
2. As the operating frequency is increased the gate charge
8
LTC1435
U
W U U
APPLICATIONS INFORMATION
MOSFET is on. Lower inductor values (higher ∆IL) will
cause this to occur at higher load currents, which can
cause a dip in efficiency in the upper range of low current
operation. In Burst Mode operation, lower inductance
values will cause the burst frequency to decrease.
ferrite. A reasonable compromise from the same manu-
facturer is Kool Mµ. Toroids are very space efficient,
especially when you can use several layers of wire. Be-
cause they generally lack a bobbin, mounting is more
difficult. However, designsforsurfacemountareavailable
which do not increase the height significantly.
The Figure 3 graph gives a range of recommended induc-
tor values vs operating frequency and VOUT
.
Power MOSFET and D1 Selection
60
Two external power MOSFETs must be selected for use
with the LTC1435: an N-channel MOSFET for the top
(main) switch and an N-channel MOSFET for the bottom
(synchronous) switch.
V
OUT
V
OUT
V
OUT
= 5.0V
= 3.3V
= 2.5V
50
40
30
20
10
0
The peak-to-peak gate drive levels are set by the INTVCC
voltage. This voltage is typically 5V during start-up (see
EXTVCC PinConnection).Consequently,logiclevelthresh-
old MOSFETs must be used in most LTC1435 applica-
tions. The only exception is applications in which EXTVCC
is powered from an external supply greater than 8V (must
be less than 10V), in which standard threshold MOSFETs
(VGS(TH) < 4V) may be used. Pay close attention to the
BVDSS specification for the MOSFETs as well; many of the
logic level MOSFETs are limited to 30V or less.
0
100
150
200
250
300
50
OPERATING FREQUENCY (kHz)
1435 F03
Figure 3. Recommended Inductor Values
Inductor Core Selection
SelectioncriteriaforthepowerMOSFETsincludethe“ON”
Once the value for L is known, the type of inductor must be
selected. High efficiency converters generally cannot af-
ford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy
or Kool Mµ® cores. Actual core loss is independent of core
size for a fixed inductor value, but it is very dependent on
inductance selected. As inductance increases, core losses
godown. Unfortunately, increasedinductancerequiresmore
turns of wire and therefore copper losses will increase.
resistance RSD(ON), reverse transfer capacitance CRSS
,
input voltage and maximum output current. When the
LTC1435 is operating in continuous mode the duty cycles
for the top and bottom MOSFETs are given by:
V
V
OUT
Main Switch Duty Cycle =
IN
V − V
(
)
IN
OUT
Synchronous Switch Duty Cycle =
V
IN
Ferrite designs have very low core loss and are preferred
at high switching frequencies, so design goals can con-
centrate on copper loss and preventing saturation. Ferrite
core material saturates “hard,” which means that induc-
tance collapses abruptly when the peak design current is
exceeded. This results in an abrupt increase in inductor
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
The MOSFET power dissipations at maximum output
current are given by:
V
V
2
OUT
P
=
I
(
1+δ R
+
) (
)
MAIN
MAX
DS ON
(
)
IN
1.85
k V
I
(
C
f
(
)
)(
)( )
IN
MAX
RSS
Molypermalloy (from Magnetics, Inc.) is a very good, low
losscorematerialfortoroids,butitismoreexpensivethan
V − V
2
IN
OUT
P
=
I
(
1+δ R
) (
)
SYNC
MAX
DS ON
(
)
V
IN
Kool Mµ is a registered trademark of Magnetics, Inc.
9
LTC1435
U
W U U
APPLICATIONS INFORMATION
where δ is the temperature dependency of RDS(ON) and k
donotoffermuchrelief.Notethatcapacitormanufacturer’s
ripple current ratings are often based on only 2000 hours
of life. This makes it advisable to further derate the
capacitor or to choose a capacitor rated at a higher
temperaturethanrequired.Severalcapacitorsmayalsobe
paralleled to meet size or height requirements in the
design. Always consult the manufacturer if there is any
question.
is a constant inversely related to the gate drive current.
Both MOSFETs have I2R losses while the topside
N-channel equation includes an additional term for tran-
sition losses, which are highest at high input voltages.
For VIN < 20V the high current efficiency generally im-
proves with larger MOSFETs, while for VIN > 20V the
transition losses rapidly increase to the point that the use
of a higher RDS(ON) device with lower CRSS actual pro-
videshigherefficiency.ThesynchronousMOSFETlosses
are greatest at high input voltage or during a short circuit
when the duty cycle in this switch is nearly 100%. Refer
to the Foldback Current Limiting section for further
applications information.
The selection of COUT is driven by the required effective
series resistance (ESR). Typically, once the ESR require-
ment is satisfied the capacitance is adequate for filtering.
The output ripple (∆VOUT) is approximated by:
1
∆V
≈ ∆I ESR +
L
OUT
4fC
OUT
The term (1 + δ) is generally given for a MOSFET in the
form of a normalized RDS(ON) vs Temperature curve, but
δ = 0.005/°C can be used as an approximation for low
voltageMOSFETs.CRSSisusuallyspecifiedintheMOSFET
characteristics. The constant k = 2.5 can be used to
estimate the contributions of the two terms in the main
switch dissipation equation.
where f = operating frequency, COUT = output capacitance
and ∆IL= ripple current in the inductor. The output ripple
is highest at maximum input voltage since ∆IL increases
with input voltage. With ∆IL = 0.4IOUT(MAX) the output
ripple will be less than 100mV at max VIN assuming:
COUT required ESR < 2RSENSE
The Schottky diode D1 shown in Figure 1 conducts during
the dead-time between the conduction of the two large
power MOSFETs. This prevents the body diode of the
bottom MOSFET from turning on and storing charge
during the dead-time, which could cost as much as 1% in
efficiency. A 1A Schottky is generally a good size for 3A
regulators.
Manufacturers such as Nichicon, United Chemicon and
Sanyoshouldbeconsideredforhighperformancethrough-
hole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest ESR(size)
product of any aluminum electrolytic at a somewhat
higher price. Once the ESR requirement for COUT has been
met, the RMS current rating generally far exceeds the
IRIPPLE(P-P) requirement.
CIN and COUT Selection
In surface mount applications multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum elec-
trolytic and dry tantalum capacitors are both available in
surfacemountconfigurations. Inthecaseoftantalum, itis
critical that the capacitors are surge tested for use in
switching power supplies. An excellent choice is the AVX
TPS series of surface mount tantalum, available in case
heights ranging from 2mm to 4mm. Other capacitor types
include Sanyo OS-CON, Nichicon PL series and Sprague
593Dand 595Dseries.Consultthemanufacturerforother
specific recommendations.
In continuous mode, the source current of the top
N-channel MOSFET is a square wave of duty cycle VOUT
/
VIN. To prevent large voltage transients, a low ESR input
capacitor sized for the maximum RMS current must be
used. The maximum RMS capacitor current is given by:
1/2
]
V
V − V
OUT
(
)
OUT IN
[
C required I
≈I
IN
RMS MAX
V
IN
This formula has a maximum at VIN = 2VOUT, where
IRMS = IOUT/2. This simple worst-case condition is com-
monlyusedfordesignbecauseevensignificantdeviations
10
LTC1435
U
W U U
APPLICATIONS INFORMATION
INTVCC Regulator
additional circuitry is required to derive INTVCC power
from the output.
An internal P-channel low dropout regulator produces the
5V supply which powers the drivers and internal circuitry
within the LTC1435. The INTVCC pin can supply up to
15mA and must be bypassed to ground with a minimum
of2.2µFtantalumorlowESRelectrolytic. Goodbypassing
isnecessarytosupplythehightransientcurrentsrequired
by the MOSFET gate drivers.
The following list summarizes the four possible connec-
tions for EXTVCC:
1. EXTVCC left open (or grounded). This will cause INTVCC
to be powered from the internal 5V regulator resulting
in an efficiency penalty of up to 10% at high input
voltages.
High input voltage applications, in which large MOSFETs
are being driven at high frequencies, may cause the
maximum junction temperature rating for the LTC1435 to
be exceeded. The IC supply current is dominated by the
gate charge supply current when not using an output
derived EXTVCC source. The gate charge is dependent on
operatingfrequencyasdiscussedintheEfficiencyConsid-
erations section. The junction temperature can be esti-
mated by using the equations given in Note 1 of the
Electrical Characteristics. For example, the LTC1435 is
limited to less than 17mA from a 30V supply:
2. EXTVCC connected directly to VOUT. This is the normal
connection for a 5V regulator and provides the highest
efficiency.
3. EXTVCC connectedtoanoutput-derivedboostnetwork.
For 3.3V and other low voltage regulators, efficiency
gains can still be realized by connecting EXTVCC to an
output-derived voltage which has been boosted to
greater than 4.8V. This can be done with either the
inductive boost winding as shown in Figure 4a or the
capacitivechargepumpshowninFigure4b.Thecharge
pump has the advantage of simple magnetics.
TJ = 70°C + (17mA)(30V)(100°C/W) = 126°C
4. EXTVCC connected to an external supply. If an external
To prevent maximum junction temperature from being
exceeded, the input supply current must be checked when
operating in continuous mode at maximum VIN.
supply is available in the 5V to 10V range (EXTVCC
≤
VIN), it may be used to power EXTVCC providing it is
compatible with the MOSFET gate drive requirements.
When driving standard threshold MOSFETs, the exter-
nal supply must always be present during operation to
prevent MOSFET failure due to insufficient gate drive.
EXTVCC Connection
The LTC1435 contains an internal P-channel MOSFET
switch connected between the EXTVCC and INTVCC pins.
The switch closes and supplies the INTVCC power when-
ever the EXTVCC pin is above 4.8V, and remains closed
until EXTVCC drops below 4.5V. This allows the MOSFET
driver and control power to be derived from the output
during normal operation (4.8V < VOUT < 9V) and from the
internal regulator when the output is out of regulation
(start-up, short circuit). Do not apply greater than 10V to
the EXTVCC pin and ensure that EXTVCC < VIN.
+
V
IN
C
IN
1N4148
V
SEC
V
IN
+
L1
1:N
1µF
TG
N-CH
N-CH
OPTIONAL
EXT V
R
CC
SENSE
EXTV
CC
CONNECTION
V
OUT
5V ≤ V
≤ 9V
SEC
+
R6
R5
LTC1435
C
OUT
SW
BG
SFB
Significant efficiency gains can be realized by powering
INTVCC from the output, since the VIN current resulting
from the driver and control currents will be scaled by a
factor of Duty Cycle/Efficiency. For 5V regulators this
PGND
SGND
LTC1435 • F04a
supply means connecting the EXTVCC pin directly to VOUT
.
Figure 4a. Secondary Output Loop and EXTVCC Connection
However, for 3.3V and other lower voltage regulators,
11
LTC1435
U
W U U
APPLICATIONS INFORMATION
1.19V ≤ V
≤ 9V
OUT
R2
+
+
V
IN
1µF
C
IN
V
OSENSE
0.22µF
BAT85
BAT85
100pF
LTC1435
SGND
R1
V
IN
LTC1435 • F05
BAT85
TG
N-CH
N-CH
VN2222LL
R
EXTV
L1
SENSE
CC
V
OUT
Figure 5. Setting the LTC1435 Output Voltage
LTC1435
+
C
OUT
SW
BG
3.3V OR 5V
RUN/SS
RUN/SS
D1
PGND
C
SS
C
SS
LTC1435 • F04b
Figure 4b. Capacitive Charge Pump for EXTVCC
LTC1435 • F06
Figure 6. RUN/SS Pin Interfacing
Topside MOSFET Driver Supply (CB, DB)
Soft start reduces surge currents from VIN by gradually
increasing the internal current limit. Power supply se-
quencing can also be accomplished using this pin.
An external bootstrap capacitor CB connected to the Boost
pinsuppliesthegatedrivevoltageforthetopsideMOSFET.
CapacitorCB intheFunctionalDiagramischargedthrough
diode DB from INTVCC when the SW pin is low. When the
topside MOSFET is to be turned on, the driver places the
CB voltage across the gate source of the MOSFET. This
enhances the MOSFET and turns on the topside switch.
The switch node voltage SW rises to VIN and the Boost pin
rises to VIN + INTVCC. The value of the boost capacitor CB
needs to be 100 times greater than the total input capaci-
tance of the topside MOSFET. In most applications 0.1µF
isadequate.ThereversebreakdownonDB mustbegreater
than VIN(MAX).
An internal 3µA current source charges up an external
capacitor CSS. When the voltage on RUN/SS reaches 1.3V
the LTC1435 begins operating. As the voltage on RUN/SS
continues to ramp from 1.3V to 2.4V, the internal current
limit is also ramped at a proportional linear rate. The
current limit begins at approximately 50mV/RSENSE (at
VRUN/SS = 1.3V) and ends at 150mV/RSENSE (VRUN/SS
>
2.7V). The output current thus ramps up slowly, charging
theoutputcapacitor.IfRUN/SShasbeenpulledalltheway
to ground there is a delay before starting of approximately
500ms/µF, followed by an additional 500ms/µF to reach
full current.
Output Voltage Programming
The output voltage is set by a resistive divider according
to the following formula:
tDELAY = 5(105)CSS Seconds
PullingtheRUN/SSpinbelow1.3VputstheLTC1435into
a low quiescent current shutdown (IQ < 25µA). This pin
can be driven directly from logic as shown in Figure 6.
Diode D1 in Figure 6 reduces the start delay but allows
CSS to ramp up slowly for the soft start function; this
diode and CSS can be deleted if soft start is not needed.
The RUN/SS pin has an internal 6V Zener clamp (See
Functional Diagram).
R2
R1
V
= 1.19V 1+
OUT
The external resistor divider is connected to the output as
shown in Figure 5 allowing remote voltage sensing.
Run/Soft Start Function
The RUN/SS pin is a dual purpose pin which provides the
softstartfunctionandameanstoshutdowntheLTC1435.
12
LTC1435
U
W U U
APPLICATIONS INFORMATION
Foldback Current Limiting
Efficiency Considerations
As described in Power MOSFET and D1 Selection, the
worst-case dissipation for either MOSFET occurs with a
short-circuited output, when the synchronous MOSFET
conducts the current limit value almost continuously. In
most applications this will not cause excessive heating,
even for extended fault intervals. However, when heat
sinking is at a premium or higher RDS(ON) MOSFETs are
being used, foldback current limiting should be added to
reducethecurrentinproportiontotheseverityofthefault.
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
oftenusefultoanalyzeindividuallossestodeterminewhat
is limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Efficiency = 100% – (L1 + L2 + L3 + ...)
whereL1, L2, etc. aretheindividuallossesasapercentage
of input power.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC1435 circuits. LTC1435 VIN current, INTVCC
current,I2Rlosses,andtopsideMOSFETtransitionlosses.
Foldback current limiting is implemented by adding diode
DFB between the output and the ITH pin as shown in the
Functional Diagram. In a hard short (VOUT = 0V) the
current will be reduced to approximately 25% of the
maximum output current. This technique may be used for
all applications with regulated output voltages of 1.8V or
greater.
1. The VIN current is the DC supply current given in the
electricalcharacteristicswhichexcludesMOSFETdriver
and control currents. VIN current results in a small
(< 1%) loss which increases with VIN.
SFB Pin Operation
2. INTVCC current is the sum of the MOSFET driver and
control currents. The MOSFET driver current results
from switching the gate capacitance of the power
MOSFETs. Each time a MOSFET gate is switched from
low to high to low again, a packet of charge dQ moves
from INTVCC to ground. The resulting dQ/dt is a current
out of INT VCC which is typically much larger than the
control circuit current. In continuous mode,
IGATECHG = f(QT + QB), where QT and QB are the gate
charges of the topside and bottom side MOSFETs.
When the SFB pin drops below its ground referenced
1.19V threshold, continuous mode operation is forced. In
continuous mode, the large N-channel main and synchro-
nous switches are used regardless of the load on the main
output.
In addition to providing a logic input to force continuous
synchronous operation, the SFB pin provides a means to
regulate a flyback winding output. Continuous synchro-
nous operation allows power to be drawn from the auxil-
iary windings without regard to the primary output load.
The SFB pin provides a way to force continuous synchro-
nous operation as needed by the flyback winding.
By powering EXTVCC from an output-derived source,
the additional VIN current resulting from the driver and
control currents will be scaled by a factor of
Duty Cycle/Efficiency. For example, in a 20V to 5V
application, 10mA of INTVCC current results in approxi-
mately3mAofVIN current. Thisreducesthemidcurrent
loss from 10% or more (if the driver was powered
directly from VIN) to only a few percent.
Thesecondaryoutputvoltageissetbytheturnsratioofthe
transformerinconjunctionwithapairofexternalresistors
returned to the SFB pin as shown in Figure 4a. The
secondaryregulatedvoltage,VSEC,inFigure4aisgivenby:
R6
R5
V
≈ N +1 V
> 1.19 1+
3. I2R losses are predicted from the DC resistances of the
MOSFET, inductor and current shunt. In continuous
mode the average output current flows through L and
RSENSE, but is “chopped” between the topside main
(
)
SEC
OUT
where N is the turns ratio of the transformer and VOUT is
the main output voltage sensed by VOSENSE
.
13
LTC1435
U
W U U
APPLICATIONS INFORMATION
only solution is to limit the rise time of the switch drive so
that the load rise time is limited to approximately
(25)(CLOAD). Thusa10µFcapacitorwouldrequirea250µs
rise time, limiting the charging current to about 200mA.
MOSFET and the synchronous MOSFET. If the two
MOSFETs have approximately the same RDS(ON), then
the resistance of one MOSFET can simply be summed
with the resistances of L and RSENSE to obtain I2R
losses. For example, if each RDS(ON) = 0.05Ω,
RL = 0.15Ω, and RSENSE = 0.05Ω, then the total
resistance is 0.25Ω. This results in losses ranging
from 3% to 10% as the output current increases from
0.5A to 2A. I2R losses cause the efficiency to drop at
high output currents.
Automotive Considerations:
Plugging into the Cigarette Lighter
As battery-powered devices go mobile, there is a natural
interest in plugging into the cigarette lighter in order to
conserveorevenrechargebatterypacksduringoperation.
But before you connect, be advised: you are plugging into
the supply from hell. The main battery line in an automo-
bileisthesourceofanumberofnastypotentialtransients,
including load dump, reverse battery and double battery.
4. Transition losses apply only to the topside MOSFET(s),
and only when operating at high input voltages (typi-
cally 20V or greater). Transition losses can be esti-
mated from:
Transition Loss = 2.5 (VIN)1.85(IMAX)(CRSS)(f)
Load dump is the result of a loose battery cable. When the
cablebreaksconnection,thefieldcollapseinthealternator
can cause a positive spike as high as 60V which takes
several hundred milliseconds to decay. Reverse battery is
just what it says, while double battery is a consequence of
tow truck operators finding that a 24V jump start cranks
cold engines faster than 12V.
Other losses, including CIN and COUT ESR dissipative
losses, Schottky conduction losses during dead-time,
and inductor core losses, generally account for less
than 2% total additional loss.
Checking Transient Response
The network shown in Figure 7 is the most straightfor-
ward approach to protect a DC/DC converter from the
ravages of an automotive battery line. The series diode
prevents current from flowing during reverse battery,
while the transient suppressor clamps the input voltage
during load dump. Note that the transient suppressor
should not conduct during double battery operation, but
muststillclamptheinputvoltagebelowbreakdownofthe
converter. Although the LT1435 has a maximum input
voltage of 36V, most applications will be limited to 30V
The regulator loop response can be checked by looking at
the load transient response. Switching regulators take
several cycles to respond to a step in DC (resistive) load
current.Whenaloadstepoccurs,VOUT immediatelyshifts
by an amount equal to (∆ILOAD)(ESR), where ESR is the
effective series resistance of COUT. ∆ILOAD also begins to
charge or discharge COUT which generates a feedback
error signal. The regulator loop then acts to return VOUT to
its steady-state value. During this recovery time VOUT can
be monitored for overshoot or ringing which would indi-
cate a stability problem. The ITH external components
shown in the Figure 1 circuit will provide adequate com-
pensation for most applications.
by the MOSFET BVDSS
.
12V
50A I RATING
PK
V
IN
A second, more severe transient is caused by switching in
loads with large (>1µF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in paral-
lel with COUT, causing a rapid drop in VOUT. No regulator
can deliver enough current to prevent this problem if the
load switch resistance is low and it is driven quickly. The
LTC1435
TRANSIENT VOLTAGE
SUPPRESSOR
GENERAL INSTRUMENT
1.5KA24A
1435 F07
Figure 7. Automotive Application Protection
14
LTC1435
U
W U U
APPLICATIONS INFORMATION
Design Example
highest at the maximum input voltage. The output voltage
ripple due to ESR is approximately:
As a design example, assume VIN = 12V(nominal), VIN =
22V(max), VOUT = 3.3V, IMAX = 3A and f = 250kHz, RSENSE
and COSC can immediately be calculated:
V
ORIPPLE = RESR(∆IL) = 0.03Ω(1.112A) = 34mVP-P
PC Board Layout Checklist
RSENSE = 100mV/3A = 0.033Ω
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1435. These items are also illustrated graphically in
the layout diagram of Figure 8. Check the following in your
layout:
COSC = 1.37(104)/250 – 11 = 43pF
Referring to Figure 3, a 10µH inductor falls within the
recommended range. To check the actual value of the
ripple current the following equation is used:
1. Are the signal and power grounds segregated? The
LTC1435 signal ground pin must return to the (–) plate
of COUT. The power ground connects to the source of
the bottom N-channel MOSFET, anode of the Schottky
diode, and (–) plate of CIN, which should have as short
lead lengths as possible.
V
f L
( )( )
V
OUT
OUT
∆I =
1–
L
V
IN
The highest value of the ripple current occurs at the
maximum input voltage:
3.3V
3.3V
22V
∆I =
1–
= 1.12A
L
2. Does the VOSENSE pin connect directly to the feedback
resistors? The resistive divider R1, R2 must be con-
nectedbetweenthe(+)plateofCOUT andsignalground.
The 100pF capacitor should be as close as possible to
the LTC1435.
3. AretheSENSE– andSENSE+ leadsroutedtogetherwith
minimum PC trace spacing? The filter capacitor be-
tween SENSE+ and SENSE– should be as close as
possible to the LTC1435.
250kHz 10µH
(
)
The power dissipation on the topside MOSFET can be
easily estimated. Choosing a Siliconix Si4412DY results
in: RDS(ON) = 0.042Ω, CRSS = 100pF. At maximum input
voltage with T(estimated) = 50°C:
2
( )
3.3V
22V
P
=
3
1+ 0.005 50°C − 25°C 0.042Ω
(
)(
) (
]
)
)
MAIN
[
1.85
+ 2.5 22V
3A 100pF 250kHz = 122mW
(
)
(
)(
)(
4. Does the (+) plate of CIN connect to the drain of the
topsideMOSFET(s)ascloselyaspossible?Thiscapaci-
tor provides the AC current to the MOSFET(s).
The most stringent requirement for the synchronous
N-channel MOSFET occurs when VOUT = 0 (i.e. short
circuit). In this case the worst-case dissipation rises to:
5. Is the INTVCC decoupling capacitor connected closely
between INTVCC and the power ground pin? This ca-
pacitor carries the MOSFET driver peak currents.
2
P
= I
1+δ R
(
DS ON
)
SYNC
(
SC AVG
)
(
)
(
)
6. KeeptheswitchingnodeSWawayfromsensitivesmall-
signal nodes. Ideally the switch node should be placed
at the furthest point from the LTC1435.
With the 0.033Ω sense resistor ISC(AVG) = 4A will result,
increasing the Si4412DY dissipation to 950mW at a die
temperature of 105°C.
7. SGND should be exclusively used for grounding exter-
nal components on COSC, ITH, VOSENSE and SFB pins.
CIN is chosen for an RMS current rating of at least 1.5A at
temperature. COUT is chosen with an ESR of 0.03Ω for low
outputripple. Theoutputrippleincontinuousmodewillbe
15
LTC1435
APPLICATIONS INFORMATION
U
W U U
+
C
M1
OSC
1
2
16
15
C
IN
C
TG
OSC
C
SS
RUN/SS
BOOST
V
C
IN
C1
R
C
3
14
I
SW
TH
C
B
0.1µF
C
4
5
13
12
C2
LTC1435
D
D1
SFB
V
B
IN
SGND
INTV
CC
–
–
100pF
+
M2
6
7
11
10
4.7µF
BG
V
OSENSE
–
SENSE
PGND
1000pF
8
9
+
SENSE
EXTV
CC
L1
R1
C
OUT
+
V
OUT
R
SENSE
R2
BOLD LINES INDICATE
HIGH CURRENT PATHS
+
LTC1435 • F08
Figure 8. LTC1435 Layout Diagram
U
TYPICAL APPLICATIONS
Dual Output 5V and Synchronous 12V Application
V
IN
5.4V TO 28V
C
IN
0.01µF
C
OSC
68pF
+
22µF
35V
× 2
IRLL014
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
M1
C
TG
BOOST
SW
OSC
Si4412DY
4.7k
C
SS
0.1µF
RUN/SS
R
C
C
C1
10k
470pF
I
TH
T1
C
C2
51pF
C
SEC
+
10µH
3.3µF
SFB
V
IN
1:1.42
35V
LTC1435
R
SENSE
0.033Ω
0.1µF
CMDSH-3
V
OUT
5V/3.5A
SGND
INTV
CC
+
100pF
R1
35.7k
1%
4.7µF
V
BG
OSENSE
M2
Si4412DY
C
MBRS140T3
OUT
+
100µF
–
10V
× 2
SENSE
SENSE
PGND
R2
20k
1%
1000pF
+
EXTV
CC
100Ω
100Ω
SGND
V
OUT2
12V
LTC1435 • TA04
11.3k
1%
100k
1%
T1: DALE LPE6562-A236
120mA
16
LTC1435
U
TYPICAL APPLICATIONS
3.3V/4.5A Converter with Foldback Current Limiting
V
IN
4.5V TO 28V
C
OSC
68pF
C
IN
+
22µF
35V
× 2
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
M1
C
TG
BOOST
SW
OSC
Si4410DY
C
SS
0.1µF
RUN/SS
R
C
C
C1
10k
330pF
I
TH
I
PIN 3
TH
C
C2
51pF
IN4148
SFB
V
INTV
IN
CC
L1
10µH
LTC1435
R
0.1µF
CMDSH-3
SENSE
0.025Ω
V
OUT
3.3V/4.5A
SGND
INTV
CC
+
100pF
R1
4.7µF
35.7k
1%
V
BG
OSENSE
M2
Si4410DY
C
OUT
+
100µF
MBRS140T3
–
10V
× 2
SENSE
SENSE
PGND
100pF
R2
20k
1%
1000pF
OPTIONAL:
CONNECT TO 5V
+
EXTV
CC
SGND
(PIN 5)
LTC1435 • TA01
Dual Output 5V and 12V Application
V
IN
5.4V TO 28V
C
IN
C
OSC
68pF
+
22µF
35V
× 2
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
M1
C
TG
BOOST
SW
OSC
IRF7403
C
SS
0.1µF
RUN/SS
MBRS1100T3
R
C
C
C1
10k
510pF
I
TH
24V
OUT
C
T1
C2
C
SEC
+
51pF
10µH
3.3µF
SFB
V
IN
1:2.2
25V
LTC1435
0.1µF
CMDSH-3
V
SGND
INTV
CC
5V/3.5A
R
SENSE
0.033Ω
+
100pF
R1
4.7µF
35.7k
1%
V
BG
OSENSE
M2
IRF7403
C
MBRS140T3
OUT
+
100µF
–
10V
× 2
SENSE
SENSE
PGND
R2
20k
1%
1000pF
+
EXTV
CC
100Ω
100Ω
SGND
10k
90.9k
V
OUT2
12V
LTC1435 • TA02
T1: DALE LPE6562-A092
17
LTC1435
TYPICAL APPLICATIONS
U
Constant-Current/Constant-Voltage High Efficiency Battery Charger
E1
V
IN
+
C1*
22µF
35V
+
C2*
22µF
35V
C4
R7
C5
0.1µF
0.1µF
1.5M
C11
E3
GND
E3
56pF
1
2
3
4
5
6
7
8
16
Q1
SHDN
C
TG
C12
0.1µF
OSC
Si4412DY
C13
0.033µF
15
14
13
12
11
10
9
R5
1k
L1
27µH
R1
0.025Ω
RUN/SS BOOST
D1
E6
U1
I
TH
SW
BATT
C6
0.33µF
C14
1000pF
LTC1435
+
C3
22µF
35V
SFB
V
D2
IN
E7
GND
SGND INTV
CC
C9
100pF
Q2
Si4412DY
V
BG
OSENSE
–
SENSE
PGND
U2
LT1620
C15
0.1µF
+
SENSE EXTV
CC
C8
C10
100pF
100pF
1
8
7
6
5
C7
SENSE
AVG
4.7µF
+
2
3
4
16V
I
PROG
OUT
R2
1M
0.1%
GND
NIN
V
CC
PIN
R3
105k
0.1%
R4
76.8k
0.1%
C16
0.33µF
C18
0.1µF
JP1A
JP1B
R6
10k
1%
C17
0.01µF
DC133 F01
E5
GND
E4
PROG
*CONSULT CAPACITOR MANUFACTURER FOR RECOMMENDED
ESR RATING FOR CONTINUOUS 4A OPERATION
I
R
PROG
Current Programming Equation
(I
PROG
)(R6) – 0.04
10(R1)
I
=
BATT
Efficiency
100
95
V
IN
= 24V
V
= 16V
= 12V
BATT
V
BATT
90
V
= 6V
BATT
85
80
75
0
1
2
3
4
5
BATTERY CHARGE CURRENT (A)
1435 TA05
18
LTC1435
U
PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
G Package
16-Lead Plastic SSOP (0.209)
(LTC DWG # 05-08-1640)
0.239 – 0.249*
(6.07 – 7.33)
16 15 14 13 12 11 10
9
0.205 – 0.212**
(5.20 – 5.38)
0.068 – 0.078
(1.73 – 1.99)
0.301 – 0.311
(7.65 – 7.90)
0° – 8°
0.0256
(0.65)
BSC
0.005 – 0.009
(0.13 – 0.22)
0.022 – 0.037
(0.55 – 0.95)
0.002 – 0.008
(0.05 – 0.21)
0.010 – 0.015
(0.25 – 0.38)
5
7
8
1
2
3
4
6
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
G16 SSOP 0795
**DIMENSIONS DO NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
S Package
16-Lead Plastic Small Outline
(Narrow 0.150)
(LTC DWG # 05-08-1610)
0.386 – 0.394*
(9.804 – 10.008)
16
15
14
13
12
11
10
9
0.150 – 0.157**
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
5
7
8
1
2
3
4
6
0.010 – 0.020
(0.254 – 0.508)
× 45°
0.053 – 0.069
(1.346 – 1.752)
0.004 – 0.010
(0.101 – 0.254)
0.008 – 0.010
(0.203 – 0.254)
0° – 8° TYP
0.050
(1.270)
TYP
0.014 – 0.019
(0.355 – 0.483)
0.016 – 0.050
0.406 – 1.270
S16 0695
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.
19
LTC1435
TYPICAL APPLICATION
U
Low Dropout 2.9V/3A Converter
V
IN
3.5V TO 25V
C
OSC
68pF
C
IN
+
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
22µF
35V
× 2
M1
C
TG
BOOST
SW
OSC
1/2 Si9925DY
C
SS
0.1µF
RUN/SS
R
C
C
C1
10k
330pF
I
TH
C
C2
51pF
SFB
V
INTV
IN
CC
L1
10µH
LTC1435
R
0.1µF
CMDSH-3
SENSE
0.033Ω
V
OUT
2.9V/3A
SGND
INTV
CC
+
100pF
R1
4.7µF
35.7k
1%
V
BG
OSENSE
M2
100pF
C
OUT
MBRS140T3
1/2 Si9925DY
+
100µF
–
10V
× 2
SENSE
SENSE
PGND
R2
24.9k
1%
1000pF
OPTIONAL:
CONNECT TO 5V
+
EXTV
CC
SGND
LTC1435 • TA03
L1: SUMIDA CDRH125-10
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC1142HV/LTC1142
LTC1148HV/LTC1148
Dual High Efficiency Synchronous Step-Down Switching Regulators Dual Synchronous, V ≤ 20V
IN
High Efficiency Sychronous Step-Down Switching
Regulator Controllers
Synchronous, V ≤ 20V
IN
LTC1159
LT®1375/LT1376
High Efficiency Synchronous Step-Down Switching Regulator
1.5A, 500kHz Step-Down Switching Regulators
Synchronous, V ≤ 40V, For Logic Threshold FETs
IN
High Frequency, Small Inductor, High Efficiency
Switchers, 1.5A Switch
LTC1430
High Power Step-Down Switching Regulator Controller
High Efficiency 5V to 3.3V Conversion at Up to 15A
Full-Featured Single Controller
LTC1436/LTC1436-PLL/ High Efficiency Low Noise Synchronous Step-Down
LTC1437
Switching Regulators
LTC1438/LTC1439
Dual High Efficiency, Low Noise, Synchronous Step-Down
Switching Regulators
Full-Featured Dual Controllers
LT1510
Constant-Voltage/ Constant-Current Battery Charger
1.3A, Li-Ion, NiCd, NiMH, Pb-Acid Charger
5V Standby in Shutdown
LTC1538-AUX
Dual High Efficiency, Low Noise, Synchronous Step-Down
Switching Regulator
LTC1539
Dual High Efficiency, Low Noise, Synchronous Step-Down
Switching Regulator
5V Standby in Shutdown
LT/GP 0896 7K • PRINTED IN USA
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
20
●
●
(408) 432-1900 FAX: (408) 434-0507 TELEX: 499-3977
LINEAR TECHNOLOGY CORPORATION 1996
相关型号：
LTC1435IG#TRPBF
LTC1435 - High Efficiency Low Noise Synchronous Step-Down Switching Regulator; Package: SSOP; Pins: 16; Temperature Range: -40°C to 85°C
Linear
LTC1435IS#PBF
LTC1435 - High Efficiency Low Noise Synchronous Step-Down Switching Regulator; Package: SO; Pins: 16; Temperature Range: -40°C to 85°C
Linear
LTC1436ACGN#TRPBF
LTC1436A - High Efficiency Low Noise Synchronous Step-Down Switching Regulators; Package: SSOP; Pins: 24; Temperature Range: 0°C to 70°C
Linear
LTC1436ACGN-PLL#PBF
LTC1436A - High Efficiency Low Noise Synchronous Step-Down Switching Regulators; Package: SSOP; Pins: 24; Temperature Range: 0°C to 70°C
Linear
LTC1436AEGN#TRPBF
LTC1436A - High Efficiency Low Noise Synchronous Step-Down Switching Regulators; Package: SSOP; Pins: 24; Temperature Range: -40°C to 85°C
Linear
©2020 ICPDF网 联系我们和版权申明