LTC1701ES5 [Linear]
1MHz Step-Down DC/DC Converter in SOT-23; 采用SOT -23为1MHz降压型DC / DC转换器
| 型号： | LTC1701ES5 |
| 厂家： | 
Linear |
| 描述： | 1MHz Step-Down DC/DC Converter in SOT-23 |
| 文件： | 总12页 (文件大小：142K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
Fina l Ele c tric a l Sp e c ific a tio ns
LTC1701
1MHz Ste p -Do wn
DC/ DC Co nve rte r in SOT-23
De c e m b e r 1999
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DESCRIPTIO
FEATURES
■
Tiny 5-Lead SOT-23 Package
Uses Tiny Capacitors and Inductor
High Frequency Operation: 1MHz
High Output Current: 500mA
Low RDS(ON) Internal Switch: 0.28Ω
High Efficiency: Up to 94%
Current Mode Operation for Excellent Line
and Load Transient Response
Short-Circuit Protected
Low Quiescent Current: 135µA
Low Dropout Operation: 100% Duty Cycle
Ultralow Shutdown Current: IQ < 1µA
Peak Inductor Current Independent of Inductor Value
Output Voltages from 5V Down to 1.25V
The LTC®1701 is the industry’s first 5-lead SOT-23 step
down, current mode, DC/DC converter. Intended for small
to medium power applications, it operates from 2.5V to
5.5V input voltage range and switches at 1MHz, allowing
the use of tiny, low cost capacitors and inductors 2mm or
less in height. The output voltage is adjustable from 1.25V
to 5V. A built-in 0.28Ω switch allows up to 0.5A of output
current at high efficiency. OPTI-LOOPTM compensation
allows the transient response to be optimized over a wide
range of loads and output capacitors.
■
■
■
■
■
■
■
■
■
The LTC1701 incorporates automatic power saving Burst
ModeTM operation to reduce gate charge losses when the
load current drops below the level required for continuous
operation. With no load, the converter draws only 135µA.
In shutdown, it draws less than 1µA, making it ideal for
current sensitive applications.
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APPLICATIO S
In dropout, the internal P-channel MOSFET switch is
turned on continuously, thereby maximizing battery life.
Its small size and switching frequency enables the com-
plete DC/DC converter function to consume less than 0.3
square inches of PC board area.
■
PDAs/Palmtop PCs
Digital Cameras
Cellular Phones
Portable Media Players
PC Cards
Handheld Equipment
■
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, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode and OPTI-LOOP are trademarks of Linear Technology Corporation.
■
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TYPICAL APPLICATIO
Efficiency Curve
L1
4.7µH
V
IN
V
OUT
100
2.5V TO
5.5V
(2.5V/
500mA)
V
SW
IN
V
IN
= 3.3V
95
90
85
80
75
70
D1
R4
1M
R2
121k
LTC1701
+
C1
10µF
+
C2
47µF
I
TH
/RUN
V
FB
R3
5.1k
C3
R1
121k
GND
330pF
C1: TAIYO YUDEN JMK316BJ106ML
C2: SANYO POSCAP 6TPA47M
D1: MBRM120L
1701 F01
L1: SUMIDA CD43-4R7
1
10
100
1000
LOAD CURRENT (mA)
Figure 1. Step-Down 2.5V/500mA Regulator
1701 F01a
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tationthattheinterconnectionofits circuits as describedhereinwillnotinfringeonexistingpatentrights.
1
LTC1701
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ABSOLUTE AXI U RATI GS
PACKAGE/ORDER I FOR ATIO
(Note 1)
ORDER PART
NUMBER
(Voltages Referred to GND Pin)
TOP VIEW
V Voltage (Pin 5).......................................–0.3V to 6V
IN
SW 1
5 V
IN
ITH/RUN Voltage (Pin 4) ..............................–0.3V to 3V
V Voltage (Pin 3) ......................................–0.3V to 3V
FB
LTC1701ES5
GND 2
V
FB
3
4 I /RUN
TH
Peak Switch Current (Pin 1) ................................... 1.3A
V – SW (Max Switch Voltage)................8.5V to –0.3V
IN
S5 PART
MARKING
S5 PACKAGE
5-LEAD PLASTIC SOT-23
Operating Temperature Range (Note 2) .. –40°C to 85°C
Junction Temperature (Note 5)............................. 125°C
Storage Temperature Range ................. –65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
TJMAX = 125°C, θJA = 110°C/W
LTKG
Consult factory for Industrial and Military grade parts.
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. V = 3.3V, RITH/RUN = 1Meg (from V to ITH/RUN) unless otherwise
IN
IN
specified. (Note 2)
SYMBOL PARAMETER
CONDITIONS
MIN
TYP
MAX
5.5
UNITS
V
V
Operating Voltage Range
Feedback Pin Input Current
Feedback Voltage
2.5
IN
I
FB
(Note 3)
(Note 3)
±0.1
1.28
0.1
µA
V
FB
●
1.22
1.25
0.04
V
∆V
Reference Voltage Line Regulation
Output Voltage Load Regulation
V
IN
= 2.5V to 5V (Note 3)
%/V
LINE REG
∆V
LOAD REG
Measured in Servo Loop, V = 1.5V, (Note 3)
Measured in Servo Loop, V = 1.9V, (Note 3)
0.01
–0.80
0.70
–1.50
%
%
ITH
ITH
Input DC Supply Current (Note 4)
Active Mode
Sleep Mode
V
= 0V
= 1.4V
185
135
0.25
300
200
1
µA
µA
µA
FB
V
FB
Shutdown
V
= 0V
ITH/RUN
V
Run Threshold High
Run Threshold Low
I
Ramping Down
Ramping Up
1.4
0.6
1.6
V
V
ITH/RUN
TH/RUN
I
0.3
50
TH/RUN
I
Run Pullup Current
V
= 1V
100
1.1
300
µA
ITH/RUN
ITH/RUN
I
Peak Switch Current Threshold
Switch ON Resistance
V
= 0V
0.9
A
SW(PEAK)
FB
R
DS(ON)
V
V
V
= 5V, V = 0V
0.28
0.30
0.35
Ω
Ω
Ω
IN
FB
= 3.3V, V = 0V
FB
= 2.5V, V = 0V
FB
IN
IN
I
Switch Leakage Current
Switch Off-Time
V
IN
= 5V, V
= 0V, V = 0V
0.01
500
1
µA
SW(LKG)
ITH/RUN
FB
t
400
600
ns
OFF
Note 1: Absolute Maximum Ratings are those values beyond which the life
Note 3: The LTC1701 is tested in a feedback loop which servos V to the
FB
of a device may be impaired.
midpoint for the error amplifier (V = 1.7V unless otherwise specified).
ITH
Note 2: The LTC1701E is guaranteed to meet performance specifications
from 0°C to 70°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.
Note 4: Dynamic supply current is higher due to the internal gate charge
being delivered at the switching frequency.
Note 5: T is calculated from the ambient T and power dissipation P
J
A
D
according to the following formula:
LTC1701ES5: T = T + (P •110°C/W)
J
A
D
2
LTC1701
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SW (Pin 1): The Switch Node Connection to the Inductor.
ITH/RUN (Pin 4): Combination of Error Amplifier Compen-
sation Point and Run Control Input. The current compara-
tor threshold increases with this control voltage. Nominal
voltage range for this pin is 1.25V to 2.25V. Forcing this
pin below 0.8V causes the device to be shut down. In
shutdown all functions are disabled.
This pin swings from V to a Schottky diode (external)
IN
voltage drop below ground. The cathode of the Schottky
diode must be closely connected to this pin.
GND (Pin 2): Ground Pin. Connect to the (–) terminal of
C
OUT, the Schottky diode and (–) terminal of C .
IN
V (Pin 5): Main Supply Pin and the (+) Input to the
CurrentComparator.Mustbecloselydecoupledtoground.
IN
V
(Pin 3): Receives the feedback voltage from the
FB
external resistive divider across the output. Nominal volt-
age for this pin is 1.25V.
Pin Limit Table
NOMINAL (V)
TYP MAX
ABSOLUTE MAX (V)
MIN MAX
PIN
1
NAME
SW
DESCRIPTION
MIN
Switch Node
–0.3
V
IN
–0.3
V
IN
+ 0.3
2
GND
Ground Pin
0
3
V
Output Feedback Pin
Error Amplifier Compensation and RUN Pin
Main Power Supply
0
0
1.25
1.35
2.25
5.5
–0.3
–0.3
–0.3
3
3
6
FB
4
I /RUN
TH
5
V
IN
2.5
W
BLOCK DIAGRA
V
IN
V
IN
V
IN
1.25V
BANDGAP
REFERENCE
V
REF
(1.25V)
50µA
+
V
REF
+
CURRENT
SENSE
AMP
I
/REF
+
TH
CLAMP
CURRENT
COMP
–
–
+
1.5V
–
I
TH
COMP
V
REF
SHDN
I /RUN
TH
–
V
REF
+
ERROR
AMP
(1.25V TO 2.25V)
V
FB
–
SW
+
OFF-TIMER
AND GATE
CONTROL LOGIC
GATE
DRIVER
OVER
VOLTAGE
COMP
–
1.4V
PULSE
GND
STRETCHER
V
FB
<0.6V
1701 BD
3
LTC1701
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OPERATIO
The LTC1701 uses a contant off-time, current mode
architecture. The operating frequency is then determined
The main control loop is shut down by pulling the ITH/RUN
pintoground.Whenthepinis releasedanexternalresistor
is used to charge the compensation capacitor. When the
voltage at the ITH/RUN pin reaches 0.8V, the main control
loop is enabled and the error amplifier drives the ITH/RUN
pin. Soft-start canbe implementedbyrampingthe voltage
on the ITH/RUN pin (see Applications Information sec-
tion).
by the off-time and the difference between V and VOUT
.
IN
Tooptimizeefficiency,theLTC1701automaticallyswitches
between continuous and Burst Mode operation.
The output voltage is set by an external divider returned to
theV pin. Anerroramplfiercompares thedividedoutput
FB
voltage with a reference voltage of 1.25V and adjusts the
peak inductor current accordingly.
Low Current Operation
Main Control Loop
When the load is relatively light, the LTC1701 automati-
cally switches to Burst Mode operation in which the
internal PMOS switch operates intermittently based on
load demand. The main control loop is interrupted when
the output voltage reaches the desired regulated value.
The hysteretic voltage comparator trips when ITH/RUN is
below 1.5V, shutting off the switch and reducing the
power consumed. The output capacitor and the inductor
supply the power to the load until the output voltage drops
slightly and the ITH/RUN pin exceeds 1.5V, turning on the
switch and the main control loop which starts another
cycle.
During normal operation, the internal PMOS switch is
turned on when the V voltage is below the reference
FB
voltage. The current into the inductor and the load in-
creases until the current limit is reached. The switch turns
off and energy stored in the inductor flows through the
external Schottky diode into the load. After the constant
off-timeinterval,theswitchturns onandthecyclerepeats.
The peak inductor current is controlled by the voltage on
the ITH/RUN pin, which is the output of the error
amplifier.This amplifier compares the V pin to the 1.25V
FB
reference. Whentheloadcurrentincreases, theFBvoltage
decreases slightly below the reference. This decrease
causes the error amplifier to increase the ITH/RUN voltage
until the average inductor current matches the new load
current.
Dropout Operation
Indropout, theinternalPMOSswitchis turnedoncontinu-
ously (100% duty cycle) providing low dropout operation
with VOUT at V . Since the LTC1701 does not incorporate
IN
an under voltage lockout, care should be taken to shut
down the LTC1701 for V < 2.5V.
IN
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APPLICATIO S I FOR ATIO
The basic LTC1701 application circuit is shown in
Figure 1. External component selection is driven by the
loadrequirementandbegins withtheselectionofL1.Once
L1 is chosen, the Schottky diode D1 can be selected
V − V
1
IN
OUT
fO =
V + V
IN
T
OFF
D
Although the inductor does not influence the operating
frequency, the inductor value has a direct effect on ripple
current. The inductor ripple current ∆IL decreases with
followed by C and COUT
.
IN
L Selection and Operating Frequency
higher inductance and increases with higher V or V
:
IN
OUT
The operating frequency is fixed by V , VOUT and the
IN
constant off-time of about 500ns. The complete expres-
sion for operating frequency is given by:
V − V
VOUT + V
D
IN
OUT
∆IL =
fL
V + V
IN D
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LTC1701
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APPLICATIO S I FOR ATIO
where VD is the output Schottky diode forward drop.
facturer is Kool Mµ core material. Toroids are very space
efficient, expecially when you can use several layers of
wire. Because they generally lack a bobbin, mounting is
more difficult. However, surface mount designs that do
not increase the height significantly are available
Accepting larger values of ∆IL allows the use of low
inductances, but results in higher output voltage ripple
and greater core losses. A reasonable starting point for
setting ripple current is ∆IL = 0.4A.
The inductor value also has an effect on low current
operation. Lower inductor values (higher ∆IL) will cause
Burst Mode operation to begin at higher load currents,
which can cause a dip in efficiency in the upper range of
low current operation. In Burst Mode operation, lower
inductance values will cause the burst frequency to de-
crease.
Catch Diode Selection
The diode D1 shown in Figure 1 conducts during the off-
time. It is important to adequately specify the diode peak
current and average power dissipation so as not to exceed
the diode ratings.
Losses in the catch diode depend on forward drop and
switching times. Therefore, Schottky diodes are a good
choice for low drop and fast switching times.
Inductor Core Selection
Once the value for L is selected, the type of inductor must
be chosen. Basically, there are two kinds of losses in an
inductor —core and copper losses.
Since the catch diode carries the load current during the
off-time, the average diode current is dependent on the
switch duty cycle. At high input voltages, the diode con-
ducts most of the time. As V approaches VOUT, the diode
IN
Core losses are dependent on the peak-to-peak ripple
current and core material. However, it is independent of
the physical size of the core. By increasing inductance, the
peak-to-peak inductor ripple current will decrease, there-
fore reducing core loss. Unfortunately, increased induc-
tance requires more turns of wire and, therefore, copper
losses will increase. When space is not a premium, larger
wire can be used to reduce the wire resistance. This also
prevents excessive heat dissipation in the inductor.
conducts only a small fraction of the time. The most
stressful condition for the diode is when the regulator
output is shorted to ground.
Under short-circuit conditions (VOUT = 0V), the diode
must safely handle ISC(PK) at close to 100% duty cycle.
Under normal load conditions, the average current con-
ducted by the diode is simply:
V − V
IN
OUT
Highefficiencyconverters generallycannotaffordthecore
loss found in low cost powdered iron cores, forcing the
use of more expensive ferrite, molypermalloy or Kool Mµ®
cores. These low core loss materials allow the user to
concentrate on reducing copper loss and preventing satu-
ration.
IDIODE(avg) = ILOAD(avg)
V + V
IN
D
Remember to keep lead lengths short and observe proper
grounding (see Board Layout Considerations) to avoid
ringing and increased dissipation.
Theforwardvoltagedropallowedinthediodeis calculated
from the maximum short-circuit current as:
Ferrite designs have very low core loss and are preferred
at high switching frequencies. Ferrite core material satu-
rates “hard,” which means that inductance collapses
abruptly when the peak design current is exceeded. This
results in an abrupt increase in inductor ripple current and
consequent output voltage ripple. Do not allow the core to
saturate!
PD
V + V
IN D
V ≈
D
ISC(avg)
V
IN
where PD is the allowable diode power dissipation and will
be determined by efficiency and/or thermal requirements
(see Efficiency Considerations).
Molypermalloy (from Magnetics, Inc.) is a very good, low
loss corematerialfortoroids,butitis moreexpensivethan
ferrite. A reasonable compromise from the same manu-
Kool Mµ is a registered trademark of Magnetics, Inc.
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LTC1701
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APPLICATIO S I FOR ATIO
Most LTC1701 circuits will be well served by either an
MBR0520L or an MBRM120L. An MBR0520L is a good
choice for IOUT(MAX) ≤ 500mA, as long as the output
doesn’t need to sustain a continuous short.
ESRCOUT < 100mΩ
Once the ESR requirements for COUT have been met, the
RMS current rating generally far exceeds the IRIPPLE(P-P)
requirement.
Input Capacitor (C ) Selection
When the capacitance of COUT is made too small, the
outputrippleatlowfrequencies willbelargeenoughtotrip
the ITH comparator. This causes Burst Mode operation to
be activated when the LTC1701 would normally be in
continuous modeoperation. Theeffectcanbeimprovedat
higher frequencies with lower inductor values.
IN
In continuous mode, the input current of the converter is
a square wave with a duty cycle of approximately VOUT
/
V . To prevent large voltage transients, a low equivalent
IN
series resistance (ESR) input capacitor sized for the maxi-
mum RMS current must be used. The maximum RMS
capacitor current is given by:
In surface mount applications, multiple capacitors may
have to be paralleled to meet the capacitance, ESR or RMS
current handling requirement of the application. Alumi-
num electrolyte and dry tantulum capacitors are both
available in surface mount configurations. The OS-CON
semiconductor dielectric capacitor available from Sanyo
has the lowest ESR(size) product of any aluminum elec-
trolytic at a somewhat higher price. In the case of tanta-
lum,itis criticalthatthecapacitors aresurgetestedforuse
in switching power supplies. An excellent choice is the
AVX TPS, AVX TPSV and KEMET T510 series of surface
mount tantalums, avalable in case heights ranging from
2mm to 4mm. Other capacitor types include Nichicon PL
series, Sanyo POSCAP and Panasonic SP.
V
OUT
V − V
IN OUT
(
)
IRMS ≈IMAX
V
IN
where the maximum average output current IMAX equals
the peak current (1 Amp) minus half the peak-to-peak
ripple current, IMAX = 1 – ∆IL/2.
This formula has a maximum at V = 2VOUT, where IRMS
IN
= IOUT/2. This simple worst-case is commonly used to
design because even significant deviations do not offer
much relief. Note that capacitor manufacturer’s ripple
current ratings are often based on only 2000 hours life-
time. This makes it advisable to further derate the capaci-
tor, or choose a capacitor rated at a higher temperature
thanrequired. Severalcapacitors mayalsobeparalleledto
meet the size or height requirements of the design. An
additional 0.1µF to 1µF ceramic capacitor is also recom-
Ceramic Capacitors
Higher value, lower cost ceramic capacitors are now
becomingavailableinsmallercasesizes.Thesearetempt-
ing for switching regulator use because of their very low
ESR. Unfortunately, the ESR is so low that it can cause
loop stability problems. Solid tantalum capacitor ESR
generates aloop“zero”at5kHzto50kHzthatis instrumen-
tal in giving acceptable loop phase margin. Ceramic ca-
pacitors remain capacitive to beyond 300kHz and usually
resonate with their ESL before ESR becomes effective.
Also, ceramic caps are prone to temperature effects which
requires the designer to check loop stability over the
operating temperature range.
mended on V for high frequency decoupling.
IN
Output Capacitor (COUT) Selection
The selection of COUT is driven by the required ESR.
Typically, once the ESR requirement is satisfied, the
capacitance is adequate for filtering. The output ripple
(∆VOUT) is determined by:
1
∆VOUT ≈ ∆IL ESR +
8fCOUT
For these reasons, most of the input and output capaci-
tance should be composed of tantalum capacitors for
stability combined with about 0.1µF to 1µF of ceramic
capacitors for high frequency decoupling.
where f = operating frequency, COUT = output capacitance
and ∆IL = ripple current in the inductor. With ∆IL = 0.4
IOUT(MAX) the output ripple will be less than 100mV with:
6
LTC1701
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APPLICATIO S I FOR ATIO
Setting the Output Voltage
(from 0.5 to 2 times their suggested values) to optimize
transient response once the final PC layout is done and the
particular output capacitor type and value have been
determined. The output capacitors need to be selected
because the various types and values determine the loop
feedback factor gain and phrase. An output current pulse
of 20% to 100% of full-load current having a rise time of
1µs to 10µs will produce output voltage and ITH pin
waveforms that will give a sense of the overall loop
stability without breaking the feedback loop.
The LTC1701 develops a 1.25V reference voltage between
the feedback pin, V , and the signal ground as shown in
FB
Figure 2. The output voltage is set by a resistive divider
according to the following formula:
R2
VOUT = 1.25V 1+
R1
To prevent stray pickup, a capacitor of about 5pF can be
added across R1, located close to the LTC1701. Unfortu-
nately, the load step response is degraded by this capaci-
tor. Using a good printed circuit board layout eliminates
the need for this capacitor. Great care should be taken to
The initial output voltage step may not be within the
bandwidth of the feedback loop, so the standard second-
order overshoot/DC ratio cannot be used to determine
phase margin. The gain of the loop increases with R3 and
the bandwidth of the loop increases with decreasing C3. If
R3 is increased by the same factor that C3 is decreased,
the zero frequency will be kept the same, thereby keeping
the phase the same in the most critical frequency range of
the feedback loop. In addition, a feed-forward capacitor,
CF, can be added to improve the high frequency response,
as shown in Figure 2. Capacitor CF provides phase lead by
creatingahighfrequencyzerowithR2whichimproves the
phase margin.
route the V line away from noise sources, such as the
FB
inductor or the SW line.
V
OUT
R2
1%
C
F
LTC1701
SGND
V
FB
R1
100k
1%
5pF
The output voltage settling behavior is related to the
stability of the closed-loop system and will demonstrate
the actual overall supply performance. For a detailed
explanation of optimizing the compensation components,
including a review of control loop theory, refer to Applica-
tion Note 76.
1701 F02
Figure 2. Setting the Output Voltage
Transient Response
The OPTI-LOOP compensation allows the transient re-
sponse to be optimized for a wide range of loads and
output capacitors. The availability of the ITH pin not only
allows optimization of the control loop behavior but also
provides a DC coupled and AC filtered closed-loop re-
sponsetestpoint.TheDCstep,risetimeandsettlingatthis
test point truly reflects the closed-loop response. Assum-
ing a predominately second order system, phase margin
and/ordampingfactorcanbeestimatedusingthepercent-
age of overshoot seen at this pin. The bandwidth can also
be estimated by examining the rise time at the pin.
RUN Function
The ITH/RUN pin is a dual purpose pin that provides the
loopcompensationandameans toshutdowntheLTC1701.
Soft-startcanalsobeimplementedwiththis pin.Soft-start
reduces surge currents from V by gradually increasing
IN
theinternalpeakinductorcurrent. Powersupplysequenc-
ing can also be accomplished using this pin.
An external pull-up is required to charge the external
capacitor C3 in Figure 1. Typically, a 1M resistor between
The ITH external components shown in the Figure 1 circuit
will provide an adequate starting point for most applica-
tions. The series R3-C3 filter sets the dominant pole-zero
loop compensation. The values can be modified slightly
V and ITH/RUN is used. When the voltage on ITH/RUN
IN
reaches about 0.8V the LTC1701 begins operating. At this
point the error amplifier pulls up the ITH/RUN pin to the
normal operating range of 1.25V to 2.25V.
7
LTC1701
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APPLICATIO S I FOR ATIO
Soft-start can be implemented by ramping the voltage on
ITH/RUN during start-up as shown in Figure 3(c). As the
voltage on ITH/RUN ramps through its operating range the
internal peak current limit is also ramped at a proportional
linear rate.
1) The V current is the DC supply current given in the
electrical characteristics which excludes MOSFET driver
IN
andcontrolcurrents.V currentresults inasmall(<0.1%)
IN
loss that increases with V , even at no load.
IN
2)Theswitchingcurrentis thesumoftheinternalMOSFET
driver and control currents. The MOSFET driver current
results from switching the gate capacitance of the power
MOSFET. Each time a MOSFET gate is switched from low
During normal operation the voltage on the ITH/RUN pin
will vary from 1.25V to 2.25V depending on the load
current. Pulling the ITH/RUN pin below 0.8V puts the
LTC1701 into a low quiescent current shutdown mode
(IQ < 1µA). This pin can be driven directly from logic as
shown in Figures 3(a) and 3(b).
tohightolowagain, apacketofchargedQmoves fromV
IN
to ground. The resulting dQ/dt is a current out of V that
IN
is typically much larger than the control circuit current. In
continuous mode, IGATECHG = f • QP, where QP is the gate
charge of the internal MOSFET switch.
3.3V OR 5V
I /RUN
TH
I /RUN
TH
D1
3) I2R Losses are predicted from the DC resistances of the
MOSFET and inductor. In continuous mode the average
output current flows through L, but is “chopped” between
the topside internal MOSFET and the Schottky diode. At
low supply voltages where the switch on-resistance is
higher and the switch is on for longer periods due to the
higher duty cycle, the switch losses will dominate. Using
a larger inductance helps minimize these switch losses. At
high supply voltages, these losses are proportional to the
load. I2R losses cause the efficiency to drop at high output
currents.
C
C
C
C
R
C
R
C
(a)
(b)
I
TH
/RUN
R1
D1
C
C
C1
R
C
4) The Schottky diode is a major source of power loss at
high currents and gets worse at low output voltages. The
diode loss is calculated by multiplying the forward voltage
drop times the diode duty cycle multiplied by the load
current.
(c)
1701 F03
Figure 3. ITH/RUN Pin Interfacing
Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and what change would
produce the most improvement. Percent efficiency can be
expressed as:
Other “hidden” losses such as copper trace and internal
battery resistances can account for additional efficiency
degradations in portable systems. It is very important to
include these “system” level losses in the design of a
system. The internal battery and fuse resistance losses
can be minimized by making sure that C has adequate
IN
%Efficiency = 100% – (L1 + L2 + L3 + ...)
charge storage and very low ESR at the switching fre-
quency.Otherlosses includingSchottkyconductionlosses
during dead-time and inductor core losses generally ac-
count for less than 2% total additional loss.
whereL1, L2, etc. aretheindividuallosses as apercentage
of input power.
Although all dissipative elements in the circuit produce
losses, 4 main sources usually account for most of the
losses in LTC1701 circuits: 1) LTC1701 V current,
IN
2) switching losses, 3) I2R losses, 4) Schottky diode
losses.
8
LTC1701
U
W
U U
APPLICATIO S I FOR ATIO
THERMAL CONSIDERATIONS
The junction temperature is given by:
T = TRISE + TAMBIENT
The power handling capability of the device at high ambi-
ent temperatures will be limited by the maximum rated
junction temperature (125°C). It is important to give
careful consideration to all sources of thermal resistance
fromjunctiontoambient.Additionalheatsources mounted
nearby must also be considered.
J
As an example, consider the case when the LTC1701 is in
dropout at an input voltage of 3.3V with a load current of
0.5A. The ON resistance of the P-channel switch is ap-
proximately 0.30Ω. Therefore, power dissipated by the
part is:
For surface mount devices, heat sinking is accomplished
by using the heat spreading capabilities of the PC board
and its copper traces. Copper board stiffeners and plated
through-holes can also be used to spread the heat gener-
ated by power devices.
PD = I2 • RDS(ON) = 75mW
The SOT package junction-to-ambient thermal resistance,
θJA, will be in the range of 125°C/W to 150°C/W. There-
fore, thejunctiontemperatureoftheregulatoroperatingin
a 25°C ambient temperature is approximately:
The following table lists thermal resistance for several
different board sizes and copper areas. All measurements
were taken in still air on 3/32" FR-4 board with one ounce
copper.
T = 0.075 • 150 + 25 = 36°C
J
Remembering that the above junction temperature is
obtained from a RDS(ON) at 25°C, we might recalculate the
junction temperature based on a higher RDS(ON) since it
increases with temperature. However, we can safely as-
sume that the actual junction temperature will not exceed
the absolute maximum junction temperature of 125°C.
Table 1. Measured Thermal Resistance
COPPER AREA
THERMAL RESISTANCE
TOPSIDE* BACKSIDE
BOARD AREA
θ
JA
2
2
2
2
2
2
2
2500mm
1000mm
2500mm
2500mm
2500mm
2500mm
2500mm
2500mm
125°C/W
125°C/W
130°C/W
135°C/W
150°C/W
2
2
2500mm
Board Layout Considerations
2
2
225mm
100mm
2500mm
2
2
2500mm
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1701. These items are also illustrated graphically in
the layout diagram of Figure 4. Check the following in your
layout:
2
2
50mm
2500mm
*Device is mounted on topside.
Calculating Junction Temperature
In a majority of applications, the LTC1701 does not
dissipate much heat due to its high efficiency. However, in
applications where the switching regulator is running at
high duty cycles or the part is in dropout with the switch
turned on continuously (DC), some thermal analysis is
required. The goal of the thermal analysis is to determine
whetherthepowerdissipatedbytheregulatorexceeds the
maximum junction temperature. The temperature rise is
given by:
1. Does the capacitor C connect to the power V (Pin 5)
IN
IN
and GND (Pin 2) as close as possible? This capacitor
provides the AC current to the internal P-channel MOSFET
and its driver.
2. Is the Schottky diode closely connected between the
ground (Pin 2) and switch output (Pin 1)?
3. Are the COUT, L1 and D1 closely connected? The
Schottky anode should connect directly to the input ca-
pacitor ground.
TRISE = PD • θJA
4. The resistor divider, R1 and R2, must be connected
between the (+) plate of COUT and a ground line terminated
nearGND(Pin2). ThefeedbacksignalFBshouldberouted
away from noisy components and traces, such as the SW
line (Pin 1).
where PD is the power dissipated by the regulator and θJA
is the thermal resistance from the junction of the die to the
ambient temperature.
9
LTC1701
U
W
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APPLICATIO S I FOR ATIO
L1
1
2
3
5
4
5. Keep sensitive components away from the SW pin. The
V
V
IN
SW
V
IN
OUT
+
+
input capacitor C , the compensation capacitor CC and all
LTC1701
IN
D1
C
C
IN
OUT
GND
the resistors R1, R2, RC and RS should be routed away
from the SW trace and the components L1 and D1.
R2
R1
R
S
V
FB
I /RUN
TH
R
C
C
C
1701 F04
BOLD LINES INDICATE HIGH CURRENT PATHS
Figure 4. LTC1701 Layout Diagram (See Board Layout Checklist)
U
TYPICAL APPLICATIO S
3- to 4-Cell NiCd/NiMH to 2.5V Converter
L1
4.7µH
V
V
OUT
(2.5V/0.5A)
IN
2.7V TO 5.5V
V
IN
SW
D1
R4
LTC1701
1M
+
+
R2
121k
C1
15µF
C4
0.1µF
C2
15µF
I
TH
/RUN
V
FB
GND
R3
5.1k
R1
121k
C3
330pF
C1, C2: AVX TAJA156M010R
1701 TA01
C4: TAIYO YUDEN EMK107BJ104MA
D1: MBR0520
L1: MURATA LQH3C4R7M24 OR TAIYO YUDEN LEMC3225B4R7M
2mm Nominal Height 1.5V Converter
L1
4.7µH
V
V
OUT
(1.5V/0.5A)
IN
V
IN
SW
2.5V TO 5.5V
D1
R4
LTC1701
1M
+
+
R2
20k
C1
15µF
C4
1µF
C2
22µF
C5
4.7µF
I
TH
/RUN
V
FB
GND
R3
5.1k
R1
100k
C3
330pF
C1: AVX TAJA156M010R
C2: AVX TAJA226M006R
1701 TA02
C4: TAIYO YUDEN LMK212BJ105MG
C5: TAIYO YUDEN JMK212BJ475MG
D1: MBRM120L
L1: MURATA LQH3C4R7M24
10
LTC1701
U
TYPICAL APPLICATIO S
5V to 3.3V Converter with Push-Button On/Off
L1
4.7µH
V
5V
V
OUT
(3.3V/0.5A)
IN
V
IN
SW
D1
ON
LTC1701
+
+
R4
1M
R2
34k
C1
15µF
C4
1µF
C2
22µF
C5
4.7µF
I /RUN
TH
V
FB
GND
R5
5.1M
R3
5.1k
R1
20.5k
OFF
C3
330pF
C1: AVX TAJA156M010R
C2: AVX TAJA226M006R
1701 TA03
C4: TAIYO YUDEN LMK212BJ105MG
C5: TAIYO YUDEN JMK212BJ475MG
D1: MBRM120L
L1: MURATA LQH3C4R7M24
U
Dimensions in inches (millimeters) unless otherwise noted.
PACKAGE DESCRIPTIO
S5 Package
5-Lead Plastic SOT-23
(LTC DWG # 05-08-1633)
2.80 – 3.00
(0.110 – 0.118)
(NOTE 3)
0.95
1.90
(0.074)
REF
(0.037)
REF
2.60 – 3.00
(0.102 – 0.118)
1.50 – 1.75
(0.059 – 0.069)
0.00 – 0.15
(0.00 – 0.006)
0.90 – 1.45
(0.035 – 0.057)
0.35 – 0.55
(0.014 – 0.022)
0.35 – 0.50
(0.014 – 0.020)
FIVE PLACES (NOTE 2)
0.90 – 1.30
(0.035 – 0.051)
S5 SOT-23 0599
0.09 – 0.20
(0.004 – 0.008)
(NOTE 2)
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DIMENSIONS ARE INCLUSIVE OF PLATING
3. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
4. MOLD FLASH SHALL NOT EXCEED 0.254mm
5. PACKAGE EIAJ REFERENCE IS SC-74A (EIAJ)
11
LTC1701
U
TYPICAL APPLICATIO
Single Cell Li-Ion to 3.3V Buck Boost Converter
C6
4.7µF
L1
4.7µH
V
V
OUT
(3.3V)
IN
V
IN
SW
2.5V TO 4.2V
R4
1M
L2
D1
+
+
LTC1701
C1
33µF
C2
15µF
V
I
OUT(MAX)
IN
R2
34k
C4
1µF
CERAMIC
2.5V
3.0V
3.5V
4.0V
4.2V
200mA
225mA
250mA
280mA
290mA
I
/RUN
V
FB
TH
GND
R3
5.1k
R1
20.5k
C3
330pF
C1: AVX TPSB336K006R0600
C2: TAJA156M010R
1701 TA04
C6: TAIYO YUDEN JMK212BJ475MG
D1: MBR0520L
L1, L2: COILTRONICS CTX5-1
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
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LTC1174/LTC1174-3.3/ High Efficiency Step-Down and Inverting DC/DC Converter
LTC1174-5
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3.5V ≤ V ≤ 36V, 16-Pin Narrow SO and SSOP
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Q
High Frequency, High Efficiency, 8-Pin MSOP
LTC1627
SO-8, 2.65V ≤ V ≤ 10V, I
Up to 500mA
IN
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IN
REF
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OUT IN
1701i LT/TP 1299 4K • PRINTED IN USA
LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
12
●
●
(408)432-1900 FAX:(408)434-0507 www.linear-tech.com
LINEAR TECHNOLOGY CORPORATION 1999
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