LTC1772BES6#TRPBF [Linear]

LTC1772B - Constant Frequency Current Mode Step-Down DC/DC Controller in SOT-23; Package: SOT; Pins: 6; Temperature Range: -40°C to 85°C;


型号：	LTC1772BES6#TRPBF
厂家：	Linear
描述：	LTC1772B - Constant Frequency Current Mode Step-Down DC/DC Controller in SOT-23; Package: SOT; Pins: 6; Temperature Range: -40°C to 85°C 开关光电二极管控制器
文件：	总12页 (文件大小：190K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

LTC1772B

Constant Frequency

Current Mode Step-Down

DC/DC Controller in SOT-23

FEATURES

■

DESCRIPTIO

Burst Mode^TMOperation Disabled for Lower Output

The LTC^®1772B is a constant frequency current mode

step-down DC/DC controller providing excellent AC and

DC load and line regulation. The device incorporates an

accurateundervoltagelockoutfeaturethatshutsdownthe

LTC1772B when the input voltage falls below 2.0V.

Ripple at Light Loads

■

High Efficiency: Up to 94%

■

High Output Currents Easily Achieved

■

Wide V_INRange: 2.5V to 9.8V

■

Constant Frequency 550kHz Operation

The LTC1772B provides a ±2.5% output voltage accuracy

and consumes only 270µA of quiescent current. In shut-

down, the device draws a mere 8µA.

■

Low Dropout: 100% Duty Cycle

■

Output Voltage down to 0.8V

■

Current Mode Operation for Excellent Line and Load

To further maximize the life of a battery source, the

external P-channel MOSFET is turned on continuously in

dropout (100% duty cycle). High constant operating

frequency of 550kHz allows the use of a small external

inductor.

Transient Response

■

Shutdown Mode Draws Only 8µA Supply Current

Tiny 6-Lead SOT-23 Package

APPLICATIO S

The LTC1772B is available in a small footprint 6-lead

SOT-23.

■

One or Two Lithium-Ion-Powered Applications

■

Cellular Telephones

For a Burst Mode operation enabled version of the

LTC1772B, please refer to the LTC1772 data sheet.

■

Wireless Devices

Portable Computers

Distributed 3.3V, 2.5V or 1.8V Power Systems

■

, LTC and LT are registered trademarks of Linear Technology Corporation.

Burst Mode is a trademark of Linear Technology Corporation.

TYPICAL APPLICATION

Efficiency vs Load Current*

100

LTC1772

BURST MODE

OPERATION

2.5V

10µF

10V

TO 9.8V

0.03Ω

4.7µH

/RUN PGATE

LTC1772B

2.5V

OUT

LTC1772B

NON-BURST MODE

OPERATION

10k

220pF

C2A

47µF

C2B

1µF

10V

GND

–

174k

SENSE

= 3.6V

OUT

C1: TAIYO YUDEN LMK325BJ106K-T

C2A: SANYO 6TPA47M

C2B: AVX 0805ZC105KAT1A

D1: MOTOROLA MBRM120T3

L1: MURATA LQN6C-4R7

M1: FAIRCHILD FDC638P

R1: IRC LRC-LR1206-01-R030F

= 2.5V

80.6k

100

1000

10000

LOAD CURRENT (mA)

*OUTPUT RIPPLE WAVEFORMS FOR THE CIRCUIT

OF FIGURE 1 APPEAR IN FIGURE 2.

1772 F01a

1772 F01b

Figure 1. High Efficiency, High Output Current 2.5V/2A Regulator

LTC1772B

W W

U W

W U

ABSOLUTE MAXIMUM RATINGS

PACKAGE/ORDER INFORMATION

(Note 1)

Input Supply Voltage (V_IN).........................–0.3V to 10V

SENSE^–, PGATE Voltages............. –0.3V to (V_IN+ 0.3V)

V_FB, I_TH/RUN Voltages ..............................–0.3V to 2.4V

PGATE Peak Output Current (<10µs) ....................... 1A

Storage Ambient Temperature Range ... –65°C to 150°C

Operating Temperature Range (Note 2) ... –40°C to 85°C

Junction Temperature (Note 3)............................. 150°C

Lead Temperature (Soldering, 10 sec).................. 300°C

ORDER PART

NUMBER

TOP VIEW

LTC1772BES6

/RUN 1

GND 2

6 PGATE

5 V

4 SENSE

–

S6 PART MARKING

LTVU

S6 PACKAGE

6-LEAD PLASTIC SOT-23

T_JMAX= 150°C, θ_JA= 230°C/ W

Consult factory for parts specified with wider operating temperature ranges.

The ● denotes specifications that apply over the full operating temperature

ELECTRICAL CHARACTERISTICS

range, otherwise specifications are at T_A= 25°C. V_IN= 4.2V unless otherwise specified. (Note 2)

PARAMETER

CONDITIONS

Typicals at V = 4.2V (Note 4)

MIN

TYP

MAX

UNITS

Input DC Supply Current

Normal Operation

Shutdown

2.4V ≤ V ≤ 9.8V, PGATE Logic High

270

420

µA

2.4V ≤ V ≤ 9.8V, V /RUN = 0V

ITH

UVLO

< UVLO Threshold

Undervoltage Lockout Threshold

Falling

Rising

●

1.55

1.85

2.00

2.10

2.35

2.40

Shutdown Threshold (at I /RUN)

0.15

0.25

0.35

0.5

0.55

0.85

Start-Up Current Source

/RUN = 0V

ITH

µA

Regulated Feedback Voltage

0°C to 70°C (Note 5)

–40°C to 85°C (Note 5)

●

0.780

0.770

0.800

0.820

0.830

Output Voltage Line Regulation

Output Voltage Load Regulation

2.4V ≤ V ≤ 9.8V (Note 5)

0.05

mV/V

/RUN Sinking 5µA (Note 5)

/RUN Sourcing 5µA (Note 5)

2.5

mV/µA

Input Current

(Note 5)

0.860

Overvoltage Protect Threshold

Overvoltage Protect Hysteresis

Oscillator Frequency

Measured at V

0.820

500

0.895

= 0.8V

= 0V

550

120

650

kHz

Gate Drive Rise Time

Gate Drive Fall Time

= 3000pF

LOAD

Peak Current Sense Voltage

(Note 6)

105

Note 1: Absolute Maximum Ratings are those values beyond which the life

T = T + (P • θ °C/W)

of a device may be impaired.

Note 4: Dynamic supply current is higher due to the gate charge being

Note 2: The LTC1772BE is guaranteed to meet specifications from 0°C to

70°C. Specifications over the –40°C to 85°C operating temperature range

are assured by design, characterization and correlation with statistical

process controls.

delivered at the switching frequency.

Note 5: The LTC1772B is tested in a feedback loop that servos V to the

output of the error amplifier.

Note 6: Peak current sense voltage is reduced dependent on duty cycle to

Note 3: T is calculated from the ambient temperature T and power

a percentage of value as given in Figure 2.

dissipation P according to the following formula:

LTC1772B

TYPICAL PERFORMANCE CHARACTERISTICS

Undervoltage Lockout Trip

Voltage vs Temperature

Reference Voltage

vs Temperature

Normalized Oscillator Frequency

vs Temperature

825

820

815

810

805

800

795

790

785

780

775

2.24

2.20

2.16

2.12

2.08

2.04

2.00

1.96

1.92

1.88

1.84

= 4.2V

FALLING

= 4.2V

–2

–4

–6

–8

–10

–55 –35 –15

25 45 65 85 105 125

–55 –35 –15

25 45 65 85 105 125

–55 –35 –15

25 45 65 85 105 125

TEMPERATURE (°C)

1772 G01

1772 G03

1772 G02

Maximum (V_IN– SENSE^–) Voltage

vs Duty Cycle

Shutdown Threshold

vs Temperature

600

560

520

480

440

400

360

320

280

240

200

120

110

100

= 4.2V

= 25°C

60 70

–55 –35 –15

85 105 125

20 30 40 50

80 90 100

TEMPERATURE (°C)

DUTY CYCLE (%)

1772 G04

1772 G05

PIN FUNCTIONS

I_TH/RUN (Pin 1): This pin performs two functions. It

servesastheerroramplifiercompensationpointaswellas

the run control input. Nominal voltage range for this pin is

0.85V to 1.9V. Forcing this pin below 0.35V causes the

device to be shut down. In shutdown all functions are

disabled and the PGATE pin is held high.

SENSE^–(Pin 4): The Negative Input to the Current Com-

parator.

V_IN(Pin5):SupplyPin. MustbecloselydecoupledtoGND

Pin 2.

PGATE (Pin 6): Gate Drive for the External P-Channel

MOSFET. This pin swings from 0V to V_IN.

GND (Pin 2): Ground Pin.

_FB(Pin 3): Receives the feedback voltage from an exter-

nal resistive divider across the output.

LTC1772B

U W

FUNCTIONAL DIAGRA

–

SENSE

–

15mV

OSC

ICMP

RS1

PGATE

SWITCHING

LOGIC AND

BLANKING

CIRCUIT

SLOPE

COMP

–

FREQ

OVP

FOLDBACK

–

0.3V

SHORT-CIRCUIT

DETECT

REF

60mV

EAMP

REF

–

0.8V

0.5µA

/RUN

–

0.35V

–

SHDN

CMP

VOLTAGE

REFERENCE

REF

0.8V

GND

UNDERVOLTAGE

LOCKOUT

1.2V

1772FD

(Refer to Functional Diagram)

OPERATIO

load current increases, it causes a slight decrease in V_FB

relative to the 0.8V reference, which in turn causes the

I_TH/RUN voltage to increase until the average inductor

current matches the new load current.

Main Control Loop

The LTC1772B is a constant frequency current mode

switchingregulator.Duringnormaloperation,theexternal

P-channel power MOSFET is turned on each cycle when

the oscillator sets the RS latch (RS1) and turned off when

the current comparator (ICMP) resets the latch. The peak

inductor current at which ICMP resets the RS latch is

controlled by the voltage on the I_TH/RUN pin, which is the

output of the error amplifier EAMP. An external resistive

divider connected between V_OUTand ground allows the

EAMPtoreceiveanoutputfeedbackvoltageV_FB.Whenthe

ThemaincontrolloopisshutdownbypullingtheI_TH/RUN

pin low. Releasing I_TH/RUN allows an internal 0.5µA

current source to charge up the external compensation

network. When the I_TH/RUN pin reaches 0.35V, the main

control loop is enabled with the I_TH/RUN voltage then

pulled up to its zero current level of approximately 0.85V.

Astheexternalcompensationnetworkcontinuestocharge

LTC1772B

(Refer to Functional Diagram)

OPERATIO

up, the corresponding output current trip level follows,

allowing normal operation.

Dropout Operation

When the input supply voltage decreases towards the

output voltage, the rate of change of inductor current

during the ON cycle decreases. This reduction means that

the external P-channel MOSFET will remain on for more

thanoneoscillatorcyclesincetheinductorcurrenthasnot

ramped up to the threshold set by EAMP. Further reduc-

tion in input supply voltage will eventually cause the

P-channel MOSFET to be turned on 100%, i.e., DC. The

outputvoltagewillthenbedeterminedbytheinputvoltage

minus the voltage drop across the MOSFET, the sense

resistor and the inductor.

Comparator OVP guards against transient overshoots

>7.5% by turning off the external P-channel power

MOSFET and keeping it off until the fault is removed.

Low Load Current Operation

Under very light load current conditions, the I_TH/RUN pin

voltage will be very close to the zero current level of 0.85V.

As the load current decreases further, an internal offset at

the current comparator input will assure that the current

comparator remains tripped (even at zero load current)

and the regulator will start to skip cycles, as it must, in

order to maintain regulation. This behavior allows the

regulator to maintain constant frequency down to very

light loads, resulting in less low frequency noise genera-

tion over a wide load current range.

Undervoltage Lockout

TopreventoperationoftheP-channelMOSFETbelowsafe

input voltage levels, an undervoltage lockout is incorpo-

rated into the LTC1772B. When the input supply voltage

drops below approximately 2.0V, the P-channel MOSFET

and all circuitry is turned off except the undervoltage

block, which draws only several microamperes.

Figure 2 illustrates this result for the circuit of Figure 1

using both an LTC1772 in Burst Mode operation and an

LTC1772B (non-Burst Mode operation). At an output

current of 100mA, the Burst Mode operation part exhibits

an output ripple of approximately 60mV_P-P, whereas the

non-Burst Mode operation part has an output ripple of

only20mV_P-P.Atloweroutputcurrentlevels,theimprove-

ment is even greater. This comes at a tradeoff of lower

efficiency for the non-Burst Mode operation part (see

Figure 1). Also notice the constant frequency operation of

the LTC1772B, even at 5% of maximum output current.

Short-Circuit Protection

Whentheoutputisshortedtoground, thefrequencyofthe

oscillator will be reduced to about 120kHz. This lower

frequency allows the inductor current to safely discharge,

thereby preventing current runaway. The oscillator’s fre-

quency will gradually increase to its designed rate when

the feedback voltage again approaches 0.8V.

V_OUTRipple for Figure 1 Circuit Using

LTC1772B Non-Burst Mode Operation.

V_OUTRipple for Figure 1 Circuit Using

LTC1772 Burst Mode Operation.

20mV /DIV

1772 F02b

1772 F02a

5µs/DIV

= 3.6V

5µs/DIV

= 3.6V

OUT

= 2.5V

= 100mA

OUT

Figure 2. Output Ripple Waveforms for the Circuit of Figure 1.

LTC1772B

(Refer to Functional Diagram)

OPERATIO

compensation begins and effectively reduces the peak

inductor current. The amount of reduction is given by the

curves in Figure 3.

Overvoltage Protection

As a further protection, the overvoltage comparator in the

LTC1772B will turn the external MOSFET off when the

feedback voltage has risen 7.5% above the reference

voltage of 0.8V. This comparator has a typical hysteresis

of 20mV.

110

100

Slope Compensation and Inductor’s Peak Current

The inductor’s peak current is determined by:

= 0.4I

RIPPLE

AT 5% DUTY CYCLE

= 0.2I

RIPPLE

_ITH– 0.85

AT 5% DUTY CYCLE

I_PK

= 4.2V

10 R_SENSE

(

)

10 20 30 40 50 60 70 80 90 100

DUTY CYCLE (%)

when the LTC1772B is operating below 40% duty cycle.

However, once the duty cycle exceeds 40%, slope

1772 F03

Figure 3. Maximum Output Current vs Duty Cycle

W U U

APPLICATIONS INFORMATION

However,foroperationthatisabove40%dutycycle,slope

compensation effect has to be taken into consideration to

selecttheappropriatevaluetoprovidetherequiredamount

of current. Using Figure 3, the value of R_SENSEis:

The basic LTC1772B application circuit is shown

in Figure 1. External component selection is driven by the

load requirement and begins with the selection of L1 and

R_SENSE(= R1). Next, the power MOSFET, M1 and the

output diode D1 is selected followed by C_IN(= C1)and

C_OUT(= C2).

(0.0875)

R_SENSE

I_OUT100

(

)

R_SENSESelection for Output Current

R_SENSEis chosen based on the required output current.

Withthecurrentcomparatormonitoringthevoltagedevel-

oped across R_SENSE, the threshold of the comparator

determines the inductor’s peak current. The output cur-

rent the LTC1772B can provide is given by:

Inductor Value Calculation

The operating frequency and inductor selection are inter-

related in that higher operating frequencies permit the use

of a smaller inductor for the same amount of inductor

ripplecurrent. However, thisisattheexpenseofefficiency

due to an increase in MOSFET gate charge losses.

0.105 I_RIPPLE

I_OUT

−

R_SENSE

The inductance value also has a direct effect on ripple

current. The ripple current, I_RIPPLE, decreases with higher

inductance or frequency and increases with higher V_INor

where I_RIPPLEis the inductor peak-to-peak ripple current

(see Inductor Value Calculation section).

V_OUT. The inductor’s peak-to-peak ripple current is given

by:

A reasonable starting point for setting ripple current is

I_RIPPLE= (0.4)(I_OUT). Rearranging the above equation, it

becomes:

V − V_OUT

_OUT+ V_D

I_RIPPLE

V + V_D

f L

( )

0.0875

I_OUT

R_SENSE

for Duty Cycle < 40%

LTC1772B

W U U

APPLICATIONS INFORMATION

P_P

wherefistheoperatingfrequency.Acceptinglargervalues

of I_RIPPLEallows the use of low inductances, but results in

higher output voltage ripple and greater core losses. A

reasonable starting point for setting ripple current is

_RIPPLE=0.4(I_OUT(MAX)).Remember,themaximumI_RIPPLE

occurs at the maximum input voltage.

R_DS(ON)

DC=100%

I_OUT(MAX)1+ δp

) (

(

)

where P_Pis the allowable power dissipation and δp is the

temperature dependency of R_DS(ON). (1 + δp) is generally

given for a MOSFET in the form of a normalized R_DS(ON)vs

temperature curve, but δp = 0.005/°C can be used as an

approximation for low voltage MOSFETs.

The ripple current is normally set such that the inductor

current is continuous down to approximately 1/4 of maxi-

mum load current. This results in:

In applications where the maximum duty cycle is less than

100% and the LTC1772B is in continuous mode, the

0.03

R_SENSE

_DS(ON)is governed by:

I_RIPPLE

≤

P_P

R_DS(ON)

This implies a minimum inductance of:

1+ δp

DC I_OUT

)

(

)

(

where DC is the maximum operating duty cycle of the

LTC1772B.

V − V_OUT

_OUT+ V_D

L_MIN

V + V_D

0.03

R_SENSE

Output Diode Selection

(Use V_IN(MAX)= V_IN)

The catch diode carries load current during the off-time.

The average diode current is therefore dependent on the

P-channel switch duty cycle. At high input voltages the

diode conducts most of the time. As V_INapproaches V_OUT

the diode conducts only a small fraction of the time. The

most stressful condition for the diode is when the output

is short-circuited. Under this condition the diode must

safelyhandleI_PEAKatcloseto100%dutycycle. Therefore,

itisimportanttoadequatelyspecifythediodepeakcurrent

and average power dissipation so as not to exceed the

diode ratings.

A smaller value than L_MINcould be used in the circuit;

however, the inductor current transitioning from continu-

ous to discontinuous will occur at a higher load current.

Power MOSFET Selection

An external P-channel power MOSFET must be selected

for use with the LTC1772B. The main selection criteria for

the power MOSFET are the threshold voltage V_GS(TH)and

the “on” resistance R_DS(ON), reverse transfer capacitance

C_RSSand total gate charge.

Under normal load conditions, the average current con-

ducted by the diode is:

SincetheLTC1772Bisdesignedforoperationdowntolow

input voltages, a logic level threshold MOSFET (R_DS(ON)

guaranteed at V_GS= 2.5V) is required for applications that

workclosetothisvoltage.WhentheseMOSFETsareused,

make sure that the input supply to the LTC1772B is less

than the absolute maximum V_GSrating, typically 8V.

V − V_OUT

V + V_D

I_D=

I_OUT

The allowable forward voltage drop in the diode is calcu-

lated from the maximum short-circuit current as:

The required minimum R_DS(ON)of the MOSFET is gov-

erned by its allowable power dissipation. For applications

that may operate the LTC1772B in dropout, i.e., 100%

duty cycle, at its worst case the required R_DS(ON)is given

by:

P_D

V_F≈

I_SC(MAX)

LTC1772B

W U U

APPLICATIONS INFORMATION

where P_Dis the allowable power dissipation and will be

determined by efficiency and/or thermal requirements.

where f is the operating frequency, C_OUTis the output

capacitance and I_RIPPLEis the ripple current in the induc-

tor. The output ripple is highest at maximum input voltage

since ∆I_Lincreases with input voltage.

A fast switching diode must also be used to optimize

efficiency. Schottky diodes are a good choice for low

forwarddropandfastswitchingtimes. Remembertokeep

lead length short and observe proper grounding (see

Board Layout Checklist) to avoid ringing and increased

dissipation.

Manufacturers such as Nichicon, United Chemicon and

Sanyoshouldbeconsideredforhighperformancethrough-

hole capacitors. The OS-CON semiconductor dielectric

capacitor available from Sanyo has the lowest ESR (size)

product of any aluminum electrolytic at a somewhat

higher price. Once the ESR requirement for C_OUThas been

met, the RMS current rating generally far exceeds the

I_RIPPLE(P-P)requirement.

C_INand C_OUTSelection

In continuous mode, the source current of the P-channel

MOSFET is a square wave of duty cycle (V_OUT+ V_D)/

(V_IN+ V_D). To prevent large voltage transients, a low ESR

input capacitor sized for the maximum RMS current must

beused. ThemaximumRMScapacitorcurrentisgivenby:

Low Supply Operation

1/2

]

Although the LTC1772B can function down to approxi-

mately 2.0V, the maximum allowable output current is

reducedwhenV_INdecreasesbelow3V. Figure4showsthe

amount of change as the supply is reduced down to 2V.

Also shown in Figure 4 is the effect of V_INon V_REFas V_IN

goes below 2.3V.

V_OUTV − V_OUT

(

)

[

C_INRequired I_RMS≈ I_MAX

This formula has a maximum value at V_IN= 2V_OUT, where

I_RMS= I_OUT/2. This simple worst-case condition is com-

monlyusedfordesignbecauseevensignificantdeviations

donotoffermuchrelief.Notethatcapacitormanufacturer’s

ripplecurrentratingsareoftenbasedon2000hoursoflife.

This makes it advisable to further derate the capacitor, or

to choose a capacitor rated at a higher temperature than

required. Several capacitors may be paralleled to meet the

size or height requirements in the design. Due to the high

operating frequency of the LTC1772B, ceramic capacitors

can also be used for C_IN. Always consult the manufacturer

if there is any question.

105

REF

100

ITH

The selection of C_OUTis driven by the required effective

series resistance (ESR). Typically, once the ESR require-

ment is satisfied, the capacitance is adequate for filtering.

The output ripple (∆V_OUT) is approximated by:

2.0

2.2

2.4

2.6

2.8

3.0

INPUT VOLTAGE (V)

1772 F03

Figure 4. Line Regulation of V_REFand V_ITH

∆V_OUT≈ I_RIPPLEESR +

4fC_OUT

LTC1772B

W U U

APPLICATIONS INFORMATION

Setting Output Voltage

1. The V_INcurrent is the DC supply current, given in the

electricalcharacteristics, thatexcludesMOSFETdriver

and control currents. V_INcurrent results in a small loss

which increases with V_IN.

The regulated output voltage is determined by:

V_OUT= 0.8 1+

2. MOSFET gate charge current results from switching

the gate capacitance of the power MOSFET. Each time

a MOSFET gate is switched from low to high to low

again,apacketofchargedQmovesfromV_INtoground.

The resulting dQ/dt is a current out of V_INwhich is

typically much larger than the DC supply current. In

continuous mode, I_GATECHG= f(Qp).

3. I²R losses are predicted from the DC resistances of the

MOSFET, inductor and current shunt. In continuous

mode the average output current flows through L but

is “chopped” between the P-channel MOSFET (in se-

ries with R_SENSE)and the output diode. The MOSFET

R_DS(ON)plus R_SENSEmultiplied by duty cycle can be

summedwiththeresistancesofLandR_SENSEtoobtain

I²R losses.

Formostapplications, an80kresistorissuggestedforR1.

To prevent stray pickup, locate resistors R1 and R2 close

to LTC1772B.

OUT

LTC1772B

1772 F04

Figure 5. Setting Output Voltage

Efficiency Considerations

The efficiency of a switching regulator is equal to the

output power divided by the input power times 100%. It is

oftenusefultoanalyzeindividuallossestodeterminewhat

is limiting the efficiency and which change would produce

the most improvement. Efficiency can be expressed as:

4. The output diode is a major source of power loss at

high currents and gets worse at high input voltages.

The diode loss is calculated by multiplying the forward

voltage times the diode duty cycle multiplied by the

load current. For example, assuming a duty cycle of

50% with a Schottky diode forward voltage drop of

0.4V, the loss increases from 0.5% to 8% as the load

current increases from 0.5A to 2A.

Efficiency = 100% – (η1 + η2 + η3 + ...)

where η1, η2, etc. are the individual losses as a percent-

age of input power.

5. Transition losses apply to the external MOSFET and

increase at higher operating frequencies and input

voltages. Transition losses can be estimated from:

Although all dissipative elements in the circuit produce

losses, four main sources usually account for most of the

losses in LTC1772B circuits: 1) LTC1772B DC bias cur-

rent, 2) MOSFET gate charge current, 3) I²R losses and 4)

voltage drop of the output diode.

Transition Loss = 2(V_IN)²I_{O(MAX) RSS}

(f)

Other losses including C_INand C_OUTESR dissipative

losses, and inductor core losses, generally account for

less than 2% total additional loss.

LTC1772B

W U U

APPLICATIONS INFORMATION

Foldback Current Limiting

LTC1772B. These items are illustrated graphically in the

layout diagram in Figure 7. Check the following in your

layout:

AsdescribedintheOutputDiodeSelection,theworst-case

dissipation occurs with a short-circuited output when the

diode conducts the current limit value almost continu-

ously. To prevent excessive heating in the diode, foldback

current limiting can be added to reduce the current in

proportion to the severity of the fault.

1. IstheSchottkydiodecloselyconnectedbetweenground

(Pin 2) and drain of the external MOSFET?

2. Does the (+) plate of C_INconnect to the sense resistor

as closely as possible? This capacitor provides AC

current to the MOSFET.

Foldbackcurrentlimitingisimplementedbyaddingdiodes

D_FB1and D_FB2between the output and the I_TH/RUN pin as

shown in Figure 6. In a hard short (V_OUT= 0V), the current

will be reduced to approximately 50% of the maximum

output current.

3. Is the input decoupling capacitor (0.1µF) connected

closely between V_IN(Pin 5) and ground (Pin 2)?

4. Connect the end of R_SENSEas close to V_IN(Pin 5) as

possible. The V_INpin is the SENSE⁺of the current

comparator.

5. Is the trace from SENSE^–(Pin 4) to the Sense resistor

OUT

LTC1772B

/RUN V

FB1

FB2

kept short? Does the trace connect close to R_SENSE

6. Keep the switching node PGATE away from sensitive

small signal nodes.

1772 F05

Figure 6. Foldback Current Limiting

7. Does the V_FBpin connect directly to the feedback

resistors? The resistive divider R1 and R2 must be

connected between the (+) plate of C_OUTand signal

ground.

PC Board Layout Checklist

When laying out the printed circuit board, the following

checklist should be used to ensure proper operation of the

/RUN PGATE

LTC1772B

SENSE

ITH

GND

OUT

0.1µF

OUT

–

SENSE

ITH

1772 F06

BOLD LINES INDICATE HIGH CURRENT PATHS

Figure 7. LTC1772B Layout Diagram (See PC Board Layout Checklist)

LTC1772B

TYPICAL APPLICATIO S

LTC1772B High Efficiency, Small Footprint 3.3V to 1.8V/0.5A Regulator

3.3V

10µF

10V

0.15Ω

10µH

/RUN PGATE

LTC1772B

1.8V

0.5A

OUT

10k

47µF

GND

–

220pF

SENSE

100k

C1: TAIYO YUDEN CERAMIC

LMK325BJ106K-T

C2: SANYO POSCAP 6TPA47M R1: DALE 0.25W

D1: MOTOROLA MBRM120T3

L1: COILTRONICS UP1B-100

M1: Si3443DV

80.6k

1772 TA02

LTC1772B 5V/0.5A Flyback Regulator

2.5V

TO 9.8V

0.033Ω

10µF

10V

/RUN PGATE

LTC1772B

10k

GND

–

220pF

SENSE

OUT

•

0.5A

100µF

10V

×2

10µH

52.3k

•

10k

C1: TAIYO YUDEN CERAMIC

LMK325BJ106K-T

M1: Si9803

R1: DALE 0.25W

C2: AVXTPSE107M010R0100 T1: COILTRONICS CTX10-4

D1: IR10BQ015

1772 TA04

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-

tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.

LTC1772B

TYPICAL APPLICATIONS

LTC1772B 3.3V to 5V/1A Boost Regulator

0.033Ω

3.3V

47µF

16V

×2

4.7µH

OUT

100µF

10V

×2

/RUN PGATE

LTC1772B

10k

GND

–

220pF

SENSE

422k

C1: AVXTPSE476M016R0047 L1: MURATA LQN6C-4R7 U1: FAIRCHILD NC7SZ04

80.6k

C2: AVXTPSE107M010R0100 M1: Si9804

ALSO SEE LTC1872

D1: IR10BQ015

R1: DALE 0.25W

FOR THIS APPLICATION

1772 TA03

PACKAGE DESCRIPTION ^{Dimensions in inches (millimeters) unless otherwise noted.}

S6 Package

6-Lead Plastic SOT-23

(LTC DWG # 05-08-1634)

2.6 – 3.0

(0.110 – 0.118)

2.80 – 3.00

(0.110 – 0.118)

(NOTE 3)

1.50 – 1.75

(0.059 – 0.069)

0.00 – 0.15

(0.00 – 0.006)

0.90 – 1.45

(0.035 – 0.057)

0.35 – 0.55

(0.014 – 0.022)

0.35 – 0.50

(0.014 – 0.020)

SIX PLACES (NOTE 2)

0.90 – 1.30

(0.035 – 0.051)

0.95

(0.037)

REF

0.09 – 0.20

(0.004 – 0.008)

(NOTE 2)

1.90

(0.074)

REF

NOTE:

S6 SOT-23 0898

1. DIMENSIONS ARE IN MILLIMETERS

2. DIMENSIONS ARE INCLUSIVE OF PLATING

3. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR

4. MOLD FLASH SHALL NOT EXCEED 0.254mm

5. PACKAGE EIAJ REFERENCE IS SC-74A (EIAJ)

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OUT OUT

Single, High Efficiency, Low Noise Synchronous Switching Controller

Ultra-Low Supply Current Step-Down DC/DC Controller

High Efficiency 5V to 3.3V Conversion at up to 15A

10µA Supply Current, 93% Efficiency,

1.23V ≤ V

≤ 18V; 2.8V ≤ V ≤ 20V

OUT

LTC1772

Constant Frequency Current Mode Step-Down

DC/DC Controller in SOT-23

With Burst Mode Operation for Higher Efficiency

at Light Load Current

LTC1773

LTC1872

95% Efficient Synchronous Step-Down Controller

SOT-23 Step-Up Controller

2.65V ≤ V ≤ 8.5V; 0.8V ≤ V

≤ V ; Current Mode; 550kHz

OUT IN

2.5V ≤ V ≤ 9.8V; 550kHz; 90% Efficiency

No R_SENSEis a trademark of Linear Technology Corporation.

1772bf LT/TP 0201 4K • PRINTED IN USA

LinearTechnology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

●

LINEAR TECHNOLOGY CORPORATION 1999

(408)432-1900 FAX:(408)434-0507 www.linear-tech.com

LTC1772BES6#TRPBF [Linear]

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