LTC1772BIS6-TR [Linear]
Constant Frequency Current Mode Step-Down DC/DC Controller in SOT-23; 恒定频率电流模式降压型DC / DC采用SOT -23控制器
| 型号： | LTC1772BIS6-TR |
| 厂家： | 
Linear |
| 描述： | Constant Frequency Current Mode Step-Down DC/DC Controller in SOT-23 |
| 文件： | 总16页 (文件大小：218K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LTC1772B
Constant Frequency
Current Mode Step-Down
DC/DC Controller in SOT-23
FEATURES
DESCRIPTION
Burst Mode® Operation Disabled for Lower Output
The LTC®1772B is a constant frequency current mode
step-down DC/DC controller providing excellent AC and
DC load and line regulation. The device incorporates an
accurateundervoltagelockoutfeaturethatshutsdownthe
LTC1772B when the input voltage falls below 2.0V.
n
Ripple at Light Loads
n
High Efficiency: Up to 94%
n
High Output Currents Easily Achieved
n
Wide V Range: 2.5V to 9.8V
IN
n
n
n
n
Constant Frequency 550kHz Operation
Low Dropout: 100% Duty Cycle
The LTC1772B provides a 2.5% output voltage accuracy
and consumes only 270μA of quiescent current. In shut-
down, the device draws a mere 8μA.
Output Voltage Down to 0.8V
Current Mode Operation for Excellent Line and Load
Transient Response
Shutdown Mode Draws Only 8µA Supply Current
Tiny 6-Lead TSOT-23 Package
To further maximize the life of a battery source, the
external P-channel MOSFET is turned on continuously
in dropout (100% duty cycle). High constant operating
frequency of 550kHz allows the use of a small external
inductor.
n
n
APPLICATIONS
The LTC1772B is available in a small footprint 6-lead
TSOT-23.
n
One or Two Lithium-Ion-Powered Applications
n
Cellular Telephones
For a Burst Mode operation enabled version of the
LTC1772B, please refer to the LTC1772 data sheet.
L, LT, LTC, LTM and Burst Mode are registered trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
n
Wireless Devices
Portable Computers
Distributed 3.3V, 2.5V or 1.8V Power Systems
n
n
TYPICAL APPLICATION
Efficiency vs Load Current*
100
V
IN
2.5V
LTC1772
Burst Mode
OPERATION
C1
10μF
10V
TO 9.8V
R1
0.03Ω
95
90
85
80
75
70
65
60
1
6
L1
4.7μH
I
/RUN PGATE
LTC1772B
M1
TH
V
2.5V
2A
OUT
10k
220pF
LTC1772B
NON-Burst Mode
OPERATION
+
C2A
47μF
6V
C2B
1μF
10V
2
3
5
4
GND
V
D1
IN
–
174k
V
SENSE
FB
C1: TAIYO YUDEN LMK325BJ106K-T
C2A: SANYO 6TPA47M
C2B: AVX 0805ZC105KAT1A
D1: MOTOROLA MBRM120T3
L1: MURATA LQN6C-4R7
M1: FAIRCHILD FDC638P
R1: IRC LRC-LR1206-01-R030F
V
V
= 3.6V
IN
OUT
80.6k
= 2.5V
10
100
1000
10000
LOAD CURRENT (mA)
*OUTPUT RIPPLE WAVEFORMS FOR THE CIRCUIT
OF FIGURE 1 APPEAR IN FIGURE 2.
1772 F01a
1772 F01b
Figure 1. High Efficiency, High Output Current 2.5V/2A Regulator
1772bfa
1
LTC1772B
ABSOLUTE MAXIMUM RATINGS
PIN CONFIGURATION
(Note 1)
Input Supply Voltage (V ).........................–0.3V to 10V
TOP VIEW
IN
–
SENSE , PGATE Voltages...............–0.3V to (V + 0.3V)
IN
I
TH
/RUN 1
GND 2
6 PGATE
V , I /RUN Voltages............................... –0.3V to 2.4V
FB TH
5 V
IN
–
PGATE Peak Output Current (<10µs) ........................ 1A
Storage Ambient Temperature Range.....–65°C to 150°C
Operating Temperature Range (Note 2).... –40°C to 85°C
Junction Temperature (Note 3) ............................ 150°C
Lead Temperature (Soldering, 10 sec) .................. 300°C
V
3
4 SENSE
FB
S6 PACKAGE
6-LEAD PLASTIC TSOT-23
T
= 150°C, θ = 230°C/W
JA
JMAX
ORDER INFORMATION
LEAD FREE FINISH
LTC1772BES6#PBF
LTC1772BIS6#PBF
LEAD BASED FINISH
LTC1772BES6
TAPE AND REEL
PART MARKING*
LTVU
PACKAGE DESCRIPTION
TEMPERATURE RANGE
–40°C to 85°C
LTC1772BES6#TRPBF
LTC1772BIS6#TRPBF
TAPE AND REEL
6-Lead Plastic TSOT Package
6-Lead Plastic TSOT Package
PACKAGE DESCRIPTION
LTVU
–40°C to 85°C
PART MARKING*
LTVU
TEMPERATURE RANGE
–40°C to 85°C
LTC1772BES6#TR
LTC1772BIS6#TR
6-Lead Plastic TSOT Package
6-Lead Plastic TSOT Package
LTC1772BIS6
LTVU
–40°C to 85°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 4.2V unless otherwise specified. (Note 2)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Input DC Supply Current
Normal Operation
Shutdown
Typicals at V = 4.2V (Note 4)
IN
2.4V ≤ V ≤ 9.8V, PGATE Logic High
270
8
6
420
22
10
μA
μA
μA
IN
2.4V ≤ V ≤ 9.8V, V /RUN = 0V
IN
ITH
UVLO
V
IN
< UVLO Threshold
l
l
Undervoltage Lockout Threshold
V
V
Falling
Rising
1.55
1.85
2.00
2.10
2.35
2.40
V
V
IN
IN
Shutdown Threshold (at I /RUN)
0.15
0.25
0.35
0.5
0.55
0.85
V
TH
Start-Up Current Source
V
/RUN = 0V
ITH
μA
l
l
Regulated Feedback Voltage
0°C to 70°C (Note 5)
–40°C to 85°C (Note 5)
0.780
0.770
0.800
0.800
0.820
0.830
V
V
Output Voltage Line Regulation
Output Voltage Load Regulation
2.4V ≤ V ≤ 9.8V (Note 5)
0.05
mV/V
IN
I /RUN Sinking 5μA (Note 5)
TH
I /RUN Sourcing 5μA (Note 5)
TH
2.5
2.5
mV/μA
mV/μA
1772bfa
2
LTC1772B
ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 4.2V unless otherwise specified. (Note 2)
PARAMETER
Input Current
CONDITIONS
(Note 5)
MIN
0.820
500
TYP
10
MAX
50
UNITS
nA
V
FB
Overvoltage Protect Threshold
Overvoltage Protect Hysteresis
Oscillator Frequency
Measured at V
0.860
20
0.895
V
FB
mV
V
FB
V
FB
= 0.8V
= 0V
550
120
650
kHz
kHz
Gate Drive Rise Time
Gate Drive Fall Time
C
= 3000pF
= 3000pF
40
40
ns
ns
LOAD
LOAD
C
Peak Current Sense Voltage
(Note 6)
105
mV
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 3: T is calculated from the ambient temperature T and power
J A
dissipation P according to the following formula:
D
T = T + (P • θ °C/W)
J
A
D
JA
Note 4: Dynamic supply current is higher due to the gate charge being
Note 2: The LTC1772BE is guaranteed to meet specifications from 0°C to
85°C. Specifications over the –40°C to 85°C operating temperature range
are assured by design, characterization and correlation with statistical
process controls. The LTC1772BI is guaranteed to meet specifications over
the full –40°C to 85°C operating temperature range.
delivered at the switching frequency.
Note 5: The LTC1772B is tested in a feedback loop that servos V to the
output of the error amplifier.
Note 6: Peak current sense voltage is reduced dependent on duty cycle to
FB
a percentage of value as given in Figure 2.
1772bfa
3
LTC1772B
TYPICAL PERFORMANCE CHARACTERISTICS
Undervoltage Lockout Trip
Voltage vs Temperature
Reference Voltage
vs Temperature
Normalized Oscillator Frequency
vs Temperature
825
820
815
810
805
800
795
790
785
780
775
10
8
2.24
2.20
2.16
2.12
2.08
2.04
2.00
1.96
1.92
1.88
1.84
V
= 4.2V
V
= 4.2V
V
FALLING
IN
IN
IN
6
4
2
0
–2
–4
–6
–8
–10
–55 –35 –15
5
25 45 65 85 105 125
–55 –35 –15
5
25 45 65 85 105 125
–55 –35 –15
5
25 45 65 85 105 125
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
1772 G01
1772 G02
1772 G03
Maximum (VIN – SENSE–) Voltage
vs Duty Cycle
Shutdown Threshold
vs Temperature
600
560
520
480
440
400
360
320
280
240
200
120
110
100
90
V
= 4.2V
V
A
= 4.2V
IN
IN
T
= 25°C
80
70
60
50
40
60 70
20 30 40 50
DUTY CYCLE (%)
80 90 100
–55 –35 –15
5
25 45 65 85 105 125
TEMPERATURE (°C)
1772 G04
1772 G05
1772bfa
4
LTC1772B
PIN FUNCTIONS
–
I /RUN (Pin 1): This pin performs two functions. It
SENSE (Pin 4): The Negative Input to the Current Com-
TH
serves as the error amplifier compensation point as well
as the run control input. Nominal voltage range for this
pin is 0.85V to 1.9V. Forcing this pin below 0.35V causes
the device to be shut down. In shutdown all functions are
disabled and the PGATE pin is held high.
parator.
V
(Pin 5): Supply Pin. Must be closely decoupled to
GND Pin 2.
IN
PGATE (Pin 6): Gate Drive for the External P-channel
MOSFET. This pin swings from 0V to V .
IN
GND (Pin 2): Ground Pin.
V (Pin3):Receivesthefeedbackvoltagefromanexternal
FB
resistive divider across the output.
FUNCTIONAL DIAGRAM
–
V
SENSE
4
IN
5
+
–
15mV
OSC
ICMP
V
IN
RS1
PGATE
6
SWITCHING
LOGIC AND
BLANKING
CIRCUIT
SLOPE
COMP
R
Q
S
–
+
FREQ
OVP
FOLDBACK
+
–
0.3V
SHORT-CIRCUIT
DETECT
V
+
REF
60mV
EAMP
V
REF
+
–
0.8V
0.5μA
V
FB
I
TH
/RUN
1
3
+
–
V
IN
V
IN
0.35V
+
–
SHDN
UV
SHDN
CMP
VOLTAGE
REFERENCE
V
REF
0.8V
GND
2
UNDERVOLTAGE
LOCKOUT
1.2V
1772FD
1772bfa
5
LTC1772B
OPERATION (Refer to Functional Diagram)
Main Control Loop
Low Load Current Operation
The LTC1772B is a constant frequency current mode
switching regulator. During normal operation, the external
P-channel power MOSFET is turned on each cycle when
the oscillator sets the RS latch (RS1) and turned off when
the current comparator (ICMP) resets the latch. The peak
inductor current at which ICMP resets the RS latch is
Under very light load current conditions, the I /RUN pin
TH
voltage will be very close to the zero current level of 0.85V.
As the load current decreases further, an internal offset at
the current comparator input will assure that the current
comparatorremainstripped(evenatzeroloadcurrent)and
the regulator will start to skip cycles, as it must, in order
to maintain regulation. This behavior allows the regulator
to maintain constant frequency down to very light loads,
resulting in less low frequency noise generation over a
wide load current range.
controlled by the voltage on the I /RUN pin, which is the
TH
output of the error amplifier EAMP. An external resistive
divider connected between V
and ground allows the
OUT
EAMP to receive an output feedback voltage V . When
FB
the load current increases, it causes a slight decrease in
Figure 2 illustrates this result for the circuit of Figure 1
using both an LTC1772 in Burst Mode operation and an
LTC1772B (non-Burst Mode operation). At an output cur-
rent of 100mA, the Burst Mode operation part exhibits
V relative to the 0.8V reference, which in turn causes the
FB
I /RUN voltage to increase until the average inductor
TH
current matches the new load current.
ThemaincontrolloopisshutdownbypullingtheITH/RUN
pinlow.ReleasingITH/RUNallowsaninternal0.5μAcurrent
source to charge up the external compensation network.
When the ITH/RUN pin reaches 0.35V, the main control
loop is enabled with the ITH/RUN voltage then pulled up
to its zero current level of approximately 0.85V. As the
external compensation network continues to charge up,
the corresponding output current trip level follows, allow-
ing normal operation.
an output ripple of approximately 60mV , whereas the
P-P
non-BurstModeoperationparthasanoutputrippleofonly
20mV . At lower output current levels, the improvement
P-P
is even greater. This comes at a tradeoff of lower efficiency
for the non-Burst Mode operation part (see Figure 1). Also
notice the constant frequency operation of the LTC1772B,
even at 5% of maximum output current.
Dropout Operation
When the input supply voltage decreases towards the
output voltage, the rate of change of inductor current
during the ON cycle decreases. This reduction means
Comparator OVP guards against transient overshoots
>7.5% by turning off the external P-channel power
MOSFET and keeping it off until the fault is removed.
VOUT Ripple for Figure 1 Circuit Using
LTC1772 Burst Mode Operation.
VOUT Ripple for Figure 1 Circuit Using
LTC1772B Non-Burst Mode Operation.
20mV /DIV
AC
20mV /DIV
AC
1772 F02b
1772 F02a
5μs/DIV
5μs/DIV
V
V
I
= 3.6V
V
V
I
= 3.6V
IN
OUT
IN
OUT
= 2.5V
= 2.5V
= 100mA
= 100mA
OUT
OUT
Figure 2. Output Ripple Waveforms for the Circuit of Figure 1.
1772bfa
6
LTC1772B
OPERATION (Refer to Functional Diagram)
quency will gradually increase to its designed rate when
the feedback voltage again approaches 0.8V.
that the external P-channel MOSFET will remain on for
more than one oscillator cycle since the inductor current
has not ramped up to the threshold set by EAMP. Further
reduction in input supply voltage will eventually cause the
P-channel MOSFET to be turned on 100%, i.e., DC. The
output voltage will then be determined by the input volt-
age minus the voltage drop across the MOSFET, the sense
resistor and the inductor.
Overvoltage Protection
As a further protection, the overvoltage comparator in
the LTC1772B will turn the external MOSFET off when
the feedback voltage has risen 7.5% above the reference
voltage of 0.8V. This comparator has a typical hysteresis
of 20mV.
Undervoltage Lockout
Slope Compensation and Inductor’s Peak Current
To prevent operation of the P-channel MOSFET below safe
input voltage levels, an undervoltage lockout is incorpo-
rated into the LTC1772B. When the input supply voltage
drops below approximately 2.0V, the P-channel MOSFET
andallcircuitryisturnedoffexcepttheundervoltageblock,
which draws only several microamperes.
The inductor’s peak current is determined by:
V
ITH – 0.85
IPK
=
10 RSENSE
(
)
when the LTC1772B is operating below 40% duty
cycle. However, once the duty cycle exceeds 40%, slope
compensation begins and effectively reduces the peak
inductor current. The amount of reduction is given by the
curves in Figure 3.
Short-Circuit Protection
When the output is shorted to ground, the frequency of
the oscillator will be reduced to about 120kHz. This lower
frequency allows the inductor current to safely discharge,
thereby preventing current runaway. The oscillator’s fre-
110
100
90
80
70
60
50
I
= 0.4I
PK
RIPPLE
AT 5% DUTY CYCLE
= 0.2I
40
30
20
10
I
RIPPLE
PK
AT 5% DUTY CYCLE
V
= 4.2V
IN
0
10 20 30 40 50 60 70 80 90 100
DUTY CYCLE (%)
1772 F03
Figure 3. Maximum Output Current vs Duty Cycle
1772bfa
7
LTC1772B
APPLICATIONS INFORMATION
ThebasicLTC1772Bapplicationcircuitisshownin Figure 1.
Externalcomponentselectionisdrivenbytheloadrequire-
The inductance value also has a direct effect on ripple
current. The ripple current, I , decreases with higher
RIPPLE
mentandbeginswiththeselectionofL1andR
(=R1).
inductance or frequency and increases with higher V or
SENSE
IN
Next, the power MOSFET, M1 and the output diode D1 is
V
. The inductor’s peak-to-peak ripple current is given
OUT
by:
selected followed by C (= C1)and C (= C2).
IN
OUT
ꢁ
ꢄ
VIN ꢀ V
V
OUT + V
D ꢆ
VIN + VD
OUT ꢃ
IRIPPLE
=
R
SENSE
Selection for Output Current
f L
( )
ꢂ
ꢅ
R
is chosen based on the required output current.
SENSE
wherefistheoperatingfrequency.Acceptinglargervalues
of I allows the use of low inductances, but results
With the current comparator monitoring the voltage de-
veloped across R
, the threshold of the comparator
SENSE
RIPPLE
in higher output voltage ripple and greater core losses.
A reasonable starting point for setting ripple current is
determinestheinductor’speakcurrent.Theoutputcurrent
the LTC1772B can provide is given by:
I
=0.4(I
).Remember,themaximumI
OUT(MAX) RIPPLE
RIPPLE
0.105 IRIPPLE
IOUT
=
−
occurs at the maximum input voltage.
RSENSE
2
The ripple current is normally set such that the induc-
tor current is continuous down to approximately 1/4 of
maximum load current. This results in:
where I
is the inductor peak-to-peak ripple current
RIPPLE
(see Inductor Value Calculation section).
A reasonable starting point for setting ripple current is
0.03
RSENSE
IRIPPLE
≤
I
= (0.4)(I ). Rearranging the above equation, it
RIPPLE
OUT
becomes:
This implies a minimum inductance of:
0.0875
IOUT
RSENSE
=
for Duty Cycle < 40%
ꢁ
ꢄ
VIN ꢀ V
VOUT + V
OUT ꢃ
D ꢆ
VIN + VD
LMIN
=
ꢁ
ꢄ
0.03
ꢂ
ꢅ
f
ꢃ
ꢂ
ꢆ
ꢅ
However, foroperationthatisabove40%dutycycle, slope
compensation effect has to be taken into consideration to
selecttheappropriatevaluetoprovidetherequiredamount
R
SENSE
(Use V
= V )
IN
IN(MAX)
of current. Using Figure 3, the value of R
is:
SENSE
A smaller value than L
could be used in the circuit;
MIN
(0.0875)
SF
however,theinductorcurrenttransitioningfromcontinuous
to discontinuous will occur at a higher load current.
RSENSE
=
IOUT 100
(
)
Power MOSFET Selection
Inductor Value Calculation
An external P-channel power MOSFET must be selected
for use with the LTC1772B. The main selection criteria for
the power MOSFET are the threshold voltage V
The operating frequency and inductor selection are inter-
related in that higher operating frequencies permit the use
ofasmallerinductorforthesameamountofinductorripple
current. However, this is at the expense of efficiency due
to an increase in MOSFET gate charge losses.
and
GS(TH)
the “on” resistance R
RSS
, reverse transfer capacitance
DS(ON)
and total gate charge.
C
1772bfa
8
LTC1772B
APPLICATIONS INFORMATION
SincetheLTC1772Bisdesignedforoperationdowntolow
Under normal load conditions, the average current con-
ducted by the diode is:
input voltages, a logic level threshold MOSFET (R
DS(ON)
guaranteed at V = 2.5V) is required for applications that
GS
ꢁ
ꢃ
ꢂ
ꢄ
ꢆ
ꢅ
VIN ꢀ VOUT
VIN + VD
workclosetothisvoltage. WhentheseMOSFETsareused,
I =
I
OUT
D
make sure that the input supply to the LTC1772B is less
than the absolute maximum V rating, typically 8V.
GS
Theallowableforwardvoltagedropinthediodeiscalculated
from the maximum short-circuit current as:
TherequiredminimumR
oftheMOSFETisgoverned
DS(ON)
by its allowable power dissipation. For applications that
may operate the LTC1772B in dropout, i.e., 100% duty
PD
VF ≈
ISC(MAX)
cycle, at its worst case the required R
is given by:
DS(ON)
PP
where P is the allowable power dissipation and will be
determined by efficiency and/or thermal requirements.
D
RDS(ON)
=
2
DC=100%
I
(
1+ꢀp
(
)
)
OUT(MAX)
A fast switching diode must also be used to optimize
efficiency. Schottky diodes are a good choice for low
forward drop and fast switching times. Remember to
keep lead length short and observe proper grounding (see
Board Layout Checklist) to avoid ringing and increased
dissipation.
where P is the allowable power dissipation and δp is the
P
temperature dependency of R
. (1 + δp) is generally
DS(ON)
given for a MOSFET in the form of a normalized R
DS(ON)
vs temperature curve, but δp = 0.005/°C can be used as
an approximation for low voltage MOSFETs.
In applications where the maximum duty cycle is less
than 100% and the LTC1772B is in continuous mode, the
C and C
IN
Selection
OUT
R
is governed by:
DS(ON)
In continuous mode, the source current of the P-chan-
nel MOSFET is a square wave of duty cycle (V + V )/
OUT
D
PP
R
DS(ON) ꢀ
(V + V ). To prevent large voltage transients, a low
2
IN
D
1+ꢁp
DC I
(
)
(
)
OUT
ESR input capacitor sized for the maximum RMS current
must be used. The maximum RMS capacitor current is
given by:
where DC is the maximum operating duty cycle of the
LTC1772B.
1/2
]
VOUT V − VOUT
(
IN
)
[
Output Diode Selection
CIN Required IRMS ≈ IMAX
V
IN
The catch diode carries load current during the off-time.
The average diode current is therefore dependent on the
P-channel switch duty cycle. At high input voltages the
This formula has a maximum value at V = 2V , where
IN
OUT
I
= I /2. This simple worst-case condition is com-
RMS
OUT
monlyusedfordesignbecauseevensignificantdeviations
donotoffermuchrelief.Notethatcapacitormanufacturer’s
ripplecurrentratingsareoftenbasedon2000hoursoflife.
This makes it advisable to further derate the capacitor, or
to choose a capacitor rated at a higher temperature than
required. Several capacitors may be paralleled to meet the
diode conducts most of the time. As V approaches V
IN
OUT
the diode conducts only a small fraction of the time. The
most stressful condition for the diode is when the output
is short-circuited. Under this condition the diode must
safelyhandleI
atcloseto100%dutycycle. Therefore,
PEAK
it is important to adequately specify the diode peak cur-
rent and average power dissipation so as not to exceed
the diode ratings.
1772bfa
9
LTC1772B
APPLICATIONS INFORMATION
size or height requirements in the design. Due to the high
operating frequency of the LTC1772B, ceramic capacitors
Efficiency Considerations
Theefficiencyofaswitchingregulatorisequaltotheoutput
power divided by the input power times 100%. It is often
useful to analyze individual losses to determine what is
limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
can also be used for C . Always consult the manufacturer
IN
if there is any question.
The selection of C
is driven by the required effective
OUT
series resistance (ESR). Typically, once the ESR require-
ment is satisfied, the capacitance is adequate for filtering.
Efficiency = 100% – (η1 + η2 + η3 + ...)
The output ripple (ΔV ) is approximated by:
OUT
where η1, η2, etc. are the individual losses as a percent-
age of input power.
ꢂ
RIPPLE ꢄ
ꢃ
ꢅ
ꢇ
ꢆ
1
ꢀVOUT ꢁI
ESR+
4fCOUT
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of
the losses in LTC1772B circuits: 1) LTC1772B DC bias
where f is the operating frequency, C
is the output
OUT
capacitance and I
is the ripple current in the induc-
RIPPLE
tor. The output ripple is highest at maximum input voltage
since ΔI increases with input voltage.
105
L
V
REF
Manufacturers such as Nichicon, United Chemicon and
Sanyoshouldbeconsideredforhighperformancethrough-
hole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest ESR (size)
productofanyaluminumelectrolyticatasomewhathigher
100
95
90
85
80
75
V
ITH
price. Once the ESR requirement for C
has been met,
OUT
the RMS current rating generally far exceeds the I
requirement.
RIPPLE(P-P)
2.0
2.2
2.4
2.6
2.8
3.0
Low Supply Operation
INPUT VOLTAGE (V)
1772 F03
Although the LTC1772B can function down to approxi-
mately 2.0V, the maximum allowable output current is
Figure 4. Line Regulation of VREF and VITH
reduced when V decreases below 3V. Figure 4 shows
IN
the amount of change as the supply is reduced down to
2V. Also shown in Figure 4 is the effect of V on V as
V goes below 2.3V.
IN
V
IN
REF
OUT
R2
LTC1772B
3
V
FB
Setting Output Voltage
R1
The regulated output voltage is determined by:
1772 F04
ꢀ
ꢃ
R2
R1
Figure 5. Setting Output Voltage
VOUT =0.8 1+
ꢂ
ꢅ
ꢁ
ꢄ
For most applications, an 80k resistor is suggested for
R1. To prevent stray pickup, locate resistors R1 and R2
close to LTC1772B.
1772bfa
10
LTC1772B
APPLICATIONS INFORMATION
current, 2) MOSFET gate charge current, 3) I R losses
and 4) voltage drop of the output diode.
2
load current. For example, assuming a duty cycle of
50% with a Schottky diode forward voltage drop of
0.4V, the loss increases from 0.5% to 8% as the load
current increases from 0.5A to 2A.
1. The V current is the DC supply current, given in the
IN
electricalcharacteristics, thatexcludesMOSFETdriver
5. Transition losses apply to the external MOSFET and
increase at higher operating frequencies and input
voltages. Transition losses can be estimated from:
and control currents. V current results in a small loss
IN
which increases with V .
IN
2. MOSFET gate charge current results from switching
thegatecapacitanceofthepowerMOSFET. Eachtimea
MOSFETgateisswitchedfromlowtohightolowagain,
2
Transition Loss = 2(V ) I
C
(f)
IN O(MAX) RSS
OtherlossesincludingC andC ESRdissipativelosses,
IN
OUT
a packet of charge dQ moves from V to ground. The
IN
and inductor core losses, generally account for less than
2% total additional loss.
resulting dQ/dt is a current out of V which is typically
IN
much larger than the DC supply current. In continuous
mode, I
2
= f(Qp).
GATECHG
Foldback Current Limiting
3. I R losses are predicted from the DC resistances of
the MOSFET, inductor and current shunt. In continu-
ous mode the average output current flows through L
but is “chopped” between the P-channel MOSFET (in
AsdescribedintheOutputDiodeSelection,theworst-case
dissipation occurs with a short-circuited output when the
diodeconductsthecurrentlimitvaluealmostcontinuously.
Topreventexcessiveheatinginthediode,foldbackcurrent
limiting can be added to reduce the current in proportion
to the severity of the fault.
series with R
and the output diode. The MOSFET
SENSE)
R
plus R
multiplied by duty cycle can be
DS(ON)
SENSE
summedwiththeresistancesofLandR
toobtain
SENSE
Foldbackcurrentlimitingisimplementedbyaddingdiodes
2
I R losses.
D
and D between the output and the I /RUN pin as
FB1
FB2
TH
4. The output diode is a major source of power loss at
high currents and gets worse at high input voltages.
The diode loss is calculated by multiplying the forward
voltage times the diode duty cycle multiplied by the
shown in Figure 6. In a hard short (V
will be reduced to approximately 50% of the maximum
output current.
= 0V), the current
OUT
V
OUT
LTC1772B
R2
R1
+
I
/RUN V
FB
TH
D
D
FB1
FB2
1772 F05
Figure 6. Foldback Current Limiting
1772bfa
11
LTC1772B
APPLICATIONS INFORMATION
PC Board Layout Checklist
4. Connect the end of R
as close to V (Pin 5) as
SENSE
IN
+
possible. The V pin is the SENSE of the current
IN
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the LTC1772B. These items are illustrated graphically in
the layout diagram in Figure 7. Check the following in
your layout:
comparator.
–
5. Is the trace from SENSE (Pin 4) to the Sense resistor
kept short? Does the trace connect close to R
?
SENSE
6. Keep the switching node PGATE away from sensitive
small signal nodes.
1. IstheSchottkydiodecloselyconnectedbetweenground
(Pin 2) and drain of the external MOSFET?
7. Does the V pin connect directly to the feedback
FB
2. Does the (+) plate of C connect to the sense resistor
resistors? The resistive divider R1 and R2 must be
IN
as closely as possible? This capacitor provides AC
current to the MOSFET.
connected between the (+) plate of C
ground.
and signal
OUT
3. Is the input decoupling capacitor (0.1μF) connected
closely between V (Pin 5) and ground (Pin 2)?
IN
V
V
IN
1
2
3
6
5
4
+
I
/RUN PGATE
LTC1772B
TH
C
IN
L1
R
SENSE
R
ITH
GND
V
IN
OUT
M1
+
0.1μF
D1
C
OUT
–
V
SENSE
C
FB
ITH
R1
R2
1772 F06
BOLD LINES INDICATE HIGH CURRENT PATHS
Figure 7. LTC1772B Layout Diagram (See PC Board Layout Checklist)
1772bfa
12
LTC1772B
TYPICAL APPLICATIONS
LTC1772B High Efficiency, Small Footprint 3.3V to 1.8V/0.5A Regulator
V
IN
3.3V
C1
10μF
10V
R1
0.15Ω
1
6
L1
10μH
I
/RUN PGATE
LTC1772B
M1
TH
V
1.8V
0.5A
OUT
R4
10k
+
C2
47μF
6V
2
3
5
4
GND
V
D1
IN
–
C3
220pF
V
SENSE
R2
100k
FB
C1: TAIYO YUDEN CERAMIC
LMK325BJ106K-T
C2: SANYO POSCAP 6TPA47M R1: DALE 0.25W
D1: MOTOROLA MBRM120T3
L1: COILTRONICS UP1B-100
M1: Si3443DV
R3
80.6k
1772 TA02
1772bfa
13
LTC1772B
TYPICAL APPLICATIONS
LTC1772B 5V/0.5A Flyback Regulator
V
IN
2.5V
TO 9.8V
R1
0.033Ω
C1
10μF
10V
1
6
I
/RUN PGATE
LTC1772B
M1
TH
R4
10k
2
3
5
4
GND
V
IN
–
C3
220pF
V
SENSE
FB
D1
V
OUT
T1
5V
C2
•
0.5A
+
100μF
10V
×2
10μH
10μH
R2
52.3k
•
R3
10k
C1: TAIYO YUDEN CERAMIC
LMK325BJ106K-T
M1: Si9803
R1: DALE 0.25W
C2: AVXTPSE107M010R0100 T1: COILTRONICS CTX10-4
D1: IR10BQ015
1772 TA04
1772bfa
14
LTC1772B
PACKAGE DESCRIPTION
S6 Package
6-Lead Plastic TSOT-23
(Reference LTC DWG # 05-08-1636)
2.90 BSC
(NOTE 4)
0.62
MAX
0.95
REF
1.22 REF
1.4 MIN
1.50 – 1.75
2.80 BSC
3.85 MAX 2.62 REF
(NOTE 4)
PIN ONE ID
RECOMMENDED SOLDER PAD LAYOUT
PER IPC CALCULATOR
0.30 – 0.45
6 PLCS (NOTE 3)
0.95 BSC
0.80 – 0.90
0.20 BSC
DATUM ‘A’
0.01 – 0.10
1.00 MAX
0.30 – 0.50 REF
1.90 BSC
0.09 – 0.20
(NOTE 3)
S6 TSOT-23 0302 REV B
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DRAWING NOT TO SCALE
3. DIMENSIONS ARE INCLUSIVE OF PLATING
4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
5. MOLD FLASH SHALL NOT EXCEED 0.254mm
6. JEDEC PACKAGE REFERENCE IS MO-193
1772bfa
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-
tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.
15
LTC1772B
TYPICAL APPLICATION
LTC1772B 3.3V to 5V/1A Boost Regulator
R1
0.033Ω
V
IN
3.3V
C1
L1
47μF
16V
×2
4.7μH
D1
V
5V
1A
OUT
U1
C2
5
3
+
1
6
2
4
100μF
10V
×2
I
/RUN PGATE
LTC1772B
M1
TH
R4
10k
2
3
5
4
GND
V
IN
–
C3
220pF
V
SENSE
R2
FB
422k
R3
C1: AVXTPSE476M016R0047 L1: MURATA LQN6C-4R7 U1: FAIRCHILD NC7SZ04
80.6k
C2: AVXTPSE107M010R0100 M1: Si9804
ALSO SEE LTC1872
D1: IR10BQ015
R1: DALE 0.25W
FOR THIS APPLICATION
1772 TA03
RELATED PARTS
PART NUMBER
LT1375/LT1376
LTC1622
DESCRIPTION
COMMENTS
1.5A, 500kHz Step-Down Switching Regulators
High Frequency, Small Inductor, High Efficiency
V 2V to 10V, I Up to 4.5A, Synchronizable to
IN
750kHz Optional Burst Mode Operation, 8-Lead MSOP
Low Input Voltage Current Mode Step-Down DC/DC Controller
OUT
LTC1624
LTC1625
LTC1627
LTC1649
High Efficiency SO-8 N-Channel Switching Regulator Controller
N-Channel Drive, 3.5V ≤ V ≤ 36V
IN
TM
No R
Synchronous Step-Down Regulator
97% Efficiency, No Sense Resistor
SENSE
Low Voltage, Monolithic Synchronous Step-Down Regulator
3.3V Input Synchronous Step-Down Controller
Low Supply Voltage Range: 2.65V to 8V, I
= 0.5A
OUT
No Need for 5V Supply, Uses Standard Logic Gate
MOSFETs; I up to 15A
OUT
LTC1702
LTC1735
550kHz, 2 Phase, Dual Synchronous Controller
Two Channels; Minimum C and C , I
up to 15A
IN
OUT OUT
Single, High Efficiency, Low Noise Synchronous Switching
Controller
High Efficiency 5V to 3.3V Conversion at up to 15A
10μA Supply Current, 93% Efficiency, 1.23V ≤ V ≤ 18V;
OUT
LTC1771
LTC1772
Ultra-Low Supply Current Step-Down DC/DC Controller
2.8V ≤ V ≤ 20V
IN
Constant Frequency Current Mode Step-Down
DC/DC Controller in SOT-23
With Burst Mode Operation for Higher Efficiency at Light Load
Current
LTC1773
LTC1872
95% Efficient Synchronous Step-Down Controller
SOT-23 Step-Up Controller
2.65V ≤ V ≤ 8.5V; 0.8V ≤ V
≤ V ; Current Mode; 550kHz
IN
OUT IN
2.5V ≤ V ≤ 9.8V; 550kHz; 90% Efficiency
IN
No R
is a trademark of Linear Technology Corporation.
SENSE
1772bfa
LT 0508 REV A • PRINTED IN USA
LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
16
●
●
© LINEAR TECHNOLOGY CORPORATION 1999
(408) 432-1900 FAX: (408) 434-0507 www.linear.com
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