LTC1871EMS-1#PBF_技术文档

LTC1871-1

Wide Input Range, No R

™

SENSE

Current Mode Boost,

Flyback and SEPIC Controller

DESCRIPTION

TheLTC^®1871-1isawideinputrange,currentmode,boost,

ﬂybackorSEPICcontrollerthatdrivesanN-channelpower

MOSFET and requires very few external components. It

eliminatestheneedforacurrentsenseresistorbyutilizing

the power MOSFET’s on-resistance, thereby maximizing

efﬁciency.Higheroutputvoltageapplicationsarepossible

with the LTC1871-1 by connecting the SENSE pin to a

resistor in the source of the power MOSFET.

FEATURES

n

High Efﬁciency (No Sense Resistor Required)

n

Wide Input Voltage Range: 2.5V to 36V

n

Current Mode Control Provides Excellent

Transient Response

n

High Maximum Duty Cycle (92% Typ)

n

2% RUN Pin Threshold with 100mV Hysteresis

n

1% Internal Voltage Reference

n

Ultra Low Pulse Skip Threshold for Wide Input

Range Applications

The IC’s operating frequency can be set with an external

resistorovera50kHzto1MHzrange, andcanbesynchro-

nized to an external clock using the MODE/SYNC pin.

n

Micropower Shutdown: I = 10μA

Q

n

Programmable Operating Frequency

(50kHz to 1MHz) with One External Resistor

Synchronizable to an External Clock Up to 1.3 × f

n

The LTC1871-1 differs from the LTC1871 by having a

lower pulse skip threshold, making it ideal for applica-

tions requiring constant frequency operation at light

loads. The lower pulse skip threshold also helps maintain

constant frequency operation in applications with a wide

input voltage range. For applications requiring primary-

to-secondary side isolation, please refer to the LTC1871

datasheet.

OSC

User-Controlled Pulse Skip or Burst Mode^®Operation

Internal 5.2V Low Dropout Voltage Regulator

Output Overvoltage Protection

Capable of Operating with a Sense Resistor for High

Output Voltage Applications

Small 10-Lead MSOP Package

n

The LTC1871-1 is available in the 10-lead MSOP package.

L, LT, LTC, LTM and Burst Mode are registered trademarks of Linear Technology Corporation.

APPLICATIONS

n

No R

is a trademark of Linear Technology Corporation. All other trademarks are the

Telecom Power Supplies

SENSE

property of their respective owners.

n

Portable Electronic Equipment

TYPICAL APPLICATION

V

IN

3.3V

Efﬁciency of Figure 1

L1

100

90

80

70

60

50

40

30

1μH

D1

RUN

SENSE

Burst Mode

OPERATION

V

OUT

5V

I

V

IN

TH

7A

C

R

OUT1

C

+

(10A PEAK)

LTC1871-1

INTV

150μF

6.3V

×4

22k

PULSE-SKIP

MODE

FB

CC

C

6.8nF

C1

R1

12.1k

1%

C

OUT2

FREQ

GATE

GND

M1

22μF

6.3V

X5R

×2

C

IN

C

4.7μF

X5R

VCC

+

R2

37.4k

1%

R

T

80.6k

1%

MODE/SYNC

22μF

6.3V

×2

C

C2

47pF

GND

18711 F01a

C

:

TAIYO YUDEN JMK325BJ226MM

: PANASONIC EEFUEOJ151R

: TAIYO YUDEN JMK325BJ226MM

D1: MBRB2515L

IN

0.001

0.01

0.1

1

10

L1: SUMIDA CEP125-H 1R0MH

M1: FAIRCHILD FDS7760A

OUT1

OUT2

OUTPUT CURRENT (A)

18711 F01b

Figure 1. High Efﬁciency 3.3V Input, 5V Output Boost Converter (Bootstrapped)

18711fb

1

LTC1871-1

ABSOLUTE MAXIMUM RATINGS

PIN CONFIGURATION

(Note 1)

TOP VIEW

V Voltage ............................................... –0.3V to 36V

IN

RUN

TH

FB

FREQ

MODE/

SYNC

1

2

3

4

5

10 SENSE

INTV Voltage............................................ –0.3V to 7V

CC

I

9

8

7

6

V

IN

INTV

INTV Output Current.......................................... 50mA

CC

GATE

GND

GATE Voltage ............................ –0.3V to V

TH

+ 0.3V

INTVCC

I , FB Voltages ....................................... –0.3V to 2.7V

MS PACKAGE

10-LEAD PLASTIC MSOP

= 125°C, θ = 120°C/W

RUN, MODE/SYNC Voltages ....................... –0.3V to 7V

FREQ Voltage ............................................ –0.3V to 1.5V

SENSE Pin Voltage.................................... –0.3V to 36V

Operating Junction Temperature Range (Note 2)

T

JMAX

JA

LTC1871E-1 ......................................... –40°C to 85°C

LTC1871I-1 ........................................ –40°C to 125°C

Junction Temperature (Note 3) ............................ 125°C

Storage Temperature Range................... –65°C to 150°C

Lead Temperature (Soldering, 10 sec) .................. 300°C

ORDER INFORMATION

LEAD FREE FINISH

LTC1871EMS-1#PBF

LTC1871IMS-1#PBF

LEAD BASED FINISH

LTC1871EMS-1

TAPE AND REEL

PART MARKING

LTCTV

PACKAGE DESCRIPTION

10-Lead Plastic MSOP

PACKAGE DESCRIPTION

10-Lead Plastic MSOP

TEMPERATURE RANGE

–40°C to 85°C

LTC1871EMS-1#TRPBF

LTC1871IMS-1#TRPBF

TAPE AND REEL

LTCTV

–40°C to 125°C

PART MARKING

LTCTV

TEMPERATURE RANGE

–40°C to 85°C

LTC1871EMS-1#TR

LTC1871IMS-1#TR

LTC1871IMS-1

LTCTV

–40°C to 125°C

Consult LTC Marketing for parts speciﬁed with wider operating temperature ranges.

For more information on lead free part marking, go to: http://www.linear.com/leadfree/

For more information on tape and reel speciﬁcations, go to: http://www.linear.com/tapeandreel/

ELECTRICAL CHARACTERISTICS

The l denotes the speciﬁcations which apply over the full operating

temperature range, otherwise speciﬁcations are at T_A= 25°C. V_IN= 5V, V_RUN= 1.5V, R_FREQ= 80k, V_MODE/SYNC= 0V, unless otherwise speciﬁed.

SYMBOL

PARAMETER

CONDITIONS

MIN

TYP

MAX

UNITS

Main Control Loop

V

Minimum Input Voltage

2.5

V

IN(MIN)

I-Grade (Note 2)

(Note 4)

●

I

Input Voltage Supply Current

Continuous Mode

Q

V

= 5V, V = 1.4V, V = 0.75V

550

1000

μA

MODE/SYNC

FB

ITH

= 5V, V = 1.4V, V = 0.75V,

I-Grade (Note 2)

●

MODE/SYNC

FB

ITH

Burst Mode Operation, No Load

Shutdown Mode

V

= 0V, V = 0V (Note 5)

250

500

μA

MODE/SYNC

ITH

V

= 0V, V = 0V (Note 5),

MODE/SYNC

ITH

I-Grade (Note 2)

V

RUN

V

RUN

= 0V

10

20

μA

= 0V, I-Grade (Note 2)

μA

18711fb

2

LTC1871-1

ELECTRICAL CHARACTERISTICS ^Thel denotes the speciﬁcations which apply over the full operating

temperature range, otherwise speciﬁcations are at T_A= 25°C. V_IN= 5V, V_RUN= 1.5V, R_FREQ= 80k, V_MODE/SYNC= 0V, unless otherwise speciﬁed.

SYMBOL

PARAMETER

CONDITIONS

MIN

TYP

MAX

UNITS

+

V

Rising RUN Input Threshold Voltage

Falling RUN Input Threshold Voltage

1.348

1.248

V

RUN

–

1.223

1.198

1.273

1.298

V

●

V

RUN Pin Input Threshold Hysteresis

50

35

100

1

150

175

60

mV

nA

RUN(HYST)

I-Grade (Note 2)

I

RUN Input Current

Feedback Voltage

RUN

V

FB

V

ITH

= 0.4V (Note 5)

1.218

1.212

1.230

1.242

1.248

V

●

V

= 0.4V (Note 5), I-Grade (Note 2)

= 0.4V (Note 5)

1.205

1.255

60

V

nA

ITH

I

FB Pin Input Current

Line Regulation

18

FB

ITH

ΔV

2.5V ≤ V ≤ 30V

0.002

–0.1

0.02

0.03

%/V

%

FB

IN

2.5V ≤ V ≤ 30V, I-Grade (Note 2)

●

IN

ΔV

Load Regulation

V

= 0V, V = 0.5V to 0.9V (Note 5)

–1

FB

MODE/SYNC

ITH

V

= 0V, V = 0.5V to 0.9V (Note 5)

%

ITH

MODE/SYNC

ITH

I-Grade (Note 2)

– V in Percent

FB(NOM)

ΔV

ΔFB Pin, Overvoltage Lockout

V

2.5

6

10

%

μmho

mV

FB(OV)

g

m

Error Ampliﬁer Transconductance

I

TH

Pin Load = 5μA (Note 5)

650

195

150

V

Burst Mode Operation I Pin Voltage

Falling I Voltage (Note 5)

ITH(BURST)

TH

Maximum Current Sense Input Threshold

Duty Cycle < 20%

120

100

180

200

50

mV

SENSE(MAX)

Duty Cycle < 20%, I-Grade (Note 2)

●

mV

I

SENSE Pin Current (GATE High)

SENSE Pin Current (GATE Low)

V

SENSE

V

SENSE

= 0V

35

μA

SENSE(ON)

SENSE(OFF)

= 30V

0.1

5

μA

Oscillator

f

Oscillator Frequency

R

= 80k

250

50

300

350

1000

97

kHz

%

OSC

FREQ

= 80k, I-Grade (Note 2)

●

FREQ

Oscillator Frequency Range

Maximum Duty Cycle

I-Grade (Note 2)

50

D

87

92

MAX

87

97

%

f

Recommended Maximum Synchronized

Frequency Ratio

f

= 300kHz (Note 6)

1.25

25

1.30

SYNC/ OSC

OSC

= 300kHz (Note 6), I-Grade (Note 2)

t

MODE/SYNC Minimum Input Pulse Width

MODE/SYNC Maximum Input Pulse Width

Low Level MODE/SYNC Input Voltage

V

SYNC

V

SYNC

= 0V to 5V

ns

V

SYNC(MIN)

SYNC(MAX)

0.8/f

OSC

V

0.3

IL(MODE)

I-Grade (Note 2)

●

V

High Level MODE/SYNC Input Voltage

1.2

V

IH(MODE)

V

R

MODE/SYNC Input Pull-Down Resistance

Nominal FREQ Pin Voltage

50

kΩ

V

MODE/SYNC

V

FREQ

0.62

18711fb

3

LTC1871-1

ELECTRICAL CHARACTERISTICS

The l denotes the speciﬁcations which apply over the full operating

temperature range, otherwise speciﬁcations are at T_A= 25°C. V_IN= 5V, V_RUN= 1.5V, R_FREQ= 80k, V_MODE/SYNC= 0V, unless otherwise speciﬁed.

SYMBOL

PARAMETER

CONDITIONS

MIN

TYP

MAX

UNITS

Low Dropout Regulator

V

INTV Regulator Output Voltage

V

= 7.5V

5.0

5.2

8

5.4

25

V

INTVCC

CC

IN

= 7.5V, I-Grade (Note 2)

●

IN

ΔV

INTV Regulator Line Regulation

7.5V ≤ V ≤ 15V

mV

INTVCC

CC

IN

ΔV

IN1

INTV Regulator Line Regulation

15V ≤ V ≤ 30V

70

200

mV

INTVCC

CC

IN

ΔV

IN2

V

INTV Load Regulation

0 ≤ I

≤ 20mA, V = 7.5V

–2

–0.2

280

10

%

mV

μA

LDO(LOAD)

DROPOUT

INTVCC

CC

INTVCC

IN

INTV Regulator Dropout Voltage

INTV Load = 20mA

CC

I

Bootstrap Mode INTV Supply

RUN = 0V, SENSE = 5V

I-Grade (Note 2)

20

30

CC

●

μA

GATE Driver

t

r

t

f

GATE Driver Output Rise Time

GATE Driver Output Fall Time

C = 3300pF (Note 7)

17

8

100

ns

L

C = 3300pF (Note 7)

L

Note 1: Stresses beyond those listed under Absolute Maximum Ratings

may cause permanent damage to the device. Exposure to any Absolute

Maximum Rating condition for extended periods may affect device

reliability and lifetime.

Note 4: The dynamic input supply current is higher due to power MOSFET

gate charging (Q • f ). See Applications Information.

G

OSC

Note 5: The LTC1871-1 is tested in a feedback loop which servos V to

FB

the reference voltage with the I pin forced to the midpoint of its voltage

TH

Note 2: The LTC1871E-1 is guaranteed to meet performance speciﬁcations

from 0°C to 85°C junction temperature. Speciﬁcations over the –40°C

to 85°C operating junction temperature range are assured by design,

characterization and correlation with statistical process controls. The

LTC1871I-1 is guaranteed over the full –40°C to 125°C operating junction

temperature range.

range (0.3V ≤ V ≤ 1.2V, midpoint = 0.75V).

ITH

Note 6: In a synchronized application, the internal slope compensation

gain is increased by 25%. Synchronizing to a signiﬁcantly higher ratio will

reduce the effective amount of slope compensation, which could result in

subharmonic oscillation for duty cycles greater than 50%.

Note 7: Rise and fall times are measured at 10% and 90% levels.

Note 3: T is calculated from the ambient temperature T and power

J

A

dissipation P according to the following formula:

D

T = T + (P • 110°C/W)

J

A

D

TYPICAL PERFORMANCE CHARACTERISTICS

FB Voltage vs Temp

FB Voltage Line Regulation

FB Pin Current vs Temperature

60

50

40

30

20

10

0

1.231

1.230

1.229

1.25

1.24

1.23

1.22

1.21

–50

0

25 50 75 100 125 150

TEMPERATURE (°C)

–25

50 75

TEMPERATURE (°C)

0

5

10

15

V

20

(V)

25

30

35

–50 –25

0

25

100 125 150

IN

18711 G03

18711 G02

18711 G01

18711fb

4

LTC1871-1

TYPICAL PERFORMANCE CHARACTERISTICS

Shutdown Mode I_Qvs V_IN

Shutdown Mode I_Qvs Temperature

Burst Mode I_Qvs V_IN

20

15

10

5

600

500

400

300

200

100

0

30

20

10

V

= 5V

IN

0

30

0

10

20

(V)

40

150

40

–50 –25

0

25 50 75 100 125 150

TEMPERATURE (°C)

0

10

20

(V)

30

40

V

IN

18711 G04

18711 G05

18711 G06

Gate Drive Rise and

Fall Time vs C_L

Burst Mode I_Qvs Temperature

Dynamic I_Qvs Frequency

500

400

300

200

100

0

18

16

14

12

10

8

60

50

40

30

20

10

0

C

= 3300pF

L

I

= 550μA + Qg • f

Q(TOT)

RISE TIME

6

FALL TIME

4

2

0

–50

50

100 125

0

4000 6000 8000 10000 12000

(pF)

–25

0

25

75

2000

0

200

400

FREQUENCY (kHz)

1000 1200

600

800

TEMPERATURE (°C)

C

L

18711 G07

18711 G09

18711 G08

RUN Thresholds vs V_IN

RUN Thresholds vs Temperature

R_Tvs Frequency

1.5

1.4

1.3

1.40

1.35

1.30

1.25

1.20

1000

100

10

1.2

30

0

10

20

(V)

50 75

25

TEMPERATURE (°C)

–50 –25

0

100 125 150

100 200

400

600 700 800

1000

900

0

300

500

V

IN

FREQUENCY (kHz)

18711 G12

18711 G10

18711 G11

18711fb

5

LTC1871-1

TYPICAL PERFORMANCE CHARACTERISTICS

SENSE Pin Current

vs Temperature

Maximum Sense Threshold

Frequency vs Temperature

vs Temperature

325

320

315

310

305

300

295

290

285

280

275

35

30

25

160

155

150

145

GATE HIGH

V

= 0V

SENSE

140

–50

50

100 125

–50 –25

0

25 50 75 100 125 150

TEMPERATURE (°C)

–25

0

25

75

150

125

100

150

–50

50

–25

0

25

75

TEMPERATURE (°C)

18711 G13

18711 G14

18711 G15

INTV_CCDropout Voltage

vs Current, Temperature

INTV_CCLoad Regulation

INTV_CCLine Regulation

500

450

400

350

300

250

200

150

100

50

5.4

5.3

5.2

V

= 7.5V

IN

150°C

5.2

125°C

75°C

25°C

5.1

5.0

0°C

–50°C

5.1

0

40

0

10 20 30

50 60 70 80

25 30

10

20

0

5

10 15 20

(V)

35 40

0

5

15

V

INTV LOAD (mA)

CC

IN

CC

18711 G16

18711 G17

18711 G18

PIN FUNCTIONS

RUN (Pin 1): The RUN pin provides the user with an

accurate means for sensing the input voltage and pro-

gramming the start-up threshold for the converter. The

falling RUN pin threshold is nominally 1.248V and the

comparatorhas100mVofhysteresisfornoiseimmunity.

When the RUN pin is below this input threshold, the IC

I

(Pin 2): Error Ampliﬁer Compensation Pin. The

TH

current comparator input threshold increases with this

control voltage. Nominal voltage range for this pin is 0V

to 1.40V.

FB(Pin3):Receivesthefeedbackvoltagefromtheexternal

resistor divider across the output. Nominal voltage for

this pin in regulation is 1.230V.

is shut down and the V supply current is kept to a low

IN

value (typ 10μA). The Absolute Maximum Rating for the

voltage on this pin is 7V.

FREQ (Pin 4): A resistor from the FREQ pin to ground

programstheoperatingfrequencyofthechip.Thenominal

voltage at the FREQ pin is 0.6V.

18711fb

6

LTC1871-1

PIN FUNCTIONS

MODE/SYNC (Pin 5): This input controls the operating

mode of the converter and allows for synchronizing the

operating frequency to an external clock. If the MODE/

SYNC pin is connected to ground, Burst Mode operation

INTV (Pin 8): The Internal 5.20V Regulator Output.

CC

The gate driver and control circuits are powered from

this voltage. Decouple this pin locally to the IC ground

with a minimum of 4.7μF low ESR tantalum or ceramic

capacitor.

isenabled.IftheMODE/SYNCpinisconnectedtoINTV ,

CC

or if an external logic-level synchronization signal is ap-

plied to this input, Burst Mode operation is disabled and

the IC operates in a continuous mode.

V (Pin 9): Main Supply Pin. Must be closely decoupled

IN

to ground.

SENSE (Pin 10): The Current Sense Input for the Control

GND (Pin 6): Ground Pin.

Loop. Connect this pin to the drain of the power MOSFET

for V sensing and highest efﬁciency. Alternatively, the

GATE (Pin 7): Gate Driver Output.

DS

SENSE pin may be connected to a resistor in the source

of the power MOSFET. Internal leading edge blanking is

provided for both sensing methods.

BLOCK DIAGRAM

RUN

+

–

1

BIAS AND

START-UP

CONTROL

SLOPE

COMPENSATION

C2

1.248V

V

FREQ

4

IN

V-TO-I

OV

OSC

9

0.6V

I

OSC

MODE/SYNC

5

INTV

CC

GATE

7

PWM LATCH

50k

LOGIC

85mV

S

Q

R

–

+

1.230V

GND

+

BURST

COMPARATOR

CURRENT

COMPARATOR

SENSE

10

+

–

0.175V

+

–

FB

C1

EA

–

+

3

g

m

1.230V

I

TH

2

V-TO-I

SLOPE

R

LOOP

INTV

8

I

CC

LOOP

5.2V

1.230V

TO

LDO

UV

1.230V

–

+

GND

START-UP

CONTROL

BIAS

V

6

REF

18711 BD

2.00V

V

IN

18711fb

7

LTC1871-1

OPERATION

Main Control Loop

which causes the current comparator C1 to trip at a higher

peak inductor current value. The average inductor current

will therefore rise until it equals the load current, thereby

maintaining output regulation.

The LTC1871-1 is a constant frequency, current mode

controller for DC/DC boost, SEPIC and ﬂyback converter

applications. The LTC1871-1 is distinguished from con-

ventional current mode controllers because the current

control loop can be closed by sensing the voltage drop

across the power MOSFET switch instead of across a

discrete sense resistor, as shown in Figure 2. This sensing

technique improves efﬁciency, increases power density,

and reduces the cost of the overall solution.

The nominal operating frequency of the LTC1871-1 is

programmedusingaresistorfromtheFREQpintoground

and can be controlled over a 50kHz to 1000kHz range. In

addition, the internal oscillator can be synchronized to

an external clock applied to the MODE/SYNC pin and can

be locked to a frequency between 100% and 130% of its

nominal value. When the MODE/SYNC pin is left open, it

is pulled low by an internal 50k resistor and Burst Mode

operation is enabled. If this pin is taken above 2V or an

externalclockisapplied, BurstModeoperationisdisabled

and the IC operates in continuous mode. With no load (or

an extremely light load), the controller will skip pulses in

order to maintain regulation and prevent excessive output

ripple.

D

L

V

C

IN

OUT

V

IN

+

SENSE

V

SW

GATE

GND

2a. SENSE Pin Connection for

Maximum Efﬁciency (V < 36V)

TheRUNpincontrolswhethertheICisenabledorisinalow

current shutdown state. A micropower 1.248V reference

and comparator C2 allow the user to program the supply

voltage at which the IC turns on and off (comparator C2

has 100mV of hysteresis for noise immunity). With the

RUN pin below 1.248V, the chip is off and the input supply

current is typically only 10μA.

SW

D

L

V

IN

OUT

V

SW

V

IN

GATE

+

SENSE

GND

C

OUT

R

S

18711 F02

An overvoltage comparator OV senses when the FB pin

exceeds the reference voltage by 6.5% and provides a

reset pulse to the main RS latch. Because this RS latch is

reset-dominant, the power MOSFET is actively held off for

the duration of an output overvoltage condition.

GND

2b. SENSE Pin Connection for Precise

Control of Peak Current or for V > 36V

SW

Figure 2. Using the SENSE Pin On the LTC1871-1

The LTC1871-1 can be used either by sensing the voltage

drop across the power MOSFET or by connecting the

SENSE pin to a conventional shunt resistor in the source

of the power MOSFET, as shown in Figure 2. Sensing the

voltage across the power MOSFET maximizes converter

efﬁciency and minimizes the component count, but limits

theoutputvoltagetothemaximumratingforthispin(36V).

By connecting the SENSE pin to a resistor in the source

of the power MOSFET, the user is able to program output

voltages signiﬁcantly greater than 36V.

For circuit operation, please refer to the Block Diagram of

theICandFigure1.Innormaloperation,thepowerMOSFET

is turned on when the oscillator sets the PWM latch and

is turned off when the current comparator C1 resets the

latch. The divided-down output voltage is compared to an

internal 1.230V reference by the error ampliﬁer EA, which

outputs an error signal at the I pin. The voltage on the

TH

I

TH

pin sets the current comparator C1 input threshold.

When the load current increases, a fall in the FB voltage

relative to the reference voltage causes the I pin to rise,

TH

18711fb

8

LTC1871-1

OPERATION

Programming the Operating Mode

MOSFET R . If the I pin drops below 0.175V, the

DS(ON) TH

BurstModecomparatorB1willturnoffthepowerMOSFET

and scale back the quiescent current of the IC to 250μA

(sleep mode). In this condition, the load current will be

For applications where maximizing the efﬁciency at very

light loads (e.g., <100μA) is a high priority, the current

in the output divider could be decreased to a few micro-

amps and Burst Mode operation should be applied (i.e.,

the MODE/SYNC pin should be connected to ground).

In applications where ﬁxed frequency operation is more

critical than low current efﬁciency, or where the lowest

outputrippleisdesired,pulse-skipmodeoperationshould

be used and the MODE/SYNC pin should be connected

supplied by the output capacitor until the I voltage rises

TH

above the 50mV hysteresis of the burst comparator. At

light loads, short bursts of switching (where the average

inductor current is 20% of its maximum value) followed

by long periods of sleep will be observed, thereby greatly

improving converter efﬁciency. Oscilloscope waveforms

illustrating Burst Mode operation are shown in Figure 3.

to the INTV pin. This allows discontinuous conduction

CC

mode (DCM) operation down to near the limit deﬁned

by the chip’s minimum on-time (about 175ns). Below

this output current level, the converter will begin to skip

cycles in order to maintain output regulation. Figures 3

and 4 show the light load switching waveforms for Burst

Mode and pulse-skip mode operation for the converter

in Figure 1.

Pulse-Skip Mode Operation

With the MODE/SYNC pin tied to a DC voltage above 2V,

Burst Mode operation is disabled. The internal, 0.525V

buffered I burst clamp is removed, allowing the I

TH

pin to directly control the current comparator from no

load to full load. With no load, the I pin is driven below

0.175V, the power MOSFET is turned off and sleep mode

is invoked. Oscilloscope waveforms illustrating this mode

of operation are shown in Figure 4.

TH

Burst Mode Operation

Burst Mode operation is selected by leaving the MODE/

SYNC pin unconnected or by connecting it to ground. In

When an external clock signal drives the MODE/SYNC

pin at a rate faster than the chip’s internal oscillator, the

oscillatorwillsynchronizetoit.Inthissynchronizedmode,

Burst Mode operation is disabled. The constant frequency

associated with synchronized operation provides a more

controlled noise spectrum from the converter, at the ex-

pense of overall system efﬁciency of light loads.

normaloperation,therangeontheI pincorrespondingto

TH

no load to full load is 0.30V to 1.2V. In Burst Mode opera-

tion, if the error ampliﬁer EA drives the I voltage below

TH

0.525V, the buffered I input to the current comparator

TH

C1 will be clamped at 0.525V (which corresponds to 25%

of maximum load current). The inductor current peak is

then held at approximately 30mV divided by the power

V

I

= 3.3V

= 5V

OUT

MODE/SYNC = 0V

(Burst Mode OPERATION)

V

I

= 3.3V

= 5V

OUT

MODE/SYNC = INTV

CC

(PULSE-SKIP MODE)

IN

OUT

IN

OUT

= 500mA

V

OUT

50mV/DIV

I

L

5A/DIV

18711 F03

18711 F04

10μs/DIV

2μs/DIV

Figure 4. LTC1871-1 Low Output Current Operation with

Burst Mode Operation Disabled (MODE/SYNC = INTV_CC

Figure 3. LTC1871-1 Burst Mode Operation

(MODE/SYNC = 0V) at Low Output Current

)

18711fb

9

LTC1871-1

APPLICATIONS INFORMATION

When the oscillator’s internal logic circuitry detects a

synchronizing signal on the MODE/SYNC pin, the in-

ternal oscillator ramp is terminated early and the slope

compensation is increased by approximately 30%. As

a result, in applications requiring synchronization, it is

recommended that the nominal operating frequency of

the IC be programmed to be about 75% of the external

clock frequency. Attempting to synchronize to too high an

0.6V, and the current that ﬂows into the FREQ pin is used

to charge and discharge an internal oscillator capacitor. A

graph for selecting the value of R for a given operating

T

frequency is shown in Figure 6.

1000

external frequency (above 1.3f ) can result in inadequate

O

slopecompensationandpossiblesubharmonicoscillation

100

(or jitter).

The external clock signal must exceed 2V for at least 25ns,

and should have a maximum duty cycle of 80%, as shown

in Figure 5. The MOSFET turn on will synchronize to the

rising edge of the external clock signal.

10

100 200

400

600 700 800

1000

900

0

300

500

FREQUENCY (kHz)

18711 F06

2V TO 7V

MODE/

SYNC

Figure 6. Timing Resistor (R_T) Value

t

= 25ns

MIN

INTV Regulator Bypassing and Operation

CC

0.8T

T

T = 1/f

O

An internal, P-channel low dropout voltage regulator pro-

duces the 5.2V supply which powers the gate driver and

logic circuitry within the LTC1871-1, as shown in Figure 7.

GATE

D = 40%

The INTV regulator can supply up to 50mA and must be

CC

bypassed to ground immediately adjacent to the IC pins

with a minimum of 4.7μF tantalum or ceramic capacitor.

Good bypassing is necessary to supply the high transient

currents required by the MOSFET gate driver.

I

L

18711 F05

For input voltages that don’t exceed 7V (the absolute

maximum rating for this pin), the internal low dropout

Figure 5. MODE/SYNC Clock Input and Switching

Waveforms for Synchronized Operation

regulator in the LTC1871-1 is redundant and the INTV

CC

pin can be shorted directly to the V pin. With the INTV

IN

Programming the Operating Frequency

pin shorted to V , however, the divider that programs the

IN

CC

The choice of operating frequency and inductor value is

a tradeoff between efﬁciency and component size. Low

frequency operation improves efﬁciency by reducing

MOSFET and diode switching losses. However, lower

frequency operation requires more inductance for a given

amount of load current.

regulated INTV voltage will draw 10μA of current from

theinputsupply, eveninshutdownmode. Forapplications

that require the lowest shutdown mode input supply cur-

rent, do not connect the INTV pin to V . Regardless of

CC

IN

whethertheINTV pinisshortedtoV ornot,itisalways

CC

IN

necessary to have the driver circuitry bypassed with a

4.7μF tantalum or low ESR ceramic capacitor to ground

TheLTC1871-1usesaconstantfrequencyarchitecturethat

can be programmed over a 50kHz to 1000kHz range with

a single external resistor from the FREQ pin to ground, as

shown in Figure 1. The nominal voltage on the FREQ pin is

immediately adjacent to the INTV and GND pins.

CC

In an actual application, most of the IC supply current is

used to drive the gate capacitance of the power MOSFET.

18711fb

10

LTC1871-1

APPLICATIONS INFORMATION

INPUT

SUPPLY

2.5V TO 30V

V

IN

–

1.230V

P-CH

5.2V

+

C

IN

R2

R1

INTV

CC

+

C

VCC

4.7μF

GATE

GND

LOGIC

DRIVER

M1

GND

PLACE AS CLOSE AS

POSSIBLE TO DEVICE PINS

18711 F07

Figure 7. Bypassing the LDO Regulator and Gate Driver Supply

As a result, high input voltage applications in which a

large power MOSFET is being driven at high frequencies

can cause the LTC1871-1 to exceed its maximum junc-

tion temperature rating. The junction temperature can be

estimated using the following equations:

Thisdemonstrateshowsigniﬁcantthegatechargecurrent

can be when compared to the static quiescent current in

the IC.

To prevent the maximum junction temperature from being

exceeded, the input supply current must be checked when

I

≈ I + f • Q

operating in a continuous mode at high V . A tradeoff

Q(TOT)

Q

G

IN

between the operating frequency and the size of the power

MOSFETmayneedtobemadeinordertomaintainareliable

IC junction temperature. Prior to lowering the operating

frequency, however, be sure to check with power MOSFET

P = V • (I + f • Q )

IC

IN

Q

G

T = T + P • R

J

A

IC

TH(JA)

The total quiescent current I

consists of the static

Q(TOT)

manufacturers for their latest-and-greatest low Q , low

G

supply current (I ) and the current required to charge and

Q

R

devices. Power MOSFET manufacturing tech-

DS(ON)

dischargethegateofthepowerMOSFET.The10-pinMSOP

nologies are continually improving, with newer and better

package has a thermal resistance of R

= 120°C/W.

TH(JA)

performance devices being introduced almost yearly.

As an example, consider a power supply with V = 5V and

IN

V = 12V at I = 1A. The switching frequency is 500kHz,

Output Voltage Programming

O

andthemaximumambienttemperatureis70°C.Thepower

The output voltage is set by a resistor divider according

to the following formula:

MOSFET chosen is the IRF7805, which has a maximum

R

of 11mΩ (at room temperature) and a maximum

DS(ON)

total gate charge of 37nC (the temperature coefﬁcient of

the gate charge is low).

R2

R1

ꢀ

ꢁ

ꢃ

ꢄ

V =1.230V • 1+

ꢂ

ꢅ

O

I

= 600μA + 37nC • 500kHz = 19.1mA

Q(TOT)

The external resistor divider is connected to the output

as shown in Figure 1, allowing remote voltage sensing.

The resistors R1 and R2 are typically chosen so that the

P = 5V • 19.1mA = 95mW

IC

T = 70°C + 120°C/W • 95mW = 81.4°C

J

18711fb

11

LTC1871-1

APPLICATIONS INFORMATION

error caused by the current ﬂowing into the FB pin dur-

ing normal operation is less than 1% (this translates to a

maximum value of R1 of about 250k).

The turn-on and turn-off input voltage thresholds are

programmed using a resistor divider according to the

following formulas:

R2

R1

ꢀ

ꢁ

ꢃ

ꢄ

V

=1.248V • 1+

ꢂ

ꢅ

IN(OFF)

Programming Turn-On and Turn-Off Thresholds

with the RUN Pin

R2

R1

ꢀ

ꢁ

ꢃ

ꢄ

V

=1.348V • 1+

ꢂ

ꢅ

IN(ON)

TheLTC1871-1containsanindependent,micropowervolt-

agereferenceandcomparatordetectioncircuitthatremains

active even when the device is shut down, as shown in

Figure 8. This allows users to accurately program an input

voltage at which the converter will turn on and off. The

falling threshold voltage on the RUN pin is equal to the

internal reference voltage of 1.248V. The comparator has

100mV of hysteresis to increase noise immunity.

The resistor R1 is typically chosen to be less than 1M.

For applications where the RUN pin is only to be used as

alogicinput,theusershouldbeawareofthe7VAbsolute

Maximum Rating for this pin! The RUN pin can be con-

nectedtotheinputvoltagethroughanexternal1Mresistor,

as shown in Figure 8c, for “always on” operation.

V

IN

+

R2

R1

RUN

COMPARATOR

RUN

+

BIAS AND

START-UP

CONTROL

6V

INPUT

SUPPLY

–

OPTIONAL

FILTER

CAPACITOR

1.248V

μPOWER

REFERENCE

GND

–

18711 F8a

Figure 8a. Programming the Turn-On and Turn-Off Thresholds Using the RUN Pin

V

IN

+

R2

1M

RUN

GND

COMPARATOR

+

–

RUN

COMPARATOR

6V

INPUT

SUPPLY

RUN

+

–

6V

1.248V

EXTERNAL

LOGIC CONTROL

1.248V

–

18711 F08b

18711 F08c

Figure 8b. On/Off Control Using External Logic

Figure 8c. External Pull-Up Resistor On

RUN Pin for “Always On” Operation

18711fb

12

LTC1871-1

APPLICATIONS INFORMATION

Application Circuits

The maximum duty cycle, D

, should be calculated at

MAX

minimum V .

IN

A basic LTC1871-1 application circuit is shown in

Figure 1. External component selection is driven by the

characteristics of the load and the input supply. The

ﬁrst topology to be analyzed will be the boost converter,

followed by SEPIC (single ended primary inductance

converter).

Boost Converter: Ripple Current ΔI and the ‘χ’ Factor

L

The constant ‘χ’ in the equation above represents the

percentage peak-to-peak ripple current in the inductor,

relative to its maximum value. For example, if 30% ripple

current is chosen, then χ = 0.30, and the peak current is

15% greater than the average.

Boost Converter: Duty Cycle Considerations

For a current mode boost regulator operating in CCM,

slope compensation must be added for duty cycles above

50% in order to avoid subharmonic oscillation. For the

LTC1871-1, thisrampcompensationisinternal. Havingan

internally ﬁxed ramp compensation waveform, however,

does place some constraints on the value of the inductor

and the operating frequency. If too large an inductor is

Foraboostconverteroperatinginacontinuousconduction

mode (CCM), the duty cycle of the main switch is:

ꢀ

_INꢃ

V_O+ V_D– V

D=

ꢂ

ꢅ

V_O+ V_D

ꢁ

ꢄ

where V is the forward voltage of the boost diode. For

D

converters where the input voltage is close to the output

voltage,thedutycycleislowandforconvertersthatdevelop

a high output voltage from a low voltage input supply,

the duty cycle is high. The maximum output voltage for a

boost converter operating in CCM is:

used, theresultingcurrentramp(ΔI )willbesmallrelative

L

to the internal ramp compensation (at duty cycles above

50%), and the converter operation will approach voltage

mode(rampcompensationreducesthegainofthecurrent

loop). If too small an inductor is used, but the converter

is still operating in CCM (near critical conduction mode),

the internal ramp compensation may be inadequate to

prevent subharmonic oscillation. To ensure good current

mode gain and avoid subharmonic oscillation, it is recom-

mended that the ripple current in the inductor fall in the

range of 20% to 40% of the maximum average current.

For example, if the maximum average input current is

V

IN(MIN)

V_O(MAX)

=

– V_D

1–D

(

)

MAX

The maximum duty cycle capability of the LTC1871-1 is

typically 92%. This allows the user to obtain high output

voltages from low input supply voltages.

Boost Converter: The Peak and Average Input Currents

1A, choose a ΔI between 0.2A and 0.4A, and a value ‘χ’

L

between 0.2 and 0.4.

ThecontrolcircuitintheLTC1871-1ismeasuringtheinput

current (either by using the R

of the power MOSFET

DS(ON)

Boost Converter: Inductor Selection

or by using a sense resistor in the MOSFET source), so

the output current needs to be reﬂected back to the input

in order to dimension the power MOSFET properly. Based

on the fact that, ideally, the output power is equal to the

input power, the maximum average input current is:

Givenanoperatinginputvoltagerange,andhavingchosen

the operating frequency and ripple current in the inductor,

the inductor value can be determined using the following

equation:

I_O(MAX)

V

IN(MIN)

I

=

IN(MAX)

L =

•D_MAX

1–D_MAX

Thepeak input current is:

ꢀI_L• f

where:

I_O(MAX)

ꢀ

ꢁ

ꢂ

ꢄ

ꢅ

ꢀI_L= ꢁ •

I

= 1+

•

ꢃ

ꢆ

IN(PEAK)

1–D_MAX

2

1–D_MAX

18711fb

13

LTC1871-1

APPLICATIONS INFORMATION

Remember that boost converters are not short-circuit

protected. Under a shorted output condition, the inductor

current is limited only by the input supply capability. For

applications requiring a step-up converter that is short-

circuit protected, please refer to the applications section

covering SEPIC converters.

Boost Converter: Inductor Core Selection

Once the value for L is known, the type of inductor must

be selected. High efﬁciency converters generally cannot

affordthecorelossfoundinlowcostpowderedironcores,

forcing the use of more expensive ferrite, molypermalloy

or Kool Mμ^®cores. Actual core loss is independent of core

size for a ﬁxed inductor value, but is very dependent on

the inductance selected. As inductance increases, core

losses go down. Unfortunately, increased inductance

requires more turns of wire and therefore, copper losses

will increase. Generally, there is a tradeoff between core

losses and copper losses that needs to be balanced.

The minimum required saturation current of the inductor

can be expressed as a function of the duty cycle and the

load current, as follows:

I_O(MAX)

ꢁ

ꢂ

ꢃ

ꢅ

ꢆ

I_L(SAT)ꢀ 1+

•

ꢄ

ꢇ

2

1–D_MAX

Ferrite designs have very low core losses and are pre-

ferred at high switching frequencies, so design goals can

concentrate on copper losses and preventing saturation.

Ferrite core material saturates “hard,” meaning that the

inductancecollapsesrapidlywhenthepeakdesigncurrent

is exceeded. This results in an abrupt increase in inductor

ripple current and consequently, output voltage ripple. Do

not allow the core to saturate!

The saturation current rating for the inductor should be

checked at the minimum input voltage (which results

in the highest inductor current) and maximum output

current.

Boost Converter: Operating in Discontinuous Mode

Discontinuous mode operation occurs when the load cur-

rent is low enough to allow the inductor current to run out

during the off-time of the switch, as shown in Figure 9.

Oncetheinductorcurrentisnearzero,theswitchanddiode

capacitancesresonatewiththeinductancetoformdamped

ringing at 1MHz to 10MHz. If the off-time is long enough,

the drain voltage will settle to the input voltage.

Molypermalloy (from Magnetics, Inc.) is a very good,

low cost core material for toroids, but is more expensive

than ferrite. A reasonable compromise from the same

manufacturer is Kool Mμ.

Boost Converter: Power MOSFET Selection

Depending on the input voltage and the residual energy

in the inductor, this ringing can cause the drain of the

power MOSFET to go below ground where it is clamped

by the body diode. This ringing is not harmful to the IC

and it has not been shown to contribute signiﬁcantly to

EMI. Any attempt to damp it with a snubber will degrade

the efﬁciency.

ThepowerMOSFETservestwopurposesintheLTC1871-1:

itrepresentsthemainswitchingelementinthepowerpath,

and its R

represents the current sensing element

DS(ON)

for the control loop. Important parameters for the power

MOSFET include the drain-to-source breakdown voltage

(BV ),thethresholdvoltage(V

),theon-resistance

GS(TH)

DSS

DS(ON)

(R

)versusgate-to-sourcevoltage,thegate-to-source

V

= 3.3V I = 200mA

OUT

IN

and gate-to-drain charges (Q and Q , respectively),

the maximum drain current (I

thermal resistances (R

GS

D(MAX)

and R

GD

= 5V

) and the MOSFET’s

MOSFET DRAIN

VOLTAGE

).

TH(JA)

TH(JC)

2V/DIV

The gate drive voltage is set by the 5.2V INTV low drop

CC

regulator. Consequently, logic-level threshold MOSFETs

should be used in most LTC1871-1 applications. If low

input voltage operation is expected (e.g., supplying power

INDUCTOR

CURRENT

2A/DIV

18711 F09

2μs/DIV

Figure 9. Discontinuous Mode Waveforms

18711fb

14

LTC1871-1

APPLICATIONS INFORMATION

from a lithium-ion battery or a 3.3V logic supply), then

sublogic-level threshold MOSFETs should be used.

Another method of choosing which power MOSFET to

use is to check what the maximum output current is for a

givenR

, sinceMOSFETon-resistancesareavailable

DS(ON)

Pay close attention to the BV

speciﬁcations for the

DSS

in discrete values.

MOSFETsrelativetothemaximumactualswitchvoltagein

theapplication.Manylogic-leveldevicesarelimitedto30V

or less, and the switch node can ring during the turn-off of

the MOSFET due to layout parasitics. Check the switching

waveforms of the MOSFET directly across the drain and

source terminals using the actual PC board layout (not

just on a lab breadboard!) for excessive ringing.

1–D_MAX

I_O(MAX)= V_SENSE(MAX)

•

ꢀ

ꢁ

ꢂ

ꢄ

1+

•R_DS(ON)•ꢇ_T

ꢃ

ꢆ

ꢅ

2

It is worth noting that the 1 – D

relationship between

MAX

I

and R

can cause boost converters with a

O(MAX)

DS(ON)

wide input range to experience a dramatic range of maxi-

mum input and output current. This should be taken into

consideration in applications where it is important to limit

the maximum current drawn from the input supply.

During the switch on-time, the control circuit limits the

maximumvoltagedropacrossthepowerMOSFETtoabout

150mV (at low duty cycle). The peak inductor current

is therefore limited to 150mV/R . The relationship

DS(ON)

Calculating Power MOSFET Switching and Conduction

Losses and Junction Temperatures

between the maximum load current, duty cycle and the

of the power MOSFET is:

R

DS(ON)

In order to calculate the junction temperature of the

power MOSFET, the power dissipated by the device must

be known. This power dissipation is a function of the

duty cycle, the load current and the junction temperature

itself (due to the positive temperature coefﬁcient of its

R_DS(ON)).Asaresult,someiterativecalculationisnormally

required to determine a reasonably accurate value. Since

the controller is using the MOSFET as both a switching

and a sensing element, care should be taken to ensure

that the converter is capable of delivering the required

load current over all operating conditions (line voltage

and temperature), and for the worst-case speciﬁcations

1–D_MAX

R_DS(ON)ꢀ V_SENSE(MAX)•

ꢁ

ꢂ

ꢃ

ꢅ

ꢆ

1+

•I_O(MAX)•ꢈ_T

ꢄ

ꢇ

2

The V_SENSE(MAX)term is typically 150mV at low duty

cycle, and is reduced to about 100mV at a duty cycle of

92% due to slope compensation, as shown in Figure 10.

The ρ_Tterm accounts for the temperature coefﬁcient of

the R_DS(ON)of the MOSFET, which is typically 0.4%/°C.

Figure 11 illustrates the variation of normalized R_DS(ON)

over temperature for a typical power MOSFET.

200

150

100

50

2.0

1.5

1.0

0.5

0

0.2

0.4

0.5

0.8

1.0

50

JUNCTION TEMPERATURE (°C)

–50

100

150

0

DUTY CYCLE

18711 F10

18711 F11

Figure 10. Maximum SENSE Threshold Voltage vs Duty Cycle

Figure 11. Normalized R_DS(ON)vs Temperature

18711fb

15

LTC1871-1

APPLICATIONS INFORMATION

for V

and the R

of the MOSFET listed in

The R

to be used in this equation normally includes

SENSE(MAX)

the manufacturer’s data sheet.

DS(ON)

TH(JA)

the R

for the device plus the thermal resistance from

TH(JC)

the board to the ambient temperature in the enclosure.

The power dissipated by the MOSFET in a boost converter

is:

Remember to keep the diode lead lengths short and to

observe proper switch-node layout (see Board Layout

Checklist) to avoid excessive ringing and increased

dissipation.

I_O(MAX)

ꢀ

ꢃ²

ꢅ

P_FET

=

• R_DS(ON)•D_MAX•ꢆ_T

ꢂ

1–D

ꢁ

_MAXꢄ

I_O(MAX)

1.85

Boost Converter: Output Capacitor Selection

+k • V_O

•

•C_RSS• f

1–D

(

)

MAX

Contributions of ESR (equivalent series resistance), ESL

(equivalent series inductance) and the bulk capacitance

mustbeconsideredwhenchoosingthecorrectcomponent

for a given output ripple voltage. The effects of these three

parameters (ESR, ESL and bulk C) on the output voltage

ripple waveform are illustrated in Figure 12e for a typical

boost converter.

2

The ﬁrst term in the equation above represents the I R

losses in the device, and the second term, the switching

losses.Theconstant,k=1.7,isanempiricalfactorinversely

related to the gate drive current and has the dimension

of 1/current.

From a known power dissipated in the power MOSFET, its

junction temperature can be obtained using the following

formula:

The choice of component(s) begins with the maximum

acceptable ripple voltage (expressed as a percentage of

theoutputvoltage), andhowthisrippleshouldbedivided

T = T + P • R

J

A

FET

TH(JA)

between the ESR step and the charging/discharging ΔV.

The R

the R

to be used in this equation normally includes

for the device plus the thermal resistance from

For the purpose of simplicity we will choose 2% for the

maximumoutputripple,tobedividedequallybetweenthe

TH(JA)

TH(JC)

the case to the ambient temperature (R

). This value

ESR step and the charging/discharging ΔV. This percent-

TH(CA)

of T can then be compared to the original, assumed value

age ripple will change, depending on the requirements

of the application, and the equations provided below can

easily be modiﬁed.

J

used in the iterative calculation process.

Boost Converter: Output Diode Selection

For a 1% contribution to the total ripple voltage, the ESR

of the output capacitor can be determined using the fol-

lowing equation:

To maximize efﬁciency, a fast switching diode with low

forwarddropandlowreverseleakageisdesired.Theoutput

diode in a boost converter conducts current during the

switch off-time. The peak reverse voltage that the diode

must withstand is equal to the regulator output voltage.

The average forward current in normal operation is equal

to the output current, and the peak current is equal to the

peak inductor current.

0.01• V_O

ESR_COUTꢀ

I_IN(PEAK)

where:

I_O(MAX)

ꢀ

ꢁ

ꢂ

ꢄ

ꢅ

I_IN(PEAK)= 1+

•

ꢃ

ꢆ

I_O(MAX)

ꢀ

ꢁ

ꢂ

ꢄ

2

1–D_MAX

I

_D(PEAK)=I_L(PEAK)= 1+

•

ꢃ

ꢆ

ꢅ

2

1–D_MAX

For the bulk C component, which also contributes 1% to

the total ripple:

The power dissipated by the diode is:

P = I • V

D

O(MAX)

D

I_O(MAX)

C_OUTꢀ

and the diode junction temperature is:

0.01• V_O• f

T = T + P • R

J

A

D

TH(JA)

18711fb

16

LTC1871-1

APPLICATIONS INFORMATION

Formanydesignsitispossibletochooseasinglecapacitor

type that satisﬁes both the ESR and bulk C requirements

forthedesign.Incertaindemandingapplications,however,

the ripple voltage can be improved signiﬁcantly by con-

necting two or more types of capacitors in parallel. For

example, using a low ESR ceramic capacitor can minimize

the ESR step, while an electrolytic capacitor can be used

to supply the required bulk C.

tested for use in switching power supplies. An excellent

choice is AVX TPS series of surface mount tantalum. Also,

ceramic capacitors are now available with extremely low

ESR, ESL and high ripple current ratings.

Boost Converter: Input Capacitor Selection

The input capacitor of a boost converter is less critical

than the output capacitor, due to the fact that the inductor

is in series with the input and the input current waveform

is continuous (see Figure 12b). The input voltage source

impedance determines the size of the input capacitor,

Once the output capacitor ESR and bulk capacitance have

been determined, the overall ripple voltage waveform

should be veriﬁed on a dedicated PC board (see Board

Layout section for more information on component place-

ment). Lab breadboards generally suffer from excessive

series inductance (due to inter-component wiring), and

these parasitics can make the switching waveforms look

signiﬁcantly worse than they would be on a properly

designed PC board.

L

D

V

OUT

V

SW

C

R

L

IN

OUT

12a. Circuit Diagram

Theoutputcapacitorinaboostregulatorexperienceshigh

RMS ripple currents, as shown in Figure 12. The RMS

output capacitor ripple current is:

I

IN

I

L

V_O– V

IN(MIN)

I_RMS(COUT)ꢀI_O(MAX)•

12b. Inductor and Input Currents

V

IN(MIN)

Note that the ripple current ratings from capacitor manu-

facturers are often based on only 2000 hours of life. This

makes it advisable to further derate the capacitor or to

choose a capacitor rated at a higher temperature than

required. Several capacitors may also be placed in parallel

to meet size or height requirements in the design.

I

SW

t

ON

12c. Switch Current

I

D

t

OFF

Manufacturers such as Nichicon, United Chemicon and

Sanyoshouldbeconsideredforhighperformancethrough-

hole capacitors. The OS-CON semiconductor dielectric

capacitor available from Sanyo has the lowest product of

ESR and size of any aluminum electrolytic, at a somewhat

higher price.

I

O

12d. Diode and Output Currents

ΔV

COUT

V

OUT

(AC)

RINGING DUE TO

TOTAL INDUCTANCE

(BOARD + CAP)

In surface mount applications, multiple capacitors may

have to be placed in parallel in order to meet the ESR or

RMS current handling requirements of the application.

Aluminum electrolytic and dry tantalum capacitors are

both available in surface mount packages. In the case of

tantalum, it is critical that the capacitors have been surge

ΔV

ESR

12e. Output Voltage Ripple Waveform

Figure 12. Switching Waveforms for a Boost Converter

18711fb

17

LTC1871-1

APPLICATIONS INFORMATION

Table 1. Recommended Component Manufacturers

VENDOR

COMPONENTS

TELEPHONE

(207) 282-5111

(952) 894-9590

(847) 639-6400

(407) 241-7876

(805) 446-4800

(408) 822-2126

(516) 847-3000

(310) 322-3331

(361) 992-7900

(408) 986-0424

(800) 245-3984

(617) 926-0404

(770) 436-1300

(847) 843-7500

(602) 244-6600

(714) 373-7334

(619) 661-6835

(847) 956-0667

(408) 573-4150

(562) 596-1212

(972) 243-4321

(408) 432-8020

(847) 699-3430

(847) 696-2000

(605) 665-9301

(800) 554-5565

(207) 324-4140

(631) 543-7100

WEB ADDRESS

avxcorp.com

AVX

Capacitors

Inductors, Transformers

Inductors

BH Electronics

Coilcraft

bhelectronics.com

coilcraft.com

Coiltronics

Diodes, Inc

Fairchild

Inductors

coiltronics.com

diodes.com

Diodes

MOSFETs

fairchildsemi.com

generalsemiconductor.com

irf.com

General Semiconductor

International Rectiﬁer

IRC

Diodes

MOSFETs, Diodes

Sense Resistors

Tantalum Capacitors

Toroid Cores

Diodes

irctt.com

Kemet

kemet.com

Magnetics Inc

Microsemi

Murata-Erie

Nichicon

mag-inc.com

microsemi.com

murata.co.jp

Inductors, Capacitors

Capacitors

nichicon.com

onsemi.com

On Semiconductor

Panasonic

Sanyo

Diodes

Capacitors

panasonic.com

sanyo.co.jp

Capacitors

Sumida

Inductors

sumida.com

Taiyo Yuden

TDK

Capacitors

t-yuden.com

Capacitors, Inductors

Heat Sinks

component.tdk.com

aavidthermalloy.com

nec-tokinamerica.com

tokoam.com

Thermalloy

Tokin

Capacitors

Toko

Inductors

United Chemicon

Vishay/Dale

Vishay/Siliconix

Vishay/Sprague

Zetex

Capacitors

chemi-com.com

vishay.com

Resistors

MOSFETs

vishay.com

Capacitors

vishay.com

Small-Signal Discretes

zetex.com

which is typically in the range of 10μF to 100μF. A low ESR

capacitor is recommended, although it is not as critical as

for the output capacitor.

Burst Mode Operation and Considerations

The choice of MOSFET R and inductor value also

determines the load current at which the LTC1871-1 en-

ters Burst Mode operation. When bursting, the controller

clamps the peak inductor current to approximately:

DS(ON)

The RMS input capacitor ripple current for a boost con-

verter is:

V

IN(MIN)

30mV

R_DS(ON)

I_RMS(CIN)= 0.3•

•D_MAX

I_BURST(PEAK)

=

L • f

Please note that the input capacitor can see a very high

surge current when a battery is suddenly connected to

the input of the converter and solid tantalum capacitors

can fail catastrophically under these conditions. Be sure

to specify surge-tested capacitors!

which represents about 20% of the maximum 150mV

SENSE pin voltage. The corresponding average current

dependsupontheamountofripplecurrent.Lowerinductor

values (higher ΔI ) will reduce the load current at which

L

18711fb

18

LTC1871-1

APPLICATIONS INFORMATION

Burst Mode operations begins, since it is the peak current

that is being clamped.

1. The supply current into V . The V current is the

IN IN

sum of the DC supply current I (given in the Electrical

Q

Characteristics) and the MOSFET driver and control

The output voltage ripple can increase during Burst Mode

currents. The DC supply current into the V pin is typi-

IN

operation if ΔI is substantially less than I

. This can

BURST

L

cally about 550μA and represents a small power loss

occur if the input voltage is very low or if a very large

inductor is chosen. At high duty cycles, a skipped cycle

causes the inductor current to quickly decay to zero.

(much less than 1%) that increases with V . The driver

IN

current results from switching the gate capacitance

of the power MOSFET; this current is typically much

larger than the DC current. Each time the MOSFET is

However, because ΔI is small, it takes multiple cycles

L

for the current to ramp back up to I

. Dur-

BURST(PEAK)

switched on and then off, a packet of gate charge Q

G

ing this inductor charging interval, the output capacitor

must supply the load current and a signiﬁcant droop in

the output voltage can occur. Generally, it is a good idea

is transferred from INTV to ground. The resulting

CC

dQ/dt is a current that must be supplied to the INTV

CC

capacitor through the V pin by an external supply. If

IN

to choose a value of inductor ΔI between 25% and 40%

L

the IC is operating in CCM:

of I

. The alternative is to either increase the value

of the output capacitor or disable Burst Mode operation

IN(MAX)

I

≈ I = f • Q

Q(TOT) Q G

using the MODE/SYNC pin.

P = V • (I + f • Q )

IC

IN

Q

G

Burst Mode operation can be defeated by connecting the

MODE/SYNC pin to a high logic-level voltage (either with

2. Power MOSFET switching and conduction losses. The

technique of using the voltage drop across the power

MOSFETtoclosethecurrentfeedbackloopwaschosen

because of the increased efﬁciency that results from

not having a sense resistor. The losses in the power

MOSFET are equal to:

a control input or by connecting this pin to INTV ). In

CC

this mode, the burst clamp is removed, and the chip can

operateatconstantfrequencyfromcontinuousconduction

mode (CCM) at full load, down into deep discontinuous

conduction mode (DCM) at light load. Prior to skipping

pulses at very light load (i.e., <5% of full load), the

controller will operate with a minimum switch on-time

in DCM. Pulse skipping prevents a loss of control of

the output at very light loads and reduces output volt-

age ripple.

ꢀ

ꢃ²

ꢅ

I_O(MAX)

P_FET

=

• R_DS(ON)•D_MAX•ꢆ_T

ꢂ

1–D

ꢁ

_MAXꢄ

I_O(MAX)

1.85

+k • V_O

•

•C_RSS• f

1–D

(

)

MAX

2

The I R power savings that result from not having a

discrete sense resistor can be calculated almost by

inspection.

Efﬁciency Considerations: How Much Does V

Sensing Help?

DS

The efﬁciency of a switching regulator is equal to the

output power divided by the input power (×100%).

Percent efﬁciency can be expressed as:

ꢀ

ꢂ

ꢃ²

ꢅ

I_O(MAX)

P_R(SENSE)

=

•R_SENSE•D_MAX

1–D

ꢁ

_MAXꢄ

% Efﬁciency = 100% – (L1 + L2 + L3 + …),

To understand the magnitude of the improvement with

this V sensing technique, consider the 3.3V input,

DS

where L1, L2, etc. are the individual loss components

as a percentage of the input power. It is often useful to

analyze individual losses to determine what is limiting

the efﬁciency and which change would produce the most

improvement. Although all dissipative elements in the

circuit produce losses, four main sources usually account

for the majority of the losses in LTC1871-1 applica-

tion circuits:

5V output power supply shown in Figure 1. The maxi-

mum load current is 7A (10A peak) and the duty cycle

is 39%. Assuming a ripple current of 40%, the peak

inductor current is 13.8A and the average is 11.5A.

With a maximum sense voltage of about 140mV, the

sense resistor value would be 10mΩ, and the power

18711fb

19

LTC1871-1

APPLICATIONS INFORMATION

dissipated in this resistor would be 514mW at maxi-

mum output current. Assuming an efﬁciency of 90%,

this sense resistor power dissipation represents 1.3%

of the overall input power. In other words, for this ap-

V

= 3.3V

OUT

MODE/SYNC = INTV

IN

= 5V

CC

I

(PULSE-SKIP MODE)

OUT

2V/DIV

plication, the use of V sensing would increase the

DS

efﬁciency by approximately 1.3%.

V

(AC)

OUT

For more details regarding the various terms in these

equations, please refer to the section Boost Converter:

Power MOSFET Selection.

100mV/DIV

18711 F13

100μs/DIV

3. The losses in the inductor are simply the DC input cur-

rent squared times the winding resistance. Expressing

this loss as a function of the output current yields:

Figure 13. Load Transient Response for a 3.3V Input,

5V Output Boost Converter Application, 0.7A to 7A Step

ꢀ

ꢃ²

ꢅ

I_O(MAX)

A second, more severe transient can occur when con-

necting loads with large (>1μF) supply bypass capacitors.

The discharged bypass capacitors are effectively put in

P_R(WINDING)

=

•R_W

ꢂ

1–D

ꢁ

_MAXꢄ

4. Losses in the boost diode. The power dissipation in the

boost diode is:

parallel with C , causing a nearly instantaneous drop in

O

V . No regulator can deliver enough current to prevent

O

P

= I

• V

O(MAX) D

DIODE

this problem if the load switch resistance is low and it is

driven quickly. The only solution is to limit the rise time

of the switch drive in order to limit the inrush current

di/dt to the load.

The boost diode can be a major source of power loss

in a boost converter. For the 3.3V input, 5V output at

7A example given above, a Schottky diode with a 0.4V

forwardvoltagewoulddissipate2.8W,whichrepresents

7%oftheinputpower.Diodelossescanbecomesigniﬁ-

cant at low output voltages where the forward voltage

is a signiﬁcant percentage of the output voltage.

Boost Converter Design Example

Thedesignexamplegivenherewillbeforthecircuitshown

in Figure 1. The input voltage is 3.3V, and the output is 5V

at a maximum load current of 7A (10A peak).

5. Other losses, including C and C ESR dissipation and

IN

O

inductor core losses, generally account for less than

1. The duty cycle is:

2% of the total additional loss.

ꢀ

_INꢃ

V_O+ V_D– V

5+ 0.4– 3.3

5+ 0.4

D=

=

= 38.9%

ꢂ

ꢅ

Checking Transient Response

V_O+ V_D

ꢁ

ꢄ

The regulator loop response can be veriﬁed by looking at

theloadtransientresponse.Switchingregulatorsgenerally

take several cycles to respond to an instantaneous step

2. Pulse-skip operation is chosen so the MODE/SYNC pin

is shorted to INTV .

CC

in resistive load current. When the load step occurs, V

3. The operating frequency is chosen to be 300kHz to

reduce the size of the inductor. From Figure 5, the

resistor from the FREQ pin to ground is 80k.

O

immediately shifts by an amount equal to (ΔI

)(ESR),

LOAD

and then C begins to charge or discharge (depending on

O

the direction of the load step) as shown in Figure 13. The

regulator feedback loop acts on the resulting error amp

4. An inductor ripple current of 40% of the maximum load

current is chosen, so the peak input current (which is

also the minimum saturation current) is:

output signal to return V to its steady-state value. During

O

this recovery time, V can be monitored for overshoot or

O

I_O(MAX)

ꢀ

7

ꢁ

ꢂ

ꢄ

ꢅ

ringing that would indicate a stability problem.

I

= 1+

•

=1.2•

= 13.8A

ꢃ

ꢆ

IN(PEAK)

2

1–D_MAX

1– 0.39

18711fb

20

LTC1871-1

APPLICATIONS INFORMATION

The inductor ripple current is:

ESRceramic.Basedonamaximumoutputripplevoltage

of 1%, or 50mV, the bulk C needs to be greater than:

I_O(MAX)

7

ꢀI_L= ꢁ •

= 0.4•

= 4.6A

1–D_MAX

1– 0.39

I_OUT(MAX)

C_OUTꢀ

=

0.01• V_OUT• f

7A

And so the inductor value is:

V

3.3V

4.6A •300kHz

IN(MIN)

= 466μF

L =

•D_MAX

=

•0.39= 0.93μH

0.01•5V •300kHz

ꢀI_L• f

The RMS ripple current rating for this capacitor needs

to exceed:

The component chosen is a 1μH inductor made by

Sumida (part number CEP125-H 1ROMH) which has

a saturation current of greater than 20A.

V_O– V

IN(MIN)

I_RMS(COUT)ꢀI_O(MAX)

•

=

V

IN(MIN)

5. With the input voltage to the IC bootstrapped to the

output of the power supply (5V), a logic-level MOSFET

can be used. Because the duty cycle is 39%, the maxi-

mum SENSE pin threshold voltage is reduced from its

low duty cycle typical value of 150mV to approximately

140mV. Assuming a MOSFET junction temperature of

5V – 3.3V

3.3V

7A •

= 5A

To satisfy this high RMS current demand, four 150μF

Panasonic capacitors (EEFUEOJ151R) are required.

In parallel with these bulk capacitors, two 22μF, low

ESR (X5R) Taiyo Yuden ceramic capacitors (JMK-

325BJ226MM) are added for HF noise reduction.

Check the output ripple with a single oscilloscope

probe connected directly across the output capacitor

terminals, where the HF switching currents ﬂow.

125°C, the room temperature MOSFET R

be less than:

should

DS(ON)

1–D_MAX

R_DS(ON)ꢀ V_SENSE(MAX)•

ꢁ

ꢂ

ꢃ

ꢅ

ꢆ

1+

•I_O(MAX)•ꢈ_T

ꢄ

ꢇ

2

1– 0.39

= 0.140V •

= 6.8mꢉ

8. The choice of an input capacitor for a boost converter

dependsontheimpedanceofthesourcesupplyandthe

amount of input ripple the converter will safely toler-

ate. For this particular design and lab setup a 100μF

Sanyo Poscap (6TPC 100M), in parallel with two 22μF

Taiyo Yuden ceramic capacitors (JMK325BJ226MM)

is required (the input and return lead lengths are kept

to a few inches, but the peak input current is close to

20A!). As with the output node, check the input ripple

with a single oscilloscope probe connected across the

input capacitor terminals.

0.4

ꢂ

ꢃ

ꢅ

ꢇ

1+

•7A •1.5

ꢄ

ꢆ

2

The MOSFET used was the Fairchild FDS7760A, which

has a maximum R of 8mΩ at 4.5V V , a BV

DSS

DS(ON)

GS

of greater than 30V, and a gate charge of 37nC at 5V

V .

GS

6. The diode for this design must handle a maximum

DC output current of 10A and be rated for a minimum

reverse voltage of V , or 5V. A 25A, 15V diode from

OUT

On Semiconductor (MBRB2515L) was chosen for its

high power dissipation capability.

7. The output capacitor usually consists of a high valued

bulk C connected in parallel with a lower valued, low

18711fb

21

LTC1871-1

APPLICATIONS INFORMATION

PC Board Layout Checklist

the power MOSFET or the bottom terminal of the sense

resistor, 4) the negative terminal of the input capacitor

and 5) at least one via to the ground plane immediately

adjacent to Pin 6. The ground trace on the top layer of

the PC board should be as wide and short as possible

to minimize series resistance and inductance.

1. Inordertominimizeswitchingnoiseandimproveoutput

load regulation, the GND pin of the LTC1871-1 should

be connected directly to 1) the negative terminal of the

INTV_CCdecoupling capacitor, 2) the negative terminal

of the output decoupling capacitors, 3) the source of

V

IN

L1

JUMPER

R3

R4

R

C

J1

C

IN

PIN 1

LTC1871-1

R2

R1

T

C

VCC

SWITCH NODE IS ALSO

THE HEAT SPREADER

FOR L1, M1, D1

M1

PSEUDO-KELVIN

SIGNAL GROUND

CONNECTION

C

OUT

C

OUT

D1

VIAS TO GROUND

PLANE

V

OUT

TRUE REMOTE

OUTPUT SENSING

18711 F14

BULK C

LOW ESR CERAMIC

Figure 14. LTC1871-1 Boost Converter Suggested Layout

V

IN

R3

R4

L1

J1

1

2

10

9

C

SWITCH

NODE

C

RUN

SENSE

R

C

I

TH

V

IN

LTC1871-1

FB

R1

D1

3

4

5

8

7

6

INTV

CC

R2

FREQ

GATE

GND

M1

R

T

+

MODE/

SYNC

C

VCC

C

IN

GND

C

PSEUDO-KELVIN

GROUND CONNECTION

OUT

+

V

OUT

18711 F15

BOLD LINES INDICATE HIGH CURRENT PATHS

Figure 15. LTC1871-1 Boost Converter Layout Diagram

18711fb

22

LTC1871-1

APPLICATIONS INFORMATION

2. Beware of ground loops in multiple layer PC boards.

Try to maintain one central ground node on the board

and use the input capacitor to avoid excess input ripple

for high output current power supplies. If the ground

plane is to be used for high DC currents, choose a path

away from the small-signal components.

7. If a sense resistor is used in the source of the power

MOSFET, minimize the capacitance between the SENSE

pin trace and any high frequency switching nodes. The

LTC1871-1 contains an internal leading edge blanking

timeofapproximately180ns,whichshouldbeadequate

for most applications.

3. Place the C

capacitor immediately adjacent to the

8. For optimum load regulation and true remote sensing,

the top of the output resistor divider should connect

independently to the top of the output capacitor (Kelvin

connection), staying away from any high dV/dt traces.

Place the divider resistors near the LTC1871-1 in order

to keep the high impedance FB node short.

VCC

INTV and GND pins on the IC package. This capaci-

CC

tor carries high di/dt MOSFET gate drive currents. A

low ESR and ESL 4.7μF ceramic capacitor works well

here.

4. The high di/dt loop from the bottom terminal of the

output capacitor, through the power MOSFET, through

the boost diode and back through the output capacitors

should be kept as tight as possible to reduce inductive

ringing. Excess inductance can cause increased stress

on the power MOSFET and increase HF noise on the

output. If low ESR ceramic capacitors are used on the

output to reduce output noise, place these capacitors

close to the boost diode in order to keep the series

inductance to a minimum.

9. Forapplicationswithmultipleswitchingpowerconvert-

ers connected to the same input supply, make sure

that the input ﬁlter capacitor for the LTC1871-1 is not

shared with other converters. AC input current from

another converter could cause substantial input volt-

age ripple, and this could interfere with the operation

of the LTC1871-1. A few inches of PC trace or wire

(L ≈ 100nH) between the C of the LTC1871-1 and the

IN

actualsourceV shouldbesufﬁcienttopreventcurrent

IN

sharing problems.

5. Check the stress on the power MOSFET by measuring

its drain-to-source voltage directly across the device

terminals(referencethegroundofasinglescopeprobe

directly to the source pad on the PC board). Beware

of inductive ringing which can exceed the maximum

speciﬁed voltage rating of the MOSFET. If this ringing

cannot be avoided and exceeds the maximum rating

of the device, either choose a higher voltage device

or specify an avalanche-rated power MOSFET. Not all

MOSFETs are created equal (some are more equal than

others).

C1

D1

C

L1

V

OUT

•

+

R

V

SW

L2

L

IN

OUT

•

16a. SEPIC Topology

V

IN

V

OUT

•

+

R

V

L

IN

•

6. Place the small-signal components away from high

frequency switching nodes. In the layout shown in

Figure 14, all of the small-signal components have

been placed on one side of the IC and all of the power

components have been placed on the other. This also

allows the use of a pseudo-Kelvin connection for the

signal ground, where high di/dt gate driver currents

ﬂow out of the IC ground pin in one direction (to the

16b. Current Flow During Switch On-Time

V

IN

D1

V

OUT

•

+

R

V

L

IN

•

16c. Current Flow During Switch Off-Time

Figures 16. SEPIC Topology and Current Flow

bottom plate of the INTV decoupling capacitor) and

CC

small-signal currents ﬂow in the other direction.

18711fb

23

LTC1871-1

APPLICATIONS INFORMATION

SEPIC Converter Applications

independentbutcanalsobewoundonthesamecoresince

identical voltages are applied to L1 and L2 throughout the

switching cycle. By making L1 = L2 and winding them on

the same core the input ripple is reduced along with cost

and size. All of the SEPIC applications information that

follows assumes L1 = L2 = L.

The LTC1871-1 is also well suited to SEPIC (single-ended

primaryinductanceconverter)converterapplications. The

SEPIC converter shown in Figure 16 uses two inductors.

The advantage of the SEPIC converter is the input voltage

may be higher or lower than the output voltage, and the

output is short-circuit protected.

SEPIC Converter: Duty Cycle Considerations

The ﬁrst inductor, L1, together with the main switch,

resembles a boost converter. The second inductor, L2,

together with the output diode D1, resembles a ﬂyback or

buck-boost converter. The two inductors L1 and L2 can be

ForaSEPICconverteroperatinginacontinuousconduction

mode (CCM), the duty cycle of the main switch is:

ꢀ

ꢃ

ꢅ

V_O+ V_D

V + V + V

D=

ꢂ

ꢁ

_Dꢄ

IN

O

I

IN

I

L1

where V is the forward voltage of the diode. For convert-

D

SW

ON

SW

OFF

ers where the input voltage is close to the output voltage

the duty cycle is near 50%.

17a. Input Inductor Current

The maximum output voltage for a SEPIC converter is:

I

O

I

L2

D_MAX

1

V_O(MAX)= V + V

– V

^D1–D_MAX

(

)

IN

D

1–D_MAX

17b. Output Inductor Current

The maximum duty cycle of the LTC1871-1 is typically

92%.

I

IN

O

I

C1

SEPIC Converter: The Peak and Average

Input Currents

17c. DC Coupling Capacitor Current

ThecontrolcircuitintheLTC1871-1ismeasuringtheinput

current (either using the R

of the power MOSFET

DS(ON)

or by means of a sense resistor in the MOSFET source),

so the output current needs to be reﬂected back to the

input in order to dimension the power MOSFET properly.

Based on the fact that, ideally, the output power is equal

to the input power, the maximum input current for a SEPIC

converter is:

I

D1

I

O

17d. Diode Current

D_MAX

1–D_MAX

V

OUT

I_IN(MAX)=I_O(MAX)•

(AC)

ΔV

COUT

ΔV

ESR

Thepeak input current is:

RINGING DUE TO

TOTAL INDUCTANCE

(BOARD + CAP)

D_MAX

1–D_MAX

ꢀ

ꢁ

ꢄ

ꢆ

I

= 1+

•I

•

ꢃ

IN(PEAK)

O(MAX)

ꢂ

ꢅ

2

17e. Output Ripple Voltage

The maximum duty cycle, D

, should be calculated at

MAX

Figure 17. SEPIC Converter Switching Waveforms

minimum V .

IN

18711fb

24

LTC1871-1

APPLICATIONS INFORMATION

Theconstant‘χ’representsthefractionofripplecurrentin

the inductor relative to its maximum value. For example, if

30% ripple current is chosen, then χ = 0.30 and the peak

current is 15% greater than the average.

By making L1 = L2 and winding them on the same core,

the value of inductance in the equation above is replace

by 2L due to mutual inductance. Doing this maintains the

sameripplecurrentandenergystorageintheinductors.For

example, aCoiltronixCTX10-4isa10μHinductorwithtwo

windings.Withthewindingsinparallel,10μHinductanceis

obtained with a current rating of 4A (the number of turns

hasn’t changed, but the wire diameter has doubled). Split-

ting the two windings creates two 10μH inductors with a

currentratingof2Aeach. Therefore, substituting2Lyields

the following equation for coupled inductors:

It is worth noting here that SEPIC converters that operate

at high duty cycles (i.e., that develop a high output volt-

age from a low input voltage) can have very high input

currents, relative to the output current. Be sure to check

that the maximum load current will not overload the input

supply.

V

SEPIC Converter: Inductor Selection

IN(MIN)

L1=L2=

•D_MAX

2• ꢀI_L• f

For most SEPIC applications the equal inductor values

will fall in the range of 10μH to 100μH. Higher values will

reduce the input ripple voltage and reduce the core loss.

Lower inductor values are chosen to reduce physical size

and improve transient response.

Specify the maximum inductor current to safely handle

I

speciﬁed in the equation above. The saturation

L(PK)

current rating for the inductor should be checked at the

minimum input voltage (which results in the highest

inductor current) and maximum output current.

Like the boost converter, the input current of the SEPIC

converter is calculated at full load current and minimum

inputvoltage.Thepeakinductorcurrentcanbesigniﬁcantly

higher than the output current, especially with smaller in-

ductors and lighter loads. The following formulas assume

CCM operation and calculate the maximum peak inductor

SEPIC Converter: Power MOSFET Selection

ThepowerMOSFETservestwopurposesintheLTC1871-1:

itrepresentsthemainswitchingelementinthepowerpath,

and its R

represents the current sensing element

DS(ON)

currents at minimum V :

IN

for the control loop. Important parameters for the power

MOSFET include the drain-to-source breakdown voltage

V_O+ V_D

ꢀ

ꢁ

ꢂ

ꢄ

ꢅ

I_L1(PEAK)= 1+

•I

•

ꢃ

ꢆ

O(MAX)

(BV ),thethresholdvoltage(V ),theon-resistance

2

V

DSS

DS(ON)

GS(TH)

IN(MIN)

(R

)versusgate-to-sourcevoltage,thegate-to-source

V

_IN(MIN)+ V_D

ꢀ

ꢁ

ꢂ

ꢄ

ꢅ

and gate-to-drain charges (Q and Q , respectively),

GS

D(MAX)

GD

I_L2(PEAK)= 1+

•I •

O(MAX)

ꢃ

ꢆ

the maximum drain current (I

thermal resistances (R

) and the MOSFET’s

2

V

IN(MIN)

and R

).

TH(JC)

TH(JA)

The ripple current in the inductor is typically 20% to 40%

Thegatedrivevoltageissetbythe5.2VINTV lowdropout

CC

(i.e., a range of ‘χ’ from 0.20 to 0.40) of the maximum

regulator. Consequently, logic-level threshold MOSFETs

should be used in most LTC1871-1 applications. If low

input voltage operation is expected (e.g., supplying power

from a lithium-ion battery), then sublogic-level threshold

MOSFETs should be used.

averageinputcurrentoccurringatV

andI

and

IN(MIN)

O(MAX)

ΔI = ΔI . Expressing this ripple current as a function of

L1

L2

the output current results in the following equations for

calculating the inductor value:

V

IN(MIN)

L =

•D_MAX

The maximum voltage that the MOSFET switch must

sustain during the off-time in a SEPIC converter is equal

ꢀI_L• f

to the sum of the input and output voltages (V + V ).

O

IN

where:

As a result, careful attention must be paid to the BV

DSS

D_MAX

1–D_MAX

speciﬁcations for the MOSFETs relative to the maximum

actual switch voltage in the application. Many logic-level

18711fb

ꢀI_L= ꢁ •I_O(MAX)

•

25

LTC1871-1

APPLICATIONS INFORMATION

devices are limited to 30V or less. Check the switching

waveforms directly across the drain and source terminals

that the converter is capable of delivering the required

load current over all operating conditions (load, line and

temperature) and for the worst-case speciﬁcations for

of the power MOSFET to ensure the V remains below

DS

the maximum rating for the device.

V

and the R

of the MOSFET listed in the

SENSE(MAX)

DS(ON)

manufacturer’s data sheet.

During the MOSFET’s on-time, the control circuit limits

the maximum voltage drop across the power MOSFET to

about150mV(atlowdutycycle).Thepeakinductorcurrent

ThepowerdissipatedbytheMOSFETinaSEPICconverter

is:

is therefore limited to 150mV/R . The relationship

DS(ON)

ꢀ

ꢃ²

ꢅ

between the maximum load current, duty cycle and the

of the power MOSFET is:

D_MAX

R

DS(ON)

P

_FET= I_O(MAX)

•

•R_DS(ON)•D_MAX•ꢆ_T

ꢂ

1–D

ꢁ

_MAXꢄ

V_SENSE(MAX)

1

+ k • V_IN(MIN)+ V_O^1.85•I_O(MAX)

•

•C_RSS• f

R_DS(ON)ꢀ

•

D_MAX

1–D_MAX

ꢁ

I_O(MAX)

ꢂ

ꢃ

ꢅ

ꢇ

ꢂ

ꢅ

(

)

V_O+ V_D

1+

•ꢈ

T

ꢄ

+1

ꢄ

ꢇ

ꢆ

2

V

ꢃ

ꢆ

IN(MIN)

2

The ﬁrst term in the equation above represents the I R

losses in the device and the second term, the switching

losses.Theconstant k=1.7isanempiricalfactorinversely

related to the gate drive current and has the dimension

of 1/current.

The V

term is typically 150mV at low duty cycle

SENSE(MAX)

and is reduced to about 100mV at a duty cycle of 92% due

toslopecompensation,asshowninFigure8.Theconstant

‘χ’ in the denominator represents the ripple current in the

inductors relative to their maximum current. For example,

From a known power dissipated in the power MOSFET, its

junction temperature can be obtained using the following

formula:

if 30% ripple current is chosen, then χ = 0.30. The ρ term

T

accounts for the temperature coefﬁcient of the R

of

DS(ON)

theMOSFET,whichistypically0.4%/°C.Figure9illustrates

T = T + P •R

J

A

FET TH(JA)

the variation of normalized R

a typical power MOSFET.

over temperature for

DS(ON)

The R

to be used in this equation normally includes

TH(JA)

the R

for the device plus the thermal resistance from

TH(JC)

Another method of choosing which power MOSFET to

use is to check what the maximum output current is for a

the board to the ambient temperature in the enclosure.

This value of T can then be used to check the original

J

given R

since MOSFET on-resistances are available

DS(ON)

assumption for the junction temperature in the iterative

in discrete values.

V_SENSE(MAX)

calculation process.

1

I

_O(MAX)ꢀ

•

ꢁ

R_DS(ON)

ꢂ

ꢃ

ꢅ

ꢇ

ꢂ

ꢅ

V_O+ V_D

SEPIC Converter: Output Diode Selection

1+

•ꢈ

T

ꢄ

+1

ꢄ

ꢇ

ꢆ

2

V

ꢃ

ꢆ

IN(MIN)

To maximize efﬁciency, a fast-switching diode with low

forwarddropandlowreverseleakageisdesired.Theoutput

diode in a SEPIC converter conducts current during the

switch off-time. The peak reverse voltage that the diode

Calculating Power MOSFET Switching and Conduction

Losses and Junction Temperatures

In order to calculate the junction temperature of the

power MOSFET, the power dissipated by the device must

be known. This power dissipation is a function of the

duty cycle, the load current and the junction temperature

itself. As a result, some iterative calculation is normally

required to determine a reasonably accurate value. Since

the controller is using the MOSFET as both a switching

and a sensing element, care should be taken to ensure

must withstand is equal to V

+ V . The average

IN(MAX)

O

forward current in normal operation is equal to the output

current, and the peak current is equal to:

ꢁ

ꢄ

ꢆ

V_O+ V

ꢀ

ꢁ

ꢂ

ꢄ

ꢅ

I_D(PEAK)= 1+

•I

•

^D+1

ꢃ

ꢆ

O(MAX)

2

V

ꢂ

ꢅ

IN(MIN)

The power dissipated by the diode is:

P = I • V

D

O(MAX)

D

18711fb

26

LTC1871-1

APPLICATIONS INFORMATION

and the diode junction temperature is:

Formanydesignsitispossibletochooseasinglecapacitor

type that satisﬁes both the ESR and bulk C requirements

forthedesign.Incertaindemandingapplications,however,

the ripple voltage can be improved signiﬁcantly by con-

necting two or more types of capacitors in parallel. For

example, using a low ESR ceramic capacitor can minimize

the ESR step, while an electrolytic or tantalum capacitor

can be used to supply the required bulk C.

T = T + P • R

J

A

D

TH(JA)

The R

the R

to be used in this equation normally includes

TH(JA)

for the device plus the thermal resistance from

TH(JC)

the board to the ambient temperature in the enclosure.

SEPIC Converter: Output Capacitor Selection

Because of the improved performance of today’s electro-

lytic, tantalum and ceramic capacitors, engineers need

to consider the contributions of ESR (equivalent series

resistance), ESL (equivalent series inductance) and the

bulk capacitance when choosing the correct component

for a given output ripple voltage. The effects of these three

parameters (ESR, ESL, and bulk C) on the output voltage

ripple waveform are illustrated in Figure 17 for a typical

coupled-inductor SEPIC converter.

Once the output capacitor ESR and bulk capacitance have

been determined, the overall ripple voltage waveform

should be veriﬁed on a dedicated PC board (see Board

Layout section for more information on component place-

ment). Lab breadboards generally suffer from excessive

series inductance (due to inter-component wiring), and

these parasitics can make the switching waveforms look

signiﬁcantly worse than they would be on a properly

designed PC board.

The choice of component(s) begins with the maximum

acceptable ripple voltage (expressed as a percentage of

the output voltage), and how this ripple should be divided

between the ESR step and the charging/discharging ΔV.

For the purpose of simplicity we will choose 2% for the

maximum output ripple, to be divided equally between the

ESRstepandthecharging/dischargingΔV.Thispercentage

ripple will change, depending on the requirements of the

application, and the equations provided below can easily

be modiﬁed.

The output capacitor in a SEPIC regulator experiences

high RMS ripple currents, as shown in Figure 17. The

RMS output capacitor ripple current is:

V_O+ V_D

I_RMS(C1)=I_O(MAX)•

V

IN(MIN)

Note that the ripple current ratings from capacitor manu-

facturers are often based on only 2000 hours of life. This

makes it advisable to further derate the capacitor or to

choose a capacitor rated at a higher temperature than

required. Several capacitors may also be placed in parallel

to meet size or height requirements in the design.

For a 1% contribution to the total ripple voltage, the ESR

of the output capacitor can be determined using the fol-

lowing equation:

Manufacturers such as Nichicon, United Chemicon and

Sanyoshouldbeconsideredforhighperformancethrough-

hole capacitors. The OS-CON semiconductor dielectric

capacitor available from Sanyo has the lowest product of

ESR and size of any aluminum electrolytic, at a somewhat

higher price.

0.01• V_O

ESR_COUTꢀ

I_IN(PEAK)

where:

ꢁ

ꢄ

V_O+ V

ꢀ

ꢁ

ꢂ

ꢄ

ꢅ

I_D(PEAK)= 1+

•I

•

^D+1

ꢃ

ꢆ

ꢃ

ꢆ

O(MAX)

2

V

ꢂ

ꢅ

In surface mount applications, multiple capacitors may

have to be placed in parallel in order to meet the ESR or

RMS current handling requirements of the application.

Aluminum electrolytic and dry tantalum capacitors are

both available in surface mount packages. In the case of

tantalum, it is critical that the capacitors have been surge

IN(MIN)

For the bulk C component, which also contributes 1% to

the total ripple:

I_O(MAX)

C_OUTꢀ

0.01• V_O• f

tested for use in switching power supplies. An excellent

18711fb

27

LTC1871-1

APPLICATIONS INFORMATION

choice is AVX TPS series of surface mount tantalum. Also,

ceramic capacitors are now available with extremely low

ESR, ESL and high ripple current ratings.

which is typically close to V

. The ripple current

IN(MAX)

through C1 is:

V_O+ V_D

I_RMS(C1)=I_O(MAX)•

V

SEPIC Converter: Input Capacitor Selection

IN(MIN)

The input capacitor of a SEPIC converter is less critical

than the output capacitor due to the fact that an inductor

is in series with the input and the input current waveform

istriangularinshape. Theinputvoltagesourceimpedance

determines the size of the input capacitor which is typi-

cally in the range of 10μF to 100μF. A low ESR capacitor

is recommended, although it is not as critical as for the

output capacitor.

The value chosen for the DC coupling capacitor normally

starts with the minimum value that will satisfy 1) the RMS

current requirement and 2) the peak voltage requirement

(typically close to V ). Low ESR ceramic and tantalum

IN

capacitors work well here.

SEPIC Converter Design Example

Thedesignexamplegivenherewillbeforthecircuitshown

in Figure 18. The input voltage is 5V to 15V and the output

is 12V at a maximum load current of 1.5A (2A peak).

The RMS input capacitor ripple current for a SEPIC con-

verter is:

1

12

I_RMS(CIN)

=

• ꢀI_L

1. The duty cycle range is:

ꢀ

ꢃ

ꢅ

V_O+ V_D

V + V + V

D=

= 45.5% to 71.4%

Please note that the input capacitor can see a very high

surge current when a battery is suddenly connected to

the input of the converter and solid tantalum capacitors

can fail catastrophically under these conditions. Be sure

to specify surge-tested capacitors!

ꢂ

ꢁ

_Dꢄ

IN

O

2. The operating mode chosen is pulse skipping, so the

MODE/SYNC pin is shorted to INTV .

CC

3. The operating frequency is chosen to be 300kHz to

reduce the size of the inductors; the resistor from the

FREQ pin to ground is 80k.

SEPIC Converter: Selecting the DC Coupling Capacitor

ThecouplingcapacitorC1inFigure16seesnearlyarectan-

gular current waveform as shown in Figure 17. During the

4. Aninductorripplecurrentof40%ischosen,sothepeak

input current (which is also the minimum saturation

current) is:

switch off-time the current through C1 is I (V /V ) while

O

IN

approximately –I ﬂows during the on-time. This current

O

waveform creates a triangular ripple voltage on C1:

V_O+ V_D

ꢀ

ꢁ

ꢂ

ꢄ

ꢅ

I_L1(PEAK)= 1+

•I

•

ꢃ

ꢆ

O(MAX)

I_O(MAX)

2

V

V_O

C1• f V + V_O+ V_D

IN(MIN)

ꢀV_C1(PꢁP)

=

•

IN

0.4

2

12+ 0.5

ꢁ

ꢂ

ꢄ

ꢅ

= 1+

•1.5•

= 4.5A

ꢃ

ꢆ

5

The maximum voltage on C1 is then:

The inductor ripple current is:

ꢀV_C1(PꢁP)

V

_C1(MAX)= V +

IN

D_MAX

2

ꢀI_L= ꢁ •I_O(MAX)

•

1–D_MAX

0.714

1– 0.714

= 0.4•1.5•

=1.5A

18711fb

28

LTC1871-1

APPLICATIONS INFORMATION

And so the inductor value is:

V_SENSE(MAX)

1

R_DS(ON)ꢀ

•

V

5

IN(MIN)

ꢁ

I_O(MAX)

ꢂ

ꢃ

ꢅ

ꢇ

ꢂ

ꢅ

L =

•D_MAX

=

•0.714= 4μH

V_O+ V_D

1+

•ꢈ

T

ꢄ

+1

2• ꢀI_L• f

2•1.5•300k

ꢄ

ꢇ

ꢆ

2

V

ꢃ

ꢆ

IN(MIN)

T

he component chosen is a BH Electronics BH510-

0.12

1.5 1.2•1.5 ꢂ

1

•

1

12.5

5

=

•

=12.7mꢉ

1007, which has a saturation current of 8A.

ꢅ

ꢆ

+1

ꢇ

ꢄ

ꢃ

5. With an minimum input voltage of 5V, only logic-level

power MOSFETs should be considered. Because the

maximum duty cycle is 71.4%, the maximum SENSE

pin threshold voltage is reduced from its low duty

cycle typical value of 150mV to approximately 120mV.

Assuming a MOSFET junction temperature of 125°C,

For a SEPIC converter, the switch BV

greater than V

anIRF7811W,whichisratedto30V,andhasamaximum

room temperature R

rating must be

DSS

+ V , or 27V. This comes close to

IN(MAX)

O

of 12mΩ at V = 4.5V.

DS(ON)

GS

the room temperature MOSFET R

than:

should be less

DS(ON)

V

IN

4.5V to 15V

•

R3

1M

C

DC

L1*

10μF

25V

1

10

X5R

RUN

SENSE

D1

V

OUT

2

9

12V

I

V

IN

TH

1.5A

C

OUT1

(2A PEAK)

R

+

LTC1871-1

INTV

C

47μF

33k

3

4

5

8

7

6

C

OUT2

20V

×2

FB

CC

C

10μF

25V

X5R

×2

R1

12.1k

1%

C1

FREQ

GATE

GND

M1

6.8nF

L2*

+

C

4.7μF

X5R

R2

105k

1%

R

T

80.6k

1%

VCC

MODE/SYNC

C

IN

47μF

•

C

C2

47pF

GND

18711 F018a

C

, C

:

KEMET T495X476K020AS

L1, L2: BH ELECTRONICS BH510-1007 (*COUPLED INDUCTORS)

M1: INTERNATIONAL RECTIFIER IRF7811W

IN OUT1

, C

: TAIYO YUDEN TMK432BJ106MM

DC OUT2

D1:

INTERNATIONAL RECTIFIER 30BQ040

Figure 18a. 4.5V to 15V Input, 12V/2A Output SEPIC Converter

100

95

V

= 12V

IN

90

85

80

75

70

65

60

55

50

45

V

= 4.5V

IN

V

= 15V

IN

V

= 12V

O

MODE = INTV

CC

0.001

0.01

0.1

1 10

OUTPUT CURRENT (A)

18711 F18b

Figure 18b. SEPIC Efﬁciency vs Output Current

18711fb

29

LTC1871-1

APPLICATIONS INFORMATION

V

= 4.5V

OUT

V

= 15V

IN

OUT

= 12V

V

(AC)

V

(AC)

OUT

200mV/DIV

OUT

200mV/DIV

I

OUT

0.5A/DIV

OUT

0.5A/DIV

18711 F19

50μs/DIV

Figure 19. LTC1871-1 SEPIC Converter Load Step Response

6. The diode for this design must handle a maximum

DC output current of 2A and be rated for a minimum

with a single oscilloscope probe connected directly

across the output capacitor terminals, where the HF

switching currents ﬂow.

reverse voltage of V + V , or 27V. A 3A, 40V diode

IN

OUT

from International Rectiﬁer (30BQ040) is chosen for its

small size, relatively low forward drop and acceptable

reverse leakage at high temp.

8. The choice of an input capacitor for a SEPIC converter

dependsontheimpedanceofthesourcesupplyandthe

amount of input ripple the converter will safely toler-

ate. For this particular design and lab setup, a single

47μF Kemet tantalum capacitor (T495X476K020AS) is

adequate.Aswiththeoutputnode,checktheinputripple

with a single oscilloscope probe connected across the

input capacitor terminals. If any HF switching noise is

observed it is a good idea to decouple the input with

7. The output capacitor usually consists of a high valued

bulkCconnectedinparallelwithalowervalued,lowESR

ceramic. Based on a maximum output ripple voltage of

1%, or 120mV, the bulk C needs to be greater than:

I_OUT(MAX)

C

_OUTꢀ

=

0.01• V_OUT• f

1.5A

a low ESR, X5R ceramic capacitor as close to the V

and GND pins as possible.

IN

= 41μF

0.01•12V •300kHz

9. The DC coupling capacitor in a SEPIC converter is cho-

sen based on its RMS current requirement and must be

The RMS ripple current rating for this capacitor needs

to exceed:

rated for a minimum voltage of V plus the AC ripple

IN

voltage. Start with the minimum value which satisﬁes

the RMS current requirement and then check the ripple

voltage to ensure that it doesn’t exceed the DC rating.

V_O

I

_RMS(COUT)ꢀI_O(MAX)

•

=

V

IN(MIN)

12V

5V

1.5A •

= 2.3A

V_O+ V_D

I_RMS(CI)ꢀI_O(MAX)•

V

IN(MIN)

To satisfy this high RMS current demand, two 47μF

Kemetcapacitors(T495X476K020AS)arerequired.Asa

result, the output ripple voltage is a low 50mV to 60mV.

Inparallelwiththesetantalums,two10μF,lowESR(X5R)

TaiyoYudenceramiccapacitors(TMK432BJ106MM)are

added for HF noise reduction. Check the output ripple

12V + 0.5V

=1.5A •

= 2.4A

5V

For this design a single 10μF, low ESR (X5R) Taiyo

Yuden ceramic capacitor (TMK432BJ106MM) is

adequate.

18711fb

30

LTC1871-1

TYPICAL APPLICATIONS

2.5V to 3.3V Input, 5V/2A Output Boost Converter

V

IN

2.5V to 3.3V

L1

1.8μH

D1

1

2

10

9

RUN

SENSE

V

5V

2A

OUT

I

TH

V

IN

C

OUT1

R

C

+

LTC1871-1

INTV

150μF

6.3V

×2

22k

3

4

5

8

7

6

C

OUT2

FB

CC

C

6.8nF

10μF

6.3V

X5R

×2

R1

12.1k

1%

C1

FREQ

GATE

GND

M1

C

4.7μF

X5R

R2

37.4k

1%

R

80.6k

1%

+

VCC

C

IN

47μF

6.3V

MODE/SYNC

T

C

C2

47pF

GND

18711 TA01a

C

:

SANYO POSCAP 6TPA47M

: SANYO POSCAP 6TPB150M

: TAIYO YUDEN JMK316BJ106ML

TAIYO YUDEN LMK316BJ475ML

D1: INTERNATIONAL RECTIFIER 30BQ015

L1: TOKO DS104C2 B952AS-1R8N

M1: SILICONIX/VISHAY Si9426

IN

OUT1

OUT2

VCC

:

Output Efﬁciency at 2.5V and 3.3V Input

100

95

90

85

80

75

70

65

60

55

50

0.001

0.01

0.1

1

10

OUTPUT CURRENT (A)

18711 TA01b

18711fb

31

LTC1871-1

TYPICAL APPLICATIONS

18V to 27V Input, 28V Output, 400W 2-Phase, Low Ripple, Synchronized RF Base Station Power Supply (Boost)

V

IN

18V to 27V

R1

93.1k

1%

L1

5.6μH

L2

R2

8.45k

1%

5.6μH

C

+

IN1

330μF

50V

1

2

10

9

RUN

SENSE

D1

I

TH

V

IN

C

C1

47pF

LTC1871-1

INTV

C

OUT1

3

4

5

8

7

6

2.2μF

35V

X5R

×3

C

+

FB

OUT2

CC

330μF

50V

FREQ

GATE

GND

M1

C

IN2

R

150k

5%

C

4.7μF

X5R

MODE/SYNC

T1

VCC1

2.2μF

35V

C

R

S1

FB1

47pF

0.007Ω

1W

X5R

GND

C

*

OUT5

330μF

50V

×4

+

EXT CLOCK

C

*

OUT6

L3

5.6μH

L4

INPUT (200kHz)

2.2μF

35V

5.6μH

X5R

1

2

10

9

RUN

SENSE

D2

V

OUT

28V

14A

I

TH

V

IN

L5*

0.3μH

LTC1871-1

INTV

C

OUT3

R

C

C2

3

4

5

8

7

6

2.2μF

35V

X5R

×3

C

+

*L5, C

AND

FB

22k

OUT4

OUT5

ARE AN

47pF

CC

330μF

50V

C

OUT6

FB2

47pF

FREQ

GATE

GND

M2

OPTIONAL SECONDARY

FILTER TO REDUCE

C

IN3

R4

R3

12.1k

1%

R

150k

5%

MODE/SYNC

C

4.7μF

X5R

T2

VCC2

2.2μF

35V

C

6.8nF

C3

R

S2

0.007Ω

1W

OUTPUT RIPPLE FROM

<500mV TO <100mV

261k

1%

P-P

X5R

18711 TA04

C

:

SANYO 50MV330AX

TAIYO YUDEN GMK325BJ225MN

SANYO 50MV330AX

TAIYO YUDEN GMK325BJ225MN

TAIYO YUDEN LMK316BJ475ML

L1 TO L4: SUMIDA CEP125-5R6MC-HD

L5: SUMIDA CEP125-0R3NC-ND

D1, D2: ON SEMICONDUCTOR MBR2045CT

M1, M2: INTERNATIONAL RECTIFIER IRLZ44NS

IN1

:

IN2, 3

OUT2, 4, 5

OUT1, 3, 6

:

VCC1, 2

5V to 12V Input, 12V/0.2A Output SEPIC Converter with Undervoltage Lockout

V

IN

5V to 12V

•

R1

127k

1%

C

DC1

R2

54.9k

1%

L1*

4.7μF

16V

1

2

10

9

X5R

RUN

SENSE

D1

V

OUT1

12V

I

V

IN

TH

0.4A

R

C

22k

LTC1871-1

INTV

C

OUT1

3

4

5

8

7

6

4.7μF

16V

X5R

×3

FB

CC

C

R4

127Ω

1%

C1

L2*

FREQ

GATE

GND

M1

6.8nF

C

IN1

IN2

VCC

MODE/SYNC

+

R3

1.10k

1%

R

T

60.4k

1%

1μF

16V

X5R

47μF

16V

•

4.7μF

10V

C

C2

R

S

100pF

0.02Ω

AVX

X5R

GND

V

C

OUT2

C

DC2

4.7μF

16V

X5R

×3

NOTE:

D1, D2: MBS120T3

L1 TO L3: COILTRONICS VP1-0076 (*COUPLED INDUCTORS)

M1: SILICONIX/VISHAY Si4840

4.7μF

+

–

D2

1. V UVLO = 4.47V

IN

16V

X5R

V

UVLO = 4.14V

IN

L3*

OUT2

•

–12V

0.4A

18711 TA03

18711fb

32

LTC1871-1

TYPICAL APPLICATIONS

4.5V to 28V Input, 5V/2A Output SEPIC Converter with Undervoltage Lockout and Soft-Start

V

IN

4.5V to 28V

R1

•

C

DC

115k

2.2μF

25V

X5R

×3

R2

54.9k

1%

L1*

C1

4.7nF

1

2

10

9

RUN

SENSE

D1

V

OUT

5V

I

V

IN

TH

2A

(3A TO 4A PEAK)

R

+

LTC1871-1

INTV

C

OUT1

12k

3

4

5

8

7

6

330μF

6.3V

FB

CC

C

OUT2

R4

49.9k

1%

C

C1

22μF

6.3V

X5R

FREQ

GATE

GND

M1

L2*

8.2nF

C

IN1

VCC

+

C

22μF

35V

IN2

R3

154k

1%

R

T

162k

1%

MODE/SYNC

4.7μF

10V

2.2μF

35V

•

C

47pF

C2

X5R

GND

18711 TA02a

R5

100Ω

C2

1μF

NOTES:

1. V UVLO = 4.17V

+

–

Q1

X5R

IN

R6

750Ω

V

UVLO = 3.86V

IN

2. SOFT-START dV /dt = 5V/6ms

OUT

C

, C : TAIYO YUDEN GMK325BJ225MN

D1:

INTERNATIONAL RECTIFIER 30BQ040

IN1 DC

IN2

:

AVX TPSE226M035R0300

SANYO 6TPB330M

TAIYO YUDEN JMK325BJ226MN

LMK316BJ475ML

L1, L2: BH ELECTRONICS BH510-1007 (*COUPLED INDUCTORS)

:

M1:

Q1:

SILICONIX/VISHAY Si4840

PHILIPS BC847BF

OUT1

OUT2

:

VCC

Soft-Start

Load Step Response at V_IN= 4.5V

V

OUT

100mV/DIV

(AC)

V

OUT

1V/DIV

2.2A

I

OUT

1A/DIV

(DC)

0.5A

18711 TA02b

18711 TA02c

1ms/DIV

250μs/DIV

Load Step Response at V_IN= 28V

V

OUT

100mV/DIV

(AC)

2.2A

I

OUT

1A/DIV

(DC)

0.5A

18711 TA02d

250μs/DIV

18711fb

33

LTC1871-1

TYPICAL APPLICATIONS

5V to 15V Input, –5V/5A Output Positive-to-Negative Converter with Undervoltage Lockout and Level-Shifted Feedback

V

IN

5V to 15V

V

–5V

5A

OUT

•

R1

154k

1%

R2

68.1k

1%

L1*

L2*

C1

1nF

1

2

10

9

RUN

SENSE

C

OUT

I

V

IN

TH

100μF

6.3V

X5R

×2

C

22μF

25V

X7R

M1

DC

LTC1871-1

INTV

3

4

5

8

7

6

R

C

FB

CC

10k

FREQ

GATE

GND

C

C2

MODE/SYNC

330pF

D1

C

IN

VCC

R

80.6k

1%

T

4.7μF

10V

47μF

16V

C

10nF

C1

X5R

GND

R4

10k

1%

C2

10nF

R5

40.2k

1%

R3

10k

1%

6

–

4

1

LT1783

2

+

3

18711 TA05

C

:

TDK C5750X5R1C476M

TDK C5750X7R1E226M

: TDK C5750X5R0J107M

: TAIYO YUDEN LMK316BJ475ML

D1:

ON SEMICONDUCTOR MBRB2035CT

IN

C

:

L1, L2: COILTRONICS VP5-0053 (*3 WINDINGS IN PARALLEL

FOR THE PRIMARY, 3 IN PARALLEL FOR SECONDARY)

DC

OUT

VCC

M1:

INTERNATIONAL RECTIFIER IRF7822

18711fb

34

LTC1871-1

PACKAGE DESCRIPTION

MS Package

10-Lead Plastic MSOP

(Reference LTC DWG # 05-08-1661 Rev E)

0.889 ± 0.127

(.035 ± .005)

5.23

(.206)

MIN

3.20 – 3.45

(.126 – .136)

3.00 ± 0.102

(.118 ± .004)

(NOTE 3)

0.497 ± 0.076

(.0196 ± .003)

0.50

0.305 ± 0.038

(.0120 ± .0015)

TYP

(.0197)

10 9

8

7 6

BSC

REF

RECOMMENDED SOLDER PAD LAYOUT

3.00 ± 0.102

(.118 ± .004)

(NOTE 4)

4.90 ± 0.152

(.193 ± .006)

DETAIL “A”

0° – 6° TYP

0.254

(.010)

GAUGE PLANE

1

2

3

4 5

0.53 ± 0.152

(.021 ± .006)

0.86

(.034)

REF

1.10

(.043)

MAX

DETAIL “A”

0.18

(.007)

SEATING

PLANE

0.17 – 0.27

(.007 – .011)

TYP

0.1016 ± 0.0508

(.004 ± .002)

0.50

(.0197)

BSC

MSOP (MS) 0307 REV E

NOTE:

1. DIMENSIONS IN MILLIMETER/(INCH)

2. DRAWING NOT TO SCALE

3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.

MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE

4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.

INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE

5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX

18711fb

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-

tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.

35

LTC1871-1

TYPICAL APPLICATION

High Power SLIC Supply with Undervoltage Lockout

(Also See the LTC3704 Data Sheet)

GND

•

4

C3

D2

10BQ060

10μF

25V

X5R

V

IN

R1

49.9k

1%

R2

150k

1%

7V TO 12V

V

OUT1

–24V

C

200mA

C

R

1nF

IN

•

5

+

C4

D3

10BQ060

220μF

16V

C

3.3μF

100V

OUT

10μF

25V

X5R

T1*

1, 2, 3

TPS

C

•

C2

100pF

RUN

SENSE

I

V

IN

TH

•

6

D4

C5

LTC1871-1

C2

10BQ060

10μF

25V

X5R

R

C

FB

INTV

CC

4.7μF

50V

82k

FREQ

GATE

GND

IRL2910

V

OUT2

X5R

–72V

C

C1

+

C1

4.7μF

X5R

MODE/SYNC

R

200mA

1nF

R

F2

196k

1%

T

F1

10k

1%

R

120k

S

f = 200kHz

0.012Ω

6

–

4

*COILTRONICS VP5-0155

(PRIMARY = 3 WINDINGS IN PARALLEL)

10k

1

LT1783

2

+

3

C8

0.1μF

18711 TA06

RELATED PARTS

PART NUMBER

LT^®1619

DESCRIPTION

COMMENTS

Current Mode PWM Controller

Current Mode DC/DC Controller

300kHz Fixed Frequency, Boost, SEPIC, Flyback Topology

LTC1624

SO-8; 300kHz Operating Frequency; Buck, Boost, SEPIC Design;

V

Up to 36V

IN

LTC1700

LTC1871

LTC1871-7

LTC1872

LT1930

No R

Synchronous Step-Up Controller

Up to 95% Efﬁciency, Operation as Low as 0.9V Input

SENSE

Wide Input Range Controller

No R

, 5V Gate Drive, Current Mode Control

, 7V Gate Drive, Current Mode Control

SENSE

Wide Input Range Controller

SENSE

SOT-23 Boost Controller

Delivers Up to 5A, 550kHz Fixed Frequency, Current Mode

1.2MHz, SOT-23 Boost Converter

Inverting 1.2MHz, SOT-23 Converter

1A/2A 3MHz Synchronous Boost Converters

Positive-to-Negative DC/DC Controller

2-Phase Step-Up DC/DC Controller

Up to 34V Output, 2.6V ≤ V ≤ 16V, Miniature Design

IN

LT1931

Positive-to-Negative DC/DC Conversion, Miniature Design

LTC3401/LTC3402

LTC3704

LT3782

Up to 97% Efﬁciency, Very Small Solution, 0.5V ≤ V ≤ 5V

IN

No R , Current Mode Control, 50kHz to 1MHz

SENSE

6V ≤ V ≤ 40V; 4A Gate Drive, 150kHz to 500kHz

IN

18711fb

LT 0108 REV B • PRINTED IN USA

LinearTechnology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

36

●

(408) 432-1900 FAX: (408) 434-0507 www.linear.com


型号：	LTC1871EMS-1#PBF
厂家：	Linear
描述：	LTC1871-1 - Wide Input Range, No RSENSE Current Mode Boost, Flyback and SEPIC Controller; Package: MSOP; Pins: 10; Temperature Range: -40°C to 85°C 稳压器开关式稳压器或控制器电源电路电阻器开关式控制器光电二极管
文件：	总36页 (文件大小：518K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

LTC1871EMS-1#PBF [Linear]

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