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LTC3129EUD-1#PBF [Linear]

LTC3129-1 - 15V, 200mA Synchronous Buck-Boost DC/DC Converter with 1.3µA Quiescent Current; Package: QFN; Pins: 16; Temperature Range: -40°C to 85°C;
LTC3129EUD-1#PBF
型号： LTC3129EUD-1#PBF
厂家： Linear    Linear
描述：

LTC3129-1 - 15V, 200mA Synchronous Buck-Boost DC/DC Converter with 1.3µA Quiescent Current; Package: QFN; Pins: 16; Temperature Range: -40°C to 85°C

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LTC3129-1  
15V, 200mA Synchronous  
Buck-Boost DC/DC Converter  
with 1.3µA Quiescent Current  
FeaTures  
DescripTion  
n
Regulates V  
Above, Below or Equal to V  
The LTC®3129-1 is a high efficiency, 200mA buck-boost  
OUT  
IN  
n
Wide V Range: 2.42V to 15V, 1.92V to 15V After  
DC/DC converter with a wide V and V  
range. It  
IN  
IN  
OUT  
Start-Up (Bootstrapped)  
includes an accurate RUN pin threshold to allow predict-  
ableregulatorturn-onandamaximumpowerpointcontrol  
(MPPC)capabilitythatensuresmaximumpowerextraction  
fromnon-idealpowersourcessuchasphotovoltaicpanels.  
n
Fixed Output Voltage with Eight User-Selectable  
Settings from 2.5V to 15V  
n
n
n
n
n
n
n
n
n
n
n
n
200mA Output Current in Buck Mode  
Single Inductor  
The LTC3129-1 employs an ultralow noise, 1.2MHz PWM  
switchingarchitecturethatminimizessolutionfootprintby  
allowing the use of tiny, low profile inductors and ceramic  
capacitors. Built-in loop compensation and soft-start  
simplify the design. For high efficiency operation at light  
loads, automatic Burst Mode operation can be selected,  
reducing the quiescent current to just 1.3µA. To further  
reduce part count and improve light load efficiency, the  
LTC3129-1 includes an internal voltage divider to provide  
eight selectable fixed output voltages.  
1.3µA Quiescent Current  
Programmable Maximum Power Point Control  
1.2MHz Ultralow Noise PWM  
Current Mode Control  
Pin Selectable Burst Mode® Operation  
Up to 95% Efficiency  
Accurate RUN Pin Threshold  
Power Good Indicator  
10nA Shutdown Current  
Thermally Enhanced 3mm × 3mm QFN and  
16-Lead MSOP Packages  
Additionalfeaturesincludeapowergoodoutput, lessthan  
10nA of shutdown current and thermal shutdown.  
applicaTions  
The LTC3129-1 is available in thermally enhanced  
3mm × 3mm QFN and 16-lead MSOP packages. For an  
adjustable output voltage, see the functionally equivalent  
LTC3129.  
n
Industrial Wireless Sensor Nodes  
n
Post-Regulator for Harvested Energy  
n
Solar Panel Post-Regulator/Charger  
n
Intrinsically Safe Power Supplies  
Wireless Microphones  
L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks  
n
of Linear Technology Corporation. All other trademarks are the property of their respective owners.  
n
Avionics-Grade Wireless Headsets  
Typical applicaTion  
Efficiency and Power Loss vs Load  
22nF  
22nF  
10µH  
100  
90  
80  
70  
60  
50  
40  
30  
20  
10  
0
1000  
100  
10  
EFFICIENCY  
BST1 SW1  
SW2 BST2  
V
OUT  
5V AT  
100mA V < V  
V
IN  
V
V
OUT  
IN  
2.42V TO 15V  
IN  
OUT  
OUT  
10µF  
LTC3129-1  
200mA V > V  
IN  
RUN  
10µF  
POWER LOSS  
1
MPPC  
PGOOD  
V
CC  
AA OR AAA  
BATTERIES  
V
IN  
V
IN  
V
IN  
V
IN  
= 2.5V  
= 3.6V  
= 5V  
PWM  
VS1  
VS2  
VS3  
0.1  
V
CC  
= 15V  
V
= 5V  
OUT  
0.01  
2.2µF  
0.01  
0.1  
1
10  
100  
1000  
GND  
PGND  
OUTPUT CURRENT (mA)  
3129 TA01b  
31291 TA01a  
31291fa  
1
For more information www.linear.com/LTC3129-1  
LTC3129-1  
absoluTe MaxiMuM raTings  
(Notes 1, 8)  
V , V  
Voltages ..................................... –0.3V to 18V  
V , PWM, MPPC, VS1, VS2,  
IN OUT  
CC  
SW1 DC Voltage.............................. –0.3V to (V + 0.3V)  
VS3 Voltages ............................................... –0.3V to 6V  
PGOOD Sink Current..............................................15mA  
Operating Junction Temperature Range  
IN  
SW2 DC Voltage............................–0.3V to (V  
+ 0.3V)  
OUT  
SW1, SW2 Pulsed (<100ns) Voltage ..............–1V to 19V  
BST1 Voltage .....................(SW1 – 0.3V) to (SW1 + 6V)  
BST2 Voltage .....................(SW2 – 0.3V) to (SW2 + 6V)  
RUN, PGOOD Voltage................................. –0.3V to 18V  
(Notes 2, 5)............................................ –40°C to 125°C  
Storage Temperature Range .................. –65°C to 150°C  
MSE Lead Temperature (Soldering, 10 sec) ..........300°C  
pin conFiguraTion  
TOP VIEW  
TOP VIEW  
16 15 14 13  
1
2
3
4
5
6
7
8
V
16 V  
IN  
CC  
RUN  
MPPC  
GND  
VS3  
15 BST1  
14 SW1  
13 PGND  
12 SW2  
11 BST2  
BST1  
1
2
3
4
12  
V
OUT  
V
11 PGOOD  
IN  
17  
PGND  
17  
PGND  
V
CC  
PWM  
VS1  
10  
9
VS2  
RUN  
VS1  
10  
9
V
OUT  
PGOOD  
PWM  
5
6
7
8
MSE PACKAGE  
16-LEAD PLASTIC MSOP  
T
= 125°C, θ = 10°C/W, θ = 40°C/W (NOTE 6)  
JMAX  
JC JA  
UD PACKAGE  
16-LEAD (3mm × 3mm) PLASTIC QFN  
= 125°C, θ = 7.5°C/W, θ = 68°C/W (NOTE 6)  
EXPOSED PAD (PIN 17) IS PGND, MUST BE SOLDERED TO PCB  
T
JMAX  
JC  
JA  
EXPOSED PAD (PIN 17) IS PGND, MUST BE SOLDERED TO PCB  
orDer inForMaTion  
LEAD FREE FINISH  
LTC3129EUD-1#PBF  
LTC3129IUD-1#PBF  
LTC3129EMSE-1#PBF  
LTC3129IMSE-1#PBF  
TAPE AND REEL  
PART MARKING*  
LGDS  
PACKAGE DESCRIPTION  
16-Lead (3mm × 3mm) Plastic QFN  
TEMPERATURE RANGE  
–40°C to 125°C  
LTC3129EUD-1#TRPBF  
LTC3129IUD-1#TRPBF  
LGDS  
16-Lead (3mm × 3mm) Plastic QFN  
16-Lead Plastic MSOP  
–40°C to 125°C  
–40°C to 125°C  
–40°C to 125°C  
LTC3129EMSE-1#TRPBF 31291  
LTC3129IMSE-1#TRPBF 31291  
16-Lead Plastic MSOP  
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.  
Consult LTC Marketing for information on non-standard lead based finish parts.  
For more information on lead free part marking, go to: http://www.linear.com/leadfree/  
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/  
31291fa  
2
For more information www.linear.com/LTC3129-1  
LTC3129-1  
elecTrical characTerisTics The l denotes the specifications which apply over the specified operating  
junction temperature range, otherwise specifications are at TA = 25°C (Note 2). Unless otherwise noted, VIN = 12V, VOUT = 5V.  
PARAMETER  
Start-Up Voltage  
CONDITIONS  
MIN  
TYP  
MAX  
2.42  
15  
UNITS  
l
l
l
l
V
2.25  
V
V
IN  
Input Voltage Range  
V
V
> 2.42V (Back-Driven)  
> 2.42V (Back-Driven)  
1.92  
1.8  
80  
CC  
CC  
V
V
V
UVLO Threshold (Rising)  
UVLO Hysteresis  
1.9  
2.0  
V
IN  
100  
130  
mV  
IN  
l
l
l
l
l
l
l
l
Voltages  
VS1 = VS2 = VS3 = 0V  
VS1 = V , VS2 = VS3 = 0V  
2.425  
3.2175  
3.998  
4.875  
6.727  
7.995  
11.64  
14.50  
2.5  
3.3  
4.1  
5.0  
6.9  
8.2  
12  
2.575  
3.383  
4.203  
5.125  
7.073  
8.405  
12.40  
15.50  
V
V
V
V
V
V
V
V
OUT  
CC  
VS2 = V , VS1 = VS3 = 0V  
CC  
VS1 = VS2 = V , VS3 = 0V  
CC  
VS1 = VS2 = 0V, VS3 = V  
VS2 = 0V, VS1 = VS3 = V  
VS1 = 0V, VS2 = VS3 = V  
CC  
CC  
CC  
VS1 = VS2 = VS3 = V  
15.0  
CC  
Quiescent Current (V ) – Shutdown  
RUN = 0V, Including Switch Leakage  
Either V or V Below Their UVLO Threshold, or  
10  
100  
3
nA  
µA  
IN  
Quiescent Current (V ) UVLO  
1.9  
IN  
IN  
CC  
RUN Below the Threshold to Enable Switching  
Quiescent Current – Burst Mode Operation  
Measured on V , V > V  
1.3  
10  
2.0  
50  
µA  
nA  
IN OUT  
REG  
PWM = 0V, RUN = V  
IN  
N-Channel Switch Leakage on V and V  
SW1 = 0V, V = 15V  
IN  
IN  
OUT  
SW2 = 0V, V  
RUN = 0V  
= 15V  
OUT  
N-Channel Switch On-Resistance  
Inductor Average Current Limit  
V
= 4V  
0.75  
Ω
CC  
l
l
V
V
> UV Threshold (Note 4)  
< UV Threshold (Note 4)  
220  
80  
275  
130  
350  
200  
mA  
mA  
OUT  
OUT  
l
l
Inductor Peak Current Limit  
Maximum Boost Duty Cycle  
(Note 4)  
< V  
400  
85  
500  
89  
680  
95  
mA  
%
V
as Set by VS1-VS3. Percentage of  
REG  
OUT  
Period SW2 is Low in Boost Mode (Note 7)  
l
l
Minimum Duty Cycle  
V
> V as Set by VS1-VS3. Percentage of  
0
%
OUT  
REG  
Period SW1 is High in Buck Mode (Note 7)  
Switching Frequency  
SW1 and SW2 Minimum Low Time  
MPPC Voltage  
PWM = V  
(Note 3)  
1.0  
1.2  
90  
1.4  
MHz  
ns  
V
CC  
l
1.12  
1.175  
1
1.22  
10  
MPPC Input Current  
MPPC = 5V  
nA  
V
l
l
RUN Threshold to Enable V  
0.5  
1.16  
50  
0.9  
1.22  
80  
1.15  
1.28  
120  
10  
CC  
RUN Threshold to Enable Switching (Rising)  
RUN (Switching) Threshold Hysteresis  
RUN Input Current  
V
> 2.4V  
V
CC  
mV  
nA  
V
RUN = 15V  
1
l
l
VS1, VS2, VS3 Input High  
VS1, VS2, VS3 Input Low  
VS1, VS2, VS3 Input Current  
PWM Input High  
1.2  
1.6  
0.4  
10  
V
VS1, VS2, VS3 = V = 5V  
1
nA  
V
CC  
l
l
PWM Input Low  
0.5  
1
V
PWM Input Current  
PWM = 5V  
0.1  
3
µA  
ms  
V
Soft-Start Time  
l
V
V
Voltage  
V
> 4.85V  
3.4  
4.1  
4.7  
CC  
CC  
IN  
Dropout Voltage (V – V  
)
V
V
= 3.0V, Switching  
35  
0
60  
2
mV  
mV  
IN  
CC  
IN  
IN  
= 2.0V (V in UVLO)  
CC  
31291fa  
3
For more information www.linear.com/LTC3129-1  
LTC3129-1  
elecTrical characTerisTics The l denotes the specifications which apply over the specified operating  
junction temperature range, otherwise specifications are at TA = 25°C (Note 2). Unless otherwise noted, VIN = 12V, VOUT = 5V.  
PARAMETER  
CONDITIONS  
MIN  
TYP  
2.25  
60  
MAX  
UNITS  
V
l
V
V
V
V
V
V
V
V
V
V
V
UVLO Threshold (Rising)  
UVLO Hysteresis  
2.1  
2.42  
CC  
mV  
mA  
V
CC  
l
l
Current Limit  
V
CC  
= 0V  
4
20  
40  
5.5  
4
CC  
Back-Drive Voltage (Maximum)  
Input Current (Back-Driven)  
CC  
V
V
= 5.5V (Switching)  
2
mA  
µA  
V
CC  
CC  
Leakage to V if V >V  
= 5.5V, V = 1.8V, Measured on V  
–27  
1.15  
150  
10  
CC  
IN  
CC IN  
CC  
IN  
IN  
l
UV Threshold (Rising)  
0.95  
1.35  
100  
OUT  
OUT  
OUT  
OUT  
OUT  
UV Hysteresis  
mV  
nA  
µA  
µA  
%
Current – Shutdown  
Current – Sleep  
Current – Active  
RUN = 0V, V  
= 15V Including Switch Leakage  
OUT  
PWM = 0V, V  
≥ V  
V
/27  
OUT  
OUT  
REG  
PWM = V , V  
= 15V (Note 4)  
5
9
CC OUT  
PGOOD Threshold, Falling  
PGOOD Hysteresis  
PGOOD Voltage Low  
PGOOD Leakage  
Referenced to Programmed V  
Referenced to Programmed V  
Voltage  
Voltage  
–5.5  
–7.5  
2.5  
250  
1
–10  
OUT  
OUT  
%
I
= 1mA  
300  
50  
mV  
nA  
SINK  
PGOOD = 15V  
Note 1: Stresses beyond those listed under Absolute Maximum Ratings  
may cause permanent damage to the device. Exposure to any Absolute  
Maximum Rating condition for extended periods may affect device  
reliability and lifetime.  
Note 3: Specification is guaranteed by design and not 100% tested in  
production.  
Note 4: Current measurements are made when the output is not switching.  
Note 5: This IC includes overtemperature protection that is intended  
to protect the device during momentary overload conditions. Junction  
temperature will exceed 125°C when overtemperature protection is active.  
Continuous operation above the specified maximum operating junction  
temperature may result in device degradation or failure.  
Note 6: Failure to solder the exposed backside of the package to the PC  
board ground plane will result in a much higher thermal resistance.  
Note 7: Switch timing measurements are made in an open-loop test  
configuration. Timing in the application may vary somewhat from these  
values due to differences in the switch pin voltage during non-overlap  
durations when switch pin voltage is influenced by the magnitude and  
duration of the inductor current.  
Note 2: The LTC3129-1 is tested under pulsed load conditions such  
that T ≈ T . The LTC3129E-1 is guaranteed to meet specifications  
J
A
from 0°C to 85°C junction temperature. Specifications over the –40°C  
to 125°C operating junction temperature range are assured by design,  
characterization and correlation with statistical process controls. The  
LTC3129I-1 is guaranteed over the full –40°C to 125°C operating junction  
temperature range. The junction temperature (T , in °C) is calculated from  
J
the ambient temperature (T , in °C) and power dissipation (P , in watts)  
A
D
according to the formula:  
T = T + (P • θ ),  
J
A
D
JA  
where θ (in °C/W) is the package thermal impedance.  
JA  
Note that the maximum ambient temperature consistent with these  
specifications is determined by specific operating conditions in  
conjunction with board layout, the rated thermal package thermal  
resistance and other environmental factors.  
Note 8: Voltage transients on the switch pin(s) beyond the DC limits  
specified in the Absolute Maximum Ratings are non-disruptive to normal  
operation when using good layout practices as described elsewhere in the  
data sheet and Application Notes and as seen on the product demo board.  
31291fa  
4
For more information www.linear.com/LTC3129-1  
LTC3129-1  
TA = 25°C, unless otherwise noted.  
Typical perForMance characTerisTics  
Efficiency, VOUT = 2.5V  
Power Loss, VOUT = 2.5V  
Efficiency, VOUT = 3.3V  
100  
90  
80  
70  
60  
50  
40  
30  
20  
10  
0
1000  
100  
10  
100  
90  
80  
70  
60  
50  
40  
30  
20  
10  
0
BURST  
BURST  
PWM  
PWM  
V
1
= 2.5V  
= 3.6V  
= 5V  
= 10V  
= 15V  
V
V
V
V
V
= 2.5V  
= 3.6V  
= 5V  
= 10V  
= 15V  
= 2.5V  
= 3.6V  
= 5V  
= 10V  
= 15V  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
BURST  
V
V
V
V
IN  
IN  
IN  
IN  
0.1  
0.01  
0.01  
0.1  
1
10  
100  
1000  
0.01  
0.1  
1
10  
100  
1000  
0.01  
0.1  
1
10  
100  
1000  
OUTPUT CURRENT (mA)  
OUTPUT CURRENT (mA)  
OUTPUT CURRENT (mA)  
31291 G01  
31291 G02  
31291 G03  
Power Loss, VOUT = 3.3V  
Efficiency, VOUT = 4.1V  
Power Loss, VOUT = 4.1V  
1000  
100  
10  
100  
90  
1000  
100  
10  
BURST  
80  
PWM  
PWM  
70  
60  
50  
40  
30  
20  
10  
0
PWM  
1
1
0.1  
BURST  
V
V
V
V
V
= 2.5V  
= 3.6V  
= 5V  
= 10V  
= 15V  
V
= 2.5V  
= 3.6V  
= 5V  
= 10V  
= 15V  
V
V
V
V
V
= 2.5V  
= 3.6V  
= 5V  
= 10V  
= 15V  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
BURST  
V
V
V
V
0.1  
0.01  
0.01  
0.01  
0.1  
1
10  
100  
1000  
0.01  
0.1  
1
10  
100  
1000  
0.01  
0.1  
1
10  
100  
1000  
OUTPUT CURRENT (mA)  
OUTPUT CURRENT (mA)  
OUTPUT CURRENT (mA)  
31291 G04  
31291 G04a  
31291 G04b  
Efficiency, VOUT = 5V  
Power Loss, VOUT = 5V  
Efficiency, VOUT = 6.9V  
100  
90  
80  
70  
60  
50  
40  
30  
20  
10  
0
1000  
100  
10  
100  
90  
BURST  
BURST  
PWM  
80  
70  
60  
50  
40  
30  
20  
10  
0
PWM  
PWM  
1
V
V
V
V
V
= 2.5V  
= 3.6V  
= 5V  
= 10V  
= 15V  
V
V
V
V
V
= 2.5V  
= 3.6V  
= 5V  
= 10V  
= 15V  
V
IN  
V
IN  
V
IN  
V
IN  
V
IN  
= 2.5V  
= 3.6V  
= 5V  
= 10V  
= 15V  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
BURST  
0.1  
0.01  
0.01  
0.1  
1
10  
100  
1000  
0.01  
0.1  
1
10  
100  
1000  
0.01  
0.1  
1
10  
100  
1000  
OUTPUT CURRENT (mA)  
OUTPUT CURRENT (mA)  
OUTPUT CURRENT (mA)  
31291 G05  
31291 G06  
31291 G06a  
31291fa  
5
For more information www.linear.com/LTC3129-1  
LTC3129-1  
TA = 25°C, unless otherwise noted.  
Typical perForMance characTerisTics  
Power Loss, VOUT = 6.9V  
Efficiency, VOUT = 8.2V  
Power Loss, VOUT = 8.2V  
1000  
100  
10  
100  
90  
1000  
BURST  
80  
100  
10  
PWM  
PWM  
70  
60  
50  
40  
30  
20  
10  
0
PWM  
1
0.1  
1
0.1  
BURST  
BURST  
V
V
V
V
V
= 2.5V  
= 3.6V  
= 5V  
= 10V  
= 15V  
V
V
V
V
V
= 2.5V  
= 3.6V  
= 5V  
= 10V  
= 15V  
V
V
V
V
V
= 2.5V  
= 3.6V  
= 5V  
= 10V  
= 15V  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
0.01  
0.01  
0.01  
0.1  
1
10  
100  
1000  
0.01  
0.1  
1
10  
100  
1000  
0.01  
0.1  
1
10  
100  
1000  
OUTPUT CURRENT (mA)  
OUTPUT CURRENT (mA)  
OUTPUT CURRENT (mA)  
31291 G06b  
31291 G06c  
31291 G06d  
Efficiency, VOUT = 12V  
Power Loss, VOUT = 12V  
Efficiency, VOUT = 15V  
100  
90  
80  
70  
60  
50  
40  
30  
20  
10  
0
1000  
100  
10  
100  
90  
80  
70  
60  
50  
40  
30  
20  
10  
0
BURST  
BURST  
PWM  
PWM  
PWM  
BURST  
1
V
V
V
V
V
= 2.5V  
= 3.6V  
= 5V  
= 10V  
= 15V  
V
V
V
V
V
= 2.5V  
= 3.6V  
= 5V  
= 10V  
= 15V  
V
IN  
V
IN  
V
IN  
V
IN  
V
IN  
= 2.5V  
= 3.6V  
= 5V  
= 10V  
= 15V  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
0.1  
0.01  
0.01  
0.1  
1
10  
100  
1000  
0.01  
0.1  
1
10  
100  
1000  
0.01  
0.1  
1
10  
100  
1000  
OUTPUT CURRENT (mA)  
OUTPUT CURRENT (mA)  
OUTPUT CURRENT (mA)  
31291 G07  
31291 G08  
31291 G09  
Maximum Output Current  
vs VIN and VOUT  
No Load Input Current  
vs VIN and VOUT (PWM = 0V)  
Power Loss, VOUT = 15V  
250  
200  
150  
100  
50  
5
4
3
2
1
0
1000  
100  
10  
V
V
V
V
V
V
V
V
= 2.5V  
OUT  
OUT  
OUT  
OUT  
OUT  
OUT  
OUT  
OUT  
= 3.3V  
= 4.1V  
= 5V  
PWM  
= 6.9V  
= 8.2V  
= 12V  
= 15V  
V
V
V
V
V
V
V
V
= 2.5V  
OUT  
OUT  
OUT  
OUT  
OUT  
OUT  
OUT  
OUT  
= 3.3V  
= 4.1V  
= 5V  
BURST  
1
V
V
V
V
V
= 2.5V  
= 3.6V  
= 5V  
= 10V  
= 15V  
IN  
IN  
IN  
IN  
IN  
= 6.9V  
= 8.2V  
= 12V  
= 15V  
0.1  
0.01  
0
2
3
4
5
6
7
8
9
10 11 12 13 14 15  
2
4
6
8
V
10  
(V)  
12  
14  
16  
0.01  
0.1  
1
10  
100  
1000  
V
(V)  
IN  
OUTPUT CURRENT (mA)  
IN  
31291 G11  
31291 G12  
31291 G10  
31291fa  
6
For more information www.linear.com/LTC3129-1  
LTC3129-1  
TA = 25°C, unless otherwise noted.  
Typical perForMance characTerisTics  
Burst Mode Threshold  
Output Voltage vs Temperature  
(Normalized to 25°C)  
vs VIN and VOUT  
Switch RDS(ON) vs Temperature  
1.3  
1.2  
1.1  
1.0  
0.9  
0.8  
0.7  
0.6  
0.5  
0.4  
1.0  
0.5  
80  
70  
60  
50  
40  
30  
V
CC  
V
CC  
V
CC  
V
CC  
= 2.5V  
= 3V  
= 4V  
= 5V  
0
V
V
V
V
V
V
V
V
= 2.5V  
= 3.3V  
= 4.1V  
= 5V  
OUT  
OUT  
OUT  
OUT  
OUT  
OUT  
OUT  
OUT  
–0.5  
20  
10  
= 6.9V  
= 8.2V  
= 12V  
= 15V  
–1.0  
0
–45 –20  
5
30  
55  
80 105 130  
–45 –20  
5
30  
55  
80 105 130  
2
4
6
8
10  
(V)  
12  
14  
16  
TEMPERATURE (°C)  
TEMPERATURE (°C)  
V
IN  
31291 G14  
31291 G15  
3129 G13  
Maximum Output vs Temperature  
(Normalized to 25°C)  
Accurate RUN Threshold  
vs Temperature (Normalized to 25°C)  
Average Input Current Limit  
vs MPPC Voltage  
2
1
100  
90  
80  
70  
60  
50  
40  
30  
20  
10  
0
15  
10  
5
0
0
–5  
–10  
–15  
–1  
–2  
–45 –20  
5
30  
55  
80 105 130  
1.13 1.135 1.14 1.145 1.15 1.155 1.16 1.165 1.17  
–45 –20  
5
30  
55  
80 105 130  
TEMPERATURE (°C)  
MPPC PIN VOLTAGE (V)  
TEMPERATURE (°C)  
31291 G17  
31291 G18  
31291 G19  
VCC Dropout Voltage vs Temperature  
(PWM Mode, Switching)  
VCC Dropout Voltage vs VIN  
(PWM Mode, Switching)  
Fixed Frequency PWM  
Waveforms  
60  
50  
40  
30  
20  
10  
0
60  
SW2  
5V/DIV  
50  
40  
30  
20  
10  
0
SW1  
5V/DIV  
I
L
200mA/DIV  
31291 G22  
500ns/DIV  
L = 10µH  
V
V
= 7V  
IN  
= 5V  
OUT  
OUT  
–45 –20  
5
30  
55  
80 105 130  
2
2.25 2.5 2.75  
3
3.25 3.5 3.75  
4
I
= 200mA  
TEMPERATURE (°C)  
V
(V)  
IN  
31291 G20  
31291 G21  
31291fa  
7
For more information www.linear.com/LTC3129-1  
LTC3129-1  
TA = 25°C, unless otherwise noted.  
Typical perForMance characTerisTics  
Burst Mode Waveforms  
Fixed Frequency Ripple on VOUT  
Burst Mode Ripple on VOUT  
SW1  
5V/DIV  
V
OUT  
V
OUT  
100mV/DIV  
20mV/DIV  
SW2  
5V/DIV  
I
L
200mA/DIV  
I
I
L
L
100mA/DIV  
200mA/DIV  
31291 G25  
31291 G24  
31291 G23  
100µs/DIV  
50µs/DIV  
L = 10µH  
200ns/DIV  
L = 10µH  
L = 10µH  
V
V
I
= 7V  
V
V
I
= 7V  
V
IN  
V
OUT  
I
= 7V  
IN  
OUT  
IN  
OUT  
= 5V  
= 5mA  
= 22µF  
= 5V  
= 5V  
= 5mA  
= 22µF  
= 200mA  
= 10µF  
OUT  
OUT  
OUT  
OUT  
C
C
C
OUT  
OUT  
Step Load Transient Response in  
Fixed Frequency  
Step Load Transient Response in  
Burst Mode Operation  
Start-Up Waveforms  
V
OUT  
5V/DIV  
V
V
OUT  
100mV/DIV  
OUT  
100mV/DIV  
V
CC  
5V/DIV  
RUN  
5V/DIV  
I
I
VOUT  
VOUT  
I
100mA/DIV  
VIN  
100mA/DIV  
200mA/DIV  
31291 G26  
31291 G27  
31291 G28  
1ms/DIV  
500µs/DIV  
500µs/DIV  
V
V
I
= 7V  
L = 10µH  
L = 10µH  
IN  
OUT  
= 5V  
V
V
C
I
= 7V  
V
V
C
I
= 7V  
IN  
OUT  
OUT  
IN  
OUT  
OUT  
= 50mA  
= 22µA  
= 5V  
= 10µF  
= 5V  
= 22µF  
OUT  
OUT  
C
= 50mA to 150mA STEP  
= 5mA to 125mA STEP  
OUT  
OUT  
PGOOD Response to a Drop  
On VOUT  
MPPC Response to a Step Load  
V
OUT  
2V/DIV  
PGOOD  
2V/DIV  
V
IN  
2V/DIV  
V
OUT  
2V/DIV  
I
VOUT  
100mA/DIV  
31291 G29  
31291 G30  
1ms/DIV  
2ms/DIV  
SET TO 3.5V  
V
= 5V  
V
V
C
V
= 5V  
OC  
MPPC  
= 22µF, R = 10Ω,  
OUT  
OUT  
OUT  
IN  
IN  
IN  
= 5V, C  
= 22µF  
OUT  
I
= 25mA to 125mA STEP  
31291fa  
8
For more information www.linear.com/LTC3129-1  
LTC3129-1  
pin FuncTions (QFN/MSOP)  
BST1 (Pin 1/Pin 15): Boot-Strapped Floating Supply for  
High Side NMOS Gate Drive. Connect to SW1 through a  
22nF capacitor, as close to the part as possible. The value  
is not critical. Any value from 4.7nF to 47nF may be used.  
VS3 (Pin 7/Pin 5): Output Voltage Select Pin. Connect this  
pin to ground or V to program the output voltage (see  
CC  
Table 1). This pin should not float or go below ground.  
If this pin is externally driven above V , a 1M resistor  
CC  
should be added in series.  
V (Pin2/Pin16):InputVoltagefortheConverter.Connect  
IN  
a minimum of 4.7µF ceramic decoupling capacitor from  
VS2 (Pin 8/Pin 6): Output Voltage Select Pin. Connect this  
thispintothegroundplane, asclosetothepinaspossible.  
pin to ground or V to program the output voltage (see  
CC  
Table 1). This pin should not float or go below ground.  
V
CC  
(Pin 3/Pin 1): Output Voltage of the Internal Voltage  
Regulator. This is the supply pin for the internal circuitry.  
Bypass this output with a minimum of 2.2µF ceramic ca-  
pacitor close to the pin. This pin may be back-driven by  
an external supply, up to a maximum of 5.5V.  
VS1 (Pin 9/Pin 7): Output Voltage Select Pin. Connect this  
pin to ground or V to program the output voltage (see  
CC  
Table 1). This pin should not float or go below ground.  
PWM (Pin 10/Pin 8): Mode Select Pin.  
RUN (Pin 4/Pin 2): Input to the Run Comparator. Pull  
PWM = Low (ground): Enables automatic Burst Mode  
operation.  
this pin above 1.1V to enable the V regulator and above  
CC  
1.28V to enable the converter. Connecting this pin to a  
PWM = High (tie to V ): Fixed frequency PWM  
resistor divider from V to ground allows programming a  
CC  
IN  
operation.  
V startthresholdhigherthanthe1.8V(typical)V UVLO  
IN  
IN  
threshold. In this case, the typical V turn-on threshold is  
IN  
This pin should not be allowed to float. It has internal 5M  
pull-down resistor.  
determinedbyV =1.22V•[1+(R3/PinR4)](seeFigure2).  
IN  
MPPC (Pin 5/Pin 3): Maximum Power Point Control  
PGOOD (Pin 11/Pin 9): Open drain output that pulls to  
ground when FB drops too far below its regulated volt-  
age. Connect a pull-up resistor from this pin to a positive  
supply. This pin can sink up to the absolute maximum  
rating of 15mA when low. Note that this pin is forced low  
Programming Pin. Connect this pin to a resistor divider  
from V to ground to enable the MPPC functionality.  
IN  
If the V  
load is greater than what the power source  
OUT  
can provide, the MPPC will reduce the inductor current  
to regulate V to a voltage determined by: V = 1.175V  
IN  
IN  
in shutdown or V UVLO.  
CC  
• [1 + (R5/R6)] (see Figure 3). By setting the V regula-  
IN  
V
(Pin 12/Pin 10): Output voltage of the converter, set  
OUT  
tion voltage appropriately, maximum power transfer from  
the limited source is assured. Note this pin is very noise  
sensitive,thereforeminimizetracelengthandstraycapaci-  
tance. Please refer to the Applications Information section  
for more detail on programming the MPPC for different  
by the VS1-VS3 programming pins according to Table 1.  
Connectaminimumvalueof4.7µFceramiccapacitorfrom  
thispintothegroundplane, asclosetothepinaspossible.  
BST2 (Pin 13/Pin 11): Boot-Strapped Floating Supply for  
High Side NMOS Gate Drive. Connect to SW2 through a  
22nF capacitor, as close to the part as possible. The value  
is not critical. Any value from 4.7nF to 47nF may be used  
sources. If this function is not needed, tie the pin to V .  
CC  
GND (Pin 6/Pin 4): Signal Ground. Provide a short direct  
PCB path between GND and the ground plane where the  
exposed pad is soldered.  
SW2 (Pin 14/Pin 12): Switch Pin. Connect to one side of  
the inductor. Keep PCB trace lengths as short and wide  
as possible to reduce EMI.  
31291fa  
9
For more information www.linear.com/LTC3129-1  
LTC3129-1  
pin FuncTions (QFN/MSOP)  
PGND(Pin15/Pin13,ExposedPadPin17/Pin17):Power  
Ground. Provide a short direct PCB path between PGND  
and the ground plane. The exposed pad must also be  
soldered to the PCB ground plane. It serves as a power  
ground connection, and as a means of conducting heat  
away from the die.  
Table 1. VOUT Program Settings  
VS3 PIN  
VS2 PIN  
VS1 PIN  
V
OUT  
0
0
0
0
0
0
0
2.5V  
3.3V  
4.1V  
5V  
V
CC  
V
CC  
V
CC  
0
V
CC  
V
0
0
0
6.9V  
8.2V  
12V  
15V  
SW1 (Pin 16/Pin 14): Switch Pin. Connect to one side of  
the inductor. Keep PCB trace lengths as short and wide  
as possible to reduce EMI.  
CC  
V
V
CC  
CC  
V
V
V
CC  
CC  
0
CC  
V
V
CC  
CC  
block DiagraM  
BST1  
SW1  
SW2  
BST2  
V
IN  
V
IN  
V
CC  
V
REF  
LDO  
V
CC_GD  
V
OUT  
START  
V
OUT  
A
B
DRIVER  
DRIVER  
V
CC  
V
4.1V  
CC  
V
CC  
I
I
SENSE  
SENSE  
D
DRIVER  
DRIVER  
1.175V  
V
START  
RUN  
V
REF  
REF  
V
REF_GD  
VS1  
VS2  
VS3  
C
V
OUT  
+
–
SELECT  
INPUTS  
START  
DRV_B  
DRV_C  
0.9V  
DRV_A  
DRV_D  
+
–
SD  
I
SENSE  
+
–
–
+
UV  
1.22V  
IN  
500mA  
I
V
1.1V  
LIM  
1.175V  
LOGIC  
ENABLE  
I
SENSE  
–
+
UVLO  
FB  
–
V
C
I
–
+
I
ZERO  
SENSE  
–
+
+
+
–
PWM  
1.175V  
20mA  
THERMAL  
SHUTDOWN  
RESET  
SOFT-START  
OSC  
MPPC  
PWM  
+
–
1.175V  
PGOOD  
–
+
–
+
600mV  
CLAMP  
5M  
–
+
SLEEP  
–7.5%  
100mV  
GND  
PGND  
31291 BD  
31291fa  
10  
For more information www.linear.com/LTC3129-1  
LTC3129-1  
operaTion  
INTRODUCTION  
voltage range, 1.3µA Burst Mode current and program-  
mable RUN and MPPC pins, the LTC3129-1 is well suited  
for many diverse applications.  
The LTC3129-1 is a 1.3µA quiescent current, monolithic,  
currentmode,buck-boostDC/DCconverterthatcanoperate  
overawideinputvoltagerangeof1.92Vto15Vandprovide  
up to 200mA to the load. Eight fixed, user-programmable  
output voltages can be selected using the three digital  
PWM MODE OPERATION  
If the PWM pin is high or if the load current on the con-  
verter is high enough to command PWM mode operation  
with PWM low, the LTC3129-1 operates in a fixed 1.2MHz  
PWM mode using an internally compensated average  
current mode control loop. PWM mode minimizes output  
voltage ripple and yields a low noise switching frequency  
spectrum. A proprietary switching algorithm provides  
seamless transitions between operating modes and  
eliminates discontinuities in the average inductor cur-  
rent, inductor ripple current and loop transfer function  
throughout all modes of operation. These advantages  
result in increased efficiency, improved loop stability and  
loweroutputvoltagerippleincomparisontothetraditional  
buck-boost converter.  
programmingpins.Internal,lowR  
N-channelpower  
DS(ON)  
switches reduce solution complexity and maximize effi-  
ciency. A proprietary switch control algorithm allows the  
buck-boostconvertertomaintainoutputvoltageregulation  
with input voltages that are above, below or equal to the  
output voltage. Transitions between the step-up or step-  
downoperatingmodesareseamlessandfreeoftransients  
and sub-harmonic switching, making this product ideal  
for noise sensitive applications. The LTC3129-1 operates  
at a fixed nominal switching frequency of 1.2MHz, which  
providesanidealtrade-offbetweensmallsolutionsizeand  
high efficiency. Current mode control provides inherent  
input line voltage rejection, simplified compensation and  
rapid response to load transients.  
Figure1showsthetopologyoftheLTC3129-1powerstage  
whichiscomprisedoffourN-channelDMOSswitchesand  
their associated gate drivers. In PWM mode operation  
both switch pins transition on every cycle independent of  
the input and output voltages. In response to the internal  
control loop command, an internal pulse width modulator  
generates the appropriate switch duty cycle to maintain  
regulation of the output voltage.  
Burst Mode capability is also included in the LTC3129-1  
and is user-selected via the PWM input pin. In Burst Mode  
operation,theLTC3129-1providesexceptionalefficiencyat  
light output loading conditions by operating the converter  
only when necessary to maintain voltage regulation. The  
BurstModequiescentcurrentisamiserly1.3µA. Athigher  
loads, the LTC3129-1 automatically switches to fixed fre-  
quencyPWM modewhenBurstModeoperationisselected.  
(Please refer to the Typical Performance Characteristic  
curves for the mode transition point at different input and  
output voltages). If the application requires extremely low  
noise, continuous PWM operation can also be selected  
via the PWM pin.  
C
BST1  
C
BST2  
L
BST1  
V
A
SW1  
SW2  
D
V
OUT  
BST2  
IN  
V
V
CC  
CC  
A MPPC (maximum power point control) function is also  
provided that allows the input voltage to the converter to  
be servo’d to a programmable point for maximum power  
when operating from various non-ideal power sources  
such as photovoltaic cells. The LTC3129-1 also features  
an accurate RUN comparator threshold with hysteresis,  
allowing the buck-boost DC/DC converter to turn on and  
V
CC  
V
CC  
B
C
PGND  
PGND  
LTC3129-1  
31291 F01  
Figure 1. Power Stage Schematic  
off at user-selected V voltage thresholds. With a wide  
IN  
31291fa  
11  
For more information www.linear.com/LTC3129-1  
LTC3129-1  
operaTion  
When stepping down from a high input voltage to a lower  
output voltage, the converter operates in buck mode and  
switch D remains on for the entire switching cycle except  
fortheminimumswitchlowduration(typically90ns).Dur-  
ing the switch low duration, switch C is turned on which  
ramps, and the comparator outputs are used to control  
the duty cycle of the switch pins on a cycle-by-cycle basis.  
The voltage error amplifier monitors the output voltage,  
V
OUT  
throughtheinternalvoltagedividerandmakesadjust-  
ments to the current command as necessary to maintain  
regulation. The voltage error amplifier therefore controls  
the outer voltage regulation loop. The average current  
amplifier makes adjustments to the inductor current as  
forces SW2 low and charges the flying capacitor, C  
.
BST2  
This ensures that the switch D gate driver power supply  
rail on BST2 is maintained. The duty cycle of switches A  
and B are adjusted to maintain output voltage regulation  
in buck mode.  
directed by the voltage error amplifier output via V and is  
C
commonly referred to as the inner current loop amplifier.  
If the input voltage is lower than the output voltage, the  
converter operates in boost mode. Switch A remains on  
for the entire switching cycle except for the minimum  
switch low duration (typically 90ns). During the switch  
low duration, switch B is turned on which forces SW1  
The average current mode control technique is similar to  
peak current mode control except that the average current  
amplifier, by virtue of its configuration as an integrator,  
controls average current instead of the peak current. This  
difference eliminates the peak to average current error  
inherent to peak current mode control, while maintaining  
most of the advantages inherent to peak current mode  
control.  
low and charges the flying capacitor, C  
. This ensures  
BST1  
that the switch A gate driver power supply rail on BST1 is  
maintained.ThedutycycleofswitchesCandDareadjusted  
to maintain output voltage regulation in boost mode.  
Average current mode control requires appropriate com-  
pensation for the inner current loop, unlike peak current  
mode control. The compensation network must have high  
DC gain to minimize errors between the actual and com-  
manded average current level, high bandwidth to quickly  
change the commanded current level following transient  
load steps and a controlled mid-band gain to provide a  
form of slope compensation unique to average current  
mode control. The compensation components required  
to ensure proper operation have been carefully selected  
and are integrated within the LTC3129-1.  
Oscillator  
The LTC3129-1 operates from an internal oscillator with a  
nominalfixedfrequencyof1.2MHz. ThisallowstheDC/DC  
converterefficiencytobemaximizedwhilestillusingsmall  
external components.  
Current Mode Control  
The LTC3129-1 utilizes average current mode control for  
the pulse width modulator. Current mode control, both  
average and the better known peak method, enjoy some  
benefits compared to other control methods including:  
simplified loop compensation, rapid response to load  
transients and inherent line voltage rejection.  
Inductor Current Sense and Maximum Output Current  
As part of the current control loop required for current  
mode control, the LTC3129-1 includes a pair of current  
sensing circuits that measure the buck-boost converter  
inductor current.  
Referring to the Block Diagram, a high gain, internally  
compensated transconductance amplifier monitors V  
OUT  
throughaninternalvoltagedivider.Theerroramplifierout-  
put is used by the current mode control loop to command  
the appropriate inductor current level. The inverting input  
of the internally compensated average current amplifier is  
connected to the inductor current sense circuit. The aver-  
agecurrentamplifier’soutputiscomparedtotheoscillator  
Thevoltageerroramplifieroutput,V ,isinternallyclamped  
C
to a nominal level of 0.6V. Since the average inductor  
current is proportional to V , the 0.6V clamp level sets  
C
the maximum average inductor current that can be pro-  
grammed by the inner current loop. Taking into account  
the current sense amplifier’s gain, the maximum average  
31291fa  
12  
For more information www.linear.com/LTC3129-1  
LTC3129-1  
operaTion  
inductor current is approximately 275mA (typical). In  
Overload Current Limit and I  
Comparator  
ZERO  
buck mode, the output current is approximately equal to  
The internal current sense waveform is also used by the  
peakoverloadcurrent(I )andzerocurrent(I )com-  
the inductor current I .  
L
PEAK  
ZERO  
I
≈ I • 0.89  
parators. The I  
current comparator monitors I  
PEAK SENSE  
OUT(BUCK)  
L
and turns off switch A if the inductor current level exceeds  
its maximum internal threshold, which is approximately  
500mA. An inductor current level of this magnitude will  
occur during a fault, such as an output short-circuit, or  
during large load or input voltage transients.  
The 90ns SW1/SW2 forced low time on each switching  
cycle briefly disconnects the inductor from V and V  
resulting in about 11% less output current in either buck  
orBoostmodeforagiveninductorcurrent.Inboostmode,  
the output current is related to average inductor current  
and duty cycle by:  
OUT  
IN  
The LTC3129-1 features near discontinuous inductor  
current operation at light output loads by virtue of the  
I
≈ I • (1 – D) • Efficiency  
L
OUT(BOOST)  
I
comparator circuit. By limiting the reverse current  
ZERO  
where D is the converter duty cycle.  
magnitude in PWM mode, a balance between low noise  
operationandimprovedefficiencyatlightloadsisachieved.  
TheI  
Since the output current in boost mode is reduced by the  
duty cycle (D), the output current rating in buck mode is  
always greater than in boost mode. Also, because boost  
mode operation requires a higher inductor current for a  
comparatorthresholdissetnearthezerocurrent  
ZERO  
level in PWM mode, and as a result, the reverse current  
magnitude will be a function of inductance value and out-  
put voltage due to the comparator's propagation delay. In  
general, higher output voltages and lower inductor values  
will result in increased reverse current magnitude.  
givenoutputcurrentcomparedtobuckmode,theefficiency  
2
in boost mode will be lower due to higher I • R  
L
DS(ON)  
losses in the power switches. This will further reduce the  
outputcurrentcapabilityinboostmode.Ineitheroperating  
mode, however, the inductor peak-to-peak ripple current  
does not play a major role in determining the output cur-  
rent capability, unlike peak current mode control.  
In automatic Burst Mode operation (PWM pin low), the  
I
comparator threshold is increased so that reverse  
ZERO  
inductor current does not normally occur. This maximizes  
efficiency at very light loads.  
With peak current mode control, the maximum output  
current capability is reduced by the magnitude of inductor  
ripplecurrentbecausethepeakinductorcurrentlevelisthe  
control variable, but the average inductor current is what  
determines the output current. The LTC3129-1 measures  
and controls average inductor current, and therefore, the  
inductor ripple current magnitude has little effect on the  
maximum current capability in contrast to an equivalent  
peak current mode converter. Under most conditions in  
buck mode, the LTC3129-1 is capable of providing a mini-  
mum of 200mA to the load. In boost mode, as described  
previously, the output current capability is related to the  
Burst Mode OPERATION  
When the PWM pin is held low, the LTC3129-1 is con-  
figured for automatic Burst Mode operation. As a result,  
the buck-boost DC/DC converter will operate with normal  
continuous PWM switching above a predetermined mini-  
mumoutputloadandwillautomaticallytransitiontopower  
saving Burst Mode operation below this output load level.  
Note that if the PWM pin is low, reverse inductor current is  
not allowed at any load. Refer to the Typical Performance  
Characteristics section of this data sheet to determine the  
Burst Mode transition threshold for various combinations  
boost ratio or duty cycle (D). For example, for a 3.6V V  
IN  
of V and V . If PWM is low, at light output loads, the  
IN  
OUT  
to 5V output application, the LTC3129-1 can provide up  
to 150mA to the load. Refer to the Typical Performance  
Characteristics section for more detail on output current  
capability.  
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LTC3129-1 will go into a standby or sleep state when the  
output voltage achieves its nominal regulation level. The  
sleep state halts PWM switching and powers down all  
nonessentialfunctionsoftheIC, significantlyreducingthe  
quiescent current of the LTC3129-1 to just 1.3µA typical.  
This greatly improves overall power conversion efficiency  
when the output load is light. Since the converter is not  
operating in sleep, the output voltage will slowly decay  
at a rate determined by the output load resistance and  
the output capacitor value. When the output voltage has  
decayed by a small amount, the LTC3129-1 will wake and  
resume normal PWM switching operation until the volt-  
protectiontosafeguardagainstaccidentalshort-circuiting  
of the V rail.  
CC  
Undervoltage Lockout (UVLO)  
Therearetwoundervoltagelockout(UVLO)circuitswithin  
theLTC3129-1thatinhibitswitching;onethatmonitorsV  
IN  
and another that monitors V . Either UVLO will disable  
CC  
operation of the internal power switches and keep other  
IC functions in a reset state if either V or V are below  
IN  
CC  
their respective UVLO thresholds.  
The V UVLO comparator has a falling voltage threshold  
IN  
of 1.8V (typical). If V falls below this level, IC operation  
IN  
age on V  
is restored to the previous level. If the load  
OUT  
is disabled until V rises above 1.9V (typical), as long as  
IN  
is very light, the LTC3129-1 may only need to switch for  
a few cycles to restore V and may sleep for extended  
the V voltage is above its UVLO threshold.  
CC  
OUT  
periods of time, significantly improving efficiency. If the  
load is suddenly increased above the burst transition  
threshold, the part will automatically resume continuous  
PWM operation until the load is once again reduced.  
The V UVLO has a falling voltage threshold of 2.19V  
CC  
(typical). If the V voltage falls below this threshold, IC  
CC  
operation is disabled until V rises above 2.25V (typical)  
CC  
as long as V is above its nominal UVLO threshold level.  
IN  
Note that Burst Mode operation is inhibited until soft-start  
Depending on the particular application, either of these  
UVLO thresholds could be the limiting factor affecting the  
minimuminputvoltagerequiredforoperation.Becausethe  
is done, the MPPC pin is greater than 1.175V and V  
OUT  
has reached regulation.  
V
regulator uses V for its power input, the minimum  
CC  
IN  
Soft-Start  
input voltage required for operation is determined by the  
V
minimum voltage, as input voltage (V ) will always  
The LTC3129-1 soft-start circuit minimizes input current  
transientsandoutputvoltageovershootoninitialpowerup.  
The required timing components for soft-start are internal  
to the LTC3129-1 and produce a nominal soft-start dura-  
tion of approximately 3ms. The internal soft-start circuit  
CC  
IN  
be higher than V in the normal (non-bootstrapped)  
CC  
configuration. Therefore, the minimum V for the part  
to start up is 2.25V (typical).  
IN  
In applications where V is bootstrapped (powered  
CC  
slowly ramps the error amplifier output, V . In doing so,  
C
through a Schottky diode by either V  
or an auxiliary  
OUT  
the current command of the IC is also slowly increased,  
starting from zero. It is unaffected by output loading or  
output capacitor value. Soft-start is reset by the UVLO on  
power rail), the minimum input voltage for operation will  
be limited only by the V UVLO threshold (1.8V typical).  
IN  
Please note that if the bootstrap voltage is derived from  
both V and V , the RUN pin and thermal shutdown.  
IN  
CC  
the LTC3129-1 V  
and not an independent power rail,  
OUT  
thentheminimuminputvoltagerequiredforinitialstart-up  
V
Regulator  
CC  
is still 2.25V (typical).  
Aninternallowdropoutregulator(LDO)generatesanomi-  
nal 4.1V V rail from V . The V rail powers the internal  
Note that if either V or V are below their UVLO  
IN  
CC  
CC  
IN  
CC  
thresholds, or if RUN is below its accurate threshold of  
controlcircuitryandthegatedriversoftheLTC3129-1.The  
regulator is disabled in shutdown to reduce quiescent  
1.22V (typical), then the LTC3129-1 will remain in a soft  
V
CC  
shutdown state, where the V quiescent current will be  
IN  
current and is enabled by raising the RUN pin above its  
only 1.9µA typical.  
logic threshold. The V regulator includes current-limit  
CC  
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V
Undervoltage  
OUT  
LTC3129-1  
1.22V  
V
ACCURATE THRESHOLD  
IN  
There is also an undervoltage comparator that monitors  
the output voltage. Until V reaches 1.15V (typical), the  
–
+
ENABLE SWITCHING  
OUT  
R3  
RUN  
average current limit is reduced by a factor of two. This  
reduces power dissipation in the device in the event of a  
shorted output. In addition, N-channel switch D, which  
+
–
R4  
ENABLE LDO AND  
CONTROL CIRCUITS  
0.9V  
feeds V , will be disabled until V  
exceeds 1.15V.  
OUT  
OUT  
LOGIC THRESHOLD  
RUN Pin Comparator  
31291 F02  
Figure 2. Accurate RUN Pin Comparator  
In addition to serving as a logic level input to enable cer-  
tain functions of the IC, the RUN pin includes an accurate  
internal comparator that allows it to be used to set custom  
rising and falling ON/OFF thresholds with the addition of  
an optional external resistor divider. When RUN is driven  
Note that once RUN is above 0.9V typical, the quiescent  
input current on V (or V if back-driven) will increase to  
IN  
CC  
about 1.9µA typical until the V and V UVLO thresholds  
IN  
CC  
are satisfied.  
above its logic threshold (0.9V typical), the V regulator  
CC  
is enabled, which provides power to the internal control  
circuitry of the IC. If the voltage on RUN is increased  
further so that it exceeds the RUN comparator’s accurate  
analog threshold (1.22V typical), all functions of the buck-  
boost converter will be enabled and a start-up sequence  
TheconverterisenabledwhenthevoltageonRUNexceeds  
1.22V (nominal). Therefore, the turn on voltage threshold  
on V is given by:  
IN  
V
= 1.22V • (1 + R3/R4)  
IN(TURN-ON)  
will ensue (assuming the V and V UVLO thresholds  
IN  
CC  
The RUN comparator includes a built-in hysteresis of  
approximately 80mV, so that the turn off threshold will  
be 1.14V.  
are satisfied).  
IfRUNisbroughtbelowtheaccuratecomparatorthreshold,  
thebuck-boostconverterwillinhibitswitching,buttheV  
CC  
There may be cases due to PCB layout, very large value  
resistorsforR3andR4, orproximitytonoisycomponents  
wherenoisepickupmaycausetheturn-onorturn-offofthe  
IC to be intermittent. In these cases, a small filter capaci-  
tor can be added across R4 to ensure proper operation.  
regulator and control circuitry will remain powered unless  
RUN is brought below its logic threshold. Therefore, in  
order to completely shut down the IC and reduce the V  
IN  
current to 10nA (typical), it is necessary to ensure that  
RUNisbroughtbelowitsworstcaselowlogicthresholdof  
0.5V. RUN is a high voltage input and can be tied directly  
PGOOD Comparator  
to V to continuously enable the IC when the input supply  
IN  
TheLTC3129-1providesanopen-drainPGOODoutputthat  
is present. Also note that RUN can be driven above V  
IN  
pulls low if V  
falls more than 7.5% (typical) below its  
OUT  
or V  
as long as it stays within the operating range of  
OUT  
programmedvalue.WhenV risestowithin5%(typical)  
OUT  
the IC (up to 15V).  
of its programmed value, the internal PGOOD pull-down  
will turn off and PGOOD will go high if an external pull-  
up resistor has been provided. An internal filter prevents  
With the addition of an optional resistor divider as shown  
in Figure 2, the RUN pin can be used to establish a user-  
programmableturn-onandturn-offthreshold.Thisfeature  
can be utilized to minimize battery drain below a certain  
input voltage, or to operate the converter in a hiccup mode  
from very low current sources.  
nuisance trips of PGOOD due to short transients on V  
.
OUT  
31291fa  
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Note that PGOOD can be pulled up to any voltage, as long  
as the absolute maximum rating of 18V is not exceeded,  
and as long as the maximum sink current rating is not  
exceeded when PGOOD is low. Note that PGOOD will  
Note that external compensation should not be required  
for MPPC loop stability if the input filter capacitor, C , is  
IN  
at least 22µF. See Typical Applications for an example of  
external compensation that can be added in applications  
also be driven low if V is below its UVLO threshold or  
where C must be less than the recommended minimum  
CC  
IN  
if the part is in shutdown (RUN below its logic threshold)  
value.  
while V is being held up (or back-driven). PGOOD is  
CC  
The divider resistor values can be in the megohm range to  
minimize the input current in very low power applications.  
However,straycapacitanceandnoisepickupontheMPPC  
pin must also be minimized.  
not affected by V UVLO or the accurate RUN threshold.  
IN  
Maximum Power-Point Control (MPPC)  
The MPPC input of the LTC3129-1 can be used with an  
optional external voltage divider to dynamically adjust  
the commanded inductor current in order to maintain  
a minimum input voltage when using high resistance  
sources, such as photovoltaic panels, so as to maximize  
The MPPC pin controls the converter in a linear fashion  
when using sources that can provide a minimum of 5mA  
to 10mA of continuous input current. For operation from  
weaker input sources, refer to the Application Information  
sectiontoseehowtheprogrammableRUNpincanbeused  
to control the converter in a hysteretic manner to provide  
an effective MPPC function for sources that can provide  
as little as 5µA or less.  
input power transfer and prevent V from dropping too  
IN  
low under load. Referring to Figure 3, the MPPC pin is  
internally connected to the noninverting input of a g  
m
amplifier,whoseinvertinginputisconnectedtothe1.175V  
reference. If the voltage at MPPC, using the external volt-  
agedivider, fallsbelowthereferencevoltage, theoutputof  
If the MPPC function is not required, the MPPC pin should  
be tied to V .  
CC  
the amplifier pulls the internal V node low. This reduces  
C
V
Programming Pins  
the commanded average inductor current so as to reduce  
OUT  
the input current and regulate V to the programmed  
IN  
The LTC3129-1 has a precision internal voltage divider on  
,eliminatingtheneedforhigh-valueexternalfeedback  
minimum voltage, as given by:  
V
OUT  
resistors. This not only eliminates two external compo-  
nents,itminimizesno-loadquiescentcurrentbyusingvery  
V
= 1.175V • (1 + R5/R6)  
IN(MPPC)  
V
IN  
*C  
IN  
V
IN  
R
R5  
R6  
LTC3129-1  
S
MPPC  
+
–
+
V
SOURCE  
–
1.175V  
+
–
V
* C SHOULD BE AT  
IN  
C
CURRENT  
LEAST 22µF FOR  
COMMAND  
MPPC APPLICATIONS  
VOLTAGE  
ERROR AMP  
31291 F03  
Figure 3. MPPC Amplifier with External Resistor Divider  
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operaTion  
highresistancevaluesthatwouldnotbepracticalduetothe  
effects of noise and board leakages that would cause V  
dissipation of the IC. As described elsewhere in this data  
sheet, bootstrapping of the V for 5V output applications  
OUT  
CC  
regulation errors. The tap point on this divider is digitally  
selected by using the VS1, VS2 and VS3 pins to program  
one of eight fixed output voltages. The VS pins should be  
can essentially eliminate the V power dissipation term  
CC  
and significantly improve efficiency. As a result, careful  
consideration must be given to the thermal environment  
of the IC in order to provide a means to remove heat from  
the IC and ensure that the LTC3129-1 is able to provide  
its full rated output current. Specifically, the exposed die  
attach pad of both the QFN and MSE packages must be  
soldered to a copper layer on the PCB to maximize the  
conduction of heat out of the IC package. This can be ac-  
complished by utilizing multiple vias from the die attach  
pad connection underneath the IC package to other PCB  
layer(s) containing a large copper plane. A typical board  
layout incorporating these concepts is shown in Figure 4.  
grounded or connected to V to select the desired output  
CC  
voltage, according to the following table. The VS1, VS2  
and VS3 pins can also be driven by external logic signals  
as long as the absolute maximum voltage ratings are not  
exceeded. Note however that driving any of the voltage  
select pins high to a voltage less than the V operating  
CC  
voltage will result in increased quiescent current. Also  
note that if the VS3 pin is driven above V , an external  
CC  
1M resistor should be added in series. For other output  
voltages, refer to the LTC3129 which has a feedback pin,  
allowing any output voltage from 1.4V to 15.75V.  
If the IC die temperature exceeds approximately 180°C,  
overtemperatureshutdownwillbeinvokedandallswitching  
will be inhibited. The part will remain disabled until the die  
temperature cools by approximately 10°C. The soft-start  
circuit is re-initialized in over temperature shutdown to  
provide a smooth recovery when the IC die temperature  
cools enough to resume operation.  
VOUT Program Settings for the LTC3129-1  
VS3 PIN  
VS2 PIN  
VS1 PIN  
V
OUT  
0
0
0
0
0
0
0
2.5V  
3.3V  
4.1V  
5.0V  
6.9V  
8.2V  
12V  
V
CC  
V
V
0
CC  
CC  
V
CC  
V
CC  
V
CC  
V
CC  
V
CC  
0
0
0
GND  
V
IN  
V
CC  
V
0
CC  
CC  
C
IN  
V
V
15V  
CC  
V
CC  
Note that in shutdown, or if V is below its UVLO thresh-  
C
C
BST1  
BST2  
CC  
old, the internal voltage divider on V  
is automatically  
OUT  
L
disconnected to eliminate any current draw on V  
.
OUT  
Thermal Considerations  
C
OUT  
The power switches of the LTC3129-1 are designed to op-  
erate continuously with currents up to the internal current  
limit thresholds. However, when operating at high current  
levels, there may be significant heat generated within the  
31291 F04  
GND  
V
OUT  
IC. In addition, the V regulator can also generate wasted  
CC  
Figure 4. Typical 2-Layer PC Board Layout (MSE Package)  
heat when V is very high, adding to the total power  
IN  
31291fa  
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LTC3129-1  
applicaTions inForMaTion  
A standard application circuit for the LTC3129-1 is shown  
on the front page of this data sheet. The appropriate selec-  
tionofexternalcomponentsisdependentupontherequired  
performance of the IC in each particular application given  
considerations and trade-offs such as PCB area, input  
and output voltage range, output voltage ripple, transient  
response, required efficiency, thermal considerations and  
cost. This section of the data sheet provides some basic  
guidelines and considerations to aid in the selection of  
external components and the design of the applications  
circuit, as well as more application circuit examples.  
only applications can use the larger inductor values as  
they are unaffected by the RHP zero, while mostly boost  
applications generally require inductance on the low end  
of this range depending on how large the step-up ratio is.  
Regardless of inductor value, the saturation current rating  
should be selected such that it is greater than the worst  
case average inductor current plus half of the ripple cur-  
rent. The peak-to-peak inductor current ripple for each  
operational mode can be calculated from the following  
formula, where f is the switching frequency (1.2MHz), L  
is the inductance in µH and t  
is the switch pin mini-  
LOW  
mum low time in µs. The switch pin minimum low time  
is typically 0.09µs.  
V
Capacitor Selection  
CC  
The V output of the LTC3129-1 is generated from V  
CC  
IN  
⎛
⎞
⎛
⎞
⎟
⎠
VOUT V – V  
1
f
IN  
⎜
OUT ⎟⎜  
by a low dropout linear regulator. The V regulator has  
ΔIL(P−P)(BUCK)  
=
– tLOW  
A
CC  
⎝
⎠
L
V
IN  
⎝
been designed for stable operation with a wide range  
of output capacitors. For most applications, a low ESR  
capacitor of at least 2.2µF should be used. The capacitor  
⎛
⎜
⎝
⎞
⎟
⎠
⎛
⎜
⎝
⎞
V
L
VOUT – V  
1
f
IN  
IN  
ΔIL(P−P)(BOOST)  
=
– tLOW A  
⎟
⎠
VOUT  
should be located as close to the V pin as possible and  
CC  
connected to the V pin and ground through the shortest  
CC  
It should be noted that the worst-case peak-to-peak in-  
ductor ripple current occurs when the duty cycle in buck  
mode is minimum (highest V ) and in boost mode when  
traces possible. V is the regulator output and is also the  
CC  
internal supply pin for the LTC3129-1 control circuitry as  
well as the gate drivers and boost rail charging diodes.  
IN  
the duty cycle is 50% (V  
= 2 • V ). As an example, if  
OUT  
IN  
The V pin is not intended to supply current to other  
CC  
V (minimum) = 2.5V and V (maximum) = 15V, V  
IN  
IN  
OUT  
external circuitry.  
= 5V and L = 10µH, the peak-to-peak inductor ripples at  
the voltage extremes (15V V for buck and 2.5V V for  
IN  
IN  
Inductor Selection  
boost) are:  
The choice of inductor used in LTC3129-1 application cir-  
cuits influences the maximum deliverable output current,  
the converter bandwidth, the magnitude of the inductor  
current ripple and the overall converter efficiency. The  
inductor must have a low DC series resistance, when  
compared to the internal switch resistance, or output  
current capability and efficiency will be compromised.  
Larger inductor values reduce inductor current ripple  
but may not increase output current capability as is the  
case with peak current mode control as described in the  
Maximum Output Current section. Larger value inductors  
also tend to have a higher DC series resistance for a given  
case size, which will have a negative impact on efficiency.  
Larger values of inductance will also lower the right half  
plane(RHP)zerofrequencywhenoperatinginboostmode,  
whichcancompromiseloopstability.NearlyallLTC3129-1  
application circuits deliver the best performance with  
an inductor value between 3.3µH and 10µH. Buck mode  
BUCK = 248mA peak-to-peak  
BOOST = 93mA peak-to-peak  
One half of this inductor ripple current must be added to  
the highest expected average inductor current in order to  
selectthepropersaturationcurrentratingfortheinductor.  
To avoid the possibility of inductor saturation during load  
transients, an inductor with a saturation current rating of  
at least 600mA is recommended for all applications.  
Inadditiontoitsinfluenceonpowerconversionefficiency,  
the inductor DC resistance can also impact the maximum  
output current capability of the buck-boost converter  
particularly at low input voltages. In buck mode, the  
output current of the buck-boost converter is primarily  
limited by the inductor current reaching the average cur-  
rent limit threshold. However, in boost mode, especially  
31291fa  
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at large step-up ratios, the output current capability can  
also be limited by the total resistive losses in the power  
stage. These losses include, switch resistances, inductor  
DC resistance and PCB trace resistance. Avoid inductors  
with a high DC resistance (DCR) as they can degrade the  
maximum output current capability from what is shown  
in the Typical Performance Characteristics section and  
from the Typical Application circuits.  
Recommended inductor values for different operating  
voltage ranges are given in Table 3. These values were  
chosen to minimize inductor size while maintaining an  
acceptable amount of inductor ripple current for a given  
V and V  
range.  
IN  
OUT  
Table 3. Recommended Inductor and Output Capacitor Values  
V
AND V  
RANGE  
RECOMMENDED MAXIMUM RECOMMENDED  
IN  
OUT  
INDUCTOR  
VALUES  
TOTAL OUTPUT CAPACITOR  
VALUE FOR PWM MODE  
OPERATION AT LIGHT LOAD  
(<15mA, PWM PIN HIGH)  
As a guideline, the inductor DCR should be significantly  
less than the typical power switch resistance of 750mΩ  
each. The only exceptions are applications that have a  
maximum output current requirement much less than  
what the LTC3129-1 is capable of delivering. Generally  
speaking, inductors with a DCR in the range of 0.15Ω to  
0.3Ωarerecommended.LowervaluesofDCRwillimprove  
the efficiency at the expense of size, while higher DCR  
values will reduce efficiency (typically by a few percent)  
while allowing the use of a physically smaller inductor.  
V
IN  
V
IN  
V
IN  
V
IN  
and V  
and V  
and V  
and V  
Both < 4.5V 3.3µH to 4.7µH  
10µF  
10µF  
10µF  
10µF  
OUT  
OUT  
OUT  
OUT  
Both < 8V  
Both < 11V 6.8µH to 8.2µH  
Up to 15V 8.2µH to 10µH  
4.7µH to 6.8µH  
Due to the fixed, internal loop compensation and feedback  
dividerprovidedbytheLTC3129-1,therearelimitationsto  
themaximumrecommendedtotaloutputcapacitorvaluein  
applications that must operate in PWM mode at light load  
(PWM pin pulled high with minimum load currents less  
than ~15mA). In these applications, a maximum output  
capacitor value, shown in Table 3, is recommended. For  
applications that must operate in PWM mode at light load  
with higher values of output capacitance, the LTC3129 is  
recommended. Its external feedback pin allows the use  
of additional feedforward compensation for improved  
light-load stability under these conditions.  
Differentinductorcorematerialsandstyleshaveanimpact  
on the size and price of an inductor at any given current  
rating. Shielded construction is generally preferred as it  
minimizesthechancesofinterferencewithothercircuitry.  
Thechoiceofinductorstyledependsupontheprice,sizing,  
and EMI requirements of a particular application. Table 2  
provides a wide sampling of inductors that are well suited  
to many LTC3129-1 applications.  
Table 2. Recommended Inductors  
Note that for applications where Burst Mode operation  
is enabled (PWM pin grounded), the output capacitor  
value can be increased without limitation regardless of  
the minimum load current or inductor value.  
VENDOR  
PART  
Coilcraft  
www.coilcraft.com  
EPL2014, EPL3012, EPL3015, XFL3012  
LPS3015, LPS3314  
Coiltronics  
www.cooperindustries.com  
SDH3812, SD3814  
SD3114, SD3118  
Murata  
www.murata.com  
LQH3NP  
LQH32P  
LQH44P  
Output Capacitor Selection  
A low effective series resistance (ESR) output capacitor  
of 4.7µF minimum should be connected at the output of  
the buck-boost converter in order to minimize output volt-  
age ripple. Multilayer ceramic capacitors are an excellent  
option as they have low ESR and are available in small  
footprints. The capacitor value should be chosen large  
enough to reduce the output voltage ripple to acceptable  
levels. Neglecting the capacitor’s ESR and ESL (effec-  
tive series inductance), the peak-to-peak output voltage  
ripple in PWM mode can be calculated by the following  
Sumida  
CDRH2D16, CDRH2D18  
CDRH3D14, CDRH3D16  
www.sumida.com  
Taiyo-Yuden  
www.t-yuden.com  
NR3012T, NR3015T, NRS4012T  
BRC2518  
TDK  
www.tdk.com  
VLS3012, VLS3015  
VLF302510MT, VLF302512MT  
Toko  
www.tokoam.com  
DB3015C, DB3018C, DB3020C  
DP418C, DP420C, DEM2815C,  
DFE322512C, DFE252012C  
Würth  
www.we-online.com  
WE-TPC 2813, WE-TPC 3816,  
WE-TPC 2828  
31291fa  
19  
For more information www.linear.com/LTC3129-1  
LTC3129-1  
applicaTions inForMaTion  
formula, where f is the frequency in MHz (1.2MHz), C  
Input Capacitor Selection  
OUT  
is the capacitance in µF, t  
low time in µs (0.09µs typical) and I  
current in amperes.  
is the switch pin minimum  
LOW  
The V pin carries the full inductor current and provides  
IN  
is the output  
LOAD  
power to internal control circuits in the IC. To minimize  
input voltage ripple and ensure proper operation of the IC,  
a low ESR bypass capacitor with a value of at least 4.7µF  
ILOAD LOW  
t
ΔV  
=
V
P−P(BUCK)  
should be located as close to the V pin as possible. The  
IN  
COUT  
ILOAD  
fCOUT  
traces connecting this capacitor to V and the ground  
IN  
⎛
⎜
⎝
⎞
⎟
⎠
VOUT – VIN + tLOWfVIN  
plane should be made as short as possible.  
ΔV  
=
V
P−P(BOOST)  
VOUT  
Whenpoweredthroughlongleadsorfromapowersource  
with significant resistance, a larger value bulk input ca-  
pacitor may be required and is generally recommended.  
Insuchapplications, a47µF to100µFlow-ESRelectrolytic  
capacitorinparallelwitha1µFceramiccapacitorgenerally  
yields a high performance, low cost solution.  
Examining the previous equations reveals that the output  
voltage ripple increases with load current and is gener-  
ally higher in boost mode than in buck mode. Note that  
these equations only take into account the voltage ripple  
that occurs from the inductor current to the output being  
discontinuous. They provide a good approximation to the  
rippleatanysignificantloadcurrentbutunderestimatethe  
output voltage ripple at very light loads where the output  
voltage ripple is dominated by the inductor current ripple.  
Note that applications using the MPPC feature should  
use a minimum C of 22µF. Larger values can be used  
IN  
without limitation.  
Recommended Input and Output Capacitor Types  
In addition to the output voltage ripple generated across  
the output capacitance, there is also output voltage ripple  
produced across the internal resistance of the output  
capacitor. The ESR-generated output voltage ripple is  
proportionaltotheseriesresistanceoftheoutputcapacitor  
and is given by the following expressions where R  
the series resistance of the output capacitor and all other  
terms as previously defined.  
The capacitors used to filter the input and output of the  
LTC3129-1musthavelowESRandmustberatedtohandle  
the AC currents generated by the switching converter.  
This is important to maintain proper functioning of the  
IC and to reduce output voltage ripple. There are many  
capacitor types that are well suited to these applications  
including multilayer ceramic, low ESR tantalum, OS-CON  
and POSCAP technologies. In addition, there are certain  
types of electrolytic capacitors such as solid aluminum  
organic polymer capacitors that are designed for low ESR  
and high AC currents and these are also well suited to  
some LTC3129-1 applications. The choice of capacitor  
technology is primarily dictated by a trade-off between  
size, leakage current and cost. In backup power applica-  
tions, the input or output capacitor might be a super or  
ultra capacitor with a capacitance value measuring in the  
farad range. The selection criteria in these applications  
are generally similar except that voltage ripple is generally  
not a concern. Some capacitors exhibit a high DC leak-  
age current which may preclude their consideration for  
applications that require a very low quiescent current in  
BurstModeoperation. Notethatultracapacitorsmayhave  
is  
ESR  
ILOAD ESR  
R
ΔV  
=
≅ILOAD ESR  
R
V
P−P(BUCK)  
1– tLOW  
f
I
LOADRESR OUT  
V
ΔV  
=
P−P(BOOST)  
V 1– t  
f
(
)
IN  
LOW  
⎛
⎜
⎞
⎟
⎠
VOUT  
≅ILOAD ESR  
R
V
V
⎝
IN  
In most LTC3129-1 applications, an output capacitor be-  
tween 10µF and 22µF will work well. To minimize output  
ripple in Burst Mode operation, values of 22µF operation  
or larger are recommended.  
31291fa  
20  
For more information www.linear.com/LTC3129-1  
LTC3129-1  
applicaTions inForMaTion  
a rather high ESR, therefore a 4.7µF (minimum) ceramic  
capacitor is recommended in parallel, close to the IC pins.  
Although the converter will be operating in bursts, it is  
enough to charge an output capacitor to power low duty  
cycle loads, such as wireless sensor applications, or to  
trickle charge a battery. In addition, note that the input  
voltage will be cycling (with a small ripple as set by the  
RUN hysteresis) about a fixed voltage, as determined by  
the divider. This allows the high impedance source to  
operate at the programmed optimal voltage for maximum  
power transfer.  
Ceramic capacitors are often utilized in switching con-  
verter applications due to their small size, low ESR and  
low leakage currents. However, many ceramic capacitors  
intended for power applications experience a significant  
loss in capacitance from their rated value as the DC bias  
voltage on the capacitor increases. It is not uncommon for  
a small surface mount capacitor to lose more than 50%  
of its rated capacitance when operated at even half of its  
maximum rated voltage. This effect is generally reduced  
as the case size is increased for the same nominal value  
capacitor. As a result, it is often necessary to use a larger  
value capacitance or a higher voltage rated capacitor than  
wouldordinarilyberequiredtoactuallyrealizetheintended  
capacitanceattheoperatingvoltageoftheapplication.X5R  
and X7R dielectric types are recommended as they exhibit  
the best performance over the wide operating range and  
temperature of the LTC3129-1. To verify that the intended  
capacitance is achieved in the application circuit, be sure  
to consult the capacitor vendor’s curve of capacitance  
versus DC bias voltage.  
When using high value divider resistors (in the MΩ range)  
to minimize current draw on V , a small noise filter ca-  
IN  
pacitor may be necessary across the lower divider resis-  
tor to prevent noise from erroneously tripping the RUN  
comparator. The capacitor value should be minimized  
so as not to introduce a time delay long enough for the  
input voltage to drop significantly below the desired V  
IN  
threshold before the converter is turned off. Note that  
larger V decoupling capacitor values will minimize this  
IN  
effect by providing more holdup time on V .  
IN  
Programming the MPPC Voltage  
As discussed in the previous section, the LTC3129-1 in-  
cludes an MPPC function to optimize performance when  
operatingfromvoltagesourceswithrelativelyhighsource  
Using the Programmable RUN Function to Operate  
from Extremely Weak Input Sources  
resistance. Using an external voltage divider from V , the  
IN  
MPPCfunctiontakescontroloftheaverageinductorcurrent  
when necessary to maintain a minimum input voltage, as  
programmed by the user. Referring to Figure 3:  
Another application of the programmable RUN pin is that  
it can be used to operate the converter in a hiccup mode  
from extremely low current sources. This allows opera-  
tion from sources that can only generate microamps of  
output current, and would be far too weak to sustain  
normal steady-state operation, even with the use of the  
MPPC pin. Because the LTC3129-1 draws only 1.9µA  
V
= 1.175V • (1 + R5/R6)  
IN(MPPC)  
This is useful for such applications as photovoltaic pow-  
ered converters, since the maximum power transfer point  
occurs when the photovoltaic panel is operated at about  
75%ofitsopen-circuitvoltage.Forexample,whenoperat-  
ing from a photovoltaic panel with an open-circuit voltage  
of 5V, the maximum power transfer point will be when  
the panel is loaded such that its output voltage is about  
3.75V. Choosing values of 2MΩ for R5 and 909k for R6  
will program the MPPC function to regulate the maximum  
typical from V until it is enabled, the RUN pin can be  
IN  
programmed to keep the IC disabled until V reaches the  
IN  
programmedvoltagelevel.Inthismanner,theinputsource  
can trickle-charge an input storage capacitor, even if it  
can only supply microamps of current, until V reaches  
IN  
the turn-on threshold set by the RUN pin divider. The  
converter will then be enabled, using the stored charge  
in the input capacitor, until V drops below the turn-off  
input current so as to maintain V at a minimum of 3.74V  
IN  
IN  
threshold, at which point the converter will turn off and  
the process will repeat.  
(typical). Note that if the panel can provide more power  
than the LTC3129-1 can draw, the input voltage will rise  
above the programmed MPPC point. This is fine as long  
as the input voltage doesn't exceed 15V.  
31291fa  
This approach allows the converter to run from weak  
sources such as thin-film solar cells using indoor lighting.  
21  
For more information www.linear.com/LTC3129-1  
LTC3129-1  
applicaTions inForMaTion  
For weak input sources with very high resistance (hun-  
Powering V in this manner is referred to as bootstrap-  
CC  
dreds of Ohms or more), the LTC3129-1 may still draw  
ping. This can be done by connecting a Schottky diode  
more current than the source can provide, causing V to  
(such as a BAT54) from V to V as shown in Figure 5.  
OUT CC  
IN  
drop below the UVLO threshold. For these applications, it  
is recommended that the programmable RUN feature be  
used, as described in the previous section.  
Withthebootstrapdiodeinstalled, thegatedrivercurrents  
aresuppliedbythebuck-boostconverterathighefficiency  
rather than through the internal linear regulator. The in-  
ternal linear regulator contains reverse blocking circuitry  
MPPC Compensation and Gain  
that allows V to be driven above its nominal regulation  
CC  
level with only a very slight amount of reverse current.  
When using MPPC, there are a number of variables that  
affect the gain and phase of the input voltage control  
loop. Primarily these are the input capacitance, the MPPC  
Please note that the bootstrapping supply (either V  
or  
OUT  
a separate regulator) must be limited to less than 5.7V so  
as not to exceed the maximum V voltage of 5.5V after  
CC  
divider ratio and the V source resistance (or current). To  
IN  
the diode drop.  
simplify the design of the application circuit, the MPPC  
control loop in the LTC3129 is designed with a relatively  
low gain, such that external MPPC loop compensation is  
By maintaining V above its UVLO threshold, bootstrap-  
CC  
ping, even to a 3.3V output, also allows operation down  
generally not required when using a V capacitor value  
to the V UVLO threshold of 1.8V (typical).  
IN  
IN  
of at least 22µF. The gain from the MPPC pin to the in-  
ternal VC control voltage is about 12, so a drop of 50mV  
on the MPPC pin (below the 1.175V MPPC threshold),  
corresponds to a 600mV drop on the internal VC voltage,  
which reduces the average inductor current all the way  
to zero. Therefore, the programmed input MPPC voltage  
will be maintained within about 4% over the load range.  
V
V
OUT  
OUT  
LTC3129-1  
C
OUT  
BAT54  
V
CC  
2.2µF  
31291 F05  
Notethatiflarge-valueV capacitorsareused(whichmay  
IN  
have a relatively high ESR) a small ceramic capacitor of  
Figure 5. Example of VCC Bootstrap  
at least 4.7µF should be placed in parallel across the V  
IN  
input, near the V pin of the IC.  
IN  
Sources of Small Photovoltaic Panels  
Bootstrapping the V Regulator  
CC  
A list of companies that manufacture small solar panels  
(sometimes referred to as modules or solar cell arrays)  
suitable for use with the LTC3129-1 is provided in Table 4.  
The high and low side gate drivers are powered through  
theV rail,whichisgeneratedfromtheinputvoltage,V ,  
CC  
IN  
through an internal linear regulator. In some applications,  
especially at high input voltages, the power dissipation  
in the linear regulator can become a major contributor to  
thermalheatingoftheICandoverallefficiency.TheTypical  
Performance Characteristics section provides data on the  
Table 4. Small Photovoltaic Panel Manufacturers  
Sanyo  
http://panasonic.net/energy/amorton/en/  
http://www.powerfilmsolar.com/  
PowerFilm  
IXYS  
Corporation  
http://www.ixys.com/ProductPortfolio/GreenEnergy.aspx  
V
current and resulting power loss versus V and V  
.
CC  
IN  
OUT  
G24  
Innovations  
http://www.g24i.com/  
Asignificantperformanceadvantagecanbeattainedinhigh  
V applications where converter output voltage (V ) is  
SolarPrint  
http://www.solarprint.ie/  
IN  
OUT  
programmed to 5V, if V  
is used to power the V rail.  
OUT  
CC  
31291fa  
22  
For more information www.linear.com/LTC3129-1  
LTC3129-1  
Typical applicaTions  
Low Noise, Fixed Frequency, Wide VIN Range 12V Converter  
22nF  
22nF  
6.8µH  
V
V
< 12V, I  
> 12V, I  
= 30mA  
IN  
IN  
OUT  
OUT  
= 200mA  
BST1 SW1  
SW2 BST2  
V
V
OUT  
12V  
IN  
V
V
OUT  
IN  
2.42V TO 15V  
10µF  
16V  
LTC3129-1  
1M  
RUN  
4.7µF  
CC  
PGOOD  
PGOOD  
V
MPPC  
PWM  
VS1  
V
CC  
VS2  
2.2µF  
VS3  
GND  
PGND  
31291 TA02  
3.3V Converter Provides Extremely Long Run Time in Low Drain Applications Using Lithium Thionyl Chloride Battery  
22nF  
22nF  
4.2µH  
BST1 SW1  
SW2 BST2  
V
OUT  
V
V
V
OUT  
IN  
IN  
3.3V  
47µF  
22µF  
LTC3129-1  
1M  
RUN  
V
CC  
MPPC  
PWM  
PGOOD  
PGOOD  
Li-SoCl  
AA  
SAFT LS14500  
TADIRAN TL-4903  
2
VS1  
VS2  
VS3  
V
CC  
2.2µF  
GND  
PGND  
31291 TA03  
RUN TIME  
> 100,000 HRS (11.4 YEARS) AT 10µA (33µW) AVERAGE LOAD  
> 34,000 HRS (3.9 YEARS) AT 50µA (165µW) AVERAGE LOAD  
31291fa  
23  
For more information www.linear.com/LTC3129-1  
LTC3129-1  
Typical applicaTions  
15V Converter Powered from Flexible Solar Panel  
IOUT vs Light Level (Daylight)  
22nF  
22nF  
10µH  
100  
10  
1
I
= 32mA IN FULL SUN  
OUT  
BST1 SW1  
SW2 BST2  
V
= 6V  
1M  
MPPC  
V
IN  
V
OUT  
15V  
V
V
OUT  
IN  
10µF  
LTC3129-1  
RUN  
47µF  
MPPC  
PGOOD  
PowerFilm  
MPT6-150  
SOLAR  
PWM  
VS1  
VS2  
VS3  
MODULE  
V
V
CC  
CC  
11.4cm × 15cm  
10000  
100000  
LIGHT LEVEL (Lx)  
1000000  
2.2µF  
243k  
GND  
PGND  
31291 TA04b  
31291 TA04a  
Hiccup Converter Keeps Li-Ion Battery Charged with Indoor Lighting  
Average IOUT vs Light Level  
(Indoors)  
22nF  
22nF  
3.3µH  
1000  
100  
10  
BST1 SW1  
SW2 BST2  
V
UVLO = 3.5V  
4.42M  
IN  
V
OUT  
V
IN  
V
OUT  
4.1V  
4.7µF  
LTC3129-1  
Li-Ion  
+
4.7µF  
470µF  
6.3V  
RUN  
V
MPPC  
PGOOD  
CC  
PV PANEL  
SANYO  
PWM  
VS1  
VS2  
VS3  
AM-1815  
V
CC  
4.9cm × 5.8cm  
10pF  
2.37M  
2.2µF  
100  
1000  
LIGHT LEVEL (Lx)  
10000  
GND  
PGND  
31291 TA05b  
31291 TA05a  
31291fa  
24  
For more information www.linear.com/LTC3129-1  
LTC3129-1  
Typical applicaTions  
5V Converter Operates from Two to Eight AA or AAA Cells Using Bootstrap Diode to Increase Efficiency  
at High VIN and Extend Operation at Low VIN  
22nF  
22nF  
8.2µH  
V
V
< 5V, I  
> 5V, I  
= 100mA  
= 200mA  
IN  
IN  
OUT  
OUT  
BST1 SW1  
SW2 BST2  
V
OUT  
V
IN  
V
V
OUT  
IN  
5V  
1.92V TO 15V  
22µF  
LTC3129-1  
AFTER STARTUP  
RUN  
BAT54  
MPPC  
PGOOD  
V
CC  
TWO TO EIGHT  
AA OR AAA  
BATTERIES  
PWM  
VS1  
VS2  
VS3  
V
CC  
10µF  
2.2µF  
GND  
PGND  
31291 TA06  
3.3V Converter Uses MPPC Function to Work with High Resistance Battery Pack  
22nF  
22nF  
3.3µH  
I
= 100mA  
OUT  
BST1 SW1  
SW2 BST2  
V
= 2.9V  
1.5M  
MPPC  
V
IN  
V
OUT  
3.3V  
V
V
OUT  
IN  
10µF  
LTC3129-1  
10Ω  
10µF  
RUN  
MPPC  
PGOOD  
PWM  
VS1  
VS2  
VS3  
1.5V  
1.5V  
1.5V  
R
C
V
V
CC  
CC  
150k  
C
33pF  
C
2.2µF  
1M  
GND  
PGND  
31291 TA07  
NOTE: R AND C HAVE BEEN ADDED FOR IMPROVED MPPC LOOP STABILITY WHEN USING AN INPUT  
C
C
CAPACITOR VALUE LESS THAN THE RECOMMENDED MINIMUM OF 22µF  
31291fa  
25  
For more information www.linear.com/LTC3129-1  
LTC3129-1  
Typical applicaTions  
Solar Powered Converter Extends Battery Life in Low Power 3V Primary Battery Applications  
22nF  
22nF  
3.3µH  
FDC6312P  
DUAL PMOS  
V
OUT  
3V TO 3.3V  
S1  
S2  
BST1 SW1  
SW2 BST2  
V
UVLO = 3.7V  
IN  
3.30V  
22µF  
D1  
D2  
2.2µF  
V
V
OUT  
IN  
LTC3129-1  
4.7µF 4.99M  
2.43M  
G1  
G2  
RUN  
CR2032  
3V COIN CELL  
V
OUT  
PV PANEL  
SANYO AM-1815  
OR  
+
V
470µF  
6.3V  
CC  
MPPC  
PWM  
VS1  
VS2  
VS3  
PowerFilm SP4.2-37  
PGOOD  
2.43M  
BAT54  
V
CC  
74LVC2G04  
10pF  
GND  
PGND  
2.2µF  
31291 TA09  
Percentage of Added Battery Life vs Light Level and Load  
(PowerFilm SP4.2-37, 30sq cm Panel)  
1000  
100  
10  
AVERAGE LOAD = 165µW  
AVERAGE LOAD = 330µW  
AVERAGE LOAD = 660µW  
AVERAGE LOAD = 1650µW  
AVERAGE LOAD = 3300µW  
1
100  
1,000  
10,000  
LIGHT LEVEL (Lx)  
31291 TA09b  
31291fa  
26  
For more information www.linear.com/LTC3129-1  
LTC3129-1  
package DescripTion  
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.  
UD Package  
16-Lead Plastic QFN (3mm × 3mm)  
(Reference LTC DWG # 05-08-1700 Rev A)  
Exposed Pad Variation AA  
0.70 ±0.05  
3.50 ±0.05  
2.10 ±0.05  
1.65 ±0.05  
(4 SIDES)  
PACKAGE OUTLINE  
0.25 ±0.05  
0.50 BSC  
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS  
BOTTOM VIEW—EXPOSED PAD  
PIN 1 NOTCH R = 0.20 TYP  
OR 0.25 × 45° CHAMFER  
R = 0.115  
TYP  
0.75 ±0.05  
3.00 ±0.10  
(4 SIDES)  
15 16  
PIN 1  
TOP MARK  
(NOTE 6)  
0.40 ±0.10  
1
2
1.65 ±0.10  
(4-SIDES)  
(UD16 VAR A) QFN 1207 REV A  
0.200 REF  
0.25 ±0.05  
0.50 BSC  
0.00 – 0.05  
NOTE:  
1. DRAWING CONFORMS TO JEDEC PACKAGE OUTLINE MO-220 VARIATION (WEED-4)  
2. DRAWING NOT TO SCALE  
3. ALL DIMENSIONS ARE IN MILLIMETERS  
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE  
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE  
5. EXPOSED PAD SHALL BE SOLDER PLATED  
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION  
ON THE TOP AND BOTTOM OF PACKAGE  
31291fa  
27  
For more information www.linear.com/LTC3129-1  
LTC3129-1  
package DescripTion  
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.  
MSE Package  
16-Lead Plastic MSOP, Exposed Die Pad  
(Reference LTC DWG # 05-08-1667 Rev F)  
BOTTOM VIEW OF  
EXPOSED PAD OPTION  
2.845 ±0.102  
(.112 ±.004)  
2.845 ±0.102  
(.112 ±.004)  
0.889 ±0.127  
(.035 ±.005)  
1
8
0.35  
REF  
5.10  
(.201)  
MIN  
1.651 ±0.102  
(.065 ±.004)  
1.651 ±0.102  
(.065 ±.004)  
3.20 – 3.45  
(.126 – .136)  
0.12 REF  
DETAIL “B”  
CORNER TAIL IS PART OF  
THE LEADFRAME FEATURE.  
FOR REFERENCE ONLY  
DETAIL “B”  
16  
9
0.305 ±0.038  
0.50  
(.0197)  
BSC  
NO MEASUREMENT PURPOSE  
4.039 ±0.102  
(.159 ±.004)  
(NOTE 3)  
(.0120 ±.0015)  
TYP  
0.280 ±0.076  
(.011 ±.003)  
RECOMMENDED SOLDER PAD LAYOUT  
16151413121110  
9
REF  
DETAIL “A”  
0.254  
(.010)  
3.00 ±0.102  
(.118 ±.004)  
(NOTE 4)  
0° – 6° TYP  
4.90 ±0.152  
(.193 ±.006)  
GAUGE PLANE  
0.53 ±0.152  
(.021 ±.006)  
1 2 3 4 5 6 7 8  
DETAIL “A”  
0.86  
(.034)  
REF  
1.10  
(.043)  
MAX  
0.18  
(.007)  
SEATING  
PLANE  
0.17 – 0.27  
(.007 – .011)  
TYP  
0.1016 ±0.0508  
(.004 ±.002)  
MSOP (MSE16) 0213 REV F  
0.50  
(.0197)  
BSC  
NOTE:  
1. DIMENSIONS IN MILLIMETER/(INCH)  
2. DRAWING NOT TO SCALE  
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.  
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE  
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.  
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE  
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX  
6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL  
NOT EXCEED 0.254mm (.010") PER SIDE.  
31291fa  
28  
For more information www.linear.com/LTC3129-1  
LTC3129-1  
revision hisTory  
REV  
DATE  
DESCRIPTION  
PAGE NUMBER  
A
5/14  
Clarified V Leakage to V if V > V : from –7µA to –27µA  
4
CC  
IN  
CC  
IN  
31291fa  
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.  
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-  
tion that the interconnection of its circuits as described h einw not infringe on existing patent rights.  
erill
29  
LTC3129-1  
Typical applicaTion  
TEG Powered Converter Operates from a 10°C Temperature Differential and Provides 3.3V at 25mA  
for 50ms Every 15 Seconds for a Wireless Sensor  
COILCRAFT  
LPR6235-123QML 33nF  
1:50  
C1A  
V
STORE  
+
+
470µF  
6.3V  
220µF  
1nF  
C1B  
C2A  
•
•
MARLOW NL1025T  
TEG MOUNTED TO  
A HEAT SINK WITH  
LESS THAN 15°C/W  
THERMAL RESISTANCE  
V
OUT2  
330k  
C2B  
LTC3109  
SWA  
SWB  
V
OUT2_EN  
V
V
INA  
INB  
V
V
OUT  
AUX  
AUX  
VS2  
VS1  
PGOOD  
VLDO  
V
V
AUX  
1µF  
22nF  
22nF  
4.7µH  
BST1 SW1  
SW2 BST2  
V
OUT  
3.3V  
V
V
OUT  
1µF  
IN  
3.01M  
LTC3129-1  
10µF  
1M  
RUN  
V
CC  
PGOOD  
PGOOD  
10pF  
1M  
MPPC  
PWM  
VS1  
BAT54  
V
CC  
1N4148  
1M  
2.2µF  
VS2  
VS3  
GND  
PGND  
31291 TA08  
relaTeD parTs  
PART NUMBER DESCRIPTION  
COMMENTS  
= 2.2V, V  
LTC3103  
LTC3104  
LTC3105  
LTC3112  
LTC3115-1  
LTC3531  
15V, 300mA Synchronous Step-Down DC/DC Converter with  
Ultralow Quiescent Current  
15V, 300mA Synchronous Step-Down DC/DC Converter with  
Ultralow Quiescent Current and 10mA LDO  
V
I
= 15V, V  
= 0.8V, I = 1.8µA,  
IN(MIN)  
IN(MAX)  
OUT(MIN) Q  
<1µA, 3mm × 3mm DFN-10, MSOP-10 Packages  
SD  
V
SD  
= 2.2V, V  
= 15V, V  
= 0.8V, I = 2.8µA,  
IN(MIN)  
IN(MAX)  
OUT(MIN) Q  
I
<1µA, 4mm × 3mm DFN-14, MSOP-16 Packages  
400mA Step-up Converter with MPPC and 250mV Start-Up  
V
SD  
= 0.2V, V  
= 5V, V  
= 0 5.25V , I = 22µA,  
MAX Q  
IN(MIN)  
IN(MAX)  
OUT(MIN)  
I
<1µA, 3mm × 3mm DFN-10/MSOP-12 Packages  
15V, 2.5A, 750kHz Monolithic Synch Buck/Boost  
V
SD  
= 2.7V, V  
= 15V, V  
= 2.7V to 14V, I = 50µA,  
IN(MIN)  
IN(MAX)  
OUT(MIN) Q  
I
<1µA, 4mm × 5mm DFN-16 TSSOP-20E Packages  
40V, 2A, 2MHz Monolithic Synch Buck/Boost  
V
SD  
= 2.7V, V  
= 40V, V  
= 2.7V to 40V, I = 50µA,  
IN(MIN)  
IN(MAX)  
OUT(MIN) Q  
I
<1µA, 4mm × 5mm DFN-16 and TSSOP-20E Packages  
5.5V, 200mA, 600kHz Monolithic Synch Buck/Boost  
20V, 50mA High Efficiency Nano Power Step-Down Regulator  
Ultralow Voltage Step-Up Converter and Power Manager  
V
SD  
= 1.8V, V  
= 5.5V, V  
= 2V to 5V, I = 16µA,  
OUT(MIN) Q  
IN(MIN)  
IN(MAX)  
I
<1µA, 3mm × 3mm DFN-8 and ThinSOT Packages  
LTC3388-1/  
LTC3388-3  
LTC3108/  
LTC3108-1  
V
= 2.7V, V  
=20V, V  
= Fixed 1.1V to 5.5V,  
IN(MIN)  
IN(MAX)  
OUT(MIN)  
I = 720nA, I = 400nA, 3mm × 3mm DFN-10, MSOP-10 Packages  
Q
SD  
V
= 0.02V, V  
= 1V, V  
= Fixed 2.35V to 5V,  
OUT(MIN)  
IN(MIN)  
IN(MAX)  
I = 6µA, I <1µA, 3mm × 4mm DFN-12, SSOP-16 Packages  
Q
SD  
LTC3109  
LTC3588-1  
LTC4070  
Auto-Polarity, Ultralow Voltage Step-Up Converter and Power  
Manager  
Piezo Electric Energy Harvesting Power Supply  
V
Q
= 0.03V, V  
SD  
= 1V, V  
= Fixed 2.35V to 5V,  
OUT(MIN)  
IN(MIN)  
IN(MAX)  
I = 7µA, I <1µA, 4mm × 4mm QFN-20, SSOP-20 Packages  
V
= 2.7V, V  
= 20V, V  
= Fixed 1.8V to 3.6V,  
IN(MIN)  
IN(MAX)  
OUT(MIN)  
I = 950nA, I 450nA, 3mm × 3mm DFN-10, MSOP-10E Packages  
Q
SD  
Li-Ion/Polymer Low Current Shunt Battery Charger System  
V
= 450nA to 50mA, V  
+ 4.0V, 4.1V, 4.2V, I = 300nA,  
IN(MIN)  
FLOAT Q  
2mm × 3mm DFN-8, MSOP-8 Packages  
31291fa  
LT 0514 REV A • PRINTED IN USA  
LinearTechnology Corporation  
30 1630 McCarthy Blvd., Milpitas, CA 95035-7417  
●
●
LINEAR TECHNOLOGY CORPORATION 2013  
(408)432-1900 FAX: (408) 434-0507 www.linear.com/3129-1  

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