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LTC3129 [Linear]

15V, 200mA Synchronous Buck-Boost DC/DC Converter with 1.3μA Quiescent Current; 15V型，200mA同步降压 - 升压型DC / DC转换器1.3μA静态电流
LTC3129
型号： LTC3129
厂家： Linear    Linear
描述：

15V, 200mA Synchronous Buck-Boost DC/DC Converter with 1.3μA Quiescent Current
15V型，200mA同步降压 - 升压型DC / DC转换器1.3μA静态电流

转换器
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LTC3129  
15V, 200mA Synchronous  
Buck-Boost DC/DC Converter  
with 1.3µA Quiescent Current  
FeaTures  
DescripTion  
n
Regulates V  
Above, Below or Equal to V  
The LTC®3129 is a high efficiency, 200mA buck-boost  
OUT  
IN  
n
Wide V Range: 2.42V to 15V, 1.92V to 15V After  
DC/DCconverterwithawideV andV range.Itincludes  
IN  
IN OUT  
Start-Up (Bootstrapped)  
anaccurateRUNpinthresholdtoallowpredictableregula-  
tor turn-on and a maximum power point control (MPPC)  
capability that ensures maximum power extraction from  
non-ideal power sources such as photovoltaic panels.  
n
n
n
n
n
n
n
n
n
n
n
n
n
Wide V  
Range: 1.4V to 15.75V  
OUT  
200mA Output Current in Buck Mode  
Single Inductor  
1.3µA Quiescent Current  
The LTC3129 employs an ultralow noise, 1.2MHz PWM  
switchingarchitecturethatminimizessolutionfootprintby  
allowing the use of tiny, low profile inductors and ceramic  
capacitors. Built-in loop compensation and soft-start  
simplify the design. For high efficiency operation at light  
loads, automatic Burst Mode operation can be selected,  
reducing the quiescent current to just 1.3µA.  
Programmable Maximum Power Point Control  
1.2MHz Ultralow Noise PWM  
Current Mode Control  
Pin Selectable Burst Mode® Operation  
Up to 95% Efficiency  
Accurate RUN Pin Threshold  
Power Good Indicator  
10nA Shutdown Current  
Thermally Enhanced 3mm × 3mm QFN and  
16-Lead MSOP Packages  
Additionalfeaturesincludeapowergoodoutput, lessthan  
10nA of shutdown current and thermal shutdown.  
The LTC3129 is available in thermally enhanced 3mm ×  
3mm QFN and 16-lead MSOP packages. For fixed output  
voltageoptions,seethefunctionallyequivalentLTC3129-1,  
which eliminates the need for an external feedback divider.  
L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks  
and PowerPath is a trademark of Linear Technology Corporation. All other trademarks are the  
property of their respective owners.  
applicaTions  
n
Industrial Wireless Sensor Nodes  
n
Post-Regulator for Harvested Energy  
n
Solar Panel Post-Regulator/Charger  
n
Intrinsically Safe Power Supplies  
n
Wireless Microphones  
n
Avionics-Grade Wireless Headsets  
Typical applicaTion  
22nF  
22nF  
Efficiency and Power Loss vs Load  
10µH  
100  
90  
80  
70  
60  
50  
40  
30  
20  
10  
0
1000  
100  
10  
5V AT 200mA, V > 5V  
IN  
EFFICIENCY  
5V AT 100mA, V < 5V  
IN  
BST1 SW1  
SW2 BST2  
2.42V TO 15V  
V
V
V
OUT  
V
IN  
IN  
OUT  
10µF  
10pF  
LTC3129  
3.32M  
RUN  
10µF  
POWER LOSS  
MPPC  
PWM  
FB  
V
1
CC  
PGOOD  
1.02M  
V
V
V
V
= 2.5V  
= 3.6V  
= 5V  
IN  
IN  
IN  
IN  
0.1  
V
CC  
= 15V  
V
= 5V  
OUT  
0.01  
2.2µF  
GND  
PGND  
0.01  
0.1  
1
10  
100  
1000  
OUTPUT CURRENT (mA)  
3129 TA01b  
3129 TA01a  
3129f  
1
For more information www.linear.com/3129  
LTC3129  
absoluTe MaxiMuM raTings  
(Notes 1, 8)  
V , V  
Voltages .................................... –0.3V to 18V  
V , FB, PWM, MPPC Voltages.................... –0.3V to 6V  
IN OUT  
CC  
SW1 DC Voltage............................ –0.3V to (V + 0.3V)  
PGOOD Sink Current .............................................15mA  
IN  
SW2 DC Voltage..........................–0.3V to (V  
+ 0.3V)  
Operating Junction Temperature Range  
OUT  
SW1, SW2 Pulsed (<100ns) Voltage ..............–1V to 19V  
BST1 Voltage ....................(SW1 – 0.3V) to (SW1 + 6V)  
BST2 Voltage ....................(SW2 – 0.3V) to (SW2 + 6V)  
RUN, PGOOD Voltages............................... –0.3V to 18V  
(Notes 2, 5)............................................ –40°C to 125°C  
Storage Temperature Range .................. –65°C to 150°C  
MSE Lead Temperature (Soldering, 10 sec) ..........300°C  
pin conFiguraTion  
TOP VIEW  
TOP VIEW  
16 15 14 13  
1
2
3
4
5
6
7
8
V
16 V  
IN  
CC  
RUN  
MPPC  
GND  
FB  
15 BST1  
14 SW1  
13 PGND  
12 SW2  
11 BST2  
BST1  
1
2
3
4
12  
V
OUT  
V
11 PGOOD  
IN  
17  
PGND  
17  
PGND  
V
PWM  
NC  
10  
9
CC  
NC  
RUN  
NC  
10  
9
V
OUT  
PGOOD  
PWM  
5
6
7
8
MSE PACKAGE  
16-LEAD PLASTIC MSOP  
T
= 125°C, θ = 10°C/W, θ = 40°C/W (NOTE 6)  
JC JA  
JMAX  
UD PACKAGE  
16-LEAD (3mm × 3mm) PLASTIC QFN  
EXPOSED PAD (PIN 17) IS PGND, MUST BE SOLDERED TO PCB  
T
= 125°C, θ = 7.5°C/W, θ = 68°C/W (NOTE 6)  
JC JA  
JMAX  
EXPOSED PAD (PIN 17) IS PGND, MUST BE SOLDERED TO PCB  
orDer inForMaTion  
LEAD FREE FINISH  
LTC3129EUD#PBF  
LTC3129IUD#PBF  
LTC3129EMSE#PBF  
LTC3129IMSE#PBF  
TAPE AND REEL  
PART MARKING*  
LGDR  
PACKAGE DESCRIPTION  
16-Lead (3mm × 3mm) Plastic QFN  
TEMPERATURE RANGE  
–40°C to 125°C  
LTC3129EUD#TRPBF  
LTC3129IUD#TRPBF  
LTC3129EMSE#TRPBF  
LTC3129IMSE#TRPBF  
LGDR  
16-Lead (3mm × 3mm) Plastic QFN  
16-Lead Plastic MSOP  
–40°C to 125°C  
–40°C to 125°C  
–40°C to 125°C  
3129  
3129  
16-Lead Plastic MSOP  
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.  
Consult LTC Marketing for information on non-standard lead based finish parts.  
For more information on lead free part marking, go to: http://www.linear.com/leadfree/  
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/  
3129f  
2
For more information www.linear.com/3129  
LTC3129  
elecTrical characTerisTics The l denotes the specifications which apply over the specified operating  
junction temperature range, otherwise specifications are at TA = 25°C (Note 2). Unless otherwise noted, VIN = 12V, VOUT = 5V.  
PARAMETER  
Start-Up Voltage  
CONDITIONS  
MIN  
TYP  
MAX  
2.42  
15  
UNITS  
V
l
l
l
l
l
l
V
2.25  
IN  
Input Voltage Range  
V
V
> 2.42V (Back-Driven)  
> 2.42V (Back-Driven)  
1.92  
1.8  
V
CC  
CC  
V
V
UVLO Threshold (Rising)  
UVLO Hysteresis  
1.9  
2.0  
V
IN  
IN  
80  
100  
130  
15.75  
1.199  
10  
mV  
V
Output Voltage Adjust Range  
Feedback Voltage  
1.4  
1.151  
1.175  
0.1  
V
Feedback Input Current  
FB = 1.25V  
nA  
nA  
µA  
Quiescent Current (V ) – Shutdown  
RUN = 0V, Including Switch Leakage  
10  
100  
3
IN  
Quiescent Current (V ) UVLO  
Either V or V Below Their UVLO Threshold, or  
1.9  
IN  
IN  
CC  
RUN Below the Threshold to Enable Switching  
Quiescent Current – Burst Mode Operation  
Measured on V , FB > 1.25V  
1.3  
10  
2.0  
50  
µA  
nA  
IN  
PWM = 0V, RUN = V  
IN  
N-Channel Switch Leakage on V and V  
SW1 = 0V, V = 15V  
IN  
IN  
OUT  
SW2 = 0V, V  
RUN = 0V  
= 15V  
OUT  
N-Channel Switch On-Resistance  
Inductor Average Current Limit  
V
= 4V  
0.75  
Ω
CC  
l
l
V
V
> UV Threshold (Note 4)  
< UV Threshold (Note 4)  
220  
80  
275  
130  
350  
200  
mA  
mA  
OUT  
OUT  
l
l
Inductor Peak Current Limit  
Maximum Boost Duty Cycle  
(Note 4)  
400  
85  
500  
89  
680  
95  
mA  
%
FB = 1.10V. Percentage of Period SW2 is Low in  
Boost Mode (Note 7)  
l
l
Minimum Duty Cycle  
FB = 1.25V. Percentage of Period SW1 is High in  
Buck Mode (Note 7)  
0
%
Switching Frequency  
SW1 and SW2 Minimum Low Time  
MPPC Voltage  
PWM = V  
(Note 3)  
1.0  
1.2  
90  
1.4  
MHz  
ns  
V
CC  
l
1.12  
1.175  
1
1.22  
10  
MPPC Input Current  
MPPC = 5V  
> 2.4V  
nA  
V
l
l
RUN Threshold to Enable V  
0.5  
1.16  
50  
0.9  
1.22  
80  
1.15  
1.28  
120  
10  
CC  
RUN Threshold to Enable Switching (Rising)  
RUN (Switching) Threshold Hysteresis  
RUN Input Current  
V
V
CC  
mV  
nA  
V
RUN = 15V  
PWM = 5V  
1
l
l
PWM Input High  
1.6  
PWM Input Low  
0.5  
1
V
PWM Input Current  
0.1  
3
µA  
ms  
V
Soft-Start Time  
l
l
V
V
Voltage  
V
IN  
> 4.85V  
3.4  
4.1  
4.7  
CC  
CC  
Dropout Voltage (V – V  
)
V
IN  
V
IN  
= 3.0V, Switching  
35  
0
60  
2
mV  
mV  
IN  
CC  
= 2.0V (V in UVLO)  
CC  
V
V
V
V
V
V
V
UVLO Threshold (Rising)  
UVLO Hysteresis  
Current Limit  
2.1  
4
2.25  
60  
2.42  
V
mV  
mA  
V
CC  
CC  
CC  
CC  
CC  
CC  
OUT  
l
l
V
CC  
= 0V  
20  
40  
5.5  
4
Back-Drive Voltage (Maximum)  
Input Current (Back-Driven)  
V
V
= 5.5V (Switching)  
2
mA  
µA  
V
CC  
Leakage to V if V > V  
= 5.5V, V = 1.8V, Measured on V  
IN  
–7  
IN  
CC  
IN  
CC  
IN  
l
UV Threshold (Rising)  
0.95  
1.15  
1.35  
3129f  
3
For more information www.linear.com/3129  
LTC3129  
elecTrical characTerisTics The l denotes the specifications which apply over the specified operating  
junction temperature range, otherwise specifications are at TA = 25°C (Note 2). Unless otherwise noted, VIN = 12V, VOUT = 5V.  
PARAMETER  
CONDITIONS  
MIN  
TYP  
150  
10  
MAX  
UNITS  
mV  
nA  
V
OUT  
V
OUT  
V
OUT  
V
OUT  
UV Hysteresis  
Current – Shutdown  
Current – Sleep  
Current – Active  
RUN = 0V, V  
= 15V Including Switch Leakage  
100  
OUT  
PWM = 0V, FB = 1.25V  
PWM = V , V = 15V (Note 4), FB = 1.25V  
V
/27  
OUT  
µA  
5
9
µA  
CC OUT  
PGOOD Threshold, Falling  
PGOOD Hysteresis  
PGOOD Voltage Low  
PGOOD Leakage  
Referenced to Programmed V  
Referenced to Programmed V  
Voltage  
Voltage  
–5.5  
–7.5  
2.5  
250  
1
–10  
%
OUT  
OUT  
%
I
= 1mA  
300  
50  
mV  
nA  
SINK  
PGOOD = 15V  
Note 1: Stresses beyond those listed under Absolute Maximum Ratings  
may cause permanent damage to the device. Exposure to any Absolute  
Maximum Rating condition for extended periods may affect device  
reliability and lifetime.  
Note 4: Current measurements are made when the output is not switching.  
Note 5: This IC includes overtemperature protection that is intended  
to protect the device during momentary overload conditions. Junction  
temperature will exceed 125°C when overtemperature protection is active.  
Continuous operation above the specified maximum operating junction  
temperature may result in device degradation or failure.  
Note 2: The LTC3129 is tested under pulsed load conditions such that  
T ≈ T . The LTC3129E is guaranteed to meet specifications from  
J
A
0°C to 85°C junction temperature. Specifications over the –40°C to  
125°C operating junction temperature range are assured by design,  
characterization and correlation with statistical process controls. The  
LTC3129I is guaranteed over the full –40°C to 125°C operating junction  
Note 6: Failure to solder the exposed backside of the package to the PC  
board ground plane will result in a much higher thermal resistance.  
Note 7: Switch timing measurements are made in an open-loop test  
configuration. Timing in the application may vary somewhat from these  
values due to differences in the switch pin voltage during non-overlap  
durations when switch pin voltage is influenced by the magnitude and  
duration of the inductor current.  
Note 8: Voltage transients on the switch pin(s) beyond the DC limits  
specified in the Absolute Maximum Ratings are non-disruptive to normal  
operation when using good layout practices as described elsewhere in the  
data sheet and application notes and as seen on the product demo board.  
temperature range. The junction temperature (T ) is calculated from the  
J
ambient temperature (T ) and power dissipation (P ) according to the  
A
D
formula: T = T + (P • θ °C/W), where θ is the package thermal  
J
A
D
JA  
JA  
impedance. Note that the maximum ambient temperature consistent  
with these specifications, is determined by specific operating conditions  
in conjunction with board layout, the rated thermal package thermal  
resistance and other environmental factors.  
Note 3: Specification is guaranteed by design and not 100% tested in  
production.  
TA = 25°C, unless otherwise noted.  
Typical perForMance characTerisTics  
Efficiency, VOUT = 2.5V  
Power Loss, VOUT = 2.5V  
Efficiency, VOUT = 3.3V  
100  
90  
80  
70  
60  
50  
40  
30  
20  
10  
0
1000  
100  
10  
100  
90  
80  
70  
60  
50  
40  
30  
20  
10  
0
BURST  
BURST  
PWM  
PWM  
PWM  
1
V
V
V
V
V
= 2.5V  
= 3.6V  
= 5V  
= 10V  
= 15V  
V
V
V
V
V
= 2.5V  
= 3.6V  
= 5V  
= 10V  
= 15V  
V
IN  
V
IN  
V
IN  
V
IN  
V
IN  
= 2.5V  
= 3.6V  
= 5V  
= 10V  
= 15V  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
BURST  
0.1  
0.01  
0.01  
0.1  
1
10  
100  
1000  
0.01  
0.1  
1
10  
100  
1000  
0.01  
0.1  
1
10  
100  
1000  
OUTPUT CURRENT (mA)  
OUTPUT CURRENT (mA)  
OUTPUT CURRENT (mA)  
3129 G01  
3129 G02  
3129 G03  
3129f  
4
For more information www.linear.com/3129  
LTC3129  
TA = 25°C, unless otherwise noted.  
Typical perForMance characTerisTics  
Power Loss, VOUT = 3.3V  
Efficiency, VOUT = 5V  
Power Loss, VOUT = 5V  
100  
90  
80  
70  
60  
50  
40  
30  
20  
10  
0
1000  
100  
10  
1000  
100  
10  
BURST  
PWM  
PWM  
PWM  
1
1
V
= 2.5V  
= 3.6V  
= 5V  
= 10V  
= 15V  
V
V
V
V
V
= 2.5V  
= 3.6V  
= 5V  
= 10V  
= 15V  
V
V
V
V
V
= 2.5V  
= 3.6V  
= 5V  
= 10V  
= 15V  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
BURST  
BURST  
V
V
V
V
0.1  
0.1  
0.01  
0.01  
0.01  
0.1  
1
10  
100  
1000  
0.01  
0.1  
1
10  
100  
1000  
0.01  
0.1  
1
10  
100  
1000  
OUTPUT CURRENT (mA)  
OUTPUT CURRENT (mA)  
OUTPUT CURRENT (mA)  
3129 G05  
3129 G06  
3129 G04  
Efficiency, VOUT = 12V  
Power Loss, VOUT = 12V  
Efficiency, VOUT = 15V  
100  
90  
80  
70  
60  
50  
40  
30  
20  
10  
0
1000  
100  
10  
100  
90  
80  
70  
60  
50  
40  
30  
20  
10  
0
BURST  
BURST  
PWM  
PWM  
BURST  
PWM  
1
V
= 2.5V  
= 3.6V  
= 5V  
= 10V  
= 15V  
V
= 2.5V  
= 3.6V  
= 5V  
= 10V  
= 15V  
V
= 2.5V  
= 3.6V  
= 5V  
= 10V  
= 15V  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
IN  
V
V
V
0.1  
V
V
V
V
V
V
V
V
V
IN  
IN  
IN  
0.01  
0.01  
0.1  
1
10  
100  
1000  
0.01  
0.1  
1
10  
100  
1000  
0.01  
0.1  
1
10  
100  
1000  
OUTPUT CURRENT (mA)  
OUTPUT CURRENT (mA)  
OUTPUT CURRENT (mA)  
3129 G07  
3129 G08  
3129 G09  
Maximum Output Current  
vs VIN and VOUT  
No Load Input Current  
vs VIN and VOUT (PWM = 0V)  
Power Loss, VOUT = 15V  
250  
200  
150  
100  
50  
30  
25  
1000  
100  
10  
V
V
V
V
= 2.5V  
OUT  
OUT  
OUT  
OUT  
= 5V  
= 10V  
= 15V  
PWM  
FB DIVIDER CURRENT = 2µA  
20  
15  
10  
5
V
V
V
V
V
V
V
V
= 2.5V  
OUT  
OUT  
OUT  
OUT  
OUT  
OUT  
OUT  
OUT  
BURST  
= 3.3V  
= 4.1V  
= 5V  
1
V
= 2.5V  
= 3.6V  
= 5V  
= 10V  
= 15V  
IN  
IN  
IN  
IN  
= 6.9V  
= 8.2V  
= 12V  
= 15V  
V
0.1  
V
V
V
IN  
0
0
0.01  
2
3
4
5
6
7
8
9
10 11 12 13 14 15  
2.5  
4.5  
6.5  
8.5 10.5 12.5 14.5  
(V)  
0.01  
0.1  
1
10  
100  
1000  
V
(V)  
V
OUTPUT CURRENT (mA)  
IN  
IN  
3129 G11  
3129 G12  
3129 G10  
3129f  
5
For more information www.linear.com/3129  
LTC3129  
TA = 25°C, unless otherwise noted.  
Typical perForMance characTerisTics  
FB Voltage vs Temperature  
(Normalized to 25°C)  
Burst Mode Threshold  
vs VIN and VOUT  
Switch RDS(ON) vs Temperature  
1.00  
0.75  
0.50  
0.25  
0
80  
70  
60  
50  
40  
30  
1.3  
1.2  
1.1  
1.0  
0.9  
0.8  
0.7  
0.6  
0.5  
0.4  
V
CC  
V
CC  
V
CC  
V
CC  
= 2.5V  
= 3V  
= 4V  
= 5V  
V
V
V
V
V
V
V
V
= 2.5V  
= 3.3V  
= 4.1V  
= 5V  
OUT  
OUT  
OUT  
OUT  
OUT  
OUT  
OUT  
OUT  
–0.25  
–0.50  
–0.75  
–1.00  
20  
10  
= 6.9V  
= 8.2V  
= 12V  
= 15V  
0
–45 –20  
5
30  
55  
80 105 130  
–45 –20  
5
30  
55  
80 105 130  
2
4
6
8
10  
(V)  
12  
14  
16  
TEMPERATURE (°C)  
TEMPERATURE (°C)  
V
IN  
3129 G15  
3129 G14  
3129 G13  
Maximum Output Current  
vs Temperature (Normalized to 25°C)  
Average Input Current Limit  
vs MPPC Voltage  
Accurate RUN Threshold  
vs Temperature (Normalized to 25°C)  
2
1
100  
90  
80  
70  
60  
50  
40  
30  
20  
10  
0
15  
10  
5
0
0
–5  
–10  
–15  
–1  
–2  
–45 –20  
5
30  
55  
80 105 130  
1.13 1.135 1.14 1.145 1.15 1.155 1.16 1.165 1.17  
–45 –20  
5
30  
55  
80 105 130  
TEMPERATURE (°C)  
MPPC PIN VOLTAGE (V)  
TEMPERATURE (°C)  
3129 G16  
3129 G17  
3129 G18  
VCC Dropout Voltage vs Temperature  
(PWM Mode, Switching)  
VCC Dropout Voltage vs VIN  
(PWM Mode, Switching)  
Fixed Frequency PWM  
Waveforms  
60  
50  
40  
30  
20  
10  
0
60  
SW2  
5V/DIV  
50  
40  
30  
20  
10  
0
SW1  
5V/DIV  
I
L
200mA/DIV  
3129 G21  
500ns/DIV  
L = 10µH  
V
V
= 7V  
IN  
= 5V  
OUT  
OUT  
–45 –20  
5
30  
55  
80 105 130  
2
2.25 2.5 2.75  
3
3.25 3.5 3.75  
4
I
= 200mA  
TEMPERATURE (°C)  
V
(V)  
IN  
3129 G19  
3129 G20  
3129f  
6
For more information www.linear.com/3129  
LTC3129  
TA = 25°C, unless otherwise noted.  
Typical perForMance characTerisTics  
Burst Mode Ripple on VOUT  
Fixed Frequency Ripple on VOUT  
Burst Mode Waveforms  
SW1  
5V/DIV  
V
OUT  
V
OUT  
100mV/DIV  
20mV/DIV  
SW2  
5V/DIV  
I
L
200mA/DIV  
I
I
L
L
200mA/DIV  
100mA/DIV  
3129 G23  
3129 G22  
3129 G24  
50µs/DIV  
L = 10µH  
200ns/DIV  
100µs/DIV  
L = 10µH  
L = 10µH  
V
V
I
= 7V  
V
V
I
= 7V  
V
V
I
= 7V  
IN  
OUT  
IN  
OUT  
IN  
OUT  
OUT  
= 5V  
= 5V  
= 5mA  
= 22µF  
= 5V  
= 5mA  
= 200mA  
= 10µF  
OUT  
OUT  
OUT  
C
C
C
= 22µF (WITH THE RECOMMENDED  
FEEDFORWARD CAPACITOR)  
OUT  
OUT  
Step Load Transient Response in  
Burst Mode Operation  
Step Load Transient Response in  
Fixed Frequency  
Start-Up Waveforms  
V
OUT  
5V/DIV  
V
V
OUT  
100mV/DIV  
OUT  
100mV/DIV  
V
CC  
5V/DIV  
RUN  
5V/DIV  
I
VOUT  
I
I
VOUT  
VIN  
100mA/DIV  
100mA/DIV  
200mA/DIV  
3129 G25  
3129 G26  
3129 G27  
1ms/DIV  
500µs/DIV  
500µs/DIV  
V
V
I
= 7V  
L = 10µH  
IN  
OUT  
= 5V  
V
V
C
I
= 7V  
= 7V  
IN  
OUT  
OUT  
= 50mA  
= 22µF  
= 5V  
= 10µF  
OUT  
OUT  
C
= 22µF (WITH THE RECOMMENDED  
= 50mA to 150mA STEP  
FEEDFORWARD CAPACITOR)  
OUT  
= 5mA to 125mA STEP  
PGOOD Response to a Drop  
On VOUT  
MPPC Response to a Step Load  
V
OUT  
2V/DIV  
PGOOD  
2V/DIV  
V
IN  
2V/DIV  
V
OUT  
2V/DIV  
I
VOUT  
100mA/DIV  
3129 G28  
3129 G29  
1ms/DIV  
2ms/DIV  
SET TO 3.5V  
V
= 5V  
V
V
C
V
= 5V  
OC  
MPPC  
= 22µF, R = 10Ω,  
OUT  
OUT  
OUT  
IN  
IN  
IN  
= 5V, C  
= 22µF  
OUT  
I
= 25mA to 125mA STEP  
3129f  
7
For more information www.linear.com/3129  
LTC3129  
pin FuncTions (QFN/MSOP)  
BST1 (Pin 1/Pin 15): Bootstrapped Floating Supply for  
High Side NMOS Gate Drive. Connect to SW1 through a  
22nF capacitor, as close to the part as possible. The value  
is not critical. Any value from 4.7nF to 47nF may be used.  
V
= 1.175V • [1+(R1/R2)]. Note this pin is very noise  
OUT  
sensitive, therefore minimize trace length and stray ca-  
pacitance.  
NC (Pins 8, 9/Pins 6, 7): Unused. These pins should be  
grounded.  
V (Pin2/Pin16):InputVoltagefortheConverter.Connect  
IN  
a minimum of 4.7µF ceramic decoupling capacitor from  
PWM (Pin 10/Pin 8): Mode Select Pin.  
thispintothegroundplane, asclosetothepinaspossible.  
PWM = Low (ground): Enables automatic Burst Mode  
operation.  
V
CC  
(Pin 3/Pin 1): Output voltage of the internal voltage  
regulator. This is the supply pin for the internal circuitry.  
Bypass this output with a minimum of 2.2µF ceramic  
capacitor close to the pin. This pin may be back-driven  
by an external supply, up to a maximum of 5.5V.  
PWM = High (tie to V ): Fixed frequency PMW opera-  
CC  
tion.  
This pin should not be allowed to float. It has an internal  
5M pull-down resistor.  
RUN (Pin 4/Pin 2): Input to the Run Comparator. Pull this  
pinabove1.1VtoenabletheV regulatorandabove1.28V  
CC  
PGOOD (Pin 11/Pin 9): Open drain output that pulls to  
ground when FB drops too far below its regulated volt-  
age. Connect a pull-up resistor from this pin to a positive  
supply. This pin can sink up to the absolute maximum  
rating of 15mA when low. Note that this pin is forced low  
to enable the converter. Connecting this pin to a resistor  
divider from V to ground allows programming a V  
IN  
IN  
start threshold higher than the 1.8V (typical) V UVLO  
IN  
threshold. In this case, the typical V turn-on threshold is  
IN  
determined by V = 1.22V • [1+(R3/R4)] (see Figure 2).  
IN  
in shutdown or V UVLO.  
CC  
MPPC (Pin 5/Pin 3): Maximum Power Point Control Pro-  
V
(Pin 12/Pin 10): Output voltage of the converter.  
OUT  
gramming Pin. Connect this pin to a resistor divider from  
Connectaminimumvalueof4.7µFceramiccapacitorfrom  
V togroundtoenabletheMPPCfunctionality. IftheV  
IN  
OUT  
thispintothegroundplane, asclosetothepinaspossible.  
load is greater than what the power source can provide,  
BST2 (Pin 13/Pin 11): Bootstrapped floating supply for  
high side NMOS gate drive. Connect to SW2 through a  
22nF capacitor, as close to the part as possible. The value  
is not critical. Any value from 4.7nF to 47nF may be used.  
the MPPC will reduce the inductor current to regulate V  
IN  
to a voltage determined by: V = 1.175V • [1+(R5/R6)]  
IN  
(see Figure 3). By setting the V regulation voltage appro-  
IN  
priately, maximum power transfer from the limited source  
is assured. Note this pin is very noise sensitive, therefore  
minimize trace length and stray capacitance. Please refer  
to the Applications Information section for more detail  
on programming the MPPC for different sources. If this  
SW2 (Pin 14/Pin 12): Switch Pin. Connect to one side of  
the inductor. Keep PCB trace lengths as short and wide  
as possible to reduce EMI.  
PGND (Pin 15, Exposed Pad Pin 17/Pin 13, Exposed  
Pad Pin 17): Power Ground. Provide a short direct PCB  
path between PGND and the ground plane. The exposed  
pad must also be soldered to the PCB ground plane. It  
serves as a power ground connection, and as a means of  
conducting heat away from the die.  
function is not needed, tie the pin to V .  
CC  
GND (Pin 6/Pin 4): Signal Ground. Provide a short direct  
PCB path between GND and the ground plane where the  
exposed pad is soldered.  
FB (Pin 7/Pin 5): Feedback Input to the Error Amplifier.  
Connect to a resistor divider from V  
to ground. The  
SW1 (Pin 16/Pin 14): Switch Pin. Connect to one side of  
the inductor. Keep PCB trace lengths as short and wide  
as possible to reduce EMI.  
OUT  
output voltage can be adjusted from 1.4V to 15.75V by:  
3129f  
8
For more information www.linear.com/3129  
LTC3129  
block DiagraM  
BST1  
SW1  
SW2  
BST2  
V
IN  
V
IN  
V
CC  
V
REF  
LDO  
V
CC_GD  
V
OUT  
START  
V
OUT  
A
B
DRIVER  
DRIVER  
V
CC  
V
4.1V  
CC  
V
CC  
I
I
SENSE  
SENSE  
D
DRIVER  
1.175V  
V
START  
RUN  
V
REF  
REF  
NC  
NC  
V
REF_GD  
C
DRIVER  
DRV_C  
+
–
START  
DRV_B  
0.9V  
DRV_A  
DRV_D  
+
–
SD  
I
SENSE  
+
–
–
+
UV  
FB  
1.22V  
500mA  
I
V
IN  
1.1V  
LIM  
1.175V  
LOGIC  
ENABLE  
I
SENSE  
–
+
UVLO  
–
+
V
C
I
–
+
I
ZERO  
SENSE  
–
+
+
–
PWM  
1.175V  
20mA  
THERMAL  
SHUTDOWN  
RESET  
SOFT-START  
OSC  
MPPC  
PWM  
+
–
1.175V  
PGOOD  
–
+
–
+
600mV  
CLAMP  
5M  
–
+
SLEEP  
–7.5%  
100mV  
GND  
PGND  
3129 BD  
3129f  
9
For more information www.linear.com/3129  
LTC3129  
operaTion  
INTRODUCTION  
PWM MODE OPERATION  
IfthePWMpinishighoriftheloadcurrentontheconverter  
is high enough to command PWM mode operation with  
PWM low, the LTC3129 operates in a fixed 1.2MHz PWM  
mode using an internally compensated average current  
mode control loop. PWM mode minimizes output voltage  
ripple and yields a low noise switching frequency spec-  
trum. A proprietary switching algorithm provides seam-  
less transitions between operating modes and eliminates  
discontinuities in the average inductor current, inductor  
ripple current and loop transfer function throughout all  
modes of operation. These advantages result in increased  
efficiency,improvedloopstabilityandloweroutputvoltage  
rippleincomparisontothetraditionalbuck-boostconverter.  
TheLTC3129isa1.3µAquiescentcurrent,monolithic,cur-  
rent mode, buck-boost DC/DC converter that can operate  
overawideinputvoltagerangeof1.92Vto15Vandprovide  
up to 200mA to the load. Internal, low R  
N-channel  
DS(ON)  
power switches reduce solution complexity and maximize  
efficiency.Aproprietaryswitchcontrolalgorithmallowsthe  
buck-boostconvertertomaintainoutputvoltageregulation  
with input voltages that are above, below or equal to the  
output voltage. Transitions between the step-up or step-  
downoperatingmodesareseamlessandfreeoftransients  
and sub-harmonic switching, making this product ideal  
for noise sensitive applications. The LTC3129 operates  
at a fixed nominal switching frequency of 1.2MHz, which  
providesanidealtrade-offbetweensmallsolutionsizeand  
high efficiency. Current mode control provides inherent  
input line voltage rejection, simplified compensation and  
rapid response to load transients.  
Figure 1 shows the topology of the LTC3129 power stage  
which is comprised of four N-channel DMOS switches  
and their associated gate drivers. In PWM mode operation  
both switch pins transition on every cycle independent of  
the input and output voltages. In response to the internal  
control loop command, an internal pulse width modulator  
generates the appropriate switch duty cycle to maintain  
regulation of the output voltage.  
Burst Mode capability is also included in the LTC3129 and  
is user-selected via the PWM input pin. In Burst Mode  
operation, the LTC3129 provides exceptional efficiency at  
light output loading conditions by operating the converter  
only when necessary to maintain voltage regulation. The  
BurstModequiescentcurrentisamiserly1.3µA. Athigher  
loads, the LTC3129 automatically switches to fixed fre-  
quencyPWMmodewhenBurstModeoperationisselected.  
(Please refer to the Typical Performance Characteristics  
curves for the mode transition point at different input and  
output voltages.) If the application requires extremely low  
noise, continuous PWM operation can also be selected  
via the PWM pin.  
C
BST1  
C
BST2  
L
BST1  
V
A
SW1  
SW2  
D
V
OUT  
BST2  
IN  
V
CC  
V
CC  
V
CC  
V
CC  
B
C
PGND  
PGND  
A MPPC (maximum power point control) function is also  
provided that allows the input voltage to the converter to  
be servo'd to a programmable point for maximum power  
when operating from various non-ideal power sources  
such as photovoltaic cells. The LTC3129 also features  
an accurate RUN comparator threshold with hysteresis,  
allowing the buck-boost DC/DC converter to turn on and  
LTC3129  
3129 F01  
Figure 1. Power Stage Schematic  
off at user-selected V voltage thresholds. With a wide  
IN  
voltage range, 1.3µA Burst Mode current and program-  
mable RUN and MPPC pins, the LTC3129 is well suited  
for many diverse applications.  
3129f  
10  
For more information www.linear.com/3129  
LTC3129  
operaTion  
When stepping down from a high input voltage to a lower  
output voltage, the converter operates in buck mode and  
switch D remains on for the entire switching cycle except  
fortheminimumswitchlowduration(typically90ns).Dur-  
ing the switch low duration, switch C is turned on which  
outputs are used to control the duty cycle of the switch  
pins on a cycle-by-cycle basis.  
The voltage error amplifier monitors the output voltage,  
V
OUT  
through a voltage divider and makes adjustments to  
thecurrentcommandasnecessarytomaintainregulation.  
The voltage error amplifier therefore controls the outer  
voltage regulation loop. The average current amplifier  
makes adjustments to the inductor current as directed by  
forces SW2 low and charges the flying capacitor, C  
.
BST2  
This ensures that the switch D gate driver power supply  
rail on BST2 is maintained. The duty cycle of switches A  
and B are adjusted to maintain output voltage regulation  
in buck mode.  
the voltage error amplifier output via V and is commonly  
C
referred to as the inner current loop amplifier.  
If the input voltage is lower than the output voltage, the  
converter operates in boost mode. Switch A remains on  
for the entire switching cycle except for the minimum  
switch low duration (typically 90ns). During the switch  
low duration, switch B is turned on which forces SW1  
The average current mode control technique is similar to  
peak current mode control except that the average current  
amplifier, by virtue of its configuration as an integrator,  
controls average current instead of the peak current. This  
difference eliminates the peak to average current error  
inherent to peak current mode control, while maintaining  
most of the advantages inherent to peak current mode  
control.  
low and charges the flying capacitor, C  
. This ensures  
BST1  
that the switch A gate driver power supply rail on BST1 is  
maintained.ThedutycycleofswitchesCandDareadjusted  
to maintain output voltage regulation in boost mode.  
Average current mode control requires appropriate com-  
pensation for the inner current loop, unlike peak current  
mode control. The compensation network must have high  
DC gain to minimize errors between the actual and com-  
manded average current level, high bandwidth to quickly  
change the commanded current level following transient  
load steps and a controlled mid-band gain to provide a  
form of slope compensation unique to average current  
mode control. The compensation components required  
to ensure proper operation have been carefully selected  
and are integrated within the LTC3129.  
Oscillator  
The LTC3129 operates from an internal oscillator with a  
nominalfixedfrequencyof1.2MHz. ThisallowstheDC/DC  
converterefficiencytobemaximizedwhilestillusingsmall  
external components.  
Current Mode Control  
The LTC3129 utilizes average current mode control for the  
pulsewidthmodulator.Currentmodecontrol,bothaverage  
and the better known peak method, enjoy some benefits  
compared to other control methods including: simplified  
loop compensation, rapid response to load transients and  
inherent line voltage rejection.  
Inductor Current Sense and Maximum Output Current  
As part of the current control loop required for current  
mode control, the LTC3129 includes a pair of current  
sensing circuits that measure the buck-boost converter  
inductor current.  
Referring to the Block Diagram, a high gain, internally  
compensated transconductance amplifier monitors Vout  
through a voltage divider connected to the FB pin. The  
error amplifier output is used by the current mode control  
loop to command the appropriate inductor current level.  
The inverting input of the internally compensated average  
current amplifier is connected to the inductor current  
sense circuit. The average current amplifier's output is  
compared to the oscillator ramps, and the comparator  
Thevoltageerroramplifieroutput,V ,isinternallyclamped  
C
to a nominal level of 0.6V. Since the average inductor  
current is proportional to V , the 0.6V clamp level sets  
C
the maximum average inductor current that can be pro-  
grammed by the inner current loop. Taking into account  
the current sense amplifier's gain, the maximum average  
3129f  
11  
For more information www.linear.com/3129  
LTC3129  
operaTion  
inductor current is approximately 275mA (typical). In  
Overload Current Limit and I  
Comparator  
ZERO  
Buck mode, the output current is approximately equal to  
The internal current sense waveform is also used by the  
peakoverloadcurrent(I )andzerocurrent(I )com-  
the inductor current I .  
L
PEAK  
ZERO  
I
≈ I • 0.89  
parators. The I  
current comparator monitors Isense  
OUT(BUCK)  
L
PEAK  
and turns off switch A if the inductor current level exceeds  
its maximum internal threshold, which is approximately  
500mA. An inductor current level of this magnitude will  
occur during a fault, such as an output short-circuit, or  
during large load or input voltage transients.  
The 90ns SW1/SW2 forced low time on each switching  
cycle briefly disconnects the inductor from V and V  
resulting in about 11% less output current in either buck  
orboostmodeforagiveninductorcurrent.Inboostmode,  
the output current is related to average inductor current  
and duty cycle by:  
OUT  
IN  
TheLTC3129featuresneardiscontinuousinductorcurrent  
operation at light output loads by virtue of the I  
com-  
ZERO  
I
≈ I • (1 – D) • Efficiency,  
L
OUT(BOOST)  
parator circuit. By limiting the reverse current magnitude  
in PWM mode, a balance between low noise operation and  
where D is the converter duty cycle.  
improved efficiency at light loads is achieved. The I  
ZERO  
Since the output current in boost mode is reduced by the  
duty cycle (D), the output current rating in buck mode is  
always greater than in boost mode. Also, because boost  
mode operation requires a higher inductor current for a  
givenoutputcurrentcomparedtobuckmode,theefficiency  
comparator threshold is set near the zero current level in  
PWMmode,andasaresult,thereversecurrentmagnitude  
will be a function of inductance value and output voltage  
due to the comparator's propagation delay. In general,  
higher output voltages and lower inductor values will  
result in increased reverse current magnitude.  
in boost mode will be lower due to higher I ² • R  
L
DS(ON)  
losses in the power switches. This will further reduce the  
outputcurrentcapabilityinboostmode.Ineitheroperating  
mode, however, the inductor peak-to-peak ripple current  
does not play a major role in determining the output cur-  
rent capability, unlike peak current mode control.  
In automatic Burst Mode operation (PWM pin low), the  
I
comparator threshold is increased so that reverse  
ZERO  
inductor current does not normally occur. This maximizes  
efficiency at very light loads.  
With peak current mode control, the maximum output  
current capability is reduced by the magnitude of inductor  
ripplecurrentbecausethepeakinductorcurrentlevelisthe  
control variable, but the average inductor current is what  
determines the output current. The LTC3129 measures  
and controls average inductor current, and therefore, the  
inductor ripple current magnitude has little effect on the  
maximum current capability in contrast to an equivalent  
peak current mode converter. Under most conditions in  
buck mode, the LTC3129 is capable of providing a mini-  
mum of 200mA to the load. In boost mode, as described  
previously, the output current capability is related to the  
Burst Mode OPERATION  
When the PWM pin is held low, the LTC3129 is configured  
for automatic Burst Mode operation. As a result, the buck-  
boost DC/DC converter will operate with normal continu-  
ous PWM switching above a predetermined minimum  
output load and will automatically transition to power  
saving Burst Mode operation below this output load level.  
Note that if the PWM pin is low, reverse inductor current is  
not allowed at any load. Refer to the Typical Performance  
Characteristics section to determine the Burst Mode  
transition threshold for various combinations of V and  
IN  
boost ratio or duty cycle (D). For example, for a 3.6V V  
IN  
V
. If PWM is low, at light output loads, the LTC3129  
OUT  
to 5V output application, the LTC3129 can provide up  
to 150mA to the load. Refer to the Typical Performance  
characteristics section for more detail on output current  
capability.  
will go into a standby or sleep state when the output volt-  
age achieves its nominal regulation level. The sleep state  
halts PWM switching and powers down all non-essential  
3129f  
12  
For more information www.linear.com/3129  
LTC3129  
operaTion  
functions of the IC, significantly reducing the quiescent  
current of the LTC3129 to just 1.3µA typical. This greatly  
improves overall power conversion efficiency when the  
output load is light. Since the converter is not operating  
in sleep, the output voltage will slowly decay at a rate  
determined by the output load resistance and the output  
capacitor value. When the output voltage has decayed by  
a small amount, typically 1%, the LTC3129 will wake and  
resume normal PWM switching operation until the volt-  
logic threshold. The V regulator includes current-limit  
CC  
protectiontosafeguardagainstaccidentalshort-circuiting  
of the V rail.  
CC  
Undervoltage Lockout (UVLO)  
Therearetwoundervoltagelockout(UVLO)circuitswithin  
the LTC3129 that inhibit switching; one that monitors V  
IN  
and another that monitors V . Either UVLO will disable  
CC  
operation of the internal power switches and keep other  
age on V  
is restored to the previous level. If the load  
OUT  
IC functions in a reset state if either V or V are below  
IN  
CC  
is very light, the LTC3129 may only need to switch for a  
few cycles to restore V and may sleep for extended  
their respective UVLO thresholds.  
OUT  
The V UVLO comparator has a falling voltage threshold  
IN  
periods of time, significantly improving efficiency. If the  
load is suddenly increased above the burst transition  
threshold, the part will automatically resume continuous  
PWM operation until the load is once again reduced.  
of 1.8V (typical). If V falls below this level, IC operation  
IN  
is disabled until V rises above 1.9V (typical), as long as  
IN  
the V voltage is above its UVLO threshold.  
CC  
The V UVLO has a falling voltage threshold of 2.19V  
CC  
A feedforward capacitor on the feedback divider can be  
(typical). If the V voltage falls below this threshold, IC  
CC  
used to reduce Burst Mode V  
ripple. This is discussed  
OUT  
operation is disabled until V rises above 2.25V (typical)  
CC  
in more detail in the Applications Information section of  
this data sheet.  
as long as V is above its nominal UVLO threshold level.  
IN  
Depending on the particular application, either of these  
UVLO thresholds could be the limiting factor affecting the  
minimuminputvoltagerequiredforoperation.Becausethe  
Note that Burst Mode operation is inhibited until soft-start  
is done, the MPPC pin is greater than 1.175V and V  
has reached regulation.  
OUT  
V
regulator uses V for its power input, the minimum  
CC  
IN  
input voltage required for operation is determined by the  
Soft-Start  
V
minimum voltage, as input voltage (V ) will always  
CC  
IN  
The LTC3129 soft-start circuit minimizes input current  
transientsandoutputvoltageovershootoninitialpowerup.  
The required timing components for soft-start are internal  
to the LTC3129 and produce a nominal soft-start dura-  
tion of approximately 3ms. The internal soft-start circuit  
be higher than V in the normal (non-bootstrapped)  
CC  
configuration. Therefore, the minimum V for the part  
to startup is 2.25V (typical).  
IN  
In applications where V is bootstrapped (powered  
CC  
through a Schottky diode by either V  
or an auxiliary  
slowly ramps the error amplifier output, V . In doing so,  
OUT  
C
power rail), the minimum input voltage for operation will  
the current command of the IC is also slowly increased,  
starting from zero. It is unaffected by output loading or  
output capacitor value. Soft-start is reset by the UVLO on  
both V and V , the RUN pin and thermal shutdown.  
be limited only by the V UVLO threshold (1.8V typical).  
IN  
Please note that if the bootstrap voltage is derived from  
theLTC3129V  
andnotanindependentpowerrail, then  
OUT  
IN  
CC  
the minimum input voltage required for initial startup is  
still 2.25V (typical).  
V
Regulator  
CC  
Aninternallowdropoutregulator(LDO)generatesanomi-  
nal 4.1V V rail from V . The V rail powers the internal  
Note that if either V or V are below their UVLO thresh-  
IN CC  
olds, or if RUN is below its accurate threshold of 1.22V  
(typical), then the LTC3129 will remain in a soft shutdown  
CC  
IN  
CC  
control circuitry and the gate drivers of the LTC3129. The  
regulator is disabled in shutdown to reduce quiescent  
V
state, where the V quiescent current will be only 1.9µA  
CC  
IN  
current and is enabled by raising the RUN pin above its  
typical.  
3129f  
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LTC3129  
operaTion  
V
OUT  
Undervoltage  
With the addition of an optional resistor divider as shown  
in Figure 2, the RUN pin can be used to establish a user-  
programmableturn-onandturn-offthreshold.Thisfeature  
can be utilized to minimize battery drain below a certain  
input voltage, or to operate the converter in a hiccup mode  
from very low current sources.  
There is also an undervoltage comparator that monitors  
the output voltage. Until V reaches 1.15V (typical), the  
OUT  
average current limit is reduced by a factor of two. This  
reduces power dissipation in the device in the event of a  
shorted output. In addition, N-channel switch D, which  
feeds V , will be disabled until V  
exceeds 1.15V.  
OUT  
OUT  
LTC3129  
V
ACCURATE THRESHOLD  
IN  
RUN Pin Comparator  
1.22V  
–
+
ENABLE SWITCHING  
R3  
RUN  
In addition to serving as a logic level input to enable cer-  
tain functions of the IC, the RUN pin includes an accurate  
internal comparator that allows it to be used to set custom  
rising and falling ON/OFF thresholds with the addition of  
an optional external resistor divider. When RUN is driven  
+
–
R4  
ENABLE LDO AND  
CONTROL CIRCUITS  
0.9V  
LOGIC THRESHOLD  
above its logic threshold (0.9V typical), the V regulator  
CC  
3129 F02  
is enabled, which provides power to the internal control  
circuitryoftheIC.IfthevoltageonRUNisincreasedfurther  
so that it exceeds the RUN comparator's accurate analog  
threshold (1.22V typical), all functions of the buck-boost  
converterwillbeenabledandastart-upsequencewillensue,  
Figure 2. Accurate RUN Pin Comparator  
Note that once RUN is above 0.9V typical, the quiescent  
input current on V (or V if back-driven) will increase to  
IN  
CC  
about 1.9µA typical until the V and V UVLO thresholds  
IN  
CC  
assuming the V and V UVLO thresholds are satisfied.  
IN  
CC  
are satisfied.  
IfRUNisbroughtbelowtheaccuratecomparatorthreshold,  
thebuck-boostconverterwillinhibitswitching,buttheV  
TheconverterisenabledwhenthevoltageonRUNexceeds  
1.22V (nominal). Therefore, the turn-on voltage threshold  
on V is given by:  
CC  
regulator and control circuitry will remain powered unless  
RUN is brought below its logic threshold. Therefore, in  
order to completely shut down the IC and reduce the Vin  
current to 10nA (typical), it is necessary to ensure that  
RUNisbroughtbelowitsworstcaselowlogicthresholdof  
0.5V. RUN is a high voltage input and can be tied directly  
IN  
V
= 1.22V • (1 + R3/R4)  
IN(TURN-ON)  
The RUN comparator includes a built-in hysteresis of  
approximately 80mV, so that the turn off threshold will  
be 1.14V.  
to V to continuously enable the IC when the input supply  
IN  
There may be cases due to PCB layout, very large value  
resistorsforR3andR4, orproximitytonoisycomponents  
wherenoisepickupmaycausetheturn-onorturn-offofthe  
IC to be intermittent. In these cases, a small filter capaci-  
tor can be added across R4 to ensure proper operation.  
is present. Also note that RUN can be driven above V  
IN  
or V  
as long as it stays within the operating range of  
OUT  
the IC (up to 15V).  
3129f  
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LTC3129  
operaTion  
PGOOD Comparator  
a minimum input voltage when using high resistance  
sources, such as photovoltaic panels, so as to maximize  
The LTC3129 provides an open-drain PGOOD output that  
input power transfer and prevent V from dropping too  
IN  
pulls low if V  
falls more than 7.5% (typical) below its  
OUT  
low under load. Referring to Figure 3, the MPPC pin is  
programmedvalue.WhenV risestowithin5%(typical)  
OUT  
internally connected to the non-inverting input of a g  
m
of its programmed value, the internal PGOOD pull-down  
will turn off and PGOOD will go high if an external pull-  
up resistor has been provided. An internal filter prevents  
amplifier,whoseinvertinginputisconnectedtothe1.175V  
reference. If the voltage at MPPC, using the external volt-  
age divider, falls below the reference voltage, the output of  
nuisance trips of PGOOD due to short transients on V  
.
OUT  
the amplifier pulls the internal V node low. This reduces  
C
Note that PGOOD can be pulled up to any voltage, as long  
as the absolute maximum rating of 18V is not exceeded,  
and as long as the maximum sink current rating is not  
exceeded when PGOOD is low. Note that PGOOD will  
the commanded average inductor current so as to reduce  
the input current and regulate V to the programmed  
IN  
minimum voltage, as given by:  
also be driven low if V is below its UVLO threshold or  
V
= 1.175V • (1 + R5/R6)  
CC  
IN(MPPC)  
if the part is in shutdown (RUN below its logic threshold)  
Note that external compensation should not be required  
while V is being held up (or back-driven). PGOOD is  
CC  
for MPPC loop stability if input filter capacitor, C , is at  
IN  
not affected by V UVLO or the accurate RUN threshold.  
IN  
least 22µF.  
Maximum Power-Point Control (MPPC)  
The divider resistor values can be in the MΩ range to  
minimize the input current in very low power applications.  
However, straycapacitanceandnoisepickupontheMPPC  
pin must also be minimized.  
The MPPC input of the LTC3129 can be used with an  
optional external voltage divider to dynamically adjust  
the commanded inductor current in order to maintain  
*C  
IN  
V
IN  
R
R5  
R6  
LTC3129  
S
MPPC  
+
–
+
V
SOURCE  
–
1.175V  
+
–
V
* C SHOULD BE AT  
IN  
C
FB  
CURRENT  
LEAST 22µF FOR  
COMMAND  
MPPC APPLICATIONS  
VOLTAGE  
ERROR AMP  
3129 F03  
Figure 3. MPPC Amplifier with External Resistor Divider  
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LTC3129  
operaTion  
The MPPC pin controls the converter in a linear fashion  
when using sources that can provide a minimum of 5mA  
to 10mA of continuous input current. For operation from  
weaker input sources, refer to the Applications Informa-  
tion section to see how the programmable RUN pin can  
be used to control the converter in a hysteretic manner to  
provide an effective MPPC function for sources that can  
provide as little as 5µA or less. If the MPPC function is not  
and significantly improve efficiency. As a result, careful  
consideration must be given to the thermal environment  
of the IC in order to provide a means to remove heat from  
the IC and ensure that the LTC3129 is able to provide its  
full rated output current. Specifically, the exposed die  
attach pad of both the QFN and MSE packages must be  
soldered to a copper layer on the PCB to maximize the  
conduction of heat out of the IC package. This can be ac-  
complished by utilizing multiple vias from the die attach  
pad connection underneath the IC package to other PCB  
layer(s) containing a large copper plane. A typical board  
layout incorporating these concepts is shown in Figure 4.  
required, the MPPC pin should be tied to V .  
CC  
Thermal Considerations  
The power switches of the LTC3129 are designed to oper-  
ate continuously with currents up to the internal current  
limit thresholds. However, when operating at high current  
levels, there may be significant heat generated within the  
IftheICdietemperatureexceedsapproximately180°C,over  
temperature shutdown will be invoked and all switching  
will be inhibited. The part will remain disabled until the die  
temperature cools by approximately 10°C. The soft-start  
circuit is re-initialized in overtemperature shutdown to  
provide a smooth recovery when the IC die temperature  
cools enough to resume operation.  
IC. In addition, the V regulator can also generate wasted  
CC  
heat when V is very high, adding to the total power  
IN  
dissipation of the IC. As described elsewhere in this data  
sheet, bootstrapping of the V for 5V output applications  
CC  
CC  
can essentially eliminate the V power dissipation term  
GND  
V
IN  
C
IN  
V
CC  
L
C
OUT  
3129 F04  
GND  
V
OUT  
Figure 4. Typical 2-Layer PC Board Layout (MSE Package)  
3129f  
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LTC3129  
applicaTions inForMaTion  
A standard application circuit for the LTC3129 is shown on  
the front page of this data sheet. The appropriate selection  
of external components is dependent upon the required  
performance of the IC in each particular application given  
considerations and trade-offs such as PCB area, input  
and output voltage range, output voltage ripple, transient  
response, required efficiency, thermal considerations and  
cost. This section of the data sheet provides some basic  
guidelines and considerations to aid in the selection of  
external components and the design of the applications  
circuit, as well as more application circuit examples.  
sible. V is the regulator output and is also the internal  
CC  
supply pin for the LTC3129 control circuitry as well as the  
gate drivers and boost rail charging diodes. The V pin is  
CC  
not intended to supply current to other external circuitry.  
Inductor Selection  
ThechoiceofinductorusedinLTC3129applicationcircuits  
influences the maximum deliverable output current, the  
converterbandwidth,themagnitudeoftheinductorcurrent  
ripple and the overall converter efficiency. The inductor  
must have a low DC series resistance, when compared to  
the internal switch resistance, or output current capabil-  
ity and efficiency will be compromised. Larger inductor  
valuesreduceinductorcurrentripplebutmaynotincrease  
output current capability as is the case with peak current  
mode control as described in the Maximum Output Cur-  
rent section. Larger value inductors also tend to have a  
higher DC series resistance for a given case size, which  
will have a negative impact on efficiency. Larger values  
of inductance will also lower the right half plane (RHP)  
zero frequency when operating in boost mode, which can  
compromise loop stability. Nearly all LTC3129 application  
circuitsdeliverthebestperformancewithaninductorvalue  
between 3.3µH and 10µH. Buck mode-only applications  
can use the larger inductor values as they are unaffected  
bytheRHPzero,whilemostlyboostapplicationsgenerally  
require inductance on the low end of this range depending  
on how large the step-up ratio is.  
Programming V  
OUT  
The output voltage of the LTC3129 is set by connecting  
the FB pin to an external resistor divider from V to  
ground, as shown in Figure 5, according to the equation:  
OUT  
V
= 1.175V • (1+ R1/R2)  
OUT  
V
OUT  
V
OUT  
LTC3129  
C
OUT  
R1  
R2  
C
FF  
FB  
3129 F05  
Figure 5. VOUT Feedback Divider  
Asmallfeedforwardcapacitorcanbeaddedinparallelwith  
R1 (in Figure 5) to reduce Burst Mode ripple and improve  
transient response. Details on selecting a feedforward  
capacitor are provided later in this data sheet.  
Regardless of inductor value, the saturation current rating  
shouldbeselectedsuchthatitisgreaterthantheworst-case  
averageinductorcurrentplushalfoftheripplecurrent.The  
peak-to-peak inductor current ripple for each operational  
modecanbecalculatedfromthefollowingformula, where  
f is the switching frequency (1.2MHz), L is the inductance  
V
Capacitor Selection  
CC  
in µH and t  
is the switch pin minimum low time in  
LOW  
The V output of the LTC3129 is generated from V by a  
CC  
IN  
µs. The switch pin minimum low time is typically 0.09µs.  
low dropout linear regulator. The V regulator has been  
CC  
designed for stable operation with a wide range of output  
capacitors. For most applications, a low ESR capacitor of  
at least 2.2µF should be used. The capacitor should be  
VOUT V – V  
1
f
IN  
OUT  
∆IL(P−P)(BUCK)  
=
– t  
A
LOW   
L
V
IN  
located as close to the V pin as possible and connected  
V
L
VOUT – V  
1
CC  
IN  
IN   
∆IL(P−P)(BOOST)  
=
– t  
A
f
LOW   
totheV pinandgroundthroughtheshortesttracespos-  
CC  
VOUT  
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It should be noted that the worst-case peak-to-peak in-  
Differentinductorcorematerialsandstyleshaveanimpact  
on the size and price of an inductor at any given current  
rating. Shielded construction is generally preferred as it  
minimizes the chances of interference with other circuitry.  
Thechoiceofinductorstyledependsupontheprice,sizing,  
and EMI requirements of a particular application. Table 1  
provides a wide sampling of inductors that are well suited  
to many LTC3129 applications.  
ductor ripple current occurs when the duty cycle in buck  
mode is minimum (highest V ) and in boost mode when  
IN  
the duty cycle is 50% (V  
IN  
= 2 • V ). As an example, if  
OUT  
IN  
V
(minimum) = 2.5V and V (maximum) = 15V, V  
IN OUT  
= 5V and L = 10µH, the peak-to-peak inductor ripples at  
the voltage extremes (15V V for buck and 2.5V V for  
IN  
IN  
boost) are:  
Buck = 248mA peak-to-peak  
Boost = 93mA peak-to-peak  
Table 1. Recommended Inductors  
VENDOR  
PART  
Coilcraft  
www.coilcraft.com  
EPL2014, EPL3012, EPL3015, LPS3015,  
LPS3314, XFL3012  
One half of this inductor ripple current must be added to  
the highest expected average inductor current in order to  
select the proper saturation current rating for the inductor.  
Coiltronics  
www.cooperindustries.com  
SDH3812, SD3814, SD3114, SD3118  
Murata  
www.murata.com  
LQH3NP, LQH32P, LQH44P  
To avoid the possibility of inductor saturation during load  
transients, an inductor with a saturation current rating of  
at least 600mA is recommended for all applications.  
Sumida  
www.sumida.com  
CDRH2D16, CDRH2D18, CDRH3D14,  
CDRH3D16  
Taiyo-Yuden  
www.t-yuden.com  
NR3012T, NR3015T, NRS4012T,  
BRC2518  
Inadditiontoitsinfluenceonpowerconversionefficiency,  
the inductor DC resistance can also impact the maximum  
output current capability of the buck-boost converter  
particularly at low input voltages. In buck mode, the  
output current of the buck-boost converter is primarily  
limited by the inductor current reaching the average cur-  
rent limit threshold. However, in boost mode, especially  
at large step-up ratios, the output current capability can  
also be limited by the total resistive losses in the power  
stage. These losses include, switch resistances, inductor  
DC resistance and PCB trace resistance. Avoid inductors  
with a high DC resistance (DCR) as they can degrade the  
maximum output current capability from what is shown  
in the Typical Performance Characteristics section and  
from the Typical Application circuits.  
TDK  
www.tdk.com  
VLS3012, VLS3015, VLF302510MT,  
VLF302512MT  
Toko  
www.tokoam.com  
DB3015C, DB3018C, DB3020C, DP418C,  
DP420C, DEM2815C, DFE322512C,  
DFE252012C  
Würth  
www.we-online.com  
WE-TPC 2813, WE-TPC 3816,  
WE-TPC 2828  
Recommended inductor values for different operating  
voltage ranges are given in Table 2. These values were  
chosen to minimize inductor size while maintaining an  
acceptable amount of inductor ripple current for a given  
V and V  
range.  
IN  
OUT  
Table 2. Recommended Inductor Values  
V
V
V
V
V
AND V  
RANGE  
RECOMMENDED INDUCTOR VALUES  
3.3µH to 4.7µH  
IN  
IN  
IN  
IN  
IN  
OUT  
and V  
and V  
and V  
and V  
Both < 4.5V  
Both < 8V  
OUT  
OUT  
OUT  
OUT  
As a guideline, the inductor DCR should be significantly  
less than the typical power switch resistance of 750mΩ  
each. The only exceptions are applications that have a  
maximumoutputcurrentrequirementmuchlessthanwhat  
the LTC3129 is capable of delivering. Generally speaking,  
inductors with a DCR in the range of 0.15Ω to 0.3Ω are  
recommended. Lower values of DCR will improve the ef-  
ficiency at the expense of size, while higher DCR values  
will reduce efficiency (typically by a few percent) while  
allowing the use of a physically smaller inductor.  
4.7µH to 6.8µH  
Both < 11V  
Up to 15.75V  
6.8µH to 8.2µH  
8.2µH to 10µH  
Feedforward Capacitor  
The use of a voltage feedforward capacitor, as shown in  
Figure 5, offers a number of performance advantages. A  
feedforward capacitor will reduce output voltage ripple in  
3129f  
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LTC3129  
applicaTions inForMaTion  
Examining the previous equations reveals that the output  
voltage ripple increases with load current and is gener-  
ally higher in boost mode than in buck mode. Note that  
these equations only take into account the voltage ripple  
that occurs from the inductor current to the output being  
discontinuous. They provide a good approximation to the  
rippleatanysignificantloadcurrentbutunderestimatethe  
output voltage ripple at very light loads where the output  
voltage ripple is dominated by the inductor current ripple.  
Burst Mode operation and improve transient response. In  
addition, due to the wide V and V  
operating range  
IN  
OUT  
of the LTC3129 and its fixed internal loop compensation,  
some applications may require the use of a feedforward  
capacitor to assure light-load stability (less than ~15mA)  
when operating in PWM mode (PWM pin pulled high).  
Therefore, to reduce Burst Mode ripple and improve  
phase margin at light load when PWM mode operation is  
selected, a feedforward capacitor is recommended for all  
applications. The recommended feedforward capacitor  
value can be calculated by:  
In addition to the output voltage ripple generated across  
the output capacitance, there is also output voltage ripple  
produced across the internal resistance of the output  
capacitor. The ESR-generated output voltage ripple is  
proportionaltotheseriesresistanceoftheoutputcapacitor  
C = 66/R1  
FF  
Where R1 is the top feedback divider resistor value in MΩ  
and C is the recommended feedforward capacitor value  
and is given by the following expressions where R  
is  
FF  
ESR  
in picofarads (use the nearest standard value). Refer to  
the series resistance of the output capacitor and all other  
terms as previously defined.  
the application circuits for examples.  
ILOAD ESR  
R
Output Capacitor Selection  
∆V  
=
≅ ILOAD ESR  
R
V
P−P(BUCK)  
1– tLOW  
f
A low effective series resistance (ESR) output capacitor  
of 4.7µF minimum should be connected at the output of  
the buck-boost converter in order to minimize output volt-  
age ripple. Multilayer ceramic capacitors are an excellent  
option as they have low ESR and are available in small  
footprints. The capacitor value should be chosen large  
enough to reduce the output voltage ripple to acceptable  
levels. Neglecting the capacitor's ESR and ESL (effec-  
tive series inductance), the peak-to-peak output voltage  
ripple in PWM mode can be calculated by the following  
ILOAD ESR OUT  
R
V
∆V  
=
P−P(BOOST)  
V 1– t  
f
(
)
IN  
LOW  
VOUT  
V
≅ ILOAD ESR  
R
V
IN  
InmostLTC3129applications,anoutputcapacitorbetween  
10µF and 22µF will work well. To minimize output ripple  
in Burst Mode operation, or transients incurred by large  
step loads, values of 22µF or larger are recommended.  
formula, where f is the frequency in MHz (1.2MHz), C  
OUT  
is the capacitance in µF, t  
is the switch pin minimum  
LOW  
low time in µs (0.09µs typical) and I  
current in amperes.  
is the output  
LOAD  
Input Capacitor Selection  
The V pin carries the full inductor current and provides  
IN  
ILOAD LOW  
t
power to internal control circuits in the IC. To minimize  
input voltage ripple and ensure proper operation of the IC,  
a low ESR bypass capacitor with a value of at least 4.7µF  
∆V  
=
V
P−P(BUCK)  
COUT  
ILOAD  
V
– VIN + tLOWfV  
IN   
OUT  
should be located as close to the V pin as possible. The  
IN  
∆V  
=
V
P−P(BOOST)  
fC  
OUT   
VOUT  
traces connecting this capacitor to V and the ground  
IN  
plane should be made as short as possible.  
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LTC3129  
applicaTions inForMaTion  
Whenpoweredthroughlongleadsorfromapowersource  
with significant resistance, a larger value bulk input ca-  
pacitor may be required and is generally recommended.  
Insuchapplications, a47µFto100µFlow-ESRelectrolytic  
capacitorinparallelwitha1µFceramiccapacitorgenerally  
yields a high performance, low cost solution.  
of its rated capacitance when operated at even half of its  
maximum rated voltage. This effect is generally reduced  
as the case size is increased for the same nominal value  
capacitor. As a result, it is often necessary to use a larger  
value capacitance or a higher voltage rated capacitor than  
wouldordinarilyberequiredtoactuallyrealizetheintended  
capacitanceattheoperatingvoltageoftheapplication.X5R  
and X7R dielectric types are recommended as they exhibit  
the best performance over the wide operating range and  
temperature of the LTC3129. To verify that the intended  
capacitance is achieved in the application circuit, be sure  
to consult the capacitor vendor's curve of capacitance  
versus DC bias voltage.  
Note that applications using the MPPC feature should  
use a minimum C of 22µF. Larger values can be used  
IN  
without limitation.  
Recommended Input and Output Capacitor Types  
The capacitors used to filter the input and output of the  
LTC3129 must have low ESR and must be rated to handle  
the AC currents generated by the switching converter.  
This is important to maintain proper functioning of the  
IC and to reduce output voltage ripple. There are many  
capacitor types that are well suited to these applications  
including multilayer ceramic, low ESR tantalum, OS-CON  
and POSCAP technologies. In addition, there are certain  
types of electrolytic capacitors such as solid aluminum  
organic polymer capacitors that are designed for low  
ESR and high AC currents and these are also well suited  
to some LTC3129 applications. The choice of capacitor  
technology is primarily dictated by a trade-off between  
size, leakage current and cost. In backup power applica-  
tions, the input or output capacitor might be a super or  
ultra capacitor with a capacitance value measuring in the  
Farad range. The selection criteria in these applications  
are generally similar except that voltage ripple is generally  
not a concern. Some capacitors exhibit a high DC leak-  
age current which may preclude their consideration for  
applications that require a very low quiescent current in  
BurstModeoperation. Notethatultracapacitorsmayhave  
a rather high ESR, therefore a 4.7µF (minimum) ceramic  
capacitor is recommended in parallel, close to the IC pins.  
Using the Programmable RUN Function to Operate  
from Extremely Weak Input Sources  
Another application of the programmable RUN pin is that  
it can be used to operate the converter in a hiccup mode  
fromextremelylowcurrentsources.Thisallowsoperation  
from sources that can only generate microamps of output  
current,andwouldbefartooweaktosustainnormalsteady-  
stateoperation,evenwiththeuseoftheMPPCpin.Because  
until it is  
the LTC3129 draws only 1.9µA typical from V  
IN  
enabled, the RUN pin can be programmed to keep the IC  
disabled until V reaches the programmed voltage level.  
IN  
Inthismanner,theinputsourcecantrickle-chargeaninput  
storage capacitor, even if it can only supply microamps of  
current, until V reaches the turn-on threshold set by the  
IN  
RUN pin divider. The converter will then be enabled using  
the stored charge in the input capacitor, until Vin drops  
below the turn-off threshold, at which point the converter  
will turn off and the process will repeat.  
This approach allows the converter to run from weak  
sources such as thin-film solar cells using indoor light-  
ing. Although the converter will be operating in bursts,  
it is enough to charge an output capacitor to power low  
duty cycle loads, such as wireless sensor applications,  
or to trickle charge a battery. In addition, note that the  
input voltage will be cycling (with a small ripple as set by  
the RUN hysteresis) about a fixed voltage, as determined  
by the divider. This allows the high impedance source to  
operate at the programmed optimal voltage for maximum  
power transfer.  
Ceramic capacitors are often utilized in switching con-  
verter applications due to their small size, low ESR and  
low leakage currents. However, many ceramic capacitors  
intended for power applications experience a significant  
loss in capacitance from their rated value as the DC bias  
voltage on the capacitor increases. It is not uncommon for  
a small surface mount capacitor to lose more than 50%  
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applicaTions inForMaTion  
When using high value divider resistors (in the MΩ  
MPPC Compensation and Gain  
range) to minimize current draw on V , a small noise  
IN  
When using MPPC, there are a number of variables that  
affect the gain and phase of the input voltage control  
loop. Primarily these are the input capacitance, the MPPC  
filter capacitor may be necessary across the lower divider  
resistor to prevent noise from erroneously tripping the  
RUNcomparator.Thecapacitorvalueshouldbeminimized  
so as not to introduce a time delay long enough for the  
divider ratio and the V source resistance (or current). To  
IN  
simplify the design of the application circuit, the MPPC  
control loop in the LTC3129 is designed with a relatively  
low gain, such that external MPPC loop compensation is  
input voltage to drop significantly below the desired V  
IN  
threshold before the converter is turned off. Note that  
larger V decoupling capacitor values will minimize this  
IN  
generally not required when using a V capacitor value  
IN  
effect by providing more holdup time on V .  
IN  
of at least 22µF. The gain from the MPPC pin to the in-  
ternal VC control voltage is about 12, so a drop of 50mV  
on the MPPC pin (below the 1.175V MPPC threshold),  
corresponds to a 600mV drop on the internal VC voltage,  
which reduces the average inductor current all the way  
to zero. Therefore, the programmed input MPPC voltage  
will be maintained within about 4% over the load range.  
Programming the MPPC Voltage  
As discussed in the previous section, the LTC3129 in-  
cludes an MPPC function to optimize performance when  
operatingfromvoltagesourceswithrelativelyhighsource  
resistance. Using an external voltage divider from V , the  
IN  
MPPCfunctiontakescontroloftheaverageinductorcurrent  
when necessary to maintain a minimum input voltage, as  
programmed by the user. Referring to Figure 3:  
Note that if large value V capacitors are used (which may  
IN  
have a relatively high ESR) a small ceramic capacitor of  
at least 4.7µF should be placed in parallel across the V  
IN  
V
= 1.175V • (1 + R5/R6)  
input, near the V pin of the IC.  
IN(MPPC)  
IN  
This is useful for such applications as photovoltaic  
powered converters, since the maximum power transfer  
point occurs when the photovoltaic panel is operated at  
about 75% of its open-circuit voltage. For example, when  
operating from a photovoltaic panel with an open-circuit  
voltage of 5V, the maximum power transfer point will be  
when the panel is loaded such that its output voltage is  
about 3.75V. Choosing values of 2MΩ for R5 and 909kΩ  
for R6 will program the MPPC function to regulate the  
Bootstrapping the V Regulator  
CC  
The high and low side gate drivers are powered through  
theV rail,whichisgeneratedfromtheinputvoltage,V ,  
CC  
IN  
through an internal linear regulator. In some applications,  
especially at high input voltages, the power dissipation  
in the linear regulator can become a major contributor to  
thermalheatingoftheICandoverallefficiency. TheTypical  
Performance Characteristics section provides data on the  
maximuminputcurrentsoastomaintainV ataminimum  
V
CC  
current and resulting power loss versus V and V  
.
IN  
IN  
OUT  
of 3.74V (typical). Note that if the panel can provide more  
power than the LTC3129 can draw, the input voltage will  
rise above the programmed MPPC point. This is fine as  
long as the input voltage doesn't exceed 15V.  
Asignificantperformanceadvantagecanbeattainedinhigh  
V applications where converter output voltage (V ) is  
IN  
OUT  
programmed to 5V, if V  
is used to power the V rail.  
OUT  
CC  
Powering V in this manner is referred to as bootstrap-  
CC  
For weak input sources with very high resistance (hun-  
dreds of Ohms or more), the LTC3129 may still draw more  
ping. This can be done by connecting a Schottky diode  
(such as a BAT54) from V  
to V as shown in Figure 6.  
OUT  
CC  
current than the source can provide, causing V to drop  
Withthebootstrapdiodeinstalled, thegatedrivercurrents  
aresuppliedbythebuck-boostconverterathighefficiency  
rather than through the internal linear regulator. The in-  
ternal linear regulator contains reverse blocking circuitry  
IN  
below the UVLO threshold. For these applications, it is  
recommended that the programmable RUN feature be  
used, as described in the previous section.  
3129f  
21  
For more information www.linear.com/3129  
LTC3129  
applicaTions inForMaTion  
that allows V to be driven above its nominal regulation  
Sources of Small Photovoltaic Panels  
CC  
level with only a very slight amount of reverse current.  
A list of companies that manufacture small solar panels  
(sometimes referred to as modules or solar cell arrays)  
suitable for use with the LTC3129 is provided in Table 3.  
Please note that the bootstrapping supply (either V  
or  
OUT  
a separate regulator) must be limited to less than 5.7V so  
as not to exceed the maximum V voltage of 5.5V after  
CC  
Table 3. Small Photovoltaic Panel Manufacturers  
the diode drop.  
Sanyo  
http://panasonic.net/energy/amorton/en/  
http://www.powerfilmsolar.com/  
By maintaining V above its UVLO threshold, bootstrap-  
CC  
PowerFilm  
ping, even to a 3.3V output, also allows operation down  
Ixys  
Corporation  
http://www.ixys.com/ProductPortfolio/GreenEnergy.aspx  
to the V UVLO threshold of 1.8V (typical).  
IN  
G24  
Innovations  
http://www.g24i.com/  
V
OUT  
V
OUT  
SolarPrint  
http://www.solarprint.ie/  
LTC3129  
C
OUT  
BAT54  
V
CC  
2.2µF  
3129 F06  
Figure 6. Example of VCC Bootstrap  
3129f  
22  
For more information www.linear.com/3129  
LTC3129  
Typical applicaTions  
Hiccup Converter Powers Wireless Sensor from Indoor Lighting  
Transmit Rate vs Light Level  
(Fluorescent)  
22nF  
22nF  
4.7µH  
4.5  
4.0  
3.5  
3.0  
2.5  
2.0  
1.5  
1.0  
PULSED I  
OUT  
25mA FOR 5ms  
BST1 SW1  
SW2 BST2  
UVLO = 3.5V  
V
IN  
V
OUT  
V
V
OUT  
IN  
3.6V  
4.7µF  
22µF  
2M  
LTC3129  
4.42M  
1M  
RUN  
+
470µF  
6.3V  
FB  
976k  
V
MPPC  
PWM  
CC  
PGOOD  
PGOOD  
PV PANEL  
SANYO AM-1815  
V
CC  
10pF  
NC  
NC  
0.5  
0
4.9cm × 5.8cm  
2.37M  
2.2µF  
0
400  
800  
1200  
1600  
2000  
GND  
PGND  
LIGHT LEVEL (Lx)  
3129 TA02b  
3129 TA02a  
Low Noise 3.6V Converter Using Bootstrap Diode to Extend Lower VIN Range  
22nF  
22nF  
6.8µH  
V
V
< 3.6V, I  
> 3.6V, I  
= 100mA  
= 200mA  
IN  
IN  
OUT  
OUT  
BST1 SW1  
SW2 BST2  
V
V
OUT  
IN  
1.8V TO 15V  
V
V
OUT  
IN  
3.6V  
10µF  
LTC3129  
2M  
33pF  
BAT54  
RUN  
FB  
V
CC  
MPPC  
976k  
PGOOD  
PWM  
V
CC  
10µF  
NC  
NC  
2.2µF  
GND  
PGND  
3129 TA03  
3129f  
23  
For more information www.linear.com/3129  
LTC3129  
Typical applicaTions  
Solar Powered Converter with MPPC Charges Storage Capacitor  
Average Output Current  
vs Light Level (Daylight)  
22nF  
22nF  
4.7µH  
100.0  
10.0  
1.0  
BST1 SW1  
SW2 BST2  
UVLO = 4.3V  
1M  
V
IN  
V
OUT  
4.8V  
V
IN  
V
OUT  
+
4.7µF  
LTC3129  
1F  
3.09M  
1M  
47µF  
CERAMIC  
RUN  
V
CC  
COOPER BUSSMANN  
PB-5R0V105-R  
MPPC  
FB  
PowerFilm  
SP4.2-37  
SOLAR  
PGOOD  
PGOOD  
PWM  
NC  
V
CC  
MODULE  
392k  
NC  
8.4cm × 3.7cm  
0.1  
2.2µF  
1000  
10000  
100000  
1000000  
GND  
PGND  
LIGHT LEVEL (Lx)  
3129 TA04b  
3129 TA04a  
Li-Ion Powered 3V Converter with 3.1V Input UVLO Reduces Low Battery IQ to 3µA  
22nF  
22nF  
4.7µH  
BST1 SW1  
SW2 BST2  
UVLO = 3.1V  
2M  
V
OUT  
V
IN  
V
OUT  
3V  
200mA  
10µF  
LTC3129  
+
33pF  
1.58M  
1.02M  
Li-Ion  
RUN  
FB  
4.7µF  
V
CC  
MPPC  
PGOOD  
PWM  
NC  
V
CC  
1.27M  
NC  
2.2µF  
10pF  
GND  
PGND  
3129 TA05  
15V Converter Operates from Three to Eight AA or AAA Cells  
22nF  
22nF  
10µH  
V
IN  
2.42V TO 15V  
BST1 SW1  
SW2 BST2  
V
OUT  
15V  
V
IN  
V
OUT  
10µF  
25V  
25mA MINIMUM  
LTC3129  
3.01M  
255k  
22pF  
RUN  
10µF  
V
FB  
THREE TO EIGHT  
AA OR AAA  
MPPC  
PWM  
NC  
CC  
PGOOD  
BATTERIES  
V
CC  
NC  
2.2µF  
GND  
PGND  
3129 TA06  
3129f  
24  
For more information www.linear.com/3129  
LTC3129  
Typical applicaTions  
Energy Harvesting Converter Operates from a Variety of Weak Sources  
22nF  
22nF  
4.7µH  
BAS 70-05  
BST1 SW1  
SW2 BST2  
UVLO = 3.3V  
V
OUT  
V
V
OUT  
IN  
5V  
10µF  
INPUT SOURCES:  
• RF  
LTC3129  
4.99M  
V
3.32M  
1.02M  
22pF  
• AC  
RUN  
• PIEZO  
• COIL-MAGNET  
FB  
100µF  
CERAMIC  
MPPC  
PWM  
NC  
CC  
PGOOD  
BAS 70-06  
3.01M  
V
CC  
NC  
10pF  
2.2µF  
GND  
PGND  
3129 TA07  
Solar Powered Converter Extends Battery Life in Low Power 3V Primary Battery Applications  
TOKO DEM2812C  
22nF  
22nF  
3.3µH  
FDC6312P  
DUAL PMOS  
V
OUT  
3V TO 3.2V  
S1  
S2  
BST1 SW1  
SW2 BST2  
V
UVLO = 3.7V  
IN  
3.20V  
4.22M  
D1  
D2  
2.2µF  
V
V
OUT  
IN  
22µF  
15pF  
LTC3129  
4.7µF 4.99M  
G1  
G2  
2.43M  
RUN  
FB  
CR2032  
3V COIN CELL  
V
OUT  
R4  
2.43M  
PV PANEL  
SANYO AM-1815  
OR  
+
470µF  
6.3V  
V
CC  
MPPC  
PowerFilm SP4.2-37  
PWM  
NC  
PGOOD  
2.43M  
BAT54  
NC  
V
CC  
74LVC2G04  
10pF  
GND  
PGND  
2.2µF  
3129 TA09  
Percentage of Added Battery Life vs Light Level and Load  
(PowerFilm SP4.2-37, 30sq cm Panel)  
1000  
100  
10  
AVERAGE LOAD = 165µW  
AVERAGE LOAD = 330µW  
AVERAGE LOAD = 660µW  
AVERAGE LOAD = 1650µW  
AVERAGE LOAD = 3300µW  
1
100  
1,000  
10,000  
LIGHT LEVEL (Lx)  
3129 TA09b  
3129f  
25  
For more information www.linear.com/3129  
LTC3129  
package DescripTion  
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.  
UD Package  
16-Lead Plastic QFN (3mm × 3mm)  
(Reference LTC DWG # 05-08-1700 Rev A)  
Exposed Pad Variation AA  
0.70 ±0.05  
3.50 ±0.05  
2.10 ±0.05  
1.65 ±0.05  
(4 SIDES)  
PACKAGE OUTLINE  
0.25 ±0.05  
0.50 BSC  
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS  
BOTTOM VIEW—EXPOSED PAD  
PIN 1 NOTCH R = 0.20 TYP  
OR 0.25 × 45° CHAMFER  
R = 0.115  
TYP  
0.75 ±0.05  
3.00 ±0.10  
(4 SIDES)  
15 16  
PIN 1  
TOP MARK  
(NOTE 6)  
0.40 ±0.10  
1
2
1.65 ±0.10  
(4-SIDES)  
(UD16 VAR A) QFN 1207 REV A  
0.200 REF  
0.25 ±0.05  
0.50 BSC  
0.00 – 0.05  
NOTE:  
1. DRAWING CONFORMS TO JEDEC PACKAGE OUTLINE MO-220 VARIATION (WEED-4)  
2. DRAWING NOT TO SCALE  
3. ALL DIMENSIONS ARE IN MILLIMETERS  
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE  
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE  
5. EXPOSED PAD SHALL BE SOLDER PLATED  
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION  
ON THE TOP AND BOTTOM OF PACKAGE  
3129f  
26  
For more information www.linear.com/3129  
LTC3129  
package DescripTion  
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.  
MSE Package  
16-Lead Plastic MSOP, Exposed Die Pad  
(Reference LTC DWG # 05-08-1667 Rev E)  
BOTTOM VIEW OF  
EXPOSED PAD OPTION  
2.845 ±0.102  
(.112 ±.004)  
2.845 ±0.102  
(.112 ±.004)  
0.889 ±0.127  
(.035 ±.005)  
1
8
0.35  
REF  
5.23  
(.206)  
MIN  
1.651 ±0.102  
(.065 ±.004)  
1.651 ±0.102  
(.065 ±.004)  
3.20 – 3.45  
(.126 – .136)  
0.12 REF  
DETAIL “B”  
CORNER TAIL IS PART OF  
THE LEADFRAME FEATURE.  
FOR REFERENCE ONLY  
DETAIL “B”  
16  
9
0.305 ±0.038  
0.50  
(.0197)  
BSC  
NO MEASUREMENT PURPOSE  
4.039 ±0.102  
(.159 ±.004)  
(NOTE 3)  
(.0120 ±.0015)  
TYP  
0.280 ±0.076  
(.011 ±.003)  
RECOMMENDED SOLDER PAD LAYOUT  
16151413121110  
9
REF  
DETAIL “A”  
0.254  
(.010)  
3.00 ±0.102  
(.118 ±.004)  
(NOTE 4)  
0° – 6° TYP  
4.90 ±0.152  
(.193 ±.006)  
GAUGE PLANE  
0.53 ±0.152  
(.021 ±.006)  
1 2 3 4 5 6 7 8  
DETAIL “A”  
0.86  
(.034)  
REF  
1.10  
(.043)  
MAX  
0.18  
(.007)  
SEATING  
PLANE  
0.17 – 0.27  
(.007 – .011)  
TYP  
0.1016 ±0.0508  
(.004 ±.002)  
MSOP (MSE16) 0911 REV E  
0.50  
(.0197)  
BSC  
NOTE:  
1. DIMENSIONS IN MILLIMETER/(INCH)  
2. DRAWING NOT TO SCALE  
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.  
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE  
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.  
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE  
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX  
6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL  
NOT EXCEED 0.254mm (.010") PER SIDE.  
3129f  
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.  
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-  
27  
tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.  
LTC3129  
Typical applicaTion  
Dual VIN Application, Using the LTC4412 PowerPath™ Controller  
22nF  
22nF  
MBR0520  
10µH  
12V WALL ADAPTER INPUT  
V
V
= 12V, I  
= 200mA  
= 50mA  
IN  
IN  
OUT  
OUT  
FDN338  
BSS314  
= 3.6V, I  
BST1 SW1  
SW2 BST2  
V
OUT  
V
V
OUT  
IN  
12V  
10µF  
25V  
LTC3129  
10pF  
3.01M  
LTC4412  
GATE  
10µF  
RUN  
V
FB  
IN  
V
CC  
MPPC  
+
SENSE  
STAT  
324k  
Li-Ion  
PGOOD  
PWM  
NC  
V
CC  
CTL  
NC  
GND  
2.2µF  
GND  
PGND  
3129 TA08  
relaTeD parTs  
PART NUMBER DESCRIPTION  
COMMENTS  
= 2.2V, V  
LTC3103  
LTC3104  
LTC3105  
LTC3112  
LTC3115-1  
LTC3531  
15V, 300mA Synchronous Step-Down DC/DC Converter with  
Ultralow Quiescent Current  
V
I
= 15V, V  
= 0.8V, I = 1.8µA,  
OUT(MIN) Q  
IN(MIN)  
IN(MAX)  
<1µA, 3mm × 3mm DFN-10, MSOP-10 Packages  
SD  
15V, 300mA Synchronous Step-Down DC/DC Converter with  
Ultralow Quiescent Current and 10mA LDO  
V
SD  
= 2.2V, V  
= 15V, V  
= 0.8V, I = 2.8µA,  
OUT(MIN) Q  
IN(MIN)  
IN(MAX)  
I
<1µA, 4mm × 3mm DFN-14, MSOP-16 Packages  
400mA Step-up Converter with MPPC and 250mV Start-Up  
V
SD  
= 0.2V, V  
= 5V, V  
= 0 5.25V , I = 22µA,  
MAX Q  
IN(MIN)  
IN(MAX)  
OUT(MIN)  
I
<1µA, 3mm × 3mm DFN-10/MSOP-12 Packages  
15V, 2.5A, 750kHz Monolithic Synch Buck/Boost  
V
SD  
= 2.7V, V  
= 15V, V  
= 2.7V to 14V, I = 50µA,  
OUT(MIN) Q  
IN(MIN)  
IN(MAX)  
I
<1µA, 4mm × 5mm DFN-16 TSSOP-20E Packages  
40V, 2A, 2MHz Monolithic Synch Buck/Boost  
V
SD  
= 2.7V, V  
= 40V, V  
= 2.7V to 40V, I = 50µA,  
OUT(MIN) Q  
IN(MIN)  
IN(MAX)  
I
<1µA, 4mm × 5mm DFN-16 and TSSOP-20E Packages  
5.5V, 200mA, 600kHz Monolithic Synch Buck/Boost  
20V, 50mA High Efficiency Nano Power Step-Down Regulator  
Ultralow Voltage Step-Up Converter and Power Manager  
V
SD  
= 1.8V, V  
= 5.5V, V  
= 2V to 5V, I = 16µA,  
OUT(MIN) Q  
IN(MIN)  
IN(MAX)  
I
<1µA, 3mm × 3mm DFN-8 and ThinSOT Packages  
LTC3388-1/  
LTC3388-3  
V
= 2.7V, V  
=20V, V  
= Fixed 1.1V to 5.5V,  
OUT(MIN)  
IN(MIN)  
IN(MAX)  
I = 720nA, I = 400nA, 3mm × 3mm DFN-10, MSOP-10 Packages  
Q
SD  
LTC3108/  
LTC3108-1  
V
= 0.02V, V  
= 1V, V  
= Fixed 2.35V to 5V,  
OUT(MIN)  
IN(MIN)  
IN(MAX)  
I = 6µA, I <1µA, 3mm × 4mm DFN-12, SSOP-16 Packages  
Q
SD  
LTC3109  
LTC3588-1  
LTC4070  
Auto-Polarity, Ultralow Voltage Step-Up Converter and Power  
Manager  
V
Q
= 0.03V, V  
SD  
= 1V, V  
= Fixed 2.35V to 5V,  
OUT(MIN)  
IN(MIN)  
IN(MAX)  
I = 7µA, I <1µA, 4mm × 4mm QFN-20, SSOP-20 Packages  
Piezo Electric Energy Harvesting Power Supply  
V
= 2.7V, V  
= 20V, V  
= Fixed 1.8V to 3.6V,  
OUT(MIN)  
IN(MIN)  
IN(MAX)  
I = 950nA, I 450nA, 3mm × 3mm DFN-10, MSOP-10E Packages  
Q
SD  
Li-Ion/Polymer Low Current Shunt Battery Charger System  
V
= 450nA to 50mA, V  
+ 4.0V, 4.1V, 4.2V, I = 300nA,  
IN(MIN)  
FLOAT Q  
2mm × 3mm DFN-8, MSOP-8 Packages  
3129f  
LT 0213 • PRINTED IN USA  
LinearTechnology Corporation  
1630 McCarthy Blvd., Milpitas, CA 95035-7417  
28  
●
●
LINEAR TECHNOLOGY CORPORATION 2013  
(408)432-1900 FAX: (408) 434-0507 www.linear.com/3129  

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SI9137DB

 Multi-Output, Sequence Selectable Power-Supply Controller for Mobile Applications

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SI9137LG

 Multi-Output, Sequence Selectable Power-Supply Controller for Mobile Applications

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SI9122E

 500-kHz Half-Bridge DC/DC Controller with Integrated Secondary Synchronous Rectification Drivers

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