LTC3801B [Linear]
Micropower Constant Frequency Step-Down DC/DC Controllers in ThinSOT; 微恒频降压型DC / DC采用ThinSOT封装的控制器型号: | LTC3801B |
厂家: | Linear |
描述: | Micropower Constant Frequency Step-Down DC/DC Controllers in ThinSOT |
文件: | 总12页 (文件大小:241K) |
中文: | 中文翻译 | 下载: | 下载PDF数据表文档文件 |
LTC3801/LTC3801B
Micropower
Constant Frequency Step-Down
DC/DC Controllers in ThinSOT
U
FEATURES
DESCRIPTIO
The LTC®3801/LTC3801B are constant frequency cur-
rent mode step-down DC/DC controllers in a low profile
■
High Efficiency: Up to 94%
■
Very Low No-Load Quiescent Current:
(1mm max) 6-lead SOT-23 (ThinSOTTM
) package. The
Only 16µA (LTC3801)
■
High Output Currents Easily Achieved
parts provide excellent AC and DC load and line regula-
tion with ±1.5% output voltage accuracy. The LTC3801
consumes only 195µA of quiescent current in normal
operation, dropping to 16µA under no-load conditions.
■
Internal Soft-Start
■
Wide VIN Range: 2.4V to 9.8V
■
Low Dropout: 100% Duty Cycle
■
Constant Frequency 550kHz Operation
TheLTC3801/LTC3801Bincorporateanundervoltagelock-
out feature that shuts down the device when the input
voltage falls below 2.2V. The LTC3801 automatically
switches into Burst Mode operation at light loads which
enhancesefficiencyatlowoutputcurrent.IntheLTC3801B,
BurstModeoperationisdisabledforloweroutputrippleat
light loads.
Burst Mode® Operation for High Efficiency
■
at Light Loads (LTC3801)
■
Burst Mode Operation Disabled for Lower Output
Ripple at Light Loads (LTC3801B)
■
Output Voltage as Low as 0.8V
■
±1.5% Voltage Reference Accuracy
■
Current Mode Operation for Excellent Line and Load
To further maximize the life of a battery source, the
external P-channel MOSFET is turned on continuously in
dropout (100% duty cycle). High switching frequency of
550kHz allows the use of a small inductor.
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode is a registered trademark of Linear Technology Corporation.
ThinSOT is a trademark of Linear Technology Corporation.
Transient Response
■
■
Only 6µA Supply Current in Shutdown (LTC3801)
Low Profile (1mm) SOT-23 Package
U
APPLICATIO S
■
1- or 2-Cell Li-Ion Battery-Powered Applications
■
Wireless Devices
■
Portable Computers
Distributed Power Systems
■
U
LTC3801 Efficiency vs Load Current*
TYPICAL APPLICATIO
100
V
= 2.5V
OUT
V
= 3.3V
95
90
85
IN
V
= 4.2V
IN
550kHz Micropower Step-Down DC/DC Controller
220pF
V
= 6.6V
IN
V
IN
10k
80
75
2.7V TO 9.8V
I
/RUN
LTC3801/
V
TH
IN
–
10µF
0.025Ω
V
= 8.4V
V
= 9.8V
IN
LTC3801B
IN
70
65
60
55
50
GND
SENSE
V
PGATE
FB
402k
4.7µH
V
OUT
866k
2.5V
2A
+
0.1
1
10
100
1000 10000
47µF
LOAD CURRENT (mA)
3801 TA02
3801 TA01
*SEE NO-LOAD IQ vs INPUT VOLTAGE ON THE LAST PAGE OF THIS DATA SHEET
3801f
1
LTC3801/LTC3801B
W W U W
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ABSOLUTE MAXIMUM RATINGS
PACKAGE/ORDER INFORMATION
(Note 1)
ORDER PART
NUMBER
Input Supply Voltage (VIN)........................ –0.3V to 10V
SENSE–, PGATE Voltages ............ –0.3V to (VIN + 0.3V)
VFB, ITH/RUN Voltages ............................. –0.3V to 2.4V
PGATE Peak Output Current (<10µs) ........................ 1A
Operating Temperature Range (Note 2) .. – 40°C to 85°C
Junction Temperature (Note 3)............................ 150°C
Storage Temperature Range ................. –65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
TOP VIEW
I
/RUN 1
GND 2
6 PGATE
5 V
TH
LTC3801ES6
LTC3801BES6
IN
4 SENSE
–
V
3
FB
S6 PART MARKING
S6 PACKAGE
6-LEAD PLASTIC TSOT-23
TJMAX = 150°C, θJA = 230°C/W
LTACR
LTAHN
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● indicates specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 4.2V unless otherwise noted. (Note 2)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Input Voltage Range
●
2.4
9.8
V
Input DC Supply Current
Normal Operation
SLEEP Mode
Typicals at V = 4.2V (Note 4)
IN
2.4V ≤ V ≤ 9.8V, V /RUN = 1.3V
195
16
6
300
30
15
17
2
µA
µA
µA
µA
µA
IN
IN
ITH
2.4V ≤ V ≤ 9.8V (LTC3801 Only)
Shutdown
2.4V ≤ V ≤ 9.8V, V /RUN = 0V (LTC3801)
IN ITH
2.4V ≤ V ≤ 9.8V, V /RUN = 0V (LTC3801B)
8
IN
ITH
UVLO
V
< UVLO Threshold
1
IN
Undervoltage Lockout Threshold
V
V
Rising
Falling
●
●
1.8
1.7
2.3
2.2
V
V
IN
IN
Start-Up Current Source
V
V
/RUN = 0V (LTC3801)
ITH
/RUN = 0V (LTC3801B)
ITH
0.5
1.0
1
2
1.5
3.0
µA
µA
Shutdown Threshold (at I /RUN)
Regulated Feedback Voltage
V
/RUN Rising
●
●
0.3
0.788
0.780
0.6
0.800
0.800
0.95
0.812
0.812
V
V
V
TH
ITH
0°C ≤ T ≤ 85°C (Note 5)
A
–40°C ≤ T ≤ 85°C (Note 5)
A
Feedback Voltage Line Regulation
Feedback Voltage Load Regulation
2.4V ≤ V ≤ 9.8V (Note 5)
0.05
2
2
mV/V
mV/µA
mV/µA
IN
I
I
/RUN Sinking 5µA (Note 5)
TH
/RUN Sourcing 5µA (Note 5)
TH
V
Input Current
(Note 5)
Measured at V
2
10
0.910
nA
V
mV
FB
Overvoltage Protect Threshold
Overvoltage Protect Hysteresis
Oscillator Frequency
Normal Operation
Output Short Circuit
0.850
500
0.880
40
FB
V
V
= 0.8V
= 0V
550
210
650
kHz
kHz
FB
FB
Gate Drive Rise Time
Gate Drive Fall Time
C
C
= 3000pF
= 3000pF
40
40
ns
ns
LOAD
LOAD
Peak Current Sense Voltage
Duty Cycle < 40% (Note 6)
LTC3801
●
●
109
95
117
104
125
113
mV
mV
LTC3801B
Peak Current Sense Voltage in Burst Mode Operation
Default Soft-Start Time
LTC3801 Only
26
0.6
mV
ms
T = T + (P • θ °C/W)
Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency.
Note 1: Absolute Maximum Ratings are those values beyond which the life
J
A
D
JA
of a device may be impaired.
Note 2: The LTC3801ES6/LTC3801BES6 are guaranteed to meet specifica-
tions from 0°C to 70°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.
Note 5: The LTC3801/LTC3801B are tested in a feedback loop that servos
V
to the output of the error amplifier while maintaining I /RUN at the
FB
TH
midpoint of the current limit range.
Note 3: T is calculated from the ambient temperature T and power
J
A
Note 6: Peak current sense voltage is reduced dependent on duty cycle as
dissipation P according to the following formula:
D
given in Figure 1.
3801f
2
LTC3801/LTC3801B
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TYPICAL PERFOR A CE CHARACTERISTICS
Input DC Supply Current (Normal)
vs Input Voltage
Input DC Supply Current (SLEEP)
vs Input Voltage (LTC3801 Only)
Input DC Supply Current
(Shutdown) vs Input Voltage
225
20
15
T
= 25°C
ITH
T
= 25°C
A
T
= 25°C
ITH
A
A
V
/RUN = 1.3V
V
/RUN = 0V
215
205
18
16
12
9
LTC3801B
LTC3801
6
195
185
175
14
12
10
3
0
6
6
2
3
4
5
7
8
9
10
2
3
4
5
7
8
9
10
6
2
3
4
5
7
8
9
10
V
(V)
V
IN
(V)
V
(V)
IN
IN
3801 G01
3801 G02
3801 G03
Undervoltage Lockout Threshold
vs Temperature
Shutdown Threshold
vs Temperature
Regulated Feedback Voltage
vs Temperature
2.2
2.0
1.8
1.6
1.4
1.2
800
700
600
500
812
808
804
800
V
IN
= 4.2V
V
IN
= 4.2V
V
IN
RISING
V
IN
FALLING
796
792
788
400
–50 –30 –10 10
30
50
70
90
–50 –30 –10 10
30
50
70
90
30
TEMPERATURE (°C)
80
90
–50 –30 –10 10
50
TEMPERATURE (°C)
TEMPERATURE (°C)
3801 G04
3801 G05
3801 G06
Regulated Feedback Voltage
vs Input Voltage
Oscillator Frequency
vs Temperature
Oscillator Frequency
vs Input Voltage
600
590
580
570
560
550
540
530
520
510
500
560
555
550
545
812
808
804
800
796
792
788
T
= 25°C
T
= 25°C
A
V
= 4.2V
A
IN
540
7
8
2
3
4
5
6
9
10
2
3
4
5
6
7
8
9
10
–50
–10 10
30
50
70
90
–30
V
IN
(V)
V
(V)
TEMPERATURE (°C)
IN
3801 G07
3801 G09
3801 G08
3801f
3
LTC3801/LTC3801B
U
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PI FU CTIO S
SENSE– (Pin 4): Current Sense Pin. An external sense
ITH/RUN (Pin 1): This pin performs two functions. It
servesastheerroramplifiercompensationpointaswellas
the run control input. Nominal voltage range for this pin is
0.7V to 1.9V. Forcing this pin below 0.6V causes the
device to be shut down. In shutdown, all functions are
disabled and the PGATE pin is held high.
resistor is connected between this pin and VIN (Pin 5).
VIN (Pin 5): Supply Pin. This pin must be closely de-
coupled to GND (Pin 2).
PGATE (Pin 6): Gate Drive for the External P-Channel
MOSFET. This pin swings from 0V to VIN.
GND (Pin 2): Ground Pin.
V
FB (Pin 3): Receives the feedback voltage from an exter-
nal resistor divider across the output.
U
U
W
FU CTIO AL DIAGRA
4
–
SENSE
15mV (LTC3801B)
V
IN
5
BURST
SLOPE
COMPENSATION
UV
UNDERVOLTAGE
LOCKOUT
VOLTAGE
REFERENCE
DEFEAT
BURST
CLAMP
0.8V
–
(LTC3801B)
+
CURRENT
COMPARATOR
1µA (LTC3801)
2µA (LTC3801B)
SHUTDOWN
COMPARATOR
I
/RUN
–
+
0.3V
TH
1
I
+
LIM
I
TH
SHDN
BUFFER
–
550kHz
OSCILLATOR
R
S
RS
LATCH
Q
V
IN
FREQUENCY
FOLDBACK
SLEEP
SWITCHING
LOGIC AND
BLANKING
CIRCUIT
COMPARATOR
PGATE
–
+
6
SLEEP
BURST
DEFEAT
(LTC3801B)
0V
SOFT-START
CLAMP
OVERVOLTAGE
COMPARATOR
SHORT-CIRCUIT
DETECT
0.15V
+
+
–
–
ERROR
AMPLIFIER
V
FB
0.225V
0.88V
0.3V
–
+
3
0.8V
1.2V
GND
2
3801 FD
3801f
4
LTC3801/LTC3801B
U
(Refer to the Functional Diagram)
OPERATIO
Main Control Loop (Normal Operation)
and the load will eventually cause the error amplifier out-
puttostartdriftinghigher. Whentheerroramplifieroutput
rises to 0.225V above its zero current level (approximately
0.925V), the sleep comparator will untrip and normal op-
eration will resume. The next oscillator cycle will turn the
external MOSFET on and the switching cycle will repeat.
The LTC3801/LTC3801B are constant frequency current
mode step-down switching regulator controllers. During
normaloperation,anexternalP-channelMOSFETisturned
on each cycle when the oscillator sets the RS latch and
turned off when the current comparator resets the latch.
Thepeakinductorcurrentatwhichthecurrentcomparator
tripsiscontrolledbythevoltageontheITH/RUNpin, which
is the output of the error amplifier. The negative input to
the error amplifier is the output feedback voltage VFB
which is generated by an external resistor divider con-
nected between VOUT and ground. When the load current
increases, it causes a slight decrease in VFB relative to the
0.8V reference, which in turn causes the ITH/RUN voltage
to increase until the average inductor current matches the
new load current.
Low Load Current Operation (LTC3801B Only)
Under very light load current conditions, the ITH/RUN pin
voltage will be very close to the zero current level of 0.85V.
As the load current decreases further, an internal offset at
the current comparator input will ensure that the current
comparator remains tripped (even at zero load current)
and the regulator will start to skip cycles, as it must, in
order to maintain regulation. This behavior allows the
regulator to maintain constant frequency down to very
light loads, resulting in less low frequency noise genera-
tion over a wide load current range.
ThemaincontrolloopisshutdownbypullingtheITH/RUN
pin to ground. Releasing the ITH/RUN pin allows an
internal1µAcurrentsource(2µAonLTC3801B)tocharge
up the external compensation network. When the ITH/
RUN pin voltage reaches approximately 0.6V, the main
control loop is enabled and the ITH/RUN voltage is pulled
up by a clamp to its zero current level of approximately
onediodevoltagedrop(0.7V). Astheexternalcompensa-
tion network continues to charge up, the corresponding
peak inductor current level follows, allowing normal op-
eration. The maximum peak inductor current attainable is
set by a clamp on the ITH/RUN pin at 1.2V above the zero
current level (approximately 1.9V).
Figure 1 illustrates this result for the circuit on the front
page of this data sheet using both an LTC3801 (in Burst
Mode operation) and an LTC3801B (with Burst Mode
operation disabled). At an output current of 100mA, the
LTC3801 exhibits an output ripple of 81.6mVP-P, whereas
the LTC3801B has an output ripple of only 17.6mVP-P. At
lower output current levels, the improvement is even
greater. This comes at a tradeoff of lower efficiency for the
non Burst Mode part at light load currents (see Figure 2).
Also notice the constant frequency operation of the
LTC3801B, even at 5% of maximum output current.
Dropout Operation
Burst Mode Operation (LTC3801 Only)
When the input supply voltage decreases towards the
output voltage, the rate of change of inductor current
during the on cycle decreases. This reduction means that
at some input-output differential, the external P-channel
MOSFET will remain on for more than one oscillator cycle
(start dropping off-cycles) since the inductor current has
not ramped up to the threshold set by the error amplifier.
Further reduction in input supply voltage will eventually
cause the external P-channel MOSFET to be turned on
100%, i.e., DC. The output voltage will then be determined
by the input voltage minus the voltage drop across the
sense resistor, the MOSFET and the inductor.
The LTC3801 incorporates Burst Mode operation at low
load currents (<25% of IMAX). In this mode, an internal
clamp sets the peak current of the inductor at a level cor-
responding to an ITH/RUN voltage 0.3V above its zero
current level (approximately 1V), even though the actual
ITH/RUN voltage is lower. When the inductor’s average
current is greater than the load requirement, the voltage at
the ITH/RUN pin will drop. When the ITH/RUN voltage falls
to0.15Vaboveitszerocurrentlevel(approximately0.85V),
the sleep comparator will trip, turning off the external
MOSFET. In sleep, the input DC supply current to the IC is
reducedto16µAfrom195µAinnormaloperation.Withthe
switch held off, average inductor current will decay to zero
3801f
5
LTC3801/LTC3801B
U
(Refer to the Functional Diagram)
OPERATIO
VOUT Ripple for Front Page Circuit Using the LTC3801
VOUT Ripple for Front Page Circuit Using the LTC3801B
(Burst Mode Operation Disabled)
(with Burst Mode Operation)
20mVAC/DIV
20mVAC/DIV
VIN = 4.2V
VOUT= 2.5V
IOUT = 100mA
5µs/DIV
3801 F01a
V
IN = 4.2V
5µs/DIV
3801 F01b
VOUT= 2.5V
IOUT = 100mA
Figure 1. Output Ripple Waveforms for the Front Page Circuit
100
V
= 2.5V
This lower frequency allows the inductor current to safely
discharge, thereby preventing current runaway. After the
short is removed, the oscillator frequency will gradually
increase back to 550kHz as VFB rises through 0.3V on its
way back to 0.8V.
OUT
95
90
85
V
= 3.3V
IN
80
75
V
V
= 4.2V
= 6.6V
IN
70
65
60
55
50
Overvoltage Protection
IN
V
= 8.4V
IN
V
= 9.8V
100
IN
If VFB exceeds its regulation point of 0.8V by more than
10% for any reason, such as an output short circuit to a
higher voltage, the overvoltage comparator will hold the
external P-channel MOSFET off. This comparator has a
typical hysteresis of 40mV.
0.1
1
10
1000
10000
LOAD CURRENT (mA)
3801 F02
Figure 2. LTC3801B Efficiency vs Load Current
Slope Compensation and Inductor’s Peak Current
Undervoltage Lockout Protection
The switch on duty cycle in normal operation is given by:
To prevent operation of the external P-channel MOSFET
with insufficient gate drive, an undervoltage lockout cir-
cuit is incorporated into the LTC3801/LTC3801B. When
the input supply voltage drops below approximately 1.7V,
the P-channel MOSFET and all internal circuitry other than
the undervoltage block itself are turned off. Input supply
current in undervoltage is approximately 1µA.
V
OUT + VD
Duty Cycle =
V + VD
IN
where VD is the forward voltage drop of the external diode
at the average inductor current. For duty cycles less than
40%, the inductor’s peak current is determined by:
V
ITH/RUN – 0.7V
10RSENSE
IMAX
=
Short-Circuit Protection
If the output is shorted to ground, the frequency of the
oscillator is folded back from 550kHz to approximately
210kHz while maintaining the same minimum on time.
However, for duty cycles greater than 40%, slope com-
pensation begins and effectively reduces the peak
3801f
6
LTC3801/LTC3801B
U
(Refer to the Functional Diagram)
OPERATIO
inductor current. The amount of reduction is given by the
curve in Figure 3.
100
115
105
95
90
LTC3801
80
LTC3801B
Soft-Start
85
70
60
50
40
30
An internal default soft-start circuit is employed at power-
up and/or when coming out of shutdown. The soft-start
circuit works by internally clamping the voltage at the
ITH/RUN pin to the corresponding zero current level and
graduallyraisingtheclampvoltagesuchthattheminimum
time required for the programmed switch current to reach
its maximum is approximately 0.6ms. After the soft-start
circuit has timed out, it is disabled until the part is put in
shutdown again or the input supply is cycled.
75
65
55
V
A
= 4.2V
45
IN
T
= 25°C
35
60 70
DUTY CYCLE (%)
20 30 40 50
80 90 100
3801 F03
Figure 3. Maximum Current Limit Trip Voltage vs Duty Cycle
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APPLICATIO S I FOR ATIO
ThebasicLTC3801/LTC3801Bapplicationcircuitisshown
on the front page of this data sheet. External component
selectionisdrivenbytheloadrequirementandbeginswith
the selection of the inductor and RSENSE. Next, the power
MOSFET and the output diode are selected followed by the
input bypass capacitor CIN and output bypass capacitor
However,foroperationthatisabove40%dutycycle,slope
compensation effect has to be taken into consideration to
selecttheappropriatevaluetoprovidetherequiredamount
of current. Using Figure 3, the value of RSENSE is:
SF
RSENSE
=
COUT
.
(10)(IOUT)(100)
where SF is the “Slope Factor.”
RSENSE Selection for Output Current
RSENSE is chosen based on the required output current.
With the current comparator monitoring the voltage
developed across RSENSE, the threshold of the compara-
tor determines the inductor’s peak current. The output
current the LTC3801 can provide is given by:
Inductor Value Calculation
The operating frequency and inductor selection are inter-
related in that higher operating frequencies permit the use
of a smaller inductor for the same amount of inductor
ripplecurrent. However, thisisattheexpenseofefficiency
due to an increase in MOSFET gate charge losses.
0.117 IRIPPLE
IOUT
=
−
RSENSE
2
The inductance value also has a direct effect on ripple cur-
rent. The ripple current, IRIPPLE, decreases with higher in-
ductance or frequency and increases with higher VIN or
VOUT.Theinductor’speak-to-peakripplecurrentisgivenby:
where IRIPPLE is the inductor peak-to-peak ripple current
(seeInductorValueCalculationsection).FortheLTC3801B
use 104mV in the previous equation and follow through
the analysis using that number.
V − VOUT
V
OUT + VD
IN
IRIPPLE
=
A reasonable starting point for setting ripple current is
f(L)
V + VD
IN
I
RIPPLE = (0.4)(IOUT). Rearranging the above equation, it
becomes:
wherefistheoperatingfrequency.Acceptinglargervalues
of IRIPPLE allows the use of low inductances, but results in
higher output voltage ripple and greater core losses. A
1
RSENSE
=
for Duty Cycle < 40%
(10)(IOUT
)
reasonable starting point for setting ripple current is
3801f
7
LTC3801/LTC3801B
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APPLICATIO S I FOR ATIO
IRIPPLE =0.4(IOUT(MAX)).Remember,themaximumIRIPPLE
manufacturerisKoolMµ. Toroidsareveryspaceefficient,
especially when you can use several layers of wire.
Because they generally lack a bobbin, mounting is more
difficult. However, new designs for surface mount that do
not increase the height significantly are available.
occurs at the maximum input voltage.
In Burst Mode operation on the LTC3801, the ripple
current is normally set such that the inductor current is
continuous during the burst periods. Therefore, the peak-
to-peak ripple current must not exceed:
Power MOSFET Selection
0.03
An external P-channel power MOSFET must be selected
for use with the LTC3801/LTC3801B. The main selection
criteria for the power MOSFET are the threshold voltage
IRIPPLE
≤
RSENSE
This implies a minimum inductance of:
V
GS(TH) and the “on” resistance RDS(ON), reverse transfer
capacitance CRSS and total gate charge.
V − VOUT
V
OUT + VD
IN
LMIN
=
Since the LTC3801/LTC3801B are designed for operation
down to low input voltages, a sublogic level threshold
MOSFET (RDS(ON) guaranteed at VGS = 2.5V) is required
for applications that work close to this voltage. When
these MOSFETs are used, make sure that the input supply
to the LTC3801/LTC3801B is less than the absolute maxi-
mum VGS rating, typically 8V.
V + VD
0.03
RSENSE
IN
f
(Use VIN(MAX) = VIN)
A smaller value than LMIN could be used in the circuit;
however, the inductor current will not be continuous
during burst periods.
TherequiredminimumRDS(ON) oftheMOSFETisgoverned
byitsallowablepowerdissipation.Forapplicationsthatmay
operatetheLTC3801/LTC3801B indropout,i.e.,100%duty
cycle, at its worst case the required RDS(ON) is given by:
Inductor Core Selection
Once the value for L is known, the type of inductor must be
selected. High efficiency converters generally cannot af-
ford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy
or Kool Mµ® cores. Actual core loss is independent of core
size for a fixed inductor value, but it is very dependent on
inductance selected. As inductance increases, core losses
go down. Unfortunately, increased inductance requires
more turns of wire and therefore copper losses will in-
crease. Ferrite designs have very low core losses and are
preferred at high switching frequencies, so design goals
canconcentrateoncopperlossandpreventingsaturation.
Ferrite core material saturates “hard,” which means that
inductance collapses abruptly when the peak design cur-
rent is exceeded. This results in an abrupt increase in
inductor ripple current and consequent output voltage
ripple. Do not allow the core to saturate!
PP
RDS(ON)
=
DC=100%
2
I
(
1+ δp
(
)
)
OUT(MAX)
where PP is the allowable power dissipation and δp is the
temperature dependency of RDS(ON). (1 + δp) is generally
given for a MOSFET in the form of a normalized RDS(ON) vs
temperature curve, but δp = 0.005/°C can be used as an
approximation for low voltage MOSFETs.
In applications where the maximum duty cycle is less than
100% and the LTC3801/LTC3801B are in continuous
mode, the RDS(ON) is governed by:
PP
RDS(ON)
2
DC I
1+ δp
(
)
(
)
OUT
Molypermalloy (from Magnetics, Inc.) is a very good, low
loss core material for toroids, but it is more expensive
than ferrite. A reasonable compromise from the same
where DC is the maximum operating duty cycle of the
LTC3801/LTC3801B.
Kool Mµ is a registered trademark of Magnetics, Inc.
3801f
8
LTC3801/LTC3801B
W U U
APPLICATIO S I FOR ATIO
U
Output Diode Selection
input capacitor sized for the maximum RMS current must
beused. ThemaximumRMScapacitorcurrentisgivenby:
The catch diode carries load current during the off-time.
The average diode current is therefore dependent on the
P-channel switch duty cycle. At high input voltages the
diode conducts most of the time. As VIN approaches VOUT
the diode conducts only a small fraction of the time. The
most stressful condition for the diode is when the output
is short-circuited. Under this condition the diode must
safelyhandleIPEAK atcloseto100%dutycycle. Therefore,
itisimportanttoadequatelyspecifythediodepeakcurrent
and average power dissipation so as not to exceed the
diode ratings.
1/2
]
V
V − V
OUT
(
)
[
OUT IN
CIN Required IRMS ≈IMAX
V
IN
This formula has a maximum value at VIN = 2VOUT, where
RMS = IOUT/2. This simple worst-case condition is com-
I
monlyusedfordesignbecauseevensignificantdeviations
donotoffermuchrelief.Notethatcapacitormanufacturer’s
ripplecurrentratingsareoftenbasedon2000hoursoflife.
This makes it advisable to further derate the capacitor, or
to choose a capacitor rated at a higher temperature than
required. Several capacitors may be paralleled to meet the
size or height requirements in the design. Due to the high
operating frequency of the LTC3801/LTC3801B, ceramic
capacitors can also be used for CIN. Always consult the
manufacturer if there is any question.
Under normal load conditions, the average current con-
ducted by the diode is:
V − VOUT
V + VD
IN
IN
ID=
IOUT
The allowable forward voltage drop in the diode is calcu-
lated from the maximum short-circuit current as:
The selection of COUT is driven by the required effective
series resistance (ESR). Typically, once the ESR require-
ment is satisfied, the capacitance is adequate for filtering.
The output ripple (∆VOUT) is approximated by:
PD
ISC(MAX)
V ≈
F
1
∆VOUT ≈IRIPPLE ESR +
8fCOUT
where PD is the allowable power dissipation and will be
determined by efficiency and/or thermal requirements.
where f is the operating frequency, COUT is the output
capacitance and IRIPPLE is the ripple current in the induc-
tor. The output ripple is highest at maximum input voltage
since ∆IL increases with input voltage.
A fast switching diode must also be used to optimize
efficiency. Schottky diodes are a good choice for low
forwarddropandfastswitchingtimes. Remembertokeep
lead length short and observe proper grounding (see
Board Layout Checklist) to avoid ringing and increased
dissipation.
Manufacturers such as Nichicon, United Chemicon and
Sanyoshouldbeconsideredforhighperformancethrough-
hole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest ESR (size)
product of any aluminum electrolytic at a somewhat
higher price. Once the ESR requirement for COUT has been
met, the RMS current rating generally far exceeds the
An additional consideration in applications where low no-
load quiescent current is critical is the reverse leakage
current of the diode at the regulated output voltage. A
leakage greater than several microamperes can represent
a significant percentage of the total input current.
IRIPPLE(P-P) requirement.
CIN and COUT Selection
In surface mount applications, multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum elec-
trolytic and dry tantalum capacitors are both available in
In continuous mode, the source current of the P-channel
MOSFET is a square wave of duty cycle (VOUT + VD)/
(VIN + VD). To prevent large voltage transients, a low ESR
3801f
9
LTC3801/LTC3801B
W U U
U
APPLICATIO S I FOR ATIO
surfacemountconfigurations. Inthecaseoftantalum, itis
critical that the capacitors are surge tested for use in
switching power supplies. An excellent choice is the AVX
TPS, AVX TPSV and KEMET T510 series of surface mount
tantalum, available in case heights ranging from 2mm to
4mm. Other capacitor types include Sanyo OS-CON,
Nichicon PL series and Panasonic SP.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC3801/LTC3801B circuits: 1) LTC3801/
LTC3801B DC bias current, 2) MOSFET gate charge cur-
rent, 3) I2R losses and 4) voltage drop of the output diode.
1. The VIN current is the DC supply current, given in the
electrical characteristics, that excludes MOSFET driver
and control currents. VIN current results in a small loss
which increases with VIN.
Setting Output Voltage
The LTC3801/LTC3801B develop a 0.8V reference voltage
between the feedback (Pin 3) terminal and ground (see
Figure 4). By selecting resistor R1, a constant current is
caused to flow through R1 and R2 to set the overall output
voltage. The regulated output voltage is determined by:
2. MOSFETgatechargecurrentresultsfromswitchingthe
gate capacitance of the power MOSFET. Each time a
MOSFET gate is switched from low to high to low again,
a packet of charge dQ moves from VIN to ground. The
resulting dQ/dt is a current out of VIN which is typically
much larger than the DC supply current. In continuous
mode, IGATECHG = (f)(dQ).
3. I2R losses are predicted from the DC resistances of the
MOSFET, inductor and current shunt. In continuous
mode the average output current flows through L but is
“chopped” between the P-channel MOSFET (in series
withRSENSE) andtheoutputdiode.TheMOSFETRDS(ON)
plusRSENSE multipliedbydutycyclecanbesummedwith
the resistances of L and RSENSE to obtain I2R losses.
R2
VOUT = 0.8 1+
R1
Formostapplications, an80kresistorissuggestedforR1.
In applications where low no-load quiescent current is
critical, R1 should be made >400k to limit the feedback
dividercurrenttoapproximately10%ofthechipquiescent
current. If R2 then results in a very high impedance, it may
bebeneficialtobypassR2witha5pFto10pFcapacitor. To
prevent stray pickup, locate resistors R1 and R2 close to
LTC3801/LTC3801B.
4. Theoutputdiodeisamajorsourceofpowerlossathigh
currents and gets worse at high input voltages. The
diode loss is calculated by multiplying the forward
voltagetimesthediodedutycyclemultipliedbytheload
current. For example, assuming a duty cycle of 50%
with a Schottky diode forward voltage drop of 0.4V, the
loss increases from 0.5% to 8% as the load current
increases from 0.5A to 2A.
V
OUT
LTC3801/
R2
LTC3801B
3
V
FB
R1
3801 F04
Figure 4. Setting Output Voltage
5. Transition losses apply to the external MOSFET and
increase at higher operating frequencies and input
voltages. Transition losses can be estimated from:
Efficiency Considerations
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
oftenusefultoanalyzeindividuallossestodeterminewhat
is limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Transition Loss = 2(VIN)2IO(MAX) RSS
(f)
C
Other losses including CIN and COUT ESR dissipative
losses, and inductor core losses, generally account for
less than 2% total additional loss.
Efficiency = 100% – (η1 + η2 + η3 + ...)
where η1, η2, etc. are the individual losses as a percent-
age of input power.
3801f
10
LTC3801/LTC3801B
W U U
APPLICATIO S I FOR ATIO
U
Foldback Current Limiting
V
LTC3801/
LTC3801B
OUT
R2
R1
As described in the Output Diode Selection, the worst-
case dissipation occurs with a short-circuited output
when the diode conducts the current limit value almost
continuously. To prevent excessive heating in the diode,
foldback current limiting can be added to reduce the
current in proportion to the severity of the fault.
+
I
/RUN V
FB
TH
D
D
FB1
FB2
3801 F05
Figure 5. Foldback Current Limiting
Foldback current limiting is implemented by adding di-
odes DFB1 and DFB2 between the output and the ITH/RUN
pin as shown in Figure 5. In a hard short (VOUT = 0V), the
current will be reduced to approximately 50% of the
maximum output current.
U
PACKAGE DESCRIPTIO
S6 Package
6-Lead Plastic TSOT-23
(Reference LTC DWG # 05-08-1636)
2.90 BSC
(NOTE 4)
0.62
MAX
0.95
REF
1.22 REF
1.4 MIN
1.50 – 1.75
2.80 BSC
3.85 MAX 2.62 REF
(NOTE 4)
PIN ONE ID
RECOMMENDED SOLDER PAD LAYOUT
PER IPC CALCULATOR
0.30 – 0.45
6 PLCS (NOTE 3)
0.95 BSC
0.80 – 0.90
0.20 BSC
DATUM ‘A’
0.01 – 0.10
1.00 MAX
0.30 – 0.50 REF
1.90 BSC
0.09 – 0.20
(NOTE 3)
S6 TSOT-23 0302
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DRAWING NOT TO SCALE
4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
5. MOLD FLASH SHALL NOT EXCEED 0.254mm
6. JEDEC PACKAGE REFERENCE IS MO-193
3. DIMENSIONS ARE INCLUSIVE OF PLATING
3801f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.
11
LTC3801/LTC3801B
U
TYPICAL APPLICATIO
550kHz Micropower Step-Down DC/DC Controller
LTC3801 No-Load IQ vs Input Voltage*
220pF
25
V
IN
10k
V
= 2.5V
OUT
2.7V TO 9.8V
I
/RUN
LTC3801/
V
FRONT PAGE APPLICATION
TH
IN
–
10µF
0.025Ω
23
21
19
17
15
LTC3801B
GND
SENSE
V
PGATE
FB
402k
4.7µH
V
2.5V
2A
866k
OUT
+
47µF
3801 TA01
3
4
5
6
7
8
9
10
V
IN
INPUT VOLTAGE (V)
3801 TA04
*SEE THE FRONT PAGE OF THIS DATA SHEET FOR THE EFFICIENCY vs LOAD CURRENT CURVE
RELATED PARTS
PART NUMBER
LTC1147 Series
LTC1622
DESCRIPTION
COMMENTS
High Efficiency Step-Down Switching Regulator Controllers
Low Input Voltage Current Mode Step-Down DC/DC Controller
100% Duty Cycle, 3.5V ≤ V ≤ 16V
IN
V
2V to 10V, I
Up to 4.5A, Synchronizable to
IN
OUT
750kHz Optional Burst Mode Operation, 8-Lead MSOP
LTC1624
LTC1625
LTC1702A
LTC1733
High Efficiency SO-8 N-Channel Switching Regulator Controller
N-Channel Drive, 3.5V ≤ V ≤ 36V
IN
No R
TM Synchronous Step-Down Regulator
97% Efficiency, No Sense Resistor
SENSE
550kHz, 2 Phase, Dual Synchronous Controller
Li-Ion Linear Battery Charger
Two Channels; Minimum C and C , I
up to 15A
IN
OUT OUT
Standalone Charger with Charge Termination, Integrated
MOSFET, Thermal Regulator Prevents Overheating
LT®1765
LTC1771
25V, 2.75A (I ), 1.25MHz Step-Down Converter
3V ≤ V ≤ 25V, V
≥ 1.2V, SO-8 and TSSOP16 Packages
OUT
OUT
IN
Ultra-Low Supply Current Step-Down DC/DC Controller
10µA Supply Current, 93% Efficiency,
1.23V ≤ V ≤ 18V; 2.8V ≤ V ≤ 20V
OUT
IN
LTC1772/LTC1772B 550kHz ThinSOT Step-Down DC/DC Controllers
LTC1778/LTC1778-1 No R Current Mode Synchronous Step-Down Controllers
2.5V ≤ V ≤ 9.8V, V
≥ 0.8V, I
≤ 6A
IN
OUT
OUT
4V ≤ V ≤ 36V, 0.8V ≤ V
≤ (0.9)(V ), I
Up to 20A
SENSE
IN
OUT
IN OUT
LTC1779
250mA Monolithic Step-Down Converter in ThinSOT
2.5V ≤ V ≤ 9.8V, 550kHz, V
≥ 0.8V
IN
OUT
LTC1872/LTC1872B 550kHz ThinSOT Step-Up DC/DC Controllers
2.5V ≤ V ≤ 9.8V; 90% Efficiency
IN
LTC3411/LTC3412
LTC3440
1.25/2.5A Monolithic Synchronous Step-Down Converter
600mA (I ), 2MHz Synchronous Buck-Boost DC/DC Converter
95% Efficiency, 2.5V ≤ V ≤ 5.5V, V
TSSOP16 Exposed Pad Package
≥ 0.8V,
OUT
IN
2.5V ≤ V ≤ 5.5V, Single Inductor
OUT
IN
No R
is a trademark of Linear Technology Corporation.
SENSE
3801f
LT/TP 1103 1K • PRINTED IN THE USA
LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
12
●
●
(408) 432-1900 FAX: (408) 434-0507 www.linear.com
LINEAR TECHNOLOGY CORPORATION 2003
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