LTC4006EGN-2 [Linear]
4A, High Efficiency, Standalone Li-Ion Battery Charger; 4A ,高效,独立锂离子电池充电器型号: | LTC4006EGN-2 |
厂家: | Linear |
描述: | 4A, High Efficiency, Standalone Li-Ion Battery Charger |
文件: | 总16页 (文件大小:221K) |
中文: | 中文翻译 | 下载: | 下载PDF数据表文档文件 |
Final Electrical Specifications
LTC4006
4A, High Efficiency,
Standalone Li-Ion Battery Charger
May 2003
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FEATURES
DESCRIPTIO
The LTC®4006 is a complete constant-current/constant-
voltage charger controller for 2-, 3- or 4-cell lithium bat-
teries in a small package using few external components.
ThePWMcontrollerisasynchronous, quasi-constantfre-
■
Complete Charger Controller for 2-, 3- or 4-Cell
Lithium-Ion Batteries
High Conversion Efficiency: Up to 96%
Output Currents Exceeding 4A
■
■
■
■
■
■
±0.8% Accurate Preset Voltages: 8.4V, 12.6V, 16.8V quency, constant off-time architecture that will not gener-
Built-In Charge Termination with Automatic Restart
AC Adapter Current Limiting Maximizes Charge Rate*
Automatic Conditioning of Deeply Discharged
Batteries
ate audible noise even when using ceramic capacitors.
TheLTC4006isavailablein8.4V,12.6Vand16.8Vversions
with ±0.8% accuracy. Charging current is programmable
withasinglesenseresistorto±4%typicalaccuracy.Charg-
ing current can be monitored as a representative voltage at
the IMON pin. A timer, programmed by an external resistor,
sets the total charge time or is reset to 25% of total charge
time after C/10 charging current is reached. Charging au-
tomaticallyresumeswhencellvoltagefallsbelow3.9V/cell.
■
■
■
■
■
Thermistor Input for Temperature Qualified Charging
Wide Input Voltage Range: 6V to 28V
0.5V Dropout Voltage; Maximum Duty Cycle: 98%
Programmable Charge Current: ±5% Accuracy
Indicator Outputs for Charging, C/10 Current
Detection and AC Adapter Present
Fully discharged cells are automatically trickle charged at
10% of the programmed current until the cell voltage ex-
ceeds 2.5V/cell. Charging terminates if the low-battery
condition persists for more than 25% of the total charge
time.
■
■
Charging Current Monitor Output
16-Pin Narrow SSOP Package
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APPLICATIO S
■
Notebook Computers
LTC4006 includes a thermistor sensor input that sus-
pends charging if an unsafe temperature condition is
detected and automatically resumes charging when bat-
tery temperature returns to within safe limits.
■
Portable Instruments
■
Battery-Backup Systems
Standalone Li-Ion Chargers
■
, LTC and LT are registered trademarks of Linear Technology Corporation.
*U.S. Patent No. 5,723,970
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TYPICAL APPLICATIO
4A Li-Ion Battery Charger
DCIN
0V TO 28V
3A
INPUT SWITCH
0.1µF
5k
V
LOGIC
DCIN
CHG
INFET
CLP
100k
0.033Ω
15nF
LTC4006
CHG
ACP
ACP/SHDN
CLN
20µF
10µH
CHARGING
I
TGATE
BGATE
PGND
CSP
MON
0.025Ω
CURRENT MONITOR
32.4k
NTC
BATTERY
20µF
R
T
0.0047µF
309k
TIMING
RESISTOR
(~2 HOURS)
THERMISTOR
I
TH
10k
NTC
6k
GND
BAT
0.47µF
4006 TA01
0.12µF
sn4006 4006is
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tation that the interconnection of its circuits as described herein will notinfringe onexisting patent rights.
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LTC4006
W W U W
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ABSOLUTE MAXIMUM RATINGS
PACKAGE/ORDER INFORMATION
(Note 1)
TOP VIEW
Voltage from DCIN, CLP, CLN, TGATE, INFET,
ACP/SHDN, CHG to GND ........................... +32V/–0.3V
CSP, BAT to GND....................................... +28V/–0.3V
RT to GND..................................................... +7V/–0.3V
NTC ............................................................ +10V/–0.3V
Operating Ambient Temperature Range
(Note 4) ............................................. –40°C to 85°C
Operating Junction Temperature ......... –40°C to 125°C
Storage Temperature Range ................. –65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
ORDER PART
DCIN
CHG
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
INFET
BGATE
PGND
TGATE
CLN
NUMBER
LTC4006EGN-2
LTC4006EGN-4
LTC4006EGN-6
ACP/SHDN
R
T
GND
NTC
CLP
GN PART MARKING
I
BAT
TH
I
CSP
MON
40062
40064
40066
GN PACKAGE
16-LEAD PLASTIC SSOP
TJMAX = 125°C, θJA = 110°C/W
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating
temperature range (Note 4), otherwise specifications are at TA = 25°C. VDCIN = 20V, VBAT = 12V unless otherwise noted.
SYMBOL PARAMETER
DCIN Operating Range
CONDITIONS
MIN
TYP
MAX
28
UNITS
V
6
I
DCIN Operating Current
Voltage Accuracy
Sum of Current from CLP, CLN, DCIN
3
5
mA
DCIN
V
(Note 2)
TOL
LTC4006-6
LTC4006-6
LTC4006-2
LTC4006-2
LTC4006-4
LTC4006-4
8.333
8.316
12.499
12.474
16.665
16.632
8.4
8.4
12.6
12.6
16.8
16.8
8.467
8.484
12.700
12.726
16.935
16.968
V
V
V
V
V
V
●
●
●
I
Current Accuracy (Note 3)
V
CSP
V
BAT
V
BAT
V
BAT
– V Target = 100mV
BAT
–4
–5
4
5
%
%
TOL
= 11.5V (LTC4006-2)
= 7.6V (LTC4006-6)
= 12V (LTC4006-4)
●
V
< 6V, V
– V Target = 10mV
BAT
–60
–40
60
40
%
%
BAT
CSP
6V ≤ V
≤ V
,
LOBAT
BAT
– V
V
CSP
Target = 10mV
BAT
T
Termination Timer Accuracy
Battery Leakage Current
R
= 270k
RT
●
–15
15
%
TOL
Shutdown
DCIN = 0V
DCIN = 0V
DCIN = 20V, V
15
20
0
30
45
10
●
●
µA
µA
= 0V
= 0V
–10
4.2
1
SHDN
UVLO
Undervoltage Lockout Threshold
Shutdown Threshold at ACP/SHDN
DCIN Current in Shutdown
DCIN Rising, V
●
●
4.7
5.5
2.5
3
V
V
BAT
V
= 0V, Sum of Current from CLP,
2
mA
SHDN
CLN, DCIN
Current Sense Amplifier, CA1
Input Bias Current Into BAT Pin
CA1/I Input Common Mode Low
11.67
µA
V
CMSL
CMSH
●
●
0
1
CA1/I Input Common Mode High
V
– 0.2
CLN
V
1
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LTC4006
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating
temperature range (Note 4), otherwise specifications are at TA = 25°C. VDCIN = 20V, VBAT = 12V unless otherwise noted.
SYMBOL PARAMETER
Current Comparators I
CONDITIONS
= 2.5V
MIN
TYP
MAX
UNITS
and I
CMP
REV
I
I
Maximum Current Sense Threshold (V
– V
)
BAT
V
ITH
●
140
165
–30
200
mV
mV
TMAX
TREV
CSP
Reverse Current Threshold (V
– V
)
BAT
CSP
Current Sense Amplifier, CA2
Transconductance
Source Current
1
mmho
µA
Measured at I , V = 1.4V
–40
40
TH ITH
Sink Current
Measured at I , V = 1.4V
µA
TH ITH
Current Limit Amplifier
Transconductance
1.5
100
100
mmho
mV
V
Current Limit Threshold
CLP Input Bias Current
●
93
107
110
CLP
CLP
I
nA
Voltage Error Amplifier, EA
Transconductance
Sink Current
1
mmho
µA
Measured at I , V = 1.4V
36
TH ITH
OVSD
Overvoltage Shutdown Threshold as a Percent
of Programmed Charger Voltage
●
●
102
0
107
%
Input P-Channel FET Driver (INFET)
DCIN Detection Threshold (V
– V
)
CLN
DCIN Voltage Ramping Up
0.17
0.25
50
V
DCIN
from V
– 0.1V
CLN
Forward Regulation Voltage (V
– V
)
CLN
●
●
●
25
–25
5.8
mV
mV
V
DCIN
Reverse Voltage Turn-Off Voltage (V
– V
)
CLN
DCIN Voltage Ramping Down
–60
5
DCIN
INFET “On” Clamping Voltage (V
INFET “Off” Clamping Voltage (V
– V
– V
)
I
I
= 1µA
6.5
DCIN
DCIN
INFET
INFET
INFET
INFET
)
= –25µA
0.25
V
Thermistor
NTCVR
Reference Voltage During Sample Time
High Threshold
4.5
V
V
V
V
V
Rising
Falling
≤ 10V
●
●
NTCVR NTCVR NTCVR
• 0.48 • 0.5 • 0.52
NTC
NTC
NTC
Low Threshold
NTCVR NTCVR NTCVR
• 0.115 • 0.125 • 0.135
V
Thermistor Disable Current
10
µA
Indicator Outputs (ACP/SHDN, CHG)
C10TOL
LBTOL
C/10 Indicator Accuracy
Voltage Falling at PROG
●
0.375
0.400
0.425
V
LOBAT Threshold Accuracy
LTC4006-6
LTC4006-2
LTC4006-4
●
●
●
4.70
7.27
9.70
4.93
7.5
10
5.14
7.71
10.28
V
V
V
RESTART Threshold Accuracy
LTC4006-6
LTC4006-2
LTC4006-4
●
●
●
7.5
11.35
15.15
7.7
11.7
15.6
7.96
11.94
15.92
V
V
V
V
V
Low Logic Level of ACP/SHDN, CHG
High Logic Level of ACP/SHDN
Pull-Up Current on ACP/SHDN
C/10 Indicator Sink Current from CHG
Off State Leakage Current of CHG
Timer Defeat Threshold at CHG
I
I
= 100µA
= –1µA
●
●
0.5
V
V
OL
OH
OL
OH
2.7
I
V = 0V
–10
25
µA
µA
µA
PO
IC10
V
V
= 3V
= 3V
●
15
–1
1
38
1
OH
OH
I
OFF
V
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LTC4006
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating
temperature range (Note 4), otherwise specifications are at TA = 25°C. VDCIN = 20V, VBAT = 12V unless otherwise noted.
SYMBOL PARAMETER
Oscillator
CONDITIONS
MIN
TYP
MAX
UNITS
f
f
Regulator Switching Frequency
255
20
300
25
345
kHz
kHz
%
OSC
MIN
Regulator Switching Frequency in Drop Out
Regulator Maximum Duty Cycle
Duty Cycle ≥ 98%
DC
V
= V
BAT
98
99
MAX
CSP
Gate Drivers (TGATE, BGATE)
V
V
V
V
High (V
High
– V
)
I
= –1mA
= 3000pF
= 3000pF
= 1mA
50
10
10
50
mV
V
TGATE
BGATE
TGATE
BGATE
CLN
TGATE
TGATE
C
C
4.5
4.5
5.6
5.6
LOAD
LOAD
Low (V
Low
– V
)
V
CLN
TGATE
I
mV
BGATE
TGATE Transition Time
TGATE Rise Time
TGATE Fall Time
TGTR
TGTF
C
C
= 3000pF, 10% to 90%
= 3000pF, 10% to 90%
50
50
110
100
ns
ns
LOAD
LOAD
BGATE Transition Time
BGATE Rise Time
BGATE Fall Time
BGTR
BGTF
C
C
= 3000pF, 10% to 90%
= 3000pF, 10% to 90%
40
40
90
80
ns
ns
LOAD
LOAD
V
V
at Shutdown (V
at Shutdown
– V
)
I
I
= –1µA, DCIN = 0V, CLN = 12V
= 1µA, DCIN = 0V, CLN = 12V
100
100
mV
mV
TGATE
BGATE
CLN
TGATE
TGATE
BGATE
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: See Test Circuit
Note 3: Does not include tolerance of current sense resistor.
Note 4: The LTC4006E is guaranteed to meet performance specifications
from 0°C to 70°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.
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PI FU CTIO S
DCIN (Pin 1): External DC Power Source Input. Bypass
this pin with at least 0.01µF. See Applications Information
section.
ACP/SHDN (Pin 3): Open-Drain Output Used to Indicate if
the AC Adapter Voltage is Adequate for Charging. Active
high digital output. Internal 10µA pull-up to 3.5V. The
charger can also be inhibited by pulling this pin below 1V.
Reset the charger by pulsing the pin low for a minimum of
0.1µs.
CHG (Pin 2): Open-Drain Charge Status Output. When the
battery is being charged, the CHG pin is pulled low by an
internal N-channel MOSFET. When the charge current
drops below 10% of programmed current, the N-channel
MOSFET turns off and a 25µA current source is connected
from the CHG pin to GND. When the timer runs out or the
inputsupplyisremoved,thecurrentsourcewillbediscon-
nected and the CHG pin is forced into a high impedance
state. A pull-up resistor is required. The timer function is
defeated by forcing this pin below 1V (or connecting it to
GND).
RT (Pin 4): Timer Resistor. The timer period is set by
placing a resistor, RRT , to GND.
The timer period is tTIMER = (1hour • RRT/154k)
If this resistor is not present, the charger will not start.
GND (Pin 5): Ground for Low Power Circuitry.
sn4006 4006is
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LTC4006
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PI FU CTIO S
NTC (Pin 6): A thermistor network is connected from NTC
to GND. This pin determines if the battery temperature is
safe for charging. The charger and timer are suspended if
the thermistor indicates a temperature that is unsafe for
charging. The thermistor function may be disabled with a
300k to 500k resistor from DCIN to NTC.
BAT (Pin 10): Battery Sense Input and the Negative
Reference for the Current Sense Resistor. A precision
internal resistor divider sets the final float potential on this
pin.Theresistordividerisdisconnectedduringshutdown.
CLP(Pin11):PositiveInputtotheSupplyCurrentLimiting
Amplifier, CL1. The threshold is set at 100mV above the
voltage at the CLN pin. When used to limit supply current,
a filter is needed to filter out the switching noise. If no
current limit function is desired, connect this pin to CLN.
ITH (Pin 7): Control Signal of the Inner Loop of the Current
Mode PWM. Higher ITH voltage corresponds to higher
charging current in normal operation. A 6.04k resistor, in
series with a capacitor of at least 0.1µF to GND, provides
loop compensation. Typical full-scale output current is
40µA. Nominal voltage range for this pin is 0V to 3V.
CLN (Pin 12): Negative Reference for the Input Current
Limit Amplifier, CL1. This pin also serves as the power
supply for the IC. A 10µF to 22µF bypass capacitor should
be connected as close as possible to this pin.
IMON (Pin 8): Current Monitoring Output. The voltage at
this pin provides a linear indication of charging current.
Peak current is equivalent to 1.19V. Zero current is ap-
proximately 0.309V. A capacitor from IMON to ground is
TGATE(Pin13):DrivesthetopexternalP-channelMOSFET
of the battery charger buck converter.
PGND (Pin 14):HighCurrentGroundReturnfortheBGATE
Driver.
required to filter higher frequency components. If VBAT
<
2.5V/cell, then IMON = 1.19V when conditioning a depleted
battery.
BGATE (Pin 15): Drives the bottom external N-channel
MOSFET of the battery charger buck converter.
CSP (Pin 9): Current Amplifier CA1 Input. This pin and the
BAT pin measure the voltage across the sense resistor,
RSENSE, to provide the instantaneous current signals re-
quired for both peak and average current mode operation.
INFET(Pin16):DrivestheGateoftheExternalInputPFET.
TEST CIRCUIT
LTC4006
11.67µA
+
V
REF
+
EA
–
3k
35mV
–
I
BAT
10
TH
7
+
LT1055
0.6V
–
4006 TC
sn4006 4006is
5
LTC4006
W
BLOCK DIAGRA
V
IN
100k
CHG
25µA
DCIN
V
1
2
LOGIC
0.01µF
5.8V
INFET
Q3
16
3
CLN
ACP/SHDN
R
RT
R
T
OSCILLATOR
4
6
TIMER/CONTROLLER
I
CL
32.4k
T
BAD
NTC
THERMISTOR
RESTART
LOBAT
10k
NTC
1.105V
708mV
0.47µF
397mV
C/10
35mV
–
+
GND
5
3k
BAT
CSP
10
9
11.67µA
R
SENSE
20µF
–
+
Ω
1.19V
+
3k
g
= 1m
m
CA1
EA
–
5.1k
CLP
CLN
Ω
11
12
–
g
= 1.5m
m
9k
R
CL
15nF
CL1
100mV
Ω
–
g
= 1m
m
+
CA2
+
1.19V
DCIN
I
TH
OSCILLATOR
WATCH DOG
DETECT
7
6.04k
0.12µF
20µF
BUFFERED I
t
TH
OV
OFF
÷5
1.28V
TGATE
S
R
+
–
+
–
13
Q1
Q2
Q
I
CMP
BGATE
PGND
CHARGE
PWM
LOGIC
15
14
–
+
I
REV
17mV
I
MON
8
L1
R
IMON1
26.44k
4.7nF
R
IMON2
52.87k
4006 BD
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OPERATIO
Overview
(LTC4006-2)or16.8V(LTC4006-4)withbetterthan±0.8%
accuracy. Charging begins when the potential at the DCIN
pin rises above the voltage at CLN (and the UVLO voltage)
and the ACP/SHDN pin is allowed to go high; the CHG pin
is set low. At the beginning of the charge cycle, if the cell
voltage is below 2.5V, the charger will trickle charge the
The LTC4006 is a synchronous current mode PWM step-
down(buck)switcherbatterychargercontroller.Thecharge
current is programmed by the sense resistor (RSENSE
between the CSP and BAT pins. The final float voltage is
internally programmed to 8.4V (LTC4006-6), 12.6V
)
battery with 10% of the maximum programmed current.
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LTC4006
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OPERATIO
If the cell voltage stays below 2.5V for 25% of the total
charge time, the charge sequence will be terminated im-
mediately and the CHG pin will be set to a high impedance.
DCIN and CLN is ever less than –25mV, then the input FET
is turned off in less than 10µs to prevent significant
reverse current from flowing in the input FET. In this
condition, theACP/SHDNpinisdrivenlowandthecharger
is disabled.
An external thermistor network is sampled at regular
intervals. If the thermistor value exceeds design limits,
charging is suspended. If the thermistor value returns to
an acceptable value, charging resumes. An external resis-
tor on the RT pin sets the total charge time. The timer can
be defeated by forcing the CHG pin to a low voltage.
Battery Charger Controller
The LTC4006 charger controller uses a constant off-time,
current mode step-down architecture. During normal op-
eration, the top MOSFET is turned on each cycle when the
oscillator sets the SR latch and turned off when the main
currentcomparatorICMP resetstheSRlatch. Whilethetop
MOSFET is off, the bottom MOSFET is turned on until
either the inductor current trips the current comparator
IREV or the beginning of the next cycle. The oscillator uses
the equation:
As the battery approaches the final float voltage, the
charge current will begin to decrease. When the current
drops to 10% of the programmed charge current, an
internal C/10 comparator will indicate this condition by
sinking25µAattheCHGpin. Thechargetimerisalsoreset
to 25% of the total charge time. If this condition is caused
by an input current limit condition, described below, then
the C/10 comparator will be inhibited. When a time-out
occurs, charging is terminated immediately and the CHG
pin changes to a high impedance. The charger will auto-
matically restart if the cell voltage is less than 3.9V. To
restartthechargecyclemanually,simplyremovetheinput
voltage and reapply it, or force the ACP/SHDN pin low
momentarily. When the input voltage is not present, the
charger goes into a sleep mode, dropping battery current
drainto15µA.Thisgreatlyreducesthecurrentdrainonthe
batteryandincreasesthestandbytime.Thechargercanbe
inhibited at any time by forcing the ACP/SHDN pin to a low
voltage.
VDCIN – VBAT
VDCIN • fOSC
tOFF
=
to set the bottom MOSFET on time. This activity is dia-
grammed in Figure 1.
OFF
TGATE
ON
ON
t
BGATE
OFF
OFF
TRIP POINT SET BY ITH VOLTAGE
INDUCTOR
CURRENT
Input FET
4006 F01
The input FET circuit performs two functions. It enables
the charger if the input voltage is higher than the CLN pin
and provides the logic indicator of AC present on the
ACP/SHDNpin. ItcontrolsthegateoftheinputFETtokeep
a low forward voltage drop when charging and also
prevents reverse current flow through the input FET.
Figure 1
The peak inductor current, at which ICMP resets the SR
latch, is controlled by the voltage on ITH. ITH is in turn
controlled by several loops, depending upon the situation
at hand. The average current control loop converts the
voltage between CSP and BAT to a representative current.
Error amp CA2 compares this current against the desired
current programmed by RIMON at the IMON pin and adjusts
ITH until:
If the input voltage is less than VCLN, it must go at least
170mVhigherthanVCLN toactivatethecharger.Whenthis
occurs the ACP/SHDN pin is released and pulled up with
an internal load to indicate that the adapter is present. The
gateoftheinputFETisdriventoavoltagesufficienttokeep
a low forward voltage drop from drain to source. If the
voltage between DCIN and CLN drops to less than 25mV,
the input FET is turned off slowly. If the voltage between
VREF
VCSP – VBAT +11.67µA •3kΩ
=
R
IMON
3kΩ
sn4006 4006is
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LTC4006
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OPERATIO
Table 1. Truth Table for LTC4006 Operation
MODE
DCIN
<BAT
>BAT
BAT VOLTAGE
>UVLO
BAT CURRENT
ACP/SHDN
LOW
TIMER STATE
Reset
CHG*
HIGH
LOW
Shut Down by Low Adapter Voltage
Conditioning a Depleted Battery
Leakage
<2.5V/Cell
10% Programmed
Current
HIGH
Running
Normal Charging
>BAT
>2.5V/Cell
Programmed
Current
HIGH
Running
LOW
LOW
Input Current Limited Charging
>BAT
>BAT
>2.5V/Cell
X
Unknown
OFF
HIGH
HIGH
Running
Paused
Charger Paused Due to Thermistor Out of Range
LOW
(Faulted)
Shut Down by ACP/SHDN Pin
>BAT
>BAT
X
OFF
OFF
Forced LOW
HIGH
Reset
HIGH
Terminated by Low-Battery Fault (Note 1)
<2.5V/Cell
>T/4 Stopped
HIGH
(Faulted)
Top-Off Charging. C/10 is Latched
>BAT
>BAT
>BAT
X
V
V
OFF
OFF
OFF
HIGH
HIGH
HIGH
<T/4 After C/10
Comparator Trip.
Running
25µA
FLOAT
FLOAT
Timer is Reset by C/10 Comparator (Latched),
then Terminates After 1/4 T
>T/4 After C/10
Comparator Trip.
Stopped
HIGH
(Waiting
for Restart)
Terminated by Expired Timer
V
FLOAT
**
>T Stopped
HIGH
(Waiting
for Restart)
Timer Defeated. (Low-Battery Conditioning Still
Functional)
X
X
X
X
Forced LOW
HIGH**
LOW
Shut Down by Undervoltage Lockout
>BAT
and <UVL
<UVL
2.5V ≤ V
OFF
HIGH
HIGH
Reset
Timer Defeated Until V
> 3.9V/Cell
>BAT
≤3.9V
BAT
Programmed
Current
Running
BAT
(V/Cell)
*Open Drain. High when used with pull-up resistor.
**Most probable condition, X = Don’t care
Note 1: If a depleted battery is inserted while the charger is in this state,
the charger must be reset to initiate charging.
will be inhibited if it is not already active. If the charging
current decreases below 10% to 15% of programmed
current, while engaged in input current limiting, BGATE
will be forced low to prevent the charger from discharging
the battery. Audible noise can occur in this mode of
operation.
therefore,
VREF
3kΩ
RSENSE
ICHARGE
=
– 11.67µA •
R
IMON
The voltage at BAT is divided down by an internal resistor
divider and is used by error amp EA to decrease ITH if the
divider voltage is above the 1.19V reference. When the
charging current begins to decrease, the voltage at IMON
will decrease in direct proportion. The voltage at IMON is
then given by:
An overvoltage comparator guards against voltage tran-
sient overshoots (>7% of programmed value). In this
case, both MOSFETs are turned off until the overvoltage
condition is cleared. This feature is useful for batteries
which “load dump” themselves by opening their protec-
tion switch to perform functions such as calibration or
pulse mode charging.
R
3kΩ
IMON
V
IMON
= ICHARGE •RSENSE +11.67µA •3kΩ •
(
)
VIMON is plotted in Figure 2.
As the voltage at BAT increases to near the input voltage
at DCIN, the converter will attempt to turn on the top
MOSFET continuously (“dropout’’). A watchdog timer
detects this condition and forces the top MOSFET to turn
The amplifier CL1 monitors and limits the input current to
a preset level (100mV/RCL). At input current limit, CL1 will
decrease the ITH voltage, thereby reducing charging cur-
rent. When this condition is detected, the C/10 indicator
sn4006 4006is
8
LTC4006
U
OPERATIO
1.2
1.0
0.8
0.6
0.4
0.2
0
LTC4006
NTC
CLK
1.19V
R9
–
32.4k
6
~4.5V
R
S1
TH
C7
0.47µF
+
10k
NTC
60k
+
–
0.309V
20
0
40
60
80
100
I
(% OF MAXIMUM CURRENT)
CHARGE
45k
15k
4006 F02
–
+
Figure 2. IMON vs ICHARGE
off for about 300ns at 40µs intervals. This is done to
prevent audible noise when using ceramic capacitors at
the input and output.
T
D
C
Q
BAD
4006 F03
Charger Startup
Figure 3
When the charger is enabled, it will not begin switching
untiltheITH voltageexceedsathresholdthatassuresinitial
current will be positive. This threshold is 5% to 15% of the
maximum programmed current. After the charger begins
switching, the various loops will control the current at a
level that is higher or lower than the initial current. The
duration of this transient condition depends upon the loop
compensation but is typically less than 100µs.
This voltage is stored by C7. Then the switch is opened for
a short period of time to read the voltage across the
thermistor.
tHOLD = 10 • RRT • 17.5pF = 54µs,
for RRT = 309k
When the tHOLD interval ends the result of the thermistor
testing is stored in the D flip-flop (DFF). If the voltage at
NTC is within the limits provided by the resistor divider
feeding the comparators, then the NOR gate output will be
low and the DFF will set TBAD to zero and charging will
continue. If the voltage at NTC is outside of the resistor
dividerlimits, thentheDFFwillsetTBAD toone, thecharger
will be shut down, and the timer will be suspended until
TBAD returns to zero (see Figure 4).
Thermistor Detection
The thermistor detection circuit is shown in Figure 3. It
requires an external resistor and capacitor in order to
function properly.
Thethermistordetectorperformsasample-and-holdfunc-
tion. An internal clock, whose frequency is determined by
the timing resistor connected to RT, keeps switch S1
closed to sample the thermistor:
CLK
(NOT TO
SCALE)
tSAMPLE = 127.5 • 20 • RRT • 17.5pF = 13.8ms,
for RRT = 309k
t
SAMPLE
t
HOLD
The external RC network is driven to approximately 4.5V
and settles to a final value across the thermistor of:
VOLTAGE ACROSS THERMISTOR
COMPARATOR HIGH LIMIT
COMPARATOR LOW LIMIT
V
NTC
4.5V •RTH
VRTH(FINAL)
=
R
TH + R9
4006 F04
Figure 4
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Alternatively, a normally closed switch can be used to
detect when the battery is present (see Figure 8).
Charger Current Programming
The basic formula for charging current is:
200
180
160
140
120
100
80
100mV
RSENSE
ICHARGE(MAX)
=
Table 2. Recommended RSENSE Resistor Values
(A) (Ω) 1%
I
R
SENSE
R
(W)
SENSE
MAX
1.0
0.100
0.25
0.25
0.5
2.0
3.0
4.0
0.050
0.033
0.025
60
40
20
0.5
0
100
300
400 450
500
150 200 250
350
Setting the Timer Resistor
R
RT
(kΩ)
4006 F05
The charger termination timer is designed for a range of
1hour to 3 hour with a ±15% uncertainty. The timer is
programmed by the resistor RRT using the following
equation:
Figure 5. tTIMER vs RRT
3.3V
tTIMER = 227 • RRT • 175pF (Refer to Figure 5)
V
DD
200k
LTC4006
CHG
µP
OUT
IN
33k
It is important to keep the parasitic capacitance on the RT
pin to a minimum. The trace connecting RT to RRT should
be as short as possible.
4006 F06
Figure 6. Microprocessor Interface
CHG Status Output Pin
When the charge cycle starts, the CHG pin is pulled down
to ground by an internal N-channel MOSFET that can drive
more than 100µA. When the charge current drops to 10%
of the full-scale current (C/10), the N-channel MOSFET is
turned off and a weak 25µA current source to ground is
connected to the CHG pin. After a time out occurs, the pin
will go into a high impedance state. By using two different
value pull-up resistors, a microprocessor can detect three
states from this pin (charging, C/10 and stop charging).
See Figure 6.
LTC4006
ADAPTER
POWER
DCIN
470k
ACP/SHDN
4006 F07
SWITCH CLOSED IF
BATTERY CONNECTED
Figure 7
Battery Detection
LTC4006
ADAPTER
POWER
DCIN
It is generally not good practice to connect a battery while
the charger is running. The timer is in an unknown state
and the charger could provide a large surge current into
the battery for a brief time. The circuit shown in Figure 7
keeps the charger shut down and the timer reset while a
battery is not connected.
ACP/SHDN
4006 F08
SWITCH OPEN WHEN
BATTERY CONNECTED
Figure 8
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Soft-Start
VBAT
VDCIN
0.29 V
1–
(
)
BAT
The LTC4006 is soft started by the 0.12µF capacitor on the
IRMS
=
I
TH pin.Onstart-up,ITH pinvoltagewillrisequicklyto0.5V,
L1 f
( )( )
then ramp up at a rate set by the internal 40µA pull-up
current and the external capacitor. Battery charging For example:
current starts ramping up when ITH voltage reaches 0.8V
VDCIN = 19V, VBAT = 12.6V, L1 = 10µH, and
and full current is achieved with ITH at 2V. With a 0.12µF
capacitor, time to reach full charge current is about 2ms
and it is assumed that input voltage to the charger will
reach full value in less than 2ms. The capacitor can be
increased up to 1µF if longer input start-up times are
needed.
f = 300kHz, IRMS = 0.41A.
EMI considerations usually make it desirable to minimize
ripple current in the battery leads, and beads or inductors
maybeaddedtoincreasebatteryimpedanceatthe300kHz
switching frequency. Switching ripple current splits be-
tween the battery and the output capacitor depending on
theESRoftheoutputcapacitorandthebatteryimpedance.
If the ESR of C3 is 0.2Ω and the battery impedance is
raised to 4Ω with a bead or inductor, only 5% of the
current ripple will flow in the battery.
Input and Output Capacitors
The input capacitor (C2) is assumed to absorb all input
switching ripple current in the converter, so it must have
adequate ripple current rating. Worst-case RMS ripple
currentwillbeequaltoonehalfofoutputchargingcurrent.
Actual capacitance value is not critical. Solid tantalum low
ESR capacitors have high ripple current rating in a rela-
tively small surface mount package, but caution must be
used when tantalum capacitors are used for input or
output bypass. High input surge currents can be created
when the adapter is hot-plugged to the charger or when a
battery is connected to the charger. Solid tantalum capaci-
tors have a known failure mechanism when subjected to
very high turn-on surge currents. Only Kemet T495 series
of “Surge Robust” low ESR tantalums are rated for high
surge conditions such as battery to ground.
Inductor Selection
Higher operating frequencies allow the use of smaller
inductor and capacitor values. A higher frequency gener-
ally results in lower efficiency because of MOSFET gate
charge losses. In addition, the effect of inductor value on
ripple current and low current operation must also be
considered. The inductor ripple current ∆IL decreases
with higher frequency and increases with higher VIN.
1
VOUT
V
IN
∆IL =
VOUT 1–
f L
( )( )
The relatively high ESR of an aluminum electrolytic for C1,
located at the AC adapter input terminal, is helpful in
reducing ringing during the hot-plug event. Refer to Appli-
cation Note 88 for more information.
Accepting larger values of ∆IL allows the use of low
inductances, but results in higher output voltage ripple
and greater core losses. A reasonable starting point for
setting ripple current is ∆IL = 0.4(IMAX). In no case should
∆IL exceed 0.6(IMAX) due to limits imposed by IREV and
CA1. Remember the maximum ∆IL occurs at the maxi-
mum input voltage. In practice 10µH is the lowest value
recommended for use.
Highest possible voltage rating on the capacitor will mini-
mize problems. Consult with the manufacturer before use.
Alternatives include new high capacity ceramic (at least
20µF) from Tokin, United Chemi-Con/Marcon, et al. Other
alternative capacitors include OS-CON capacitors from
Sanyo.
Lower charger currents generally call for larger inductor
values. Use Table 3 as a guide for selecting the correct
inductor value for your application.
The output capacitor (C3) is also assumed to absorb
output switching current ripple. The general formula for
capacitor current is:
sn4006 4006is
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APPLICATIO S I FOR ATIO
Table 3
highest at high input voltages. For VIN < 20V the high
currentefficiencygenerallyimproveswithlargerMOSFETs,
while for VIN > 20V the transition losses rapidly increase
to the point that the use of a higher RDS(ON) device with
lower CRSS actually provides higher efficiency. The syn-
chronous MOSFET losses are greatest at high input volt-
age or during a short circuit when the duty cycle in this
switch in nearly 100%. The term (1 + δ∆T) is generally
given for a MOSFET in the form of a normalized RDS(ON) vs
temperature curve, but δ = 0.005/°C can be used as an
approximation for low voltage MOSFETs. CRSS is usually
specified in the MOSFET characteristics; if not, then CRSS
can be calculated using CRSS = QGD/∆VDS. The constant
k = 2 can be used to estimate the contributions of the two
terms in the main switch dissipation equation.
MAXIMUM
AVERAGE CURRENT (A)
INPUT
VOLTAGE (V)
MINIMUM INDUCTOR
VALUE (µH)
1
1
2
2
3
3
4
4
≤20
>20
≤20
>20
≤20
>20
≤20
>20
40 ±20%
56 ±20%
20 ±20%
30 ±20%
15 ±20%
20 ±20%
10 ±20%
15 ±20%
Charger Switching Power MOSFET
and Diode Selection
Two external power MOSFETs must be selected for use
with the charger: a P-channel MOSFET for the top (main)
switch and an N-channel MOSFET for the bottom (syn-
chronous) switch.
If the charger is to operate in low dropout mode or with a
high duty cycle greater than 85%, then the topside
P-channel efficiency generally improves with a larger
MOSFET.UsingasymmetricalMOSFETsmayachievecost
savings or efficiency gains.
The peak-to-peak gate drive levels are set internally. This
voltageistypically6V.Consequently,logic-levelthreshold
MOSFETs must be used. Pay close attention to the BVDSS
specification for the MOSFETs as well; many of the logic
level MOSFETs are limited to 30V or less.
The Schottky diode D1, shown in the Typical Application
on the back page, conducts during the dead-time between
the conduction of the two power MOSFETs. This prevents
thebodydiodeofthebottomMOSFETfromturningonand
storing charge during the dead-time, which could cost as
much as 1% in efficiency. A 1A Schottky is generally a
good size for 4A regulators due to the relatively small
average current. Larger diodes can result in additional
transition losses due to their larger junction capacitance.
SelectioncriteriaforthepowerMOSFETsincludethe“ON”
resistance RDS(ON), total gate capacitance QG, reverse
transfer capacitance CRSS, input voltage and maximum
output current. The charger is operating in continuous
mode at moderate to high currents so the duty cycles for
the top and bottom MOSFETs are given by:
The diode may be omitted if the efficiency loss can be
tolerated.
Main Switch Duty Cycle = VOUT/VIN
Synchronous Switch Duty Cycle = (VIN – VOUT)/VIN.
Calculating IC Power Dissipation
The MOSFET power dissipations at maximum output
current are given by:
The power dissipation of the LTC4006 is dependent upon
the gate charge of the top and bottom MOSFETs (QG1 and
QG2 respectively) The gate charge is determined from the
manufacturer’sdatasheetandisdependentuponboththe
gate voltage swing and the drain voltage swing of the
MOSFET. Use 6V for the gate voltage swing and VDCIN for
the drain voltage swing.
PMAIN = VOUT/VIN(I2MAX)(1 + δ∆T)RDS(ON)
+ k(V2IN)(IMAX)(CRSS)(fOSC
)
PSYNC = (VIN – VOUT)/VIN(I2MAX)(1 + δ∆T)RDS(ON)
Where δ is the temperature dependency of RDS(ON) and k
is a constant inversely related to the gate drive current.
Both MOSFETs have I2R losses while the PMAIN equation
includesanadditionaltermfortransitionlosses,whichare
PD = VDCIN • (fOSC (QG1 + QG2) + IDCIN
)
sn4006 4006is
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LTC4006
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APPLICATIO S I FOR ATIO
Example:
U
Table 5. Common RCL Resistor Values
ADAPTER
RATING (A)
R
VALUE*
R
POWER
R
POWER
CL
(Ω) 1%
CL
CL
VDCIN = 19V, fOSC = 345kHz, QG1 = QG2 = 15nC.
DISSIPATION (W)
RATING (W)
1.5
1.8
2
0.06
0.05
0.135
0.25
PD = 292mW
0.162
0.25
Adapter Limiting
0.045
0.039
0.036
0.033
0.03
0.18
0.25
2.3
2.5
2.7
3
0.206
0.25
An important feature of the LTC4006 is the ability to
automatically adjust charging current to a level which
avoids overloading the wall adapter. This allows the prod-
uct to operate at the same time that batteries are being
charged without complex load management algorithms.
Additionally, batteries will automatically be charged at the
maximum possible rate of which the adapter is capable.
0.225
0.5
0.241
0.5
0.27
0.5
* Values shown above are rounded to nearest standard value.
As is often the case, the wall adapter will usually have at
least a +10% current limit margin and many times one can
simply set the adapter current limit value to the actual
adapter rating (see Table 5).
This feature is created by sensing total adapter output
current and adjusting charging current downward if a
preset adapter current limit is exceeded. True analog
control is used, with closed-loop feedback ensuring that
adapter load current remains within limits. Amplifier CL1
in Figure 9 senses the voltage across RCL, connected
between the CLP and DCIN pins. When this voltage ex-
ceeds 100mV, the amplifier will override programmed
charging current to limit adapter current to 100mV/RCL. A
lowpass filter formed by 5kΩ and 15nF is required to
eliminate switching noise. If the current limit is not used,
CLP should be connected to CLN.
Designing the Thermistor Network
There are several networks that will yield the desired
function of voltage vs temperature needed for proper
operation of the thermistor. The simplest of these is the
voltage divider shown in Figure 10. Unfortunately, since
theHIGH/LOWcomparatorthresholdsarefixedinternally,
there is only one thermistor type that can be used in this
network; the thermistor must have a HIGH/LOW resis-
tance ratio of 1:7. If this happy circumstance is true for
you, then simply set R9 = RTH(LOW)
LTC4006
100mV
If you are using a thermistor that doesn’t have a 1:7 HIGH/
LOW ratio, or you wish to set the HIGH/LOW limits to
different temperatures, then the more generic network in
Figure 11 should work.
CLP
+
18
–
15nF
5k
CL1
+
AC ADAPTER
INPUT
R *
CL
CLN
V
IN
19
Once the thermistor, RTH, has been selected and the
thermistor value is known at the temperature limits, then
resistors R9 and R9A are given by:
+
C
IN
100mV
ADAPTER CURRENT LIMIT
*R
CL
=
4006 F09
Figure 9. Adapter Current Limiting
For NTC thermistors:
R9 = 6 RTH(LOW) • RTH(HIGH)/(RTH(LOW) – RTH(HIGH)
)
Setting Input Current Limit
R9A=6RTH(LOW) •RTH(HIGH)/(RTH(LOW) –7•RTH(HIGH)
For PTC thermistors:
)
)
To set the input current limit, you need to know the
minimum wall adapter current rating. Subtract 5% for the
input current limit tolerance and use that current to deter-
mine the resistor value.
R9 = 6 RTH(LOW) • RTH(HIGH)/(RTH(HIGH) – RTH(LOW)
)
R9A =6RTH(LOW)•RTH(HIGH)/(RTH(HIGH)–7• RTH(LOW)
RCL = 100mV/ILIM
ILIM = Adapter Min Current –
(Adapter Min Current • 5%)
sn4006 4006is
13
LTC4006
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APPLICATIO S I FOR ATIO
LTC4006
LTC4006
NTC
R9
R9
NTC
C7
R
C7
R9A
R
TH
TH
4006 F10
4006 F11
Figure 10. Voltage Divider Thermistor Network
Figure 11. General Thermistor Network
Example #1: 10kΩ NTC with custom limits
Disabling the Thermistor Function
TLOW = 0°C, THIGH = 50°C
RTH = 10k at 25°C,
If the thermistor is not needed, connecting a resistor
between DCIN and NTC will disable it. The resistor should
besizedtoprovideatleast10µAwiththeminimumvoltage
applied to DCIN and 10V at NTC. Do not exceed 30µA into
NTC. Generally, a 301k resistor will work for DCIN less
than 15V. A 499k resistor is recommended for DCIN
between 15V and 24V.
R
TH(LOW) = 32.582k at 0°C
RTH(HIGH) = 3.635k at 50°C
R9 = 24.55k → 24.3k (nearest 1% value)
R9A = 99.6k → 100k (nearest 1% value)
Example #2: 100kΩ NTC
TLOW = 5°C, THIGH = 50°C
PCB Layout Considerations
RTH = 100k at 25°C,
Formaximumefficiency,theswitchnoderiseandfalltimes
should be minimized. To prevent magnetic and electrical
field radiation and high frequency resonant problems,
proper layout of the components connected to the IC is
essential. (See Figure 12.) Here is a PCB layout priority list
forproperlayout.LayoutthePCBusingthisspecificorder.
RTH(LOW) = 272.05k at 5°C
RTH(HIGH) = 33.195k at 50°C
R9 = 226.9k → 226k (nearest 1% value)
R9A = 1.365M → 1.37M (nearest 1% value)
Example #3: 22kΩ PTC
TLOW = 0°C, THIGH = 50°C
RTH = 22k at 25°C,
RTH(LOW) = 6.53k at 0°C
RTH(HIGH) = 61.4k at 50°C
R9 = 43.9k → 44.2k (nearest 1% value)
R9A = 154k
1. Inputcapacitorsneedtobeplacedascloseaspossible
to switching FET’s supply and ground connections.
Shortest copper trace connections possible. These
parts must be on the same layer of copper. Vias must
not be used to make this connection.
2. ThecontrolICneedstobeclosetotheswitchingFET’s
gate terminals. Keep the gate drive signals short for a
clean FET drive. This includes IC supply pins that con-
nect to the switching FET source pins. The IC can be
placedontheoppositesideofthePCBrelativetoabove.
Sizing the Thermistor Hold Capacitor
During the hold interval, C7 must hold the voltage across
the thermistor relatively constant to avoid false readings.
A reasonable amount of ripple on NTC during the hold
interval is about 10mV to 15mV. Therefore, the value of C7
is given by:
3. Place inductor input as close as possible to switching
FET’s output connection. Minimize the surface area of
this trace. Make the trace width the minimum amount
needed to support current—no copper fills or pours.
Avoid running the connection using multiple layers in
parallel. Minimize capacitance from this node to any
other trace or plane.
C7 = tHOLD/(R9/7 • –ln(1 – 8 • 15mV/4.5V))
= 10 • RRT • 17.5pF/(R9/7 • –ln(1 – 8 • 15mV/4.5V)
Example:
R9 = 24.3k
RRT = 309k (~2 hour timer)
C7 = 0.58µF → 0.56µF (nearest value)
4. Place the output current sense resistor right next to
the inductor output but oriented such that the IC’s
sn4006 4006is
14
LTC4006
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APPLICATIO S I FOR ATIO
U
currentsensefeedbacktracesgoingtoresistorarenot
long. The feedback traces need to be routed together
asasinglepaironthesamelayeratanygiventimewith
smallest trace spacing possible. Locate any filter
componentonthesetracesnexttotheICandnotatthe
sense resistor location.
any other ground. Avoid using the system ground
plane. CAD trick: make analog ground a separate
ground net and use a 0Ω resistor to tie analog ground
to system ground.
9. A good rule of thumb for via count for a given high
current path is to use 0.5A per via. Be consistent.
5. Place output capacitors next to the sense resistor
output and ground.
10. If possible, place all the parts listed above on the same
PCB layer.
6. Output capacitor ground connections need to feed
into same copper that connects to the input capacitor
ground before tying back into system ground.
11. Copper fills or pours are good for all power connec-
tionsexceptasnotedaboveinRule3.Youcanalsouse
copper planes on multiple layers in parallel too—this
helps with thermal management and lower trace in-
ductance improving EMI performance further.
General Rules
7. Connection of switching ground to system ground or
internal ground plane should be single point. If the
system has an internal system ground plane, a good
way to do this is to cluster vias into a single star point
to make the connection.
12. For best current programming accuracy provide a
Kelvin connection from RSENSE to CSP and BAT. See
Figure 12 as an example.
It is important to keep the parasitic capacitance on the RT,
CSP and BAT pins to a minimum. The traces connecting
these pins to their respective resistors should be as short
as possible.
8. Route analog ground as a trace tied back to IC ground
(analog ground pin if present) before connecting to
SWITCH NODE
L1
DIRECTION OF CHARGING CURRENT
V
BAT
R
SNS
HIGH
FREQUENCY
CIRCULATING
PATH
C2
D1
V
IN
C3
BAT
4006 F13
CSP
BAT
4006 F12
Figure 12. High Speed Switching Path
Figure 13. Kelvin Sensing of Charging Current
U
PACKAGE DESCRIPTION
GN Package
16-Lead Plastic SSOP (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1641)
.189 – .196*
(4.801 – 4.978)
.009
(0.229)
REF
.015 ± .004
(0.38 ± 0.10)
.045 ±.005
16 15 14 13 12 11 10 9
.004 – .0098
× 45°
.053 – .068
(1.351 – 1.727)
(0.102 – 0.249)
.007 – .0098
(0.178 – 0.249)
0° – 8° TYP
.229 – .244
.0250
(0.635)
BSC
.150 – .157**
(3.810 – 3.988)
.016 – .050
(0.406 – 1.270)
.008 – .012
(0.203 – 0.305)
.254 MIN
.150 – .165
(5.817 – 6.198)
NOTE:
1. CONTROLLING DIMENSION: INCHES
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
INCHES
2. DIMENSIONS ARE IN
(MILLIMETERS)
GN16 (SSOP) 0502
3. DRAWING NOT TO SCALE
4
5
1
2
3
6
7
8
.0165 ±.0015
.0250 TYP
RECOMMENDED SOLDER PAD LAYOUT
sn4006 4006is
15
LTC4006
U
TYPICAL APPLICATIO
2A Li-Ion Battery Charger
Q3
DCIN
0V TO 20V
2.5A
INPUT SWITCH
D1: MBRM140T3
Q1, Q3: Si3457
Q2: Si3454
R1
5k
C1
0.1µF
V
LOGIC
R3
100k
C4
15nF
DCIN
CHG
INFET
CLP
R
CL
LTC4006
0.04Ω
CHG
ACP
TO LOAD
C2
20µF
ACP/SHDN
CLN
L1
22µH 2A
CHARGING
R
SENSE
0.05Ω
I
TGATE
BGATE
PGND
CSP
Q1
Q2
MON
CURRENT MONITOR
R9 32.4k
NTC
BATTERY
R
I
T
D1
C3
20µF
C5
THERMISTOR
R
T
0.0047µF
TH
10k
NTC
R4
6.04k
C6
309k
TIMING
GND
BAT
C7
RESISTOR
(~2 HOURS)
4006 TA02
0.47µF
0.12µF
RELATED PARTS
PART NUMBER DESCRIPTION
COMMENTS
LT®1511
3A Constant-Current/Constant-Voltage Battery Charger High Efficiency, Minimum External Components to Fast Charge Lithium,
NIMH and NiCd Batteries
LT1513
Sepic Constant- or Programmable- Current/Constant-
Voltage Battery Charger
Charger Input Voltage May be Higher, Equal to or Lower Than Battery Voltage,
500kHz Switching Frequency
LT1571
1.5A Switching Charger
1- or 2-Cell Li-Ion, 500kHz or 200kHz Switching Frequency, Termination Flag
LTC1628-PG
LTC1709
2-Phase, Dual Synchronous Step-Down Controller
Minimizes C and C , Power Good Output, 3.5V ≤ V ≤ 36V
IN OUT IN
2-Phase, Dual Synchronous Step-Down Controller
with VID
Up to 42A Output, Minimum C and C , Uses Smallest Components for
IN OUT
Intel and AMD Processors
LTC1729
Li-Ion Battery Charger Termination Controller
Trickle Charge Preconditioning, Temperature Charge Qualification, Time or
Charge Current Termination, Automatic Charger and Battery Detection, and
Status Output
LT1769
2A Switching Battery Charger
Constant-Current/Constant-Voltage Switching Regulator, Input Current
Limiting Maximizes Charge Current
LTC1778
LTC1960
LTC3711
LTC4007
LTC4008
Wide Operating Range, No R
Step-Down Controller
TM Synchronous
2% to 90% Duty Cycle at 200kHz, Stable with Ceramic C
SENSE
OUT
Dual Battery Charger/Selector with SPI Interface
Simultaneous Charge or Discharge of Two Batteries, DAC Programmable
Current and Voltage, Input Current Limiting Maximizes Charge Current
No R Synchronous Step-Down Controller
3.5V ≤ V ≤ 36V, 0.925V ≤ V
≤ 2V, for Transmeta, AMD and Intel
SENSE
IN
OUT
with VID
Mobile Processors
High Efficiency, Programmable Voltage,
Battery Charger with Termination
Complete Charger for 3- or 4-Cell Li-Ion Batteries, AC Adapter
Current Limit, Thermistor Sensor and Indicator Outputs
High Efficiency, Programmable Voltage/Current
Battery Charger
Constant-Current/Constant-Voltage Switching Regulator, Resistor Voltage/
Current Programming, AC Adapter Current Limit and Thermistor Sensor and
Indicator Outputs
No R
is a trademark of Linear Technology Corporation.
SENSE
sn4006 4006is
LT/TP 0503 1K • PRINTED IN USA
LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
16
LINEAR TECHNOLOGY CORPORATION 2003
●
●
(408) 432-1900 FAX: (408) 434-0507 www.linear.com
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