MAX15004D [MAXIM]
4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers;
| 型号： | MAX15004D |
| 厂家： | 
MAXIM INTEGRATED PRODUCTS |
| 描述： | 4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers |
| 文件： | 总27页 (文件大小：1654K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
EVALUATION KIT AVAILABLE
Click here for production status of specific part numbers.
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
General Description
Benefits and Features
● Wide Supply Voltage Range Meets Automotive
Power-Supply Operating Requirement Including
“Cold Crank” Conditions
The MAX15004/MAX15005 high-performance, current-
mode PWM controllers operate at an automotive input
voltage range from 4.5V to 40V (load dump). The input
voltage can go lower than 4.5V after startup if IN is boot-
strapped to a boosted output voltage. The controllers
integrate all the building blocks necessary for implement-
ing fixed-frequency isolated/nonisolated power supplies.
The general-purpose boost, flyback, forward, and SEPIC
converters can be designed with ease around the
MAX15004/MAX15005.
• 4.5V to 40V Operating Input Voltage Range
(Can Operate at Lower Voltage After Startup if
Input is Bootstrapped to a Boosted Output)
● Control Architecture Offers Excellent Performance
While Simplifying the Design
• Current-Mode Control
• 300mV, 5% Accurate Current-Limit Threshold
Voltage
• Programmable Slope Compensation
• 50% (MAX15004) or Adjustable (MAX15005)
Maximum Duty Cycle
The current-mode control architecture offers excellent line-
transient response and cycle-by-cycle current limit while
simplifying the frequency compensation. Programmable
slope compensation simplifies the design further. A fast
60ns current-limit response time, low 300mV current-limit
threshold makes the controllers suitable for high-efficiency,
high-frequency DC-DC converters. The devices include
an internal error amplifier and 1% accurate reference to
facilitate the primary-side regulated, single-ended flyback
converter or nonisolated converters.
● Accurate, Adjustable Switching Frequency and
Synchronization Avoids Interference with Sensitive
Radio Bands
• Switching Frequency Adjustable from 15kHz to
500kHz (1MHz for the MAX15005A/B/C/D)
• RC Programmable 4% Accurate Switching
Frequency
An external resistor and capacitor network programs the
switching frequency from 15kHz to 500kHz (1MHz for
the MAX15005). The MAX15004A/MAX15005 provide a
SYNC input for synchronization to an external clock. The
maximum FET-driver duty cycle for the MAX15004A/B/C/D
is 50%. The maximum duty cycle can be set on the
MAX15005A/B/C/D by selecting the right combination of
RT and CT.
• External Frequency Synchronization
● Built-In Protection Capability for Improved System
Reliability
• Cycle-by-Cycle and Hiccup Current-Limit
Protection
• Overvoltage and Thermal-Shutdown Protection
• -40°C to +125°C Automotive Temperature Range
• AEC-Q100 Qualified
The input undervoltage lockout (ON/OFF) programs the
input-supply startup voltage and can be used to shutdown
the converter to reduce the total shutdown current down
to 10µA. Protection features include cycle-by-cycle and
hiccup current limit, output overvoltage protection, and
thermal shutdown.
Ordering Information
PART
PIN-PACKAGE MAX DUTY CYCLE
MAX15004AAUE+
MAX15004AAUE/V+
MAX15004BAUE+
MAX15004BAUE/V+
MAX15004CAUE/V+
MAX15004DAUE/V+
MAX15005AAUE+
MAX15005AAUE/V+
MAX15005BAUE+
MAX15005BAUE/V+
MAX15005CAUE/V+
MAX15005DAUE/V+
16 TSSOP-EP*
16 TSSOP-EP*
16 TSSOP
50%
50%
50%
The MAX15004/MAX15005 are available in space-saving
16-pin TSSOP and thermally enhanced 16-pin TSSOP-EP
packages. All devices operate over the -40°C to +125°C
automotive temperature range.
16 TSSOP
50%
16 TSSOP-EP*
16 TSSOP
50%
50%
16 TSSOP-EP*
16 TSSOP-EP*
16 TSSOP
Programmable
Programmable
Programmable
Programmable
Programmable
Programmable
Applications
● Automotive
● Vacuum Fluorescent Display (VFD) Power Supply
● Isolated Flyback, Forward, Nonisolated SEPIC,
Boost Converters
16 TSSOP
16TSSOP-EP*
16 TSSOP
Note: All devices are specified over the -40°C to +125°C
temperature range.
+Denotes a lead(Pb)-free/RoHS-compliant package.
/V denotes an automotive qualified part.
*EP = Exposed pad.
Pin Configuration appears at end of data sheet.
** Future product—contact
factory for availability
19-0723; Rev 10; 6/20
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Absolute Maximum Ratings
IN to SGND ...........................................................-0.3V to +45V
Continuous Power Dissipation* (T = +70°C)
A
IN to PGND ...........................................................-0.3V to +45V
16-Pin TSSOP-EP (derate 21.3mW/°C
ON/OFF to SGND ......................................-0.3V to (V + 0.3V)
above +70°C) ............................................................1702mW
16-Pin TSSOP (derate 9.4mW/°C above +70°C)........754mW
Operating Junction Temperature Range .......... -40°C to +125°C
Junction Temperature......................................................+150°C
Storage Temperature Range............................ -60°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
Soldering Temperature (reflow).......................................+260°C
IN
OVI, SLOPE, RTCT, SYNC, SS, FB, COMP,
CS to SGND.....................................-0.3V to (V
+ 0.3V)
REG5
V
to PGND........................................................-0.3V to +12V
CC
REG5 to SGND.......................................................-0.3V to +6V
OUT to PGND .......................................... -0.3V to (V + 0.3V)
CC
SGND to PGND....................................................-0.3V to +0.3V
Sink Current (clamped mode) ....................................35mA
V
CC
OUT Current (< 10μs transient) .........................................±1.5A
*As per JEDEC51 Standard, Multilayer Board.
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these
or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect
device reliability.
Package Information
PACKAGE TYPE: 16 TSSOP
Package Code
U16+2
Outline Number
21-0066
90-0117
Land Pattern Number
THERMAL RESISTANCE, FOUR-LAYER BOARD
Junction to Ambient (θ
)
90°C/W
27°C/W
JA
Junction to Case (θ
)
JC
PACKAGE TYPE: 16 TSSOP-EP
Package Code
U16E+3
21-0108
90-0120
Outline Number
Land Pattern Number
THERMAL RESISTANCE, FOUR-LAYER BOARD
Junction to Ambient (θ
)
38.3°C/W
3°C/W
JA
Junction to Case (θ
)
JC
For the latest package outline information and land patterns (footprints), go to www.maximintegrated.com/packages. Note that a “+”,
“#”, or “-” in the package code indicates RoHS status only. Package drawings may show a different suffix character, but the drawing
pertains to the package regardless of RoHS status.
Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-layer board.
For detailed information on package thermal considerations, refer to www.maximintegrated.com/thermal-tutorial.
Maxim Integrated
│ 2
www.maximintegrated.com
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Electrical Characteristics
(V = 14V, C = 0.1μF, C
= 0.1μF // 1μF, C
= 1μF, V
= 5V, C = 0.01μF, C
= 100pF, RT = 13.7kΩ, CT = 560pF,
IN
IN
VCC
REG5
ON/OFF
SS
SLOPE
V
= V
= V = V
= 0V, COMP = unconnected, OUT = unconnected. T = T = -40°C to +125°C, unless otherwise noted.
SYNC
OVI
FB
CS A J
Typical values are at T = +25°C. All voltages are referenced to PGND, unless otherwise noted.) (Note 1) (Figure 5)
A
PARAMETER
POWER SUPPLY
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
Input Supply Range
Operating Supply Current
ON/OFF CONTROL
Input-Voltage Threshold
Input-Voltage Hysteresis
Input Bias Current
V
4.5
40.0
3.1
V
IN
I
Q
V
V
= 40V, f
= 150kHz
2
mA
IN
OSC
V
ON
rising
1.05
1.23
75
1.40
V
ON/OFF
V
mV
µA
µA
HYST-ON
I
V
V
= 40V
= 0V
0.5
20
B-ON/OFF
ON/OFF
ON/OFF
Shutdown Current
I
10
SHDN
INTERNAL 7.4V LDO (V
)
CC
Output (V ) Voltage Set Point
CC
V
I
= 0 to 20mA (sourcing)
= 8V to 40V
7.15
3.15
7.4
1
7.60
3.75
0.5
V
mV/V
V
VCC
VCC
Line Regulation
V
V
IN
UVLO Threshold Voltage
UVLO Hysteresis
V
rising
3.5
500
0.25
45
UVLO-VCC
CC
V
mV
V
HYST-UVLO
Dropout Voltage
V
= 4.5V, I
= 20mA (sourcing)
IN
VCC
Output Current Limit
Internal Clamp Voltage
INTERNAL 5V LDO (REG5)
Output (REG5) Voltage Set Point
Line Regulation
I
I
sourcing
mA
V
VCC-ILIM
VCC
V
I
= 30mA (sinking)
10.0
4.75
10.4
10.8
5.05
0.5
VCC-CLAMP VCC
V
V
V
V
= 7.5V, I
= 0 to 15mA (sourcing)
= 15mA (sourcing)
REG5
4.95
2
V
mV/V
V
REG5
CC
REG5
= 5.5V to 10V
= 4.5V, I
CC
Dropout Voltage
0.25
32
CC
Output Current Limit
OSCILLATOR (RTCT)
I
I
sourcing
mA
REG5-ILIM
REG5
f
f
= 2 x f
for MAX15004A/B/C/D,
for MAX15005A/B/C/D
OSC
OSC
OUT
Oscillator Frequency Range
f
15
1000
kHz
OSC
= f
OUT
RTCT Peak Trip Level
RTCT Valley Trip Level
RTCT Discharge Current
V
0.55 x V
V
V
TH,RTCT
REG5
0.1 x V
REG5
V
TL,RTCT
I
V
= 2V
1.30
-4
1.33
1.36
+4
mA
DIS,RTCT
RTCT
RT = 13.7kΩ, CT = 4.7nF,
(typ) = 18kHz
f
OSC
RT = 13.7kΩ, CT = 560pF,
(typ) = 150kHz
-4
-5
-7
+4
+5
f
Oscillator Frequency Accuracy
(Note 2)
OSC
%
RT = 21kΩ, CT = 100pF,
(typ) = 500kHz
f
OSC
RT = 7kΩ, CT = 100pF,
(typ) = 1MHz
+7
50
f
OSC
MAX15004A/B/C/D
Maximum PWM Duty Cycle
(Note 3)
MAX15005A/B/C/D
RT = 13.7kΩ, CT = 560pF,
D
%
MAX
78.5
80
81.5
170
f
(typ) = 150kHz
= 14V
IN
OSC
Minimum On-Time
t
V
110
ns
ON-MIN
Maxim Integrated
│ 3
www.maximintegrated.com
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Electrical Characteristics (continued)
(V = 14V, C = 0.1μF, C
= 0.1μF // 1μF, C
= 1μF, V
= 5V, C = 0.01μF, C
= 100pF, RT = 13.7kΩ, CT = 560pF,
IN
IN
VCC
REG5
ON/OFF
SS
SLOPE
V
= V
= V = V
= 0V, COMP = unconnected, OUT = unconnected. T = T = -40°C to +125°C, unless otherwise noted.
SYNC
OVI
FB
CS A J
Typical values are at T = +25°C. All voltages are referenced to PGND, unless otherwise noted.) (Note 1) (Figure 5)
A
PARAMETER
SYMBOL
CONDITIONS
MIN
102
2
TYP
MAX
UNITS
SYNC Lock-In Frequency Range
(Note 4)
RT = 13.7kΩ, CT = 560pF,
200
%f
OSC
f
(typ) = 150kHz
OSC
SYNC High-Level Voltage
SYNC Low-Level Voltage
V
V
V
IH-SYNC
V
0.8
IL-SYNC
SYNC Input Current
I
V
= 0 to 5V
-0.5
+0.5
µA
ns
SYNC
SYNC
SYNC Minimum Input Pulse Width
ERROR AMPLIFIER/SOFT-START
Soft-Start Charging Current
SS Reference Voltage
50
I
V
V
= 0V
8
15
1.228
1.1
21
µA
V
SS
SS
V
1.215
1.240
SS
SS Threshold for HICCUP Enable
rising
V
SS
COMP = FB,
= -500µA to +500µA
FB Regulation Voltage
V
1.215
-5
1.228
1.240
V
REF-FB
I
COMP
COMP = 0.25V to 4.5V,
FB Input Offset Voltage
V
I
V
= -500µA to +500µA,
= 0 to 1.5V
+5
mV
OS-FB
COMP
SS
FB Input Current
V
V
= 0 to 1.5V
-300
3
+300
nA
FB
COMP Sink Current
I
= 1.5V, V
= 0.25V
5.5
2.8
mA
COMP-SINK
FB
COMP
I
COMP-
SOURCE
COMP Source Current
COMP High Voltage
V
= 1V, V
= 4.5V
1.3
mA
V
FB
COMP
V
REG5
- 0.5
V
REG5
- 0.2
V
V
V
= 1V, I
= 1mA (sourcing)
= 1mA (sinking)
OH-COMP
FB
COMP
COMP Low Voltage
Open-Loop Gain
V
= 1.5V, I
0.1
100
1.6
75
0.25
V
dB
OL-COMP
FB
COMP
A
EAMP
Unity-Gain Bandwidth
Phase Margin
UGF
MHz
degrees
V/µs
V/µs
EAMP
PM
EAMP
COMP Positive Slew Rate
COMP Negative Slew Rate
PWM COMPARATOR
Current-Sense Gain
SR+
0.5
-0.5
SR-
A
ΔV
/ΔV (Note 5)
CS
2.85
3
3.15
V/V
ns
CS-PWM
PD-PWM
COMP
CS = 0.15V, from V
3V to 0.5V to OUT falling (excluding
falling edge:
COMP
PWM Propagation Delay to OUT
t
60
leading-edge blanking time)
PWM Comparator Current-Sense
Leading-Edge Blanking Time
t
50
ns
CS-BLANK
CURRENT-LIMIT COMPARATOR
Current-Limit Threshold Voltage
Current-Limit Input Bias Current
V
290
-2
305
317
+2
mV
µA
ILIM
I
OUT= high, 0 ≤ V
≤ 0.3V
B-CS
CS
From CS rising above V
(50mV
ILIM
ILIMIT Propagation Delay to OUT
t
overdrive) to OUT falling (excluding
60
ns
PD-ILIM
leading-edge blanking time)
Maxim Integrated
│ 4
www.maximintegrated.com
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Electrical Characteristics (continued)
(V = 14V, C = 0.1μF, C
= 0.1μF // 1μF, C
= 1μF, V
= 5V, C = 0.01μF, C
= 100pF, RT = 13.7kΩ, CT = 560pF,
IN
IN
VCC
REG5
ON/OFF
SS
SLOPE
V
= V
= V = V
= 0V, COMP = unconnected, OUT = unconnected. T = T = -40°C to +125°C, unless otherwise noted.
SYNC
OVI
FB
CS A J
Typical values are at T = +25°C. All voltages are referenced to PGND, unless otherwise noted.) (Note 1) (Figure 5)
A
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
ILIM Comparator Current-Sense
Leading-Edge Blanking Time
t
50
ns
CS-BLANK
Number of Consecutive ILIMIT
Events to HICCUP
7
Clock
periods
HICCUP Timeout
512
SLOPE COMPENSATION (Note 6)
Slope Capacitor Charging Current
Slope Compensation
I
V
= 100mV
9.8
-4
10.5
25
11.2
+4
µA
SLOPE
SLOPE
C
= 100pF
= 100pF
mV/µs
SLOPE
SLOPE
Slope Compensation Tolerance
(Note 2)
C
%
C
C
= 22pF
110
2.5
SLOPE
Slope Compensation Range
mV/µs
= 1000pF
SLOPE
OUTPUT DRIVER
V
= 8V (applied externally),
= 100mA (sinking)
CC
R
R
1.7
3
3.5
5
OUT-N
I
OUT
Driver Output Impedance
Ω
V
= 8V (applied externally),
= 100mA (sourcing)
CC
OUT-P
I
OUT
C
C
= 10nF, sinking
= 10nF, sourcing
1000
750
OUT
Driver Peak Output Current
I
mA
OUT-PEAK
OUT
OVERVOLTAGE COMPARATOR
Overvoltage Comparator Input
Threshold
V
V
rising
1.20
-0.5
1.228
125
1.26
+0.5
V
OV-TH
OVI
Overvoltage Comparator
Hysteresis
V
mV
OV-HYST
From OVI rising above 1.228V to OUT
falling, with 50mV overdrive
Overvoltage Comparator Delay
TD
1.6
µs
OVI
OVI Input Current
I
V
= 0 to 5V
µA
OVI
OVI
THERMAL SHUTDOWN
Shutdown Temperature
Thermal Hysteresis
T
Temperature rising
160
15
°C
°C
SHDN
T
HYST
Note 1: 100% production tested at +125°C. Limits over the temperature range are guaranteed by design.
Note 2: Guaranteed by design; not production tested.
Note 3: For the MAX15005A/B/C/D, D
Synchronization section.
depends upon the value of RT. See Figure 3b and the Oscillator Frequency/External
MAX
Note 4: The external SYNC pulse triggers the discharge of the oscillator ramp. See Figure 2. During external SYNC, D
= 50%
MAX
for the MAX15004A/B/C/D; for the MAX15005A/B/C/D, there is a shift in D
with f
/f ratio (see the Oscillator
MAX
SYNC OSC
Frequency/External Synchronization section).
Note 5: The parameter is measured at the trip point of latch, with 0 ≤ V
≤ 0.3V, and FB = COMP.
CS
-9
Note 6: Slope compensation = (2.5 x 10 )/C
mV/μs. See the Applications Information section.
SLOPE
Maxim Integrated
│ 5
www.maximintegrated.com
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Typical Operating Characteristics
V
= 14V, C = 0.1μF, C
= 0.1μF // 1μF, C
= 1μF, V
= 5V, C
= 0.01μF, C
= 100pF, RT = 13.7kΩ,
SLOPE
IN
IN
VCC
REG5
ON/OFF
SS
CT = 560pF. T = +25°C, unless otherwise noted.)
A
V
IN
UVLO HYSTERESIS
vs. TEMPERATURE
V
SUPPLY CURRENT (I
)
SHUTDOWN SUPPLY CURRENT
vs. SUPPLY VOLTAGE
IN
SUPPLY
vs. OSCILLATOR FREQUENCY (f
)
OSC
120
110
100
90
80
70
60
50
40
30
20
10
0
20
19
18
17
16
15
14
13
12
11
10
9
31
28
25
22
19
16
13
10
7
MAX15005
V
IN
= 14V
CT = 220pF
T
A
= +135°C
C
= 10nF
OUT
T
= +25°C
A
8
7
6
5
C
OUT
= 0nF
4
T
A
= -40°C
1
-40 -15
10
35
60
85 110 135
10 60 110 160 210 260 310 360 410 460 510
FREQUENCY (kHz)
5
10 15 20 25 30 35 40 45
SUPPLY VOLTAGE (V)
TEMPERATURE (°C)
V
OUTPUT VOLTAGE
IN
REG5 OUTPUT VOLTAGE
REG5 DROPOUT VOLTAGE
CC
vs. V SUPPLY VOLTAGE
vs. V VOLTAGE
vs. I
CC
REG5
7.5
7.0
6.5
6.0
5.5
5.0
5.000
4.975
4.950
4.925
4.900
4.875
4.850
4.825
4.800
4.775
4.750
4.725
4.700
0.30
0.28
0.25
0.23
0.20
0.18
0.15
0.13
0.10
0.08
0.05
0.03
0
V
= 4.5
CC
I
= 1mA (SOURCING)
REG5
T
= +125°C
A
V
IN
= V
ON/OFF
I
= 0mA
VCC
I
= 1mA
VCC
I
= 20mA
VCC
T
= +135°C
A
I
= 15mA (SOURCING)
REG5
T
A
= +25°C
T
A
= -40°C
5
10 15 20 25 30 35 40 45
5.5 6.0 6.5 7.0 7.5 8.0 8.5 9.0 9.5 10.0 10.5
VOLTAGE (V)
0
2
4
6
8
10 12 14
V
SUPPLY VOLTAGE (V)
V
I
(mA)
REG5
IN
CC
OSCILLATOR FREQUENCY (f
)
OSCILLATOR FREQUENCY (f
vs. RT/CT
)
OSC
OSC
vs. V SUPPLY VOLTAGE
IN
150
1000
100
10
CT = 100pF
CT = 220pF
CT = 560pF
RT = 13.7kΩ
CT = 560pF
MAX15005
149
148
147
146
145
144
143
142
141
140
T
A
= -40°C
T
A
= +25°C
CT = 1000pF
CT = 1500pF
CT = 2200pF
T
A
= +125°C
T
= +135°C
A
CT = 3300pF
5.5 10.5 15.5 20.5 25.5 30.5 35.5 40.5 45.5
SUPPLY VOLTAGE (V)
1
10
100
1000
V
IN
RT (kΩ)
Maxim Integrated
│ 6
www.maximintegrated.com
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Typical Operating Characteristics (continued)
V
= 14V, C = 0.1μF, C
= 0.1μF // 1μF, C
= 1μF, V
= 5V, C
= 0.01μF, C
= 100pF, RT = 13.7kΩ,
SLOPE
IN
IN
VCC
REG5
ON/OFF
SS
CT = 560pF. T = +25°C, unless otherwise noted.)
A
MAX15004 MAXIMUM DUTY CYCLE
vs. TEMPERATURE
MAX15005 MAXIMUM DUTY CYCLE
vs. TEMPERATURE
MAX15005 MAXIMUM DUTY CYCLE
vs. OUTPUT FREQUENCY (f
)
OUT
100
95
90
85
80
75
70
65
60
55
50
55
54
53
52
51
50
49
48
47
46
45
85
83
CT = 100pF
f
= 75kHz
CT = 560pF
RT = 13.7kΩ
OUT
f
= f
= 150kHz
OSC OUT
81
79
77
75
73
71
69
67
65
CT = 3300pF
CT = 2200pF
CT = 1500pF
CT = 560pF
100
CT = 1000pF
CT = 220pF
10
1000
-40 -15
10
35
60
85 110 135
-40 -15
10
35
60
85 110 135
OUTPUT FREQUENCY (kHz)
TEMPERATURE (°C)
TEMPERATURE (°C)
MAXIMUM DUTY CYCLE
ERROR AMPLIFIER OPEN-LOOP GAIN
vs. f
/f
RATIO
AND PHASE vs. FREQUENCY
CS-TO-OUT DELAY vs. TEMPERATURE
SYNC OSC
MAX15004 toc14
MAX15004 toc15
110
100
90
80
70
60
50
40
30
20
10
0
100
90
80
70
60
50
40
30
20
10
0
80
75
70
65
60
55
50
340
MAX15005
V
CS
OVERDRIVE = 50mV
CT = 560pF
RT = 10kΩ
GAIN
300
260
220
180
140
100
60
f
= f
= 180kHz
OSC OUT
V
CS
OVERDRIVE = 190mV
C
R
f
= 220pF
= 10kΩ
RTCT
RTCT
= f
PHASE
= 418kHz
OSC OUT
-10
1.0 1.1 1.2 1.3 1.4 1.5 1.6 1.7 1.8 1.9 2.0
/f RATIO
0.1
1
10 100 1k 10k 100k 1M 10M
FREQUENCY (Hz)
-40 -15
10
35
60
85 110 135
f
TEMPERATURE (°C)
SYNC OSC
OVI TO OUT DELAY THROUGH
OVERVOLTAGE COMPARATOR
DRIVER OUTPUT PEAK SOURCE
AND SINK CURRENT
POWER-UP SEQUENCE THROUGH V
IN
MAX15004 toc18
MAX15004 toc17
MAX15004 toc16
C
OUT
= 10nF
V
V
V
= 5V
IN
OUT
ON/OFF
10V/div
5V/div
V
OUT
V
OUT
V
CC
2V/div
5V/div
V
OVI
REG5
5V/div
I
V
OUT
OVI
1A/div
500mV/div
V
OUT
5V/div
2ms/div
400ns/div
1µs/div
Maxim Integrated
│ 7
www.maximintegrated.com
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Typical Operating Characteristics (continued)
V
= 14V, C = 0.1μF, C
= 0.1μF // 1μF, C
= 1μF, V
= 5V, C
= 0.01μF, C
= 100pF, RT = 13.7kΩ,
SLOPE
IN
IN
VCC
REG5
ON/OFF
SS
CT = 560pF. T = +25°C, unless otherwise noted.)
A
POWER-UP SEQUENCE
POWER-DOWN SEQUENCE
POWER-DOWN SEQUENCE THROUGH V
IN
THROUGH ON/OFF
THROUGH ON/OFF
MAX15004 toc20
MAX15004 toc21
MAX15004 toc19
V
= 5V
ON/OFF
ON/OFF
5V/div
ON/OFF
5V/div
V
IN
V
CC
10V/div
5V/div
V
CC
V
REG5
5V/div
CC
5V/div
5V/div
REG5
5V/div
REG5
5V/div
V
OUT
5V/div
V
OUT
V
OUT
5V/div
5V/div
1ms/div
400ms/div
4ms/div
LINE TRANSIENT FOR V STEP
IN
LINE TRANSIENT FOR V STEP
IN
FROM 14V TO 5.5V
FROM 14V TO 40V
MAX15004 toc22
MAX15004 toc23
V
IN
V
IN
10V/div
20V/div
V
CC
V
CC
5V/div
5V/div
REG5
5V/div
REG5
5V/div
V
OUT
V
OUT
5V/div
5V/div
100µs/div
100µs/div
HICCUP MODE FOR FLYBACK CIRCUIT
DRAIN WAVEFORM IN
(FIGURE 7)
FLYBACK CONVERTER (FIGURE 7)
MAX15004 toc24
MAX15004 toc25
I
= 10mA
LOAD
V
CS
200mV/div
10V/div
V
ANODE
1V/div
I
SHORT
500mA/div
10µs/div
4µs/div
Maxim Integrated
│ 8
www.maximintegrated.com
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Pin Description
PIN
NAME
FUNCTION
1
IN
Input Power Supply. Bypass IN with a minimum 0.1µF ceramic capacitor to PGND.
ON/OFF Input. Connect ON/OFF to IN for always-on operation. To externally program the UVLO threshold of
the IN supply, connect a resistive divider between IN, ON/OFF, and SGND. Pull ON/OFF to SGND to disable the
ON/OFF controller. To guarantee proper startup using MAX15004A/B or MAX15005A/B, ensure IN voltage is > 6V before
asserting ON/OFF signal high. Use MAX15004C/D or MAX15005C/D to ensure proper startup at lower
IN voltages.
2
Overvoltage Comparator Input. Connect a resistive divider between the output of the power supply, OVI, and
SGND to set the output overvoltage threshold.
3
4
OVI
Programmable Slope Compensation Capacitor Input. Connect a capacitor (C
) to SGND to set the amount
SLOPE
SLOPE
of slope compensation.
-9
Slope compensation = (2.5 x 10 )/C
mV/µs with C
in farads.
SLOPE
SLOPE
5
6
N.C.
No Connection. Not internally connected.
Oscillator-Timing Network Input. Connect a resistor from RTCT to REG5 and a capacitor from RTCT to SGND to
set the oscillator frequency (see the Oscillator Frequency/External Synchronization section).
RTCT
7
8
SGND
SYNC
SS
Signal Ground. Connect SGND to SGND plane.
External-Clock Synchronization Input. Connect SYNC to SGND when not using an external clock.
Soft-Start Capacitor Input. Connect a capacitor from SS to SGND to set the soft-start time interval.
Internal Error-Amplifier Inverting Input. The noninverting input is internally connected to SS.
Error-Amplifier Output. Connect the frequency compensation network between FB and COMP.
9
10
11
FB
COMP
Current-Sense Input. The current-sense signal is compared to a signal proportional to the error-amplifier output
voltage.
12
CS
13
14
15
REG5
PGND
OUT
5V Low-Dropout Regulator Output. Bypass REG5 with a 1µF ceramic capacitor to SGND.
Power Ground. Connect PGND to the power ground plane.
Gate Driver Output. Connect OUT to the gate of the external n-channel MOSFET.
7.4V Low-Dropout Regulator Output—Driver Power Source. Bypass V
with 0.1µF and 1µF or higher ceramic
CC
.
16
—
V
CC
capacitors to PGND. Do not connect external supply or bootstrap to V
CC
Exposed Pad (MAX15004A/C/MAX15005A/C only). Connect EP to the SGND plane to improve thermal
performance. Do not use the EP as an electrical connection.
EP
Maxim Integrated
│ 9
www.maximintegrated.com
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Functional Diagram
IN
1
MAꢀꢁꢂꢃꢃꢄAꢅꢆꢅꢇꢅꢈ
MAꢀꢁꢂꢃꢃꢂAꢅꢆꢅꢇꢅꢈ
16
V
CC
OFF
7.4V LDO
REG
PREREGULATOR
REFERENCE
1.228V
2
OFF
ON/OFF
ON/OFF
COMP
UVB
3.5V
UVLO
15 OUT
DRIVER
14 PGND
V
CC
THERMAL
SHUTDOWN
SET
UVB
13
REG5
5V LDO
REG
RESET
ILIMIT
COMP
OV-COMP
OVI
3
0.3V
50ns
LEAD
DELAY
1.228V
12 CS
PWM-
COMP
R
OVRLD
SLOPE
RTCT
4
6
SLOPE
COMPENSATION
2R
OSCILLATOR
11 COMP
10 FB
SS_OK
CLK
SGND 7
EAMP
7
RESET
CONSECUTIVE
EVENTS
1.228V
COUNTER
9
SS
SYNC
8
REF-AMP
OVRLD
Maxim Integrated
│ 10
www.maximintegrated.com
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
age protection. During continuous high input operation, the
power dissipation into the MAX15004/MAX15005 could
exceed its limit. Internal thermal shutdown protection safely
turns off the converter when the junction heats up to 160°C.
Detailed Description
The MAX15004/MAX15005 are high-performance,
current-mode PWM controllers for wide input-voltage
range isolated/nonisolated power supplies. These con-
trollers are for use as general-purpose boost, flyback,
and SEPIC controllers. The input voltage range of 4.5V
to 40V makes it ideal in automotive applications such as
vacuum fluorescent display (VFD) power supplies. The
Current-Mode Control Loop
The advantages of current-mode control overvoltage-
mode control are twofold. First, there is the feed-forward
characteristic brought on by the controller’s ability to adjust
for variations in the input voltage on a cycle-by-cycle
basis. Secondly, the stability requirements of the current-
mode controller are reduced to that of a single-pole
system unlike the double pole in voltage-mode control.
internal low-dropout regulator (V
regulator) enables the
CC
MAX15004/MAX15005 to operate directly from an auto-
motive battery input. The input voltage can go lower than
4.5V after startup if IN is bootstrapped to a boosted output
voltage.
The MAX15004/MAX15005 offer peak current-mode
control operation to make the power supply easy to design
with. The inherent feed-forward characteristic is useful
especially in an automotive application where the input
voltage changes fast during cold-crank and load dump con-
ditions. While the current-mode architecture offers many
advantages, there are some shortcomings. For higher duty-
cycle and continuous conduction mode operation where
the transformer does not discharge during the off duty
cycle, subharmonic oscillations appear. The MAX15004/
MAX15005 offer programmable slope compensation using
a single capacitor. Another issue is noise due to turn-on
of the primary switch that may cause the premature end
of the on cycle. The current-limit and PWM comparator
inputs have leading-edge blanking. All the shortcomings of
the current-mode control are addressed in the MAX15004/
MAX15005, making it ideal to design for automotive power
conversion applications.
The undervoltage lockout (ON/OFF) allows the devices
to program the input-supply startup voltage and ensures
predictable operation during brownout conditions.
The devices contain two internal regulators, V
and
CC
REG5. The V
regulator output voltage is set at 7.4V
CC
and REG5 regulator output voltage at 5V ±2%. The input
undervoltage lockout (UVLO) circuit monitors the V
CC
voltage and turns off the converter when the V
drops below 3.5V (typ).
voltage
CC
An external resistor and capacitor network programs
the switching frequency from 15kHz to 500kHz. The
MAX15004/MAX15005 provide a SYNC input for syn-
chronization to an external clock. The OUT (FET-driver
output) duty cycle for the MAX15004A/B/C/D is 50%. The
maximum duty cycle can be set on MAX15005A/B/C/D by
selecting the right combination of RT and CT. The RTCT
discharge current is trimmed to 2%, allowing accurate
setting of the duty cycle for the MAX15005. An internal
slope-compensation circuit stabilizes the current loop when
operating at higher duty cycles and can be programmed
externally.
Internal Regulators V
and REG5
CC
The internal LDO converts the automotive battery voltage
input to a 7.4V output voltage (V ). The V output is
CC
CC
set at 7.4V and operates in a dropout mode at input volt-
ages below 7.5V. The internal LDO is capable of delivering
20mA current, enough to provide power to internal control
The MAX15004/MAX15005 include an internal error
amplifier with 1% accurate reference to regulate the output
in nonisolated topologies using a resistive divider. The
internal reference connected to the noninverting input
of the error amplifier can be increased in a controlled
manner to obtain soft-start. A capacitor connected at SS
to ground programs soft-start to reduce inrush current and
prevent output overshoot.
circuitry and the gate drive. The regulated V
keeps the
CC
driver output voltage well below the absolute maximum
gate voltage rating of the MOSFET especially during the
double battery and load dump conditions.
The second 5V LDO regulator from V
to REG5 provides
CC
power to the internal control circuits. This LDO can also be
The MAX15004/MAX15005 include protection features like
hiccup current limit, output overvoltage, and thermal shut-
down. The hiccup current-limit circuit reduces the power
delivered to the electronics powered by the MAX15004/
MAX15005 converter during severe fault conditions. The
overvoltage circuit senses the output using the path differ-
ent from the feedback path to provide meaningful overvolt-
used to source 15mA of external load current.
Bypass V
and REG5 with a parallel combination of 1µF
CC
and 0.1µF low-ESR ceramic capacitors. Additional capaci-
tors (up to 22µF) at V can be used although they are
CC
not necessary for proper operation of the MAX15004/
MAX15005.
Maxim Integrated
│ 11
www.maximintegrated.com
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Startup Operation/UVLO/ON/OFF
The MAX15004A/B/MAX15005A/B feature two undervolt-
age lockouts (UVLO). The internal UVLO monitors the
V
IN
MAꢀꢁꢂꢃꢃꢄAꢅꢆꢅꢇꢅꢈ
MAꢀꢁꢂꢃꢃꢂAꢅꢆꢅꢇꢅꢈ
V
CC
-regulator and turns on the converter once V
rises
CC
R1
R2
above 3.5V. The internal UVLO circuit has about 0.5V
hysteresis to avoid chattering during turn-on.
ON/OFF
An external undervoltage lockout can be achieved by
controlling the voltage at the ON/OFF input. The ON/
OFF input threshold is set at 1.23V (rising) with 75mV
hysteresis.
1.23V
Before any operation can commence, the ON/OFF volt-
age must exceed the 1.23V threshold.
Calculate R1 in Figure 1 by using the following formula:
Figure 1. Setting the MAX15004A/B/MAX15005A/B
Undervoltage-Lockout Threshold
V
ON
R1 =
−1 × R2
V
UVLO
Oscillator Frequency/External Synchronization
where V
is the ON/OFF’s 1.23V rising threshold,
Use an external resistor and capacitor at RTCT to program
the MAX15004A/B/C/D/MAX15005A/B/C/D internal oscil-
lator frequency from 15kHz to 1MHz. The MAX15004A/
B/C/D output switching frequency is one-half the pro-
grammed oscillator frequency with a 50% maximum duty-
cycle limit. The MAX15005A/B/C/D output switching fre-
quency is the same as the oscillator frequency. The RC
network connected to RTCT controls both the oscillator
frequency and the maximum duty cycle. The CT capaci-
UVLO
and V
is the desired input startup voltage. Choose an
ON
R2 value in the 100kΩ range. The UVLO circuits keep
the PWM comparator, ILIM comparator, oscillator, and
output driver shut down to reduce current consumption
(see the Functional Diagram). The ON/OFF input can be
used to disable the MAX15004/MAX15005 and reduce
the standby current to less than 20μA.
Soft-Start
tor charges and discharges from (0.1 x V ) to (0.55 x
REG5
The MAX15004/MAX15005 are provided with an
externally adjustable soft-start function, saving a number of
external components. The SS is a 1.228V reference bypass
connection for the MAX15004A/B/C/D/MAX15005A/B/C/D
and also controls the soft-start period. At startup, after
V
). It charges through RT and discharges through an
REG5
internal trimmed controlled current sink. The maximum
duty cycle is inversely proportional to the discharge time
(t
). See Figure 3a and Figure 3b for a coarse
DISCHARGE
selection of capacitor values for a given switching frequen-
cy and maximum duty cycle and then use the following
equations to calculate the resistor value to fine-tune the
switching frequency and verify the worst-case maximum
duty cycle.
V
IN
is applied and the UVLO thresholds are reached, the
device enters soft-start. During soft-start, 15μA is sourced
into the capacitor (C ) connected from SS to GND
SS
causing the reference voltage to ramp up slowly. The
HICCUP mode of operation is disabled during soft-start.
When V
reaches 1.228V, the output as well as the
SS
D
MAX
OSC
t
=
HICCUP mode become fully active. Set the soft-start time
(t ) using following equation:
CHARGE
f
SS
t
CHARGE
0.7× CT
2.25(V)×RT × CT
(1.33×10−3(A)×RT) − 3.375(V)
RT =
1.23(V)× C
SS
A
t
=
SS
−6
15×10
( )
t
=
DISCHARGE
where t is in seconds and C is in farads.
SS
SS
1
f
=
The soft-start programmability is important to control
the input inrush current issue and also to avoid the
MAX15004/MAX15005 power supply from going into the
unintentional hiccup during the startup. The required soft-
start time depends on the topology used, current-limit
setting, output capacitance, and the load condition.
OSC
t
+ t
DISCHARGE
CHARGE
where f
is the oscillator frequency, RT is the
resistance connected from RTCT to REG5, and CT is
the capacitor connected from RTCT to SGND. For the
OSC
Maxim Integrated
│ 12
www.maximintegrated.com
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
most accuracy, CT should include all additional stray
capacitance (typically 25pF to 35pF).
is lost, the internal oscillator takes control of the switching
rate, returning the switching frequency to that set by RC
network connected to RTCT. This maintains output regu-
lation even with intermittent SYNC signals.
The MAX15004A/B/C/D is a 50% maximum duty-cycle
part, while the MAX15005A/B/C/D is a 100% maximum
duty-cycle part:
n-Channel MOSFET Driver
OUT drives the gate of an external n-channel MOSFET.
1
f
=
f
OUT
OSC
The driver is powered by the internal regulator (V ),
CC
2
internally set to approximately 7.4V. The regulated V
CC
voltage keeps the OUT voltage below the maximum gate
voltage rating of the external MOSFET. OUT can source
750mA and sink 1000mA peak current. The average
current sourced by OUT depends on the switching
frequency and total gate charge of the external MOSFET.
for the MAX15004A/B/C/D and:
f
= f
OUT
OSC
for the MAX15005A/B/C/D.
The MAX15004A/B/C/D/MAX15005A/B/C/D can be syn-
chronized using an external clock at the SYNC input.
For proper frequency synchronization, SYNC’s input fre-
quency must be at least 102% of the programmed internal
oscillator frequency. Connect SYNC to SGND when not
using an external clock. A rising clock edge on SYNC is
interpreted as a synchronization input. If the SYNC signal
Error Amplifier
The MAX15004A/B/C/D/MAX15005A/B/C/D include an
internal error amplifier. The noninverting input of the error
amplifier is connected to the internal 1.228V reference
and feedback is provided at the inverting input. High
100dB open-loop gain and 1.6MHz unity-gain bandwidth
allow good closed-loop bandwidth and transient response.
MAX15004A/B (D
= 50%)
MAX
WITHOUT
SYNC INPUT
WITH SYNC
INPUT
RTCT
CLKINT
SYNC
OUT
D = 50%
D = 50%
WITHOUT
SYNC INPUT
MAX15005A/B (D
= 81%)
MAX
WITH SYNC
INPUT
RTCT
CLKINT
SYNC
OUT
D = 81.25%
D = 80%
Figure 2. Timing Diagram for Internal Oscillator vs. External SYNC and D
Behavior
MAX
Maxim Integrated
│ 13
www.maximintegrated.com
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
MAX15005 MAXIMUM DUTY CYCLE
vs. OUTPUT FREQUENCY (f
OSCILLATOR FREQUENCY (f
)
OSC
)
vs. RT/CT
OUT
100
95
90
85
80
75
70
65
60
55
50
1000
100
10
CT = 100pF
CT = 220pF
CT = 560pF
CT = 100pF
CT = 1000pF
CT = 3300pF
CT = 1500pF
CT = 2200pF
CT = 2200pF
CT = 1500pF
CT = 560pF
100
CT = 1000pF
CT = 3300pF
CT = 220pF
10
1000
1
10
100
1000
OUTPUT FREQUENCY (kHz)
RT (kΩ)
Figure 3b. Oscillator Frequency vs. RT/CT
Figure 3a. MAX15005 Maximum Duty Cycle vs. Output
Frequency.
Moreover, the source and sink current capability of 2mA
provides fast error correction during the output load
transient. For Figure 5, calculate the power-supply output
voltage using the following equation:
Current Limit
The current-sense resistor (R ), connected between the
source of the MOSFET and ground, sets the current limit.
The CS input has a voltage trip level (V ) of 305mV. The
CS
CS
current-sense threshold has 5% accuracy. Set the current-
limit threshold 20% higher than the peak switch current at
the rated output power and minimum input voltage. Use
R
R
A
V
= 1+
V
REF
OUT
B
the following equation to calculate the value of R :
S
where V
= 1.228V. The amplifier’s noninverting input
REF
is internally connected to a soft-start circuit that gradu-
ally increases the reference voltage during startup. This
forces the output voltage to come up in an orderly and
well-defined manner under all load conditions.
R
= V
I
×1.2
(
)
S
CS PK
where I
is the peak current that flows through the
PRI
MOSFET at full load and minimum V .
IN
Slope Compensation
When the voltage produced by this current (through the
current-sense resistor) exceeds the current-limit com-
parator threshold, the MOSFET driver (OUT) quickly
terminates the on-cycle. In most cases, a short-time
constant RC filter is required to filter out the leading-edge
spike on the sense waveform. The amplitude and width
of the leading edge depends on the gate capacitance,
drain capacitance (including interwinding capacitance),
and switching speed (MOSFET turn-on time). Set the RC
time constant just long enough to suppress the leading
edge. For a given design, measure the leading spike at
the highest input and rated output load to determine the
value of the RC filter.
The MAX15004A/B/C/D/MAX15005A/B/C/D use an inter-
nal ramp generator for slope compensation. The internal
ramp signal resets at the beginning of each cycle and
slews at the rate programmed by the external capacitor
connected to SLOPE. The amount of slope compensation
needed depends on the downslope of the current wave-
form. Adjust the MAX15004A/B/C/D/MAX15005A/B/C/D
slew rate up to 110mV/μs using the following equation:
2.5×10−9(A)
Slope compensation(mV µs) =
C
SLOPE
where C
farads.
is the external capacitor at SLOPE in
SLOPE
Maxim Integrated
│ 14
www.maximintegrated.com
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Inductor Selection in Boost Configuration
V
IN
Using the following equation, calculate the minimum
inductor value so that the converter remains in continuous
mode operation at minimum output current (I
):
OMIN
REG5
2
V
×D× η
×I
OUT OMIN
IN
L
=
MIN
MAꢀꢁꢂꢃꢃꢄAꢅꢆꢅꢇꢅꢈ
MAꢀꢁꢂꢃꢃꢂAꢅꢆꢅꢇꢅꢈ
2× f
× V
OUT
N
R1
where:
and
V
+ V − V
D
+ V − V
D DS
OUT
IN
R
CS
D =
V
OUT
0.3V
R
S
C
CS
CURRENT-LIMIT
COMPARATOR
I
= (0.1×I ) to (0.25×I )
OMIN
O
O
The higher value of I
reduces the required
OMIN
inductance; however, it increases the peak and RMS
currents in the switching MOSFET and inductor. Use
Figure 4. Reducing Current-Sense Threshold
I
from 10% to 25% of the full load current. The V is
OMIN
D
The low 305mV current-limit threshold reduces the power
dissipation in the current-sense resistor. The current-limit
threshold can be further reduced by adding a DC offset
to the CS input from REG5 voltage. Do not reduce the
current-limit threshold below 150mV as it may cause
noise issues. See Figure 4. For a new value of the
the forward voltage drop of the external Schottky diode,
D is the duty cycle, and V is the voltage drop across
DS
the external switch. Select the inductor with low DC
resistance and with a saturation current (I ) rating
SAT
higher than the peak switch current limit of the converter.
Input Capacitor Selection in Boost
Configuration
current-limit threshold (V
), calculate the value of
ILIM_LOW
R1 using the following equation:
The input current for the boost converter is continuous
and the RMS ripple current at the input capacitor is
low. Calculate the minimum input capacitor value and
maximum ESR using the following equations:
4.75×R
R1 =
CS
0.290 − V
ILIM_LOW
Applications Information
∆I ×D
L
C
=
IN
4× f
∆V
× ∆V
Q
Boost Converter
OUT
The MAX15004A/B/C/D/MAX15005A/B/C/D can be con-
figured for step-up conversion. The boost converter
output can be fed back to IN through a Schottky diode
(see Figure 5) so the controller can function during low
voltage conditions such as cold-crank. Use a Schottky
diode (D ) in the V path to avoid backfeeding the
ESR
ESR =
∆I
L
where :
(V − V )×D
IN
DS
∆I
=
L
L × f
OUT
VIN
IN
input source. Use the equations in the following sections
V
is the total voltage drop across the external MOSFET
to calculate inductor (L
), input capacitor (C ), and
DS
MIN
IN
plus the voltage drop across the inductor ESR. ΔI is
peak-to-peak inductor ripple current as calculated above.
output capacitor (C
) when using the converter in
L
OUT
boost operation.
ΔV is the portion of input ripple due to the capacitor
Q
Maxim Integrated
│ 15
www.maximintegrated.com
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
V
IN
D
VIN
C
IN
C
1µF
VIN
L
D
VBS
C
REG5
0.1µF
V
OUT
18V
D3
13
1
REG5
IN
16
RT
CT
V
C
OUT
CC
C
4.7µF
VCC
6
RTCT
15
12
OUT
Q
MAꢀꢁꢂꢃꢃꢄAꢅꢆꢅꢇꢅꢈ
MAꢀꢁꢂꢃꢃꢂAꢅꢆꢅꢇꢅꢈ
C
FF
R
CS
CS
RA
RB
CF
RF
11
10
COMP
RS
C
CS
FB
SLOPE
SS
9
PGND
4
C
SLOPE
C
SS
Figure 5. Application Schematic
discharge and ΔV
the capacitor. Assume the input capacitor ripple contribu-
tion due to ESR (ΔV ) and capacitor discharge (ΔV )
is equal when using a combination of ceramic and alu-
minum capacitors. During the converter turn-on, a large
current is drawn from the input source especially at high
output to input differential. The MAX15004/MAX15005
are provided with a programmable soft-start; however, a
large storage capacitor at the input may be necessary to
avoid chattering due to finite hysteresis.
is the contribution due to ESR of
Output Capacitor Selection in
Boost Configuration
ESR
ESR
Q
For the boost converter, the output capacitor supplies the
load current when the main switch is on. The required out-
put capacitance is high, especially at higher duty cycles.
Also, the output capacitor ESR needs to be low enough to
minimize the voltage drop due to the ESR while support-
ing the load current. Use the following equations to calcu-
late the output capacitor, for a specified output ripple. All
ripple values are peak-to-peak.
∆V
I
ESR
O
ESR =
I
×D
O
MAX
C
=
OUT
∆V × f
Q
OUT
I is the load current, ΔV is the portion of the ripple due to
O
Q
the capacitor discharge, and ΔV
is the contribution due
is the maximum duty
ESR
to the ESR of the capacitor. D
MAX
cycle at the minimum input voltage. Use a combination
of low-ESR ceramic and high-value, low-cost aluminum
capacitors for lower output ripple and noise.
Maxim Integrated
│ 16
www.maximintegrated.com
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
MOSFET must be greater than the maximum output volt-
age setting plus a diode drop. The 10V additional margin
is recommended for spikes at the MOSFET drain due to
the inductance in the rectifier diode and output capacitor
Calculating Power Loss in Boost Converter
The MAX15004A/C/MAX15005A/C devices are available
in a thermally enhanced package and can dissipate up
to 1.7W at +70°C ambient temperature. The total power
dissipation in the package must be limited so that the
junction temperature does not exceed its absolute maxi-
mum rating of +150°C at maximum ambient temperature;
however, Maxim recommends operating the junction at
about +125°C for better reliability.
path. In addition, Q helps predict the current needed to
g
drive the gate at the selected operating frequency when
the internal LDO is driving the MOSFET.
Slope Compensation in Boost Configuration
The MAX15004A/B/MAX15005A/B use an internal
ramp generator for slope compensation to stabilize the
current loop when operating at duty cycles above 50%. It is
advisable to add some slope compensation even at lower
than 50% duty cycle to improve the noise immunity. The
slope compensations should be optimized because too
much slope compensation can turn the converter into the
voltage-mode control. The amount of slope compensa-
tion required depends on the downslope of the inductor
current when the main switch is off. The inductor downslope
depends on the input to output voltage differential of the
boost converter, inductor value, and the switching frequen-
cy. Theoretically, the compensation slope should be equal
to 50% of the inductor downslope; however, a little higher
than 50% slope is advised.
The average supply current (I
the switch driver is:
) required by
DRIVE-GATE
I
= Q × f
g OUT
DRIVE−GATE
where Q is total gate charge at 7.4V, a number available
g
from MOSFET data sheet.
ThesupplycurrentintheMAX15004A/B/C/D/MAX15005A/
B/C/D is dependent on the switching frequency. See the
Typical Operating Characteristics to find the supply current
I
of the MAX15004A/B/C/D/MAX15005A/B/C/D at
SUPPLY
a given operating frequency. The total power dissipation
(P ) in the device due to supply current (I ) and the
T
SUPPLY
current required to drive the switch (I
calculated using following equation.
) is
DRIVEGATE
Use the following equation to calculate the required
compensating slope (mc) for the boost converter:
P
= V
×(I
+ I
)
DRIVE−GATE
T
INMAX
SUPPLY
(V
− V )×R ×10−3
IN
OUT
S
MOSFET Selection in Boost Converter
The MAX15004A/B/C/D/MAX15005A/B/C/D drive a wide
variety of n-channel power MOSFETs. Since V limits
the OUT output peak gate-drive voltage to no more than
11V, a 12V (max) gate voltage-rated MOSFET can be
used without an additional clamp. Best performance,
mc =
mV µs
(
)
2L
The internal ramp signal resets at the beginning of each
cycle and slews at the rate programmed by the external
capacitor connected to SLOPE. Adjust the MAX15004A/
B/C/D/MAX15005A/B/C/D slew rate up to 110mV/μs
using the following equation:
CC
especially at low-input voltages (5V ), is achieved
IN
with low-threshold n-channel MOSFETs that specify on-
2.5×10−9
mc(mV µs)
resistance with a gate-source voltage (V ) of 2.5V or
GS
C
=
SLOPE
less. When selecting the MOSFET, key parameters can
include:
where C
farads.
is the external capacitor at SLOPE in
SLOPE
1) Total gate charge (Q ).
g
2) Reverse-transfer capacitance or charge (C
).
RSS
Flyback Converter
3) On-resistance (R
).
DS(ON)
The choice of the conversion topology is the first stage
in power-supply design. The topology selection criteria
include input voltage range, output voltage, peak currents
in the primary and secondary circuits, efficiency, form fac-
tor, and cost.
4) Maximum drain-to-source voltage (V
).
DS(MAX)
5) Maximum gate frequencies threshold voltage
(V ).
TH(MAX)
At high switching, dynamic characteristics (parameters 1
and 2 of the above list) that predict switching losses have
For an output power of less than 50W and a 1:2 input
voltage range with small form factor requirements, the
flyback topology is the best choice. It uses a minimum
of components, thereby reducing cost and form factor.
more impact on efficiency than R
, which predicts
DS(ON)
DC losses. Q includes all capacitances associated
g
with charging the gate. The V
of the selected
DS(MAX)
Maxim Integrated
│ 17
www.maximintegrated.com
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
The flyback converter can be designed to operate either
in continuous or discontinuous mode of operation. In
discontinuous mode of operation, the transformer core
completes its energy transfer during the off-cycle, while
in continuous mode of operation, the next cycle begins
before the energy transfer is complete. The discontinuous
mode of operation is chosen for the present example for
the following reasons:
to discharge the core. Use the following equations to
calculate the secondary inductance:
2
V
+ V × D
D
(
≤
)
(
)
OUT
OFFMIN
× f
OUT(MAX)
L
S
2×I
OUT
t
OFF
D
=
OFF
t
+ t
ON
OFF
● It maximizes the energy storage in the magnetic com-
where:
ponent, thereby reducing size.
D
= Minimum D
OFF
OFFMIN
● Simplifies the dynamic stability compensation design
V
= Secondary diode forward voltage drop
D
(no right-half plane zero).
I
= Maximum output rated current
OUT
● Higher unity-gain bandwidth.
Step 2) The rising current in the primary builds the energy
stored in the core during on-time, which is then released
to deliver the output power during the off-time. Primary
inductance is then calculated to store enough energy
during the on-time to support the maximum output power.
A major disadvantage of discontinuous mode opera-
tion is the higher peak-to-average current ratio in the
primary and secondary circuits. Higher peak-to-average
current means higher RMS current, and therefore, higher
loss and lower efficiency. For low-power converters, the
advantages of using discontinuous mode easily surpass
the possible disadvantages. Moreover, the drive capabil-
ity of the MAX15004/MAX15005 is good enough to drive
a large switching MOSFET. With the presently available
MOSFETs, power output of up to 50W is easily achievable
with a discontinuous mode flyback topology using the
MAX15004/MAX15005 in automotive applications.
2
2
V
×D
× f
× η
MAX
INMIN
L
=
P
2×P
OUT
OUT(MAX)
t
ON
+ t
OFF
D =
t
ON
D
= Maximum D.
MAX
Step 3) Calculate the secondary to primary turns ratio
Transformer Design
Step-by-step transformer specification design for a dis-
continuous flyback example is explained below.
(N ) and the bias winding to primary turns ratio (N
)
BP
SP
using the following equations:
Follow the steps below for the discontinuous mode trans-
former:
N
N
L
L
S
P
S
P
N
=
=
SP
Step 1) Calculate the secondary winding inductance for
guaranteed core discharge within a minimum off-
time.
and
N
11.7
+ 0.35
OUT
BIAS
N
=
=
BP
Step 2) Calculate primary winding inductance for suf-
ficient energy to support the maximum load.
N
V
P
Step 3) Calculate the secondary and bias winding turns
ratios.
The forward bias drops of the secondary diode and the
bias rectifier diode are assumed to be 0.35V and 0.7V,
respectively. Refer to the diode manufacturer’s data sheet
to verify these numbers.
Step 4) Calculate the RMS current in the primary and
estimate the secondary RMS current.
Step 4) The transformer manufacturer needs the RMS
current maximum values in the primary, secondary,
and bias windings to design the wire diameter for the
different windings. Use only wires with a diameter smaller
than 28AWG to keep skin effect losses under control. To
Step 5) Consider proper sequencing of windings and
transformer construction for low leakage.
Step 1) As discussed earlier, the core must be discharged
during the off-cycle for discontinuous mode operation.
The secondary inductance determines the time required
Maxim Integrated
│ 18
www.maximintegrated.com
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
achieve the required copper cross-section, multiple wires
must be used in parallel. Multifilar windings are common
in high-frequency converters. Maximum RMS currents
in the primary and secondary occur at 50% duty cycle
(minimum input voltage) and maximum output power.
Use the following equations to calculate the primary and
secondary RMS currents:
There are no transition losses during turn-on since the
primary current starts from zero in the discontinuous con-
duction mode. MOSFET derating may be necessary to
avoid damage during system turn-on and any other fault
conditions. Use the following equation to estimate the
power dissipation due to the power MOSFET:
2
P
= (1.4 ×R
×I
) + (Q × V × f ) +
IN OUTMAX
MOS
DSON
PRMS
g
P
D
OUT
MAX
3
V
×I × t
× f
OFF OUTMAX
I
I
=
=
×
INMAX PK
PRMS
SRMS
(
)
0.5×D
× η× V
INMIN
MAX
4
2
I
D
OUT
OFFMAX
3
C
× V × f
DS OUTMAX
DS
+
0.5×D
OFFMAX
2
The bias current for most MAX15004/MAX15005 applica-
tions is about 20mA and the selection of wire depends
more on convenience than on current capacity.
where:
Q = Total gate charge of the MOSFET (C) at 7.4V
g
V
= Input voltage (V)
IN
Step 5) The winding technique and the windings sequence
is important to reduce the leakage inductance spike at
switch turn-off. For example, interleave the secondary
between two primary halves. Keep the bias winding close
to the secondary, so that the bias voltage tracks the out-
put voltage.
t
= Turn-off time (s)
OFF
C
= Drain-to-source capacitance (F)
DS
Output Filter Design
The output capacitance requirements for the flyback con-
verter depend on the peak-to-peak ripple acceptable at
the load. The output capacitor supports the load current
during the switch on-time. During the off-cycle, the trans-
former secondary discharges the core replenishing the lost
charge and simultaneously supplies the load current. The
output ripple is the sum of the voltage drop due to charge
loss during the switch on-time and the ESR of the output
capacitor. The high switching frequency of the MAX15004/
MAX15005 reduces the capacitance requirement.
MOSFET Selection
MOSFET selection criteria include the maximum drain
voltage, peak/RMS current in the primary and the maxi-
mum-allowable power dissipation of the package without
exceeding the junction temperature limits. The voltage
seen by the MOSFET drain is the sum of the input
voltage, the reflected secondary voltage through trans-
former turns ratio and the leakage inductance spike. The
MOSFET’s absolute maximum V
than the worst-case (maximum input voltage and output
load) drain voltage.
rating must be higher
DS
An additional small LC filter may be necessary to
suppress the remaining low-energy high-frequency spikes.
The LC filter also helps attenuate the switching frequency
ripple. Care must be taken to avoid any compensation
problems due to the insertion of the additional LC filter.
Design the LC filter with a corner frequency at more than
a decade higher than the estimated closed-loop, unity-
gain bandwidth to minimize its effect on the phase margin.
Use 1μF to 10μF low-ESR ceramic capacitors and calcu-
late the inductance using following equation:
N
P
S
V
= V
+
×(V
+ V ) + V
DSMAX
INMAX
OUT
D
SPIKE
N
Lower maximum V
requirement means a shorter
, lower gate charge, and smaller
DS
channel, lower R
DS(ON)
package. A lower N /N ratio allows a low V speci-
P
S
DSMAX
fication and keeps the leakage inductance spike under
control. A resistor/diode/capacitor snubber network can
be also used to suppress the leakage inductance spike.
1
L ≤
3
2
4×10 × f
× C
C
where f = estimated converter closed-loop unity-gain
C
frequency.
The DC losses in the MOSFET can be calculated using the
value for the primary RMS maximum current. Switching
losses in the MOSFET depend on the operating frequency,
total gate charge, and the transition loss during turn-off.
Maxim Integrated
│ 19
www.maximintegrated.com
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
SEPIC Converter
Inductor Selection in SEPIC Converter
The MAX15004A/B/C/D/MAX15005A/B/C/D can be con-
figured for SEPIC conversion when the output voltage
must be lower and higher than the input voltage when
the input voltage varies through the operating range. The
duty-cycle equation:
Use the following equations to calculate the inductance
values. Assume both L1 and L2 are equal and that the
inductor ripple current (ΔI ) is equal to 20% of the input
L
current at nominal input voltage to calculate the induc-
tance value.
V
V
D
V
×D
MAX
O
IN−MIN
=
L = L = L2 =
1
1− D
2× f
× ∆I
IN
OUT
L
0.2×I
×D
MAX
) × η
indicates that the output voltage is lower than the input for
a duty cycle lower than 0.5 while V is higher than the
OUT−MAX
(1− D
∆I
=
L
OUT
MAX
input at a duty cycle higher than 0.5. The inherent advan-
tage of the SEPIC topology over the boost converter is a
complete isolation of the output from the source during a
fault at the output. The SEPIC converter output can be fed
back to IN through a Schottky diode (see Figure 6) so the
controller can function during low voltage conditions such
where f
is the converter switching frequency and
OUT
η is the targeted system efficiency. Use the coupled
inductors MSD-series from Coilcraft or PF0553-series from
Pulse Engineering, Inc. Make sure the inductor saturating
current rating (I
) is 30% higher than the peak inductor
SAT
as cold-crank. Use a Schottky diode (D ) in the V
path to avoid backfeeding the input source.
current calculated using the following equation. Use the
current-sense resistor calculated based on the I value
VIN
IN
LPK
from the equation below (see the Current Limit section).
The SEPIC converter design includes sizing of inductors,
a MOSFET, series capacitance, and the rectifier diode.
The inductance is determined by the allowable ripple
current through all the components mentioned above.
Lower ripple current means lower peak and RMS currents
and lower losses. The higher inductance value needed
for a lower ripple current means a larger-sized inductor,
which is a more expensive solution. The inductors (L1 and
L2) can be independent, however, winding them on the
same core reduces the ripple currents.
I
×D
MAX
OUT−MAX
(1− D
I
=
+ I
+ ∆I
OUT−MAX L
LPK
) × η
MAX
MOSFET, Diode, and Series Capacitor
Selection in a SEPIC Converter
For the SEPIC configuration, choose an n-channel
MOSFET with a V rating at least 20% higher than the
DS
sum of the output and input voltages. When operating at
a high switching frequency, the gate charge and switch-
ing losses become significant. Use low gate-charge
MOSFETs. The RMS current of the MOSFET is:
Calculate the maximum duty cycle using the following
equation and choose the RT and CT values accord-
ingly for a given switching frequency (see the Oscillator
Frequency/External Synchronization section).
D
2
2
MAX
3
×I ) ×
LPK LDC
I
(A) = (I
)
+ (I )
LDC
+ (I
MOS−RMS
LPK
V
+ V
D
OUT
D
=
MAX
V
+ V
+ V − (V
DS
+ V
)
CS
where I
= (I - ΔI ).
LPK L
IN−MIN
OUT
D
LDC
Use Schottky diodes for higher conversion efficiency. The
reverse voltage rating of the Schottky diode must be high-
where V is the forward voltage of the Schottky diode,
D
V
(0.305V) is the current-sense threshold of the
CS
er than the sum of the maximum input voltage (V
)
IN-MAX
MAX15004/MAX15005, and V
is the voltage drop
DS
and the output voltage. Since the average current flowing
through the diode is equal to the output current, choose
across the switching MOSFET during the on-time.
the diode with forward current rating of I
. The
OUT-MAX
Maxim Integrated
│ 20
www.maximintegrated.com
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
current sense (R ) can be calculated using the current-
CS
limit threshold (0.305V) of MAX15004/MAX15005 and
Power Dissipation
The MAX15004/MAX15005 maximum power dissipation
depends on the thermal resistance from the die to the
ambient environment and the ambient temperature. The
thermal resistance depends on the device package, PCB
copper area, other thermal mass, and airflow.
I
. Use a diode with a forward current rating more than
LPK
the maximum output current limit if the SEPIC converter
needs to be output short-circuit protected.
0.305
R
=
CS
Calculate the temperature rise of the die using following
equation:
I
LPK
Select R
20% below the value calculated above.
CS
T = T + (P x θ )
JC
J
C
T
Calculate the output current limit using the following
equation:
or:
T = T + (P x θ )
JA
J
A
T
D
where θ
is the junction-to-case thermal impedance
(3°C/W) of the 16-pin TSSOP-EP package and P is
I
=
× I
(
− ∆I
LPK L
)
JC
OUT−LIM
(1− D)
T
power dissipated in the device. Solder the exposed
pad of the package to a large copper area to spread
heat through the board surface, minimizing the case-to-
ambient thermal impedance. Measure the temperature of
where D is the duty cycle at the highest input voltage
(V ).
IN-MAX
The series capacitor should be chosen for minimum ripple
voltage (ΔV ) across the capacitor. We recommend
CP
the copper area near the device (T ) at worst-case condi-
C
using a maximum ripple ΔV
to be 5% of the minimum
CP
tion of power dissipation and use 3°C/W as θ thermal
JC
input voltage (V
) when operating at the minimum
IN-MIN
impedance. The case-to-ambient thermal impedance
input voltage. The multilayer ceramic capacitor X5R and
X7R series are recommended due to their high ripple
current capability and low ESR. Use the following
equation to calculate the series capacitor CP value.
(θ ) is dependent on how well the heat is transferred
JA
from the PCB to the ambient. Use a large copper area
to keep the PCB temperature low. The θ is 38°C/W for
JA
TSSOP-16-EP and 90°C/W for TSSOP-16 package with
the condition specified by the JEDEC51 standard for a
multilayer board.
I
×D
OUT−MAX
MAX
CP =
∆V × f
CP OUT
where ΔV
is 0.05 x V
.
CP
IN-MIN
For a further discussion of SEPIC converters, go to
http://pdfserv.maximintegrated.com/en/an/AN1051.
pdf.
Maxim Integrated
│ 21
www.maximintegrated.com
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
V
IN
2.5V TO 16V
L1
L11 = L22 = 7.5mH
V
OUT
C7
6.8µF
D2
STP745G
(8V/2A)
V
OUT
D1
LL4148
C4
22µF
C5
22µF
C6
22µF
D3
BAT54C
C1
C2
C3
1
16
6.8µF
6.8µF
6.8µF
IN
V
CC
C
VCC
C1
100nF
1µF
MAꢀꢁꢂꢃꢃꢂAꢄꢅꢄꢆꢄꢇ
2
ON
ON/OFF
OFF
RG
1Ω
15
14
STD20NF06L
OUT
PGND
REG5
3
4
OVI
SLOPE
REG5
C
SLOPE
47pF
13
12
C10
1µF
5
6
REG5
N.C.
RT
R
100Ω
CS
15kΩ
RTCT
CS
C
100pF
CS
CT
150pF
R
S
0.025Ω
7
8
SGND
SYNC
11
COMP
SYNC
V
OUT
R3
1.8kΩ
C4
680pF
R2
15kΩ
R
10kΩ
SYNC
C3
47nF
10
9
FB
SS
R1
2.7kΩ
C
SS
150nF
EP
Figure 6. SEPIC Application Circuit
Maxim Integrated
│ 22
www.maximintegrated.com
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
3) Isolate the power components and high-current path
from the sensitive analog circuitry.
Layout Recommendations
Typically, there are two sources of noise emission in a
switching power supply: high di/dt loops and high dv/dt
surfaces. For example, traces that carry the drain current
often form high di/dt loops. Similarly, the heatsink of the
MOSFET connected to the device drain presents a dv/dt
source; therefore, minimize the surface area of the heat-
sink as much as possible. Keep all PCB traces carrying
switching currents as short as possible to minimize cur-
rent loops. Use a ground plane for best results.
4) Keep the high-current paths short, especially at the
ground terminals. This practice is essential for stable,
jitter-free operation.
5) Connect SGND and PGND together close to the
device at the return terminal of V
bypass capacitor.
CC
Do not connect them together anywhere else.
6) Keep the power traces and load connections short.
This practice is essential for high efficiency. Use
thick copper PCBs (2oz vs. 1oz) to enhance full-load
efficiency.
Careful PCB layout is critical to achieve low switch-
ing losses and clean, stable operation. Refer to the
MAX15005 EV kit data sheet for a specific layout exam-
ple. Use a multilayer board whenever possible for better
noise immunity. Follow these guidelines for good PCB
layout:
7) Ensure that the feedback connection to FB is short and
direct.
8) Route high-speed switching nodes away from the
sensitive analog areas. Use an internal PCB layer for
SGND as an EMI shield to keep radiated noise away
from the device, feedback dividers, and analog bypass
capacitors.
1) Use a large copper plane under the package and
solder it to the exposed pad. To effectively use this
copper area as a heat exchanger between the PCB
and ambient, expose this copper area on the top and
bottom side of the PCB.
9) Connect SYNC pin to SGND when not used.
2) Do not connect the connection from SGND (pin
7) to the EP copper plane underneath the IC. Use
midlayer-1 as an SGND plane when using a multilayer
board.
Maxim Integrated
│ 23
www.maximintegrated.com
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Typical Operating Circuits
C12
R7
220pF
510Ω
V
IN
V
ANODE
(5.5V TO 16V)
(110V/55mA)
C13
10µF
200V
D2
C1
330µF
50V
C11
2200pF
100V
R16
10Ω
R2
560Ω
R8
100kΩ
V
GRID
C18
4700pF
100V
(60V/12mA)
D2
D1
C15
22µF
60V
1
16
IN
V
CC
C3
1µF
16V
R9
NU
R10
36kΩ
C2
0.1mF
C14
NU
MAꢀꢁꢂꢃꢃꢂAꢄꢅꢄꢆꢄꢇ
50V
R17
100kΩ
1%
FILAMENT+
(3V/650mA)
D4
2
3
ON/OFF
C16
330µF
6.3V
V
R18
47.5kΩ
1%
IN
R15
100Ω
R3
50Ω
R11
182kΩ
1%
15
14
N
OUT
PGND
REG5
FILAMENT-
C17
2.2µF
10V
OVI
D5
R12
12.1kΩ
1%
REG5
4
13
12
SLOPE
C4
100pF
C10
1µF
R1
8.45kΩ
1%
5
6
R5
1kΩ
REG5
N.C.
CS
RTCT
C9
R6
0.06Ω
1%
560pF
C5
1200pF
7
8
SGND
SYNC
11
COMP
V
R2
402kΩ
1%
ANODE
C7
47pF
R13
118kΩ
1%
1
R19
C6
4700pF
JU1
10kΩ
10
9
2
FB
SS
R14
1.3kΩ
1%
C8
EP
0.1µF
Figure 7. VFD Flyback Application Circuit
Maxim Integrated
│ 24
www.maximintegrated.com
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Typical Operating Circuits (continued)
V
IN
(4.5V TO 16V)
C1
10µF
25V
L1
10µH/IHLP5050
VISHAY
1
16
IN
V
CC
C10
C11
1µF/16V
CERAMIC
0.1µF
MAꢀꢁꢂꢃꢃꢂAꢄꢅꢄꢆꢄꢇ
V
OUT
D1
B340LB
R11
301kΩ
(18V/2A)
2
C6
56µF/25V
ON/OFF
V
OUT
R10
100kΩ
SVP-SANYO
R1
5Ω
15
14
R8
153kΩ
Q
OUT
PGND
REG5
Si736DP
3
4
OVI
R9
10kΩ
REG5
13
12
SLOPE
C2
100pF
C10
1µF
5
6
R3
1kΩ
N.C.
R2
13kΩ
REG5
CS
RTCT
C4
R4
0.025Ω
100pF
C3
180pF
7
8
SGND
SYNC
11
COMP
SYNC
V
R5
100kΩ
OUT
C8
R6
136kΩ
330pF
1
2
C9
0.1µF
JU1
10
9
FB
SS
R7
10kΩ
C7
EP
0.1µF
Figure 8. Boost Application Circuit
Maxim Integrated
│ 25
www.maximintegrated.com
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Pin Configurations
TOP VIEW
+
+
IN
ON/OFF
OVI
1
2
3
4
5
6
7
8
16 V
CC
IN
ON/OFF
OVI
1
2
3
4
5
6
7
8
16
V
CC
15 OUT
14 PGND
13 REG5
12 CS
15 OUT
14 PGND
13 REG5
12 CS
MAꢀꢁꢂꢃꢃꢄꢊ
MAꢀꢁꢂꢃꢃꢂꢊ
MAꢀꢁꢂꢃꢃꢄꢋ
MAꢀꢁꢂꢃꢃꢂꢋ
MAꢀꢁꢂꢃꢃꢄA
SLOPE
N.C.
SLOPE
N.C.
MAꢀꢁꢂꢃꢃꢂA
MAꢀꢁꢂꢃꢃꢄꢅ
MAꢀꢁꢂꢃꢃꢂꢅ
RTCT
SGND
SYNC
11 COMP
10 FB
RTCT
SGND
SYNC
11 COMP
10 FB
9
SS
9
SS
EP
ꢆꢇꢇꢈP
ꢆꢇꢇꢈPꢉEP
Chip Information
PROCESS: BiCMOS
Maxim Integrated
│ 26
www.maximintegrated.com
MAX15004A/B/C/D-
MAX15005A/B/C/D
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Revision History
REVISION REVISION
PAGES
CHANGED
DESCRIPTION
NUMBER
DATE
0
1/07
Initial release
—
Updated Features, revised equations on pages 13, 20, and 21, revised Figure 8 with
correct MOSFET, and updated package outline
1, 13, 20, 21,
25, 28
1
2
11/07
12/10
Added MAX15005BAUE/V+ automotive part, updated Features, updated Package
Information, style edits
1–5, 9, 13, 21,
25–29
Added MAX15004AAUE/V+, MAX15004BAUE/V+, MAX15005AAUE/V+ automotive parts
to the Ordering Information
3
4
1/11
1/15
1
1
Updated Benefits and Features section
1, 6, 9–11,
14–16, 18,
20–22
5
9/15
Miscellaneous updates
6
7
12/15
2/17
Deleted last sentence in the Startup Operation/UVLO/ON/OFF section
12
13
Corrected f
formula and moved section to page 12
OSC
Added part number to header on all pages, updated part number in General Description,
Benefits and Features, Ordering Information, Electrical Characteristics table and Notes,
updated Pin Description table, Functional Diagram, Detailed Description section, Soft-
Start section, Oscillator Frequency/External Synchronization section, Error Amplifier
section, Slope Compensation section,and Boost Converter section, updated Figure
1, Figure 4, Figure 5, Calculating Power Loss in Boost Converter section, MOSFET
Selection in Boost Converter section, Slope Compensation in Boost Configuration
section, and SEPIC Converter section, updated Figure 6, Figure 7, Figure 8, and Pin
Configuration figures
1, 3, 5, 9–17,
20, 22, 24–26
8
1/20
9
2/20
6/20
GFT?Moved Package Information to page 2 and added thermal characteristics
2, 26
1
10
Removed all future-product notation from Ordering Information
For pricing, delivery, and ordering information, please visit Maxim Integrated’s online storefront at https://www.maximintegrated.com/en/storefront/storefront.html.
Maxim Integrated cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim Integrated product. No circuit patent licenses
are implied. Maxim Integrated reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and max limits)
shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.
©
Maxim Integrated and the Maxim Integrated logo are trademarks of Maxim Integrated Products, Inc.
2020 Maxim Integrated Products, Inc.
│ 27
相关型号：
SI9130DB
5- and 3.3-V Step-Down Synchronous ConvertersWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9135LG-T1
SMBus Multi-Output Power-Supply ControllerWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9135LG-T1-E3
SMBus Multi-Output Power-Supply ControllerWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9135_11
SMBus Multi-Output Power-Supply ControllerWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9136_11
Multi-Output Power-Supply ControllerWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9130CG-T1-E3
Pin-Programmable Dual Controller - Portable PCsWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9130LG-T1-E3
Pin-Programmable Dual Controller - Portable PCsWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9130_11
Pin-Programmable Dual Controller - Portable PCsWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9137
Multi-Output, Sequence Selectable Power-Supply Controller for Mobile ApplicationsWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9137DB
Multi-Output, Sequence Selectable Power-Supply Controller for Mobile ApplicationsWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9137LG
Multi-Output, Sequence Selectable Power-Supply Controller for Mobile ApplicationsWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9122E
500-kHz Half-Bridge DC/DC Controller with Integrated Secondary Synchronous Rectification DriversWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
©2020 ICPDF网 联系我们和版权申明