MAX16929_技术文档

�� ꢀMaxim Integrated Productsꢀ ꢀ 1

19-5857; Rev 2; 1/12

E V A L U A T I O N K I T A V A I L A B L E

MAX16929

Automotive TFT-LCD Power Supply with Boost

Converter and Gate Voltage Regulators

General Description

Features

The MAX16929 is a highly integrated power supply for

automotive TFT-LCD applications. The device integrates

one buck converter, one boost converter, one 1.8V/3.3V

regulator controller, and two gate voltage regulators.

The device comes in several versions to satisfy com-

mon automotive power-supply requirements (see the

Ordering Information/Selector Guide table).

SꢀOperatingꢀVoltageꢀRangeꢀofꢀ4Vꢀtoꢀ28Vꢀ(Buck)ꢀorꢀ

3Vꢀtoꢀ5.5Vꢀ(Boost)

SꢀIndependentꢀ28VꢀInputꢀBuckꢀConverterꢀPowersꢀ

TFTꢀBiasꢀSupplyꢀCircuitryꢀandꢀExternalꢀCircuitry

SꢀHigh-Powerꢀ(Upꢀtoꢀ6W)ꢀBoostꢀOutputꢀProvidingꢀUpꢀ

toꢀ18V

Sꢀ1.8Vꢀorꢀ3.3VꢀRegulatorꢀProvidesꢀ500mAꢀwithꢀ

Designed to operate from a single 4V to 28V supply or

5.5V to 28V supply, the device is ideal for automotive

TFT-LCD applications.

ExternalꢀnpnꢀTransistor

SꢀOneꢀPositive-GateꢀVoltageꢀRegulatorꢀCapableꢀofꢀ

Deliveringꢀ20mAꢀatꢀ28V

Both the buck and boost converters use spread-spec-

trum modulation to reduce peak interference and to opti-

mize EMI performance.

SꢀOneꢀNegative-GateꢀVoltageꢀRegulator

SꢀHigh-FrequencyꢀOperation

ꢀ2.1MHzꢀ(BuckꢀConverter)

ꢀ2.2MHzꢀ(BoostꢀConverter)

The sequencing input (SEQ) allows flexible sequencing

of the positive-gate and negative-gate voltage regulators.

The power-good indicator (PGOOD) indicates a failure

on any of the converters or regulator outputs. Integrated

thermal shutdown circuitry protects the device from over-

heating.

SꢀFlexibleꢀStand-AloneꢀSequencing

SꢀTrueꢀShutdown™ꢀBoostꢀConverter

Sꢀ6µAꢀLow-CurrentꢀShutdownꢀModeꢀ(Buck)

SꢀInternalꢀSoft-Start

The MAX16929 is available in a 28-pin TSSOP pack-

age with exposed pad, and operates over the -40NC to

+105NC temperature range.

SꢀOvertemperatureꢀShutdown

Sꢀ-40NCꢀtoꢀ+105NCꢀOperation

Applications

Ordering Information/Selector Guide appears at end of data

sheet.

Automotive Dashboards

Automotive Central Information Displays

Automotive Navigation Systems

Typical Application Circuit appears at end of data sheet.

True Shutdown is a trademark of Maxim Integrated Products, Inc.

For related parts and recommended products to use with this part, refer to: www.maxim-ic.com/MAX16929.related

Forꢀpricing,ꢀdelivery,ꢀandꢀorderingꢀinformation,ꢀpleaseꢀcontactꢀMaximꢀDirectꢀatꢀ1-888-629-4642,ꢀ

orꢀvisitꢀMaxim’sꢀwebsiteꢀatꢀwww.maxim-ic.com.

�� ꢀMaxim Integrated Productsꢀ ꢀ 2

MAX16929

Automotive TFT-LCD Power Supply with Boost

Converter and Gate Voltage Regulators

ABSOLUTEꢀMAXIMUMꢀRATINGS

INB, ENB to GND..................................................-0.3V to +42V

BST to GND...........................................................-0.3V to +47V

BST to LXB ..............................................................-0.3V to +6V

LXB to GND..............................................................-6V to +42V

AVL, PGOOD to GND .............................................-0.3V to +6V

FBB to GND...........................................................-0.5V to +12V

CP, GH to GND.....................................................-0.3V to +31V

ENP, DR, FB, GATE, COMPI, FBGH,

FBGL, REF, SEQ to GND .....................-0.3V to (V

GND to PGNDP....................................................-0.3V to +0.3V

+ 0.3V)

INA

Continuous Power Dissipation (T = +70NC)

A

TSSOP (derate 27mW/NC above +70NC)...................2162mW

Operating Temperature Range........................ -40NC to +105NC

Junction Temperature Range........................... -40NC to +150NC

Storage Temperature Range............................ -65NC to +150NC

Lead Temperature (soldering, 10s) ................................+300NC

Soldering Temperature (reflow) ......................................+260NC

CP, GH to GND (V

= 3.3V) ..............................-0.3V to +29V

INA

LXP to GND...........................................................-0.3V to +20V

DRVN to GND........................................................-25V to +0.3V

INA, COMPV, FBP to GND......................................-0.3V to +6V

Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional opera-

tion of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute

maximum rating conditions for extended periods may affect device reliability.

PACKAGEꢀTHERMALꢀCHARACTERISTICS (Noteꢀ1)

TSSOP

Junction-to-Ambient Thermal Resistance (B ) ..........37NC/W

JA

Junction-to-Case Thermal Resistance (B ).................2NC/W

JC

Noteꢀ1: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-

layer board. For detailed information on package thermal considerations, refer to www.maxim-ic.com/thermal-tutorial.

ELECTRICALꢀCHARACTERISTICS

(V

= 12V, V

= 5V, V

= V

= 0V, T = T = -40NC to +105NC, typical values are at T = +25NC, unless otherwise noted.)

INB

INA

GND

PGNDP A J A

(Note 2)

PARAMETER

SYMBOL

CONDITIONS

MIN

TYP

MAX

UNITS

BUCKꢀCONVERTER

V

= 5V (Note 3)

5.5

4

28

42

9

OUTB

Supply Voltage Range

V

= 3.3V (Note 3)

V

INB

OUTB

t < 500ms

V

= 0V

6

ENB

Supply Current

I

FA

INB

= V , no load, T = +25NC

70

ENB

INB

A

Undervoltage Lockout

V

AVL rising

3.1

0.5

2.1

3.5

2.3

V

INB,UVLO

Undervoltage Lockout Hysteresis

PWM Switching Frequency

Spread-Spectrum Range

f

1.9

MHz

%

SWB

SSR

+6

5

5V, continuous mode

5V, skip mode

P 18V,

-3%

+3%

+6%

+3%

+6%

400

5

6V P V

INB

Output-Voltage Accuracy

V

OUTB

I

< full load

3.3V, continuous mode

3.3V, skip mode

3.3

180

LOAD

High-Side DMOS R

R

I = 1A

mI

DS_ON

DS_ON(LXB) LXB

Skip-Current Threshold

I

16

2

%I

SKIP

MAX

I

= 1.2A option

= 2.0A option

1.6

2.7

2.4

OUTB

Current-Limit Threshold

I

A

MAX

3.4

4.08

�� ꢀMaxim Integrated Productsꢀ ꢀ 3

MAX16929

Automotive TFT-LCD Power Supply with Boost

Converter and Gate Voltage Regulators

ELECTRICALꢀCHARACTERISTICSꢀ(continued)

(V

= 12V, V

= 5V, V

= V

= 0V, T = T = -40NC to +105NC, typical values are at T = +25NC, unless otherwise noted.)

INB

INA

GND

PGNDP A J A

(Note 2)

PARAMETER

SYMBOL

CONDITIONS

MIN

TYP

3.9

80

MAX

UNITS

ms

%

Soft-Start Ramp Time

Maximum Duty Cycle

Minimum Duty Cycle

Continuous mode

Dropout

20

%

Maximum Duty Cycle in Dropout

Thermal Shutdown Temperature

Thermal Shutdown Hysteresis

POWERꢀGOODꢀ(PGOOD)

95

%

+175

15

NC

Rising

Falling

94

92

13

PGOOD Threshold

%

90

95

PGOOD Debounce Time

Output High-Leakage Current

Output Low Level

Fs

FA

V

0.2

0.4

LOGICꢀLEVELS

ENB Threshold

ENB rising

1.4

3

1.8

0.2

5

2.2

9

V

ENB Hysteresis

ENB Input Current

FA

BOOST,ꢀPOSITIVEꢀ(GH),ꢀNEGATIVEꢀ(GL),ꢀ1.8V/3.3VꢀCONVERTERS

INA Input Supply Range

V

3

5.5

2.0

V

INA

V

= V

= 1.3V, V

= 0V,

FBP

FBGH

FBGL

INA Supply Current

I

1.5

2.7

mA

INA

LXP not switching

V rising, hysteresis = 200mV,

INA

INA Undervoltage Lockout

Threshold

V

2.5

2.9

V

INA,UVLO

T

= +25NC

A

INA Shutdown Current

I

V

= 0V, T = +25NC

0.5

+165

15

FA

NC

SHDN

ENP

A

Thermal Shutdown Temperature

Thermal Shutdown Hysteresis

T

Temperature rising

SHDN

T

H

V

old

, V

, or V

below its thresh-

FBP FBGH

FBGL

Duration to Trigger Fault Condition

238

1.9

ms

s

Autoretry Time

REFERENCEꢀ(REF)

REF Output Voltage

REF Load Regulation

V

No output current

0 < I < 80FA, REF sourcing

1.236

-2

1.25

1.264

+2

V

REF

%

REF

REF Undervoltage Lockout

Threshold

Rising edge, hysteresis = 200mV

1.165

V

OSCILLATOR

Internal Oscillator Frequency

f

T

= +25NC

A

3.96

4.40

4.84

MHz

OSC

Spread-Spectrum Modulation

Frequency

f

/2

OSC

SS

�� ꢀMaxim Integrated Productsꢀ ꢀ 4

MAX16929

Automotive TFT-LCD Power Supply with Boost

Converter and Gate Voltage Regulators

ELECTRICALꢀCHARACTERISTICSꢀ(continued)

(V

= 12V, V

= 5V, V

= V

= 0V, T = T = -40NC to +105NC, typical values are at T = +25NC, unless otherwise noted.)

INB

INA

GND

PGNDP A J A

(Note 2)

PARAMETER

SYMBOL

SSR

CONDITIONS

As a percentage of switching frequency,

MIN

TYP

MAX

UNITS

Spread-Spectrum Factor

Q4

%

f

SW

BOOSTꢀCONVERTER

Switching Frequency

Maximum Duty Cycle

f

1.98

82

2.20

2.42

93.5

MHz

%

SW

Low boost current-

limit option

0.625

1.25

0.78

1.56

Duty cycle = 70%,

= 220pF

LXP Current Limit

I

A

LIM

C

COMPI

High boost current-

limit option

1.87

LXP On-Resistance

LXP Leakage Current

Soft-Start Time

R

I

= 200mA

110

8.5

30

250

20

mI

FA

ms

V

DS_ON(LXP) LXP

I

V

= 20V, T =+25NC

LK_LXP

LXP A

(Note 4)

Output Voltage Range

V

18

SH

INA

T

= +3V to +5.5V,

= +25NC

0.985

0.98

0.74

1.0

0.85

-1

1.015

1.02

0.96

V

0 < I

A

INA

FBP Regulation Voltage

V

FBP

< full load

T =-40NCto+105NC

LOAD

A

PGOOD Threshold

V

Measured at FBP

0 < I < full load

V

%

PG_FBP

FBP Load Regulation

FBP Line Regulation

FBP Input Bias Current

LOAD

V

= +3V to +5.5V

0.1

%/V

FA

FS

INA

= +1V, T = +25NC

Q1

FBP

A

FBP to COMPV Transconductance

POSITIVE-GATEꢀVOLTAGEꢀREGULATORꢀ(GH)

DI = Q2.5FA at COMPV, T = +25NC

400

A

With external charge pump, T = +25NC

A

Output Voltage Range

V

5

29

V

GH

(maximum V = 29.5V)

CP

CP Overvoltage Threshold

FBGH Regulation Voltage

PGOOD Threshold

T

= +25NC (Note 6)

29.5

0.98

0.83

30.5

1.0

0.85

2

V

A

V

I

= 1mA

1.034

0.87

FBGH

GH

V

Measured at FBGH

V

PG_FBGH

FBGH Load Regulation

I

= 0 to 20mA

%

GH

V

= 12V to 20V at V

= 10mA

= 10V,

GH

CP

FBGH Line Regulation

2

%

I

GH

FBGH Input Bias Current

GH Output Current

GH Current Limit

V

= 1V, T = +25NC

Q1

FA

mA

ms

FBGH

A

I

- V = 2V

GH

20

35

GH

CP

I

56

LIM_GH

GH Soft-Start Time

7.45

NEGATIVE-GATEꢀVOLTAGEꢀREGULATORꢀ(GL)

Output Voltage Range

V

-24

-2

V

DRVN

FBGL Regulation Voltage

V

I

= 100FA

0.212

0.242

0.271

FBGL

DRVN

�� ꢀMaxim Integrated Productsꢀ ꢀ 5

MAX16929

Automotive TFT-LCD Power Supply with Boost

Converter and Gate Voltage Regulators

ELECTRICALꢀCHARACTERISTICSꢀ(continued)

(V

= 12V, V

= 5V, V

= V

= 0V, T = T = -40NC to +105NC, typical values are at T = +25NC, unless otherwise noted.)

INB

INA

GND

PGNDP A J A

(Note 2)

PARAMETER

SYMBOL

CONDITIONS

Measured at FBGL

MIN

TYP

MAX

0.42

Q1

UNITS

V

PGOOD Threshold

V

0.38

0.4

PG_FBGL

FBGL Input Bias Current

DRVN Source Current

V

= +0.25V

FA

FBGL

V

= +0.5V, V

= -10V

2

mA

ms

DRVN

DRVN Source Current Limit

GL Soft-Start Time

2.5

4

7.45

1.8V/3.3VꢀREGULATORꢀCONTROLLER

3.3V regulator option

1.8V regulator option

3.18

3.3

1.8

3.38

Output Voltage

V

= V

FB

V

FB

DR

1.746

1.854

3.3V regulator option,

FB rising

2.4

2.57

2.7

Measured at FB

(Notes 5, 7)

FB PGOOD Threshold

V

PG_FB

1.8V regulator option

1.364

1.38

2.5

4.5

6

1.396

V

= 1.8V

= 3.3V

= 1.8V

FB

FB Input Bias Current

FA

DR Drive Current

4.5

33

mA

INPUTꢀSERIESꢀSWITCHꢀCONTROL

p-Channel FET GATE Sink Current

V

= 0.5V

55

75

FA

GATE

Measured at GATE; below this voltage, the

external p-channel FET is considered on

GATE Voltage Threshold

1.25

V

DIGITALꢀLOGIC

ENP, SEQ Input Pulldown Resistor

Value

R

V

500

kI

PD

ENP, SEQ Input-Voltage Low

ENP, SEQ Input-Voltage High

PGOOD Leakage Current

V

0.3 x V

V

IL

INA

0.7 x V

IH

INA

I

T

= +25NC

A

Q1

FA

V

LK_IN

PGOOD Output-Voltage Low

V

2mA sink current, T = +25NC

0.4

OL

A

Noteꢀ2: Specifications over temperature are guaranteed by design and not production tested.

Noteꢀ3: Operation in light-load conditions or at extreme duty cycles result in skipped cycles, resulting in lower operating frequency

and possibly limited output accuracy and load response.

Noteꢀ4: 50% of the soft-start voltage time is due to the soft-start ramp, and the other 50% is due to the settling of the output voltage.

Noteꢀ5: Guaranteed by design; not production tested.

Noteꢀ6: After the voltage at CP exceeds this overvoltage threshold, the entire circuit switches off and autoretry is started.

Noteꢀ7: FB power good is indicated by PGOOD. The condition V < V

does not shutdown/restart the device.

FB

PG_FB

�� ꢀMaxim Integrated Productsꢀ ꢀ 6

MAX16929

Automotive TFT-LCD Power Supply with Boost

Converter and Gate Voltage Regulators

Typical Operating Characteristics

(V

= 5V, V

= 12V, measurements taken on “A” version, unless otherwise noted; V = 12V, V

= 18V, V = -6V, V

= 3.3V,

REG

INA

INB

A

SH

GH

GL

V

= 5V, T = +25NC, unless otherwise noted.)

OUTB

SHUTDOWN SUPPLY CURRENT (BUCK)

EFFICIENCY vs. LOAD CURRENT (BUCK)

20

100

90

80

70

60

50

40

30

20

10

0

18

16

14

12

10

8

V

=12V

INB

V

=18V

INB

V

= 28V

INB

6

4

2

0

4

8

12

16

20

24

28

0

0.4

0.8

1.2

1.6

2.0

SUPPLY VOLTAGE (V)

LOAD CURRENT (A)

LINE REGULATION (BUCK)

LOAD REGULATION (BUCK)

4.0

3.2

6

5

4

I

= 0A

OUTB

2.4

3

1.6

V

= 12V

INB

2

0.8

I

= 1A

OUTB

1

0

-1

-2

-3

-4

-5

-6

V

= 18V

INB

-0.8

-1.6

-2.4

-3.2

-4.0

V

= 28V

INB

I

= 2A

OUTB

4

6

8

10 12 14 16 18 20 22 24 26 28

INPUT VOLTAGE (V)

0

0.4

0.8

1.2

1.6

2.0

LOAD CURRENT (A)

STARTUP WAVEFORMS (BUCK)

LOAD-TRANSIENT RESPONSE (BUCK)

MAX16929 toc06

MAX16929 toc05

V

ENB

1.8A

0.2A

5V/div

I

OUTB

1A/div

I

LXB

2A/div

V

OUTB

V

OUTB

2V/div

(AC-COUPLED)

100mV/div

1ms/div

400µs/div

�� ꢀMaxim Integrated Productsꢀ ꢀ 7

MAX16929

Automotive TFT-LCD Power Supply with Boost

Converter and Gate Voltage Regulators

Typical Operating Characteristics (continued)

(V

= 5V, V

= 12V, measurements taken on “A” version, unless otherwise noted; V = 12V, V

= 18V, V = -6V, V = 3.3V,

REG

INA

INB

A

SH

GH

GL

V

= 5V, T = +25NC, unless otherwise noted.)

OUTB

LINE-TRANSIENT RESPONSE (BUCK)

SHORT-CIRCUIT BEHAVIOR (BUCK)

MAX16929 toc08

MAX16929 toc07

V

OUTB

28V

12V

5V/div

V

INB

10V/div

I

LXB

2A/div

V

OUTB

V

PGOOD

(AC-COUPLED)

50mV/div

5V/div

1ms/div

100ms/div

LOAD DUMP RESPONSE

INA SHUTDOWN SUPPLY CURRENT

MAX16929 toc09

10

9

8

7

6

5

4

3

2

1

0

42V

12V

V

INB

20V/div

I

LXB

2A/div

V

OUTB

5V/div

100ms/div

3.0

3.5

4.0

4.5

5.0

5.5

INPUT VOLTAGE (V)

EFFICIENCY vs. LOAD CURRENT

(BOOST)

LOAD REGULATION (BOOST)

LINE REGULATION (BOOST)

100

90

80

70

60

50

40

30

20

10

0

1.0

0.8

I

= 0mA

LOAD

0.8

0.6

V

= 5V

INA

0.4

0.2

0.4

V

= 3.3V

INA

0.2

V

= 3.3V

INA

0

-0.2

-0.4

-0.6

-0.8

-1.0

-0.2

-0.4

-0.6

-0.8

-1.0

V

= 5V

INA

0

100

200

300

400

500

0

100

200

300

400

500

3.0

3.5

4.0

4.5

5.0

5.5

LOAD CURRENT (mA)

INPUT VOLTAGE (V)

�� ꢀMaxim Integrated Productsꢀ ꢀ 8

MAX16929

Automotive TFT-LCD Power Supply with Boost

Converter and Gate Voltage Regulators

Typical Operating Characteristics (continued)

(V

= 5V, V

= 12V, measurements taken on “A” version, unless otherwise noted; V = 12V, V

= 18V, V = -6V, V

= 3.3V,

REG

INA

INB

A

SH

GH

GL

V

= 5V, T = +25NC, unless otherwise noted.)

OUTB

STARTUP WAVEFORMS (BOOST)

LOAD-TRANSIENT RESPONSE (BOOST)

MAX16929 toc14

MAX16929 toc15

V

ENP

5V/div

450mA

50mA

V

LXP

I

SH

10V/div

500mA/div

I

LXP

1A/div

V

SH

100mV/div

V

SH

10V/div

4ms/div

100µs/div

SUPPLY SEQUENCING WAVEFORMS

(V

SEQ

= 0V)

(V

SEQ

= V

)

INA

MAX16929 toc16

MAX16929 toc17

V

ENP

5V/div

V

GH

5V/div

V

SH

5V/div

V

REG

V

REG

5V/div

V

GL

5V/div

10ms/div

LOAD REGULATION (GH REGULATOR)

LINE REGULATION (GH REGULATOR)

0

-0.4

-0.8

-1.2

-1.6

-2.0

-2.4

-2.6

-3.2

-3.6

-4.0

1.0

0.8

0.6

0.4

I

= 10mA

LOAD

0.2

0

-0.2

-0.4

-0.6

-0.8

-1.0

I

= 20mA

LOAD

0

2

4

6

8

10 12 14 16 18 20

18 19 20 21 22 23 24 25 26 27 28 29 30

VOLTAGE (V)

LOAD CURRENT (mA)

V

CP

�� ꢀMaxim Integrated Productsꢀ ꢀ 9

MAX16929

Automotive TFT-LCD Power Supply with Boost

Converter and Gate Voltage Regulators

Typical Operating Characteristics (continued)

(V

= 5V, V

= 12V, measurements taken on “A” version, unless otherwise noted; V = 12V, V

= 18V, V = -6V, V

= 3.3V,

REG

INA

INB

A

SH

GH

GL

V

= 5V, T = +25NC, unless otherwise noted.)

OUTB

LINE REGULATION

(GL REGULATOR)

LOAD REGULATION

(GL REGULATOR)

1.0

0.8

3.0

2.7

2.4

2.1

1.8

1.5

1.2

0.9

0.6

0.3

0

0.6

0.4

0.2

I

= 20mA

= 10mA

LOAD

0

-0.2

-0.4

-0.6

-0.8

-1.0

I

LOAD

-24 -22 -20 -18 -16 -14 -12 -10 -8 -6

VOLTAGE (V)

0

2

4

6

8

10 12 14 16 18 20

V

LOAD CURRENT (mA)

CN

LOAD REGULATION

(3.3V LINEAR REGULATOR)

LOAD-TRANSIENT RESPONSE

(3.3V LINEAR REGULATOR)

MAX16929 toc23

0

-0.02

-0.04

-0.06

-0.08

-0.10

-0.12

-0.14

-0.16

-0.18

-0.20

450mA

50mA

I

OUTB

500mA/div

V

REG

(AC-COUPLED)

100mV/div

0

50 100 150 200 250 300 350 400 450 500

LOAD CURRENT (mA)

100µs/div

�� ꢀMaxim Integrated Productsꢀ ꢀ 10

MAX16929

Automotive TFT-LCD Power Supply with Boost

Converter and Gate Voltage Regulators

Pin Configuration

TOP VIEW

1

2

28

27

26

25

24

23

22

21

20

19

18

17

16

15

ENP

DR

SEQ

REF

3

FB

FBGL

FBGH

COMPI

GND

DRVN

GH

4

GATE

PGNDP

LXP

5

MAX16929

6

7

INA

8

COMPV

FBP

9

CP

10

11

12

13

14

FBB

PGOOD

GND

ENB

AVL

BST

LXB

INB

EP

LXB

INB

TSSOP

Pin Description

NAME

FUNCTION

Boost Circuitry and 1.8V/3.3V Regulator Controller Enable Input. ENP has an internal 500kI pulldown

resistor. Drive high for normal operation and drive low to place the device (except buck converter) in

shutdown.

1

ENP

1.8V or 3.3V Regulator Output. DR has a 4.5mA (min) drive capability. For greater output current capa-

bility, use an external npn bipolar transistor whose base is connected to DR.

2

3

DR

FB

1.8V or 3.3V Regulator Feedback Input. FB is regulated to 1.8V or 3.3V. Connect FB to DR when power-

ing loads demanding less than 4.5mA. For greater output current capability, use an external npn bipo-

lar transistor whose emitter is connected to FB.

External p-Channel FET Gate Drive. GATE is an open-drain driver connected to the gate of the external

input series p-channel FET. Connect a pullup resistor between GATE and INA. During a fault condition,

the gate driver turns off and the pullup resistor turns off the FET.

4

GATE

5

6

7

PGNDP

LXP

Boost Converter Power Ground

Boost Converter Switching Node. Connect LXP to the inductor and catch diode of the boost converter.

Boost Circuitry and 1.8V/3.3V Regulator Controller Power Input. Connect INA to a 3V to 5.5V supply.

INA

Boost Error Amplifier Compensation Connection. Connect a compensation network between COMPV to

GND.

8

9

COMPV

FBP

Boost Converter Feedback Input. FBP is regulated to 1V. Connect FBP to the center of a resistive divid-

er connected between the boost output and GND.

�� ꢀMaxim Integrated Productsꢀ ꢀ 11

MAX16929

Automotive TFT-LCD Power Supply with Boost

Converter and Gate Voltage Regulators

Pin Description (continued)

NAME

FUNCTION

Buck Converter Feedback Input. FBB is regulated to either 3.3V or 5V. Connect FBB to the output-volt-

age node, OUTB, as shown in the Typical Application Circuit.

10

FBB

Buck Converter Internal 5V Regulator. Connect a 1FF capacitor between AVL and the analog ground

plane. Do not use AVL to power external circuitry.

11

12

AVL

BST

LXB

Buck Converter Bootstrap Capacitor Connection. Connect a 0.1FF capacitor between BST and LXB.

Buck Converter Inductor Connection. Connect the inductor, boost capacitor, and catch diode at this

node.

13, 14

Buck Converter Power Input. Connect to a 4V to 28V supply. Connect a 1FF or larger ceramic capaci-

tor in parallel with a 47FF bulk capacitor from INB to the power ground plane. Connect both INB power

inputs together.

15, 16

INB

Buck Converter Enable Input. ENB is a high-voltage compatible input. Connect to INB for normal opera-

tion and connect to ground to disable the buck converter.

17

ENB

18, 23

19

GND

Analog Ground

PGOOD Open-Drain Power-Good Output. Connect PGOOD to INA through an external pullup resistor.

Positive-Gate Voltage Regulator Power Input. Connect CP to the positive output of the external charge

20

CP

pump. Ensure that V does not exceed the CP overvoltage threshold as given in the Electrical

CP

Characteristics table.

21

22

GH

Positive-Gate Voltage Regulator Output

Negative-Gate Voltage Regulator Driver Output. DRVN is the open drain of an internal p-channel FET.

Connect DRVN to the base of an external npn pass transistor.

DRVN

Boost Slope Compensation Connection. Connect a capacitor between COMPI and GND to set the

slope compensation.

24

25

COMPI

FBGH

Positive-Gate Voltage Regulator Feedback Input. FBGH is regulated to 1V. Connect FBGH to the center

of a resistive divider connected between GH and GND.

Negative-Gate Voltage Regulator Feedback Input. FBGL is regulated to 0.25V. Connect FBGL to the

center of a resistive divider connected between REF and the output of the negative-gate voltage

regulator.

26

FBGL

27

28

REF

SEQ

1.25V Reference Output. Bypass REF to GND with a 0.1FF ceramic capacitor.

Sequencing Input. SEQ has an internal 500kI pulldown resistor. SEQ determines the sequence in

which V

and V power-up. See Table 1 for supply sequencing options.

GH

GL

Exposed Pad. Connect to a large contiguous copper ground plane for optimal heat dissipation. Do not

use EP as the only electrical ground connection.

—

EP

�� ꢀMaxim Integrated Productsꢀ ꢀ 12

MAX16929

Automotive TFT-LCD Power Supply with Boost

Converter and Gate Voltage Regulators

enabling the buck converter, V

begins to rise. Once

AVL

Detailed Description

V

exceeds the undervoltage lockout voltage of 3.5V

AVL

(max), LXB starts switching. Bypass AVL to GND with a

The MAX16929 is a highly integrated power supply for

automotive TFT-LCD applications. The device integrates

one buck converter, one boost converter, one 1.8V/3.3V

regulator controller, one positive-gate voltage regulator,

and one negative-gate voltage regulator.

1FF ceramic capacitor.

Spread-Spectrum Modulation

The buck converter features spread-spectrum operation

that varies the internal operating frequency of the buck

converter by +6% relative to the switching frequency of

2.1MHz (typ).

The device achieves enhanced EMI performance through

spread-spectrum modulation. Digital input control allows

the device to be placed in a low-current shutdown mode

and provides flexible sequencing of the gate voltage

regulators.

Soft-Start

The buck converter features an internal soft-start timer.

The output voltage takes 3.9ms to ramp up to its set

voltage. If a short circuit or undervoltage is encountered

after the soft-start timer has expired, the device enters

hiccup mode, during which soft-start is reattempted

every 16ms. This process repeats until the short circuit

has been removed.

Internal thermal shutdown circuitry protects the device

from overheating. The buck converter is designed to

shut down when its die temperature reaches +175NC

(typ), while the boost circuitry does so at +165NC (typ).

Each resumes normal operation once its die temperature

has fallen 15NC below its respective thermal shutdown

temperature.

Overcurrent Protection

The device enters hiccup mode in one of three ways. If

eight consecutive current limits are detected and the out-

put is below 77% of its nominal value, the buck converter

enters hiccup mode. The converter enters hiccup mode

immediately if the output is short circuited to ground

(output below 1V). Additionally, the device enters hiccup

mode if 256 consecutive overcurrent events are detected

when the output is greater than 77% of its nominal value.

In hiccup mode, the buck controller idles for 16ms before

reattempting soft-start.

The device is factory-trimmed to provide a variety of

power options to meet the most common automotive

TFT-LCD display power requirements, as outlined in the

Ordering Information/Selector Guide table.

Buck Converter

The device features a current-mode buck converter with

an integrated high-side FET, which requires no external

compensation network. The buck converter regulates the

output voltage to within Q3% in continuous mode over

line and load conditions. The high 2.1MHz (typ) switching

frequency allows for small external components, reduced

output ripple, and guarantees no AM interference.

Power Good (PGOOD)

When an overcurrent condition causes the buck output to

fall below 92% of its set voltage, the open-drain power-

good indicator output (PGOOD) asserts low. PGOOD

deasserts once the output voltage has risen above 95%

of its set voltage.

A power-good (PGOOD) indicator is available to moni-

tor output-voltage quality. The enable input allows the

device to be placed in shutdown, reducing supply cur-

rent to 70FA.

PGOOD serves as a general fault indicator for all the

converters and regulators. Besides indicating an under-

voltage on the buck output, it also indicates any of the

faults listed in the Fault Conditions and PGOOD section.

The buck converter comes with a preset output voltage

of either 3.3V or 5V, and can deliver either 1.2A or 2A to

the output.

Enable (ENB)

Connect ENB to INB for always-on operation. ENB is also

compatible with 3.3V logic systems and can be con-

trolled through a microcontroller or by automotive KEY or

CAN inhibit signals.

Boost Converter

The boost converter employs a current-mode, fixed-

frequency PWM architecture to maximize loop bandwidth

and provide fast transient response to pulsed loads typical

of TFT-LCD panel source drivers. The 2.2MHz switching

frequency allows the use of low-profile inductors and

ceramic capacitors to minimize the thickness of LCD panel

designs. The integrated low on-resistance MOSFET and

Internal 5V Regulator (AVL)

AVL is an internal 5V regulator that supplies power to the

buck controller and charges the boost capacitor. After

�� ꢀMaxim Integrated Productsꢀ ꢀ 13

MAX16929

Automotive TFT-LCD Power Supply with Boost

Converter and Gate Voltage Regulators

the device’s built-in digital soft-start functions reduce the

number of external components required while control-

ling inrush currents. The output voltage can be set from

to 1V and changes the COMPV output. The voltage at

COMPV sets the peak inductor current. As the load var-

ies, the error amplifier sources or sinks current to the

COMPV output accordingly to produce the peak induc-

tor current necessary to service the load. To maintain

stability at high duty cycles, a slope-compensation sig-

nal (set by the capacitor at COMPI) is summed with the

current-sense signal. On the rising edge of the internal

clock, the controller turns on the n-channel MOSFET and

applies the input voltage across the inductor. The current

through the inductor ramps up linearly, storing energy in

its magnetic field. Once the sum of the current feedback

signal and the slope compensation exceeds the COMPV

voltage, the controller turns off the MOSFET. The inductor

current then flows through the diode to the output. The

MOSFET remains off for the rest of the clock cycle.

V

INA

to 18V with an external resistive voltage-divider.

The regulator controls the output voltage by modulat-

ing the duty cycle (D) of the internal power MOSFET in

each switching cycle. The duty cycle of the MOSFET is

approximated by:

ηV

IN

D=1−

V

O

where V is the voltage at INA, V = V (the boost

SH

IN

O

output voltage), and E is the efficiency of the boost con-

verter, as shown in the Typical Operating Characteristics.

Figure 1 shows the functional diagram of the boost

regulator. An error amplifier compares the signal at FBP

LXP

CLOCK

LOGIC AND

DRIVER

PGNDP

I

LIM

COMPARATOR

SOFT-

START

V

LIMIT

PWM

COMPARATOR

CURRENT

SENSE

Σ

2.2MHz

OSCILLATOR

SLOPE

COMP

COMPI

FBP

ERROR

AMP

TO FAULT LOGIC

0.85V

FAULT

COMPARATOR

1V

MAX16929

COMPV

Figure 1. Boost Converter Functional Diagram

�� ꢀMaxim Integrated Productsꢀ ꢀ 14

MAX16929

Automotive TFT-LCD Power Supply with Boost

Converter and Gate Voltage Regulators

The external p-channel FET controlled by GATE protects

the output during fault conditions and provides True

Shutdown of the converter. Connect a pullup resistor

between GATE and INA (see the Boost Converter section

to select the value for the pullup resistor). Under normal

operation, GATE turns on the p-channel FET, connecting

the supply to the boost input. During a fault condition or

in shutdown, GATE is off and the pullup resistor turns off

the p-channel FET, disconnecting the supply from the

boost input.

4) The LXP voltage is greater than 21V (typ).

5) The positive charge-pump voltage (V ) is greater

CP

than 30.5V (typ).

6) The 1.8V/3.3V regulator output voltage falls below

85% of its nominal value.

7) The buck output voltage falls below 92% of its nominal

value.

If any of the first three fault conditions persists for longer

than the 238ms fault blanking period, the device pulls

PGOOD low, turns off all outputs, and starts the autoretry

timer.

Spread-Spectrum Modulation

The high-frequency 2.2MHz operation of the boost con-

verter keeps switching noise outside of the AM band. The

device achieves enhanced EMI performance by modu-

lating the switching frequency by Q4%. The modulating

signal is pseudorandom and changes each switching

If either condition 4 or 5 occurs, the device pulls PGOOD

low and turns off all outputs immediately. The device initi-

ates startup only after the fault has cleared.

If condition 6 occurs, the device pulls PGOOD low, but

does not turn off any of the outputs.

period (i.e., f = 2.2MHz).

SS

Startup

During startup, PGOOD is masked and goes high as

soon as the 1.8V/3.3V regulator controller turns on. This

Immediately after power-up, coming out of shutdown,

or going into autoretry, the boost converter performs a

short-circuit detection test on the output by connecting

the input (INA) to the switching node (LXP) through an

internal 50I resistor.

regulator turns on as soon as V

undervoltage lockout threshold.

exceeds the INA

INA

Autoretry

When the autoretry counter finishes incrementing after

1.9s, the device attempts to turn on the boost converter

and gate voltage regulators in the order shown in

Table 1. The device continues to autoretry as long as the

fault condition persists. A fault on the 1.8V/3.3V regulator

output causes PGOOD to go low, but does not result in

the device shutting down and going into autoretry.

If the resulting voltage on LXP exceeds 1.2V, the device

turns on the external pMOS switch by pulling GATE low.

The boost output ramps to its final value in 15ms.

An overloaded or shorted output is detected if the result-

ing voltage on LXP is below 1.2V. The external pMOS

switch remains off and the converter does not switch.

After the fault blanking period of 238ms, the device pulls

PGOOD low and starts the autoretry timer.

Current Limit

The effective current limit of the boost converter is

reduced by the internally injected slope compensation by

an amount dependent on the duty cycle of the converter.

The effective current limit is given by:

The short-circuit detection feature places a lower limit

on the output load of approximately 46I when the input

voltage is 3V.

Fault Conditions and PGOOD

PGOOD signals whether all the regulators and the boost

converter are operating normally. PGOOD is an open-

drain output that pulls low if any of the following faults

occur:

D

-12

I

=192 ×10

×I

×

LIM(EFF)

LIM_DC_0

C

COMPI

where I

is the effective current limit, I

=

LIM(EFF)

LIM_DC_0

1.1A or 2.2A depending on the boost converter current-

limit option, D is the duty cycle of the boost converter,

1) The boost output voltage falls below 85% of its set

value.

and C

is the value of the capacitor at the COMPI

input. Estimate the duty cycle of the converter using the

formulas shown in the Design Procedure section.

COMPI

2) The positive-gate voltage regulator output (V ) falls

GH

below 85% of its set value.

3) The negative-gate voltage regulator output (V ) falls

GL

below 85% of its set value.

�� ꢀMaxim Integrated Productsꢀ ꢀ 15

MAX16929

Automotive TFT-LCD Power Supply with Boost

Converter and Gate Voltage Regulators

dependent bypassing requirements. Connect a ceramic

capacitor between the collector and ground with the

value shown in Table 3.

1.8V/3.3V Regulator Controller

The 1.8V/3.3V regulator controller delivers 4.5mA (min)

to an external load. Connect FB to DR for a regulated

1.8V/3.3V output.

The regulator derives its negative supply voltage from an

inverting charge pump, a single-stage example of which

is shown in the Typical Application Circuit. A more nega-

tive voltage using a multistage charge pump is possible

as described in the Charge Pumps section.

For higher output capability, use an external npn transis-

tor as shown in the Typical Application Circuit. The drive

capability of the regulator is then increased by the cur-

rent gain of the transistor (h ). When using an external

FE

transistor, use DR as the base drive and connect FB to

the transistor’s emitter. Bypass the base to ground with a

0.1FF ceramic capacitor.

The external npn transistor is not short-circuit protected.

To maintain proper pulldown capability of external npn

transistor and optimal regulation, a minimum load of at

least 500µA is recommended on the output of the GL

regulator.

If the boost output current is greater than 300mA, con-

nect a 30kI resistor between DR and GND.

Enable (ENP)

Use the enable input (ENP) to enable and disable the

boost section of the device. Connect ENP to INA for

normal operation and to GND to place the device in shut-

down. In shutdown, the INA supply current is reduced to

0.5FA.

Positive-Gate Voltage Regulator (GH)

The positive-gate voltage regulator includes a p-channel

FET output stage to generate a regulated output between

+5V and V - 2V. The regulator maintains accuracy over

CP

wide line and load conditions. It is capable of at least

20mA of output current and includes current-limit protec-

tion. V

drivers’ gate-on voltage.

is typically used to provide the TFT-LCD gate

GH

Soft-Start and Supply Sequencing (SEQ)

When enabled, the boost output ramps up from V

to

INA

The regulator derives its positive supply voltage from a

noninverting charge pump, a single-stage example of

which is shown in the Typical Application Circuit. A high-

er voltage using a multistage charge pump is possible,

as described in the Charge Pumps section.

its set voltage. Once the boost output reaches 85% of the

set voltage and the soft-start timer expires, the gate volt-

age regulators turn on in the order shown in Table 1. The

1.8V/3.3V regulator controller is enabled at the beginning

of the boost converter’s soft-start.

Both gate voltage regulators have a 7.45ms soft-start

time. The second one turns on as soon as the output of

the first reaches 85% of its set voltage.

Negative-Gate Voltage Regulator (GL)

The negative-gate voltage regulator is an analog gain

block with an open-drain p-channel output. It drives an

external npn pass transistor with a 6.8kI base-to-emitter

resistor (see the Pass Transistor Selection section). Its

guaranteed base drive source current is at least 2mA.

Thermal Shutdown

Internal thermal shutdown circuitry shuts down the

device immediately when the die temperature exceeds

+165NC. A 15NC thermal shutdown hysteresis prevents

the device from resuming normal operation until the die

temperature falls below +150NC.

V

is typically used to provide the TFT-LCD gate driv-

GL

ers’ gate-off voltage.

The output of the negative-gate voltage regulator (i.e.,

the collector of the external npn pass transistor) has load-

Tableꢀ1.ꢀSupplyꢀSequencing

CONTROLꢀINPUTS

SUPPLYꢀSEQUENCING

ENP

SEQ

FIRST

SECOND

THIRD

0

1

X

0

1

Device is in shutdown

V

GL

SH

GH

V

GH

SH

GL

�� ꢀMaxim Integrated Productsꢀ ꢀ 16

MAX16929

Automotive TFT-LCD Power Supply with Boost

Converter and Gate Voltage Regulators

Tableꢀ2.ꢀMinimumꢀBuckꢀInductorꢀValueꢀ

RequiredꢀforꢀNormalꢀOperationꢀDuringꢀ

LoadꢀDump

Design Procedure

Buck Converter

Inductor Selection

Three key inductor parameters must be specified for

operation with the device: inductance value (L), induc-

BUCKꢀV

(V)

BUCKꢀI

(A)

L

(µH)

MINꢀ

OUTBꢀ

3.3

1.2

3.3

tor saturation current (I

), and DC resistance (R ).

SAT

DC

3.3

5

2

6.8

3.3

4.7

To determine the inductance value, select the ratio of

inductor peak-to-peak ripple current to average output

current (LIR) first. For LIR values that are too high, the

RMS currents are high, and therefore I²R losses are high.

Use high-valued inductors to achieve low LIR values.

Typically, inductance is proportional to resistance for a

given package type, which again makes I²R losses high

for very low LIR values. A good compromise between

size and loss is to select a 30%-to-60% peak-to-peak

ripple current to average-current ratio. If extremely thin

high-resistance inductors are used, as is common for

LCD-panel applications, the best LIR can increase

between 0.5 and 1.0. The size of the inductor is deter-

mined as follows:

1.2

2

5

Capacitor Selection

The input and output filter capacitors should be of a low-

ESR type (tantalum, ceramic, or low-ESR electrolytic) and

should have I

ratings greater than:

RMS

2

LIR

12

I

= I D× (1-D +

)

for the input capacitor

INB(RMS)

O

LIR×I

O

I

=

for the output capacitor

OUTB(RMS)

12

where D is the duty cycle given above.

(V

-V )×D

O

INB

L =

and

The output voltage contains a ripple component whose

peak-to-peak value depends on the value of the ESR

and capacitance of the output capacitor, and is approxi-

mately given by:

LIR×I × f

O

SWB

V

O

D =

η× V

INB

DV

= DV

+ DV

RIPPLE

ESR CAP

where V

is the input voltage, V is the output volt-

O

INB

DV

= LIR x I x R

O ESR

ESR

age, I is the output current, E is the efficiency of the

O

LIR×I

O

buck converter, D is the duty cycle, and f

is 2.1MHz

SWB

∆V

=

CAP

8 × C× f

(the switching frequency of the buck converter). The

efficiency of the buck converter can be estimated from

the Typical Operating Characteristics and accounts for

SWB

Diode Selection

The catch diode should be a Schottky type to minimize

its voltage drop and maximize efficiency. The diode must

be capable of withstanding a reverse voltage of at least

the maximum input voltage in the application. The diode

should have an average forward current rating greater

than:

losses in the internal switch, catch diode, inductor R

and capacitor ESR.

,

DC

To ensure the buck converter does not shut down

during load dump input-voltage transients to 42V, an

inductor value larger than calculated above should be

used. Table 2 lists the minimum inductance that should

be used for proper operation during load dump. The

I

= I × (1-D)

O

D

saturation current rating (I

) must be high enough to

where D is the duty cycle given above. In addition, ensure

that the peak current rating of the diode is greater than:

SAT

ensure that saturation can occur only above the maxi-

mum current-limit value. Find a low-loss inductor having

the lowest possible DC resistance that fits in the allotted

dimensions.

LIR

2





I

× 1+

OUTB









�� ꢀMaxim Integrated Productsꢀ ꢀ 17

MAX16929

Automotive TFT-LCD Power Supply with Boost

Converter and Gate Voltage Regulators

The output voltage contains a ripple component whose

peak-to-peak value depends on the value of the ESR and

capacitance of the output capacitor and is approximately

given by:

Boost Converter

Inductor Selection

Three key inductor parameters must be specified for

operation with the device: inductance value (L), induc-

tor saturation current (I

), and DC resistance (R ).

DV

= DV + DV

ESR CAP

SAT

DC

RIPPLE

To determine the inductance value, select the ratio of

inductor peak-to-peak ripple current to average input

current (LIR) first. For LIR values that are too high, the

RMS currents are high, and therefore I²R losses are high.

Use high-valued inductors to achieve low LIR values.

Typically, inductance is proportional to resistance for a

given package type, which again makes I²R losses high

for very low LIR values. A good compromise between

size and loss is to select a 30%-to-60% peak-to-peak

ripple current to average-current ratio. If extremely thin

high-resistance inductors are used, as is common for

LCD-panel applications, the best LIR can increase

between 0.5 and 1.0. The size of the inductor is deter-

mined as follows:

LIR

2

∆V

=I

× (1+

)×R

ESR

ESR INP

I

×D

O

∆V

=

CAP

C

×f

OUT SW

where I

given above.

and D are the input current and duty cycle

INP

Rectifier Diode

The catch diode should be a Schottky type to minimize

its voltage drop and maximize efficiency. The diode must

be capable of withstanding a reverse voltage of at least

V

SH

. The diode should have an average forward current

rating greater than:

V

×D

V ×I

O O

INA

L =

and I

=

I

= I

× (1-D)

D

INP

LIR×I

× f

ηV

INP SW

INA

where I

and D are the input current and duty cycle

INP

given above. In addition ensure that the peak current rat-

ing of the diode is greater than:

ηV

V

INA

D=1−

O

LIR

2





I

× 1+

INP







where V

is the input voltage, V is the output voltage,

O

INA



I

is the output current, I

is the average boost input

O

INP

current, E is the efficiency of the boost converter, D is the

duty cycle, and f is 2.2MHz (the switching frequency

of the boost converter). The efficiency of the boost

converter can be estimated from the Typical Operating

Characteristics and accounts for losses in the internal

Output-Voltage Selection

The output voltage of the boost converter can be adjust-

ed by using a resistive voltage-divider formed by R

SW

TOP

and R

FBP and connect R

Select R

. Connect R

between the output and

between FBP and GND.

BOTTOM

TOP

BOTTOM

switch, catch diode, inductor R , and capacitor ESR.

DC

in the 10kI to 50kI range. Calculate

BOTTOM

R

with the following equation:

TOP

Capacitor Selection

The input and output filter capacitors should be of a low-

ESR type (tantalum, ceramic, or low-ESR electrolytic) and

V

O

R

=R

× (

BOTTOM

−1)

TOP

V

FBP

should have I

ratings greater than:

RMS

where V

, the boost converter’s feedback set point, is

FBP

LIR×I

INP

1V. Place both resistors as close as possible to the device

and connect R to the analog ground plane.

I

=

for the input capacitor

RMS

12

BOTTOM

Loop Compensation

to set the high-frequency integrator

COMPV

2

LIR

Choose R

D+

12

gain for fast transient response. Choose C

to set

I

=I

O

COMPV

for the output capacitor

RMS

1− D

the integrator zero to maintain loop stability. For low-ESR

output capacitors, use Table 3 to select the initial values

where I

given above.

and D are the input current and duty cycle

INP

for R

lel with R

and C

. Use a 22pF capacitor in paral-

COMPV

+ C

.

COMPV

�� ꢀMaxim Integrated Productsꢀ ꢀ 18

MAX16929

Automotive TFT-LCD Power Supply with Boost

Converter and Gate Voltage Regulators

55FA and the resulting gate source voltage (V ) turns

on the FET. When the gate drive is removed under a fault

GS

Tableꢀ3.ꢀCompensationꢀComponentꢀValues

V

(V)

8

18

200

5

SHꢀ

condition or in shutdown, R

bleeds off charge to turn

SG

off the FET. Size R

on the FET.

to produce the V

needed to turn

I

(mA)

200

3.3

1.75

5

SG

GS

SHꢀ

V

(V)

INAꢀ

P

(W)

3.75

5

1.8V/3.3V Regulator Controller

INꢀ

npn Bipolar Transistor Selection

Lꢀ(µH)

ꢀ(kI)

There are two important considerations in selecting the

pass npn bipolar transistor: current gain (h ) and power

R

33

39

COMPV

FE

C

ꢀ(pF)

220

820

180

330

COMPV

dissipation. Select a transistor with an h high enough to

FE

C

ꢀ(pF)

COMPI

ensure adequate drive capability. This condition is satis-

fied when I

x (h + 1) is greater than the maximum

DR

FE

load current. The regulator can source I = 4.5mA (min).

The transistor should be capable of dissipating:

DR

V

SH

V

CP

P

= (V

- V

) × I

NPN_REG

INA

REG_OUT LOAD(MAX)

where V

= 1.8V or 3.3V. Bypass DR to ground

REG_OUT

LXP

with a 0.1FF ceramic capacitor. For applications in which

the boost output current exceeds 300mA, connect a

30kI resistor from DR to ground.

Supply Considerations

INA needs to be at least 4.5V for the 3.3V regulator to

operate properly.

Figure 2. Multistage Charge Pump for Positive Output Voltage

V

CN

Charge Pumps

Selecting the Number of Charge-Pump Stages

For most applications, a single charge-pump stage is

sufficient, as shown in the Typical Application Circuit.

Connect the flying capacitors to LXP. The output voltages

generated on the storage capacitors are given by:

LXP

V

= 2 x V + V

- 2 x V

CP

SH

SCHOTTKY D

Figure 3. Multistage Charge Pump for Negative Output Voltage

V

= -(V + V

- 2 x V )

CN

SH

SCHOTTKY D

where V is the positive supply for the positive-gate volt-

To further optimize transient response, vary R

COMPV

CP

age regulator, and V

is the negative supply for the neg-

in 20% steps and C

in 50% steps while observ-

CN

COMPV

ative-gate voltage regulator. Where larger output voltages

are needed, use multistage charge pumps (however, the

maximum charge-pump voltage is limited by the absolute

maximum ratings of CP and DRVN). Figure 2 and Figure 3

show the configuration of a multistage charge pump for

both positive and negative output voltages.

ing transient-response waveforms. The ideal transient

response is achieved when the output settles quickly with

little or no overshoot. Connect the compensation network

to the analog ground plane.

Use the following formula to calculate the value for C

-6

:

COMPI

C

≤ 950 × 10 × L/(V + V

- V

)

COMPI

SH

SCHOTTKY

INA

For mutistage charge pumps the output voltages are:

p-Channel FET Selection

The p-channel FET used to gate the boost converter’s

input should have low on-resistance. Connect a resistor

V

CP

V

= V + n × (V + V

- 2 x V )

SH

SCHOTTKY D

= -n × (V + V

- 2 x V )

D

CN

SH

SCHOTTKY

(R ) between the source and gate of the FET. Under

For highest efficiency, choose the lowest number of

charge-pump stages that meets the output requirement.

SG

normal operation, R

carries a gate drive current of

SG

�� ꢀMaxim Integrated Productsꢀ ꢀ 19

MAX16929

Automotive TFT-LCD Power Supply with Boost

Converter and Gate Voltage Regulators

The number of positive charge-pump stages needed is

given by:

switching frequency of the boost converter, and V

RIPPLE_

is the peak-to-peak value of the output ripple.

CP

V

+V

− V

− 2 × V

For the inverting charge pump connected to CN, use the

following equation to approximate the required output

capacitance:

GH DROPOUT

SH

n

=

CP

V

+V

SH SCHOTTKY D

and the number of negative charge-pump stages is

given by:

(1-D)×I

LOAD_CN

C

≥

OUT_CN

f

× V

RIPPLE_CN

SW

|V |+V

GL

DROPOUT

n

=

CN

where C

is the output capacitor of the charge

pump, D is the duty cycle of the boost converter,

OUT_CN

V

+ V

− 2 × V

SH

SCHOTTKY D

where n

is the number of positive charge-pump stag-

is the number of negative charge-pump stages,

I

is the load current of the charge pump, f

CP

LOAD_CN SW

es, n

is the switching frequency of the boost converter, and

V

ripple.

CN

V

GH

is the positive-gate voltage regulator output volt-

is the peak-to-peak value of the output

RIPPLE_CN

age, V

is the negative-gate voltage regulator output

GL

voltage, V

is the boost converter’s output voltage, V

SH

D

Charge-Pump Rectifier Diodes

is the forward-voltage drop of the charge-pump diode,

is the forward drop of the Schottky diode

Use high-speed silicon switching diodes with a current

rating equal to or greater than two times the average

charge-pump input current. If it helps avoid an extra

stage, some or all of the diodes can be replaced with

Schottky diodes with an equivalent current rating.

V

SCHOTTKY

of the boost converter, and V

is the dropout

= 0.3V for the

= 2V at 20mA

DROPOUT

margin for the regulator. Use V

negative voltage regulator and V

for the positive-gate voltage regulator.

Positive-Gate Voltage Regulator

Flying Capacitors

Increasing the flying capacitor (C ) value lowers the

Output-Voltage Selection

The output voltage of the positive-gate voltage regula-

tor can be adjusted by using a resistive voltage-divider

X

effective source impedance and increases the output

current capability. Increasing the capacitance indefi-

nitely has a negligible effect on output current capability

because the internal switch resistance and the diode

impedance place a lower limit on the source impedance.

A 0.1FF ceramic capacitor works well in most low-current

applications. The voltage rating of the flying capacitors

formed by R

and R

. Connect R

between

TOP

BOTTOM

TOP

the output and FBGH, and connect R

FBGH and GND. Select R

BOTTOM

in the 10kI to 50kI

BOTTOM

range. Calculate R

with the following equation:

TOP

V

GH

R

= R

× (

BOTTOM

−1)

for the positive charge pump should exceed V , and

TOP

CP

V

FBGH

that for the negative charge pump should exceed the

magnitude of V

.

CN

where V

is the desired output voltage and V

= 1V

FBGH

GH

(the regulated feedback voltage for the regulator). Place

both resistors as close as possible to the device.

Charge-Pump Output Capacitor

Increasing the output capacitance or decreasing the ESR

reduces the output-ripple voltage and the peak-to-peak

transient voltage. With ceramic capacitors, the output-

voltage ripple is dominated by the capacitance value.

Use the following equation to approximate the required

output capacitance for the noninverting charge pump

connected to CP:

Avoid excessive power dissipation within the internal

pMOS device of the regulator by paying attention to the

voltage drop across the drain and source. The amount of

power dissipation is given by:

P

GL

= (V - V ) × I

CP GH LOAD(MAX)

where V

age applied to the drain, V

voltage, and I is the maximum load current.

is the noninverting charge-pump output volt-

CP

D×I

LOAD_CP

is the regulated output

GH

C

≥

OUT_CP

f

× V

RIPPLE_CP

SW

LOAD(MAX)

Stability Requirements

The positive-gate voltage regulator (GH) requires a

minimum output capacitance for stability. For an output

where C

is the output capacitor of the charge

OUT_CP

pump, D is the duty cycle of the boost converter, I

LOAD_

is the

SW

is the load current of the charge pump, f

CP

�� ꢀMaxim Integrated Productsꢀ ꢀ 20

MAX16929

Automotive TFT-LCD Power Supply with Boost

Converter and Gate Voltage Regulators

voltage of 5V to (V

10mA to 15mA, use a minimum capacitance of 0.47FF.

- 2V) and an output current of

output voltage applied to the emitter of the transistor,

and I is the maximum load current. Note

that the external transistor is not short-circuit protected.

CP

LOAD(MAX)_GL

Negative-Gate Voltage Regulator

Stability Requirements

The device’s negative-gate voltage regulator uses an

internal transconductance amplifier to drive an external

pass transistor. The transconductance amplifier, the

pass transistor, the base-emitter resistor, and the output

capacitor determine the loop stability.

Output-Voltage Selection

The output voltage of the negative-gate voltage regula-

tor can be adjusted by using a resistive voltage-divider

formed by R

REF and FBGL, and connect R

and R

. Connect R

between

TOP

BOTTOM

TOP

between FBGL

BOTTOM

and the collector of the external npn transistor. Select

greater than 20kI to avoid loading down the ref-

R

TOP

The transconductance amplifier regulates the output volt-

age by controlling the pass transistor’s base current. The

total DC loop gain is approximately:

erence output. Calculate R

equation:

with the following

BOTTOM

V

− V

GL

FBGL

R

= R

×

TOP

I

×h

I

LOAD

BOTTOM

4

BIAS

FE

V

− V

A

≅ ( )× (1+

)× V

REF

FBGL

V_GL

V

T

where V

is the desired output voltage, V

= 0.25V (the regulated feedback voltage of

= 1.25V,

GL

REF

where V is 26mV at room temperature, and I

current through the base-to-emitter resistor (R ). For

is the

BIAS

BE

T

and V

FBGL

the regulator).

the device, the bias current for the negative-gate voltage

regulator is 0.1mA. Therefore, the base-to-emitter resistor

should be chosen to set 0.1mA bias current:

Pass Transistor Selection

The pass transistor must meet specifications for current

gain (h ), input capacitance, collector-emitter saturation

FE

V

0.7V

BE

voltage, and power dissipation. The transistor’s current

gain limits the guaranteed maximum output current to:

R

=

= 7kΩ

BE

0.1mA 0.1mA

Use the closest standard resistor value of 6.8kI. The

output capacitor and the load resistance create the

dominant pole in the system. However, the internal

amplifier delay, pass transistor’s input capacitance,

and the stray capacitance at the feedback node create

additional poles in the system, and the output capacitor’s

ESR generates a zero. For proper operation, use the fol-

lowing equations to verify that the regulator is properly

compensated:

V

BE

I

= (I

−

)×h

FE(MIN)

LOAD(MAX)

DRVN

R

BE

where I

rent, V

is the minimum guaranteed base-drive cur-

is the transistor’s base-to-emitter forward volt-

DRVN

BE

age drop, and R

is the pulldown resistor connected

BE

between the transistor’s base and emitter. Furthermore,

the transistor’s current gain increases the regulator’s DC

loop gain (see the Stability Requirements section), so

excessive gain destabilizes the output.

1) First, determine the dominant pole set by the regula-

tor’s output capacitor and the load resistor:

The transistor’s saturation voltage at the maximum output

current determines the minimum input-to-output volt-

age differential that the regulator can support. Also, the

package’s power dissipation limits the usable maximum

input-to-output voltage differential. The maximum power-

dissipation capability of the transistor’s package and

mounting must exceed the actual power dissipated in

the device. The power dissipated equals the maximum

I

LOAD(MAX)_GL

f

=

POLE_GL

2π × C

× V

OUT_GL

The unity-gain crossover frequency of the regulator is:

= A × f

f

CROSSOVER

V_GL

POLE_GL

2) The pole created by the internal amplifier delay is

approximately 1MHz:

load current (I

) multiplied by the maximum

LOAD(MAX)_GL

input-to-output voltage differential:

= (V - V ) × I

LOAD(MAX)_GL

f

= 1MHz

POLE_AMP

P

NPN_GL

GL

CN

3) Next, calculate the pole set by the transistor’s input

capacitance, the transistor’s input resistance, and the

base-to-emitter pullup resistor:

where V

tor of the transistor, V

is the regulated output voltage on the collec-

GL

is the inverting charge-pump

CN

�� ꢀMaxim Integrated Productsꢀ ꢀ 21

MAX16929

Automotive TFT-LCD Power Supply with Boost

Converter and Gate Voltage Regulators

Tableꢀ4.ꢀMinimumꢀOutputꢀCapacitanceꢀvs.ꢀ

OutputꢀVoltageꢀRangeꢀforꢀNegative-Gateꢀ

1

f

=

POLE_IN

2π × C × (R /R

)

IN

BE IN

VoltageꢀRegulatorꢀ(I ꢀ=ꢀ10mAꢀtoꢀ15mA)

OUT

where:

g

h

FE

OUTPUTꢀVOLTAGEꢀ

RANGE

MINIMUMꢀOUTPUTꢀ

CAPACITANCEꢀ(µF)

m

C

=

, R

=

IN

2πf

g

T

m

-2V R V Rꢀ-4V

2.2

1.5

1

GL

g

is the transconductance of the pass transistor, and

m

-5V R V R -7V

GL

f is the transition frequency. Both parameters can be

T

-8V R V R -13V

found in the transistor’s data sheet. Because R

is

GL

BE

much greater than R , the above equation can be

simplified:

IN

Applications Information

1

2π × C ×R

f

=

POLE_IN

Power Dissipation

An IC’s maximum power dissipation depends on the ther-

mal resistance from the die to the ambient environment

and the ambient temperature. The thermal resistance

depends on the IC package, PCB copper area, other

thermal mass, and airflow. More PCB copper, cooler

ambient air, and more airflow increase the possible dis-

sipation, while less copper or warmer air decreases the

IC’s dissipation capability. The major components of

power dissipation are the power dissipated in the buck

converter, boost converter, positive-gate voltage regula-

tor, negative-gate voltage regulator, and the 1.8V/3.3V

regulator controller.

IN

Substituting for C and R yields:

IN

f

T

f

=

POLE

h

FE

4) Next, calculate the pole set by the regulator’s feed-

back resistance and the capacitance between FBGL

and GND (including stray capacitance):

1

× (R

f

=

POLE_FBGL

2π × C

/R

)

FBGL

TOP BOTTOM

where C

GND and is equal to 30pF, R

of the regulator’s feedback divider, and R

the lower resistor of the divider.

is the capacitance between FBGL and

FBGL

is the upper resistor

TOP

Buck Converter

In the buck converter, conduction and switching losses

in the internal MOSFET are dominant. Estimate these

losses using the following formula:

is

BOTTOM

5) Next, calculate the zero caused by the output capaci-

tor’s ESR:

2

P

≈ [(I

× √D) × R

OUTB

] + [0.5 × V

×

LXB

OUTB

DS_ON(LXB)

INB

1

I

× (t + t ) × f

]

f

=

R

F

SWB

ZERO_ESR

2π × C

×R

ESR

OUT_LR

where I

is the output current, D is the duty cycle

OUTB

of the buck converter, R

is the on-resistance

is the input voltage,

(t + t is the time is takes for the switch current and

where R

C

is the equivalent series resistance of

DS_ON(LXB)

ESR

of the internal high-side FET, V

. To ensure stability, make C

large

INB

OUT_LR

enough so the crossover occurs well before the poles

and zero calculated in steps 2 to 5. The poles in steps

3 and 4 generally occur at several MHz and using

ceramic capacitors ensures the ESR zero also occurs

at several MHz. Placing the crossover frequency below

500kHz is sufficient to avoid the amplifier delay pole

and generally works well, unless unusual component

choices or extra capacitances move one of the other

poles or the zero below 1MHz.

R

F)

voltage to settle to their final values during the rising and

falling transitions, and f

of the buck converter. R

is the switching frequency

is 180mI (typ) and

(t + t is 4.4ns + 4.6ns = 9ns at V = 12V.

SWB

DS_ON(LXB)

R

F)

INB

Table 4 is a list of recommended minimum output capaci-

tance for the negative-gate voltage regulator and are

applicable for output currents in the 10mA to 15mA range.

�� ꢀMaxim Integrated Productsꢀ ꢀ 22

MAX16929

Automotive TFT-LCD Power Supply with Boost

Converter and Gate Voltage Regulators

Boost Converter

Power dissipation in the boost converter is primarily due

to conduction and switching losses in the low-side FET.

Conduction loss is produced by the inductor current

flowing through the on-resistance of the FET during the

on-time. Switching loss occurs during switching transi-

tions and is a result of the finite time needed to fully turn

on and off the FET. Power dissipation in the boost con-

verter can be estimated with the following formula:

Positive-Gate Voltage Regulator

Use the lowest number of charge-pump stages possible

in supplying power to the positive-gate voltage regulator.

Doing so minimizes the drain-source voltage of the inte-

grated pMOS switch and power dissipation. The power

dissipated in the switch is given as:

P

GH

= (V - V ) × I

CP GH LOAD(MAX)_GH

Ensure that the voltage on CP does not exceed the

CP overvoltage threshold as given in the Electrical

Characteristics table.

2

P

≈ [(I

IN(DC,MAX)

× √D) × R

] + V

×

LXP

IN(DC,MAX)

DS_ON(LXP)

SH

I

× f

× [(t

+ t ) + (t + t )]

SW

R-V F-I R-I F-V

Negative-Gate Voltage Regulator

Use the lowest number of charge-pump stages possible

to provide the negative voltage to the negative-gate

voltage regulator. Estimate the power dissipated in the

negative-gate voltage regulator using the following:

where I

is the maximum expected average

IN(DC,MAX)

input (i.e., inductor) current, D is the duty cycle of the

boost converter, R

the internal low-side FET, V

is the on-resistance of

is the output voltage, and

DS_ON(LXP)

SH

f

R

is the switching frequency of the boost converter.

SW

P

GL

= (V + |V | - V ) × I

INA CN BE DRVN

is 110mI (typ) and f

is 2.2MHz.

DS_ON(LXP)

SW

where V is the base-emitter voltage of the external npn

BE

The voltage and current rise and fall times at the LXP

node are equal to t (voltage rise time), t (voltage fall

bipolar transistor, and I

is the current sourced from

DRVN

R-V

F-V

DRVN to the R

bias resistor and to the base of the

BE

time), t

(current rise time), and t (current fall time),

R-I

F-I

transistor, which is given by:

and are determined as follows:

V

I

GL

BE

I

=

+

DRVN

V

+ V

SCHOTTKY

SH

R

h

+1

t

=

BE

FE

R-V

F-V

K

R-V

1.8V/3.3V Regulator Controller

The power dissipated in the 1.8V/3.3V regulator controller

is given by:

V

+ V

K

SH

SCHOTTKY

F-V

P

= (V

- V

- V ) × I

REG

INA

OUT_REG BE DR

I

IN(DC,MAX)

where V

= 1.8V or 3.3V, V is the base-emitter

BE

t

=

OUT_REG

R-I

F-I

K

voltage of the external npn bipolar transistor, and I

is

R-I

DR

the current sourced from DR to the base of the transistor.

is given by:

I

IN(DC,MAX)

I

DR

t

K

I

F-I

LOAD

I

=

DR

h

+1

FE

K

R-V

, K , K , and K

are the voltage and current

F-I

slew rates of the LXP node and are supply dependent.

Use Table 5 to determine their values.

F-V

R-I

where I

is load current of the 1.8V/3.3V regulator

FE

LOAD

controller, and h is the current gain of the transistor.

Tableꢀ5.ꢀLXPꢀVoltageꢀandꢀCurrentꢀSlewꢀRatesꢀvs.ꢀSupplyꢀVoltage

LXPꢀVOLTAGEꢀANDꢀCURRENTꢀSLEWꢀRATES

RISINGꢀVOLTAGE

SLEWꢀRATE

FALLINGꢀVOLTAGE

SLEWꢀRATE

RISINGꢀCURRENT

SLEWꢀRATE

FALLINGꢀCURRENT

SLEWꢀRATE

V

ꢀ(V)

INA

K

ꢀ(V/ns)

K

ꢀ(V/ns)

K

ꢀ(A/ns)

K

ꢀ(A/ns)

R-V

F-V

R-I

F-I

3.3

0.52

1.35

1.7

2

0.13

0.3

0.38

0.44

5

�� ꢀMaxim Integrated Productsꢀ ꢀ 23

MAX16929

Automotive TFT-LCD Power Supply with Boost

Converter and Gate Voltage Regulators

Total Power Dissipation

The total power dissipated in the package is the sum of

the losses previously calculated. Therefore, total power

dissipation can be estimated as follows:

2) Connect input and output capacitors to the power

ground planes; connect all other capacitors to the

analog ground plane.

3) Keep the high-current paths as short and wide as

possible. Keep the path of switching currents short.

P = P

+ P

+ P + P + P

GH GL REG

T

LXB

LXP

Achieve maximum heat transfer by connecting the exposed

pad to a thermal landing pad and connecting the thermal

landing pad to a large ground plane through thermal vias.

4) Place the feedback resistors as close to the IC as

possible. Connect the negative end of the resistive

divider and the compensation network to the analog

ground plane.

Layout Considerations

Careful PCB layout is critical in achieving stable and

optimized performance. Follow the following guidelines

for good PCB layout:

5) Route the high-speed switching node LXB and LXP

away from sensitive analog nodes (FB, FBP, FBGH,

FBGL, FBB, and REF).

Refer to the MAX16929 Evaluation Kit data sheet for a

recommended PCB layout.

1) Place decoupling capacitors as close as possible to

the device. Connect the power ground planes and the

analog ground plane together at one point close to the

device.

Ordering Information/Selector Guide

REGULATOR

ꢀ(V)

BUCK

BOOST

(A)

PART

PIN-PACKAGE

V

(V)

I

ꢀ(A)

OUTB

2

I

LIM

REG

OUTBꢀ

MAX16929AGUI/V+

MAX16929BGUI/V+

MAX16929CGUI/V+

MAX16929DGUI/V+

MAX16929EGUI/V+

MAX16929FGUI/V+

MAX16929GGUI/V+

MAX16929HGUI/V+

MAX16929IGUI/V+

3.3

1.8

3.3

1.8

5

1.5

28 TSSOP-EP*

2

1.5

3.3

5

2

1.5

2

0.75

5

1.2

2

5

1.2

2

3.3

1.2

Note: All devices are specified over the -40°C to +105°C operating temperature range.

/V denotes an automotive qualified part.

+Denotes a lead(Pb)-free/RoHS-compliant package.

*EP = Exposed pad.

Chip Information

Package Information

For the latest package outline information and land patterns

(footprints), go to www.maxim-ic.com/packages. Note that a

“+”, “#”, or “-” in the package code indicates RoHS status only.

Package drawings may show a different suffix character, but

the drawing pertains to the package regardless of RoHS status.

PROCESS: BiCMOS

PACKAGEꢀ

TYPE

PACKAGEꢀ

CODE

OUTLINEꢀ

NO.

LANDꢀ

PATTERNꢀNO.

28 TSSOP-EP

U28ME+1

21-0108

90-0147

�� ꢀMaxim Integrated Productsꢀ ꢀ 24

MAX16929

Automotive TFT-LCD Power Supply with Boost

Converter and Gate Voltage Regulators

Typical Application Circuit

OUTB

R

COMPV

C

COMPI

C

COMPV

COMPI

COMPV

GATE

INA

L

P

OPTIONAL

LXP

V

SH

DR

FB

V TO 18V

INA

BOOST

1.8V/3.3V

REGULATOR

CONTROLLER

PGNDP

FBP

V

REG

1.8V/3.3V

LXP

V

OSCILLATOR

CN

V

CN

CP

DRVN

FBGL

V

SH

POSITIVE

GATE

VOLTAGE

REGULATOR

NEGATIVE

GATE

VOLTAGE

REGULATOR

GH

V

GH

V

GL

FBGH

BST

INB

4V TO 28V

REF

BANDGAP

REFERENCE

3.3V/5V

BUCK

GND

LXB

OUTB

ENP

SEQ

INA

CONTROL

FBB

ENB

AVL

GND

PGOOD

MAX16929

Automotive TFT-LCD Power Supply with Boost

Converter and Gate Voltage Regulators

Revision History

REVISION REVISION

PAGES

CHANGED

DESCRIPTION

NUMBER

DATE

5/11

9/11

1/12

0

1

2

Initial release

Removed ENB from the FBB to GND range in the Absolute Maximum Ratings

Corrected the C formula in the Loop Compensation section

—

2

18

COMPI

Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied.

Maxim reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and max limits) shown in the Electrical

Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.

Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600

25

©

2012 Maxim Integrated Products

Maxim is a registered trademark of Maxim Integrated Products, Inc.


型号：	MAX16929
厂家：	MAXIM INTEGRATED PRODUCTS
描述：	Automotive TFT-LCD Power Supply with Boost Converter and Gate Voltage Regulators 车载TFT -LCD电源与升压转换器和栅极电压稳压器转换器栅极稳压器 CD 升压转换器
文件：	总25页 (文件大小：2347K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

MAX16929 [MAXIM]

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