MAX17292ETCE [MAXIM]
2.5V to 36V, 2.5MHz, PWM Boost Controller with 4μA Shutdown Current and Reduced EMI;
| 型号： | MAX17292ETCE |
| 厂家： | 
MAXIM INTEGRATED PRODUCTS |
| 描述： | 2.5V to 36V, 2.5MHz, PWM Boost Controller with 4μA Shutdown Current and Reduced EMI |
| 文件： | 总18页 (文件大小：1340K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
EVALUATION KIT AVAILABLE
MAX17290/MAX17292
2.5V to 36V, 2.5MHz, PWM Boost Controller
with 4µA Shutdown Current and Reduced EMI
General Description
Benefits and Features
● Reduces Solution Size and Cost
• All-Ceramic Capacitor Solution Allows Ultra-Compact
Solution Size
The MAX17290/MAX17292 high-efficiency, synchronous
step-up DC-DC controllers operate over a 4.5V to 36V
input voltage range with 42V input transient protection.
The input operating range can be extended to as low as
2.5V in Bootstrapped mode.
• 100kHz to 1MHz (MAX17290) and 1MHz to
2.5MHz (MAX17292) Switching-Frequency with
External Synchronization
The MAX17290 and MAX17292 use a constant-frequency,
pulse-width modulating (PWM), peak current-mode
control architecture. There are multiple versions of the
devices offering one or more of the following functions:
a synchronization output (SYNCO) for 180° out-of-phase
operation, an overvoltage protection function using a
separate input pin (OVP), and a reference input pin
(REFIN) to allow on-the-fly output voltage adjustment.
● Increases Design Flexibility
• Bootstrapped Mode Allows Input Voltage to be 2.5V
• Adjustable Slope Compensation
● Reduces Power Dissipation
• >90% Peak Efficiency
• Low 4μA (typ.) Shutdown Current
● Operates Reliably
The MAX17290 and MAX17292 operate in different
frequency ranges. All versions can be synchronized to an
external master clock using the FSET/SYNC input.
• 42V Input Voltage Transient Protection
• Fixed 9ms Internal Software Start Reduces Input
Inrush Current
• PGOOD Output and Hiccup Mode for Enhanced
System Protection
• Overtemperature Shutdown
• Reduced EMI Emission with Spread-Spectrum
Control
The devices are available in a compact 12-pin (3mm x
3mm) TQFN and 10-pin µMAX packages. Both packages
have exposed pads. -40°C to +85°C Operation.
Applications
● Distributed Supply Regulation
Typical Application Circuit
● Offline Power Supplies
● Telecom Hardware
● General-Purpose Point-of-Load
BOOTSTRAPPED 2.2MHz APPLICATION WITH LOW OPERATING VOLTAGE
22µF
0.47µH
BATTERY INPUT
2.5V to 40V
SW_OUT
8V/2A
47µF
CERAMIC
PVL
SUP
DRV
N
10kΩ
91kΩ
1kΩ
PGOOD
ISNS
22mΩ
PVL
MAX17292EUBA/B
2.2µF
FB
FSET/SYNC
COMP
13kΩ
12kΩ
EN
GND
Ordering Information appears at end of data sheet.
ENABLE
µMAX is a registered trademark of Maxim Integrated Products, Inc.
19-8544; Rev 0; 8/16
MAX17290/MAX17292
2.5V to 36V, 2.5MHz, PWM Boost Controller
with 4µA Shutdown Current and Reduced EMI
Absolute Maximum Ratings
EN, SUP, OVP, FB to GND....................................-0.3V to +42V
DRV, SYNCO, FSET/SYNC, COMP,
PGOOD, ISNS, REFIN to GND............ -0.3V to (V
Operating Temperature Range.......................... -40NC to +85NC
Maximum Junction Temperature.....................................+150NC
Storage Temperature Range............................ -65NC to +150NC
Lead Temperature (soldering, 10s) ................................+300NC
Soldering Temperature (reflow) ......................................+260NC
+ 0.3V)
PVL
PVL to GND............................................................... -0.3V to 6V
Continuous Power Dissipation (T = +70NC)
A
μMAX on SLB (derate 10.3mW/NC above +70NC) ......825mW
μMAX on MLB (derate 12.9mW/NC above +70NC)....1031mW
TQFN on SLB (derate 13.2mW/NC above +70NC).....1053mW
TQFN on MLB (derate 14.7mW/NC above +70NC)....1176mW
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation
of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum
rating conditions for extended periods may affect device reliability.
(Note 1)
Package Thermal Characteristics
μMAX (Single-Layer Board)
Junction-to-Ambient Thermal Resistance (B ) ..........97NC/W
TQFN (Single-Layer Board)
Junction-to-Ambient Thermal Resistance (B ) ..........76NC/W
JA
JA
Junction-to-Case Thermal Resistance (B ).................5NC/W
Junction-to-Case Thermal Resistance (B )...............11NC/W
JC
JC
μMAX (Four-Layer Board)
TQFN (Four-Layer Board)
Junction-to-Ambient Thermal Resistance (B ) ..........78NC/W
Junction-to-Ambient Thermal Resistance (B ) ..........68NC/W
JA
JA
Junction-to-Case Thermal Resistance (B ).................5NC/W
Junction-to-Case Thermal Resistance (B )...............11NC/W
JC
JC
Note 1: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-layer
board. For detailed information on package thermal considerations, refer to www.maximintegrated.com/thermal-tutorial.
Electrical Characteristics
(V
= 14V, T = T = -40NC to +85NC, unless otherwise noted. Typical values are at T =+25NC.) (Note 2)
SUP
A
J
A
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
POWER SUPPLY
SUP Operating Supply Range
V
4.5
36
1.3
2
V
SUP
MAX17290
MAX17292
0.75
1.25
4
V
= 1.1V, no
FB
SUP Supply Current in Operation
I
mA
CC
switching
SUP Supply Current in Shutdown
OVP Threshold Voltage
I
V
= 0V
7
FA
SHDN
EN
% of
V
OVP rising
105
-1
110
2.5
115
OVP
V
FB
OVP Threshold Voltage
Hysteresis
% of
V
OVPH
V
FB
OVP Input Current
I
+1
FA
OVP
PVL REGULATOR
PVL Output Voltage
PVL Undervoltage Lockout
V
4.7
3.8
5
4
5.45
4.3
V
V
PVL
V
SUP rising
UV
PVL Undervoltage-Lockout
Hysteresis
V
0.4
V
UVH
Maxim Integrated
│ 2
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MAX17290/MAX17292
2.5V to 36V, 2.5MHz, PWM Boost Controller
with 4µA Shutdown Current and Reduced EMI
Electrical Characteristics (continued)
(V
= 14V, T = T = -40NC to +85NC, unless otherwise noted. Typical values are at T =+25NC.) (Note 2)
SUP
A
J
A
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
OSCILLATOR
R
R
= 69kI
= 12kI
360
400
440
FSET
Switching Frequency
f
kHz
SW
2000
2200
2400
FSET
Spread-Spectrum Spreading
Factor
% of
SS
B, D, and F versions
Q6
f
SW
MAX17290
MAX17292
MAX17290
MAX17292
100
1000
220
1000
2500
1000
2500
When set with
resistor on pin
Switching Frequency Range
FSET/SYNC Frequency Range
f
kHz
kHz
SWR
Using external
SYNC signal
f
SYNC
1000
FSET Regulation Voltage
Soft-Start Time
V
12kI < R
< 69kI
0.9
9
V
FSET
FSET
t
Internally set
6
12
ms
ms
SS
Hiccup Period
t
55
HICCUP
MAX17290, R
MAX17292, R
= 69kI
93
85
50
FSET
Maximum Duty Cycle
DC
%
MAX
= 12kI
FSET
Minimum On-Time
t
80
110
ns
ON
THERMAL SHUTDOWN
Thermal-Shutdown Temperature
Thermal-Shutdown Hysteresis
GATE DRIVERS
T
Temperature rising
165
10
NC
NC
S
T
H
I
I
DRV Pullup Resistance
DRV Pulldown Resistance
R
I
I
= 100mA
= -100mA
3
1.4
0.75
1
5.5
2.5
DRVH
DRV
DRV
R
DRVL
Sourcing, C
= 10nF
DRV
DRV Output Peak Current
I
A
DRV
Sinking, C
= 10nF
DRV
REGULATION/CURRENT SENSE
V
V
V
= VPVL
= 2V
0.99
1.98
0.495
-0.5
1
2
1.01
2.02
0.505
+0.5
288
REFIN
REFIN
REFIN
Across full line, load,
and temperature
range
FB Regulation Voltage
V
V
FB
= 0.5V
0.5
FB Input Current
ISNS Threshold
I
FA
FB
212
250
60
40
8
mV
MAX16990
MAX16992
ISNS Leading-Edge Blanking
Time
t
ns
BLANK
Current-Sense Gain
A
V/V
VI
Peak Slope Compensation
Current-Ramp Magnitude
Added to ISNS input
40
50
60
FA
Rising
Falling
85
80
90
85
95
90
Percentage of final
value
PGOOD Threshold
V
%
PG
Maxim Integrated
│ 3
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MAX17290/MAX17292
2.5V to 36V, 2.5MHz, PWM Boost Controller
with 4µA Shutdown Current and Reduced EMI
Electrical Characteristics (continued)
(V
= 14V, T = T = -40NC to +85NC, unless otherwise noted. Typical values are at T =+25NC.) (Note 2)
SUP
A
J
A
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
ERROR AMPLIFIER
REFIN Input Voltage Range
0.5
2
V
V
REFIN Threshold for 1V FB
Regulation
V
-
V
PVL
0.4
-
V
-
PVL
0.8
PVL
0.1
Error-Amplifier gm
A
700
FS
MI
VEA
Error-Amplifier Output
Impedance
R
50
OEA
COMP Output Current
I
140
3
μA
V
COMP
COMP Clamp Voltage
2.7
3.3
LOGIC-LEVEL INPUTS/OUTPUTS
PGOOD/SYNCO Output Leakage
Current
V
/V
= 5V
0.5
FA
PGOOD SYNCO
PGOOD/SYNCO Output Low
Level
Sinking 1mA
EN rising
0.4
1.2
V
EN High Input Threshold
1.7
2.5
-1
V
V
EN Low Input Threshold
FSET/SYNC High Input Threshold
FSET/SYNC Low Input Threshold
EN and REFIN Input Current
V
1
V
+1
FA
Note 2: All devices 100% production tested at T = +25NC. Limits over temperature are guaranteed by design.
A
Maxim Integrated
│ 4
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MAX17290/MAX17292
2.5V to 36V, 2.5MHz, PWM Boost Controller
with 4µA Shutdown Current and Reduced EMI
Typical Operating Characteristics
(V
= 14V, T = +25NC, unless otherwise noted.)
SUP
A
SHUTDOWN SUPPLY CURRENT
vs. SUPPLY VOLTAGE
SHUTDOWN SUPPLY CURRENT
vs. TEMPERATURE
SUPPLY CURRENT vs. SUPPLY VOLTAGE
toc01
toc02
toc03
1.4
1.2
1.0
0.8
0.6
0.4
0.2
0
10
5.2
5.0
4.8
4.6
4.4
4.2
4.0
3.8
3.6
9
8
7
2.2MHz
400kHz
6
5
4
3
3
2
1
0
V
V
= V
SUP
= 1.1V
EN
FB
V
= 0V
V
EN
= 0V
EN
4
12
20
SUPPLY VOLTAGE (V)
28
36
4
12
20
28
36
-40 -20
0
20 40 60 80 100 120
TEMPERATURE (°C)
SUPPLY VOLTAGE (V)
MAX17290 INTERNAL OSCILLATOR
FREQUENCY vs. SUPPLY VOLTAGE
PVL VOLTAGE vs. SUPPLY VOLTAGE
PVL VOLTAGE vs. SUPPLY VOLTAGE
toc05
toc06
toc04
410
408
406
404
402
400
398
396
394
392
390
5.2
5.1
5.0
4.9
4.8
4.7
4.6
4.5
4.4
4.3
4.2
4.1
4.0
5.2
I
= 1mA
PVL
5.0
4.8
4.6
4.4
4.2
4.0
3.8
3.6
3.4
3.2
3.0
I
= 1mA
PVL
I
= 10mA
PVL
I
= 10mA
PVL
R
= 68.1kI
SET
28
4
12
20
36
4
12
20
SUPPLY VOLTAGE (V)
28
36
3
4
5
6
7
SUPPLY VOLTAGE (V)
SUPPLY VOLTAGE (V)
MAX17290 INTERNAL OSCILLATOR
FREQUENCY vs. TEMPERATURE
MAX17292 INTERNAL OSCILLATOR
FREQUENCY vs. SUPPLY VOLTAGE
MAX17292 INTERNAL OSCILLATOR
FREQUENCY vs. TEMPERATURE
toc08
toc09
toc07
420
415
410
405
400
395
390
385
380
2400
2350
2300
2250
2200
2150
2100
2050
2000
2200
2190
2180
2170
2160
2150
2140
2130
2120
2110
2100
R
= 68.1kI
R
= 12.1kI
R
= 12.1kI
SET
SET
SET
28
-40 -20
0
20 40 60 80 100 120
TEMPERATURE (°C)
4
12
20
36
-40 -20
0
20 40 60 80 100 120
TEMPERATURE (°C)
SUPPLY VOLTAGE (V)
Maxim Integrated
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MAX17290/MAX17292
2.5V to 36V, 2.5MHz, PWM Boost Controller
with 4µA Shutdown Current and Reduced EMI
Typical Operating Characteristics (continued)
(V
= 14V, T = +25NC, unless otherwise noted.)
SUP
A
POWER-UP RESPONSE
POWER-UP RESPONSE
toc10
toc11
5V/div
0V
5V/div
0V
V
V
V
SUP
OUT
SUP
5V/div
0V
5V/div
0V
V
V
OUT
DRV
5V/div
5V/div
0V
V
PVL
0V
5V/div
0V
5V/div
0V
V
V
PGOOD
PGOOD
2ms/div
2ms/div
STARTUP RESPONSE
STARTUP RESPONSE
toc12
toc13
5V/div
0V
5V/div
0V
V
V
V
SUP
OUT
PGOOD
5V/div
0V
5V/div
0V
5V/div
0V
V
V
OUT
DRV
V
PVL
5V/div
0V
5V/div
0V
5V/div
0V
V
EN
V
EN
2ms/div
2ms/div
STARTUP RESPONSE
(WITH SWITCHED OUTPUT)
OUTPUT LOAD TRANSIENT
toc15
toc14
5V/div
0V
5V/div
0V
V
SUP
V
PGOOD
5V/div
0V
5V/div
0V
5V/div
0V
V
V
OUT
OUT
V
OUT
500mV/div
(AC-COUPLED)
V
SW_OUT
5V/div
0V
V
EN
1A/div
0A
I
LOAD
50ms/div
2ms/div
Maxim Integrated
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MAX17290/MAX17292
2.5V to 36V, 2.5MHz, PWM Boost Controller
with 4µA Shutdown Current and Reduced EMI
Typical Operating Characteristics (continued)
(V
= 14V, T = +25NC, unless otherwise noted.)
SUP
A
LINE TRANSIENT
MAX17292 V vs. V
SYNC SYNCO
toc16
toc17
5V/div
0V
V
V
V
SUP
OUT
OUT
2V/div
0V
V
SYNC
5V/div
0V
500mV/div
(AC-COUPLED)
2V/div
0V
V
SYNCO
1A/div
0A
I
LOAD
20ms/div
200ns/div
SWITCHING WAVEFORM
OUTPUT VOLTAGE vs. REFIN VOLTAGE
toc19
toc18
30
25
20
15
10
5
5V/div
V
OUT
0V
5V/div
0V
V
IN
5V/div
0V
V
LX
1A/div
0A
I
LOAD
I
= 0A
OUT
0
500ns/div
0.5
1.0
1.5
2.0
2.5
3.0
REFIN VOLTAGE (V)
OVP SHUTDOWN
HICCUP MODE
toc20
toc21
V
OUT
5V/div
0V
V
V
OUT
DRV
5V/div
0V
1V/div
0V
V
V
OVP
DRV
5V/div
0V
5V/div
0V
V
PGOOD
5V/div
0V
5V/div
0V
V
PGOOD
1s/div
20ms/div
Maxim Integrated
│ 7
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MAX17290/MAX17292
2.5V to 36V, 2.5MHz, PWM Boost Controller
with 4µA Shutdown Current and Reduced EMI
Typical Operating Characteristics (continued)
(V
= 14V, T = +25NC, unless otherwise noted.)
SUP
A
MAX17292 INTERNAL OSCILLATOR
MAX17290 EFFICIENCY
MAX17292 EFFICIENCY
FREQUENCY vs. R
SET
toc23
toc22
toc24
100
100
2600
2400
2200
2000
1800
1600
1400
1200
1000
800
95
90
85
80
75
70
65
60
55
50
I
= 1A
95
90
85
80
75
70
65
60
55
50
OUT
I
= 2A
OUT
I
= 2A
OUT
I
= 1A
OUT
I
= 100mA
OUT
I
= 100mA
OUT
4
5
6
7
8
4
5
6
7
8
10
15
20
(kI)
25
30
SUPPLY VOLTAGE (V)
SUPPLY VOLTAGE (V)
R
SET
CURRENT-LIMIT THRESHOLD
vs. TEMPERATURE
MAX17292 MAXIMUM DUTY
CYCLE vs. TEMPERATURE
MAX17290 INTERNAL OSCILLATOR
FREQUENCY vs. RSET
toc27
toc25
toc28
260
258
256
254
252
250
248
246
244
242
240
1100
1000
900
800
700
600
500
400
300
200
100
0
91.0
90.5
90.0
89.5
89.0
88.5
88.0
87.5
87.0
R
= 12.1kI
SET
0
100
200
RSET(kΩ)
300
-40 -20
0
20 40 60 80 100 120
TEMPERATURE (°C)
-40 -20
0
20 40 60 80 100 120
TEMPERATURE (°C)
MAX17290 MAXIMUM DUTY
CYCLE vs. TEMPERATURE
INPUT VOLTAGE TRANSIENT
toc29
toc30
95.9
95.7
95.5
95.3
95.1
94.9
94.7
94.5
5V/div
V
IN
0V
5V/div
V
OUT
0V
1A/div
0A
5V/div
0V
I
LOAD
V
PGOOD
R
= 68.1kI
SET
-40 -20
0
20 40 60 80 100 120
TEMPERATURE (°C)
20ms/div
Maxim Integrated
│ 8
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MAX17290/MAX17292
2.5V to 36V, 2.5MHz, PWM Boost Controller
with 4µA Shutdown Current and Reduced EMI
Pin Configurations
TOP VIEW
TOP VIEW
TOP VIEW
9
8
7
9
8
7
+
SUP
EN
1
2
3
4
5
10
9
FB
FSET/SYNC 10
COMP 11
6
5
ISNS
PVL
FSET/SYNC 10
COMP 11
6
5
ISNS
PVL
MAX17290EUBA/B
MAX17292EUBA/B
COMP
FSET/SYNC
PGOOD
ISNS
MAX17290ETCC/D
MAX17292ETCC/D
MAX17290ETCE/F
MAX17292ETCE/F
GND
DRV
PVL
8
7
EP
FB 12
4
DRV
FB 12
4
DRV
EP
EP
6
+
+
1
2
3
1
2
3
µMAX
TQFN
(3mm x 3mm)
TQFN
(3mm x 3mm)
Pin Descriptions
MAX17290EUBA/B, MAX17290ETCC/D, MAX17290ETCE/F,
MAX17292EUBA/B MAX17292ETCC/D MAX17292ETCE/F
NAME
FUNCTION
μMAX-EP
TQFN-EP
TQFN-EP
Power-Supply Input. Place a bypass capacitor of at
least 1FF between this pin and ground.
1
1
1
SUP
Active-High Enable Input. This input is high-voltage
capable or can alternatively be driven from a logic-
level signal.
2
3
4
3
2
4
3
2
4
EN
GND
DRV
Ground Connection
Drive Output for Gate of nMOS Boost Switch. The
nominal voltage swing of this output is between PVL
and GND.
Output of 5V Internal Regulator. Connect a ceramic
capacitor of at least 2.2FF from this pin to ground,
placing it as close as possible to the pin.
5
6
5
6
5
6
PVL
Current-Sense Input to Regulator. Connect a sense
resistor between the source of the external switching
FET and GND. Then connect another resistor
between ISNS and the source of the FET for slope
compensation adjustment.
ISNS
Open-Drain Synchronization Output. SYNCO outputs
a square-wave signal which is 180N out-of-phase
—
—
7
SYNCO with the device’s operational clock. Connect a pullup
resistor from this pin to PVL or to a 5V or lower
supply when used.
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│ 9
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MAX17290/MAX17292
2.5V to 36V, 2.5MHz, PWM Boost Controller
with 4µA Shutdown Current and Reduced EMI
Pin Descriptions (continued)
MAX17290EUBA/B, MAX17290ETCC/D, MAX17290ETCE/F,
MAX17292EUBA/B MAX17292ETCC/D MAX17292ETCE/F
NAME
FUNCTION
μMAX-EP
TQFN-EP
TQFN-EP
Overvoltage Protection Input. When this pin goes
above 110% of the FB regulation voltage, all
switching is disabled. Operation resumes normally
when OVP drops below 107.5% of the FB regulation
point. Connect a resistor-divider between the output,
OVP, and GND to set the overvoltage protection
level.
—
7
—
OVP
Reference Input. When using the internal reference
connect REFIN to PVL. Otherwise, drive this pin with
an external voltage between 0.5V and 2V to set the
boost output voltage.
—
8
9
8
9
REFIN
Open-Drain Power-Good Output. Connect a resistor
from this pin to PVL or to another voltage less than or
equal to 5V. PGOOD goes high after soft-start when
the output exceeds 90% of its final value. When EN is
low PGOOD is also low. After soft-start is complete,
if PGOOD goes low and 16 consecutive current-limit
cycles occur, the devices enter hiccup mode and a
new soft-start is initiated after a delay of 44ms.
7
PGOOD
Frequency Set/Synchronization. To set a switching
frequency between 100kHz and 1000kHz
(MAX16990) or between 1000kHz and 2500kHz
(MAX16992), connect a resistor from this pin to GND.
To synchronize the converter, connect a logic signal
in the range 220kHz to 1000kHz (MAX16990) or
1000kHz to 2500kHz (MAX16992) to this input. The
external nMOSFET is turned on (i.e., DRV goes high)
after a short delay (60ns for 2.2MHz operation, 125ns
for 400kHz) when SYNC transitions low.
FSET/
SYNC
8
10
10
Output of Error Amplifier. Connect the compensation
network between COMP and GND.
9
11
12
11
12
COMP
FB
Boost Converter Feedback. This pin is regulated to
1V when REFIN is tied to PVL or otherwise regulated
to REFIN during boost operation. Connect a resistor-
divider between the boost output, the FB pin and
GND to set the boost output voltage. In a two-phase
converter connect the FB pin of the slave IC to PVL.
10
Exposed Pad. Internally connected to GND.
Connect to a large ground plane to maximize
thermal performance. Not intended as an electrical
connection point.
—
—
—
EP
Maxim Integrated
│ 10
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MAX17290/MAX17292
2.5V to 36V, 2.5MHz, PWM Boost Controller
with 4µA Shutdown Current and Reduced EMI
Functional Diagram
5V REGULATOR
PVL
SUP
+ REFERENCE
(OVP)
UVLO
REF.
EN
EN
DRV
THERMAL
THERMAL
50µA x f
SW
GND
250mV
BLANKING
TIME
ISNS
CONTROL
LOGIC
FSET/SYNC
(SYNCO)
OSCILLATOR
8
AGND
AGND
PGOOD
COMP
FB
PGOOD
COMPARATOR
OTA
V
- 0.4V
PVL
MAX17290
MAX17292
1V
(REFIN)
input. The input operating range can be as low as 2.5V
when the converter output supplies the SUP input.
Detailed Description
The MAX17290/MAX17292 are high-performance,
current-mode PWM controllers for wide input voltage
range boost converters. The input operating voltage
range of 4.5V to 36V makes these devices ideal in
battery operated harsh environment applications such as
for front-end “preboost” for the first boost stage in high-power
LED lighting applications. An internal low-dropout regulator
(PVL regulator) with an output voltage of 5V enables the
devices to operate directly from an automotive battery
The input undervoltage lockout (UVLO) circuit monitors
the PVL voltage and turns off the converter when the
voltage drops below 3.6V (typ). An external resistor
programs the switching frequency in two ranges from
100kHz to 1000kHz (MAX17290) or between 1000kHz
and 2500kHz (MAX17292). The FSET/SYNC input can
also be used for synchronization to an external clock. The
SYNC pulse width should be greater than 70ns.
Maxim Integrated
│ 11
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MAX17290/MAX17292
2.5V to 36V, 2.5MHz, PWM Boost Controller
with 4µA Shutdown Current and Reduced EMI
Inductor current information is obtained by means of an
external sense resistor connected from the source of the
external nMOSFET to GND.
Oscillator Frequency/External Synchronization/
Spread Spectrum
Use an external resistor at FSET/SYNC to program
the MAX17290 internal oscillator frequency from 100kHz
to 1MHz and the MAX17292 frequency between 1MHz
and 2.5MHz. See TOCs 24 and 25 in the Typical Operating
Characteristics section for resistor selection.
The devices include an internal transconductance error
amplifier with 1% accurate reference. At startup, the
internal reference is ramped in a time of 9ms to obtain
soft-start.
The SYNCO output is a 180N phase-shifted version of
the internal clock and can be used to synchronize other
converters in the system or to implement a two-phase
boost converter with a second MAX17290/MAX17292.
The advantages of a two-phase boost topology are
lower input and output ripple and simpler thermal
management as the power dissipation is spread over more
components. See the Multiphase Operation section for
further details.
The devices also include protection features such as
hiccup mode and thermal shutdown as well as an optional
overvoltage-detection circuit (OVP pin, C and D versions).
Current-Mode Control Loop
The MAX17290/MAX17292 offers peak current-mode
control operation for best load step performance and
simpler compensation. The inherent feed-forward
characteristic is useful especially in applications where
the input voltage changes quickly. While the current-mode
architecture offers many advantages, there are some
shortcomings. In high duty-cycle operation, subharmonic
oscillations can occur. To avoid this, the device offers
programmable slope compensation using a single resistor
between the ISNS pin and the current-sense resistor. To
avoid premature turn-off at the beginning of the on-cycle
the current-limit and PWM comparator inputs have
leading-edge blanking.
The devices can be synchronized using an external clock
at the FSET/SYNC input. A falling clock edge on FSET/
SYNC turns on the external MOSFET by driving DRV high
after a short delay.
The B, D, and F versions of the devices have spread-
spectrum oscillators. In these parts the internal
oscillator frequency is varied dynamically ±6% around
the switching frequency. Spread spectrum can improve
system EMI performance by reducing the height of peaks
due to the switching frequency and its harmonics in the
spectrum. The SYNCO output includes spread-spectrum
modulation when the internal oscillator is used on the B,
D, and F versions. Spread spectrum is not active when an
external clock is applied to the FSET/SYNC pin.
Startup Operation/UVLO/EN
The devices feature undervoltage lockout on the PVL-
regulator and turn on the converter once PVL rises
above 4V. The internal UVLO circuit has about 400mV
hysteresis to avoid chattering during turn-on. Once the
converter is operating and if SUP is fed from the output,
the converter input voltage can drop below 4.5V. This
feature allows operation at voltages as low as 2.5V or
even lower with careful selection of external components.
The EN input can be used to disable the device and
reduce the standby current to less than 4μA (typ).
nMOSFET Driver
DRV drives the gate of an external nMOSFET. The
driver is powered by the internal regulator (PVL), which
provides approximately 5V. This makes both the devices
suitable for use with logic-level MOSFETs. DRV can
source 750mA and sink 1000mA peak current. The
average current sourced by DRV depends on the
switching frequency and total gate charge of the external
MOSFET (see the Power Dissipation section).
Soft-Start
The devices are provided with an internal soft-start time
of 9ms. At startup, after voltage is applied and the UVLO
threshold is reached, the device enters soft-start. During
soft-start, the reference voltage ramps linearly to its final
value in 9ms.
Maxim Integrated
│ 12
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MAX17290/MAX17292
2.5V to 36V, 2.5MHz, PWM Boost Controller
with 4µA Shutdown Current and Reduced EMI
Error Amplifier
Applications Information
The devices include an internal transconductance error
amplifier. The noninverting input of the error amplifier is
connected to the internal 1V reference and feedback is
provided at the inverting input. High 700μS open-loop
transconductance and 50MΩ output impedance allow
good closed-loop bandwidth and transient response.
Moreover, the source and sink current capability of 140μA
provides fast error correction during output load transients.
Inductor Selection
Using the following equation, calculate the minimum
inductor value so that the converter remains in continuous
mode operation at minimum output current (I
):
OMIN
2
L
= (V
x D x E)/(2 x f
x V
x I
)
MIN
IN
SW
OUT
OMIN
where:
D = (V
+ V - V )/(V
+ V - V
)
OUT
D
IN
OUT
D
DS
Slope Compensation
and:
The devices use an internal current-ramp generator for
slope compensation. The internal ramp signal resets at
the beginning of each cycle and slews at a typical rate of
I
is between 10% and 25% of I
OUT
OMIN
A higher value of I
however, it increases the peak and RMS currents in the
switching MOSFET and inductor. Select I between
10% to 25% of the full load current. V is the forward
voltage drop of the external Schottky diode, D is the duty
reduces the required inductance;
OMIN
50μA x f . The amount of slope compensation needed
SW
OMIN
depends on the slope of the current ramp in the inductor.
See the Current-Sense Resistor Selection and Setting
Slope Compensation section for further information.
D
cycle, and V
is the voltage drop across the external
DS
Current Limit
switch. Select an inductor with low DC resistance and
with a saturation current (I ) rating higher than the peak
switch current limit of the converter.
The current-sense resistor (R ) connected between
SAT
CS
the source of the MOSFET and ground sets the current
limit. The ISNS input has a voltage trip level (V ) of
CS
Input and Output Capacitors
250mV. When the voltage produced by the current in the
inductor exceeds the current-limit comparator threshold, the
MOSFET driver (DRV) quickly terminates the on-cycle.
In some cases, a short time-constant RC filter could be
required to filter out the leading-edge spike on the sense
waveform in addition to the internal blanking time. The
amplitude and width of the leading edge spike depends
on the gate capacitance, drain capacitance, and switching
speed (MOSFET turn-on time).
The input current to a boost converter is almost
continuous and the RMS ripple current at the input capacitor
is low. Calculate the minimum input capacitor value and
maximum ESR using the following equations:
C
= DI x D/(4 x f
x DV )
IN
L
SW Q
ESR
= DV
/DI
MAX
ESR L
where DI = ((V - V ) x D)/(L x f ).
L
IN
DS
SW
V
is the total voltage drop across the external
DS
Hiccup Operation
MOSFET plus the voltage drop across the inductor
ESR. DI is peak-to-peak inductor ripple current as
calculated above. DV is the portion of input ripple due
to the capacitor discharge and DV
due to ESR of the capacitor. Assume the input capacitor
ripple contribution due to ESR (DV ) and capacitor
The devices incorporate a hiccup mode in an effort to
protect the external power components when there is
an output short-circuit. If PGOOD is low (i.e., the output
voltage is less than 85% of its set value) and there are
16 consecutive current-limit events, switching is stopped.
There is then a waiting period of 44ms before the device
tries to restart by initiating a soft-start. Note that a
short-circuit on the output places considerable stress on
all the power components even with hiccup mode, so that
careful component selection is important if this condition
is encountered. For more complete protection against
output short-circuits, a series pMOS switch driven from
PGOOD through a level-shifter can be employed.
L
Q
is the contribution
ESR
ESR
discharge (DV ) are equal when using a combination of
Q
ceramic and aluminium capacitors. During the converter
turn-on, a large current is drawn from the input source
especially at high output to input differential. The devices
have an internal soft-start, however, a larger input capacitor
than calculated above could be necessary to avoid
chattering due to finite hysteresis during turn-on.
Maxim Integrated
│ 13
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MAX17290/MAX17292
2.5V to 36V, 2.5MHz, PWM Boost Controller
with 4µA Shutdown Current and Reduced EMI
In a boost converter, the output capacitor supplies the
load current when the main switch is on. The required
output capacitance is high, especially at lower duty
cycles. Also, the output capacitor ESR needs to be low
enough to minimize the voltage drop due to the ESR while
supporting the load current. Use the following equations
to calculate the output capacitor, for a specified output
ripple. All ripple values are peak-to-peak.
The internal ramp signal resets at the beginning of each
cycle and slews at the rate of 50μA x f . Adjust the
SW
amount of slope compensation by choosing R
satisfy the following equation:
to
SCOMP
R
= (mc x R )/(50e-6 x f
)
SCOMP
CS
SW
In some applications a filter could be needed between the
current-sense resistor and the ISNS pin to augment the
internal blanking time. Set the RC time constant just long
enough to suppress the leading-edge spike of the MOSFET
current. For a given design, measure the leading spike
at the lowest input and rated output load to determine
the value of the RC filter which can be formed from the
slope-compensation resistor and an added capacitor from
ISNS to GND.
ESR = DV
/I
ESR OUT
C
= (I
x D
)/(DV x f
)
OUT
OUT
MAX
Q
SW
where I
is the output current, DV is the portion of the
Q
OUT
ripple due to the capacitor discharge, and DV
is the
ESR
ripple contribution due to the ESR of the capacitor. D
MAX
is the maximum duty cycle (i.e., the duty cycle at the
minimum input voltage). Use a combination of low-ESR
ceramic and high-value, low-cost aluminium capacitors
for lower output ripple and noise.
MOSFET Selection
The devices drive a wide variety of logic-level n-channel
power MOSFETs. The best performance is achieved with
low-threshold nMOSFETs that specify on-resistance with
Current-Sense Resistor Selection and Setting
Slope Compensation
Set the current-limit threshold 20% higher than the peak
switch current at the rated output power and minimum
input voltage. Use the following equation to calculate an
a gate-source voltage (V ) of 5V or less. When selecting
GS
the MOSFET, key parameters can include:
1) Total gate charge (Q ).
g
2) Reverse-transfer capacitance or charge (C
).
RSS
initial value for R
:
CS
3) On-resistance (R
).
DS(ON)
R
= 0.2/{1.2 x [((V
x I
) x (V
)/E)/V
+ 0.5 x
x L))]}
CS
((V
OUT
OUT
INMIN
4) Maximum drain-to-source voltage (V
).
DS(MAX)
– V
)/V
/(f
OUT
INMIN OUT
INMIN SW
5) Maximumgatefrequenciesthresholdvoltage(V
).
TH(MAX)
where E is the estimated efficiency of the converter (use
0.85 as an initial value or consult the graph in the Typical
Non-Synchronous Diode Selection
Operating Characteristics section); V
and I
are
OUT
OUT
The average diode current for a Boost converter is equal
to the output load current. The peak diode current depends
on how much ripple current is implemented in the design.
Therefore at minimum, choose a diode with average forward
current rating that is higher than the output current and
ensure the peak forward current rating is higher than the
output current plus one half the ripple current. As a rule
of thumb, choose I_AVG_DIODE at least equal to two
the output voltage and current, respectively; V
is the
INMIN
minimum value of the input voltage; f
is the switching
SW
frequency; and L is the minimum value of the chosen
inductor.
The devices use an internal ramp generator for slope
compensation to stabilize the current loop when
operating at duty cycles above 50%. The amount of slope
compensation required depends on the down-slope of
the inductor current when the main switch is off. The
inductor down-slope in turn depends on the input to output
voltage differential of the converter and the inductor value.
Theoretically, the compensation slope should be equal to
50% of the inductor downslope; however, a little higher
than 50% slope is advised. Use the following equation to
calculate the required compensating slope (mc) for the
boost converter:
times I
for minimum power loss and proper component
OUT
thermal dissipation. Once that is met the diode’s peak
specification will be more than enough.
I_AVG_DIODE = I
x 2
OUT
V_DIODE >> V
OUT
mc = 0.5 x (V
– V )/L A/s
IN
OUT
Maxim Integrated
│ 14
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MAX17290/MAX17292
2.5V to 36V, 2.5MHz, PWM Boost Controller
with 4µA Shutdown Current and Reduced EMI
At high switching frequencies, dynamic characteristics
(parameters 1 and 2 of the above list) that predict switching
In addition, the current-limit of the devices must be set
high enough so that the limit is not reached during the on-
time of the MOSFET which would result in output power
limitation and eventually entering hiccup mode. Estimate
the maximum input current using the following equation:
losses have more impact on efficiency than R
),
DS(ON
which predicts DC losses. Qg includes all capacitances
associated with charging the gate. The V of the
DS(MAX)
selected MOSFET must be greater than the maximum
output voltage setting plus a diode drop (or the maximum
input voltage if greater) plus an additional margin to allow
for spikes at the MOSFET drain due to the inductance in
the rectifier diode and output capacitor path. In addition,
Qg determines the current needed to drive the gate at the
selected operating frequency via the PVL linear regulator
and thus determines the power dissipation of the IC (see
the Power Dissipation section).
I
= ((V
x I
)/V
)/E)/V
+ 0.5 x
INMAX
OUT
OUT
OUT
INMIN
((V
– V
) x (V
/(f
x L))
INMIN OUT
INMIN SW
where I
is the maximum input current; V
and
INMAX
OUT
I
are the output voltage and current, respectively;
E is the estimated efficiency (which is lower at low input
voltages due to higher resistive losses); V is the
minimum value of the input voltage; f
frequency; and L is the minimum value of the chosen
OUT
INMIN
is the switching
SW
inductor.
Low-Voltage Operation
Boost Converter Compensation
Refer to Application Note 5587.
The devices operate down to a voltage of 4.5V or less on
their SUP pins. If the system input voltage is lower than
this the circuit can be operated from its own output as
shown in the Typical Application Circuit. At very low input
voltages it is important to remember that input current will
be high and the power components (inductor, MOSFET,
and diode) must be specified for this higher input current.
Overvoltage Protection
The “C” and “D” variants of the devices include the
overvoltage protection input. When the OVP pin goes
above 110% of the FB regulation voltage, all switching is
disabled. For an example application circuit, see Figure 2.
INPUT
V
OUT
INPUT
V
OUT
SUP
DRV
N
SUP
EN
DRV
N
ISNS
REFIN
PVL
ISNS
PVL
MAX17290ETCC/D
MAX17292ETCC/D
MAX17290
MAX17292EUBA
PVL
OVP
PGOOD
SYNCO
FB
FB
COMP
FSET/SYNC
FSET/SYNC
COMP
EN
ENABLE
GND
GND
Figure 2. Application with Independent Output Overvoltage
Protection
Figure 1. Standard Boost Application Circuit.
Maxim Integrated
│ 15
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MAX17290/MAX17292
2.5V to 36V, 2.5MHz, PWM Boost Controller
with 4µA Shutdown Current and Reduced EMI
1µF
10µH
VIN
50V/1A
2x47µF
CERAMIC
22µF
SUP
DRV
N
FSET/SYNC
2200Ω
69kΩ
ISNS
MAX17290ETCE/F
20mΩ
PGOOD
PVL
FB
2.2µF
COMP
EN
SYNCO GND
10kΩ
10µH
22µF
1µF
FSET/
SYNC
SUP
DRV
N
75kΩ
2200Ω
ISNS
REFIN
20mΩ
MAX17290ETCE/F
PGOOD
1500Ω
PVL
2.2µF
COMP
SYNCO
FB
EN
N
GND
ENABLE
Figure 3. Two-Phase 400kHz Boost Application with Minimum Component Count
multiphase converter it is important to protect the COMP
trace in the layout from noisy signals by placing it on an
inner layer and surrounding it with ground traces.
Multiphase Operation
Two boost phases can be implemented with no extra
components using two ICs as shown in Figure 3. In this
circuit the SYNCO output of the master device drives the
SYNC input of the slave forcing it to operate 180N out-
of-phase. The FB pin of the slave device is connected to
PVL, thus disabling its error amplifier. In this way the error
amplifier of the master controls both devices by means
of the COMP signal and good current-sharing is attained
between the two phases. When designing the PCB for a
Using REFIN to Adjust the Output Voltage
The REFIN pin can be used to directly adjust the
reference voltage of the boost converter, thus altering
the output voltage. When not used, REFIN should be
connected to PVL. Because REFIN is a high-impedance
pin, it is simple to drive it by means of an external digital-
to-analog converter (DAC) or a filtered PWM signal.
Maxim Integrated
│ 16
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MAX17290/MAX17292
2.5V to 36V, 2.5MHz, PWM Boost Controller
with 4µA Shutdown Current and Reduced EMI
where V
is the voltage at the SUP pin of the IC,
Power Dissipation
SUP
I
is the IC quiescent current consumption or typically
CC
The power dissipation of the IC comes from two sources:
the current consumption of the IC itself and the current
required to drive the external MOSFET, of which the latter
is usually dominant. The total power dissipation can be
estimated using the following equation:
0.75mA (MAX17290) or 1.25mA (MAX17292), Q is the
total gate charge of the chosen MOSFET at 5V, and f
is the switching frequency. P reaches it maximum at
maximum V
g
SW
IC
.
SUP
P
IC
= V
x I
+ (V
– 5) x (Q x f
)
SUP
CC
SUP
g
SW
Ordering Information
FREQUENCY
OVP/
SYNCO
SPREAD
SPECTRUM
PART
TEMP RANGE
PIN-PACKAGE
RANGE
MAX17290EUBA+
MAX17290EUBB+
MAX17290ETCC+
MAX17290ETCD+
MAX17290ETCE+
MAX17290ETCF+
MAX17292EUBA+
MAX17292EUBB+
MAX17292ETCC+
MAX17292ETCD+
MAX17292ETCE+
MAX17292ETCF+
220kHz to 1MHz
220kHz to 1MHz
220kHz to 1MHz
220kHz to 1MHz
220kHz to 1MHz
220kHz to 1MHz
1MHz to 2.5MHz
1MHz to 2.5MHz
1MHz to 2.5MHz
1MHz to 2.5MHz
1MHz to 2.5MHz
1MHz to 2.5MHz
None
None
Off
On
Off
On
Off
On
Off
On
Off
On
Off
On
-40NC to +85NC
-40NC to +85NC
-40NC to +85NC
-40NC to +85NC
-40NC to +85NC
-40NC to +85NC
-40NC to +85NC
-40NC to +85NC
-40NC to +85NC
-40NC to +85NC
-40NC to +85NC
-40NC to +85NC
10 FMAX-EP*
10 FMAX-EP*
12 TQFN-EP*
12 TQFN-EP*
12 TQFN-EP*
12 TQFN-EP*
10 FMAX-EP*
10 FMAX-EP*
12 TQFN-EP*
12 TQFN-EP*
12 TQFN-EP*
12 TQFN-EP*
OVP
OVP
SYNCO
SYNCO
None
None
OVP
OVP
SYNCO
SYNCO
+Denotes a lead(Pb)-free/RoHS-compliant package.
*EP = Exposed pad.
Chip Information
PROCESS: BiCMOS
Package Information
For the latest package outline information and land patterns (foot-
prints), go to www.maximintegrated.com/packages. Note that
a “+”, “#”, or “-” in the package code indicates RoHS status only.
Package drawings may show a different suffix character, but the
drawing pertains to the package regardless of RoHS status.
PACKAGE
TYPE
PACKAGE
CODE
OUTLINE
NO.
LAND
PATTERN NO.
21-0136
21-0109
90-0019
90-0148
12 TQFN-EP
T1233+4
U10E+3
10 μMAX-EP
Maxim Integrated
│ 17
www.maximintegrated.com
MAX17290/MAX17292
2.5V to 36V, 2.5MHz, PWM Boost Controller
with 4µA Shutdown Current and Reduced EMI
Revision History
REVISION REVISION
PAGES
DESCRIPTION
CHANGED
NUMBER
DATE
0
8/16
Initial release
—
For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim Integrated’s website at www.maximintegrated.com.
Maxim Integrated cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim Integrated product. No circuit patent licenses
are implied. Maxim Integrated reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and max limits)
shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.
©
Maxim Integrated and the Maxim Integrated logo are trademarks of Maxim Integrated Products, Inc.
2016 Maxim Integrated Products, Inc.
│ 18
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