MAX25202MATEAVY [MAXIM]
36V HV Synchronous Boost Controller for Automotive Infotainment Applications;
| 型号： | MAX25202MATEAVY |
| 厂家： | 
MAXIM INTEGRATED PRODUCTS |
| 描述： | 36V HV Synchronous Boost Controller for Automotive Infotainment Applications |
| 文件： | 总22页 (文件大小：1372K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
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MAX25201/MAX25202
36V HV Synchronous Boost Controller
for Automotive Infotainment Applications
General Description
Benefits and Features
● Meets Stringent OEM Module Power Consumption
The MAX2501/MAX25202 are high-performance, current-
mode PWM controllers with 1.5μA (typ) shutdown cur-
rent for wide input voltage range boost converters. The
4.5V to 36V input operating voltage range makes these
devices ideal in automotive applications, such as front-
end preboost or general-purpose boost power supply, for
the first boost stage in high-power LED lighting applica-
tions or to generate audio amplifier voltages. An internal
low-dropout regulator with a 5V output voltage enables
the MAX25201/MAX25202 to operate directly from an
automotive battery input. The input operating range can
be extended to as low as 1.8V after startup.
and Performance Specifications
• 20µA Quiescent Current in Skip Mode
• ±1.5% FB Voltage Accuracy
• Output Voltage Range: Fixed or Adjustable
Between 3.5V and 60V
● Enables Crank-Ready Designs
• Operates Down to 1.8V After Startup
• Wide Input Supply Range from 4.5V to 36V
● EMI Reduction Features Reduce Interference with
Sensitive Radio Bands Without Sacrificing Wide Input
Voltage Range
The MAX25201/MAX25202’s switching frequency opera-
tion (up to 2.2MHz) reduces output ripple, avoids AM band
interference, and allows for the use of smaller external
components. The switching frequency is resistor adjust-
able from 220kHz to 2.2MHz. Alternatively, the frequency
can be synchronized to an external clock. A spread-
spectrum option is available to improve system EMI per-
formance. For high-current applications the dual-phase
MAX25202 is available. The MAX25202 operates at a
fixed 400kHz switching frequency and can be synchro-
nized to an external clock.
• Spread-Spectrum Option
• Frequency-Synchronization Input
• Resistor-Programmable Frequency Between
200kHz and 2.2MHz
● Integration and Thermally Enhanced Packages Save
Board Space and Cost
• Current-Mode Controllers with Forced-Continuous
and Skip Modes
• Thermally Enhanced 16-Pin TQFN-EP Package
● Protection Features Improve System Reliability
• Supply Undervoltage Lockout
The controllers feature a power-OK monitor and under-
voltage lockout. Protection features include cycle-by-
cycle current limit and thermal shutdown. The MAX25201/
MAX25202 operate over the -40°C to +125°C automotive
temperature range.
• Overtemperature and Short-Circuit Protection
Applications
Infotainment Systems
Cluster Systems
E-Call
Ordering Information appears at end of data sheet.
19-100588; Rev 3; 2/20
MAX25201/MAX25202
36V HV Synchronous Boost Controller
for Automotive Infotainment Applications
Simplified Block Diagram
PGOOD
SS
COMP
EN
FB
THRES
SOFT START
EAMP
REF
BST
SUP
BIAS
EN
OUT
DH
LX
BIAS LDO
GATE
DRIVE
PWM
SUP
CS
CSA
PWM
ILIM
DL
ZX
LX
ILIM THRES
SLOPE COMP
LOGIC
GND
ZERO CROSS
FOSC
OSCILLATOR
SPS OTP
MODE/
FSYNC
(SKIP MODE )
(PWM MODE )
FSYNC
SELECT
LOGIC
Maxim Integrated
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MAX25201/MAX25202
36V HV Synchronous Boost Controller
for Automotive Infotainment Applications
Absolute Maximum Ratings
SUP, EN to GND .....................................................-0.3V to 42V
OUT, FB, LX to GND...............................................-0.3V to 65V
SUP to CS..............................................................-0.3V to 0.3V
BIAS, MODE/FSYNC, PGOOD, SS to GND.............-0.3V to 6V
DL, FOSC, COMP to GND........................ -0.3V to BIAS + 0.3V
BST to LX..................................................................-0.3V to 6V
DH to LX........................................................-0.3V to BST+0.3V
Continuous Power Dissipation
Operating Temperature Range......................... -40°C to +125°C
Junction Temperature......................................................+150°C
Storage Temperature Range............................ -65°C to +150°C
Soldering Temperature (reflow).......................................+260°C
Lead Temperature (soldering, 10s) .................................+300°C
TQFN (derate 28.8mW/°C* above +70°C) ................1666mW
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these
or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect
device reliability.
Recommended Operating Conditions
PARAMETER
SYMBOL
CONDITION
TYPICAL RANGE
UNIT
Ambient Temperature
Range
-40 to 125
°C
Note: These limits are not guaranteed.
Package Information
TQFN
Package Code
T1633Y+5C
Outline Number
21-100150
90-100064
Land Pattern Number
Thermal Resistance, Four-Layer Board:
Junction to Ambient (θ
)
44.5°C/W
5°C/W
JA
Junction to Case (θ
)
JC
For the latest package outline information and land patterns (footprints), go to www.maximintegrated.com/packages. Note that a “+”,
“#”, or “-” in the package code indicates RoHS status only. Package drawings may show a different suffix character, but the drawing
pertains to the package regardless of RoHS status.
Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-layer board.
For detailed information on package thermal considerations, refer to www.maximintegrated.com/thermal-tutorial.
Maxim Integrated
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MAX25201/MAX25202
36V HV Synchronous Boost Controller
for Automotive Infotainment Applications
Electrical Characteristics
(V
= 14V, V
= 14V, C
= 1μF, C
= 0.1μF, T = -40°C to +150°C, unless otherwise noted. Typical values are at T =
BST J A
SUP
EN
BIAS
+25°C.) (Note 2)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
STEP UP CONTROLLER
Initial startup, V
= V
4.5
1.8
36
36
OUT
BATT
Supply Voltage Range
V
V
Operation after initial startup condition is satisfied
SUP
Output Over-Voltage
Threshold
Detected with respect to V rising
102.0
105
25
107.5
%
FB
V
V
= V
, V = V
(fixed output voltage),
EN
SUP FB
BIAS
> V
, no load (MAX25201)
SUP
OUT
V
EN
= V
, V
> V
, adjustable output, no
SUP SUP
OUT
load. Excludes current through external FB divider
(MAX25201)
20
Supply Current
I
IN
µA
Shutdown, V
= 0V, fixed output voltage
1.5
1.5
3
3
EN
EN
Shutdown, V
current through external FB divider
= 0V, adjustable output, excludes
V
= V , PWM mode, MAX25201ATEA/VY+
FB
BIAS
9.85
10.04
10.04
10.25
and MAX25201ATEB/VY+ only
Fixed Output Voltage
V
V
V
OUT
V
= V , skip mode, MAX25201ATEA/VY+
FB
BIAS
9.70
10.30
36
and MAX25201ATEB/VY+ only
MAX25201ATEA/VY+ and MAX25201ATEB/VY+
3.5
Output Voltage
Adjustable Range
MAX25201ATEC/VY+, MAX25201ATED/
VY+, MAX25202MATEA/VY+,
MAX25202SATEA/VY+
20
60
Regulated Feedback
Voltage
V
0.99
1.005
0.01
0.01
250
1.02
1
V
FB
Feedback Leakage
Current
I
T
= 25°C
µA
FB
A
Feedback Line
Regulation Error
V
= 3.5V to 36V, V = 1V
%/V
µS
IN
FB
Transconductance
(from FB to COMP)
gm_boost
V
= 1V, V
= 5V (Note 1)
165
345
FB
BIAS
DL low to DH rising
DH low to DL rising
20
20
Dead Time
ns
DH Pullup Resistance
V
V
V
V
= 5V, I
= -100mA
1.5
2.6
2
Ω
Ω
BIAS
BIAS
BIAS
BIAS
DH
DH
DH Pulldown
Resistance
= 5V, I
= 100mA
1
1.5
1
DL Pullup Resistance
= 5V, I = -100mA
2.8
2
Ω
DL
DL Pulldown
Resistance
= 5V, I = 100mA
Ω
DL
Minimum Off Time
t
80
ns
MHz
OFFBST
PWM Switching
Frequency Range
f
MAX25201, programmable with R
0.22
2.2
SW
FOSC
R
= 70kΩ, V
= 5V, 3.3V (MAX25201)
380
400
400
420
425
Switching Frequency
Accuracy
FOSC
BIAS
kHz
MAX25202M/MAX25202S
375
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MAX25201/MAX25202
36V HV Synchronous Boost Controller
for Automotive Infotainment Applications
Electrical Characteristics (continued)
(V
= 14V, V
= 14V, C
= 1μF, C
= 0.1μF, T = -40°C to +150°C, unless otherwise noted. Typical values are at T =
SUP
EN
BIAS
BST J A
+25°C.) (Note 2)
PARAMETER
SYMBOL
CONDITIONS
MIN
40
TYP
50
MAX
60
UNITS
MAX25201
MAX25202M/S
CS Current-Limit
Voltage Threshold
V
5V, V
- V ; V
=
SUP
CS BIAS
V
mV
LIMIT
> 2.5V
36
48
60
SUP
Soft-Start Current
Source
I
V
= 5V
BIAS
8
10
12
µA
µA
SS
LX Leakage Current
V
= V
or V
, T = 25°C
0.001
94.5
92.5
5
LX
PGND
SUP
A
PGOOD_H % of V , rising
92.5
90.5
96.5
94.5
FB
PGOOD Threshold
%
PGOOD_F % of V , falling
FB
PGOOD Leakage
Current
V
= 5V, T = 25°C
A
1
µA
V
PGOOD
PGOOD Output Low
Voltage
I
= 1mA
0.2
PGOOD
PGOOD Debounce
Time
Fault detection, rising and falling
150
1.5
µs
PGOOD Timeout
Output in regulation to PGOOD high
ms
FSYNC INPUT
Minimum sync pulse of 100ns, f
OSC
= 2.2MHz
= 400kHz
1.8
250
1.4
2.6
MHz
kHz
FSYNC Frequency
Range
Minimum sync pulse of 100ns, f
High threshold
550
OSC
FSYNC Switching
Thresholds
V
Low threshold
0.4
INTERNAL LDO BIAS
Internal BIAS Voltage
V
V
V
> 6V
5
V
V
IN
rising
falling
3.1
2.6
3.25
BIAS
BIAS
BIAS UVLO Threshold
2.4
Minimum Current
Capability
V
= 5V
150
mA
BIAS
THERMAL OVERLOAD
Thermal Shutdown
Temperature
(Note 1)
(Note 1)
170
°C
°C
Thermal Shutdown
Hysteresis
20
EN LOGIC INPUT
High Threshold
1.8
V
V
Low Threshold
0.8
1
EN Input Bias Current
SPREAD SPECTRUM
EN logic inputs only, T = 25°C
0.01
µA
A
f
±
OSC
6%
Spread Spectrum
Note 1: Limits are 100% tested at +25°C. Limits over operating temperature range and relevant supply voltage are guaranteed by
design and characterization. Typical values are at +25°C.
Note 2: The device is designed for continuous operation up to T = +125°C for 95,000 hours and T = +150°C for 5,000 hours.
J
J
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MAX25201/MAX25202
36V HV Synchronous Boost Controller
for Automotive Infotainment Applications
Typical Operating Characteristics
(V
= 14V, T = 25°C, unless otherwise noted.)
SUP
A
OUTPUT VOLTAGE
vs. INPUT VOLTAGE
OUTPUT VOLTAGE
vs. INPUT VOLTAGE
MAX25201 EFFICIENCY
vs. LOAD CURRENT
toc02
toc03
toc01
8.16
8.12
8.08
8.04
8
100
95
90
85
80
75
70
65
60
55
50
24.5
7V INPUT
2.1MHz FPWM
8V OUTPUT
24.4
24.3
24.2
24.1
24
0A LOAD
5V INPUT
0A LOAD
4A LOAD
3V INPUT
CURRENT
LIMIT
2A LOAD
4A LOAD
23.9
23.8
23.7
23.6
23.5
7.96
7.92
7.88
7.84
8V OUT
400kHz FPWM
24V OUTPUT
2.1MHz FPWM
RCS = 3mΩ
1
3
0
0
4
5
6
7
8
0
0
6
2
3
4
5
6
4
8
12
16
20
24
INPUT VOLTAGE (V)
LOAD CURRENT (A)
INPUT VOLTAGE (V)
MAX25201 EFFICIENCY
vs. LOAD CURRENT
MAX25201 EFFICIENCY
vs. LOAD CURRENT
MAX25201 EFFICIENCY
vs. LOAD CURRENT
toc05
toc06
toc04
100
95
90
85
80
75
100
95
90
85
80
75
100
95
90
85
80
75
70
65
60
55
50
7V INPUT
21V INPUT
14V INPUT
14V INPUT
5V INPUT
21V INPUT
3V INPUT
4.5V INPUT
4.5V INPUT
CURRENT
LIMIT
CURRENT
LIMIT
8V OUT
2.1MHz
SKIP
24V OUT
400kHz
SKIP
24V OUT
RCS = 1.5mΩ
RCS = 1.5mΩ
400kHz FPWM
RCS = 3mΩ
1
1
2
3
4
5
6
7
8
1
2
3
4
5
6
7
8
0
2
3
4
5
6
LOAD CURRENT (A)
LOAD CURRENT (A)
LOAD CURRENT (A)
OUTPUT VOLTAGE
vs. LOAD CURRENT
QUIESCENT CURRENT
vs. SUPPLY VOLTAGE
OUTPUT VOLTAGE
vs. LOAD CURRENT
toc07
toc09
toc08
8.15
8.1
50
40
30
20
10
0
24.5
24.4
24.3
24.2
24.1
24
21V INPUT
7V INPUT
14V INPUT
5V INPUT
8.05
8
4.5V INPUT
23.9
23.8
23.7
23.6
23.5
3V INPUT
7.95
7.9
8V OUT
24V OUT
400kHz FPWM
2.1MHz FPWM
VFB = 1.15V
30
7.85
0
0.5
1
1.5
2
2.5
3
3.5
4
12
18
24
36
0.5
1
1.5
2
2.5
3
3.5
4
LOAD CURRENT (A)
SUPPLY VOLTAGE (V)
LOAD CURRENT (A)
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MAX25201/MAX25202
36V HV Synchronous Boost Controller
for Automotive Infotainment Applications
Typical Operating Characteristics (continued)
(V
= 14V, T = 25°C, unless otherwise noted.)
SUP
A
COLD-CRANK INPUT
VOLTAGE TRANSIENT
SHUTDOWN SUPPLY CURRENT
vs. SUPPLY VOLTAGE
toc11
toc10
4
VSUP
8V/div
3.5
3
0V
10V/div
2.5
2
VOUT
0V
1.5
1
5V/div
0V
VPGOOD
0.5
0
5V/div
VBIAS
0V
0
4
8
12 16 20 24 28 32 36
SUPPLY VOLTAGE (V)
50ms/div
INPUT UNDERVOLTAGE PULSE
SUPPLY VOLTAGE RAMP
toc12
toc13
10V/div
10V/div
VSUP
VSUP
0V
0V
10V/div
10V/div
VOUT
VOUT
0V
0V
5V/div
0V
5V/div
0V
VPGOOD
VPGOOD
5V/div
5V/div
0V
VBIAS
VBIAS
0V
5s/div
500ms/div
POWER-UP RESPONSE
POWER-UP RESPONSE
toc15
toc14
10V/div
10V/div
VSUP
VSUP
0V
0V
12V/div
12V/div
VOUT
VOUT
0V
0V
5V/div
0V
5V/div
0V
VPGOOD
VPGOOD
5V/div
5V/div
VBIAS
VDL
0V
0V
3ms/div
3ms/div
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MAX25201/MAX25202
36V HV Synchronous Boost Controller
for Automotive Infotainment Applications
Typical Operating Characteristics (continued)
(V
= 14V, T = 25°C, unless otherwise noted.)
SUP
A
STARTUP RESPONSE
STARTUP RESPONSE
toc16
toc17
10V/div
10V/div
VSUP
VSUP
0V
0V
12V/div
12V/div
VOUT
VOUT
0V
0V
5V/div
0V
5V/div
0V
VPGOOD
VBIAS
5V/div
5V/div
VEN
VEN
0V
0V
3ms/div
3ms/div
SWITCHING WAVEFORM
LOAD TRANSIENT RESPONSE
toc19
toc18
10V/div
10V/div
VSUP
VOUT
VSUP
0V
500mV/div
(AC)
0V
20V/div
VOUT
0V
12V/div
VOUT
14V/div
0V
VLX
0V
2A/div
ILOAD
4A/div
ILOAD
0A
0A
5µs/div
1ms/div
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MAX25201/MAX25202
36V HV Synchronous Boost Controller
for Automotive Infotainment Applications
Pin Configurations
MAX25201
MAX25202M
DUAL PHASE MASTER
TOP VIEW
TOP VIEW
16
15
14
13
16
15
14
13
CS
SUP
OUT
FB
+
1
2
3
4
12
11
10
9
DL
+
CS
SUP
OUT
FB
1
2
3
4
12
11
10
9
DL
GND
BIAS
FOSC
GND
MAX25201
MAX25202M
BIAS
FSYNCOUT
5
6
7
8
5
6
7
8
SW TQFN
3mm x 3mm
SW TQFN
3mm x 3mm
MAX25202S
DUAL PHASE SLAVE
TOP VIEW
16
15
14
13
+
CS
1
2
3
4
12
11
10
9
DL
SUP
OUT
FB
GND
BIAS
NC
MAX25202S
5
6
7
8
SW TQFN
3mm x 3mm
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MAX25201/MAX25202
36V HV Synchronous Boost Controller
for Automotive Infotainment Applications
Pin Description
PIN
NAME
FUNCTION
MAX25201 MAX25202M MAX25202S
Negative Current-Sense Input for Boost Controller. Connect CS to the
negative side of the current-sense element. See the Current Limiting
and Current-Sense Inputs (SUP and CS) and Current-Sense Resistor
Selection sections.
1
1
1
CS
Supply Input and Positive Current-Sense Input for Boost Controller. Con-
nect SUP to the positive terminal of the current-sense element. See the
Current Limiting and Current-Sense Inputs (SUP and CS) and Current
Sense Measurement sections.
2
3
2
3
2
3
SUP
OUT
Input for the BIAS LDO. Connect OUT to the boost output when the
output voltage is set at 24V or below. For V
greater than 24V,
OUT
connect OUT to the input supply.
Boost Converter Feedback Input. To set the output voltage between
3.5V and 60V, connect FB to the center tap of a resistive divider be-
tween the boost regulator output. FB regulates to 1V (typ). To use the
factory set fixed output voltage on applicable parts (see the Ordering
Information section, connect FB to BIAS and connect OUT to the output.
For more information, see the Setting the Output Voltage section.
4
5
4
5
4
5
FB
Boost Controller Error Amplifier Output. Connect a RC network to COMP
to compensate boost converter.
COMP
Programmable Soft-Start. Connect a capacitor from SS to GND to set
the soft-start time. To select the value, see the Typical Operating Char-
acteristics section.
6
6
—
6
SS
—
—
MODE
Connect to FSYNCIN of the MAX25202M.
Open-Drain Power-Good Output for Buck Controller One. PGOOD goes
low if OUT drops below 92.5% (typ falling) of the normal regulation
point. PGOOD asserts low during soft-start and in shutdown. PGOOD
becomes high impedance when OUT is in regulation. To obtain a logic
signal, pull up PGOOD with an external resistor connected to a positive
voltage lower than 5.5V.
7
7
—
PGOOD
NC
—
8
—
—
7, 9
Do Not Connect
External Clock Synchronization Input. To use the internal oscillator con-
nect MODE/FSYNC high for forced-PWM or low for skip-mode opera-
tion. To synchronize with an external clock, connect the clock to MODE/
FSYNC. See the Light-Load Efficiency Skip Mode and Forced-PWM
Mode sections.
MODE/
FSYNC
—
Synchronization Input. Connect to an external clock for synchronization.
Connect to ground for internal frequency setting. When an external
signal is connected, the spread spectrum is disabled.
—
8
—
FSYNCIN
Slave Input Synchronization. For dual-phase operation, connect FSYN-
CINS of the MAX25202S to FSYNCOUT of the MAX25202M.
—
9
—
—
8
FSYNCINS
FOSC
Frequency Setting Input. Connect a resistor to FOSC to set the
switching frequency of the DC-DC converters.
—
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MAX25201/MAX25202
36V HV Synchronous Boost Controller
for Automotive Infotainment Applications
Pin Description (continued)
PIN
NAME
FUNCTION
MAX25201 MAX25202M MAX25202S
Clock Synchronization Output. Connect FSYNCOUT to FSYNCINS of
the MAX25202S.
—
9
—
FSYNCOUT
5V Internal Linear Regulator Output. Bypass BIAS to GND with a low-
ESR ceramic capacitor of 1µF minimum value. BIAS provides the power
to the internal circuitry and external loads. See the Fixed 5V Linear
Regulator (BIAS) section.
10
10
10
BIAS
11
12
11
12
11
12
GND
Ground
DL
Low-Side N-Channel MOSFET Gate Driver Output
Inductor Connection for Boost Controller. Connect LX to the switched
side of the inductor. LX serves as the lower supply rail for the DH high-
side gate driver.
13
14
13
14
13
14
LX
High-Side MOSFET Gate Driver Output for Boost Controller. DH output
DH
voltage swings from V to V
.
LX
BST
Boost Flying Capacitor Connection for High-Side Gate Voltage of Boost
Controller. Connect a high-voltage diode between BIAS and BST. Connect
a ceramic capacitor between BST and LX. See the High-Side Gate-
Driver Supply (BST) section.
15
16
15
16
15
16
BST
EN
High-Voltage Tolerant, Active-High Digital Enable Input for Controller
Exposed Pad. Connect the exposed pad to ground. Connecting the
exposed pad to ground does not remove the requirement for proper
ground connections to GND. The exposed pad is attached with epoxy to
the substrate of the die, making it an excellent path to remove heat from
the IC.
—
—
—
EP
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MAX25201/MAX25202
36V HV Synchronous Boost Controller
for Automotive Infotainment Applications
The OUT pin is the input to the linear regulator. OUT is
typically connected to the boost output for applications
with the output voltage set to 24V or less and applica-
tions that require operation with a supply voltage below
5.2V. To reduce power dissipation in applications with
higher output voltages, OUT should be connected to
SUP. Bypass OUT with a 1µF or greater ceramic capaci-
tor to GND.
Detailed Description
The MAX25201/MAX25202 automotive controller main-
tains regulation during cold crank or start-stop operations
down to a battery input of 1.8V, and operates with only
20μA I . The devices generate backlight voltages, audio
Q
amplifier voltages, stand-alone preboost, as well as a
standby voltage in telematics applications. The devices
can start up with an input voltage supply from 3.5V to 42V
and can operate down to 1.8V after startup.
Startup Operation/UVLO/EN
The MAX25201/MAX25202’s 2.2MHz switching frequency
reduces output ripple, avoids AM band interference, and
allows for the use of smaller external components. The
switching frequency is resistor adjustable from 220kHz to
2.2MHz. Alternatively, the frequency can be synchronized
to an external clock. A spread-spectrum option is avail-
able to improve system EMI performance.
The BIAS undervoltage lockout (UVLO) circuitry inhibits
switching if the 5V bias supply (BIAS) is below its 2.6V
(typ) UVLO falling threshold. Once BIAS rises above its
UVLO rising threshold and EN is high, the boost controller
starts switching and the output voltage begins to ramp up
using soft-start. Driving EN low disables the device and
reduces the standby current to less than 10μA.
These controllers feature a power-OK monitor as well as
overvoltage and undervoltage lockout. Protection features
include cycle-by-cycle current limit and thermal shutdown.
The MAX25201/MAX25202 are specified for operation
over the -40°C to +125°C automotive temperature range.
Soft-Start
Soft-start ramps up the internal reference during startup
to reduce input surge current. Connect a capacitor from
SS to GND to set the soft-start time. Select the capacitor
value as follows:
Current-Mode Control Loop
C
[nF] = 10 × t [ms]
ss
SS
Peak current-mode control operation provides excellent
load step performance and simple compensation. The
inherent feed-forward characteristic is useful especially in
automotive applications where the input voltage changes
quickly during cold-crank and load dump conditions.
To avoid premature turn-off at the beginning of the on
cycle the current-limit and PWM comparator inputs have
leading-edge blanking.
Soft-start begins when EN is logic-high and V
above the undervoltage lockout threshold.
is
BIAS
Oscillator Frequency/External
Synchronization
The MAX25201's internal oscillator is set by a resistor
connected from FOSC to GND with an adjustment range
of 220kHz to 2.2MHz. High-frequency operation optimizes
the application for the smallest component size, trading
off efficiency to higher switching losses. Low-frequency
operation offers the best overall efficiency at the expense
of component size and board space.
Fixed 5V Linear Regulator (BIAS)
An internal 5V linear regulator (BIAS) is used to power
the controller's internal circuitry. Connect a 1μF or greater
ceramic capacitor from BIAS to GND as close as possible
to the IC pins to guarantee stability under the full-load
condition. The internal linear regulator can provide up to
150mA (typ) total. The internal bias current requirements
can be estimated as follows:
R
FOSC
24500 +
√
0.006
F
=
SW
R
FOSC
I
= I
+ f
(QG_DL + QG_DH)
BIAS
CC
SW
where:
The MAX25202's internal oscillator is fixed at 400kHz.
I
= the internal supply current
= the switching frequency
The devices can also be synchronized to an external
clock by connecting the external clock signal to MODE/
FSYNC (MAX25201) or FSYNCIN (MAX25202M). The
internal oscillator is synchronized on the rising edge of the
external clock. See the Electrical Characteristics table for
the FSYNC frequency range and voltage levels.
CC
f
SW
QG_ = the low- and high-side MOSFET total gate charge
(specification limits at V = 5V).
GS
To reduce the internal power dissipation, BIAS can option-
ally be connected to an external 5V rail, bypassing the
internal linear regulator.
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MAX25201/MAX25202
36V HV Synchronous Boost Controller
for Automotive Infotainment Applications
voltage of the external MOSFETs. A low-resistance, low-
inductance path from DL and DH to the MOSFET gates
is required in order for the protection circuits to work
properly.
Light-Load Efficiency Skip Mode
The skip mode feature of the MAX25201/MAX25202 is
used to improve light-load efficiency. Drive MODE/FSYNC
low to enable skip mode.
In skip mode, once the output reaches regulation, the
MAX25201/MAX25202 stop switching until the FB voltage
drops below the reference voltage. Once the FB voltage
has dropped below the reference voltage, the devices
resume switching until the inductor current reaches 30%
(skip threshold) of the maximum current set by the induc-
tor DCR or current-sense resistor.
High-Side Gate-Driver Supply (BST)
The high-side MOSFET is turned on by closing an inter-
nal switch between BST and DH and transferring the
bootstrap capacitor’s (at BST) charge to the gate of the
high-side MOSFET. This charge refreshes when the high-
side MOSFET turns off and the LX voltage drops down
to ground potential, taking the negative terminal of the
capacitor to the same potential. The bootstrap diode then
recharges the positive terminal of the bootstrap capacitor.
Forced-PWM Mode
Drive MODE/FSYNC of the MAX25201/MAX25202 high
(connect to BIAS) for forced-PWM operation. This pre-
vents the devices from entering skip mode by disabling
the zero-crossing detection of the inductor current, and
forces the low-side gate-drive waveform to the comple-
ment of the high-side gate-drive waveform. Under light-
load the inductor current reverses, discharging the output
capacitor. The benefit of forced-PWM mode is that it
keeps the switching frequency constant under all load
conditions. This reduces ripple and makes it predict-
able and easier to filter. Forced-PWM mode is useful
for improving load-transient response and eliminating
unknown frequency harmonics that can interfere with AM
radio bands. The disadvantage with forced-PWM opera-
tion is that it reduces light-load efficiency.
The selected n-channel high-side MOSFET determines
the appropriate boost capacitance values according to the
following equation:
C
BST
= Q /∆V
G BST
where:
= the total gate charge of the high-side MOSFET
Q
G
∆V
= the voltage variation allowed on the high-
BST
side MOSFET driver after turn-on. Choose ∆V
such
BST
that the available gate-drive voltage is not significantly
degraded (e.g., ∆V = 100mV to 300mV) when deter-
BST
mining C
. The boost capacitor should be a low-ESR
BST
ceramic capacitor. A minimum value of 0.1μF works well
in most cases.
Forced-PWM is always used when synchronizing to an
Current Limiting and Current-Sense Inputs
(SUP and CS)
The current-limit circuit uses differential current-sense
inputs (SUP and CS) to limit the peak inductor current.
If the magnitude of the current-sense signal exceeds the
external clock and in multiphase applications.
Spread Spectrum
Spread spectrum reduces peak emission noise at the
clock frequency and its harmonics, making it easier to
meet stringent EMI limits. This is done by dithering the
switching frequency ±6%. Using an external clock source
(i.e. driving the MODE/FSYNC input with an external
clock) disables spread spectrum.
current-limit threshold (V
> 50mV (typ)), the PWM
LIMIT
controller turns off the high-side MOSFET.
For the most accurate current sensing, use a current-
sense resistor between the inductor and the input capaci-
Spread spectrum is a factory set option. See the Ordering
Information section to determine which part numbers
have spread spectrum enabled.
tor. Connect CS to the inductor side of R
and SUP
CS
to the capacitor side. See the Current-Sense Resistor
Selection section to determine the resistor value.
MOSFET Drivers (DH and DL)
To improve efficiency, the current can also be measured
directly across the inductor, eliminating the power loss
from the sense resistor. However, this method is sig-
nificantly less accurate and requires a filter network in
the current-sense circuit. See the Inductor DCR Current
Sense section for more information.
The DH high-side n-channel MOSFET driver is pow-
ered from BST. The low-side driver (DL) is powered
from BIAS. To prevent a MOSFET from turning on before
a complementary switch is fully off, each driver has
shoot-through protection that monitors the gate-to-source
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MAX25201/MAX25202
36V HV Synchronous Boost Controller
for Automotive Infotainment Applications
Voltage Monitoring (PGOOD)
V
OUT
PGOOD is the open-drain output of the output voltage
monitor. PGOOD is high impedance when the output
voltage is in regulation. PGOOD pulls low when the out-
put voltage drops below the PGOOD threshold. See the
Electrical Characteristics table. Typically, a pullup resistor
is connected from PGOOD to the relevant logic rail to
provide a logic-level output. PGOOD asserts low during
soft-start and when disabled (EN is low).
R1
=
R2
− 1
V
FB
where R1 is the resistor connected from FB to the output,
R2 is the resistor connected from FB to ground, V
the desired output voltage, and V is the regulated feed-
is
OUT
FB
back voltage (1.005V typ).
Parts with a fixed output voltage option (see the Ordering
Information section) can also be used without the external
FB divider. To use the preset output voltage, connect FB
to BIAS, and connect OUT to the regulator output.
Protection Features
Overcurrent Protection
Inductor Selection
If the inductor current exceeds the maximum current
limit set by R
or inductor DCR sensing, the respective
CS
Duty cycle and frequency are important when calculat-
ing the inductor size because the inductor current ramps
up during the on-time of the switch and ramps down
during its off-time. A higher switching frequency gener-
ally improves transient response and reduces component
size; however, if the boost components are used as the
input filter components during non-boost operation, a low
frequency is advantageous.
MOSFET driver turns off. Increasing the output current
further results in shorter and shorter high-side pulses. A
hard short results in a minimum on-time pulse every clock
cycle. When required, choose components that can with-
stand the short-circuit current.
Thermal Overload Protection
Thermal-overload protection limits total power dissipa-
tion in the MAX25201/MAX25202. When the junction
temperature exceeds +170°C (typ), an internal thermal
sensor shuts the devices off, allowing them to cool down.
The thermal sensor turns the devices on again after the
junction temperature cools by 20°C (typ).
The duty-cycle range of the boost converter depends on
the effective input-to-output voltage ratio. In the following
calculations, the duty cycle refers to the on-time of the
boost MOSFET:
V
− V
OUT(MAX)
V
SUP(MIN)
D
=
MAX
Overvoltage Protection
OUT(MAX)
The devices limit the output voltage by turning off the
high-side gate driver if the output voltage exceeds 105%
(typ) of the nominal output voltage. The output volt-
age must come back into regulation before the devices
resume switching.
or including losses in the inductor and high-side MOSFET
(VON,FET):
V
− V
+ I
(
× (R
OUT DC
+ R
HSRDSON
)
)
OUT(MAX)
SUP(MIN)
V
D
=
MAX
OUT(MAX)
Slope Compensation
The devices use an internal current-ramp generator for
slope compensation. The slope compensation for the
MAX25201A and MAX25201B is optimized for operation
with output voltage set to 36V or lower. The MAX25201C,
MAX25201D, and MAX25202 are optimized for output
voltages between 20V and 60V.
The ratio of the inductor peak-to-peak AC current to DC
average current (LIR) must be selected first. A good initial
value is a 30% peak-to-peak ripple current to average
current ratio (LIR = 0.3). The switching frequency, input
voltage, output voltage, and selected LIR determine the
inductor value as follows:
V
× D
SUP
MHz × LIR
Applications Information
L μH =
[
]
f
[
]
SW
Setting the Output Voltage
where:
All versions of the MAX25201/MAX25202 support an
adjustable output voltage. See the Ordering Information
section for the adjustable output voltage range. To set the
output voltage, connect FB to the center a resistor divider
from the output to ground. Calculate the resistor values as
follows:
D = (V
-V
)/V
OUT SUP OUT
V
V
= Typical input voltage
SUP
OUT
= Typical output voltage
/(1-D)
LIR = 0.3 x I
OUT
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MAX25201/MAX25202
36V HV Synchronous Boost Controller
for Automotive Infotainment Applications
Select the inductor with a saturation current rating higher
than the peak switch current limit of the converter:
enough to minimize the voltage drop while supporting the
load current. Use the following equations to calculate the
output capacitor for a specified output ripple. All ripple
values are peak-to-peak:
∆ I
L_RIP_MAX
I
> I
+
L_PEAK
L_MAX
2
Running a boost converter in continuous-conduction
mode introduces a right-half plane zero into the transfer
function. To avoid the effect of this right-half plane zero,
the crossover frequency for the control loop should be ≤
∆ V
ESR
ESR =
I
OUT
I
× D
MAX
1/3 x f
. If faster bandwith is required, a smaller
OUT
RHP_ZERO
C =
∆ V × f
inductor and higher switching frequency is recommended.
Q
SW
Input Capacitor Selection
I
is the load current in A, f
is in MHz, C
is in
OUT
OUT
SW
The input current for the boost converter is continuous
and the RMS ripple current at the input capacitor is low.
Calculate the minimum input capacitor value and the
maximum ESR using the following equations:
μF, ∆V is the portion of the ripple due to the capacitor
discharge, and ∆V
of the capacitor. D
minimum input voltage. Use a combination of low-ESR
ceramic and high-value, low-cost aluminum capacitors for
lower output ripple and noise.
Q
is the contribution due to the ESR
is the maximum duty cycle at the
ESR
MAX
∆ I × D
L
C
=
4 × f
SUP
× ∆ V
SW
Q
Current-Sense Resistor Selection
∆ V
ESR
The current-sense resistor (R ), connected between the
CS
ESR =
∆ I
L
battery and the inductor, sets the current limit. The CS
input has a voltage trip level (V ) of 50mV (typ).
CS
where:
Set the current-limit threshold high enough to accommo-
date the component variations. Use the following equa-
V
(
− V
× D
)
SUP
L × f
DS
∆ I =
tion to calculate the value of R
:
L
CS
SW
V
CS
V
is the total voltage drop across the external MOSFET
DS
R
=
I
CS
SUP(MAX)
plus the voltage drop across the inductor ESR. ∆I is the
L
peak-to-peak inductor ripple current as calculated above.
where I
is the peak current that flows through the
IN(MAX)
∆V is the portion of input ripple due to the capacitor
Q
MOSFET at full load and minimum V .
IN
discharge and ∆V
is the contribution due to ESR of
ESR
the capacitor. Assume the input capacitor ripple contribu-
I
LOAD(MAX)
tion due to ESR (∆V ) and capacitor discharge (∆V )
ESR
Q
I
=
SUP(MAX)
1 − D
MAX
are equal when using a combination of ceramic and alu-
minum capacitors. During the converter turn-on, a large
current is drawn from the input source, especially at high
output-to-input differential.
When the voltage produced by this current (through the
current-sense resistor) exceeds the current-limit com-
parator threshold, the MOSFET driver (DL) quickly termi-
nates the on-cycle.
Output Capacitor Selection
In a boost converter, the output capacitor supplies the
load current when the boost MOSFET is on. The required
output capacitance is high, especially at higher duty
cycles. Also, the output capacitor ESR needs to be low
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MAX25201/MAX25202
36V HV Synchronous Boost Controller
for Automotive Infotainment Applications
L
R
CS
BATTERY
CS
SUP
CURRENT SENSE RESISTOR
L
R
DC
BATTERY
R2
R1
C
EQ
CS
SUP
INDUCTOR DCR CURRENT SENSE
Figure 1. Current-Sense Configurations
Inductor DCR Current Sense
Boost Converter Compensation
High-power applications that do not require accurate cur-
rent sense can use the inductor's DC resistance as the
current sense element instead of the current-sense resis-
tor. This is done by connecting an RC network across the
inductor. The equivalent sense resistance of the network
is:
The basic regulator loop is modeled as a power modula-
tor, output feedback-divider, and an error amplifier, as
shown in the Synchronous Boost Application Circuit. The
power modulator has a DC gain set by gmc x R
, with
LOAD
a pole and zero pair set by R
, the output capacitor
LOAD
(C ), and its ESR. The loop response is set by the fol-
OUT
lowing equations:
R2
R1 + R2
R
=
× R
f
CS_EQ
DC
(
)
1 + j
f
1 − D
2
zMOD
f
G
= g
× R ×
LOAD
×
MOD
MC
(
)
1 + j
where R
is the DC resistance of the inductor, R1 is
connected from the switch side of the inductor to CS, and
R2 is connected from the battery side of the inductor to
CS. The capacitor C
culated as follows:
f
DC
pMOD
f
× 1 − j
(connected parallel to R2) is cal-
f
EQ
Rph_zMOD
where R
= V
/I
in Ω and g
is the voltage gain of the
current-sense amplifier and is typically 12V/V. R
DC resistance of the inductor or the current-sense resis-
=1/
mc
LOAD
OUT LOUT(MAX)
L
1
1
(A
V_CS
x R ) in S. A
DC
V_CS
C
=
+
R1 R2
EQ
(
)
R
DC
is the
DC
tor in Ω.
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MAX25201/MAX25202
36V HV Synchronous Boost Controller
for Automotive Infotainment Applications
In a current-mode step-down converter, the output capaci-
tor and the load resistance introduce a pole at the follow-
ing frequency:
The loop gain crossover frequency (f ) should be ≤ 1/3 of
right-half plane zero frequency.
C
f
Rph_zMOD
f
≤
C
3
1
f
=
pMOD
π × R
× C
OUT
At the crossover frequency, the total loop gain must be
equal to 1. So:
LOAD
The output capacitor and its ESR also introduce a zero at:
V
FB
GAIN
×
× GAIN
= 1
)
MOD(f
)
EA(f
V
C
C
1
OUT
f
=
zMOD
2π × ESR × C
OUT
GAIN
= g
× R
EA(f
)
m, EA
C
C
The right-half plane zero is at:
f
pMOD
R
LOAD
GAIN
= GAIN
×
MOD(dc)
MOD(f
)
f
=
× 1 − D × 1 − D
)
f
(
)
(
C
Rph_zMOD
C
2π × L
Therefore:
GAIN
When C
parallel, the resulting C
= ESR(EACH)/n. Note that the capacitor zero for a paral-
lel combination of similar capacitors is the same as for an
individual capacitor.
is composed of “n” identical capacitors in
OUT
= n x C
, and ESR
OUT
OUT(EACH)
V
FB
×
× g
× R = 1
MOD(f
)
m, EA C
V
C
OUT
Solving for R :
C
The feedback voltage-divider has a gain of GAIN
=
FB
V
V
/V
, where V is 1.0V (typ).
FB OUT FB
OUT
× GAIN
R
=
C
g
× V
m, EA
FB
MOD(f )
The transconductance error amplifier has a DC gain of
GAIN = gm x R , where gm is the
C
EA(DC)
,EA
OUT,EA
,EA
Set the error-amplifier compensation zero formed by R
error-amplifier transconductance, which is 345μS (max),
C
and C at the f
lows:
. Calculate the value of C as fol-
and R is the output resistance of the error amplifi-
er, which is 10MΩ (typ). See the Electrical Characteristics
C
pMOD
C
OUT,EA
table.
1
C
C
=
2π × f
Adominant pole (f
itor (CC) and the amplifier output resistance (R
) is set by the compensation capac-
× R
dpEA
pMOD
C
). A
OUT,EA
zero (f
) is set by the compensation resistor (RC) and
ZEA
If f
is less than 5 x f , add a second capacitor (C )
C F
zMOD
the compensation capacitor (CC). There is an optional
from COMP to GND. The value of C is:
F
pole (f ) set by CF and RC to cancel the output capaci-
PEA
tor ESR zero if it occurs near the crossover frequency
(f ), where the loop gain equals 1 (0dB). Thus:
C
1
C =
F
2π × f
× R
zMOD
C
1
f
=
MOSFET Selection
pEA
2π × R
(
+ R × C
)
OUTEA
C
C
The key selection parameters to choose the n-channel
MOSFET used in the boost converter are as follows.
1
f
=
zEA
2π × R × C
C
C
Threshold Voltage
The boost n-channel MOSFETs must be a logic-level
1
f
=
p2EA
type with guaranteed on-resistance specifications at V
2π × R × C
GS
C
F
= 4.5V.
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MAX25201/MAX25202
36V HV Synchronous Boost Controller
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Maximum Drain-to-Source Voltage (VDS(MAX))
Multiphase Operation
Dual-Phase (MAX25202)
The MOSFET must be chosen with an appropriate VDS
rating to handle all VIN voltage conditions.
Dual-phase operation uses a MAX25202M as the mas-
ter controller and MAX25202S as the slave. Connect
these devices as shown in the Dual-Phase Application
Circuit. In this configuration, the master outputs a clock
from SYNCOUT that is 180° out-of-phase for driving the
slave FSYNCINS input. When synchronizing to an exter-
nal clock, connect the clock to FSYNCIN of the master
and MODE of the slave. The external clock must have
50% duty-cycle to ensure the 180° phase shift. To use
the internal oscillator from the master, drive FSYNCIN
of the master and MODE of the slave high (connect
to BIAS). Dual-phase solutions allow spread spectrum
operation on both the master and slave.
Current Capability
The n-channel MOSFET must deliver the input current
(I
):
IN(MAX)
D
MAX
×
LOAD(MAX)
1 − D
I
= I
IN(MAX)
MAX
Choose MOSFETs with the appropriate average current
at V = 4.5V.
GS
Low Voltage Operation
The devices start with a supply voltage as low as 4.5V,
and can operate after initial start up with a supply voltage
as low as 1.8V. At very low input voltages it is important
to remember that input current will be high and the power
components (inductor, MOSFET, and diode) must be
specified for this higher input current.
Layout Recommendations
Careful PCB layout is critical to achieve low switch-
ing losses and clean, stable operation. Layout of the
switching power components requires particular attention.
Follow these guidelines for good PCB layout:
In addition, the current-limit must be set high enough
so that the limit is not reached during the MOSFET's on
time, which would limit output power and eventually force
the MAX25201/MAX25202 into hiccup mode. Estimate
the maximum input current using the following equation:
● Keep high-current paths short, especially at the
ground terminals.
● Minimize resistance in high-current paths by keeping
the traces short and wide. Using thick (2oz vs. 1oz
copper) improves full load efficiency.
V
× I
OUT OUT
V
− V
V
OUT
SUPMIN
SUPMIN
● Connect the CS and SUP connections used for cur-
rent sensing directly across the sense resistor using
a Kelvin sense connection.
I
=
+ 0.5 ×
×
SUPMAX
η × V
V
f
× L
SUPMIN
OUT
SW
where I
is the maximum input current; V
and
SUPMAX
OUT
● Route noisy switching and clock traces away from
sensitive analog areas (FB, CS).
I
are the output voltage and current, respectively; η
OUT
is the estimated efficiency (which is lower at low input
voltages due to higher resistive losses); V is the
SUPMIN
minimum value of the input voltage; f
is the switching
SW
frequency; and L is the minimum value of the chosen
inductor.
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MAX25201/MAX25202
36V HV Synchronous Boost Controller
for Automotive Infotainment Applications
Typical Application Circuits
Synchronous Boost Application Circuit
BATTERY INPUT
3.5V TO 36V
OUTPUT
BST
BIAS
LX
DL
MAX25201
CS
SUP
EN
DH
OUT
MODE/
FSYNC
FB
FOSC
PGOOD
SS
C OM P
GND
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MAX25201/MAX25202
36V HV Synchronous Boost Controller
for Automotive Infotainment Applications
Typical Application Circuits (continued)
Dual-Phase Application Circuit
BATTERY INPUT
3.5V TO 36V
OUTPUT
BST
BIAS
LX
DL
MAX25202M
CS
EXTERNAL CLOCK
(OPTIONAL)
SUP
DH
OUT
FB
FSYNCIN
FSYNCOUT
EN
PGOOD
GND
COMP
SS
BST
BIAS
LX
DL
MAX25202S
CS
SUP
DH
OUT
COMP
EN
FB
FSYNCINS
MODE
GND
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MAX25201/MAX25202
36V HV Synchronous Boost Controller
for Automotive Infotainment Applications
Ordering Information
INTERNAL
SWITCHING
FREQUENCY
TEMP
RANGE
PIN-
PACKAGE
V
FIXED
SPREAD
SPECTRUM
OUT
PART
TOPOLOGY
RANGE
V
OUT
-40°C to
MAX25201ATEA/VY+
+125°C
16 SW
TQFN-EP*
SINGLE
PHASE
3.5V to 36V
3.5V to 36V
20V to 60V
20V to 60V
20V to 60V
20V to 60V
10
Adjustable
Adjustable
Adjustable
Adjustable
400kHz
OFF
ON
-40°C to
MAX25201ATEB/VY+
+125°C
16 SW
TQFN-EP*
SINGLE
PHASE
10
-40°C to
MAX25201ATEC/VY+
+125°C
16 SW
TQFN-EP*
SINGLE
PHASE
N/A
N/A
N/A
N/A
OFF
ON
-40°C to
MAX25201ATED/VY+
+125°C
16 SW
TQFN-EP*
SINGLE
PHASE
-40°C to
MAX25202MATEA/VY+
+125°C
16 SW
TQFN-EP*
2-PHASE
MASTER
ON
-40°C to
MAX25202SATEA/VY+
+125°C
16 SW
TQFN-EP*
2-PHASE
SLAVE
400kHz
ON
*EP = Exposed pad.
Maxim Integrated
│ 21
www.maximintegrated.com
MAX25201/MAX25202
36V HV Synchronous Boost Controller
for Automotive Infotainment Applications
Revision History
REVISION
NUMBER
REVISION
DATE
PAGES
DESCRIPTION
CHANGED
0
1
2
3
7/19
7/19
Initial release
—
Updated Ordering Information section
21
12/19
2/20
Updated Electrical Chracteristics table and Ordering Information
Removed remaining future-product notation in Ordering Information
4. 5, 21
21
For information on other Maxim Integrated products, visit Maxim Integrated’s website at www.maximintegrated.com.
Maxim Integrated cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim Integrated product. No circuit patent licenses
are implied. Maxim Integrated reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and max limits)
shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.
©
Maxim Integrated and the Maxim Integrated logo are trademarks of Maxim Integrated Products, Inc.
2019 Maxim Integrated Products, Inc.
│ 22
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