MAX746CSE [MAXIM]
High-Efficiency, PWM, Step-Down, N-Channel DC-DC Controller; 高效率， PWM ，降压型， N沟道DC- DC控制器
| 型号： | MAX746CSE |
| 厂家： | 
MAXIM INTEGRATED PRODUCTS |
| 描述： | High-Efficiency, PWM, Step-Down, N-Channel DC-DC Controller |
| 文件： | 总16页 (文件大小：158K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
19-0192; Rev 1; 11/93
Hig h -Effic ie n c y, P WM, S t e p -Do w n ,
N-Ch a n n e l DC-DC Co n t ro lle r
MAX746
_______________Ge n e ra l De s c rip t io n
____________________________Fe a t u re s
The MAX746 is a high-efficiency, high-current, step-down
DC-DC power-supply controller that drives external N-chan-
nel FETs. It provides 93% to 96% efficiency from a 6V supply
voltage with load currents ranging from 50mA up to 3A. It
uses a pulse-width-modulating (PWM) current-mode control
scheme to provide precise output regulation and low output
noise. The MAX746's 4V to 15V input voltage range, fixed
♦ 93% to 96% Efficiency for 50mA to 3A
Output Currents
♦ 4V to 15V Input Voltage Range
♦ Low 950µA Supply Current
♦ 1.4µA Shutdown Current
♦ Drives External N-Channel FETs
♦ Fixed-Frequency Current-Mode PWM (Heavy Loads)
♦ Idle-Mode PFM (Light Loads)
TM
5V/adjustable (Dual-Mode ) output, and adjustable current
limit make this device ideal for a wide range of applications.
High efficiency is maintained with light loads due to a propri-
TM
etary automatic pulse-skipping control (Idle-Mode ) scheme
♦ Cycle-by-Cycle Current Limiting
♦ 2V ±1.5% Accurate Reference Output
♦ Adjustable Soft-Start
that minimizes switching losses by reducing the switching fre-
quency at light loads. The low 950µA quiescent current and
ultra-low 1.4µA shutdown current further extend battery life.
♦ Undervoltage Lockout
External components are protected by the MAX746's cycle-
by-cycle current limit. The MAX746 also features a 2V ±1.5%
reference, a comparator for low-battery detection or level
translating, and soft-start and shutdown capability.
♦ Precision Comparator for Power-Fail or
Low-Battery Warning
______________Ord e rin g In fo rm a t io n
The MAX747—d is c us s e d in a s e p a ra te d a ta s he e t—
functions similarly to the MAX746, but drives P-channel logic
level FETs.
PART
TEMP. RANGE
0°C to +70°C
PIN-PACKAGE
16 Plastic DIP
16 Narrow SO
Dice*
MAX746CPE
MAX746CSE
MAX746C/D
MAX746EPE
MAX746ESE
MAX746MJE
________________________Ap p lic a t io n s
0°C to +70°C
5V-to-3.3V Green PC Applications
Notebook/Laptop Computers
Personal Digital Assistants
Battery-Operated Equipment
Cellular Phones
0°C to +70°C
-40°C to +85°C
-40°C to +85°C
-55°C to +125°C
16 Plastic DIP
16 Narrow SO
16 CERDIP
* Contact factory for dice specifications.
__________Typ ic a l Op e ra t in g Circ u it
__________________P in Co n fig u ra t io n
INPUT 6V TO 15V
TOP VIEW
V+
AV+
40mΩ
LBO
LBI
SS
GND
V+
CP
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
MAX746
CS
HIGH
CP
SHDN
EXT
OUTPUT
5V
ON/OFF
39µH
REF
HIGH
EXT
AGND
CS
MAX746
SHDN
FB
LOW-BATTERY
DETECTOR INPUT
440µF
LBI
OUT
CC
LOW-BATTERY
DETECTOR OUTPUT
LBO
REF
SS CC
AGND
GND
FB
AV+
OUT
DIP/SO
™Dual-Mode and Idle-Mode are trademarks of Maxim Integrated Products.
________________________________________________________________ Maxim Integrated Products
1
Ca ll t o ll fre e 1 -8 0 0 -9 9 8 -8 8 0 0 fo r fre e s a m p le s o r lit e ra t u re .
Hig h -Effic ie n c y, P WM, S t e p -Do w n ,
N-Ch a n n e l DC-DC Co n t ro lle r
ABSOLUTE MAXIMUM RATINGS
Operating Temperature Ranges:
Supply Voltage V+, AV+ to GND..............................-0.3V to 17V
HIGH, EXT to GND....................................................-0.3V to 21V
AGND to GND..........................................................-0.3V to 0.3V
All Other Pins ................................................-0.3V to (V+ + 0.3V)
MAX746C_E........................................................0°C to +70°C
MAX746E_E .....................................................-40°C to +85°C
MAX746MJE ..................................................-55°C to +125°C
Junction Temperatures:
Reference Current (I
) ....................................................±2mA
REF
MAX746C_E/E_E..........................................................+150°C
MAX746MJE.................................................................+175°C
Storage Temperature Range .............................-65°C to +160°C
Lead Temperature (soldering, 10sec) .............................+300°C
Continuous Power Dissipation (T = +70°C)
A
Plastic DIP (derate 10.53mW/°C above +70°C) ..........842mW
Narrow SO (derate 8.70mW/°C above +70°C) ............696mW
CERDIP (derate 10.00mW/°C above +70°C)...............800mW
MAX746
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(V+ = 10V, I
= 0A, I
= 0µA, T = T
to T , unless otherwise noted.)
MAX
LOAD
REF
A
MIN
PARAMETER
Input Voltage
SYMBOL
V+
CONDITIONS
MIN
TYP
MAX
UNITS
4
15
V
V+ = 6V to 15V, 0V < (V+ - CS) < 0.125V,
FB = 0V (includes line and load regulation)
Output Voltage
V
4.85
5.08
5.25
V
V
OUT
MAX746C
1.96
1.95
2.00
2.00
0.05
2.04
2.05
(V+ - CS) = 0V,
Feedback Voltage
V
FB
external feedback mode
MAX746E/M
V+ = 6V to 15V, FB = 0V
Line Regulation
Load Regulation
Efficiency
%/V
%
V+ = 4V to 15V, external feedback mode
0V < (V+ - CS) < 0.125V
0.1
2.5
1.3
94
50
Circuit of Figure 1, I
V+ = 6V
= 0.5A to 2.5A,
LOAD
%
µA
mV
nA
OUT Leakage Current
FB Input Logic Low
V
OUT
= 5V
80
40
For dual-mode switchover
FB = 2V
FB Input Leakage Current
1
100
2.03
2.04
20
MAX746C
1.97
1.96
2.00
2.00
9
I
= 0µA
Reference Voltage
V
V
REF
REF
MAX746E/M
I
= 0µA to 100µA
Reference Load Regulation
Soft-Start Source Current
mV
µA
µA
REF
SS = 0V
SS = 2V
0.5
1.0
500
1.1
1.5
Soft-Start Fault Current (Note 1)
100
MAX746C
1.4
1.7
Operating, V+ = 15V
MAX746E/M
mA
Supply Current (Note 2)
Oscillator Frequency
I
SUPP
Operating, V+ = 10V
Shutdown mode
0.95
1.4
µA
20
MAX746C
85
80
100
100
115
120
f
kHz
OSC
MAX746E/M
2
_______________________________________________________________________________________
Hig h -Effic ie n c y, P WM, S t e p -Do w n ,
N-Ch a n n e l DC-DC Co n t ro lle r
MAX746
ELECTRICAL CHARACTERISTICS (continued)
(V+ = 10V, I
= 0A, I
= 0µA, T = T
to T , unless otherwise noted.)
MAX
LOAD
REF
A
MIN
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
Maximum Duty Cycle
V+ = 6V
91
96
%
V
Charge-Pump Output Voltage
V
HIGH
I
= 0mA to 10mA
V+ + 4 V+ + 5 V+ + 6
HIGH
Current-Sense Amplifier
Current-Limit Threshold
V
LIMIT
V+ – CS
forced to 15V, I
125
150
175
mV
EXT Output High
V
= -1mA
V
- 0.1
V
V
HIGH
EXT
EXT
HIGH
EXT Output Low
V
forced to 15V, I
= 1mA
0.25
HIGH
EXT Sink Current
V
= 15V, V
= 12.5V
= 2.5V
160
270
24
mA
mA
kΩ
HIGH
EXT
EXT Source Current
Compensation Pin Impedance
V
HIGH
= 15V, V
EXT
MAX746C
1.97
1.96
2.00
2.00
2.03
2.04
0.4
100
1
LBI Threshold Voltage
LBI falling
= 0.5mA
V
MAX746E/M
LBO Output Voltage Low
LBI Input Leakage Current
LBO Output Leakage Current
SHDN Input Voltage Low
SHDN Input Voltage High
SHDN Input Leakage Current
V
I
V
nA
µA
V
OL
SINK
LBI = 2.5V
V+ = 15V, LBO = 15V, LBI = 2.5V
V
0.4
IL
V
2.0
V
IH
SHDN = 10V
0.1
100
nA
Note 1: The soft-start fault current is the current sink capability of SS when V
< 1V or when the device is in shutdown.
REF
Note 2:
I
is the supply current drawn by V+, which includes the current drawn by the charge pump. The charge pump
SUPP
doubles the current drawn by HIGH from the V+ input, so I
= I + 2I
.
SUPP
V+
HIGH
_______________________________________________________________________________________
3
Hig h -Effic ie n c y, P WM, S t e p -Do w n ,
N-Ch a n n e l DC-DC Co n t ro lle r
__________________________________________Typ ic a l Op e ra t in g Ch a ra c t e ris t ic s
(Circuit of Figure 1a, T = +25°C, unless otherwise noted.)
A
CONTINUOUS-CONDUCTION MODE
BOUNDARY AND CORRESPONDING
PEAK INDUCTOR CURRENT
N0-LOAD SUPPLY CURRENT
vs. SUPPLY VOLTAGE
NO-LOAD SUPPLY CURRENT
vs. TEMPERATURE
15
13
1.2
1.1
4
3
DISCONTINUOUS-
CONDUCTION REGION
V+ = 9V
= 5V
V
OUT
MAX746
PEAK
INDUCTOR
CURRENT
11
2
1
0
1.0
0.9
ENTIRE
CIRCUIT
9
7
CONTINUOUS-
CONDUCTION
REGION
SCHOTTKY DIODE
LEAKAGE EXCLUDED
5
0.8
0.7
0.9
1.1
1.3
1.5
1.7
-75 -50 -25
0
25 50 75 100 125
5
7
9
11
13
15
OUTPUT CURRENT (A)
TEMPERATURE (°C)
SUPPLY VOLTAGE (V)
EFFICIENCY vs. OUTPUT CURRENT
EFFICIENCY vs. OUTPUT CURRENT
EFFICIENCY vs. OUTPUT CURRENT
100
100
100
CIRCUIT OF FIGURE 1c
= 5V
CIRCUIT OF FIGURE 1b
V
V
= 3.3V
OUT
OUT
V+ = 5V
V
= 6V
IN
90
90
90
V
IN
= 6V
V
IN
= 12V
V
IN
= 9V
V
IN
= 12V
80
70
80
70
80
70
CIRCUIT OF FIGURE 1a
= 5V
V
OUT
0.01
0.1
1
10
0.01
0.1
1
10
0.01
0.1
1
10
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
PEAK INDUCTOR CURRENT
vs. OUTPUT CURRENT
PEAK INDUCTOR CURRENT
vs. OUTPUT CURRENT
PEAK INDUCTOR CURRENT
vs. OUTPUT CURRENT
1.5
1.0
4
4
3
2
1
CIRCUIT OF FIGURE 1a
CIRCUIT OF FIGURE 1c
= 5V
CIRCUIT OF FIGURE 1b
V
OUT
= 5V
V
= 3.3V
V
OUT
OUT
V+ = 5V
3
2
1
V
IN
= 12V
V
IN
= 12V
0.5
0
V
IN
= 9V
V
IN
= 6V
V
IN
= 6V
0
0
0.01
0.1
1
10
0.01
0.1
1
10
0.01
0.1
1
10
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
4
_______________________________________________________________________________________
Hig h -Effic ie n c y, P WM, S t e p -Do w n ,
N-Ch a n n e l DC-DC Co n t ro lle r
MAX746
____________________________Typ ic a l Op e ra t in g Ch a ra c t e ris t ic s (c o n t in u e d )
(Circuit of Figure 1a, T = +25°C, unless otherwise noted.)
A
LOAD-TRANSIENT RESPONSE
LINE-TRANSIENT RESPONSE
LOAD-TRANSIENT RESPONSE
10V
A
8V
A
B
A
B
B
200µs/div
500ms/div
1ms/div
A: LOAD CURRENT, 0.1A TO 1.5A, 1A/div
A: V+ = 8V TO 10V, 2V/div
A: LOAD CURRENT, 0.1A TO 1.5A, 1A/div
B: V RIPPLE, 50mV/div, AC-COUPLED
OUT
B: V RIPPLE, 100mV/div
OUT
B: V RIPPLE, 50mV/div, AC COUPLED
OUT
V+ = 10V
I
= 3A
V+ = 10V
OUT
MODERATE-LOAD, IDLE-MODE
WAVEFORMS
CONTINUOUS-CONDUCTION MODE
WAVEFORMS
DISCONTINUOUS-CONDUCTION
IDLE-MODE WAVEFORMS
A
A
B
A
B
C
B
C
0V
C
20µs/div
20µs/div
5µs/div
A: EXT VOLTAGE, 10V/div
A: EXT VOLTAGE, 10V/div
A : EXT VOLTAGE, 20V/div
B: INDUCTOR CURRENT, 500mA/div
B: INDUCTOR CURRENT, 500mA/div
B : INDUCTOR CURRENT 1A/div
C: V RIPPLE, 50mV/div, AC-COUPLED
OUT
C: V RIPPLE, 50mV/div, AC-COUPLED
OUT
C : V RIPPLE, 50mV/div
OUT
V+ = 6V, I = 480mA
OUT
V+ = 10V, I = 75mA
OUT
V+ = 10V, I = 3A
OUT
_______________________________________________________________________________________
5
Hig h -Effic ie n c y, P WM, S t e p -Do w n ,
N-Ch a n n e l DC-DC Co n t ro lle r
______________________________________________________________P in De s c rip t io n
PIN
1
NAME
LBO
LBI
FUNCTION
Low-battery output is an open-drain output that goes low when LBI is less than 2V. Connect to V+ through a
pull-up resistor. Leave floating if not used. LBO is disabled in shutdown mode.
2
Input to the low-battery comparator. Tie to V+ or GND if not used.
MAX746
Soft-start limits start-up surge currents. On power-up, it charges the soft-start capacitor, slowly raising the peak
current limit to the level set by the sense resistor.
3
SS
2V reference output can source 100µA for external loads. Bypass with 1µF. The reference is disabled in shutdown mode.
4
REF
Active-high logic input. In shutdown mode, V
Connect to GND for normal operation.
= 0V and the supply current is reduced to less than 20µA.
OUT
5
SHDN
Feedback input for adjustable-output operation. Connect to GND for fixed 5V output. Use a resistor-divider net-
work to adjust the output voltage (see Setting the Output Voltage section).
6
7
8
9
FB
CC
AC compensation input for the error amplifier. Connect a capacitor between CC and GND for fixed 5V-output
operation (see Compensation Capacitor section).
Quiet supply voltage for sensitive analog circuitry. Also the noninverting input to the current-sense amplifier. A
separate bypass capacitor is not recommended for AV+.
AV+
OUT
Output voltage sense that connects to the internal resistor divider. Bypass with 0.1µF to AGND, close to the IC
for fixed output operation. Leave unconnected for adjustable-output operation.
10
11
CS
Inverting input to the current-sense amplifier. Connect the current-sense resistor (R
) from AV+ to CS.
SENSE
AGND
Quiet analog ground.
Power MOSFET gate-drive output that swings between HIGH and GND. EXT is not protected against short cir-
cuits to V+ or AGND.
12
EXT
13
14
15
16
HIGH
CP
Regulated high-side voltage, 5V above the V+ supply voltage.
Charge-pump output that generates a 0V to V+, 50kHz square wave (see Charge Pump section).
High-current supply voltage for the charge pump.
V+
GND
High-current ground return for the output driver and charge pump.
current-mode pulse-width-modulating (PWM) control
____________________Ge t t in g S t a rt e d
scheme that results in tight output-voltage regulation,
excellent load- and line-transient response, low noise,
and high efficiency over a wide range of load currents.
Efficiency at light loads is further enhanced by a propri-
etary idle-mode switching control scheme that skips
oscillator cycles in order to reduce switching losses.
Other features include undervoltage lockout, shutdown,
and a low-battery detection comparator.
Figure 1a shows the 5V-output 3A standard application
circuit, Figure 1b shows the 3.3V-output 3A standard
application circuit, and Figure 1c shows the 5V-output
1.5A standard application circuit. Most applications will
be served by these circuits. To learn more about compo-
nent selection for particular applications, refer to the
Design Procedure section. To learn more about the oper-
ation of the MAX746, refer to the Detailed Description.
Op e ra t in g P rin c ip le
Figure 2 is the MAX746 block diagram. The MAX746
regulates using an inner current-feedback loop and an
outer voltage-feedback loop. A slope-compensation
scheme stabilizes the current loop; the dominant pole,
forme d b y the outp ut filte r c a p a c itor a nd the loa d ,
stabilizes the voltage loop.
_______________De t a ile d De s c rip t io n
The MAX746 monolithic, CMOS, step-down, switch-
mode power-supply controller provides high-side drive
for external logic-level N-channel FETs. A charge pump
generates a voltage 5V above the supply voltage for
high-side drive capability. The MAX746 uses a unique
6
_______________________________________________________________________________________
Hig h -Effic ie n c y, P WM, S t e p -Do w n ,
N-Ch a n n e l DC-DC Co n t ro lle r
MAX746
V
IN
6V TO 15V
C9
4.7µF
C3
0.1µF
C2
100µF
D4
1N5817
D3
1N914
D2
1N914
R2
R1
15
V+
*
*
C8
0.1µF
2
3
14
13
8
LBI
CP
HIGH
AV+
C5
0.1µF
SS
R
SENSE
40mΩ
C6
10
1.0µF
4
6
MAX746
CS
REF
Q1
Si9410DY
12
7
L1
39µH
EXT
N
FB
5V
AT 3A
5
CC
SHDN
AGND
D1
NSQ03A03
C7
2.7nF
C1
430µF
11
9
1
OUT
C4
0.1µF
R3
100k
LB0
GND
16
SEE TABLE 2 FOR DIODE SELECTION.
*
Figure 1a. 5V Standard Application Circuit (15W)
Under these conditions, the inductor must be scaled to the
current-sense resistor value.
Dis c o n t in u o u s -/Co n t in u o u s -
Co n d u c t io n Mo d e s
The MAX746 is designed to operate in continuous-con-
duction mode (CCM) but can also operate in discontinu-
ous-conduction mode (DCM), making it ideal for variable-
load applications. In DCM, the current starts at zero and
returns to zero on each cycle. In CCM, the inductor current
never returns to zero; it consists of a small AC component
superimposed on a DC offset. This results in higher current
capability because the AC component in the inductor cur-
rent waveform is small. It also results in lower output noise,
since the inductor does not exhibit the ringing that would
occur if the current reached zero (see inductor waveforms
in the Typical Operating Characteristics). To transfer equal
amounts of energy to the load in one cycle, the peak cur-
rent level for the discontinuous waveform must be much
larger than the peak current for the continuous waveform.
Overcompensation adds a pole to the outer voltage feed-
back-loop response, degrading loop stability. This may cause
voltage-mode pulse-frequency-modulation instead of PWM
operation. Undercompensation results in inner current feed-
back-loop instability, and may cause the inductor current to
staircase. Ideal matching between the sense resistor and
inductor is not required; it can differ by ±30% or more.
Os c illa t o r a n d EXT Co n t ro l
The oscillator frequency is nominally 100kHz, and the duty
cycle varies from 5% to 96%, depending on the input/out-
put voltage ratio. EXT, which provides the gate drive for the
external logic-level N-FET, is switched between HIGH and
GND at the switching frequency. EXT is controlled by a
unique two-comparator control scheme consisting of a PWM
comparator and an idle-mode comparator (Figure 2). The
PWM comparator determines the cycle-by-cycle peak cur-
rent with heavy loads, and the idle-mode comparator sets
S lo p e Co m p e n s a t io n
Slope compensation stabilizes the inner current-feedback
loop by adding a ramp signal to the current-sense amplifier
output. Ideal slope compensation can be achieved by
adding a linear ramp, with the same slope as the declining
inductor current, to the rising inductor current-sense voltage.
the light-load peak current. As V
begins to drop, EXT
OUT
goes high and remains high until both comparators trip.
With heavy loads, the idle-mode comparator trips first and
the PWM control comparator determines the EXT on-time;
_______________________________________________________________________________________
7
Hig h -Effic ie n c y, P WM, S t e p -Do w n ,
N-Ch a n n e l DC-DC Co n t ro lle r
V
IN
4.5V TO 6V
C11
1µF
C3
0.1µF
C2
100µF
D2
D3
D5
D6
D4
1N5817
1N914 1N914 1N914 1N914
R2
R1
15
V+
C8
0.1µF
C9
1µF
C10
0.1µF
MAX746
2
14
13
8
LBI
CP
HIGH
AV+
C5
0.1µF
3
4
SS
R
SENSE
C6
1µF
40mΩ
10
MAX746
CS
REF
Q1
Si9410DY
12
7
L1
22µH
EXT
N
*
3.3V
AT 3A
CC
5
SHDN
AGND
D1
C4
0.1µF
C3
660µF
9
6
NSQ03A03
R5
13k (1%)
OUT
11
FB
C7
2nF
R4
20k (1%)
R3
100k
1
LB0
GND
16
SUMIDA CDR125 22µH SURFACE-MOUNT INDUCTOR
*
Figure 1b. 3.3V Standard Application Circuit (9.9W)
with light loads, the PWM comparator trips quickly and the
idle-mode comparator sets the EXT on-time.
quency. When the voltage at HIGH exceeds AV+ by
5V, the charge-pump oscillator is inhibited (Figure 2).
When the voltage at HIGH is less than 4.3V below V+,
undervoltage lockout occurs. Use the voltage tripler
(Figure 3b) when V+ ≤ 6V; otherwise, use the voltage
doubler (Figure 3a).
Traditional PWM converters continue to switch on every
cycle, even when the inductor current is discontinuous
due to smaller loads, decreasing light-load efficiency.
In contrast, the MAX746’s idle-mode comparator increas-
es the switch on-time, allowing more energy to be trans-
ferred per cycle. Since fewer cycles are required, the
switching frequency is reduced, resulting in minimal
switching losses and increased efficiency.
S o ft -S t a rt a n d Cu rre n t Lim it in g
The MAX746 draws its highest current at power-up. If
the power source to the MAX746 cannot provide this
initial elevated current, the circuit may not function cor-
rectly. For example, after prolonged use the increased
series resistance of a battery may prevent it from pro-
vid ing a d e q ua te initia l s urg e c urre nts whe n the
MAX746 is brought out of shutdown. Using soft-start
(SS) minimizes the possibility of overloading the incom-
ing supply at power-up by gradually increasing the
peak current limit. Connect an external capacitor from
SS to AGND to reduce the initial peak currents drawn
from the supply.
The light-load output noise spectrum widens due to the
variable switching frequency in idle-mode, but output
ripple remains low. Using the Typical Operating Circuit,
with a 9V input and a 125mA load current, output ripple
is less than 40mV.
Ch a rg e P u m p
The MAX746 contains all the control circuitry required
to p rovid e a re g ula te d c ha rg e -p ump volta g e 5V
above V+ for high-side driving N-channel logic FETs.
The charge pump operates with a nominal 50kHz fre-
8
_______________________________________________________________________________________
Hig h -Effic ie n c y, P WM, S t e p -Do w n ,
N-Ch a n n e l DC-DC Co n t ro lle r
MAX746
V
IN
6V TO 15V
C9
4.7µF
C3
0.1µF
C2
47µF
D4
1N5817
D3
1N914
D2
1N914
R2
R1
15
V+
*
*
C8
0.1µF
2
3
14
13
8
LBI
CP
HIGH
AV+
C5
0.1µF
SS
R
SENSE
C6
75mΩ
10
1µF
4
6
MAX746
CS
REF
Q1
Si9410DY
12
7
L1
N
82µH
EXT
**
FB
5V
AT 1.5A
5
CC
SHDN
AGND
D1
NSQ03A03
C7
1nF
C1
220µF
11
9
1
OUT
C4
0.1µF
R3
100k
LB0
GND
16
SEE TABLE 2 FOR DIODE SELECTION.
*
SUMIDA CDR125 SURFACE-MOUNT INDUCTOR.
**
Figure 1c. 5V Standard Application Circuit (7.5W)
The steady-state SS pin voltage is typically 3.8V. On
power-up, SS sources 1µA until its voltage reaches
3.8V. The current-limit comparator inhibits EXT switch-
ing until the SS voltage reaches 1.8V. The peak current
limit is set by:
S h u t d o w n Mo d e
When SHDN is high, the MAX746 is shut down. In this
mode, the internal biasing circuitry (including EXT) is
turned off, V
drops to 0V, and the supply current
OUT
drops to 1.4µA (20µA max). This excludes external
c o m p o n e n t le a ka g e , wh ic h m a y a d d s e ve ra l
mic roa mp s to the s hutd own s up p ly c urre nt for the
entire circuit. SHDN is a logic input. Connect SHDN to
GND for normal operation.
V
LIMIT
150mV (typ)
___________
_________
I
PK
=
=
R
R
SENSE
SENSE
where V
is the differential voltage across the current-
LIMIT
sense amplifier inputs. Figure 4 shows how the SS peak
current limit increases as the voltage on SS rises for two
Lo w -Ba t t e ry De t e c t o r
The MAX746 provides a low-battery comparator that
compares the voltage on LBI to the reference voltage.
LBO, an open-drain output, goes low when the LBI volt-
R
values.
SENSE
Un d e rvo lt a g e Lo c k o u t
age is below V
. Use a resistor-divider network, as
REF
Undervoltage lockout inhibits operation of EXT until the
charge pump is capable of generating a voltage greater
than 4.3V above the supply voltage (Figure 2). When
the undervoltage-lockout comparator detects an under-
voltage condition, the switching action at EXT is halted.
shown in the Input Voltage Monitor Circuit (Figure 5),
to set the trip voltage (V ) at the desired level. In
TRIP
this circuit, LBO goes low when V+ ≤ V
. LBO is high
TRIP
impedance in shutdown mode.
_______________________________________________________________________________________
9
Hig h -Effic ie n c y, P WM, S t e p -Do w n ,
N-Ch a n n e l DC-DC Co n t ro lle r
LBO
EXT
HIGH
V+
PUMP
FROM AV+
LBI
CHARGE-PUMP CONTROL
COMPARATOR
4.3V
N
MAX746
LOW-BATTERY
COMPARATOR
T
T FLIP-
FLOP
5V
Q
+2V
REFERENCE
UNDERVOLTAGE-
LOCKOUT
COMPARATOR
REF
100kHz
OSCILLATOR
OUT
EXT
CONTROL
SHDN
CC
FB
PWM
COMPARATOR
ERROR
AMPLIFIER
DUAL-MODE
COMPARATOR
100mV
CURRENT-SENSE
AMPLIFIER
AV+
CS
LIGHT-LOAD
COMPARATOR
Σ
SLOPE-
COMPENSATION
RAMP
V
RAMP
50mV
SOFT-START
CIRCUITRY
SS
CURRENT-LIMIT
COMPARATOR
AGND
GND
Figure 2. Block Diagram
10 ______________________________________________________________________________________
Hig h -Effic ie n c y, P WM, S t e p -Do w n ,
N-Ch a n n e l DC-DC Co n t ro lle r
MAX746
3a. CHARGE-PUMP
VOLTAGE DOUBLER
PEAK CURRENT LIMIT
vs. SOFT-START VOLTAGE
3
2
D2
D3
D4
V
IN
1N914 1N914 1N5817
R
SENSE
= 50mΩ
15
C8
0.1µF
C9
1µF
V+
V+ - V = 150mV
CS
MAX746
T FLIP-
CP 14
1
0
T
Q
FLOP
CLK
HIGH 13
R
SENSE
= 100mΩ
100kHz
OSCILLATOR
0
1
2
3
4
SOFT-START VOLTAGE (V)
5V
3b. CHARGE-PUMP
VOLTAGE TRIPLER
Figure 4. Peak Current Limit vs. Soft-Start Voltage
AV+
GND
16
D4
1N5817
V
IN
D2
D3
D5
D6
V
IN
15
1N914 1N914
1N914 1N914
…TO V OR V
OUT IN
V+
C8
0.1µF
C9
1µF
C10
0.1µF 1µF
C11
R2
R1
R3
100k
15
V+
MAX746
14
13
2
1
CP
LBO
LBI
LOW-BATTERY
OUTPUT
MAX746
HIGH
GND
16
GND
16
V
TRIP
(
)
R = R1
-1
2
V
REF
V
REF
= 2.0V
Figure 3. Charge-Pump Configurations
Figure 5. Input Voltage Monitor Circuit
the two modes while operating. If two different output
voltages are required, use external feedback mode
with a resistor network similar to the 3.3V/5V adjustable
output circuit shown in Figure 7.
__________________De s ig n P ro c e d u r e
S e t t in g t h e Ou t p u t Vo lt a g e
The MAX746’s dual-mode output voltage can be set
to 5V by grounding FB, or it can be adjusted from
2V to 14V using external resistors R4 and R5 config -
ured as shown in Figure 6. Select feedback resistor
R4 in the 10kΩ to 60kΩ range. R5 is given by:
S e le c t in g R
S ENS E
), firs t
To s e le c t the s e ns e -re s is tor va lue (R
SENSE
a p p roxima te the p e a k c urre nt a s s uming
I
is
PK
(1.1) (I ), where I is the maximum load cur-
LOAD
LOAD
rent. Once all component values have been deter-
mined, the actual peak current is given by:
V
OUT
_______
2V
R5 = (R4)
– 1
)
(
V
V
OUT
OUT
___________
_______
The MAX746 is designed to use either internal or exter-
nal feedback mode, but should not be toggled between
I
PK
= I
LOAD
+
1–
)(
(
)
(2L) (f
)
V
IN
OSC
______________________________________________________________________________________ 11
Hig h -Effic ie n c y, P WM, S t e p -Do w n ,
N-Ch a n n e l DC-DC Co n t ro lle r
V
IN
12
15
N
EXT
L
V+
R5
V
OUT
6
9
V
FB
OUT
C1
MAX746
D1
R5
C7*
26.1k (1%)
R4
6
9
FB
MAX746
C7
MAX746
R4a
OUT
17.4k (1%)
OUT
SELECT WITH FET OFF:
R5
R4 = 10kΩ TO 60kΩ
5V/3.3V
R4b
22.6k (1%)
GND
16
V
= V
+1
R5
OUT REF
N
(
(
)
V
R4a
OUT
R5 = R4
-1
(
)
V
REF
SELECT WITH FET OFF:
V
= V
OUT REF
V
= 2.0V NOMINAL
+1
REF
)
R4a + R4b
*SEE COMPENSATION CAPACITOR SECTION.
V
REF
= 2.0V NOMINAL
Figure 6. Adjustable Output Circuit
Figure 7. 3.3V/5V Ajustable Output Circuit
where V
compensation linear ramp signal.
is the 50mV peak value of the slope-
RAMP(max)
Next, determine the value of R
such that:
SENSE
V
125mV
LIMIT(min)
_____________
________
Although 38µH is the calculated value, the component
used may have a tolerance of ±30% or more.
R
=
=
SENSE
I
I
PK
PK
Inductors with molypermalloy powder (MPP), Kool Mµ,
or ferrite are recommended. Inexpensive iron-powder
core inductors are not suitable, due to their increased
core losses, especially at switching frequencies in the
100kHz range. MPP and Kool Mµ cores have low per-
meability, allowing larger currents.
For e xa mp le , to ob ta in 5V a t 3A, I
= 3.3A a nd
PK
R
= 125mV/3.3A = 38mΩ.
SENSE
The sense resistor should have a power rating greater
2)
than (I
(R
) with an adequate safety margin.
PK
SENSE
With a 3A load current, I = 3.3A and R
= 38mΩ.
SENSE
PK
The power dissipated by the resistor (assuming an 80%
duty cycle) is 331mW. Metal-film resistors are recom-
me nd e d . Do not us e wire -wound re s is tors b e c a us e
their inductance will adversely affect circuit operation.
For highest efficiency, use a coil with low DC resis-
tance. To minimize radiated noise, use a toroid, a pot
core, or a shielded coil.
The duty cycle (for continuous conduction) is determined
from the following equation:
It is customary to select an inductor with a saturation
rating that exceeds the peak current set by R
,
SENSE
V
V
but inductors are often specified very conservatively.
If the inductor’s core losses do not cause excessive
temperature rise (inductor wire insulation is usually
rated for +125°C) and the associated efficiency loss-
es are minimal, inductors with lower current ratings
are acceptable.
OUT
DIODE
+
_____________________
Duty Cycle (%) =
x 100%
V+ - V + V
SW
DIODE
where V is the voltage drop across the external
N-FET and sense resistor. V
SW
can be approximated
SW
as [I
x (r
+ R
)].
LOAD
DS(ON)
SENSE
In the 3.3V Standard Application Circuit (Figure 1b), the
ind uc tor s e le c te d ha s a 2.2A c urre nt ra ting e ve n
though the peak current is 3.3A. This inductor was
selected for two reasons: it is the highest-rated readily
available surface-mount inductor of its size, and lab
tests have verified that the core-loss increase is mini-
mal. With a 3A load current, the inductor current does
not begin showing significant losses due to saturation
until the supply voltage increases to 10V (the maximum
supply for this circuit is 6V).
In d u c t o r S e le c t io n
Once the sense-resistor value is determined, calculate
the inductor value (L) using the following equation. The
correct inductor value ensures proper slope compen-
sation. Continuing from the equations above:
(
) (
)
OUT
R
V
SENSE
______________________
L =
(VRAMP(max)) (f
)
OSC
(
) (
)
38mΩ 5V
_____________________
(50mV) (100kHz)
=
= 38µH
12 ______________________________________________________________________________________
Hig h -Effic ie n c y, P WM, S t e p -Do w n ,
N-Ch a n n e l DC-DC Co n t ro lle r
MAX746
To ensure stability, the minimum capacitance and max-
imum ESR values are:
Ex t e rn a l Lo g ic -Le ve l N-FET S e le c t io n
To ensure the external N-FET is turned on hard, use
logic-level or low-threshold N-FETs. Three important
parameters to note when selecting the N-FET are the
(5) (V
)
REF
______________________________
C1
>
(min)
(2π) (GBW) (VOUT) (R
)
SENSE
total gate charge (Q ), on resistance (r
), and
g
DS(ON)
and,
reverse transfer capacitance (C
).
RSS
(VOUT ) (R
)
SENSE
___________________
Q
includes all capacitances associated with charging
g
ESR
<
C1
the gate. Use the typical Q value for best results; the
(V
REF
)
g
maximum value is usually grossly overspecified, since
it is a guaranteed limit and not the measured value.
The typical total gate charge should be 50nC or less;
with la rg e r numbe rs, EXT ma y not be a b le to a de-
q ua te ly d rive the g a te . EXT s ink/s ourc e c a p a b ility
where GBW = the loop gain-bandwidth product, 15kHz.
Sprague 595D surface-mount solid tantalum capacitors
and Sanyo OS-CON through-hole capacitors are rec-
ommended due to their extremely low ESR. OS-CON
capacitors are particularly useful at low temperatures.
For best results when using other capacitors, increase
the output filter capacitor’s size or use capacitors in
parallel to reduce the ESR.
(I ) is typically 210mA.
EXT
The two mos t s ig nific a nt los s e s c ontrib uting to the
2
N-FET’s power dissipation are I R losses and switching
losses. CCM power dissipation (P ), is approximated by:
D
Bypass OUT with a 0.1µF (C4) capacitor to GND when using
a fixed 5V output (Figures 1a and 1c). With adjustable-output
operation, place C4 between the output voltage and AGND
as close to the IC as possible (Figure 1b).
P
= (Duty Cycle) (I 2) (rDS(ON)) +
D
PK
2
(V+ ) (CRSS) (IPK) (f
)
OSC
__________________________
(I
)
EXT
The circuit load-step response is improved by using a
larger output filter capacitor or by placing a low-cost
bulk capacitor in parallel with the required low-ESR
output filter capacitor. The output voltage sag under a
whe re the d uty c yc le is a p p roxima te ly V
/V+ ,
are given in the
OUT
f
= 100kHz, and r
and C
OSC
DS(ON)
RSS
d a ta s he e t of the c hos e n N-FET. In the e q ua tion,
is assumed constant, but is actually a function
load step (I
) is approximated by:
STEP
r
DS(ON)
of temperature. The equation given does not account
for losses incurred by charging and discharging the
gate capacitance, because that energy is dissipated
by the gate-drive circuitry, not the N-FET.
2
(I
) (L)
STEP
_____________________________________
V
=
SAG
(2) (C1) (VIN(MIN) (D
- V
)
MAX
OUT
where DMAX is the maximum duty cycle (91% worst
case). The equation assumes an input/output voltage
differential of 2V or more. Table 1 gives measured val-
ues of output voltage sag with a 30mA to 3A load step
for various input voltages and output filter capacitors.
Refer also to the AC Stability with Low Input/Output
Differentials section.
The Standard Application Circuits (Figure 1) use an
8-pin, Si9410DY, surface-mount N-FET that has 0.05Ω
on resistance with a 4.5V V . Optimum efficiency is
GS
obtained when the voltage at the source swings between
the supply rails (within a few hundred millivolts).
Dio d e S e le c t io n
The MAX746’s high switching frequency demands a
high-speed rectifier. Schottky diodes are recommend-
ed. Ensure that the Schottky diode average current
rating exceeds the maximum load current.
Input Bypass Capacitor
The input bypass capacitor C2 reduces peak currents
drawn from the voltage source, and also reduces the
amount of noise at the voltage source caused by the
MAX746’s fa s t s witc hing a c tion (this is e s p e c ia lly
important when other circuitry is operated from the
same source). The input capacitor ripple current rating
must exceed the RMS input ripple current.
Ca p a c it o r S e le c t io n
Output Filter Capacitor
The output filter capacitor C1 should have a low effec-
tive series resistance (ESR), and its capacitance should
remain fairly constant over temperature. This is espe-
cially true when in CCM, since the output filter capaci-
tor a nd the loa d form the d omina nt p ole tha t
stabilizes the voltage loop.
I
= RMS AC input current
RMS
(
√
) (
)
V
V
V
OUT
IN - OUT
_______________________
= I
(
)
LOAD
V
IN
______________________________________________________________________________________ 13
Hig h -Effic ie n c y, P WM, S t e p -Do w n ,
N-Ch a n n e l DC-DC Co n t ro lle r
Table 1. Measured Output Voltage Sag
with 30mA to 3A Load Step*
Table 2. Charge-Pump Configuration
V+
CHARGE-PUMP CONFIGURATION
OUTPUT
FILTER
CAPACITOR
C1 (µF)
OUTPUT VOLTAGE SAG (mV)
FOR VARIOUS INPUT VOLTAGES
Voltage tripler with 1N914 diodes for D2,
D3, D5, and D6
V+ ≤ 6V
V
IN
=6V
V
IN
=6.5V
V
=7V
IN
V =9V
IN
V =10V
IN
Voltage doubler with 1N5817 Schottky
diodes for D2 and D3
6V < V+ < 6.5V*
MAX746
440
660
880
400
260
200
250
190
100
210
160
90
140
70
90
50
25
Voltage doubler with 1N914 diodes for
D2 and D3
V+ ≥ 6.5V*
40
* When using the voltage-doubler circuit over the military
temperature range, increase the 6.5V limit to 7V.
*Circuit of Figure 1a.
For load currents up to 3A, 100µF (C2) in parallel with
0.1µF (C3) is adequate. Smaller bypass capacitors may
also be acceptable for lighter loads. The input voltage
source impedance determines the size of the capacitor
required at the V+ input. As with the output filter capaci-
tor, a low-ESR capacitor (Sanyo OS-CON, Sprague 595D
or equivalent) is recommended for input bypassing.
voltage and load current. With a 3A load current, a 10V
input voltage, and a 0.1µF soft-start capacitor, it typi-
cally takes 240ms for the MAX746 to power up.
A
0.47µF soft-start capacitor increases the start-up time
to approximately 2.3sec.
Bypass REF with a 1µF capacitor (C6).
Compensation Capacitor
With a fixed 5V output, connect a compensation capac-
itor (C7) between CC and AGND to optimize transient
response. Appropriate compensation is determined by
the size and ESR of the output filter capacitor (C1), and
by the load current.
Charge-Pump Capacitors
Figure 3a shows the charge-pump doubler circuit con-
figured with a 0.1µF charge-pump capacitor C8 and a
1.0µF reservoir capacitor C9. The ratio of the capaci-
tors , a long with the inp ut volta g e , d e te rmine s the
amount of ripple on HIGH. If the input supply range
e xc e e d s 12V, inc re a s e C9 to 4.7µF to re d uc e the
c ha rg e -p ump rip p le . C9 s hould b e 10µF for le s s .
Figure 3b shows the charge-pump tripler circuit.
In the standard 5V application circuit, 2.7nF is appro-
priate for load currents up to 3A; for lighter loads,
C7’s value can be reduced. If 2.7nF does not com-
pensate adequately, use the following equations to
determine C7.
Refer to Table 2 to determine the proper charge-pump
configuration (which is based on the minimum expect-
ed supply voltage at V+).
For fixed 5V-output operation:
(
) (
)
C1 ESR
C1
_____________
Some interaction occurs between the switch oscillator
and the charge-pump oscillator. This interaction modu-
lates the inductor-current waveform, but has negligible
impact on the output.
C7 =
12kΩ
For adjustable-output operation, FB becomes the
compensation input pin, and CC and OUT are left
unconnected. Connect C7 between FB and GND in
parallel with R4 (Figure 6). C7 is determined by:
Soft-Start and Reference Capacitors
Soft-start provides a ramp to the full current limit. A typi-
c a l va lue for the s oft-s ta rt c a p a c itor (C5) is 0.1µF,
which provides a 380ms soft-start time. Use values in
the 0.001µF to 1µF range. The nominal time for C5 to
reach its steady-state value is given by:
(2) (C1) (ESR
)
C1
___________________
C7 =
R4
For example, with a fixed 5V output with C1 = 470µF
and an ESR of 0.04Ω (at a frequency of 100kHz):
R5
6
t
SS
(sec) = (C5) (3.8 x 10 )
C1
Note that t does NOT equal the time it takes for the
SS
MAX746 to power-up, although it does affect the start-
up time. The start-up time is also a function of the input
(
) (
12kΩ
)
C1 ESR
C1
_____________
C7 =
= 1560pF
14 ______________________________________________________________________________________
Hig h -Effic ie n c y, P WM, S t e p -Do w n ,
N-Ch a n n e l DC-DC Co n t ro lle r
MAX746
Inc re a s ing C7 b y up to 50% e nha nc e s oute r-loop
s ta b ility b y a d d ing s ta b ility to the ind uc tor c urre nt
wa ve form. But inc re a s ing C7 too muc h c a us e s
FB’s response time to decrease (due to the larger
RC time constant caused by the feedback resistors
a nd the c omp e ns a tion c a p a c itor), whic h re d uc e s
load-transient stability.
V
IN
V+
AV+
KELVIN SENSE
CONNECTION
R
SENSE
MAX746
S e t t in g t h e Lo w -Ba t t e ry
De t e c t o r Vo lt a g e
CS
Select R1 between 10kΩ and 1MΩ. Determine R2 using
the following equation:
N
EXT
(V
- V
)
TRIP
REF
L1
V
OUT
________________
R2 = R1
(
)
V
REF
where V
is typically 2.0V. Connect a pull-up resistor
REF
(e.g., 100kΩ) between LBO and V
(Figure 5).
OUT
Figure 8. Kelvin Connection for Current-Sense Amplifier
Us in g a S e c o n d S u p p ly in
P la c e o f t h e Ch a rg e P u m p
If a secondary power supply (a minimum of 5V above
the main supply) is available, it can be substituted for
the c ha rg e -p ump hig h-s id e s up p ly. In this c a s e ,
b yp a s s HIGH with a 1µF c a p a c itor a nd le a ve CP
unc onne c te d. Sinc e this se c ond a ry sup ply volta ge
tor, any noise at the CS input will also appear at the
AV+ inp ut, a nd will b e inte rp re te d b y the c urre nt-
sense amplifier as a common-mode signal . A sepa-
rate AV+ capacitor causes the noise to appear on
only one inp ut, a nd this d iffe re ntia l nois e will b e
amplified, adversely affecting circuit operation.
is a p p lie d to the g a te , V
mus t not e xc e e d the
GS
gate-source breakdown voltage of the external N-FET.
Also, the voltage at HIGH must not exceed 20V. If
a secondary supply is used, the shutdown function
cannot be used because HIGH is internally tied to
V+ in shutdown mode. In this case, SHDN must be
tied low. With the main supply off and HIGH at 12V,
HIGH will typically sink 130µA.
Ad d it io n a l No t e s
Whe n p rob ing the MAX746 c irc uit, a void s horting
V+ to GND (the two pins are adjacent) as this may
cause the IC to malfunction because of large ground
currents. Because of its fast switching and high drive-
capability requirements, EXT is a low-impedance point
that is not short-circuit protected. Therefore, do not
short EXT to any node (including AGND and V+, which
are adjacent to EXT).
La yo u t Co n s id e ra t io n s
Because high current levels and fast switching wave-
forms radiate noise, proper PC board layout is essen-
tial. Use a ground plane, and minimize ground noise by
connecting GND, the anode of the steering Schottky
diode, the input bypass-capacitor ground lead, and the
output filter capacitor ground lead to a single point (star
ground configuration). Also minimize lead lengths to
reduce stray capacitance, trace resistance, and radiat-
ed noise. Place bypass capacitor C3 as close to V+
and GND as possible.
Similarly, CC (or FB in adjustable-output operation) is a
sensitive input that should not be shorted to any node.
Avoid shorting CC when probing the circuit, as this may
damage the device.
The MAX746 may continue to operate with AV+ discon-
nected, but erratic switching waveforms will appear at EXT.
Switching Waveforms
There is a region between CCM and DCM where the
ind uc tor c urre nt op e ra te s in b oth mod e s , a s s hown
in the Idle-Mode Moderate Current EXT waveform in
the Typ ic a l Op e ra ting Cha ra c te ris tic s . As the out-
p ut volta g e va rie s , it is fe d b a c k into CC a nd the
d uty c yc le a d jus ts to c omp e ns a te for this c ha ng e .
AV+ and CS are the inputs to the differential-input
c urre nt-s e ns e a mp lifie r. Us e a Ke lvin c onne c tion
a c ros s the s e ns e re s is tor, a s s hown in Fig ure 8.
Although AV+ also functions as the supply voltage
for sensitive analog circuitry, a separate AV+ bypass
capacitor should not be used. By not using a capaci-
The switch is considered off when V
is less than
EXT
______________________________________________________________________________________ 15
Hig h -Effic ie n c y, P WM, S t e p -Do w n ,
N-Ch a n n e l DC-DC Co n t ro lle r
or equal to the N-FET’s V
threshold voltage. Once
AC Stability with Low Input/Output Differentials
At low input/output differentials, the inductor current
c a nnot s le w q uic kly e noug h to re s p ond to loa d
changes, so the output filter capacitor must hold up the
voltage as the load transient is applied. In Figure 1a’s
circuit, for V+ = 6V, increase the output filter capacitor
to 900µF (Sprague 595D low-ESR capacitors) to obtain
a transient response less than 250mV with a load step
from 0.1A to 3A. As V+ increases, the inductor current
slews faster, so the size of the output filter capacitor can
be reduced (see Table 1).
GS
the switch is off, the voltage at EXT is pulled to GND
a nd the N-FET s ourc e volta g e is a Sc hottky d iod e
drop below GND. However, this is not always the case
in the “in-between” mode, due to the changing duty
cycle inherent with DCM. When the device is at maxi-
mum duty cycle, EXT turns off at V , but the switch
GS
sometimes turns on again after the minimum off-time
before EXT can be pulled to GND. This results in short
spikes, which can be seen on the EXT waveform in the
Typical Operating Characteristics .
MAX746
___________________Ch ip To p o g ra p h y
Table 3. Component Suppliers
SUPPLIER
INDUCTORS
PHONE
FAX
LBI LBO GND
V+
Coiltronics
(305) 781-8900
(716) 532-2234
(708) 956-0666
81-3-3607-511
(305) 782-4163
(716) 532-2702
(708) 956-0702
81-3-3607-5428
SS
CP
Gowanda
Sumida USA
Sumida Japan
CAPACITORS
Kemet
HIGH
EXT
(803) 963-6300
(714) 969-2491
(708) 843-7500
(603) 224-1961
(619) 661-6322
81-3-3837-6242
(714) 255-9500
(803) 963-6322
(714) 960-6492
(708) 843-2798
(603) 224-1430
REF
Matsuo
SHDN
0. 130"
(3. 30mm)
Nichicon
Sprague
Sanyo USA
Sanyo Japan
United Chemi-Con
DIODES
(714) 255-9400
(805) 867-2698
AGND
Motorola
(800) 521-6274
(805) 867-2555
Nihon USA
Nihon Japan
POWER TRANSISTORS
Harris
CC
OUT CS
FB
AV+
81-3-3494-7411 81-3-3494-7414
0. 080"
(2. 03mm)
(407) 724-3739
(213) 772-2000
(408) 988-8000
(407) 724-3937
(213) 772-9028
(408) 727-5414
International Rectifier
Siliconix
TRANSISTOR COUNT: 508;
SUBSTRATE CONNECTED TO HIGH.
RESISTORS
IRC
(512) 992-7900
(512) 992-3377
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
16 __________________Ma x im In t e g ra t e d P ro d u c t s , 1 2 0 S a n Ga b rie l Drive , S u n n yva le , CA 9 4 0 8 6 (4 0 8 ) 7 3 7 -7 6 0 0
© 1993 Maxim Integrated Products
Printed USA
is a registered trademark of Maxim Integrated Products.
相关型号：
MAX7472UTI
Switched Capacitor Filter, 1 Func, Lowpass, BICMOS, 4 X 4 MM, 0.80 MM HEIGHT, MO-220WGGE, TQFN-28,
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Switched Capacitor Filter, 1 Func, Lowpass, BICMOS, 4 X 4 MM, 0.80 MM HEIGHT, ROHS COMPLIANT, MO-220WGGE, TQFN-28,
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