MAX8664BEEP+T [MAXIM]
Dual Switching Controller, Voltage-mode, 1000kHz Switching Freq-Max, BICMOS, PDSO20, 0.150 INCH, 0.025 INCH PITCH, LEAD FREE, MO-137AD, QSOP-20;
| 型号： | MAX8664BEEP+T |
| 厂家： | 
MAXIM INTEGRATED PRODUCTS |
| 描述： | Dual Switching Controller, Voltage-mode, 1000kHz Switching Freq-Max, BICMOS, PDSO20, 0.150 INCH, 0.025 INCH PITCH, LEAD FREE, MO-137AD, QSOP-20 控制器 |
| 文件： | 总26页 (文件大小：350K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
19-0796; Rev 0; 4/07
Low-Cost, Dual-Output, Step-Down
Controller with Fast Transient Response
MAX864
General Description
Features
The MAX8664 dual-output PWM controller is a low-cost,
high-performance solution for systems requiring dual
power supplies. It provides two individual outputs that
operate 180° out-of-phase to minimize input current
ripple, and therefore, capacitance requirements. Built-in
drivers are capable of driving external MOSFETs to
deliver up to 25A output current from each channel.
The MAX8664 operates from a 4.5V to 28V input volt-
age source and generates output voltages from 0.6V
up to 90% of the input voltage on each channel. Total
output regulation error is less than 0.8% over load,
line, and temperature.
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The MAX8664 operates with a constant switching fre-
quency adjustable from 100kHz to 1MHz. Built-in boost
diodes reduce external component count. Digital soft-
start eliminates input inrush current during startup. The
second output has an optional external REFIN2, facili-
tating tracking supply applications. Each output is
capable of sourcing and sinking current, making the
device a great solution for DDR applications.
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Ordering Information
PIN-
PꢀCKꢀGE
PKG
C DE
E20-1
FꢀUꢃT
ꢀCTI N
The MAX8664 employs Maxim’s proprietary peak volt-
age-mode control architecture that provides superior
transient response during either load or line transients.
This architecture is easily stabilized using two resistors
and one capacitor for any type of output capacitors. Fast
transient response requires less output capacitance,
consequently reducing total system cost. The MAX8664B
latches off both controllers during a fault condition, while
the MAX8664A allows one controller to continue to func-
tion when there is a fault in the other controller.
PꢀRT
MꢀX.664ꢀEEP+
MꢀX.664BEEP+
20 QSOP
Independent
20 QSOP
E20-1
Joint
NLuv: This device operates over the -40°C to +85°C operating
temperature range.
+Denotes lead-free package.
Typical Operating Circuit
IN2
VL
IN
Applications
ILIM2
ILIM1
IN1
Desktop and Notebook PCs
OUT2
OUT1
Graphic Cards
DH1
DH2
BST2
BST1
ASIC/CPU/DSP Power Supplies
Set-Top Box Power Supply
Printer Power Supply
Network Power Supply
POL Power Supply
LX1
LX2
DL2
DL1
MAX8664
GND
FB2
PGND
FB1
PWRGD
REFIN2
OSC/EN12
V
CC
Pꢅꢄ%CLꢄfꢅgOꢁruꢅLꢄ%rttvrꢁs%ru%vꢄa%Lf%arur%shvvu0
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1
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Low-Cost, Dual-Output, Step-Down
Controller with Fast Transient Response
ꢀBS ꢃUTE%MꢀXIMUM%RꢀTINGS
IN to GND ...........................................................…-0.3V to +30V
VL to GND...................................................................-0.3 to +8V
IN, BST_ to VL ........................................................-0.3V to +30V
ILIM_ to GND...............................................-0.3V to (V + 0.3V)
ILIM_ to LX_............................................................-0.6V to +30V
IN
OSC/EN12, REFIN2 to GND.....................-0.3V to (V
+ 0.3V)
VCC
V
, FB_, PWRGD to GND.......................................-0.3V to +6V
VL Continuous Current ..............................................125mA
CC
RMS
RMS
VL to V
....................................................................-2V to +8V
V
CC
Continuous Current..............................................10mA
CC
PGND to GND .......................................................-0.3V to +0.6V
DL_ to PGND...............................................-0.3V to (V + 0.3V)
Continuous Power Dissipation (T = +70°C) (Note 1)
A
20-Pin QSOP (derate 11.0mW/°C above +70°C).........884mW
Operating Temperature Range ...........................-40°C to +85°C
Junction Temperature......................................................+150°C
Storage Temperature Range.............................-65°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
VL
BST_
DH_ to PGND............................................-0.3V to (V
+ 0.3V)
BST_ to GND.............................................................-0.3V to 38V
BST_ to LX................................................................-0.3V to +8V
LX_ to PGND .................-1V (-2.5V for < 50ns transient) to +30V
MAX864
DH_ to LX_................................................-0.3V to (V
+ 0.3V)
BST_
NLuv%1: Package mounted on a multilayer PCB.
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
EꢃECTRICꢀꢃ%CHꢀRꢀCTERISTICS
(V = 12V, R
IN
to GND = 56.1kΩ, REFIN2 = V , T = -40°C to +85°C, unless otherwise noted. Typical values are at
CC A
OSC/EN12
T
A
= +25°C.) (Note 2)
PꢀRꢀMETER
C NDITI NS
MIN
TYP
MꢀX
UNITS
SUPPꢃY%V ꢃTꢀGES
IN Supply Voltage
VL Output Voltage
7.2
4.5
28.0
5.5
V
IN = VL = V
CC
7.2V < V < 28V, 0 < I < 60mA
6.10
4.5
6.6
5.0
6.75
5.5
V
V
IN
VL
V
Output Voltage
7.2V < V < 28V, 0 < I
IN
< 5mA
CC
CC
Rising
3.4
3.5
3.6
V
V
Undervoltage Lockout
CC
(UVLO)
Hysteresis
350
0.095
0.08
1.4
mV
V
V
V
V
= 12V, I
IN
0.2
0.2
2.5
1.8
OSC/EN12 not
connected
IN
Standby Supply Current
mA
mA
= V = V = 5V, I + I + I
VCC
CC
IN
VL
IN
VL
= 12V, I
IN
No switching,
IN
Operating Supply Current
V
= 0.65V
FB_
= V = V = 5V, I + I + I
VL VCC
1.1
CC
IN
VL
IN
REGUꢃꢀT R%SPECIFICꢀTI NS
T
T
= 0°C to +85°C
0.5955 0.600 0.6045
A
Reference Accuracy
V
= -40°C to +85°C
0.5930 0.600 0.6070
0.5952 0.600 0.6048
0.5925 0.600 0.6075
A
T
T
= 0°C to +85°C
A
V
V
= V
VCC
REFIN2
REFIN2
= -40°C to +85°C
FB_ Regulation Accuracy
V
V
A
= 1.000V
0.995
1.000
1.005
REFIN2 to Internal Reference
Switchover Threshold
V
-
V
-
VCC
0.7
VCC
0.3
Not to be switched during operation
2
REFIN2 Maximum Program Voltage
REFIN2 Disable Threshold
FB Input Bias Current
1.3
50
3
V
mV
nA
nA
ns
V
V
= 0.5V
FB
REFIN2 Bias Current
= 0.65V
3
REFIN2
FB Propagation Delay
FB rising to DH falling
90
2
_______________________________________________________________________________________
Low-Cost, Dual-Output, Step-Down
Controller with Fast Transient Response
MAX864
EꢃECTRICꢀꢃ%CHꢀRꢀCTERISTICS%(ALꢄuꢅꢄOva)
(V = 12V, R
IN
to GND = 56.1kΩ, REFIN2 = V , T = -40°C to +85°C, unless otherwise noted. Typical values are at
CC A
OSC/EN12
T
A
= +25°C.) (Note 2)
PꢀRꢀMETER
C NDITI NS
MIN
TYP
MꢀX
UNITS
PR TECTI N%FEꢀTURES
V
V
rising
0.75
FB1
Overvoltage Protection (OVP)
Threshold
V
REFIN2
+ 0.15
rising, V
≤ 1.3V
REFIN2
FB2
V
V
= V
, V
rising, MAX8664B
REFIN2
VCC FB_
0.500
44
0.525
0.550
V
Power-Good (PWRGD) Threshold
rising, MAX8664A
FB1
Hysteresis
5
%
T
A
T
A
T
A
T
A
= +85oC
= +25oC
= +25°C
= +85°C
60
50
0.1
0.1
High-Side Current-Sense Program
Current (Note 3)
µA
60
1.0
ILIM Leakage
µA
V
High-Side Current-Sense
Overcurrent Trip Adjustment Range
0.05
0.40
20
Internal Soft-Start Time
R
= 56.1kΩ, 400kHz
2.5
10
ms
Ω
OSC/EN12
REFIN2 Internal Pulldown Resistance Engaged momentarily at startup
Thermal-Shutdown Threshold
Junction temperature
+160
°C
DRIVER%SPECIFICꢀTI NS
V
V
V
V
V
V
V
V
V
V
= 6.5V
1.35
1.55
0.9
1.0
1.3
1.5
0.6
0.7
25
2.1
1.4
2
VL
IN
Sourcing current,
I
= -50mA
DH
= V = V
VL
= 5V
= 5V
= 5V
= 5V
VCC
VCC
VCC
VCC
DH_ Driver Resistance
DL_ Driver Resistance
Ω
= 6.5V
VL
IN
Sinking current,
= 50mA
I
DH
= V = V
VL
= 6.5V
VL
IN
Sourcing current,
= -50mA
I
DL
= V = V
VL
Ω
= 6.5V
1.1
43
VL
IN
Sinking current,
= 50mA
I
DL
= V = V
VL
= 6.5V
= 5V
13
70
VL
VL
Dead Time for Low-Side to
High-Side Transition
DL_ falling to DH_ rising
ns
28
DH_ Minimum On-Time
108
1.25
0.001
6
149
2.3
ns
mA
µA
Ω
V
- V = 7V, V = 28V, V
= 0.55V
BST
LX
LX
FB_
BST Current
OSC/EN12 not connected
Internal Boost Switch Resistance
PWM%Cꢃ CK% SCIꢃꢃꢀT R
PWM Clock-Frequency Accuracy
-15
+15
1000
2.5
%
kHz
µA
PWM Clock-Frequency Adjustment
Range
R
= 226kΩ to 22.6kΩ
100
OSC/EN12
OSC/EN12 Disable Current
1.5
NLuv%2: Specifications at -40°C are guaranteed by design and not production tested.
NLuv%3: This current linearly compensates for the MOSFET temperature coefficient.
_______________________________________________________________________________________
3
Low-Cost, Dual-Output, Step-Down
Controller with Fast Transient Response
Typical Operating Characteristics
(Circuit of Figure 2, 600kHz, V = 12V, V
IN
= 2.5V, V
= 1.8V, T = +25°C, unless otherwise noted.)
OUT2 A
OUT1
EFFICIENCY vs. LOAD CURRENT
EFFICIENCY vs. LOAD CURRENT
(1MHz, FIGURE 4)
LOAD REGULATION
(600kHz, FIGURE 2)
(600kHz, FIGURE 2)
100
100
90
80
70
60
50
40
30
20
10
0
2.55
2.54
2.53
2.52
2.51
2.50
2.49
2.48
2.47
2.46
2.45
90
80
70
60
V
= 2.5V
OUT1
MAX864
I
= 8A
OUT2
I
= 4A
OUT2
V
= 1.8V
OUT1
V
= 2.5V
OUT
50
40
30
20
10
0
V
= 1.8V
OUT
I
= 0A
OUT2
V
V
= 3.3V
= 5V
IN
VL
NO LOAD ON THE
OTHER REGULATOR
NO LOAD ON OUT2
0.1
1
10
0.1
1
10
0
2
4
6
8
10
LOAD CURRENT (A)
LOAD CURRENT (A)
OUT1 LOAD CURRENT (A)
LINE REGULATION
(600kHz, FIGURE 2)
R
vs. SWITCHING FRQUENCY
OUT1 LOAD TRANSIENT (FIGURE 2)
MAX8664 toc06
OSC/EN12
250
200
150
100
50
2.55
2.54
2.53
2.52
2.51
2.50
2.49
2.48
2.47
2.46
2.45
8A LOAD
V
I
100mV/div
2A/div
OUT2
OUT2
5A
2.5A
2.5A
NO LOAD
0
100
400
700
1000
20μs/div
6
8
10
12
14
16
18
20
SWITCHING FREQUENCY (kHz)
INPUT VOLTAGE (V)
LOAD TRANSIENT
-3A TO +3A TO -3A (FIGURE 3)
POWER-UP WAVEFORMS
MAX8664 toc07
MAX8664 toc08
10V/div
50mV/div
V
V
I
OUT1
OUT2
OUT2
V
IN
2V/div
2V/div
V
V
50mV/div
5A/div
OUT1
OUT2
+3A
5V/div
-3A
-3A
V
PRWGD
100μs/div
1ms/div
4
_______________________________________________________________________________________
Low-Cost, Dual-Output, Step-Down
Controller with Fast Transient Response
MAX864
Typical Operating Characteristics (continued)
(Circuit of Figure 2, 600kHz, V = 12V, V
= 2.5V, V = 1.8V, T = +25°C, unless otherwise noted.)
OUT2 A
IN
OUT1
POWER-DOWN WAVEFORMS
ENABLE WAVEFORMS (FIGURE 2)
MAX8664 toc09
MAX8664 toc10
V
IN
10V/div
5V/div
ENABLE
V
V
OUT1
OUT2
V
V
OUT1
OUT2
2V/div
2V/div
2V/div
2V/div
V
PRWGD
V
PRWGD
5V/div
5V/div
1ms/div
1ms/div
FEEDBACK VOLTAGE
vs. TEMPERATURE
ENABLE WAVEFORMS (FIGURE 4)
SWITCHING WAVEFORMS
MAX8664 toc12
MAX8664 toc11
605
604
603
602
601
600
599
598
597
596
595
ENABLE
5V/div
V
V
10V/div
5A/div
LX1
I
L1
V
OUT1
1V/div
1V/div
10V/div
5A/div
V
OUT2
LX2
V
I
L2
PRWGD
5V/div
NO LOAD
60 80 100
400μs/div
2μs/div
-40 -20
0
20
40
TEMPERATURE (°C)
SHORT-CIRCUIT WAVEFORMS
OVERVOLTAGE PROTECTION
MAX8664 toc15
MAX8664 toc14
V
OUT1
V
5V/div
OUT1
2V/div
2A/div
I
I
L1
IN
10A/div
I
L1
V
DH1
10V/div
10V/div
5A/div
5V/div
V
DL1
V
PRWGD
10μs/div
20μs/div
_______________________________________________________________________________________
5
Low-Cost, Dual-Output, Step-Down
Controller with Fast Transient Response
Pin Description
PIN
NꢀME
FUNCTI N
High-Side MOSFET Driver Output for Controller 1. Connect DH1 to the gate of the high-side MOSFET. DH1 is
low in shutdown and UVLO.
1
DH1
External Inductor Connection for Controller 1. Connect LX1 to the switching node of the MOSFETs and
inductor. Make sure LX1 is close to the source of the high-side MOSFET(s) to form a Kelvin connection for
high-side current sensing. LX1 is high impedance during monotonic startup and shutdown.
2
LX1
MAX864
Boost Capacitor Connection for the High-Side MOSFET Driver for Controller 1. Connect a 0.22µF ceramic
capacitor from BST1 to LX1.
3
4
BST1
DL1
Low-Side MOSFET Driver Output for Controller 1. Connect DL1 to the gate of the low-side MOSFET(s) for
controller 1. DL1 is low in shutdown and UVLO.
Low-Side Gate Drive Supply and Output of the 6.5V Linear Regulator. Connect a 4.7µF ceramic capacitor from
5
VL
VL to PGND. When using a 4.5V to 5.5V supply, connect VL to IN. VL is the input to the V
load VL when IC is disabled.
supply. Do not
CC
Power Ground. Connect to the power ground plane. Connect power and analog grounds at a single point near
the output capacitor’s ground.
6
7
8
PGND
DL2
Low-Side MOSFET Driver Output for Controller 2. Connect DL2 to the gate of the low-side MOSFET(s) for
controller 2. DL2 is low in shutdown and UVLO.
Boost Capacitor Connection for the High-Side MOSFET Driver for Controller 2. Connect a 0.22µF ceramic
capacitor from BST2 to LX2.
BST2
External Inductor Connection for Controller 2. Connect LX2 to the switching node of the MOSFETs and
inductor. Make sure LX2 is close to the source of the high-side MOSFET(s) to form a Kelvin connection for
high-side current sensing. LX2 is high impedance during monotonic startup and shutdown.
9
LX2
High-Side MOSFET Driver Output for Controller 2. Connect DH2 to the gate of the high-side MOSFET(s) for
controller 2. DH2 is low in shutdown and UVLO.
10
11
DH2
Current-Limit Set for Controller 2. Connect a resistor from the drain of the high-side MOSFET(s) to ILIM2. See
the Setting the Overcurrent Threshold section.
ILIM2
Feedback Input for Controller 2. Connect FB2 to the center of a resistor-divider connected between the output
12
13
FB2
of controller 2 and GND to set the desired output voltage. V
reference. To use the internal reference, connect REFIN2 to V
regulates to V
or the internal 0.6V
FB2
REFIN2
.
CC
External Reference Input for Controller 2. To use the internal 0.6V reference, connect REFIN2 to V . To use
CC
an external reference, connect REFIN2 through a resistor (> 1kΩ) to a reference voltage between 0V and
1.3V. An RC lowpass filter is recommended when using an external reference and soft-start is not provided by
the external reference. For tracking applications, connect REFIN2 to the center of a resistor voltage-divider
between the output of controller 1 and GND (see Figure 3). Connect REFIN2 to GND to disable controller 2.
REFIN2
Switching Frequency Set Input. Connect a 22.6kΩ to 226kΩ resistor from OSC/EN12 to GND to set the
switching frequency between 1000kHz and 100kHz. Connect a switch in series with this resistor for
enable/shutdown control. When the switch is open, the IC enters low-power shutdown mode. In shutdown,
OSC/EN12 is internally driven to approximately 800mV.
14
OSC/EN12
Internal 6.5V Linear Regulator Input. Connect IN to a 7.2V to 28V supply, and connect a 0.47µF or larger
ceramic capacitor from IN to PGND. When using a 4.5V to 5.5V supply, connect IN to VL.
15
16
IN
Analog Ground. Connect to the analog ground plane. Connect the analog and power ground planes at a
single point near the output capacitor’s ground.
GND
6
_______________________________________________________________________________________
Low-Cost, Dual-Output, Step-Down
Controller with Fast Transient Response
MAX864
Pin Description (continued)
PIN
NꢀME
FUNCTI N
Internal Analog Supply. V
regulates to 1.5V below V
Connect a 1µF ceramic capacitor from V
to GND.
CC
VL.
CC
17
V
When using a 4.5V to 5.5V supply, connect a 10Ω resistor from V
to IN. V
is used to power the IC’s
CC
CC
CC
internal circuitry.
Open-Drain Power-Good Output. PWRGD is high impedance when controllers 1 and 2 (using the internal
reference) are in regulation. PWRGD is low if the outputs are out of regulation, if there is a fault condition, or if
the IC is shut down. PWRGD does not reflect the status of output 2 in the MAX8664A or when REFIN2 is
connected to an external reference in the MAX8664B.
18
PWRGD
Feedback Input for Controller 1. Connect FB1 to the center of a resistor-divider connected between the output
19
20
FB1
of controller 1 and GND to set the desired output voltage. V
regulates to 0.6V.
FB1
Current-Limit Set for Controller 1. Connect a resistor from the drain of the high-side MOSFET(s) to ILIM1. See
the Setting the Overcurrent Threshold section.
ILIM1
_______________________________________________________________________________________
7
Low-Cost, Dual-Output, Step-Down
Controller with Fast Transient Response
V
CC
CURRENT-LIMIT
COMPARATOR
ILIM1
BST1
UVLO
CIRCUITRY
BIAS
GENERATOR
50μA
BST CAP
CHARGING SWITCH
LX1
THERMAL
EN SHUTDOWN
VOLTAGE
REFERENCE
MAX864
REF
REF
DH1
LX1
EN
SHUTDOWN
CONTROL
LOGIC
SOFT-START 1
CONTROL
LOGIC
SHUTDOWN 1
SHUTDOWN 2
DL1
CLOCK 1 SHUTDOWN 1
PGND
VL
FB1
PWM
COMPARATOR 1
CURRENT-LIMIT
COMPARATOR
ILIM2
0.6V
BST2
50μA
BST CAP
CHARGING SWITCH
LX2
S2
PWM
COMPARATOR 2
FB2
DH2
LX2
REF2
S1
CONTROL
LOGIC
ENABLE2
REFIN2
DL2
SHUTDOWN 2
CLOCK 2
50mV
SOFT-START
CLOCK 1
IF V
> 2.0V
REFIN2
OPEN S1 AND CLOSE S2.
OTHERWISE, CLOSE S1
AND OPEN S2.
OSC/EN12
OSCILLATOR
CLOCK 2
ENABLE
FB1
THERMAL
SHUTDOWN
THERMAL
SHUTDOWN
4μA
REF
PWRGD
REF1 - 0.1V
IN
VL
6.5V LDO
FB2
REF2 - 0.1V
1.5V
GND
V
CC
Figure 1. Functional Diagram
_______________________________________________________________________________________
.
Low-Cost, Dual-Output, Step-Down
Controller with Fast Transient Response
MAX864
Internal Linear Regulators
Detailed Description
The internal VL low-dropout linear regulator of the
The MAX8664 dual-output PWM controller is a low-cost
MAX8664A and MAX8664B provides the 6.5V supply
used for the gate drive. Connect a 4.7µF ceramic
capacitor from VL to PGND. When using a 4.5V to 5.5V
input supply, connect VL directly to IN.
solution for dual power-supply systems. It provides two
individual outputs that operate 180° out-of-phase to
minimize input capacitance requirements. Built-in dri-
vers are capable of driving external MOSFETs to deliv-
er up to 25A of current from each output. The MAX8664
operates from a 4.5V to a 5.5V or a 7.2V to 28V input
and generates output voltages from 0.6V up to 90% of
the input voltage on each channel. Total output error is
less than 0.8% over load, line, and temperature.
The 5V supply used to power IC functions (V ) is gen-
CC
erated by an internal 1.5V shunt regulator from VL.
Connect a 2.2µF ceramic capacitor from V
to GND.
CC
When using a 4.5V to 5.5V input supply, connect V
CC
to IN through a 10Ω resistor.
The MAX8664 operates with a constant switching fre-
quency adjustable from 100kHz to 1MHz. Built-in boost
diodes reduce external component count. Digital soft-
start eliminates input inrush current during startup. The
second output has an optional REFIN2 input that takes
an external reference voltage, facilitating tracking supply
applications. Each output is capable of sourcing and
sinking current. Internal 6.5V and 5V linear regulators
provide power for gate drive and internal IC functions.
The MAX8664 has built-in protection against output over-
voltage, overcurrent, and thermal faults. The MAX8664B
latches off both controllers during a fault condition, while
the MAX8664A allows one controller to continue to func-
tion when there is a fault in the other controller.
High-Side Gate-Drive Supply (BST_)
The gate-drive voltage for the high-side MOSFETs is
generated using a flying capacitor boost circuit. The
capacitor between BST_ and LX_ is charged to the VL
voltage through the integrated BST_ diode during the
low-side MOSFET on-time. When the low-side MOSFET
is switched off, the BST_ voltage is shifted above the
LX_ voltage to provide the necessary turn-on voltage
(V ) for the high-side MOSFET. The controller closes
GS
a switch between BST_ and DH_ to turn the high-side
MOSFET on.
Voltage Reference
An internal 0.6V reference sets the feedback regulation
voltage. Controller 1 always uses the internal reference.
An external reference input is provided for controller 2.
To use the external reference, connect a 0 to 1.3V sup-
ply to REFIN2. This facilitates tracking applications. To
use the internal 0.6V reference for controller 2, connect
The MAX8664 employs Maxim’s proprietary peak-volt-
age mode control architecture that provides superior
transient response during either load or line transients.
This architecture is easily stabilized using two resistors
and one capacitor for any type of output capacitors.
Fast transient response requires less output capaci-
tance, consequently reducing total system cost.
REFIN2 to V
.
CC
Undervoltage Lockout (UVLO)
supply voltage drops below the UVLO
When the V
CC
DC-DC Controller Architecture
The peak-voltage mode PWM control scheme ensures
stable operation, simple compensation for any output
capacitor, and fast transient response. An on-chip inte-
grator removes any DC error due to the ripple voltage.
This control scheme is simple: when the output voltage
falls below the regulation threshold, the error compara-
tor begins a switching cycle by turning on the high-side
switch at the rising edge of the following clock cycle.
This switch remains on until the minimum on-time
expires and the output voltage is in regulation or the
current-limit threshold is exceeded. At this point, the
low-side synchronous rectifier turns on and remains on
until the rising edge of the first clock cycle after the out-
put voltage falls below the regulation threshold.
threshold (3.15V falling typ), the undervoltage lockout
(UVLO) circuitry inhibits the switching of both con-
trollers, and forces the DL and DH gate drivers low.
When V
rises above the UVLO threshold (3.5V rising
CC
typ), the controllers begin the startup sequence and
resume normal operation.
Output Overcurrent Protection
When the MAX8664 detects an overcurrent condition,
DH is immediately pulled low. If the overcurrent condition
persists for four consecutive cycles, the controller latch-
es off and both DH_ and DL_ are pulled low. During soft-
start, when FB_ is less than 300mV, the controller latches
off on the first overcurrent condition. The protection cir-
cuit detects an overcurrent condition by sensing the
drain-source voltage across the high-side MOSFET(s).
_______________________________________________________________________________________
9
Low-Cost, Dual-Output, Step-Down
Controller with Fast Transient Response
The threshold that trips overcurrent protection is set by a
resistor connected from ILIM_ to the drain of the high-
side MOSFET(s). ILIM_ sinks 50µA (typ) through this
resistor. When the drain-source voltage exceeds the volt-
age drop across this resistor during the high-side
MOSFET(s) on-time, an overcurrent fault is triggered. To
prevent glitches from falsely tripping the overcurrent pro-
tection, connect a filter capacitor (0.01µF typically) in
parallel with the overcurrent-setting resistor.
Power-Good Output (PWRGD)
PWRGD is an open-drain output that is pulled low when
the output voltage rises above the PWRGD upper
threshold or falls below the PWRGD falling threshold.
PWRGD is held low in shutdown, when V
is below the
CC
UVLO threshold, during soft-start, and during fault con-
ditions. PWRGD does not reflect the status of controller
2 in the MAX8664A, or when REFIN2 is connected to an
external reference with either version. See Table 1 for
PWRGD operation of the circuits of Figures 2–5 during
fault conditions. For logic-level output voltages, con-
nect an external pullup resistor between PWRGD and
the logic power supply. A 100kΩ resistor works well in
most applications.
MAX864
Output Overvoltage Protection (OVP)
During an overvoltage event on one or both of its out-
puts, the MAX8664 latches off the controller. This
occurs when the feedback voltage exceeds its normal
regulation voltage by 150mV for 10µs. In this state, the
low-side MOSFET(s) are on and the high-side MOS-
FET(s) are off to discharge the output. To clear the
latch, cycle EN or the input power.
Fault-Shutdown Modes
When an overvoltage or overcurrent fault occurs on one
controller of the MAX8664A, the second controller con-
tinues to operate. With the MAX8664B, a fault in one
controller latches off both controllers automatically, and
PWRGD is pulled low. See Table 1 for the fault-shut-
down modes of the circuits shown in Figures 2–5.
Thermal-Overload Protection
Thermal-overload protection limits total power dissipa-
tion in the MAX8664. When the junction temperature
exceeds +160°C, an internal thermal sensor shuts down
the device, pulling DH_ and DL_ low for both controllers.
To restart the controller, cycle EN or input power.
Trblv 10%FrOlu%ShOuaLwꢄ%MLavs%fLꢁ%CꢅꢁAOꢅus%Lf%FꢅgOꢁvs%2–5
MꢀX.664ꢀ%(INDEPENDENT)
CIRCUIT
MꢀX.664B%(J INT)
C NTR ꢃꢃER%1%FꢀUꢃT C NTR ꢃꢃER%2%FꢀUꢃT C NTR ꢃꢃER%1%FꢀUꢃT
C NTR ꢃꢃER%2%FꢀUꢃT
Figure 2,
Figure 5
(Independent)
Controller 2 remains on.
PWRGD is pulled low.
Controller 1 remains on.
PWRGD remains high.
Controller 2 is shut down.
PWRGD is pulled low.
Controller 1 is shut down.
PWRGD is pulled low.
Figure 3
(Tracking)
Controller 2 shuts down.
PWRGD is pulled low.
Controller 1 remains on.
PWRGD remains high.
Controller 2 is shut down.
PWRGD is pulled low.
Controller 1 is shut down.
PWRGD is pulled low.
Figure 4
(Sequenced)
Controller 2 shuts down.
PWRGD is pulled low.
Controller 1 remains on.
PWRGD remains high.
Controller 2 is shut down.
PWRGD is pulled low.
Controller 1 is shut down.
PWRGD is pulled low.
1± ______________________________________________________________________________________
Low-Cost, Dual-Output, Step-Down
Controller with Fast Transient Response
MAX864
C19
0.01μF
C20
10μF
C1
10μF
C4
1000μF
R1
2.7kΩ
INPUT
10.8V TO 13.2V
ILIM1
DH1
FB1
IN
N1
N2
C5
1500pF
R3
51.1kΩ
C17
1μF
R4
R5
3.92kΩ
1.15kΩ
OUT1
LX1
V
CC
2.5V/8A
C18
1μF
C13
0.22μF
REFIN2
VL
L1
1μH
C6
47μF
C7
47μF
C8
47μF
C23
0.1μF
BST1
DL1
R37
3Ω
C14
4.7μF
C25
680pF
MAX8664
PGND
GND
C16
0.01μF
VCC
C21
10μF
C3
10μF
R9
10kΩ
ILIM2
POWER-GOOD
TO SYSTEM
R2
3.01kΩ
N3
PWRGD
DH2
LX2
R10
39.2kΩ
L2
1μH
OUT2
1.8V/8A
OSC/EN12
C15
0.22μF
ENABLE
ON
OFF
N9
2N7002
C12
1500μF
C9
47μF
C10
47μF
C11
47μF
C22
0.1μF
R6
51.1kΩ
BST2
DL2
FB2
R7
3.92kΩ
R38
3Ω
N4
C27
0.47μF
C26
680pF
R8
1.82kΩ
Figure 2. Low-Cost, 600kHz Typical Application Circuit
______________________________________________________________________________________ 11
Low-Cost, Dual-Output, Step-Down
Controller with Fast Transient Response
Trblv 20%CLmtLꢄvꢄu%ꢃꢅsu%fLꢁ%FꢅgOꢁv 2
DESIGNꢀTI N QTY
DESCRIPTI N
DESIGNꢀTI N QTY
DESCRIPTI N
680pF, 50V C0G ceramic capacitors
(0603)
C1, C3,
4
10µF 20%, 16V X5R ceramic
capacitors (1206)
C25, C26
C27
2
1
2
4
1
C20, C21
0.47µF 10%, 16V ceramic
capacitor (0603)
1000µF 20%, 16V electrolytic
capacitor (8mm diameter,
20mm height)
C4
1
1µH inductors
TOKO FDV0630-1R0M
L1, L2
N1–N4
N9
MAX864
1500pF, 50V C0G ceramic
capacitors (0603)
C5, C12
C6–C11
C13, C15
C14
2
6
2
1
2
1
1
2
n-channel MOSFETs (8-pin SO)
International Rectifier IRF7821
47µF 20%, 6.3V X5R ceramic
capacitors (1206)
n-channel MOSFET (SOT23)
Central 2N7002
0.22µF 10%, 25V X7R ceramic
capacitors (0603)
R1
R2
1
1
2
2
1
1
1
1
2
1
2.74kΩ 1% resistor (0603)
301kΩ 1% resistor (0603)
51.1kΩ 1% resistors (0603)
3.92kΩ 1% resistors (0603)
1.15kΩ 1% resistor (0603)
1.82kΩ 1% resistor (0603)
10kΩ 5% resistor (0603)
39.2kΩ 1% resistor (0603)
3Ω 5% resistors (0805)
4.7µF 10%, 6.3V X5R ceramic
capacitor (0805)
R3, R6
R4, R7
R5
0.01µF 10%, 50V X7R ceramic
capacitors (0603)
C16, C19
C17
1µF 20%, 16V X5R ceramic
capacitor (0603)
R8
R9
1µF 20%, 6.3V X5R ceramic
capacitor (0603)
R10
C18
R37, R38
U1
0.1µF 20%, 16V X7R ceramic
capacitors (0603)
MAX8664 (20-pin QSOP)
C22, C23
12 ______________________________________________________________________________________
Low-Cost, Dual-Output, Step-Down
Controller with Fast Transient Response
MAX864
C1
0.01μF
C2
C3
C4
10μF
10μF
1000μF
R1
3.16kΩ
ILIM1
DH1
FB1
IN
INPUT
10V TO 14V
N1
N2
R3
10kΩ
R4
3.57kΩ
C8
0.015μF
C5
1μF
R2
24.3kΩ
V
CC
C6
1μF
LX1
OUT1
1.8V/20A
L1
0.56μH
R6
1kΩ
OUT1
C7
0.22μF
REFIN2
VL
C11
10μF
C9
470μF
C10
470μF
R5
3Ω
BST1
C12
1000pF
C13
4.7μF
R7
1kΩ
N3
N4
C14
2200pF
MAX8664 DL1
GND
VCC
PGND
C15
0.01μF
R9
10kΩ
C16
10μF
C17
10μF
R8
2.74kΩ
POWER-GOOD
TO SYSTEM
PWRGD
ILIM2
DH2
R10
44.2kΩ
N5
OSC/EN12
L2
0.47μH
OUT2
0.9V/6A
LX2
ENABLE
ON
N7
2N7002
C18
0.22μF
C19
4700pF
C20
680μF
C21
680μF
C22
10μF
R11
14.7kΩ
R12
2Ω
OFF
BST2
DL2
N6
C23
2200pF
FB2
R13
10kΩ
Figure 3. 500kHz Tracking Circuit for DDR2 Applications
______________________________________________________________________________________ 13
Low-Cost, Dual-Output, Step-Down
Controller with Fast Transient Response
Trblv 30%CLmtLꢄvꢄu%ꢃꢅsu%fLꢁ%FꢅgOꢁv 3
DESIGNꢀTI N
QTY
DESCRIPTI N
DESIGNꢀTI N
QTY
DESCRIPTI N
0.47µH, 1.2mΩ inductor
TOKO FDV0603-R47M
0.01µF, 10V X7R ceramic
capacitors
L2
1
C1, C15
2
4
n-channel MOSFETs
IRLR7821 (D-Pak)
C2, C3, C16, C17
10µF, 16V X5R ceramic capacitors
N1, N2
N3, N4
N5
2
2
1
1
1
1000µF/16V aluminum electrolytic
capacitor
Rubycon 16MBZ1000M
n-channel MOSFETs
IRLR3907Z (D-Pak)
C4
1
MAX864
n-channel MOSFET
IRF7807Z (8-pin SO)
C5
C6
1
1
1µF, 16V X5R ceramic capacitor
1µF, 10V X5R ceramic capacitor
n-channel MOSFET
IRF7821 (8-pin SO)
N6
0.22µF, 10V X7R ceramic
capacitors
C7, C18
C8
2
1
n-channel MOSFET
2N7002 (SOT23)
N7
0.015µF, 10V X7R ceramic
capacitor
R1
R2
1
1
2
1
1
2
1
1
1
1
1
3.16kΩ 1% resistor (0402 or 0603)
24.3kΩ 1% resistor (0402 or 0603)
10kΩ 1% resistors (0402 or 0603)
3.57kΩ 5% resistor (0402 or 0603)
3.0Ω 5% resistor (0603)
470µF, 2.5V POS capacitors
Sanyo 2R5TPD470M6
C9, C10
C11, C22
C12
2
2
1
R3, R13
R4
10µF, 6.3V X5R ceramic capacitors
1000pF, 10V X7R ceramic
capacitor
R5
R6, R7
R8
1kΩ 1% resistors (0402 or 0603)
2.74kΩ 1% resistor (0402 or 0603)
10kΩ 5% resistor (0402 or 0603)
44.2kΩ 1% resistor (0402 or 0603)
14.7kΩ 1% resistor (0402 or 0603)
2.0Ω 5% resistor (0402 or 0603)
C13
C14, C23
C19
1
2
1
4.7µF, 10V X5R ceramic capacitor
2200pF, 25V X7R capacitors
4700pF, 10V X7R capacitor
R9
R10
R11
R12
680µF, 2.5V POS capacitors
Sanyo 2R5TPD680M6
C20, C21
L1
2
1
0.56µH, 4.6mΩ inductor
Panasonic ETQP4LR56WFL
14 ______________________________________________________________________________________
Low-Cost, Dual-Output, Step-Down
Controller with Fast Transient Response
MAX864
C1
0.01μF
C2
1μF
C3
10μF
C4
10μF
R1
3.32kΩ
N1
IRF7821
ILIM1
DH1
LX1
FB1
5V
IN
C8
820pF
R2
17.4kΩ
C5
1μF
VL
R3
R4
C6
4.7μF
10kΩ
3.16kΩ
L1
OUT1
1.8V/10A
R5
10Ω
0.6V
EXT REF
0.2μH
C7
0.22μF
V
MAX8664
C9
47μF
C10
47μF
C11
0.1μF
CC
C12
1μF
BST1
N2
VCC
R6
2Ω
R7
10kΩ
IRF7821
DL1
REFIN2
GND
C13
2200pF
C14
0.01μF
R8
10kΩ
PGND
C15
0.01μF
N5
INPUT
2.97V TO 3.63V
2N7002
ILIM2
Q1
CMST3904
VCC
R9
47kΩ
C16
1μF
C17
10μF
C18
10μF
R11
3.32kΩ
R10
10kΩ
DH2
LX2
POWER-GOOD
TO SYSTEM
N3
IRF7821
PWRGD
R12
L2
0.2μH
OUT2
1.2V/10A
22.6kΩ
OSC/EN12
BST2
DL2
C19
0.22μF
C21
820pF
C22
47μF
C23
47μF
C24
0.1μF
R14
R13
2Ω
17.4kΩ
FB2
N4
IRF7821
R15
10kΩ
C20
2200pF
R16
6.34kΩ
Figure 4. 1MHz Application Circuit with All Ceramic Capacitors and Sequenced Outputs
______________________________________________________________________________________ 15
Low-Cost, Dual-Output, Step-Down
Controller with Fast Transient Response
Trblv 40%CLmtLꢄvꢄu%ꢃꢅsu%fLꢁ%FꢅgOꢁv 4
DESIGNꢀTI N
QTY
DESCRIPTI N
DESIGNꢀTI N
C1, C14, C15
C2, C16
QTY
DESCRIPTI N
0.2µH, 2.4mΩ inductors
TOKO FDV0603-R20M
0.01µF, 10V X7R ceramic
capacitors
L1, L2
2
2
2
4
1µF, 6.3V X5R ceramic capacitors
n-channel MOSFETs
IRF7821 (8-pin SO)
N1–N4
N5
4
1
1
10µF, 6.3V X5R ceramic
capacitors
C3, C4, C17, C18
n-channel MOSFET
2N7002 (SOT23)
MAX864
C5, C12
C6
2
1
1µF, 10V X5R ceramic capacitors
4.7µF, 10V X5R ceramic capacitor
Transistor, bipolar, npn
Central CMST3904
Q1
0.22µF, 10V X7R ceramic
capacitors
C7, C19
C8, C21
2
2
4
2
2
R1, R11
R2, R14
R3, R15
R4
2
2
2
1
1
2
3
1
1
1
3.32kΩ 1% resistors (0402 or 0603)
17.4kΩ 1% resistors (0402 or 0603)
10kΩ 1% resistors (0402 or 0603)
3.16kΩ 1% resistor (0402 or 0603)
10.0Ω 5% resistor (0402 or 0603)
2.0Ω 5% resistors (0603)
820pF,10V X7R ceramic
capacitors
47µF, 6.3V X5R ceramic
capacitors
C9, C10, C22, C23
C11, C24
R5
R6, R13
R7, R8, R10
R9
0.1µF, 10V X7R ceramic
capacitors
10kΩ 5% resistors (0402 or 0603)
47kΩ 5% resistor (0402 or 0603)
22.6kΩ 1% resistor (0402 or 0603)
6.34kΩ 1% resistor (0402 or 0603)
2200pF, 25V X7R ceramic
capacitors
C13, C20
R12
R16
16 ______________________________________________________________________________________
Low-Cost, Dual-Output, Step-Down
Controller with Fast Transient Response
MAX864
C1
0.01μF
C2
10μF
C3
10μF
C4
OPEN
R1
2.87kΩ
FB1
IN
ILIM1
DH1
N1
INPUT
7.2V TO 20V
C8
4700pF
R2
40.2kΩ
R3
10kΩ
R4
5.36kΩ
C5
1μF
LX1
L1
1.43μH
C7
0.22μF
OUT1
1.5V/10A
BST1
V
CC
C6
1μF
C9
470μF
C10
10μF
N2
R5
2Ω
DL1
MAX8664
REFIN2
VL
C11
1000pF
C12
4.7μF
PGND
C13
0.01μF
GND
C14
10μF
C15
10μF
ILIM2
N3
R6
2.26kΩ
VCC
DH2
LX2
R7
10kΩ
L2
1.43μH
POWER-GOOD
TO SYSTEM
OUT2
1.05V/8A
C16
0.22μF
PWRGD
R9
25.5kΩ 4700pF
C17
C18
470μF
C19
10μF
BST2
DL2
R10
2Ω
R8
75kΩ
N4
OSC/EN12
FB2
R11
10kΩ
C20
1000pF
N5
2N7002
ENABLE
R12
9.53kΩ
Figure 5. 300kHz Circuit with 7.2V to 20V Input
______________________________________________________________________________________ 17
Low-Cost, Dual-Output, Step-Down
Controller with Fast Transient Response
Trblv 50%CLmtLꢄvꢄu%ꢃꢅsu%fLꢁ%FꢅgOꢁv 5
DESIGNꢀTI N
QTY
DESCRIPTI N
DESIGNꢀTI N
QTY
DESCRIPTI N
0.01µF, 10V X7R ceramic
capacitors
1.43µH, 4.52mΩ inductors
Panasonic ETQP3H1E4BFA
C1, C13
2
L1, L2
2
C2, C3, C14, C15
4
1
1
10µF, 25V X5R ceramic capacitors
1µF, 25V X5R ceramic capacitor
1µF, 10V X5R ceramic capacitor
n-channel MOSFETs
IRF7821 (8-pin SOs)
N1–N4
N5
4
1
C5
C6
n-channel MOSFET
2N7002 (SOT23)
MAX864
0.22µF, 10V X7R ceramic
capacitors
C7, C16
C8, C17
2
2
R1
R2
1
1
2
1
2
1
1
1
1
1
2.87kΩ 1% resistor (0402 or 0603)
40.2kΩ 1% resistor (0402 or 0603)
10kΩ 1% resistors (0402 or 0603)
5.36kΩ 1% resistor (0402 or 0603)
2.0Ω 5% resistors (1206)
4700pF, 10V X7R ceramic
capacitors
R3, R11
R4
470µF/2.5V POSCAP capacitors
Sanyo 2R5TPD470M6
C9, C18
C10, C19
C11, C20
C12
2
2
2
1
R5, R10
R6
10µF, 6.3V X5R ceramic capacitors
2.26kΩ 1% resistor (0402 or 0603)
10kΩ 5% resistor (0402 or 0603)
75kΩ 1% resistor (0402 or 0603)
25.5kΩ 1% resistor (0402 or 0603)
9.53kΩ 1% resistor (0402 or 0603)
1000pF, 25V X7R ceramic
capacitors
R7
R8
4.7µF, 10V X5R ceramic capacitor
R9
R12
For tracking applications, connect REFIN2 to the center
of a resistive voltage-divider between the output of con-
troller 1 and GND. See Figure 6b. In this application,
the output of regulator 2 tracks the output voltage of
controller 1. The voltage-divider resistors set the
Power-Up and Sequencing
The MAX8664 features an OSC/EN12 input that is used
both for setting the switching frequency and as an
enable input for both controllers. A resistor from
OSC/EN12 to GND sets the switching frequency, and
when OSC/EN12 is high impedance, both controllers
enter low-power shutdown mode. This is easily
achieved with a transistor between the resistor and
GND. Figure 6a shows the startup configuration with
V
/V
ratio. A typical tracking application is for
OUT2 OUT1
the VTT supply of DDR memory.
Figure 6c shows one method of sequencing the out-
puts. Output 1 rises first. When PWRGD goes high, the
transistors allow the external reference to drive REFIN2
and output 2 rises. The circuit in Figure 6d functions
similarly, except the enable signal is supplied externally
instead of being driven by the PWRGD signal.
independent outputs. With REFIN2 connected to V
both controllers use the internal reference.
,
CC
CHIP
ENABLE
V
CC
V
OUT1
REFIN2
V
OUT2
MAX8664
ON
OSC/EN12
OFF
PWRGD
CHIP
ENABLE
Figure 6a. Two Independent Output Startup and Shutdown Waveforms
1. ______________________________________________________________________________________
Low-Cost, Dual-Output, Step-Down
Controller with Fast Transient Response
MAX864
V
OUT1
CHIP
ENABLE
REFIN2
V
V
OUT1
MAX8664
OUT2
ON
OSC/EN12
OFF
PWRGD
CHIP
ENABLE
Figure 6b. Ratiometric Tracking Startup and Shutdown Waveforms
V
CC
EXTERNAL
REF
CHIP
ENABLE
PWRGD
REFIN2
V
V
OUT1
MAX8664
OUT2
ON
OSC/EN12
OFF
PWRGD
CHIP
ENABLE
Figure 6c. Sequencing Startup and Shutdown Waveforms
______________________________________________________________________________________ 19
Low-Cost, Dual-Output, Step-Down
Controller with Fast Transient Response
EXTERNAL
REF
V
CC
CHIP
ENABLE
OUT2
ENABLE
REFIN2
MAX864
ON
OFF
V
OUT1
OUT2
ENABLE
MAX8664
V
OUT2
ON
OSC/EN12
OFF
PWRGD
CHIP
ENABLE
Figure 6d. Sequencing Startup and Shutdown Waveforms with System Enable 2 Signal
inductor value is not critical and can be adjusted to make
trade-offs among size, cost, and efficiency. Lower induc-
tor values minimize size and cost, but they also increase
the output ripple and reduce the efficiency due to higher
peak currents. On the other hand, higher inductor values
increase efficiency, but eventually resistive losses due to
extra turns of wire exceed the benefit gained from lower
AC current levels. This is especially true if the inductance
is increased without also increasing the physical size of
the inductor. Find a low-loss inductor having the lowest
possible DC resistance that fits the allotted dimensions.
The chosen inductor’s saturation current rating must
exceed the peak inductor current determined as:
Design Procedure
Setting the Switching Frequency
Connect a resistor from OSC/EN12 to GND to set the
switching frequency between 100kHz and 1000kHz.
Calculate the resistor value (R10 in Figures 2–5) as follows:
10
2.24 ×10 (Hz)
R10 =
(Ω)
f
S
Inductor Selection
There are several parameters that must be examined
when determining which inductor is to be used. Input
voltage, output voltage, load current, switching fre-
quency, and LIR. LIR is the ratio of inductor-current rip-
LIR
I
=I
+
×I
LOAD(MAX)
PEAK LOAD(MAX)
2
ple to maximum DC load current (I
). A higher
LOAD(MAX)
LIR value allows for a smaller inductor, but results in
higher losses and higher output ripple. A good compro-
mise between size and efficiency is an LIR of 0.3. Once
all the parameters are chosen, the inductor value is
determined as follows:
Output Capacitor
The key selection parameters for the output capacitor
are the actual capacitance value, the equivalent series
resistance (ESR), the equivalent series inductance
(ESL), and the voltage-rating requirements. These
parameters affect the overall stability, output voltage
ripple, and transient response. The output ripple has
three components: variations in the charge stored in
the output capacitor, the voltage drop across the
capacitor’s ESR, and ESL caused by the current into
and out of the capacitor. The maximum output voltage
ripple is estimated as follows:
V
×(V − V
)
OUT
IN
×LIR
LOAD(MAX)
OUT
L =
V
× f ×I
IN
S
where f is the switching frequency. Choose a standard
S
value inductor close to the calculated value. The exact
2± ______________________________________________________________________________________
Low-Cost, Dual-Output, Step-Down
Controller with Fast Transient Response
MAX864
V
= V
+ V
+ V
RIPPLE
RIPPLE(ESR)
RIPPLE(C) RIPPLE(ESL)
output voltage instantly changes by ESR x ΔI
.
LOAD
Before the controller can respond, the output voltage
deviates further depending on the inductor and output
capacitor values. After a short period of time (see the
Typical Operating Characteristics), the controller
responds by regulating the output voltage back to its
nominal state. The controller response time depends on
its closed-loop bandwidth. With a higher bandwidth,
the response time is faster, thus preventing the output
voltage from further deviation from its regulating value.
The output voltage ripple as a consequence of the
ESR, ESL, and output capacitance is:
V
=I
×ESR
RIPPLE(ESR) P−P
V
IN
V
=
×ESL
RIPPLE(ESL)
L +ESL
I
P−P
V
=
RIPPLE(C)
8×C
× f
S
OUT
Setting the Output Voltages and Voltage
Positioning
where I
is the peak-to-peak inductor current:
Figure 7 shows the feedback network used on the
MAX8664. With this configuration, a portion of the feed-
back signal is sensed on the switched side of the
inductor (LX), and the output voltage droops slightly as
the load current is increased due to the DC resistance
of the inductor (DCR). This allows the load regulation to
be set to match the voltage droop during a load tran-
sient (voltage positioning), reducing the peak-to-peak
output voltage deviation during a load transient, and
reducing the output capacitance requirements.
P-P
V
IN
− V
V
OUT
OUT
I
=
×
P−P
f ×L
S
V
IN
These equations are suitable for initial capacitor selec-
tion, but final values should be chosen based on a pro-
totype or evaluation circuit. As a general rule, a smaller
ripple current results in less output-voltage ripple. Since
the inductor ripple current is a factor of the inductor
value and input voltage, the output-voltage ripple
decreases with larger inductance, and increases with
higher input voltages. Ceramic, tantalum, or aluminum
polymer electrolytic capacitors are recommended. The
aluminum electrolytic capacitor is the least expensive;
however, it has higher ESR and ESL. To compensate for
this, use a ceramic capacitor in parallel to reduce the
switching ripple and noise. For reliable and safe opera-
tion, ensure that the capacitor’s voltage and ripple-cur-
rent ratings exceed the calculated values.
To set the magnitude of the voltage positioning, select
a value for R2 in the 8kΩ to 24kΩ range, then calculate
the value of R1 as follows:
⎛
⎞
I
×DCR
OUT(MAX)
R1=R2×
−1
⎟
⎜
ΔV
⎝
OUT(MAX)
⎠
where I
is the maximum output current and
OUT(MAX)
Δ V
is the maximum allowable droop in the
OUT(MAX)
output voltage at full load.
The response to a load transient depends on the
selected output capacitors. After a load transient, the
L
DCR
LX_
OUT
R
LOAD
ESR
R1
Cr
R2
R3
C
OUT
FB_
Figure 7. Feedback Network
______________________________________________________________________________________ 21
Low-Cost, Dual-Output, Step-Down
Controller with Fast Transient Response
To set the no-load output voltage (V
value of R3 as follows:
), calculate the
Finally, calculate the value of Cr as follows:
OUT
V
OUT
V
IN
− V
OUT
(
V
IN
)
⎛
⎞
⎟
V
R1 × R2
R1 + R2
⎛
⎞
FB
Cr =
R3 =
⎜
⎝
⎟
⎠
⎜
R1× f ×| V
− V
|
(
)
S
FB_RIPPLE
OUT_RIPPLE
V
− V
⎝
⎠
OUT
FB
where V
is the feedback regulation voltage (0.6V
MOSFET Selection
FB
when using the internal reference or V
for exter-
Each output of the MAX8664 is capable of driving two to
four external, logic-level, n-channel MOSFETs as the cir-
cuit switch elements. The key selection parameters are:
REFIN2
nal reference). If the desired output voltage is equal to
the reference voltage (typical for tracking applications),
R3 is not installed.
MAX864
•
•
On-resistance (R
)—the lower, the better.
DS(ON)
To achieve the lowest possible load regulation in appli-
cations where voltage positioning is not desired, R1 is
not installed and R3 is calculated as follows:
Maximum Drain-to-Source Voltage (V
)—should
DSS
be at least 20% higher than the input supply rail at
the high-side MOSFET’s drain.
•
Gate charges (Q , Q , Q )— the lower, the better.
g gd gs
⎛
V
⎞
FB
− V
FB
R3 =
×R2
⎜
⎝
⎟
⎠
For a 5V input application, choose MOSFETs with rated
at V ≤ 4.5V. With higher input voltages, the
V
OUT
R
DS(ON)
GS
internal VL regulator provides 6.5V for gate drive in
order to minimize the on-resistance for a wide range of
MOSFETs.
Compensation
To ensure stable operation, connect a compensation
capacitor (Cr) across the upper feedback resistor as
shown in Figure 7. To find the value of this capacitor,
follow the compensation design procedure below.
For a good compromise between efficiency and cost,
choose the high-side MOSFETs that have conduction
losses equal to switching losses at nominal input voltage
Choose a closed-loop bandwidth (f ) that is less than
C
and output current. Low R
is preferred for low-
DS(ON)
1/3 the switching frequency (f ). Calculate the output
S
side MOSFETs. Make sure that the low-side MOSFET(s)
does not spuriously turn on due to dV/dt caused by the
high-side MOSFET(s) turning on, as this would result in
shoot-through current and degrade the efficiency.
double pole (f ) as follows:
O
1
f
O
=
R
R
+ESR
+DCR
LOAD
MOSFETs with a lower Q
/ Q ratio have higher
gs
gd
2π L ×C
×
OUT
LOAD
immunity to dV/dt. For high-current applications, it is
often preferable to parallel two MOSFETs rather than to
use a single large MOSFET.
The FB peak-to-peak voltage ripple is:
For proper thermal management, the power dissipation
must be calculated at the desired maximum operating
junction temperature, maximum output current, and
worst-case input voltage. For the-low side MOSFET(s),
the worst-case power dissipation occurs at the highest
⎛
⎞
R2
R1
⎛
⎞
1+
⎜
⎜
⎜
⎟
⎟
⎟
V
⎜
⎟
OUT
V
=
×
FB_RIPPLE
⎜
⎟
R2 R2
⎛
DCR ⎞
f
C
f
O
1+
+
⎜
⎝
⎟
⎠
1+
×
⎜
⎝
⎟
⎜
⎝
⎟
⎠
R3 R1
⎠
R
LOAD
duty cycle (V
). The low-side MOSFET(s) operate
IN(MAX)
The output ripple voltage due to the ESR of the output
as zero voltage switches; therefore, major losses are
capacitor, C
is:
OUT,
the channel conduction loss (P
) and the body
LSCC
diode conduction loss (P
):
LSDC
V
OUT
V
− V
OUT
(
)
IN
⎛
⎞
⎟
V
V
OUT
2
IN
V
=
×
P
= 1−
×I
×R
OUT_RIPPLE
LSCC(MAX)
DS(ON)
⎜
LOAD(MAX)
L × f
V
S
⎝
IN(MAX) ⎠
⎛
1
⎞
ESR+
⎜
⎝
⎟
⎠
Use R
at T
:
J(MAX)
DS(ON)
8×C × f
O
S
P
= 2 x I
V x t x f
LSDC(MAX)
LOAD(MAX) F DT S
Target the feedback ripple in the 25mV to 60mV range.
For high duty-cycle applications (> 70%), a feedback
ripple of 25mV is recommended.
where V is the body diode forward-voltage drop, t is the
F
DT
dead time between high-side and low-side switching tran-
sitions (25ns typical), and f is the switching frequency.
S
22 ______________________________________________________________________________________
Low-Cost, Dual-Output, Step-Down
Controller with Fast Transient Response
MAX864
The high-side MOSFET(s) operate as duty-cycle control
switches and have the following major losses: the chan-
nel conduction loss (P ), the overlapping switching
interfere with circuit performance and generate EMI. To
dampen this ringing, a series RC snubber circuit is
added across each low-side switch. Below is the pro-
cedure for selecting the value of the series RC circuit.
HSCC
loss (P
), and the drive loss (P
). The maxi-
HSSW
HSDR
mum power dissipation could occur either at V
IN(MAX)
Connect a scope probe to measure V
to GND and
LX_
or V
:
IN(MIN)
observe the ringing frequency, f .
R
Find the capacitor value (connected from LX_ to GND)
that reduces the ringing frequency by half.
V
OUT
2
P
=
×I
×R
DS(ON)
LOAD(MAX)
HSCC(MAX)
V
IN(MIN)
The circuit parasitic capacitance (C
) at LX_ is then
PAR
equal to 1/3 the value of the added capacitance above.
Use R
at T
:
J(MAX)
DS(ON)
The circuit parasitic inductance (L
) is calculated by:
PAR
Q
GD
1
P
= V
×I
×
× f
S
HSSW(MAX)
IN(MAX) LOAD(MAX)
L
=
PAR
I
GATE
2
2πf
(
×C
PAR
)
R
where I
capability determined by:
is the average DH driver output-current
GATE
The resistor for critical dampening (R
) is equal to
SNUB
2π x f x L
. Adjust the resistor value up or down to
R
PAR
tailor the desired damping and the peak-voltage excur-
sion.
0.5× V
VL
+R
GATE
I
≅
GATE
R
DS(ON)(DR)
The capacitor (C
) should be at least 2 to 4 times
to be effective. The power loss of
SNUB
the value of the C
PAR
where R
is the DH_ driver’s on-resistance
DS(ON)(DR)
(see the Electrical Characteristics) and R
the snubber circuit is dissipated in the resistor
(P ) and can be calculated as:
is the
GATE
RSNUB
internal gate resistance of the MOSFET (~ 2Ω):
2
P
= C
× V
× f
SW
(
)
RSNUB
SNUB
IN
R
GATE
P
HSDR
= Q × V × f ×
G GS S
R
GATE
+R
DS(ON)(DR)
where V is the input voltage and f
frequency. Choose an R
is the switching
IN
SW
power rating that meets
SNUB
the specific application’s derating rule for the power
dissipation calculated.
where V ≈ V
.
GS
VL
The high-side MOSFET(s) do not have body diode con-
duction loss, unless the converter is sinking current.
When sinking current, calculate this loss as
Setting the Overcurrent Threshold
Connect a resistor from ILIM_ to the drain of the high-
side MOSFET(s) to set the overcurrent protection
threshold. ILIM_ sinks 50µA (typ) through this resistor.
When the drain-source voltage exceeds the voltage
drop across this resistor during the high-side MOS-
FET(s) on-time, overcurrent protection is triggered. To
set the output current level where overcurrent protec-
P
= I
x V x (2 x t + t ) x f ,
LOAD(MAX) F DT WD S
HSDC(MAX)
where t
is about 130ns.
WD
Allow an additional 20% for losses due to MOSFET out-
put capacitances and low-side MOSFET body diode
reverse-recovery charge dissipated in the high-side
MOSFET(s). Refer to the MOSFET data sheet for ther-
mal resistance specifications to calculate the PCB area
needed to maintain the desired maximum operating
junction temperature with the above calculated power
dissipations.
tion is triggered (I
), calculate the value of the ILIM_
LIMIT
resistor as follows:
R
×I
DS(ON)HS LIMIT
R
=
ILIM_
50μA
MOSFET Snubber Circuit
Fast switching transitions cause ringing because of res-
onating circuit parasitic inductance and capacitance at
the switching nodes. This high-frequency ringing
occurs at LX’s rising and falling transitions and can
where R
is the maximum on-resistance of the
DS(ON)HS
high-side MOSFET(s) at +25°C. At higher tempera-
tures, the ILIM current increases to compensate for the
temperature coefficient of the high-side MOSFET(s).
______________________________________________________________________________________ 23
Low-Cost, Dual-Output, Step-Down
Controller with Fast Transient Response
way that the high-side MOSFET’s drain is close and
Input Capacitor
The input filter capacitors reduce peak currents drawn
from the power source and reduce noise and voltage
ripple on the input caused by the circuit’s switching.
The input capacitors must meet the ripple current
near the low-side MOSFET’s source. This allows the
input ceramic decoupling capacitor to be placed
directly across and as close as possible to the
high-MOSFET’s drain and the low-side MOSFET’s
source. This helps contain the high switching cur-
rent within this small loop.
requirement (I
) imposed by the switching currents.
RMS
The ripple current requirement can be estimated by the
following equation:
3) Pour an analog ground plane in the second layer
underneath the IC to minimize noise coupling.
1
V
IN
2
2
MAX864
I
=
I
× V
OUT1
× V − V
+ I
× V
× V − V
(
)
(
)
(
)
(
)
RMS
OUT1
IN
OUT1
OUT2
OUT2 IN OUT2
4) Connect input, output, and VL capacitors to the
power ground plane; connect all other capacitors to
the signal ground plane.
Choose a capacitor that exhibits less than 10°C tem-
perature rise at the maximum operating RMS current for
optimum long-term reliability.
5) Place the MOSFETs as close as possible to the IC
to minimize trace inductance of the gate drive loop.
If parallel MOSFETs are used, keep the trace
lengths to both gates equal and short.
Applications Information
PCB Layout Guidelines
Careful PCB layout is an important factor in achieving
low switching losses and clean, stable operation. The
switching power stage requires particular attention.
Follow these guidelines for good PCB layout:
6) Connect the drain leads of the power MOSFET to a
large copper area to help cool the device. Refer to
the power MOSFET data sheet for recommended
copper area.
7) Place the feedback network components as close
as possible to the IC pins.
1) A multilayer PCB is recommended.
2) Place IC decoupling capacitors as close as possi-
ble to the IC pins. Keep separate power ground
and signal ground planes. Place the low-side
MOSFETs near the PGND pin. Arrange the high-
side MOSFETs and low-side MOSFETs in such a
8) The current-limit setting RC should be Kelvin con-
nected to the high-side MOSFETs’ drain.
Refer to the MAX8664 evaluation kit for an example layout.
24 ______________________________________________________________________________________
Low-Cost, Dual-Output, Step-Down
Controller with Fast Transient Response
MAX864
Pin Configuration
Chip Information
PROCESS: BiCMOS
TOP VIEW
DH1
LX1
1
2
3
4
5
6
7
8
9
20 ILIM1
19 FB1
BST1
DL1
18 PWRGD
MAX8664
17 V
CC
VL
16 GND
PGND
DL2
15 IN
14 OSC/EN12
13 REFIN2
12 FB2
BST2
LX2
DH2 10
11 ILIM2
QS P
______________________________________________________________________________________ 25
Low-Cost, Dual-Output, Step-Down
Controller with Fast Transient Response
Package Information (continued)
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information
go to www0mrxꢅm-ꢅA0ALm/trAkrgvs.)
MAX864
PACKAGE OUTLINE, QSOP .150", .025" LEAD PITCH
1
21-0055
F
1
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
26 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600
© 2007 Maxim Integrated Products
is a registered trademark of Maxim Integrated Products, Inc.
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