MIC2101 [MICREL]
38V, Synchronous Buck Controllers Featuring Adaptive On-Time Control; 38V ，同步降压控制器配备自适应导通时间控制
| 型号： | MIC2101 |
| 厂家： | 
MICREL SEMICONDUCTOR |
| 描述： | 38V, Synchronous Buck Controllers Featuring Adaptive On-Time Control |
| 文件： | 总37页 (文件大小：1594K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
MIC2101/02
38V, Synchronous Buck Controllers
Featuring Adaptive On-Time Control
Hyper Speed Control Family
General Description
Features
Hyper Speed Control architecture enables:
- High Delta V operation (VIN = 38V and VOUT = 1.2V)
- Any Capacitor stable
The
Micrel MIC2101/02
are constant-frequency,
synchronous buck controllers featuring a unique adaptive
ON-time control architecture. The MIC2101/02 operates
over an input supply range from 4.5V to 38V and can be
used to supply up to 15A of output current. The output
voltage is adjustable down to 0.8V with a guaranteed
accuracy of ±1%. The device operates with programmable
switching frequency from 200kHz to 600kHz.
4.5V to 38V input voltage
0.8V reference voltage with ±1% accuracy
200kHz to 600kHz, programmable switching frequency
Hyper Light Load Control (MIC2101)
Hyper Speed Control (MIC2102)
Enable input and Power Good output
Built-in 5V regulator for single-supply operation
Programmable current limit and fold-back “hiccup”
mode short-circuit protection
5ms internal soft-start, internal compensation, and
thermal shutdown
Micrel’s Hyper Light Load™ architecture provides the same
high-efficiency and ultra-fast transient response as the Hyper
Speed Control architecture under the medium to heavy loads,
but also maintains high efficiency under light load conditions
by transitioning to variable frequency, discontinuous-mode
operation.
The MIC2101/02 offers a full suite of protection features to
ensure protection of the IC during fault conditions. These
include undervoltage lockout to ensure proper operation
under power-sag conditions, internal soft-start to reduce
inrush current, fold-back current limit, “hiccup” mode short-
circuit protection and thermal shutdown.
Supports safe start-up into a pre-biased output
–40C to +125C junction temperature range
Available in 16-pin 3mm x 3mm MLF® package
All support documentation can be found on Micrel’s web
site at: www.micrel.com.
Applications
Distributed power systems
Networking/telecom Infrastructure
Printers, scanners, graphic cards, and video cards
__________________________________________________________________________________________________________
Typical Application
MIC2101 Efficiency (VIN = 12V)
vs. Output Current (MIC2101)
100
5.0V
3.3V
90
2.5V
1.8V
80
1.5V
1.2V
1.0V
0.9V
0.8V
70
60
50
40
30
20
10
fSW = 600kHz (CCM)
0
1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16
OUTPUT CURRENT (A)
MIC2101/02 Wide Input, Hyper Light Load Buck Converter
Hyper Speed Control, Hyper Light Load, and Any Capacitor are trademarks of Micrel, Inc.
MLF and MicroLeadFrame are registered trademarks of Amkor Technology, Inc.
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
M9999-080712-A
August 2012
Micrel, Inc.
MIC2101/02
Ordering Information
Junction
Temperature
Range
Switching
Part Number
Features
Package
Lead Finish
Frequency
MIC2101YML
MIC2102YML
200kHz to 600kHz
200kHz to 600kHz
Hyper Light Load
16-Pin 3mm x 3mm MLF
16-Pin 3mm x 3mm MLF
–40°C to +125°C
–40°C to +125°C
Pb-Free
Pb-Free
Hyper Speed Control
Pin Configuration
16-Pin 3mm x 3mm MLF (ML)
(TOP VIEW)
Pin Description
Pin Number
Pin Name Pin Function
Internal +5V Linear Regulator Output. VDD is the internal supply bus for the device. A 4.7μF ceramic
capacitor from VDD to AGND is required for decoupling. In the applications with VIN < +5.5V, VDD
should be tied to VIN to by-pass the linear regulator.
1
VDD
5V supply input for the low-side N-channel MOSFET driver, which can be tied to VDD externally. A
4.7μF ceramic capacitor from PVDD to PGND is recommended for decoupling.
2
3
PVDD
ILIM
Current Limit Setting. Connect a resistor from SW to ILIM to set the over-current threshold for the
converter.
Low-Side Drive output. High-current driver output for external low-side MOSFET of a buck converter.
The DL driving voltage swings from ground to VDD. Adding a small resistor between DL pin and the
gate of the low-side N-channel MOSFET can slow down the turn-on and turn-off speed of the
MOSFET.
4
DL
Power Ground. PGND is the return path for the buck converter power stage. The PGND pin
connects to the sources of low-side N-Channel external MOSFET, the negative terminals of input
capacitors, and the negative terminals of output capacitors. The return path for the power ground
should be as small as possible and separate from the signal ground (AGND) return path.
5
6
PGND
FREQ
Switching Frequency Adjust input. Tie this pin to VIN to operate at 600kHz and place a resistor
divider to reduce the frequency.
2
M9999-080712-A
August 2012
Micrel, Inc.
MIC2101/02
Pin Configuration (Continued)
Pin Number
Pin Name
Pin Function
High-Side Drive Output. High-current driver output for external high-side MOSFET of a buck
converter. The DH driving voltage is floating on the switch node voltage (VSW). Adding a small
resistor between DH pin and the gate of the high-side N-channel MOSFET can slow down the turn-
on and turn-off speed of the MOSFET.
7
DH
Switch Node and Current-Sense input. High current output driver return. The SW pin connects
directly to the switch node. Due to the high-speed switching on this pin, the SW pin should be
routed away from sensitive nodes. The SW pin also senses the current by monitoring the voltage
across the low-side MOSFET during OFF time. In order to sense the current accurately, connect the
low-side MOSFET drain to the SW pin using a Kelvin connection.
8
SW
9, 11
10
NC
No Connection.
Voltage supply input for the high-side N-channel MOSFET driver, which can be powered by a
bootstrapped circuit connected between VDD and SW, using a Schottky diode and a 0.1μF ceramic
capacitor. Adding a small resistor at BST pin can slow down the turn-on speed of the high-side
MOSFET.
BST
Signal ground for VDD and the control circuitry, which is connected to thermal pad electronically.
The signal ground return path should be separate from the power ground (PGND) return path.
12
13
14
15
AGND
FB
Feedback Input. Input to the transconductance amplifier of the control loop. The FB pin is regulated
to 0.8V. A resistor divider connecting the feedback to the output is used to set the desired output
voltage.
Power Good Output. Open drain output, an external pull-up resistor to VDD or external power rails
is required.
PG
Enable Input. A logic signal to enable or disable the buck converter operation. The EN pin is CMOS
compatible. Logic high enables the device, logic low shutdowns the regulator. In the disable mode,
the VDD supply current for the device is minimized to 0.7mA typically.
EN
Supply Voltage. The VIN operating voltage range is from 4.5V to 38V. A 1μF ceramic capacitor from
VIN to AGND is required for decoupling.
16
VIN
EP
ePad
Exposed Pad. Connect the EPAD to PGND plain on the PCB to improve the thermal performance.
3
M9999-080712-A
August 2012
Micrel, Inc.
MIC2101/02
Absolute Maximum Ratings(1)
Operating Ratings(3)
Supply Voltage (VIN).......................................... 4.5V to 38V
Enable Input (VEN).................................................. 0V to VIN
VSW, VFREQ, VILIM, VEN ............................................. 0V to VIN
Junction Temperature (TJ) ........................40C to +125C
Junction Thermal Resistance
VIN ................................................................ 0.3V to +40V
VDD, VPVDD........................................................ 0.3V to +6V
VSW, VFREQ, VILIM, VEN............................ 0.3V to (VIN +0.3V)
VBST to VSW ........................................................ 0.3V to 6V
VBST ................................................................ 0.3V to 46V
VPG .....................................................0.3V to (VDD + 0.3V)
VFB .....................................................0.3V to (VDD + 0.3V)
PGND to AGND ........................................... 0.3V to +0.3V
Junction Temperature ..............................................+150°C
Storage Temperature (TS).........................65C to +150C
Lead Temperature (soldering, 10s)............................ 260°C
ESD Rating(2)................................................. ESD Sensitive
3mm x 3mm MLF-16 (JA) .......................50.8°C/W
3mm x 3mm MLF-16 (JC) .......................25.3°C/W
Electrical Characteristics(4)
VIN = 12V, VOUT =1.2V; VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate 40°C ≤ TJ ≤ +125°C.
Parameter
Condition
Min.
Typ.
Max.
Units
Power Supply Input
Input Voltage Range (VIN)(5)
Quiescent Supply Current (MIC2101)
Quiescent Supply Current (MIC2102)
Shutdown Supply Current
4.5
38
750
3
V
VFB = 1.5V
400
2.1
0.1
µA
mA
µA
VFB = 1.5V
SW unconnected, VEN = 0V
10
VDD Supply
VDD Output Voltage
VDD UVLO Threshold
VDD UVLO Hysteresis
Load Regulation
Reference
VIN = 7V to 38V, IDD = 10mA
VDD rising
4.8
3.8
5.2
4.2
400
2
5.4
4.6
V
V
mV
%
IDD = 0 to 40mA
0.6
3.6
TJ = 25°C (±1.0%)
40°C ≤ TJ ≤ 125°C (±2%)
VFB = 0.8V
0.792
0.8
0.8
5
0.808
0.816
500
Feedback Reference Voltage
V
0.784
FB Bias Current
Enable Control
EN Logic Level High
EN Logic Level Low
EN Hysteresis
nA
1.8
V
V
0.6
30
200
6
mV
µA
EN Bias Current
Oscillator
VEN = 12V
V
FREQ = VIN
400
600
300
85
750
Switching Frequency
kHz
VFREQ = 50%VIN
Maximum Duty Cycle
Minimum Duty Cycle
Minimum Off-Time
%
%
ns
VFB > 0.8V
0
140
200
260
4
M9999-080712-A
August 2012
Micrel, Inc.
MIC2101/02
Electrical Characteristics(4) (Continued)
VIN = 12V, VOUT = 1.2V; VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate 40°C ≤ TJ ≤ +125°C.
Parameter
Condition
Min.
Typ.
Max.
Units
Soft-Start
Soft-Start time
5
ms
Short-Circuit Protection
Current-Limit Threshold
Short-Circuit Threshold
Current-Limit Source Current
Short-Circuit Source Current
FET Drivers
VFB = 0.79V
VFB = 0V
0
9
mV
mV
µA
30
23
60
14
7
VFB = 0.79V
VFB = 0V
80
100
47
27
37
µA
DH, DL Output Low Voltage
ISINK = 10mA
0.1
V
V
VPVDD 0.1V
or
DH, DL Output High Voltage
ISOURCE = 10mA
VBST 0.1V
DH On-Resistance, High State
DH On-Resistance, Low State
DL On-Resistance, High State
DL On-Resistance, Low State
SW, BST Leakage Current
Power Good (PG)
2.1
1.8
1.8
1.2
3.3
3.3
3.3
2.3
50
Ω
Ω
Ω
Ω
µA
PG Threshold Voltage
PG Hysteresis
Sweep VFB from Low to High
Sweep VFB from High to Low
Sweep VFB from Low to High
VFB < 90% x VNOM, IPG = 1mA
85
90
6
95
%VOUT
%VOUT
µs
PG Delay Time
100
70
PG Low Voltage
200
mV
Thermal Protection
Over-Temperature Shutdown
Over-Temperature Shutdown Hysteresis
Notes:
TJ Rising
160
7
°C
°C
1. Exceeding the absolute maximum rating may damage the device.
2. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5kΩ in series with 100pF.
3. The device is not guaranteed to function outside operating range.
4. Specification for packaged product only.
5. The application is fully functional at low VDD (supply of the control section) if the external MOSFETs have low voltage VTH
.
5
M9999-080712-A
August 2012
Micrel, Inc.
MIC2101/02
Typical Characteristics
VIN Operating Supply Current
vs. Input Voltage (MIC2101)
VIN Operating Supply Current
vs. Input Voltage (MIC2101)
Output Regulation
vs. Input Voltage (MIC2101)
1.00
0.90
0.80
0.70
0.60
0.50
0.40
0.30
0.20
0.10
0.00
1.0%
0.8%
0.6%
0.4%
0.2%
0.0%
-0.2%
-0.4%
-0.6%
-0.8%
-1.0%
1.00
VOUT = 1.2V
IOUT = 0A
VOUT = 3.3V
VOUT = 3.3V
0.90
IOUT = 0A
0.80
IOUT = 0A to 12A
0.70
0.60
0.50
0.40
0.30
0.20
0.10
0.00
4
9
14
19
24
29
34
5
10
15
20
25
30
35
5
10
15
20
25
30
35
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Output Voltage
vs. Input Voltage (MIC2101)
Feedback Voltage
vs. Input Voltage (MIC2101)
Output Regulation
vs. Input Voltage (MIC2101)
1.0%
0.8%
0.6%
0.4%
0.2%
0.0%
-0.2%
-0.4%
-0.6%
-0.8%
-1.0%
3.333
3.316
3.300
3.283
3.267
3.250
3.234
3.217
0.824
0.816
0.808
0.800
0.792
0.784
0.776
VOUT = 3.3V
IOUT = 0A
VOUT = 1.2V
VOUT = 3.3V
IOUT = 0A to 12A
IOUT = 0A
5
10
15
20
25
30
35
5
10
15
20
25
30
35
5
10
15
20
25
30
35
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Output Voltage
vs. Input Voltage (MIC2101)
VIN Operating Supply Current
vs. Temperature (MIC2101)
Feedback Voltage
vs. Temperature (MIC2101)
1.212
1.210
1.208
1.206
1.204
1.202
1.200
1.198
1.196
1.00
0.90
0.80
0.70
0.60
0.50
0.40
0.30
0.20
0.10
0.00
0.808
0.804
0.800
0.796
0.792
VOUT = 1.2V
VIN = 12V
VIN = 12V
VOUT = 3.3V
IOUT = 0A
IOUT = 0A
VOUT = 3.3V
IOUT = 0A
5
10
15
20
25
30
35
-50
-25
0
25
50
75
100
125
-50 -25
0
25
50
75 100 125
INPUT VOLTAGE (V)
TEMPERATURE (°C)
TEMPERATURE (°C)
6
M9999-080712-A
August 2012
Micrel, Inc.
MIC2101/02
Typical Characteristics (Continued)
Load Regulation
Line Regulation
Feedback Voltage
vs. Temperature (MIC2101)
vs. Temperature (MIC2101)
vs. Output Current (MIC2101)
0.808
0.804
0.800
0.796
0.792
0.4%
0.0%
-0.2%
-0.4%
-0.6%
-0.8%
-1.0%
-1.2%
-1.4%
-1.6%
-1.8%
VIN = 12V
0.3%
VOUT = 3.3V
IOUT = 0 to 12A
0.2%
0.1%
0.0%
-0.1%
-0.2%
-0.3%
VIN = 5V to 38V
V
OUT = 3.3V
VIN = 12V
IOUT = 0A
VOUT = 3.3V
-50
-25
0
25
50
75
100
125
-50 -25
0
25 50 75 100 125
0
1
2
3
4
5
6
7
8
9
10 11 12
TEMPERATURE (°C)
TEMPERATURE (°C)
OUTPUT CURRENT (A)
Line Regulation
vs. Output Current (MIC2101)
Efficiency (VIN = 5V)
vs. Output Current (MIC2101)
Efficiency (VIN =12V)
vs. Output Current (MIC2101)
100
3.0%
100
90
80
70
60
50
40
30
20
10
0
90
80
70
60
50
40
30
20
10
0
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
5.0V
2.0%
1.0%
0.0%
-1.0%
-2.0%
-3.0%
3.3V
2.5V
1.8V
1.2V
0.8V
VIN = 5V to 38V
V
OUT = 3.3V
fSW = 600kHz (CCM)
fSW = 600kHz (CCM)
0
2
4
6
8
10
12
14
16
0
4
8
12
16
0
1
2
3
4
5
6
7
8
9
10 11 12
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Efficiency (VIN = 24V)
vs. Output Current (MIC2101)
Efficiency (VIN = 38V)
vs. Output Current (MIC2101)
Efficiency (VIN = 18V)
vs. Output Current (MIC2101)
100
90
80
70
60
50
40
30
20
10
0
100
100
90
80
70
60
50
40
30
20
10
0
90
80
70
60
50
40
30
20
10
0
5.0V
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
3.3V
2.5V
1.8V
1.2V
0.8V
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
fSW = 600kHz (CCM)
fSW = 600kHz (CCM)
fSW = 600kHz (CCM)
0
2
4
6
8
10
12
14
16
0
2
4
6
8
10
12
14
16
0
2
4
6
8
10
12
14
16
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
7
M9999-080712-A
August 2012
Micrel, Inc.
MIC2101/02
Typical Characteristics (Continued)
Feedback Voltage
vs. Input Voltage (MIC2102)
Output Regulation
vs. Input Voltage (MIC2102)
VIN Operating Supply Current
vs. Input Voltage (MIC2102)
1.0%
0.8%
0.6%
0.4%
0.2%
0.0%
-0.2%
-0.4%
-0.6%
-0.8%
-1.0%
0.808
0.804
0.800
0.796
0.792
60
48
36
24
12
0
VOUT = 3.3V
VOUT = 3.3V
IOUT = 0A
IOUT = 0A to 12A
fSW = 600kHz
fSW = 600kHz
VOUT = 3.3V
OUT = 0A
fSW = 600kHz
I
4
9
14
19
24
29
34
39
4
9
14
19
24
29
34
39
5
10
15
20
25
30
35
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Output Regulation
vs. Input Voltage (MIC2102)
VIN Operating Supply Current
vs. Input Voltage (MIC2102)
VIN Operating Supply Current
vs. Temperature (MIC2102)
60
48
36
24
12
1.0%
0.8%
0.6%
0.4%
0.2%
0.0%
-0.2%
-0.4%
-0.6%
-0.8%
-1.0%
50
40
30
20
10
`
VOUT = 1.2V
VIN = 12V
IOUT = 0A to 12A
fSW = 600kHz
VOUT = 1.2V
V
OUT = 3.3V
OUT = 0A
fSW = 600kHz
IOUT = 0A
I
fSW = 600kHz
0
5
0
5
10
15
20
25
30
35
10
15
20
25
30
35
-50
-25
0
25
50
75
100
125
INPUT VOLTAGE (V)
TEMPERATURE (°C)
INPUT VOLTAGE (V)
Load Regulation
vs. Temperature (MIC2102)
Line Regulation
vs. Temperature (MIC2102)
Feedback Voltage
vs. Temperature (MIC2102)
0.808
0.804
0.800
0.796
0.792
0.4%
0.3%
0.2%
0.1%
0.0%
-0.1%
-0.2%
-0.3%
0.4%
0.3%
0.2%
0.1%
0.0%
-0.1%
-0.2%
-0.3%
VIN = 5V to 38V
V
OUT = 3.3V
VIN = 12V
IOUT = 0A
VOUT = 3.3V
IOUT = 0A to 12A
fSW = 600kHz
VIN = 12V
V
OUT = 3.3V
IOUT = 0A
-50
-25
0
25
50
75
100
125
-50
-25
0
25
50
75
100
125
-50
-25
0
25
50
75
100
125
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
8
M9999-080712-A
August 2012
Micrel, Inc.
MIC2101/02
Typical Characteristics (Continued)
Switching Frequency
Feedback Voltage
Line Regulation
vs. Output Current (MIC2102)
vs. Output Current (MIC2102)
vs. Output Current (MIC2102)
700
0.808
0.3%
0.2%
0.1%
0.0%
-0.1%
-0.2%
-0.3%
650
25°C
600
550
500
450
400
350
300
250
200
150
100
0.804
0.800
0.796
0.792
-40°C
125°C
VIN = 5V to 38V
VOUT = 3.3V
VDD = 5V
VIN = 12V
VIN = 12V
V
OUT = 3.3V
VOUT = 3.3V
fSW = 600kHz
fSW = 600kHz
0
1
2
3
4
5
6
7
8
9
10 11 12
0
1
2
3
4
5
6
7
8
9
10 11 12
0
2
4
6
8
10
12
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Efficiency (VIN = 12V)
vs. Output Current (MIC2102)
Efficiency (VIN = 5V)
Efficiency (VIN = 18V)
vs. Output Current (MIC2102)
vs. Output Current (MIC2102)
100
90
80
70
60
50
40
30
20
10
0
100
90
80
70
60
50
40
30
20
10
100
90
80
70
60
50
40
30
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
5.0V
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
3.3V
2.5V
1.8V
1.2V
0.8V
fSW = 600kHz
VSW = 600kHz
fSW = 600kHz
0
0
4
8
12
16
0
4
8
12
16
0
4
8
12
16
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Efficiency (VIN = 24V)
Efficiency (VIN = 38V)
vs. Output Current (MIC2102)
vs. Output Current (MIC2102)
100
90
80
70
60
50
40
30
20
10
100
90
80
70
60
50
40
30
20
10
0
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
fSW = 600kHz
12 16
fSW = 600kHz
0
0
0
4
8
12
16
4
8
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
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MIC2101/02
Typical Characteristics (Continued)
Die Temperature* (VIN = 5.0V)
vs. Output Current
Die Temperature* (VIN = 12V)
vs. Output Current
Die Temperature* (VIN = 24V)
vs. Output Current
80
60
40
20
0
120
100
80
60
40
20
0
100
80
60
40
20
0
VIN = 5.0V
VIN = 12V
VIN = 24V
VOUT = 3.3V
fSW = 600kHz
V
OUT = 3.3V
VOUT = 3.3V
fSW = 600kHz
fSW = 600kHz
0
1
2
3
4
5
6
7
8
9 10 11 12
0
1
2
3
4
5
6
7
8
9
10 11 12
0
1
2
3
4
5
6
7
8
9
10 11 12
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Die Temperature* (VIN = 5.0V)
vs. Output Current
Die Temperature* (VIN = 12V)
vs. Output Current
Die Temperature* (VIN = 38V)
vs. Output Current
100
80
60
40
20
0
80
60
40
20
0
160
140
120
100
80
`
60
VIN = 38V
40
VIN = 5.0V
VOUT = 3.3V
fSW = 600kHz
VIN = 12V
VOUT = 1.2V
fSW = 600kHz
20
VOUT = 1.2V
fSW = 600kHz
0
0
1
2
3
4
5
6
7
8
9 10 11 12
0
1
2
3
4
5
6
7
8
9
10 11 12
0
1
2
3
4
5
6
7
8
9
10 11 12
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Die Temperature* (VIN = 24V)
vs. Output Current
Die Temperature* (VIN = 38V)
vs. Output Current
120
160
100
80
60
40
20
0
140
120
100
80
60
VIN = 24V
VIN = 38V
40
V
OUT = 1.2V
V
OUT = 1.2V
fSW = 600kHz
20
fSW = 600kHz
0
0
1
2
3
4
5
6
7
8
9
10 11 12
0
1
2
3
4
5
6
7
8
9 10 11 12
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Case Temperature* : The temperature measurement was taken at the hottest point on the MIC2101/02 case mounted on a 5 square inch PCB, see
Thermal Measurement section. Actual results will depend upon the size of the PCB, ambient temperature and proximity to other heat emitting
components.
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MIC2101/02
Typical Characteristics (Continued)
VIN Shutdown Current
vs. Input Voltage
VDD Voltage
vs. Input Voltage
Enable Threshold
vs. Input Voltage
30
1.50
1.40
1.30
1.20
1.10
1.00
0.90
0.80
0.70
0.60
0.50
0.40
0.30
0.20
0.10
0.00
10
RISING
24
8
IDD = 10mA
FALLING
18
12
6
6
4
2
0
VOUT = 3.3V
IDD = 40mA
fSW = 600kHz
HYST
VEN = 0V
0
4
9
14
19
24
29
34
39
4
9
14
19
24
29
34
39
INPUT VOLTAGE (V)
4
9
14
19
24
29
34
39
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Switching Frequency
vs. Input Voltage
Switching Frequency
vs. Input Voltage
Output Peak Current Limit
vs. Input Voltage
800
750
700
650
600
550
500
450
400
350
300
250
200
800
25
750
700
650
600
550
500
450
400
350
300
250
200
VOUT = 1.2V
OUT = 2A
VOUT = 3.3V
OUT = 2A
I
I
20
15
10
5
VOUT = 3.3V
fSW = 600kHz
0
5
10
15
20
25
30
35
5
10
15
20
25
30
35
5
10
15
20
25
30
35
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
VDD Voltage
vs. Temperature
VDD UVLO Threshold
vs. Temperature
VIN Shutdown Current
vs. Temperature
15
5.0
4.5
4.0
3.5
3.0
2.5
2.0
6.0
5.5
5.0
4.5
4.0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0.0
IDD = 10mA
VIN =12V
VIN =12V
VEN = 0V
IOUT = 0A
RISING
12
9
IOUT = 0A
IDD = 40mA
FALLING
6
VIN = 12V
OUT = 0A
I
3
-50
-25
0
25
50
75
100
125
-50
-25
0
25
50
75
100
125
0
-50
-25
0
25
50
75
100
125
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
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MIC2101/02
Typical Characteristics (Continued)
Output Peak Current Limit
Enable Threshold
vs. Temperature
EN Bias Current
vs. Temperature
vs. Temperature
25
2.0
1.7
1.4
1.1
0.8
0.5
8
VIN = 12V
VIN =12V
VEN = 0V
20
15
10
5
7
6
5
4
3
2
1
0
RISING
FALLING
VIN =12V
VOUT = 3.3V
fSW = 600kHz
-50
-25
0
25
50
75
100
125
0
-50
-25
0
25
50
75
100
125
-50
-25
0
25
50
75
100
125
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
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M9999-080712-A
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MIC2101/02
Functional Characteristics
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MIC2101/02
Functional Characteristics (Continued)
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MIC2101/02
Functional Characteristics (Continued)
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MIC2101/02
Functional Characteristics (Continued)
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MIC2101/02
Functional Characteristics (Continued)
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MIC2101/02
Functional Characteristics (Continued)
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MIC2101/02
Functional Diagram
Note:
ZC Detection* MIC2101 Only.
Figure 1. MIC2101/02 Functional Diagram
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Micrel, Inc.
MIC2101/02
The maximum duty cycle is obtained from the 200ns
Functional Description
tOFF(min)
:
The MIC2101/02 are adaptive on-time synchronous buck
controllers built for high-input voltage to low output
voltage applications. It is designed to operate over a
wide input voltage range from, 4.5V to 38V and the
output is adjustable with an external resistive divider. An
adaptive on-time control scheme is employed to obtain a
constant switching frequency and to simplify the control
compensation. Over-current protection is implemented
by sensing low-side MOSFET’s RDS(ON). The device
features internal soft-start, enable, UVLO, and thermal
shutdown.
tS tOFF(MIN)
200ns
tS
DMAX
1
Eq. 2
tS
where tS = 1/fSW. It is not recommended to use
MIC2101/02 with a OFF-time close to tOFF(min) during
steady-state operation.
The adaptive ON-time control scheme results in a
constant switching frequency in the MIC2101/02. The
actual ON-time and resulting switching frequency will
vary with the different rising and falling times of the
external MOSFETs. Also, the minimum tON results in a
lower switching frequency in high VIN to VOUT
applications. During load transients, the switching
frequency is changed due to the varying OFF-time.
Theory of Operation
Figure 1 illustrates the block diagram of the MIC2101/02.
The output voltage is sensed by the MIC2101/02
feedback pin FB via the voltage divider R1 and R2, and
compared to a 0.8V reference voltage VREF at the error
comparator through a low-gain transconductance (gm)
amplifier. If the feedback voltage decreases and the
amplifier output is below 0.8V, then the error comparator
will trigger the control logic and generate an ON-time
period. The ON-time period length is predetermined by
the “Fixed tON Estimator” circuitry:
To illustrate the control loop operation, we will analyze
both the steady-state and load transient scenarios. For
easy analysis, the gain of the gm amplifier is assumed to
be 1. With this assumption, the inverting input of the
error comparator is the same as the feedback voltage.
Figure 2 shows the MIC2101/02 control loop timing
during steady-state operation. During steady-state, the
gm amplifier senses the feedback voltage ripple, which is
proportional to the output voltage ripple plus injected
voltage ripple, to trigger the ON-time period. The ON-
time is predetermined by the tON estimator. The
termination of the OFF-time is controlled by the feedback
voltage. At the valley of the feedback voltage ripple,
which occurs when VFB falls below VREF, the OFF period
ends and the next ON-time period is triggered through
the control logic circuitry.
V
OUT
t
Eq. 1
ON(ESTIMATED)
V
f
IN
SW
where VOUT is the output voltage, VIN is the power stage
input voltage, and fSW is the switching frequency.
At the end of the ON-time period, the internal high-side
driver turns off the high-side MOSFET and the low-side
driver turns on the low-side MOSFET. The OFF-time
period length depends upon the feedback voltage in
most cases. When the feedback voltage decreases and
the output of the gm amplifier is below 0.8V, the ON-time
period is triggered and the OFF-time period ends. If the
OFF-time period determined by the feedback voltage is
less than the minimum OFF-time tOFF(min), which is about
200ns, the MIC2101/02 control logic will apply the
tOFF(min) instead. tOFF(min) is required to maintain enough
energy in the boost capacitor (CBST) to drive the high-
side MOSFET.
Figure 2. MIC2101/02 Control Loop Timing
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MIC2101/02
Figure 3a shows the operation of the MIC2101/02 during
a load transient. The output voltage drops due to the
sudden load increase, which causes the VFB to be less
than VREF. This will cause the error comparator to trigger
an ON-time period. At the end of the ON-time period, a
minimum OFF-time tOFF(min) is generated to charge CBST
since the feedback voltage is still below VREF. Then, the
next ON-time period is triggered due to the low feedback
voltage. Therefore, the switching frequency changes
during the load transient, but returns to the nominal fixed
frequency once the output has stabilized at the new load
current level. With the varying duty cycle and switching
frequency, the output recovery time is fast and the
output voltage deviation is small in MIC2101/02
converter.
Discontinuous Mode (MIC2101 only)
In continuous mode, the inductor current is always
greater than zero; however, at light loads the MIC2101 is
able to force the inductor current to operate in
discontinuous mode. Discontinuous mode is where the
inductor current falls to zero, as indicated by trace (IL)
shown in Figure 3b. During this period, the efficiency is
optimized by shutting down all the non-essential circuits
and minimizing the supply current. The MIC2101 wakes
up and turns on the high-side MOSFET when the
feedback voltage VFB drops below 0.8V.
The MIC2101 has a zero crossing comparator (ZC
Detection) that monitors the inductor current by sensing
the voltage drop across the low-side MOSFET during its
ON-time. If the VFB > 0.8V and the inductor current goes
slightly negative, then the MIC2101 automatically
powers down most of the IC circuitry and goes into a
low-power mode.
Once the MIC2101 goes into discontinuous mode, both
LSD and HSD are low, which turns off the high-side and
low-side MOSFETs. The load current is supplied by the
output capacitors and VOUT drops. If the drop of VOUT
causes VFB to go below VREF, then all the circuits will
wake up into normal continuous mode. First, the bias
currents of most circuits reduced during the
discontinuous mode are restored, then a tON pulse is
triggered before the drivers are turned on to avoid any
possible glitches. Finally, the high-side driver is turned
on. Figure 3b shows the control loop timing in
discontinuous mode.
Figure 3a. MIC2101/02 Load Transient Response
Unlike true current-mode control, the MIC2101/02 uses
the output voltage ripple to trigger an ON-time period.
The output voltage ripple is proportional to the inductor
current ripple if the ESR of the output capacitor is large
enough.
In order to meet the stability requirements, the
MIC2101/02 feedback voltage ripple should be in phase
with the inductor current ripple and large enough to be
sensed by the gm amplifier and the error comparator.
The recommended feedback voltage ripple is
20mV~100mV over full input voltage range. If a low-ESR
output capacitor is selected, then the feedback voltage
ripple may be too small to be sensed by the gm amplifier
and the error comparator. Also, the output voltage ripple
and the feedback voltage ripple are not necessarily in
phase with the inductor current ripple if the ESR of the
output capacitor is very low. In these cases, ripple
injection is required to ensure proper operation. Please
refer to “Ripple Injection” subsection in Application
Information for more details about the ripple injection
technique.
Figure 3b. MIC2101 Control Loop Timing
(Discontinuous Mode)
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MIC2101/02
During discontinuous mode, the bias current of most
circuits are reduced. As a result, the total power supply
current during discontinuous mode is only about 400μA,
allowing the MIC2101 to achieve high efficiency in light
load applications.
The small capacitor (CCL) connected from ILIM pin to
PGND filters the switching node ringing during the off
time allowing a better short limit measurement. The time
constant created by RCL and CCL should be much less
than the minimum off time.
The VCL drop allows programming of short limit through
the value of the resistor (RCL), If the absolute value of the
voltage drop on the bottom FET is greater than VCL’ in
that case the V(ILIM) is lower than PGND and a short
circuit event is triggered. A hiccup cycle to treat the short
event is generated. The hiccup sequence including the
soft start reduces the stress on the switching FETs and
protects the load and supply for severe short conditions.
Soft-Start
Soft-start reduces the power supply input surge current
at startup by controlling the output voltage rise time. The
input surge appears while the output capacitor is
charged up. A slower output rise time will draw a lower
input surge current.
The MIC2101/02 implements an internal digital soft-start
by making the 0.8V reference voltage VREF ramp from 0
to 100% in about 6ms with 9.7mV steps. Therefore, the
output voltage is controlled to increase slowly by a stair-
case VFB ramp. Once the soft-start cycle ends, the
related circuitry is disabled to reduce current
consumption. VDD must be powered up at the same time
or after VIN to make the soft-start function correctly.
The short circuit current limit can be programmed by
using the formula illustrated in Equation 3:
(I
∆ 0.5)R
V
DS(ON) CL
CLIM
PP
R
Eq. 3
CL
I
CL
Current Limit
Where ISH = Desired current limit
The MIC2101/02 uses the RDS(ON) and external resistor
connected from ILIM pin to SW node to decides the
current limit.
∆
PP = Inductor current peak-to-peak
R
DS (ON) = On-resistance of low-side power MOSFET
VCL = Current-limit threshold, the typical value is 14mV in
EC table
ICL = Current-limit source current, the typical value is
80µA in EC table.
In case of hard short, the short limit is folded down to
allow an indefinite hard short on the output without any
destructive effect. It is mandatory to make sure that the
inductor current used to charge the output capacitance
during soft start is under the folded short limit, otherwise
the supply will go in hiccup mode and may not be
finishing the soft start successfully.
The MOSFET RDS(ON) varies 30% to 40% with
temperature; therefore, it is recommended to add a 50%
margin to ICL in the above equation to avoid false current
limiting due to increased MOSFET junction temperature
rise. It is also recommended to connect SW pin directly
to the drain of the low-side MOSFET to accurately sense
the MOSFETs RDS(ON)
.
Figure 4. MIC2101/02 Current Limiting Circuit
In each switching cycle of the MIC2101/02 converter, the
inductor current is sensed by monitoring the low-side
MOSFET in the OFF period. The sensed voltage V(ILIM) is
compared with the power ground (PGND) after a
blanking time of 150nS. In this way the drop voltage over
the resistor RCL (VCL) is compared with the drop over the
bottom FET generating the short current limit.
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MIC2101/02
MOSFET Gate Drive
The MIC2101/02 high-side drive circuit is designed to
switch an N-Channel MOSFET. Figure 1 shows a
bootstrap circuit, consisting of D1 (a Schottky diode is
recommended) and CBST. This circuit supplies energy to
the high-side drive circuit. Capacitor CBST is charged
while the low-side MOSFET is on and the voltage on the
SW pin is approximately 0V. When the high-side
MOSFET driver is turned on, energy from CBST is used to
turn the MOSFET on. As the high-side MOSFET turns
on, the voltage on the SW pin increases to
approximately VIN. Diode D1 is reverse biased and CBST
floats high while continuing to keep the high-side
MOSFET on. The bias current of the high-side driver is
less than 10mA so a 0.1μF to 1μF is sufficient to hold
the gate voltage with minimal droop for the power stroke
(high-side switching) cycle, i.e. ∆BST = 10mA x
3.33μs/0.1μF = 333mV. When the low-side MOSFET is
turned back on, CBST is recharged through D1. A small
resistor RG, which is in series with CBST, can be used to
slow down the turn-on time of the high-side N-channel
MOSFET.
The drive voltage is derived from the VDD supply voltage.
The nominal low-side gate drive voltage is VDD and the
nominal high-side gate drive voltage is approximately
VDD – VDIODE, where VDIODE is the voltage drop across
D1. An approximate 30ns delay between the high-side
and low-side driver transitions is used to prevent current
from simultaneously flowing unimpeded through both
MOSFETs.
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MIC2101/02
MOSFET Selection
Application Information
The MIC2101/02 controllers work from input voltages of
4.5V to 38V and has internal 5V VDD LDO. This internal
VDD LDO provides power to turn the external N-Channel
power MOSFETs for the high-side and low-side switches.
For applications where VDD < 5V, it is necessary that the
power MOSFETs used are sub-logic level and are in full
conduction mode for VGS of 2.5V. For applications when
VDD > 5V; logic-level MOSFETs, whose operation is
specified at VGS = 4.5V must be used.
Setting the Switching Frequency
The MIC2101/02 are adjustable-frequency, synchronous
buck controllers featuring a unique adaptive on-time
control architecture. The switching frequency can be
adjusted between 200kHz and 600kHz by changing the
resistor divider network consisting of R19 and R20.
There are different criteria for choosing the high-side and
low-side MOSFETs. These differences are more
significant at lower duty cycles. In such an application,
the high-side MOSFET is required to switch as quickly
as possible to minimize transition losses, whereas the
low-side MOSFET can switch slower, but must handle
larger RMS currents. When the duty cycle approaches
50%, the current carrying capability of the high-side
MOSFET starts to become critical.
It is important to note that the on-resistance of a
MOSFET increases with increasing temperature. A 75°C
rise in junction temperature will increase the channel
resistance of the MOSFET by 50% to 75% of the
resistance specified at 25°C. This change in resistance
must be accounted for when calculating MOSFET power
dissipation and in calculating the value of current limit.
Total gate charge is the charge required to turn the
MOSFET on and off under specified operating conditions
(VDS and VGS). The gate charge is supplied by the
MIC2101/02 gate-drive circuit. At 600kHz switching
frequency, the gate charge can be a significant source of
power dissipation in the MIC2101/02. At low output load,
this power dissipation is noticeable as a reduction in
efficiency. The average current required to drive the
high-side MOSFET is:
Figure 5. Switching Frequency Adjustment
The following formula gives the estimated switching
frequency:
R20
fSW_ADJ fO
Eq. 4
R19R20
Where fO = Switching Frequency when R19 is 100k and
R20 being open, fO is typically 600kHz. For more precise
setting, it is recommended to use the following graph:
I
(AVG) Q f
SW
Eq. 5
G[HIGH-SIDE]
G
Switching Frequency
700.00
where:
R19 = 100k, IOUT =12A
IG[HIGHSIDE](avg) = Average high-side MOSFET gate
current
600.00
VIN = 12V
500.00
QG = Total gate charge for the high-side MOSFET taken
from the manufacturer’s data sheet for VGS = VDD.
400.00
VIN =38V
300.00
200.00
100.00
0.00
fSW = Switching Frequency
The low-side MOSFET is turned on and off at VDS = 0
because an internal body diode or external freewheeling
diode is conducting during this time. The switching loss
for the low-side MOSFET is usually negligible. Also, the
gate-drive current for the low-side MOSFET is more
accurately calculated using CISS at VDS = 0 instead of
gate charge.
10.00
100.00
1000.00
10000.00
R20 (k Ohm)
Figure 6. Switching Frequency vs. R20
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MIC2101/02
For the low-side MOSFET:
Making the assumption that the turn-on and turn-off
transition times are equal; the transition times can be
approximated by:
I
(AVG) C
V f
SW
Eq. 6
G[LOW-SIDE]
ISS
GS
CISS VIN COSS VHSD
tT
Eq.11
Since the current from the gate drive comes from the
VDD, the power dissipated in the MIC2101/02 due to gate
drive is:
IG
where:
CISS and COSS are measured at VDS = 0
IG = Gate-drive current
P
V (I
(AVG)
G[HIGH-SIDE]
GATEDRIVE
DD
Eq. 7
I
(AVG))
G[LOW-SIDE]
The total high-side MOSFET switching loss is:
A convenient figure of merit for switching MOSFETs is
the on resistance times the total gate charge RDS(ON)
×
PAC (VHSD VD )IPK tT fSW
Eq. 12
QG. Lower numbers translate into higher efficiency. Low
gate-charge logic-level MOSFETs are a good choice for
use with the MIC2101/02. Also, the RDS(ON) of the low-
side MOSFET will determine the current-limit value.
Please refer to “Current Limit” subsection is Functional
Description for more details.
where:
tT = Switching transition time
VD = Body diode drop (0.5V)
fSW = Switching Frequency
Parameters that are important to MOSFET switch
selection are:
Voltage rating
On-resistance
Total gate charge
The high-side MOSFET switching losses increase with
the switching frequency and the input voltage VHSD. The
low-side MOSFET switching losses are negligible and
can be ignored for these calculations.
Inductor Selection
The voltage ratings for the high-side and low-side
MOSFETs are essentially equal to the power stage input
voltage VHSD. A safety factor of 20% should be added to
the VDS(MAX) of the MOSFETs to account for voltage
spikes due to circuit parasitic elements.
Values for inductance, peak, and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine the
peak-to-peak inductor ripple current. Generally, higher
inductance values are used with higher input voltages.
Larger peak-to-peak ripple currents will increase the
power dissipation in the inductor and MOSFETs. Larger
output ripple currents will also require more output
capacitance to smooth out the larger ripple current.
Smaller peak-to-peak ripple currents require a larger
inductance value and therefore a larger and more
expensive inductor.
The power dissipated in the MOSFETs is the sum of the
conduction losses during the on-time (PCONDUCTION) and
the switching losses during the period of time when the
MOSFETs turn on and off (PAC).
PSW PCONDUCTION PAC
Eq.8
A good compromise between size, loss and cost is to set
the inductor ripple current to be equal to 20% of the
maximum output current.
PCONDUCTION ISW(RMS)2 RDS(ON)
Eq. 9
PAC PAC(off ) PAC(on)
Eq. 10
where:
RDS(ON) = On-resistance of the MOSFET switch
D = Duty Cycle = VOUT / VHSD
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MIC2101/02
The inductance value is calculated by Equation 13:
Copper loss in the inductor is calculated by Equation 17:
P
INDUCTOR(Cu) = IL(RMS)2 RWINDING
Eq. 17
VOUT (VIN(MAX) VOUT
)
L
Eq. 13
VIN(MAX) fsw 20%IOUT(MAX)
The resistance of the copper wire, RWINDING, increases
with the temperature. The value of the winding
resistance used should be at the operating temperature.
where:
fSW = Switching frequency
20% = Ratio of AC ripple current to DC output current
VIN(MAX) = Maximum power stage input voltage
P
WINDING(Ht) = RWINDING(20°C)
(1 + 0.0042 × (TH – T20°C))
Eq. 18
The peak-to-peak inductor current ripple is:
where:
TH = temperature of wire under full load
T20°C = ambient temperature
VOUT (VIN(MAX) VOUT
VIN(MAX) fsw L
)
∆IL(PP)
Eq. 14
RWINDING(20°C) = room temperature winding resistance
(usually specified by the manufacturer)
Output Capacitor Selection
The peak inductor current is equal to the average output
current plus one half of the peak-to-peak inductor current
ripple.
The type of the output capacitor is usually determined by
its equivalent series resistance (ESR). Voltage and RMS
current capability are two other important factors for
selecting the output capacitor. Recommended capacitor
types are tantalum, low-ESR aluminum electrolytic, OS-
CON and POSCAP. The output capacitor’s ESR is
usually the main cause of the output ripple. The output
capacitor ESR also affects the control loop from a
stability point of view. The maximum value of ESR is
calculated:
IL(pk) =IOUT(MAX) 0.5 ∆IL(PP)
Eq. 15
The RMS inductor current is used to calculate the I2R
losses in the inductor.
2
∆IL(PP)
2
IL(RMS) IOUT(MAX)
Eq. 16
12
∆VOUT(pp)
ESRC
Eq. 19
OUT
∆IL(PP)
Maximizing efficiency requires the proper selection of
core material and minimizing the winding resistance. The
high frequency operation of the MIC2101/02 requires the
use of ferrite materials for all but the most cost sensitive
applications. Lower cost iron powder cores may be used
but the increase in core loss will reduce the efficiency of
the power supply. This is especially noticeable at low
output power. The winding resistance decreases
efficiency at the higher output current levels. The
winding resistance must be minimized although this
usually comes at the expense of a larger inductor. The
power dissipated in the inductor is equal to the sum of
the core and copper losses. At higher output loads, the
core losses are usually insignificant and can be ignored.
At lower output currents, the core losses can be a
significant contributor. Core loss information is usually
available from the magnetics vendor.
where:
ΔVOUT(pp) = peak-to-peak output voltage ripple
∆IL(PP) = peak-to-peak inductor current ripple
26
M9999-080712-A
August 2012
Micrel, Inc.
MIC2101/02
The total output ripple is a combination of the ESR and
output capacitance. The total ripple is calculated in
Equation 20:
Input Capacitor Selection
The input capacitor for the power stage input VIN should
be selected for ripple current rating and voltage rating.
Tantalum input capacitors may fail when subjected to
high inrush currents, caused by turning the input supply
on. A tantalum input capacitor’s voltage rating should be
at least two times the maximum input voltage to
maximize reliability. Aluminum electrolytic, OS-CON, and
multilayer polymer film capacitors can handle the higher
inrush currents without voltage de-rating. The input
voltage ripple will primarily depend on the input
capacitor’s ESR. The peak input current is equal to the
peak inductor current, so:
2
∆IL(PP)
2
∆VOUT(pp)
∆IL(PP) ESRC
OUT
COUT fSW 8
Eq. 20
where:
D = duty cycle
COUT = output capacitance value
fsw = switching frequency
∆VIN = IL(pk) × ESRCIN
Eq. 23
As described in the “Theory of Operation” subsection in
Functional Description, the MIC2101/02 requires at least
20mV peak-to-peak ripple at the FB pin to make the gm
amplifier and the error comparator behave properly. Also,
the output voltage ripple should be in phase with the
inductor current. Therefore, the output voltage ripple
caused by the output capacitors value should be much
smaller than the ripple caused by the output capacitor
ESR. If low-ESR capacitors, such as ceramic capacitors,
are selected as the output capacitors, a ripple injection
method should be applied to provide the enough
feedback voltage ripple. Please refer to the “Ripple
Injection” subsection for more details.
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming
the peak-to-peak inductor current ripple is low:
ICIN(RMS) IOUT(max) D(1D)
Eq. 24
The power dissipated in the input capacitor is:
P
DISS(CIN) = ICIN(RMS)2 × ESRCIN
Eq. 25
The voltage rating of the capacitor should be twice the
output voltage for a tantalum and 20% greater for
aluminum electrolytic or OS-CON. The output capacitor
RMS current is calculated in Equation 21:
Voltage Setting Components
The MIC2101/02 requires two resistors to set the output
voltage as shown in Figure 7:
∆IL(PP)
IC
Eq. 21
OUT (RMS)
12
The power dissipated in the output capacitor is:
2
PDISS(C
IC
ESRC
Eq. 22
)
OUT (RMS)
OUT
OUT
Figure 7. Voltage-Divider Configuration
27
M9999-080712-A
August 2012
Micrel, Inc.
MIC2101/02
The output voltage is determined by the equation:
2. Inadequate ripple at the feedback voltage due to the
small ESR of the output capacitors.
The output voltage ripple is fed into the FB pin
through a feedforward capacitor Cff in this situation,
as shown in Figure 8b. The typical Cff value is
between 1nF and 100nF. With the feedforward
capacitor, the feedback voltage ripple is very close
to the output voltage ripple:
R1
VOUT VFB (1
)
Eq. 26
R2
where, VFB = 0.8V. A typical value of R1 can be between
3kꢀ and 10kꢀ. If R1 is too large, it may allow noise to be
introduced into the voltage feedback loop. If R1 is too
small in value, it will decrease the efficiency of the power
supply, especially at light loads. Once R1 is selected, R2
can be calculated using:
∆VFB(pp) ESR ∆IL
Eq. 29
(pp)
3. Virtually no ripple at the FB pin voltage due to the
very-low ESR of the output capacitors:
VFB R1
VOUT VFB
R2
Eq. 27
Ripple Injection
The VFB ripple required for proper operation of the
MIC2101/02 gm amplifier and error comparator is 20mV
to 100mV. However, the output voltage ripple is
generally designed as 1% to 2% of the output voltage.
For a low output voltage, such as a 1V, the output
voltage ripple is only 10mV to 20mV, and the feedback
voltage ripple is less than 20mV. If the feedback voltage
ripple is so small that the gm amplifier and error
comparator can’t sense it, then the MIC2101/02 will lose
control and the output voltage is not regulated. In order
to have some amount of VFB ripple, a ripple injection
method is applied for low output voltage ripple
applications.
Figure 8a. Enough Ripple at FB
The applications are divided into three situations
according to the amount of the feedback voltage ripple:
1. Enough ripple at the feedback voltage due to the
large ESR of the output capacitors.
As shown in Figure 8a, the converter is stable
without any ripple injection. The feedback voltage
ripple is:
Figure 8b. Inadequate Ripple at FB
R2
∆VFB(pp)
ESRC
∆IL
Eq. 28
(pp)
OUT
R1 R2
where ∆IL(pp) is the peak-to-peak value of the
inductor current ripple.
Figure 8c. Invisible Ripple at FB
28
M9999-080712-A
August 2012
Micrel, Inc.
MIC2101/02
In this situation, the output voltage ripple is less than
20mV. Therefore, additional ripple is injected into the FB
pin from the switching node SW via a resistor RINJ and a
capacitor Cinj, as shown in Figure 8c. The injected ripple
is:
The process of sizing the ripple injection resistor and
capacitors is:
Step 1. Select Cff to feed all output ripples into the
feedback pin and make sure the large time constant
assumption is satisfied. Typical choice of Cff is 1nF to
100nF if R1 and R2 are in kꢀ range.
1
Step 2. Select Rinj according to the expected feedback
voltage ripple using Equation 35:
∆VFB(pp) VIN Kdiv D(1- D)
Eq.30
fSW
∆VFB(pp)
fSW
D (1 D)
R1//R2
Kdiv
Eq. 33
Kdiv
Eq. 31
V
IN
RINJ R1//R2
Then the value of RINJ is obtained as:
1
where:
VIN = Power stage input voltage
D = Duty cycle
RINJ (R1//R2)(
1)
Eq. 34
Kdiv
fSW = Switching frequency
τ = (R1//R2//Rinj) Cff
Step 3. Select Cinj as 100nF, which could be considered
as short for a wide range of the frequencies.
In Equations 30 and 32, it is assumed that the time
constant associated with Cff must be much greater than
the switching period:
1
T
1
Eq. 32
fSW
If the voltage divider resistors R1 and R2 are in the kꢀ
range, a Cff of 1nF to 100nF can easily satisfy the large
time constant requirements. Also, a 100nF injection
capacitor CINJ is used in order to be considered as short
for a wide range of the frequencies.
29
M9999-080712-A
August 2012
Micrel, Inc.
MIC2101/02
Inductor
PCB Layout Guidelines
Keep the inductor connection to the switch node
(SW) short.
Warning!!! To minimize EMI and output noise, follow
these layout recommendations.
Do not route any digital lines underneath or close to
the inductor.
PCB Layout is critical to achieve reliable, stable and
efficient performance. A ground plane is required to
control EMI and minimize the inductance in power,
signal and return paths.
Keep the switch node (SW) away from the feedback
(FB) pin.
The following guidelines should be followed to insure
proper operation of the MIC2101/02 converter.
The SW pin should be connected directly to the
drain of the low-side MOSFET to accurate sense the
voltage across the low-side MOSFET.
IC
To minimize noise, place a ground plane underneath
the inductor.
The 4.7µF ceramic capacitors, which are connected
to the VDD and PVDD pins, must be located right at
the IC. The VDD pin is very noise sensitive and
placement of the capacitor is very critical. Use wide
traces to connect to the VDD, PVDD and AGND,
PGND pins respectively.
Output Capacitor
Use a wide trace to connect the output capacitor
ground terminal to the input capacitor ground
terminal.
Phase margin will change as the output capacitor
value and ESR changes. Contact the factory if the
output capacitor is different from what is shown in
the BOM.
The signal ground pin (AGND) must be connected
directly to the ground planes. Do not route the
AGND pin to the PGND pin on the top layer.
Place the IC close to the point of load (POL).
The feedback trace should be separate from the
power trace and connected as close as possible to
the output capacitor. Sensing a long high-current
load trace can degrade the DC load regulation.
Use fat traces to route the input and output power
lines.
Signal and power grounds should be kept separate
and connected at only one location.
MOSFETs
Input Capacitor
Low-side MOSFET gate drive trace (DL pin to
Place the input capacitor next.
MOSFET gate pin) must be short and routed over a
ground plane. The ground plane should be the
connection between the MOSFET source and PGND.
Place the input capacitors on the same side of the
board and as close to the MOSFETs as possible.
Chose a low-side MOSFET with a high CGS/CGD ratio
and a low internal gate resistance to minimize the
effect of dv/dt inducted turn-on.
Place several vias to the ground plane close to the
input capacitor ground terminal.
Use either X7R or X5R dielectric input capacitors.
Do not use Y5V or Z5U type capacitors.
Do not put a resistor between the Low-side
MOSFET gate drive output and the gate.
Do not replace the ceramic input capacitor with any
other type of capacitor. Any type of capacitor can be
placed in parallel with the input capacitor.
Use a 4.5V VGS rated MOSFET. Its higher gate
threshold voltage is more immune to glitches than a
2.5V or 3.3V rated MOSFET. MOSFETs that are
rated for operation at less than 4.5V VGS should not
be used.
If a Tantalum input capacitor is placed in parallel
with the input capacitor, it must be recommended for
switching regulator applications and the operating
voltage must be derated by 50%.
In “Hot-Plug” applications, a Tantalum or Electrolytic
bypass capacitor must be used to limit the over-
voltage spike seen on the input supply with power is
suddenly applied.
RC Snubber
Place the RC snubber on the same side of the board
and as close to the SW pin as possible.
30
M9999-080712-A
August 2012
Micrel, Inc.
MIC2101/02
Evaluation Board Schematic
Figure 9. Schematic of MIC2101/02 Evaluation Board
(J1, J9, J12, R14, and R21 are for testing purposes)
31
M9999-080712-A
August 2012
Micrel, Inc.
MIC2101/02
Bill of Materials
Item
Part Number
Manufacturer
Panasonic(1)
AVX(2)
TDK(3)
Murata(4)
AVX
Description
Qty
C1
EEU-FC1J221S
220µF Aluminum Capacitor, 63V
1
12105C225KAT2A
C3225X7R1H225K
GRM32ER60J107ME20L
12106D107MAT2A
C3225X5ROJ107M
GRM188R71H104KA93D
06035C104KAT2A
C1608X7R1H104K
GRM188R60J475KE19D
06036D475KAT2A
C1608X5R0J475K
GRM188R70J105KA01D
06036C105KAT2A
C1608X5R0J105K
GRM21BR72A474KA73
08051C474KAT2A
GRM188R71H102KA01D
06035C102KAT2A
C1608X7R1H102K
GRM188R71H472MA01D
06035C472KAT2A
C1608X7R1H472K
6SEPC470MX
C2, C3, C4
2.2µF/50V Ceramic Capacitor, X7R, Size 1210
3
C14
100µF/6.3V Ceramic Capacitor, X7R, Size 1210
1
TDK
Murata
AVX
C6, C16, C10
C7, C17
0.1µF/50V Ceramic Capacitor, X7R, Size 0603
4.7µF/6.3V Ceramic Capacitor, X7R, Size 0603
3
2
TDK
Murata
AVX
TDK
Murata
AVX
C8
1µF/6.3V Ceramic Capacitor, X7R, Size 0603
0.47µF/100V,X7R,0805
1
1
1
TDK
Murata
AVX
C9
Murata
AVX
C11
1nF/50V Cermiac Capacitor, X7R, Size 0603
TDK
Murata
AVX
C12
C13
4.7nF/50V Cermiac Capacitor, X7R, Size 0603
1
1
TDK
Sanyo(5)
Sanyo
Sanyo
Murata
Murata
AVX
MCC(6)
Sumida(7)
Infineon(8)
470µF/6.3V, 7m, OSCON
6SEPC470M
470µF/6.3V, 7m, OSCON
C15 (OPEN)
C5 (OPEN)
6TPB470M
470µF/6.3V, POSCAP
GRM32ER60J107ME20L
GRM1885C1H150JA01D
06035A150JAT2A
BAT46W-TP
100µF/6.3V Ceramic Capacitor, X7R, Size 1210
C18
15pF, 50V, 0603, NPO
1
D1
L1
100V Small Signal Schottky Diode, SOD123
1.5µH, 27/22Asat, 20Arms for 40C rise
MOSFET, N-CH, Power SO-8
1
1
2
CDEP147NP- 1R5M-95
Q1, Q3
BSC067N06LS3
Notes:
1. Panasonic: www.panasonic.com.
2. AVX: www.avx.com
3. TDK: www.tdk.com.
4. Murata: www.murata.com.
5. Sanyo: www.sanyo.com.
6. MCC.: www.mccsemi.com.
7. Sumida: www.sumida.com
8. Infineon: www.infineon.com.
32
M9999-080712-A
August 2012
Micrel, Inc.
MIC2101/02
Bill of Materials (Continued)
Item
Part Number
Manufacturer
Description
Qty.
Q2, Q4 (OPEN)
R1
CRCW060310K0FKEA
CRCW08051R21FKEA
CRCW06035K23FKEA
CRCW060380K6FKEA
CRCW060340K2FKEA
CRCW060320K0FKEA
CRCW060311K5FKEA
CRCW06038K06FKEA
CRCW06034K75FKEA
CRCW06033K24FKEA
CRCW06031K91FKEA
CRCW0603715R0FKEA
CRCW0603348R0FKEA
CRCW06030000FKEA
CRCW08052R0FKEA
CRCW06031K65FKEA
CRCW060349K9FKEA
No Load
Vishay Dale(9)
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay/Dale
10kꢀ Resistor, Size 0603, 1%
1.21ꢀ Resistor, Size 0805, 5%
5.23K,1%,1/10W,0603.
1
2
1
1
1
1
1
1
1
1
1
R2, R23
R3
R4
80.6kꢀ Resistor, Size 0603, 1%
40.2kꢀ Resistor, Size 0603, 1%
20kꢀ Resistor, Size 0603, 1%
11.5kꢀ Resistor, Size 0603, 1%
8.06kꢀ Resistor, Size 0603, 1%
4.75kꢀ Resistor, Size 0603, 1%
3.24kꢀ Resistor, Size 0603, 1%
1.91kꢀ Resistor, Size 0603, 1%
715ꢀ Resistor, Size 0603, 1%
348ꢀ Resistor, Size 0603, 1%
0ꢀ Resistor, Size 0603, 5%
2ꢀ Resistor, Size 0805, 5%
1.65kꢀ Resistor, Size 0603, 1%
49.9K,1%,1/10W,0603
R5
R6
R7
R8
R9
R10
R11
R12 (OPEN)
R13 (OPEN)
R14, R15, R19
R16
3
1
1
1
R17
R18
R20 (OPEN)
R21
CRCW060349R9FKEA
CRCW0603100KFKEA
MIC2101YML
Vishay Dale
Vishay Dale
49.9ꢀ Resistor, Size 0603, 1%
100kꢀ Resistor, Size 0603, 1%
1
1
R22
U1
Micrel. Inc.(10)
38V Synchronous Buck DC/DC Controller
1
MIC2102YML
Notes:
9. Vishay: www.vishay.com.
10. Micrel, Inc.: www.micrel.com.
33
M9999-080712-A
August 2012
Micrel, Inc.
MIC2101/02
PCB Layout
Figure 10. MIC2101/02 Evaluation Board Top Layer
Figure 11. MIC2101/02 Evaluation Board Mid-Layer 1 (Ground Plane)
34
M9999-080712-A
August 2012
Micrel, Inc.
MIC2101/02
PCB Layout (Continued)
Figure 12. MIC2101/02 Evaluation Board Mid-Layer 2
Figure 13. MIC2101/02 Evaluation Board Bottom Layer
35
M9999-080712-A
August 2012
Micrel, Inc.
MIC2101/02
Recommended Land Pattern
`
Red circle indicates Thermal Via. Size should be .300mm .350mm in diameter
and it should be connected to GND plane for maximum thermal performance.
ALL UNITS ARE IN mm, TOLERANCE 0.05, IF NOT NOTED
LP # MLF33D-16LD-LP-1
36
M9999-080712-A
August 2012
Micrel, Inc.
MIC2101/02
Package Information
16-Pin 3mm 3mm MLF (ML)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
Micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this data sheet. This
information is not intended as a warranty and Micrel does not assume responsibility for its use. Micrel reserves the right to change circuitry,
specifications and descriptions at any time without notice. No license, whether express, implied, arising by estoppel or otherwise, to any intellectual
property rights is granted by this document. Except as provided in Micrel’s terms and conditions of sale for such products, Micrel assumes no liability
whatsoever, and Micrel disclaims any express or implied warranty relating to the sale and/or use of Micrel products including liability or warranties
relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright or other intellectual property right
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant
into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully
indemnify Micrel for any damages resulting from such use or sale.
© 2012 Micrel, Incorporated.
37
M9999-080712-A
August 2012
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