MIC2171WUTR [MICREL]
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| 型号： | MIC2171WUTR |
| 厂家： | 
MICREL SEMICONDUCTOR |
| 描述： | 暂无描述 稳压器 开关 |
| 文件： | 总10页 (文件大小：84K) |
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MIC2171
100kHz 2.5A Switching Regulator
Preliminary Information
General Description
Features
The MIC2171 is a complete 100kHz SMPS current-mode
controller with an internal 65V 2.5A power switch.
• 2.5A, 65V internal switch rating
• 3V to 40V input voltage range
• Current-mode operation, 2.5A peak
• Internal cycle-by-cycle current limit
• Thermal shutdown
• Twice the frequency of the LM2577
• Low external parts count
Although primarily intended for voltage step-up applications,
the floating switch architecture of the MIC2171 makes it
practical for step-down, inverting, and Cuk configurations as
well as isolated topologies.
Operating from 3V to 40V, the MIC2171 draws only 7mA of
quiescent current, making it attractive for battery operated
supplies.
• Operates in most switching topologies
• 7mA quiescent current (operating)
• Fits LT1171/LM2577 TO-220 and TO-263 sockets
Applications
• Laptop/palmtop computers
• Battery operated equipment
• Hand-held instruments
The MIC2171 is available in a 5-pin TO-220 or TO-263 for
–40°C to +85°C operation.
• Off-line converter up to 50W
(requires external power switch)
• Predriver for higher power capability
4
Typical Applications
+5V
VOUT
5V, 0.5A
VIN
4V to 6V
(4.75V min.)
C1*
L1
T1
47µF
15µH
D2
1N5818
C1
R4*
C3*
D1*
47µF
VOUT
+12V, 0.25A
R1
D1
IN
C4
470µF
3.74k
1%
SW
FB
IN
R1
10.7k
1%
1N5822
MIC2171
SW
1:1.25
PRI = 12µH
L
MIC2171
COMP
R2
R3
1k
C2
470µF
GND
1.24k
COMP
FB
1%
R2
C3
1µF
R3
1k
GND
1.24k
1%
C2
1µF
* Locate near MIC2171 when supply leads > 2"
* Optional voltage clipper (may be req’d if T1 leakage inductance too high)
Figure 1.
Figure 2.
MIC2171 5V to 12V Boost Converter
MIC2171 5V Flyback Converter
1997
4-3
MIC2171
Micrel
Ordering Information
Part Number
MIC2171BT
MIC2171BU
Temperature Range
Package
–40°C to +85°C
–40°C to +85°C
5-lead TO-220
5-lead TO-263
Pin Configuration
5 IN
5 IN
4 SW
3 GND
2 FB
4 SW
3 GND
2 FB
1 COMP
1 COMP
Tab GND
Tab GND
5-lead TO-220 (BT)
5-lead TO-263 (BU)
Pin Description
Pin Number
Pin Name
Pin Function
1
COMP
Frequency Compensation: Output of transconductance-type error amplifier.
Primary function is for loop stabilization. Can also be used for output voltage
soft-start and current limit tailoring.
2
3
4
5
FB
GND
SW
IN
Feedback: Inverting input of error amplifier. Connect to external resistive
divider to set power supply output voltage.
Ground: Connect directly to the input filter capacitor for proper operation
(see applications info).
Power Switch Collector: Collector of NPN switch. Connect to external
inductor or input voltage depending on circuit topology.
Supply Voltage: 3.0V to 40V
4-4
1997
MIC2171
Micrel
Junction Temperature ................................ –55°C to 150°C
Thermal Resistance
Absolute Maximum Ratings
Input Voltage (V ) ........................................................40V
IN
θ
θ
5-lead TO-220, Note 1.................................45°C/W
5-lead TO-263, Note 2.................................45°C/W
JA
JA
Switch Voltage (V ) ....................................................65V
SW
Feedback Voltage (transient, 1ms) (V )................... ±15V
FB
Storage Temperature ............................... –65°C to +150°C
Soldering (10 sec.) .................................................. +300°C
Operating Temperature Range ...................... –40 to +85°C
Electrical Characteristics
VIN = 5V; TA = 25°C, bold values indicate –40°C ≤ TA ≤ +85°C; unless noted.
Parameter
Conditions
Min
Typ
Max
Units
Reference Section
Feedback Voltage (VFB
)
VCOMP = 1.24V
1.220 1.240 1.264
V
V
1.214
1.274
Feedback Voltage
Line Regulation
3V ≤ VIN ≤ 40V
VCOMP = 1.24V
.06
%/V
Feedback Bias Current (IFB
)
VFB = 1.24V
310
750
1100
nA
nA
Error Amplifier Section
Transconductance (gm)
∆ICOMP = ±25µA
3.0
2.4
3.9
6.0
7.0
µA/mV
µA/mV
Voltage Gain (AV)
Output Current
0.9V ≤ VCOMP ≤ 1.4V
400
800
175
2000
V/V
4
VCOMP = 1.5V
125
100
350
400
µA
µA
Output Swing
High Clamp, VFB = 1V
Low Clamp, VFB = 1.5V
1.8
0.25
2.1
0.35
2.3
0.52
V
V
Compensation Pin
Threshold
Duty Cycle = 0
0.8
0.6
0.9
1.08
1.25
V
V
Output Switch Section
ON Resistance
I
SW = 2A, VFB = 0.8V
0.37
0.50
0.55
Ω
Ω
Current Limit
Duty Cycle = 50%, TJ ≥ 25°C
Duty Cycle = 50%, TJ < 25°C
Duty Cycle = 80%, Note 3
2.5
2.5
2.0
3.6
4.0
3.0
5
5.5
5
A
A
A
Breakdown Voltage (BV)
3V ≤ VIN ≤ 40V
65
75
V
ISW = 5mA
Oscillator Section
Frequency (fO)
88
85
100
90
112
115
kHz
kHz
Duty Cycle [δ(max)]
80
95
%
Input Supply Voltage Section
Minimum Operating Voltage
Quiescent Current (IQ)
2.7
7
3.0
9
V
3V ≤ VIN ≤ 40V, VCOMP = 0.6V, ISW = 0
mA
mA
Supply Current Increase (∆IIN)
∆ISW = 2A, VCOMP = 1.5V, during swich on-time
9
20
General Note Devices are ESD sensitive. Handling precautions required.
Note 1 Mounted vertically, no external heat sink, 1/4 inch leads soldered to PC board containing approximently 4 inch squared copper area
surrounding leads.
Note 2 All ground leads soldered to approximently 2 inches squared of horizontal PC board copper area.
Note 3 For duty cycles (δ) between 50% and 95%, minimum guaranteed switch current is I = 1.66 (2-δ) Amp (Pk).
CL
1997
4-5
MIC2171
Micrel
Typical Performance Characteristics
Feedback Voltage
Line Regulation
Minimum
Operating Voltage
Feedback Bias Current
5
4
2.9
2.8
2.7
2.6
2.5
2.4
2.3
800
700
600
500
400
300
200
100
0
T
= 125°C
3
J
2
1
0
T
= 25°C
J
Switch Current = 2A
-1
-2
-3
-4
-5
T
= -40°C
J
0
10
V
20
30
40
-100 -50
0
50
100 150
-100 -50
0
50
100 150
Operating (V)
Temperature (°C)
Temperature (°C)
IN
Supply Current
Supply Current
Supply Current
15
14
13
12
11
10
9
10
9
8
7
6
5
4
3
2
1
0
50
40
30
20
10
0
I
= 0
VCOMP = 0.6V
SW
D.C. = 90%
δ = 90%
D.C. = 50%
D.C. = 0%
8
δ = 50%
7
6
5
0
10
20
30
40
-100 -50
0
50
100 150
0
1
2
3
4
Temperature (°C)
V
Operating Voltage (V)
Switch Current (A)
IN
Oscillator Frequency
Current Limit
Switch On-Voltage
1.6
1.4
1.2
1.0
0.8
0.6
0.4
0.2
0
120
110
100
90
8
6
T
= 25°C
J
T
= –40°C
J
25°C
–40°C
4
2
0
80
125°C
T
= 125°C
J
70
60
0
1
2
3
-50
0
50
100
150
0
20
40
60
80
100
Temperature (°C)
Switch Current (A)
Duty Cycle (%)
Error Amplifier Gain
Error Amplifier Gain
Error Amplifier Phase
5.0
4.5
4.0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0.0
7000
6000
5000
4000
3000
2000
1000
0
-30
0
30
60
90
120
150
180
210
-100 -50
0
50
100 150
1
10
100
1000 10000
1
10
100
1000 10000
Temperature (°C)
Frequency (kHz)
Frequency (kHz)
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1997
MIC2171
Micrel
Block Diagram MIC2171
D1
IN
2.3V
SW
Reg.
Anti-Sat.
Driver
100kHz
Osc.
Logic
Q1
Com-
parator
FB
Current
Amp.
Error
Amp.
1.24V
Ref.
COMP
GND
4
response. Inherent cycle-by-cycle current limiting greatly
improves the power switch reliability and provides automatic
output current limiting. Finally, current-mode operation pro-
vides automatic input voltage feed forward which prevents
instantaneous input voltage changes from disturbing the
output voltage setting.
Functional Description
Refer to “Block Diagram MIC2171”.
Internal Power
The MIC2171 operates when V is ≥ 2.6V. An internal 2.3V
IN
regulator supplies biasing to all internal circuitry including a
precision 1.24V band gap reference.
Anti-Saturation
The anti-saturation diode (D1) increases the usable duty
cycle range of the MIC2171 by eliminating the base to
collector stored charge which would delay Q1’s turnoff.
PWM Operation
The 100kHz oscillator generates a signal with a duty cycle of
approximately 90%. The current-mode comparator output is
used to reduce the duty cycle when the current amplifier
output voltage exceeds the error amplifier output voltage.
The resulting PWM signal controls a driver which supplies
base current to output transistor Q1.
Compensation
Loop stability compensation of the MIC2171 can be accom-
plished by connecting an appropriate network from either
COMP to circuit ground (see typical Applications) or COMP
to FB.
Current-Mode Advantages
The error amplifier output (COMP) is also useful for soft start
and current limiting. Because the error amplifier output is a
transconductance type, the output impedance is relatively
high which means the output voltage can be easily clamped
or adjusted externally.
The MIC2171 operates in current mode rather than voltage
mode. There are three distinct advantages to this technique.
Feedback loop compensation is greatly simplified because
inductor current sensing removes a pole from the closed loop
1997
4-7
MIC2171
Micrel
The device operating losses are the dc losses associated
with biasing all of the internal functions plus the losses of the
power switch driver circuitry. The dc losses are calculated
from the supply voltage (V ) and device supply current (I ).
The MIC2171 supply current is almost constant regardless of
the supply voltage (see “Electrical Characteristics”). The
driver section losses (not including the switch) are a function
of supply voltage, power switch current, and duty cycle.
Applications Information
Soft Start
A diode-coupled capacitor from COMP to circuit ground
slows the output voltage rise at turn on (Figure 3).
IN
Q
VIN
IN
P
= V
I
+ V
(
× I
× ∆I
(
)
)
(bias+driver)
IN Q
IN(min)
SW
IN
MIC2171
where:
P
V
= device operating losses
COMP
(bias+driver)
= supply voltage = V – V
SW
IN(min)
IN
D1
D2
C1
I = typical quiescent supply current
Q
R1
C2
I
= power switch current limit
CL
∆I = typical supply current increase
IN
As a practical example refer to Figure 1.
Figure 3. Soft Start
V
= 5.0V
IN
Theadditionaltimeittakesfortheerroramplifiertochargethe
capacitor corresponds to the time it takes the output to reach
regulation. Diode D1 discharges C1 when V is removed.
I = 0.007A
Q
I
= 2.21A
CL
IN
δ = 66.2% (0.662)
Current Limit
Then:
V
= 5 – (2.21 × 0.37) = 4.18V
VIN
IN(min)
IN
P
= (5 × 0.007) + (4.18 × 2.21 × .009)
SW
(bias + driver)
P
= 0.1W
MIC2171
(bias+driver)
Power switch dissipation calculations are greatly simplified
bymakingtwoassumptionswhichareusuallyfairlyaccurate.
First, the majority of losses in the power switch are due to
on-losses. To find these losses, assign a resistance value to
the collector/emitter terminals of the device using the satura-
tion voltage versus collector current curves (see Typical
Performance Characteristics). Power switch losses are
calculatedbymodelingtheswitchasaresistorwiththeswitch
duty cycle modifying the average power dissipation.
VOUT
FB
COMP
GND
R1
C1
R2
ICL ≈ 0.6V/R2
R3
C2
Q1
Note: Input and output
returns not common.
2
P
= (I ) R
δ
SW
SW
SW
Figure 4. Current Limit
where:
δ = duty cycle
ThemaximumcurrentlimitoftheMIC2171canbereducedby
adding a voltage clamp to the COMP output (Figure 4). This
feature can be useful in applications requiring either a com-
plete shutdown of Q1’s switching action or a form of current
fold-back limiting. This use of the COMP output does not
disable the oscillator, amplifiers or other circuitry, therefore
the supply current is never less than approximately 5mA.
VOUT + VF – V
IN(min)
δ =
VOUT + VF
= I (R
V
V
)
CL SW
SW
= output voltage
OUT
Thermal Management
V = D1 forward voltage drop at I
F
OUT
Although the MIC2171 family contains thermal protection
circuitry, for best reliability, avoid prolonged operation with
junction temperatures near the rated maximum.
From the Typical performance Characteristics:
R
= 0.37Ω
SW
Then:
The junction temperature is determined by first calculating
the power dissipation of the device. For the MIC2171, the
total power dissipation is the sum of the device operating
losses and power switch losses.
2
P
= (2.21) × 0.37 × 0.662
= 1.2W
SW
P
P
P
SW)
= 1.2 + 0.1
(total)
(total)
= 1.3W
4-8
1997
MIC2171
Micrel
Thejunctiontemperatureforanysemiconductoriscalculated
using the following:
mode is preferred because the feedback control of the
converter is simpler.
T = T + P θ
(total) JA
WhenL1dischargesitscurrentcompletelyduringtheMIC2171
off-time, it is operating in discontinuous mode.
J
A
Where:
L1 is operating in continuous mode if it does not discharge
completely before the MIC2171 power switch is turned on
again.
T = junction temperature
J
T = ambient temperature (maximum)
A
P
= total power dissipation
(total)
Discontinuous Mode Design
θ
= junction to ambient thermal resistance
JA
Given the maximum output current, solve equation (1) to
determine whether the device can operate in discontinuous
mode without initiating the internal device current limit.
For the practical example:
T = 70°C
A
θ
= 45°C/W (TO-220)
JA
ICL
Then:
V
δ
IN(min)
2
T = 70 + (1.24 × 45)
J
(1)
IOUT ≤
VOUT
T = 126°C
J
This junction temperature is below the rated maximum of
150°C.
VOUT + VF – V
IN(min)
(1a) δ =
VOUT + VF
Grounding
Where:
Refer to Figure 5. Heavy lines indicate high current paths.
I
= internal switch current limit
CL
VIN
I
I
= 2.5A when δ < 50%
= 1.67 (2 – δ) when δ ≥ 50%
CL
4
IN
CL
(Refer to Electrical Characteristics.)
SW
I
= maximum output current
OUT
MIC2171
V
= minimum input voltage = V – V
SW
IN(min)
IN
FB
COMP
δ = duty cycle
GND
V
= required output voltage
OUT
V = D1 forward voltage drop
F
For the example in Figure 1.
I
I
= 0.25A
OUT
Single point ground
= 1.67 (2–0.662) = 2.24A
= 4.18V
CL
V
IN(min)
Figure 5. Single Point Ground
δ = 0.662
= 12.0V
A single point ground is strongly recommended for proper
operation.
V
OUT
V = 0.36V (@ .26A, 70°C)
F
The signal ground, compensation network ground, and feed-
back network connections are sensitive to minor voltage
variations. The input and output capacitor grounds and
power ground conductors will exhibit voltage drop when
carrying large currents. Keep the sensitive circuit ground
traces separate from the power ground traces. Small voltage
variations applied to the sensitive circuits can prevent the
MIC2171 or any switching regulator from functioning prop-
erly.
Then:
2.235
× 4.178 × 0.662
12
2
I
≤
OUT
I
≤ 0.258A
OUT
This value is greater than the 0.25A output current require-
ment, so we can proceed to find the minimum inductance
value of L1 for discontinuous operation at P
.
OUT
Boost Conversion
2
Refer to Figure 1 for a typical boost conversion application
where a +5V logic supply is available but +12V at 0.25A is
required.
V
δ
(
)
IN
(2)
L1 ≥
2 P
f
OUT SW
Where:
The first step in designing a boost converter is determining
whether inductor L1 will cause the converter to operate in
either continuous or discontinuous mode. Discontinuous
P
= 12 × 0.25 = 3W
OUT
5
f
= 1×10 Hz (100kHz)
SW
1997
4-9
MIC2171
Micrel
down (failure) of the MIC2171’s internal power switch.
For our practical example:
Discontinuous Mode Design
2
4.178 × 0.662
(
)
L1 ≥
When designing a discontinuous flyback converter, first de-
termine whether the device can safely handle the peak
primary current demand placed on it by the output power.
Equation (8) finds the maximum duty cycle required for a
given input voltage and output power. If the duty cycle is
greater than 0.8, discontinuous operation cannot be used.
5
2 × 3.0 × 1×10
L1 ≥ 12.4µH (use 15µH)
Equation (3) solves for L1’s maximum current value.
V
T
IN ON
I
=
(3)
L1(peak)
L1
2 POUT
Where:
(8)
δ ≥
-6
ICL
V
– VSW
T
= δ / f
= 6.62×10 sec
SW
(
)
IN(min)
ON
-6
For a practical example let: (see Figure 2)
4.178 × 6.62 ×10
I
=
L1(peak)
-6
15 ×10
P
= 5.0V × 0.5A = 2.5W
OUT
I
= 1.84A
V
I
= 4.0V to 6.0V
= 2.5A when δ < 50%
1.67 (2 – δ) when δ ≥ 50%
L1(peak)
IN
Use a 15µH inductor with a peak current rating of at least 2A.
CL
Flyback Conversion
Then:
Flyback converter topology may be used in low power appli-
cations where voltage isolation is required or whenever the
input voltage can be less than or greater than the output
voltage. As with the step-up converter the inductor (trans-
former primary) current can be continuous or discontinuous.
Discontinuous operation is recommended.
V
= V – I
× R
(
)
IN
CL SW
IN min
(
)
V
V
= 4 – 0.78V
IN(min)
= 3.22V
IN(min)
δ ≥ 0.74 (74%), less than 0.8 so discontinous is
permitted.
Figure 2 shows a practical flyback converter design using the
MIC2171.
A few iterations of equation (8) may be required if the duty
cycle is found to be greater than 50%.
Switch Operation
Calculate the maximum transformer turns ratio a, or
During Q1’s on time (Q1 is the internal NPN transistor—see
block diagrams), energy is stored in T1’s primary inductance.
DuringQ1’sofftime,storedenergyispartiallydischargedinto
C4 (output filter capacitor). Careful selection of a low ESR
capacitor for C4 may provide satisfactory output ripple volt-
age making additional filter stages unnecessary.
N
/N
, thatwillguaranteesafeoperationoftheMIC2171
PRI SEC
power switch.
V
F
– V
CE CE
IN(max)
(9)
a ≤
V
SEC
Where:
C1 (input capacitor) may be reduced or eliminated if the
MIC2171 is located near a low impedance voltage source.
a = transformer maximum turns ratio
V
= power switch collector to emitter
maximum voltage
CE
Output Diode
The output diode allows T1 to store energy in its primary
inductance (D2 nonconducting) and release energy into C4
(D2 conducting). The low forward voltage drop of a Schottky
diode minimizes power loss in D2.
F
= safety derating factor (0.8 for most
commercial and industrial applications)
CE
V
= maximum input voltage
IN(max)
V
= transformer secondary voltage (V
+ V )
OUT F
SEC
Frequency Compensation
For the practical example:
A simple frequency compensation network consisting of R3
and C2 prevents output oscillations.
V
F
= 65V max. for the MIC2171
CE
= 0.8
CE
High impedance output stages (transconductance type) in
theMIC2171oftenpermitsimplifiedloop-stabilitysolutionsto
beconnectedtocircuitground, althoughamoreconventional
technique of connecting the components from the error
amplifier output to its inverting input is also possible.
V
= 5.6V
SEC
Then:
65 × 0.8 – 6.0
a ≤
5.6
Voltage Clipper
a ≤ 8.2 (N /N
)
PRI SEC
Care must be taken to minimize T1’s leakage inductance,
otherwise it may be necessary to incorporate the voltage
clipper consisting of D1, R4, and C3 to avoid second break-
Next, calculate the maximum primary inductance required to
store the needed output energy with a power switch duty
cycle of 55%.
4-10
1997
MIC2171
Micrel
2
2
L
0.5 f
V
T
PRI
SW IN(min)
ON
a ≤
a ≤
(12)
(10)
L
≥
PRI
L
P
SEC
OUT
Where:
Then:
L
= maximum primary inductance
PRI
11.4
7.9
= 1.20
f
= device switching frequency (100kHz)
= minimum input voltage
SW
V
T
IN(min)
This ratio is less than the ratio calculated in equation (9).
When specifying the transformer it is necessary to know the
primary peak current which must be withstood without satu-
rating the transformer core.
= power switch on time
ON
Then:
2
2
5
-6
0.5 × 1×10 × 3.22 × 7.4 ×10
(
)
(
)
L
≥
V
T
PRI
IN(min) ON
2.5
I
=
(13)
So:
PEAK(pri)
L
PRI
L
≥ 11.4µH
PRI
Use an 12µH primary inductance to overcome circuit ineffi-
ciencies.
3.22 × 7.6 ×10-6
12µH
IPEAK(pri)
=
To complete the design the inductance value of the second-
ary is found which will guarantee that the energy stored in the
transformer during the power switch on time will be com-
pleted discharged into the output during the off-time. This is
necessary when operating in discontinuous-mode.
I
= 2.1A
PEAK(pri)
Now find the minimum reverse voltage requirement for the
output rectifier. This rectifier must have an average current
rating greater than the maximum output current of 0.5A.
4
2
2
0.5 fSW VSEC TOFF
VIN(max) + V
a
(
)
OUT
LSEC
Where:
≤
(11)
VBR
Where:
≥
(14)
POUT
FBR
a
L
= maximum secondary inductance
= power switch off time
V
= output rectifier maximum peak
reverse voltage rating
SEC
BR
T
OFF
Then:
a = transformer turns ratio (1.2)
F
= reverse voltage safety derating factor (0.8)
BR
2
2
5
-6
0.5 × 1×10 × 5.41 × 2.6 ×10
(
)
(
)
Then:
L
≤
SEC
2.5
6.0 + 5.0 × 1.2
(
)
L
≤ 7.9µH
V
≥
SEC
BR
0.8 × 1.2
Finally, recalculate the transformer turns ratio to insure that
it is less than the value earlier found in equation (9).
V
≥ 12.5V
BR
A 1N5817 will safely handle voltage and current require-
ments in this example.
1997
4-11
MIC2171
Micrel
Forward Converters
core is reset by the tertiary winding discharging T1’s peak
magnetizing current through D2.
Micrel’s MIC2171 can be used in several circuit configura-
tionstogenerateanoutputvoltagewhichislessthantheinput
voltage (buck or step-down topology). Figure 7 shows the
MIC2171 in a voltage step-down application. Because of the
internal architecture of these devices, more external compo-
nents are required to implement a step-down regulator than
with other devices offered by Micrel (refer to the LM257x or
MIC457x family of buck switchers). However, for step-down
conversion requiring a transformer (forward), the MIC2171 is
a good choice.
For most forward converters the duty cycle is limited to 50%,
allowing the transformer flux to reset with only two times the
input voltage appearing across the power switch. Although
during normal operation this circuit’s duty cycle is well below
50%, the MIC2172 has a maximum duty cycle capability of
90%. If90%wasrequiredduringoperation(start-upandhigh
load currents), a complete reset of the transformer during the
off-time would require the voltage across the power switch to
be ten times the input voltage. This would limit the input
voltage to 6V or less for forward converter applications.
A 12V to 5V step-down converter using transformer isolation
(forward) is shown in Figure 7. Unlike the isolated flyback
converter which stores energy in the primary inductance
during the controller’s on-time and releases it to the load
during the off-time, the forward converter transfers energy to
the output during the on-time, using the off-time to reset the
transformer core. In the application shown, the transformer
To prevent core saturation, the application given here uses a
duty cycle limiter consisting of Q1, C4 and R3. Whenever the
MIC2171 exceeds a duty cycle of 50%, T1’s reset winding
current turns Q1 on. This action reduces the duty cycle of the
MIC2171 until T1 is able to reset during each cycle.
T1
1:1:1
D3
1N5819
L1 100µH
VOUT
VIN
12V
5V, 1A
R4
D4
1N5819
C5
470µF
3.74k
1%
R1*
C2*
D1*
IN
SW
FB
MIC2171
C1
22µF
D2
1N5819
COMP
GND
R5
1.24k
1%
R2
1k
Q1†
R3†
C3
1µF
C4†
* Voltage clipper
†
Duty cycle limiter
Figure 7. MIC2171 Forward Converter
4-12
1997
相关型号：
MIC2172BM
Switching Regulator, Current-mode, 3.5A, 115kHz Switching Freq-Max, BIPolar, PDSO8, SOIC-8
MICROCHIP
MIC2172BMT&R
Switching Regulator, Current-mode, 3.5A, 115kHz Switching Freq-Max, PDSO8, SOIC-8
MICREL
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