MIC26603 [MICREL]
28V, 6A Hyper Light Load Synchronous DC/DC Buck Regulator; 28V ，6A超轻型载同步DC / DC降压稳压器
| 型号： | MIC26603 |
| 厂家： | 
MICREL SEMICONDUCTOR |
| 描述： | 28V, 6A Hyper Light Load Synchronous DC/DC Buck Regulator |
| 文件： | 总30页 (文件大小：1054K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
MIC26603
28V, 6A Hyper Light Load™
Synchronous DC/DC Buck Regulator
SuperSwitcher IIG™
General Description
Features
•
•
Hyper Light Load™ efficiency – up to 80% at 10mA
Hyper Speed Control™ architecture enables
− High Delta V operation (VIN = 28V and VOUT = 0.8V)
− Small output capacitance
Input voltage range: 4.5V to 28V
Output current up to 6A
The Micrel MIC26603 is
a
constant-frequency,
synchronous DC/DC buck regulator featuring adaptive on-
time control architecture. The MIC26603 operates over a
supply range of 4.5V to 28V. It has an internal linear
regulator which provides a regulated 5V to power the
internal control circuitry. MIC26603 operates at a constant
600kHz switching frequency in continuous-conduction
mode and can be used to provide up to 6A of output
current. The output voltage is adjustable down to 0.8V.
•
•
•
•
•
•
•
•
•
Up to 95% Efficiency
Adjustable output voltage from 0.8V to 5.5V
±1% FB accuracy
Any CapacitorTM stable − zero-to-high ESR
600kHz switching frequency
Micrel’s Hyper Light Load™ architecture provides the same
high-efficiency and ultra-fast transient response as the
Hyper Speed Control™ architecture under medium to heavy
loads, but also maintains high efficiency under light load
conditions
by
transitioning
to
variable-frequency,
Power Good (PG) output
discontinuous-mode operation.
Foldback current-limit and “hiccup mode” short-circuit
protection
The MIC26603 offers a full suite of protection features to
ensure protection of the IC during fault conditions. These
include undervoltage lockout to ensure proper operation
under power-sag conditions, thermal shutdown, internal
soft-start to reduce the inrush current, foldback current
limit and “hiccup mode” short-circuit protection. The
MIC26603 includes a Power Good (PG) output to allow
simple sequencing.
•
•
•
Safe start-up into pre-biased loads
5mm x 6mm MLF® package
–40°C to +125°C junction temperature range
Applications
•
•
•
Distributed power systems
All support documentation can be found on Micrel’s web
site at: www.micrel.com.
Telecom/networking infrastructure
Printers, scanners, graphic cards, and video cards
___________________________________________________________________________________________________________
Typical Application
Efficiency (VIN = 12V)
vs. Output Current
100
95
5.0V
3.3V
2.5V
90
85
80
75
70
65
60
55
50
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
0
1
2
3
4
5
6
7
8
OUTPUT CURRENT (A)
Hyper Speed Control, Hyper Light Load, SuperSwitcher II, and Any Capacitor are trademarks of Micrel, Inc.
MLF and MicroLeadFrame are registered trademarks of Amkor Technology, Inc.
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
M9999-071311-A
July 2011
Micrel, Inc.
MIC26603
Ordering Information
Junction Temperature
Range
Part Number
Voltage
Switching Frequency
Package
Lead Finish
28-Pin 5mm × 6mm MLF®
MIC26603YJL
Adjustable
600kHz
Pb-Free
–40°C to +125°C
Pin Configuration
28-Pin 5mm x 6mm MLF® (YJL)
Pin Description
Pin Number
Pin Name
PVDD
NC
Pin Function
5V Internal Linear Regulator (Output): PVDD supply is the power MOSFET gate drive supply voltage
and created by internal LDO from VIN. When VIN < +5.5V, PVDD should be tied to PVIN pins.
A 2.2µF ceramic capacitor from the PVDD pin to PGND (pin 2) must be place next to the IC.
1
3
No Connect.
Switch Node (Output): Internal connection for the high-side MOSFET source and low-side MOSFET
drain. Due to the high speed switching on this pin, the SW pin should be routed away from sensitive
nodes.
4, 9, 10, 11, 12
SW
Power Ground. PGND is the ground path for the MIC26603 buck converter power stage. The PGND
pins connect to the low-side N-Channel internal MOSFET gate drive supply ground, the sources of
the MOSFETs, the negative terminals of input capacitors, and the negative terminals of output
capacitors. The loop for the power ground should be as small as possible and separate from the
Signal ground (SGND) loop.
2, 5, 6, 7, 8, 21
PGND
High-Side N-internal MOSFET Drain Connection (Input): The PVIN operating voltage range is from
4.5V to 26V. Input capacitors between the PVIN pins and the power ground (PGND) are required
and keep the connection short.
13,14,15,
PVIN
BST
16,17,18,19
Boost (output): Bootstrapped voltage to the high-side N-channel MOSFET driver. A Schottky diode is
connected between the PVDD pin and the BST pin. A boost capacitor of 0.1μF is connected between
the BST pin and the SW pin. Adding a small resistor at the BST pin can slow down the turn-on time
of high-side N-Channel MOSFETs.
20
2
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July 2011
Micrel, Inc.
MIC26603
Pin Description (Continued)
Pin Number
Pin Name
Pin Function
Current Sense (input): The CS pin senses current by monitoring the voltage across the low-side
MOSFET during the OFF-time. The current sensing is necessary for short circuit protection and zero
current cross comparator. In order to sense the current accurately, connect the low-side MOSFET
drain to SW using a Kelvin connection. The CS pin is also the high-side MOSFET’s output driver
return.
22
CS
Signal ground. SGND must be connected directly to the ground planes. Do not route the SGND pin
to the PGND Pad on the top layer, see PCB layout guidelines for details.
23
24
25
SGND
FB
Feedback (Input): Input to the transconductance amplifier of the control loop. The FB pin is
regulated to 0.8V. A resistor divider connecting the feedback to the output is used to adjust the
desired output voltage.
Power Good (Output): Open Drain Output. The PG pin is externally tied with a resistor to VDD. A
high output is asserted when VOUT > 92% of nominal.
PG
Enable (input): A logic level control of the output. The EN pin is CMOS-compatible. Logic high =
enable, logic low = shutdown. In the off state, supply current of the device is greatly reduced
(typically 5µA). The EN pin should not be left open.
26
27
28
EN
VIN
Power Supply Voltage (Input): Requires bypass capacitor to SGND.
5V Internal Linear Regulator (Output): VDD supply is the supply bus for the IC control circuit. VDD is
created by internal LDO from VIN. When VIN < +5.5V, VDD should be tied to PVIN pins. A 1.0µF
ceramic capacitor from the VDD pin to SGND pins must be place next to the IC.
VDD
3
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July 2011
Micrel, Inc.
MIC26603
Absolute Maximum Ratings(1,2)
Operating Ratings(3)
Supply Voltage (PVIN, VIN)................................. 4.5V to 28V
PVDD, VDD Supply Voltage (PVDD, VDD)......... 4.5V to 5.5V
Enable Input (VEN).................................................. 0V to VIN
Junction Temperature (TJ) ........................−40°C to +125°C
Maximum Power Dissipation......................................Note 4
Package Thermal Resistance(4)
PVIN to PGND................................................ −0.3V to +29V
VIN to PGND ....................................................−0.3V to PVIN
PVDD, VDD to PGND......................................... −0.3V to +6V
VSW, VCS to PGND..............................−0.3V to (PVIN +0.3V)
VBST to VSW ........................................................ −0.3V to 6V
VBST to PGND.................................................. −0.3V to 35V
VFB, VPG to PGND...............................−0.3V to (VDD + 0.3V)
VEN to PGND ........................................ −0.3V to (VIN +0.3V)
PGND to SGND ........................................... −0.3V to +0.3V
Junction Temperature ..............................................+150°C
Storage Temperature (TS).........................−65°C to +150°C
Lead Temperature (soldering, 10sec)........................ 260°C
5mm x 6mm MLF®-24L (θJA) .............................28°C/W
Electrical Characteristics(5)
PVIN = VIN = VEN = 12V, VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate -40°C ≤ TJ ≤ +125°C.
Parameter
Condition
Min.
Typ.
Max.
Units
Power Supply Input
Input Voltage Range (VIN, PVIN)
Quiescent Supply Current
Shutdown Supply Current
VDD Supply Voltage
VDD Output Voltage
4.5
28
750
10
V
VFB = 1.5V (non-switching)
VEN = 0V
450
5
µA
µA
VIN = 7V to 28V, IDD = 40mA
VDD Rising
4.8
3.7
5
5.4
4.5
V
V
VDD UVLO Threshold
VDD UVLO Hysteresis
4.2
400
380
mV
mV
Dropout Voltage (VIN – VDD
)
IDD = 25mA
600
5.5
DC/DC Controller
Output-Voltage Adjust Range
(VOUT
0.8
V
)
Reference
0.792
0.788
0.8
0.8
0.808
0.812
0°C ≤ TJ ≤ 85°C (±1.0%)
−40°C ≤ TJ ≤ 125°C (±1.5%)
IOUT = 1A to 6A (Continuous Mode)
VIN = 4.5V to 28V
V
Load Regulation
0.25
0.25
50
%
%
Line Regulation
FB Bias Current
VFB = 0.8V
500
nA
Enable Control
EN Logic Level High
EN Logic Level Low
EN Bias Current
1.8
V
V
0.6
30
VEN = 12V
6
µA
Oscillator
Switching Frequency (6)
Maximum Duty Cycle (7)
Minimum Duty Cycle
Minimum Off-Time
450
600
82
750
kHz
%
VFB = 0V
VFB = 1.0V
0
%
300
ns
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July 2011
Micrel, Inc.
MIC26603
Electrical Characteristics(5) (Continued)
PVIN = VIN = VEN = 12V, VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C ≤ TJ ≤ +125°C.
Parameter
Condition
Min.
Typ.
Max.
Units
ms
Soft-Start
Soft-Start time
5
Short-Circuit Protection
Current-Limit Threshold
Current-Limit Threshold
Short-Circuit Current
Internal FETs
7.5
6.6
13
13
17
17
A
A
A
V
FB = 0.8V, TJ = 25°C
FB = 0.8V, TJ = 125°C
V
VFB = 0V
2.7
Top-MOSFET RDS (ON)
Bottom-MOSFET RDS (ON)
SW Leakage Current
VIN Leakage Current
Power Good
ISW = 3A
ISW = 3A
VEN = 0V
VEN = 0V
42
mΩ
mΩ
µA
12.5
60
25
µA
Power Good Threshold Voltage
Power Good Hysteresis
Power Good Delay Time
Power Good Low Voltage
Thermal Protection
Over-Temperature Shutdown
Sweep VFB from Low to High
Sweep VFB from High to Low
Sweep VFB from Low to High
Sweep VFB < 0.9 × VNOM, IPG = 1mA
85
92
5.5
100
70
95
%VOUT
%VOUT
µs
200
mV
TJ Rising
160
15
°C
°C
Over-Temperature Shutdown
Hysteresis
Notes:
1. Exceeding the absolute maximum rating may damage the device.
2. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5kΩ in series with 100pF.
3. The device is not guaranteed to function outside operating range.
4. PD(MAX) = (TJ(MAX) – TA)/ θJA, where θJA depends upon the printed circuit layout. A 5 square inch 4 layer, 0.62”, FR-4 PCB with 2oz finish copper weight
per layer is used for the θJA.
5. Specification for packaged product only.
6. Measured in test mode.
7. The maximum duty-cycle is limited by the fixed mandatory off-time tOFF of typically 300ns.
5
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July 2011
Micrel, Inc.
MIC26603
Typical Characteristics
VDD Output Voltage
vs. Input Voltage
VIN Shutdown Current
vs. Input Voltage
VIN Operating Supply Current
vs. Input Voltage
1.0
10
8
60
40
20
0
VOUT = 1.8V
OUT = 0A
SWITCHING
VEN = 0V
0.8
I
REN = Open
0.6
0.4
0.2
0.0
6
4
VFB = 0.9V
IDD = 10mA
2
0
4
10
16
22
28
4
10
16
22
28
28
28
4
10
16
22
28
28
28
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Feedback Voltage
vs. Input Voltage
Current Limit
vs. Input Voltage
Total Regulation
vs. Input Voltage
0.808
0.804
0.800
0.796
0.792
20
15
10
5
1.0%
0.5%
0.0%
-0.5%
-1.0%
VOUT = 1.8V
IOUT = 1A to 6A
VOUT = 1.8V
IOUT = 1A
VOUT = 1.8V
0
4
10
16
22
4
10
16
22
28
4
10
16
22
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Switching Frequency
vs. Input Voltage
Enable Input Current
vs. Input Voltage
PG Threshold/VREF Ratio
vs. Input Voltage
700
650
600
550
500
100%
95%
90%
85%
80%
16
12
8
VEN = VIN
VOUT = 1.8V
IOUT = 1A
4
VREF = 0.7V
0
4
4.0
10.0
16.0
22.0
28.0
4
10
16
22
10
16
22
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
6
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July 2011
Micrel, Inc.
MIC26603
Typical Characteristics (Continued)
VIN Operating Supply Current
vs. Temperature
VIN Shutdown Current
vs. Temperature
VDD UVLO Threshold
vs. Temperature
1.0
0.8
0.6
0.4
0.2
0.0
5
4
3
2
1
0
8
6
4
2
0
Rising
Falling
VIN = 12V
VOUT = 1.8V
IOUT = 0A
VIN = 12V
SWITCHING
IOUT = 0A
VEN = 0V
Hyst
0
-50
-25
0
25
50
75
100
100
100
125
125
125
-50
-25
25
50
75
100
125
-50
-25
0
25
50
75
100
125
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
Line Regulation
vs. Temperature
Load Regulation
vs. Temperature
Feedback Voltage
vs. Temperature
0.808
0.804
0.800
0.796
0.792
0.2%
0.1%
0.0%
-0.1%
-0.2%
0.4%
0.2%
0.0%
-0.2%
-0.4%
VIN = 12V
V
OUT = 1.8V
I
OUT = 1A
VIN = 4.5V to 28V
VOUT = 1.8V
OUT = 1A
I
VIN = 12V
VOUT = 1.8V
IOUT =1A to 6A
-50
-25
0
25
50
75
-50
-25
0
25
50
75
100
125
-50
-25
0
25
50
75
100
125
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
Current Limit
vs. Temperature
Switching Frequency
vs. Temperature
VDD
vs. Temperature
700
650
600
550
500
25
20
15
10
5
6
5
4
3
VIN = 12V
OUT = 1.8V
IOUT = 1A
V
VIN = 12V
OUT = 1.8V
V
VIN = 12V
IOUT = 0A
0
2
-50
-25
0
25
50
75
100
125
-50
-25
0
25
50
75
-50
-25
0
25
50
75
100
125
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
7
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July 2011
Micrel, Inc.
MIC26603
Typical Characteristics (Continued)
Feedback Voltage
vs. Output Current
Efficiency
vs. Output Current
Output Voltage
vs. Output Current
100
1.819
1.814
1.810
1.805
1.800
1.796
1.791
1.787
1.782
0.808
12VIN
VIN = 12V
90
V
OUT = 1.8V
0.804
80
24VIN
0.800
70
0.796
VIN = 12V
OUT = 1.8V
60
V
VOUT = 1.8V
50
0.792
0
1
2
3
4
5
6
0
1
2
3
4
5
6
0
1
2
3
4
5
6
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Output Voltage (VIN = 5V)
vs. Output Current
Switching Frequency
vs. Output Current
Line Regulation
vs. Output Current
5
4.6
4.2
3.8
3.4
700
650
600
550
500
1.0%
0.5%
0.0%
-0.5%
-1.0%
VIN = 5V
VFB < 0.8V
VIN = 4.6V to 28V
OUT = 1.8V
V
TA
25ºC
85ºC
125ºC
VIN = 12V
VOUT = 1.8V
3
0
1
2
3
4
5
6
7
8
1
2
3
4
5
6
0
1
2
3
4
5
6
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Die Temperature* (VIN = 5V)
vs. Output Current
Efficiency (VIN = 5V)
vs. Output Current
IC Power Dissipation
vs. Output Current
100
95
90
85
80
75
70
65
60
55
60
50
40
30
20
10
2.5
2.0
1.5
1.0
0.5
0.0
VIN = 5V
VOUT = 0.8,V, 1.0V, 1.2V, 1.5V, 1.8V,2.5V, 3.3V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
VIN = 5V
3.3V
VIN
=
5V
2
V
OUT = 1.8V
0.8V
5
0
0
50
0
1
2
3
4
5
6
1
3
4
5
6
7
8
0
1
2
3
4
6
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
8
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July 2011
Micrel, Inc.
MIC26603
Typical Characteristics (Continued)
Efficiency (VIN = 12V)
vs. Output Current
Die Temperature* (VIN = 12V)
vs. Output Current
IC Power Dissipation (VIN = 12V)
vs. Output Current
100
95
90
85
80
75
70
65
60
55
50
60
50
40
30
20
10
0
2.5
2.0
1.5
1.0
0.5
0.0
VIN = 12V
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
VOUT = 0.8,V, 1.0V, 1.2V, 1.5V, 1.8V,2.5V, 3.3V, 5.0V
5.0V
VIN = 12V
OUT = 1.8V
V
0.8V
0
1
2
3
4
5
6
0
1
2
3
4
5
6
7
8
0
1
2
3
4
5
6
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Die Temperature* (VIN = 24V)
vs. Output Current
Efficiency (VIN = 24V)
vs. Output Current
IC Power Dissipation (VIN = 24V)
vs. Output Current
95
90
85
80
75
70
65
60
55
50
60
50
40
30
20
10
0
2.5
2.0
1.5
1.0
0.5
0.0
VIN = 24V
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
VOUT = 0.8,V, 1.0V, 1.2V, 1.5V, 1.8V,2.5V, 3.3V, 5.0V
5.0V
VIN = 24V
V
OUT = 1.8V
0.8V
0
1
2
3
4
5
6
0
1
2
3
4
5
6
7
8
0
1
2
3
4
5
6
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Thermal Derating*
Thermal Derating*
Thermal Derating*
vs. Ambient Temperature
vs. Ambient Temperature
vs. Ambient Temperature
12
10
8
12
10
8
12
10
8
0.8V
1.8V
0.8V
1.8V
1.5V
6
6
6
3.3V
4
4
4
VIN = 12V
OUT = 0.8, 1.2, 1.8V
VIN = 5V
OUT = 0.8, 1.2, 1.5V
VIN = 5V
V
V
VOUT = 1.8, 2.5, 3.3V
2
2
2
0
0
0
-50
-25
0
25
50
75
100
125
-50
-25
0
25
50
75
100
125
-50
-25
0
25
50
75
100
125
AMBIENT TEMPERATURE (°C)
AMBIENT TEMPERATURE (°C)
AMBIENT TEMPERATURE (°C)
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Micrel, Inc.
MIC26603
Typical Characteristics (Continued)
Thermal Derating*
Thermal Derating*
vs. Ambient Temperature
vs. Ambient Temperature
12
10
8
12
10
8
2.5V
0.8V
5V
6
6
2.5V
4
4
VIN = 12V
OUT = 2.5, 3.3, 5V
VIN = 24V
V
V
OUT = 0.8, 1.2, 2.5V
2
2
0
0
-50
-25
0
25
50
75
100
125
-50
-25
0
25
50
75
100
125
AMBIENT TEMPERATURE (°C)
AMBIENT TEMPERATURE (°C)
Die Temperature* : The temperature measurement was taken at the hottest point on the MIC26603 case mounted on a 5 square inch 4 layer, 0.62”,
FR-4 PCB with 2oz finish copper weight per layer; see Thermal Measurement section. Actual results will depend upon the size of the PCB, ambient
temperature and proximity to other heat emitting components.
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MIC26603
Functional Characteristics
11
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MIC26603
Functional Characteristics (Continued)
12
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MIC26603
Functional Characteristics (Continued)
13
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MIC26603
Functional Diagram
Figure 1. MIC26603 Block Diagram
14
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Micrel, Inc.
MIC26603
The maximum duty cycle is obtained from the 300ns
Functional Description
tOFF(min)
:
The MIC26603 is an adaptive, ON-time, synchronous
step-down, DC/DC regulator with an internal 5V linear
regulator and a Power Good (PG) output. It is designed
to operate over a wide input voltage range from 4.5V to
28V and provides a regulated output voltage at up to 7A
of output current. An adaptive ON-time control scheme is
employed in to obtain a constant switching frequency
and to simplify the control compensation. Over-current
protection is implemented without the use of an external
sense resistor. The device includes an internal soft-start
function which reduces the power supply input surge
current at start-up by controlling the output voltage rise
time.
tS - tOFF(min)
300ns
tS
Dmax
=
= 1-
Eq. 2
tS
where tS = 1/600kHz = 1.66μs.
It is not recommended to use MIC26603 with a OFF-time
close to tOFF(min) during steady-state operation. Also, as
VOUT increases, the internal ripple injection will increase
and reduce the line regulation performance. Therefore,
the maximum output voltage of the MIC26603 should be
limited to 5.5V and the maximum external ripple injection
should be limited to 200mV.Please refer to “Setting
Output Voltage” subsection in Application Information for
more details.
Theory of Operation
The MIC26603 is able to operate in either continuous
mode or discontinuous mode. The operating mode is
determined by the output of the Zero Cross comparator
(ZC) as shown in Figure 1.
The actual ON-time and resulting switching frequency
will vary with the part-to-part variation in the rise and fall
times of the internal MOSFETs, the output load current,
and variations in the VDD voltage. Also, the minimum tON
results in a lower switching frequency in high VIN to VOUT
applications, such as 24V to 1.0V. The minimum tON
measured on the MIC26603 evaluation board is about
100ns. During load transients, the switching frequency is
changed due to the varying OFF-time.
Continuous Mode
In continuous mode, the output voltage is sensed by the
MIC26603 feedback pin FB via the voltage divider R1
and R2, and compared to a 0.8V reference voltage VREF
at the error comparator through
a
low gain
transconductance (gm) amplifier. If the feedback voltage
decreases and the output of the gm amplifier is below
0.8V, then the error comparator will trigger the control
logic and generate an ON-time period. The ON-time
period length is predetermined by the “FIXED tON
ESTIMATION” circuitry:
To illustrate the control loop operation, we will analyze
both the steady-state and load transient scenarios.
Figure 2 shows the MIC26603 control loop timing during
steady-state operation. During steady-state, the gm
amplifier senses the feedback voltage ripple, which is
proportional to the output voltage ripple and the inductor
current ripple, to trigger the ON-time period. The ON-
time is predetermined by the tON estimator. The
termination of the OFF-time is controlled by the feedback
voltage. At the valley of the feedback voltage ripple,
which occurs when VFB falls below VREF, the OFF period
ends and the next ON-time period is triggered through
the control logic circuitry.
VOUT
tON(estimated)
=
Eq. 1
V ×600kHz
IN
where VOUT is the output voltage and VIN is the power
stage input voltage.
At the end of the ON-time period, the internal high-side
driver turns off the high-side MOSFET and the low-side
driver turns on the low-side MOSFET. The OFF-time
period length depends upon the feedback voltage in
most cases. When the feedback voltage decreases and
the output of the gm amplifier is below 0.8V, the ON-time
period is triggered and the OFF-time period ends. If the
OFF-time period determined by the feedback voltage is
less than the minimum OFF-time tOFF(min), which is about
300ns, the MIC26603 control logic will apply the tOFF(min)
instead. tOFF(min) is required to maintain enough energy in
the boost capacitor (CBST) to drive the high-side
MOSFET.
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MIC26603
Unlike true current-mode control, the MIC26603 uses the
output voltage ripple to trigger an ON-time period. The
output voltage ripple is proportional to the inductor
current ripple if the ESR of the output capacitor is large
enough. The MIC26603 control loop has the advantage
of eliminating the need for slope compensation.
In order to meet the stability requirements, the
MIC26603 feedback voltage ripple should be in phase
with the inductor current ripple and large enough to be
sensed by the gm amplifier and the error comparator.
The recommended feedback voltage ripple is
20mV~100mV. If a low-ESR output capacitor is selected,
then the feedback voltage ripple may be too small to be
sensed by the gm amplifier and the error comparator.
Also, the output voltage ripple and the feedback voltage
ripple are not necessarily in phase with the inductor
current ripple if the ESR of the output capacitor is very
low. In these cases, ripple injection is required to ensure
proper operation. Please refer to “Ripple Injection”
subsection in Application Information for more details
about the ripple injection technique.
Figure 2. MIC26603 Control Loop Timing
Figure 3 shows the operation of the MIC26603 during a
load transient. The output voltage drops due to the
sudden load increase, which causes the VFB to be less
than VREF. This will cause the error comparator to trigger
an ON-time period. At the end of the ON-time period, a
minimum OFF-time tOFF(min) is generated to charge CBST
since the feedback voltage is still below VREF. Then, the
next ON-time period is triggered due to the low feedback
voltage. Therefore, the switching frequency changes
during the load transient, but returns to the nominal fixed
frequency once the output has stabilized at the new load
current level. With the varying duty cycle and switching
frequency, the output recovery time is fast and the
output voltage deviation is small in MIC26603 converter.
Discontinuous Mode
In continuous mode, the inductor current is always
greater than zero; however, at light loads the MIC26603
is able to force the inductor current to operate in
discontinuous mode. Discontinuous mode is where the
inductor current falls to zero, as indicated by trace (IL)
shown in Figure 4. During this period, the efficiency is
optimized by shutting down all the non-essential circuits
and minimizing the supply current. The MIC26603 wakes
up and turns on the high-side MOSFET when the
feedback voltage VFB drops below 0.8V.
The MIC26603 has a zero crossing comparator that
monitors the inductor current by sensing the voltage
drop across the low-side MOSFET during its ON-time. If
the VFB > 0.8V and the inductor current goes slightly
negative, then the MIC26603 automatically powers down
most of the IC circuitry and goes into a low-power mode.
Once the MIC26603 goes into discontinuous mode, both
LSD and HSD are low, which turns off the high-side and
low-side MOSFETs. The load current is supplied by the
output capacitors and VOUT drops. If the drop of VOUT
causes VFB to go below VREF, then all the circuits will
wake up into normal continuous mode. First, the bias
currents of most circuits reduced during the
discontinuous mode are restored, then a tON pulse is
triggered before the drivers are turned on to avoid any
possible glitches. Finally, the high-side driver is turned
on. Figure 4 shows the control loop timing in
discontinuous mode.
Figure 3. MIC26603 Load Transient Response
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MIC26603
Current Limit
The MIC26603 uses the RDS(ON) of the internal low-side
power MOSFET to sense over-current conditions. This
method will avoid adding cost, board space and power
losses taken by a discrete current sense resistor. The
low-side MOSFET is used because it displays much
lower parasitic oscillations during switching than the
high-side MOSFET.
In each switching cycle of the MIC26603 converter, the
inductor current is sensed by monitoring the low-side
MOSFET in the OFF period. If the inductor current is
greater than 13A, then the MIC26603 turns off the high-
side MOSFET and a soft-start sequence is triggered.
This mode of operation is called “hiccup mode” and its
purpose is to protect the downstream load in case of a
hard short. The load current-limit threshold has a fold
back characteristic related to the feedback voltage as
shown in Figure 5.
Figure 4. MIC26603 Control Loop Timing
(Discontinuous Mode)
Current Limit Threshold
vs. Feedback Voltage
During discontinuous mode, the zero crossing
comparator and the current-limit comparator are turned
off. The bias current of most circuits are reduced. As a
result, the total power supply current during
discontinuous mode is only about 450μA, allowing the
MIC26603 to achieve high efficiency in light load
applications.
20
16
12
8
4
VDD Regulator
The MIC26603 provides a 5V regulated output for input
voltage VIN ranging from 5.5V to 28V. When VIN < 5.5V,
VDD should be tied to PVIN pins to bypass the internal
linear regulator.
0
0.0
0.2
0.4
0.6
0.8
1.0
FEEDBACK VOLTAGE (V)
Soft-Start
Figure 5. MIC26603 Current Limit
Foldback Characteristic
Soft-start reduces the power supply input surge current
at startup by controlling the output voltage rise time. The
input surge appears while the output capacitor is
charged up. A slower output rise time will draw a lower
input surge current.
Power Good (PG)
The Power Good (PG) pin is an open drain output which
indicates logic high when the output is nominally 92% of
its steady state voltage. A pull-up resistor of more than
10kꢀ should be connected from PG to VDD.
The MIC26603 implements an internal digital soft-start
by making the 0.8V reference voltage VREF ramp from 0
to 100% in about 5ms with 9.7mV steps. Therefore, the
output voltage is controlled to increase slowly by a stair-
case VFB ramp. Once the soft-start cycle ends, the
related circuitry is disabled to reduce current
consumption. VDD must be powered up at the same time
or after VIN to make the soft-start function correctly.
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MIC26603
MOSFET Gate Drive
The Block Diagram (Figure 1) shows a bootstrap circuit,
consisting of D1 (a Schottky diode is recommended) and
C
BST. This circuit supplies energy to the high-side drive
circuit. Capacitor CBST is charged, while the low-side
MOSFET is on, and the voltage on the SW pin is
approximately 0V. When the high-side MOSFET driver is
turned on, energy from CBST is used to turn the MOSFET
on. As the high-side MOSFET turns on, the voltage on
the SW pin increases to approximately VIN. Diode D1 is
reverse biased and CBST floats high while continuing to
keep the high-side MOSFET on. The bias current of the
high-side driver is less than 10mA so a 0.1μF to 1μF is
sufficient to hold the gate voltage with minimal droop for
the power stroke (high-side switching) cycle, i.e. ΔBST =
10mA × 1.67μs/0.1μF = 167mV. When the low-side
MOSFET is turned back on, CBST is recharged through
D1. A small resistor RG, which is in series with CBST, can
be used to slow down the turn-on time of the high-side
N-channel MOSFET.
The drive voltage is derived from the VDD supply voltage.
The nominal low-side gate drive voltage is VDD and the
nominal high-side gate drive voltage is approximately
VDD – VDIODE, where VDIODE is the voltage drop across
D1. An approximate 30ns delay between the high-side
and low-side driver transitions is used to prevent current
from simultaneously flowing unimpeded through both
MOSFETs.
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MIC26603
Maximizing efficiency requires the proper selection of
core material and minimizing the winding resistance. The
high-frequency operation of the MIC26603 requires the
use of ferrite materials for all but the most cost sensitive
applications. Lower cost iron powder cores may be used
but the increase in core loss will reduce the efficiency of
the power supply. This is especially noticeable at low
output power. The winding resistance decreases
efficiency at the higher output current levels. The
winding resistance must be minimized although this
usually comes at the expense of a larger inductor. The
power dissipated in the inductor is equal to the sum of
the core and copper losses. At higher output loads, the
core losses are usually insignificant and can be ignored.
At lower output currents, the core losses can be a
significant contributor. Core loss information is usually
available from the magnetics vendor. Copper loss in the
inductor is calculated by Equation 7:
Application Information
Inductor Selection
Values for inductance, peak, and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine the
peak-to-peak inductor ripple current. Generally, higher
inductance values are used with higher input voltages.
Larger peak-to-peak ripple currents will increase the
power dissipation in the inductor and MOSFETs. Larger
output ripple currents will also require more output
capacitance to smooth out the larger ripple current.
Smaller peak-to-peak ripple currents require a larger
inductance value and therefore a larger and more
expensive inductor. A good compromise between size,
loss and cost is to set the inductor ripple current to be
equal to 20% of the maximum output current. The
inductance value is calculated in Equation 3:
P
INDUCTOR(Cu) = IL(RMS)2 × RWINDING
Eq. 7
VOUT × (V
− VOUT )
IN(max)
L =
Eq. 3
V
IN(max) × fsw × 20%×IOUT(max)
The resistance of the copper wire, RWINDING, increases
with the temperature. The value of the winding
resistance used should be at the operating temperature.
where:
SW = switching frequency, 600kHz
20% = ratio of AC ripple current to DC output current
IN(max) = maximum power stage input voltage
The peak-to-peak inductor current ripple is:
f
P
WINDING(Ht) = RWINDING(20°C) × (1 + 0.0042 × (TH – T20°C))
Eq. 8
V
where:
TH = temperature of wire under full load
VOUT ×(V
− VOUT )
IN(max)
ΔIL(pp)
=
Eq. 4
T20°C = ambient temperature
VIN(max) × fsw ×L
RWINDING(20°C) = room temperature winding resistance
(usually specified by the manufacturer)
The peak inductor current is equal to the average output
current plus one half of the peak-to-peak inductor current
ripple.
Output Capacitor Selection
The type of the output capacitor is usually determined by
its equivalent series resistance (ESR). Voltage and RMS
current capability are two other important factors for
selecting the output capacitor. Recommended capacitor
types are tantalum, low-ESR aluminum electrolytic, OS-
CON and POSCAP. The output capacitor’s ESR is
usually the main cause of the output ripple. The output
capacitor ESR also affects the control loop from a
stability point of view.
IL(pk) =IOUT(max) + 0.5 × ΔIL(pp)
Eq. 5
The RMS inductor current is used to calculate the I2R
losses in the inductor.
2
ΔIL(PP)
2
IL(RMS) = IOUT(max)
+
Eq. 6
12
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MIC26603
The power dissipated in the output capacitor is:
The maximum value of ESR is calculated:
2
ΔVOUT(pp)
PDISS(C
= IC
× ESRC
OUT (RMS)
Eq. 12
)
OUT
OUT
ESRC
≤
Eq. 9
OUT
ΔIL(PP)
Input Capacitor Selection
where:
The input capacitor for the power stage input VIN should
be selected for ripple current rating and voltage rating.
Tantalum input capacitors may fail when subjected to
high inrush currents, caused by turning the input supply
on. A tantalum input capacitor’s voltage rating should be
at least two times the maximum input voltage to
maximize reliability. Aluminum electrolytic, OS-CON, and
multilayer polymer film capacitors can handle the higher
inrush currents without voltage de-rating. The input
voltage ripple will primarily depend on the input
capacitor’s ESR. The peak input current is equal to the
peak inductor current, so:
ΔVOUT(pp) = peak-to-peak output voltage ripple
ΔIL(PP) = peak-to-peak inductor current ripple
The total output ripple is a combination of the ESR and
output capacitance. The total ripple is calculated in
Equation 10:
2
ΔIL(PP)
⎛
⎜
⎜
⎝
⎞
⎟
⎟
⎠
2
ΔVOUT(pp)
=
+
(
ΔIL(PP) × ESRC
)
OUT
COUT × fSW × 8
Eq. 10
ΔVIN = IL(pk) × ESRCIN
Eq. 13
where:
D = duty cycle
OUT = output capacitance value
SW = switching frequency
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming
the peak-to-peak inductor current ripple is low:
C
f
As described in the “Theory of Operation” subsection in
Functional Description, the MIC26603 requires at least
20mV peak-to-peak ripple at the FB pin to make the gm
amplifier and the error comparator behave properly. Also,
the output voltage ripple should be in phase with the
inductor current. Therefore, the output voltage ripple
caused by the output capacitors value should be much
smaller than the ripple caused by the output capacitor
ESR. If low-ESR capacitors, such as ceramic capacitors,
are selected as the output capacitors, a ripple injection
method should be applied to provide the enough
feedback voltage ripple. Please refer to the “Ripple
Injection” subsection for more details.
ICIN(RMS) ≈ IOUT(max) × D×(1−D)
Eq. 14
The power dissipated in the input capacitor is:
P
DISS(CIN) = ICIN(RMS)2 × ESRCIN
Eq. 15
Ripple Injection
The VFB ripple required for proper operation of the
MIC26603 gm amplifier and error comparator is 20mV to
100mV. However, the output voltage ripple is generally
designed as 1% to 2% of the output voltage. For a low
output voltage, such as a 1V, the output voltage ripple is
only 10mV to 20mV, and the feedback voltage ripple is
less than 20mV. If the feedback voltage ripple is so small
that the gm amplifier and error comparator can’t sense it,
then the MIC26603 will lose control and the output
voltage is not regulated. In order to have some amount
of VFB ripple, a ripple injection method is applied for low
output voltage ripple applications.
The voltage rating of the capacitor should be twice the
output voltage for a tantalum and 20% greater for
aluminum electrolytic or OS-CON. The output capacitor
RMS current is calculated in Equation 11:
ΔIL(PP)
IC
=
Eq. 11
OUT (RMS)
12
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MIC26603
The applications are divided into three situations
according to the amount of the feedback voltage ripple:
1. Enough ripple at the feedback voltage due to the
large ESR of the output capacitors.
As shown in Figure 6a, the converter is stable without
any ripple injection. The feedback voltage ripple is:
Figure 6c. Invisible Ripple at FB
R2
ΔVFB(pp)
=
× ESRC
× ΔIL
Eq. 16
(pp)
OUT
R1+ R2
In this situation, the output voltage ripple is less than
20mV. Therefore, additional ripple is injected into the FB
pin from the switching node SW via a resistor Rinj and a
capacitor Cinj, as shown in Figure 6c. The injected ripple
is:
where ΔIL(pp) is the peak-to-peak value of the inductor
current ripple.
2. Inadequate ripple at the feedback voltage due to the
small ESR of the output capacitors.
1
ΔVFB(pp) = V ×Kdiv ×D×(1-D)×
Eq. 18
Eq. 19
The output voltage ripple is fed into the FB pin through a
feedforward capacitor Cff in this situation, as shown in
Figure 6b. The typical Cff value is between 1nF and
100nF. With the feedforward capacitor, the feedback
voltage ripple is very close to the output voltage ripple:
IN
fSW ×τ
R1//R2
Kdiv
=
Rinj + R1//R2
where
ΔVFB(pp) ≈ ESR × ΔIL
Eq. 17
(pp)
VIN = Power stage input voltage
D = duty cycle
3. Virtually no ripple at the FB pin voltage due to the
very-low ESR of the output capacitors.
f
SW = switching frequency
τ = (R1//R2//Rinj) × Cff
In Equations 21 and 22, it is assumed that the time
constant associated with Cff must be much greater than
the switching period:
1
T
=
<<1
Eq. 20
fSW ×τ
τ
If the voltage divider resistors R1 and R2 are in the kꢀ
range, a Cff of 1nF to 100nF can easily satisfy the large
time constant requirements. Also, a 100nF injection
capacitor Cinj is used in order to be considered as short
for a wide range of the frequencies.
Figure 6a. Enough Ripple at FB
The process of sizing the ripple injection resistor and
capacitors is:
Step 1. Select Cff to feed all output ripples into the
feedback pin and make sure the large time constant
assumption is satisfied. Typical choice of Cff is 1nF to
100nF if R1 and R2 are in kꢀ range.
Figure 6b. Inadequate Ripple at FB
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MIC26603
Step 2. Select Rinj according to the expected feedback
voltage ripple using Equation 24:
In addition to the external ripple injection added at the
FB pin, internal ripple injection is added at the inverting
input of the comparator inside the MIC26603, as shown
in Figure 8. The inverting input voltage VINJ is clamped to
1.2V. As VOUT is increased, the swing of VINJ will be
clamped. The clamped VINJ reduces the line regulation
because it is reflected as a DC error on the FB terminal.
Therefore, the maximum output voltage of the MIC26603
should be limited to 5.5V to avoid this problem.
ΔVFB(pp)
fSW ×τ
D×(1-D)
Kdiv
=
×
Eq. 21
V
IN
Then the value of Rinj is obtained as:
1
Rinj = (R1//R2) × (
− 1)
Eq. 22
Kdiv
Step 3. Select Cinj as 100nF, which could be considered
as short for a wide range of the frequencies.
Setting Output Voltage
The MIC26603 requires two resistors to set the output
voltage as shown in Figure 7.
The output voltage is determined by the equation:
R1
VOUT = VFB × (1+
)
Eq. 23
R2
Figure 8. Internal Ripple Injection
Thermal Measurements
where, VFB = 0.8V.
A typical value of R1 can be between 3kꢀ and 10kꢀ. If
R1 is too large, it may allow noise to be introduced into
the voltage feedback loop. If R1 is too small, it will
decrease the efficiency of the power supply, especially
at light loads. Once R1 is selected, R2 can be calculated
using:
Measuring the IC’s case temperature is recommended to
insure it is within its operating limits. Although this might
seem like a very elementary task, it is easy to get
erroneous results. The most common mistake is to use
the standard thermal couple that comes with a thermal
meter. This thermal couple wire gauge is large, typically
22 gauge, and behaves like a heatsink, resulting in a
lower case measurement.
VFB ×R1
VOUT − VFB
R2 =
Eq. 24
Two methods of temperature measurement are using a
smaller thermal couple wire or an infrared thermometer.
If a thermal couple wire is used, it must be constructed
of 36 gauge wire or higher then (smaller wire size) to
minimize the wire heat-sinking effect. In addition, the
thermal couple tip must be covered in either thermal
grease or thermal glue to make sure that the thermal
couple junction is making good contact with the case of
the IC. Omega brand thermal couple (5SC-TT-K-36-36)
is adequate for most applications.
Wherever possible, an infrared thermometer is
recommended. The measurement spot size of most
infrared thermometers is too large for an accurate
reading on a small form factor ICs. However, a IR
thermometer from Optris has a 1mm spot size, which
makes it a good choice for measuring the hottest point
on the case. An optional stand makes it easy to hold the
beam on the IC for long periods of time.
Figure 7. Voltage-Divider Configuration
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MIC26603
Inductor
PCB Layout Guidelines
•
•
•
•
Keep the inductor connection to the switch node
(SW) short.
Warning!!! To minimize EMI and output noise, follow
these layout recommendations.
Do not route any digital lines underneath or close to
the inductor.
PCB Layout is critical to achieve reliable, stable and
efficient performance. A ground plane is required to
control EMI and minimize the inductance in power,
signal and return paths.
Keep the switch node (SW) away from the feedback
(FB) pin.
The following guidelines should be followed to insure
proper operation of the MIC26603 regulator.
The CS pin should be connected directly to the SW
pin to accurate sense the voltage across the low-
side MOSFET.
IC
•
•
To minimize noise, place a ground plane underneath
the inductor.
•
A 2.2µF ceramic capacitor, which is connected to
the PVDD pin, must be located right at the IC. The
PVDD pin is very noise sensitive and placement of
the capacitor is very critical. Use wide traces to
connect to the PVDD and PGND pins.
The inductor can be placed on the opposite side of
the PCB with respect to the IC. It does not matter
whether the IC or inductor is on the top or bottom as
long as there is enough air flow to keep the power
components within their temperature limits. The
input and output capacitors must be placed on the
same side of the board as the IC.
•
A 1.0uF ceramic capacitor must be placed right
between VDD and the signal ground SGND. The
SGND must be connected directly to the ground
planes. Do not route the SGND pin to the PGND
Pad on the top layer.
Output Capacitor
•
Use a wide trace to connect the output capacitor
ground terminal to the input capacitor ground
terminal.
•
•
Place the IC close to the point-of-load (POL).
Use fat traces to route the input and output power
lines.
•
Phase margin will change as the output capacitor
value and ESR changes. Contact the factory if the
output capacitor is different from what is shown in
the BOM.
•
Signal and power grounds should be kept separate
and connected at only one location.
Input Capacitor
•
The feedback trace should be separate from the
power trace and connected as close as possible to
the output capacitor. Sensing a long high current
load trace can degrade the DC load regulation.
•
•
Place the input capacitor next.
Place the input capacitors on the same side of the
board and as close to the IC as possible.
•
•
•
•
Keep both the PVIN pin and PGND connections
short.
Optional RC Snubber
•
Place the RC snubber on either side of the board
and as close to the SW pin as possible.
Place several vias to the ground plane close to the
input capacitor ground terminal.
Use either X7R or X5R dielectric input capacitors.
Do not use Y5V or Z5U type capacitors.
Do not replace the ceramic input capacitor with any
other type of capacitor. Any type of capacitor can be
placed in parallel with the input capacitor.
•
•
If a Tantalum input capacitor is placed in parallel
with the input capacitor, it must be recommended for
switching regulator applications and the operating
voltage must be derated by 50%.
In “Hot-Plug” applications, a Tantalum or Electrolytic
bypass capacitor must be used to limit the over-
voltage spike seen on the input supply with power is
suddenly applied.
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MIC26603
Evaluation Board Schematic
Figure 9. Schematic of MIC26603 Evaluation Board
(J11, R13, R15 are for testing purposes)
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MIC26603
Bill of Materials
Item
Part Number
Manufacturer
Description
Qty.
C1
Open
12105C475KAZ2A
GRM32ER71H475KA88L
C3225X7R1H475K
Open
AVX(1)
Murata(2)
TDK(3)
C2, C3
4.7µF Ceramic Capacitor, X7R, Size 1210, 50V
2
C4, C13, C15
C5
12106D107MAT2A
GRM32ER60J107ME20L
C3225X5R0J107M
06035C104KAT2A
GRM188R71H104KA93D
C1608X7R1H104K
0603ZC105KAT2A
GRM188R71A105KA61D
C1608X7R1A105K
0603ZD225KAT2A
GRM188R61A225KE34D
C1608X5R1A225K
06035C472KAZ2A
GRM188R71H472K
C1608X7R1H472K
B41851F7227M
AVX(1)
Murata(2)
TDK(3)
AVX(1)
Murata(2)
TDK(3)
AVX(1)
Murata(2)
TDK(3)
AVX(1)
Murata(2)
TDK(3)
100µF Ceramic Capacitor, X5R, Size 1210, 6.3V
0.1µF Ceramic Capacitor, X7R, Size 0603, 50V
1.0µF Ceramic Capacitor, X7R, Size 0603, 10V
2.2µF Ceramic Capacitor, X5R, Size 0603, 10V
1
3
1
1
C6, C7, C10
C8
C9
AVX(1)
Murata(2)
TDK(3)
C12
4.7nF Ceramic Capacitor, X7R, Size 0603, 50V
220µF Aluminum Capacitor, 35V
1
1
C14
EPCOS(4)
C11, C16
Open
SD103AWS
MCC(5)
Diodes Inc(6)
Vishay(7)
D1
40V, 350mA, Schottky Diode, SOD323
1
SD103AWS-7
SD103AWS
Cooper Bussmann(8) 2.2µH Inductor, 15A Saturation Current
1
1
1
1
1
1
3
1
1
1
1
1
1
L1
HCF1305-2R2-R
R1
CRCW06032R21FKEA
Vishay Dale(7)
Vishay Dale(7)
Vishay Dale(7)
Vishay Dale(7)
Vishay Dale(7)
Vishay Dale(7)
Vishay Dale(7)
Vishay Dale(7)
Vishay Dale(7)
Vishay Dale(7)
Vishay Dale(7)
Vishay Dale(7)
2.21ꢀ Resistor, Size 0603, 1%
2.00ꢀ Resistor, Size 0603, 1%
19.6kꢀ Resistor, Size 0603, 1%
2.49kꢀ Resistor, Size 0603, 1%
20.0kꢀ Resistor, Size 0603, 1%
10.0kꢀ Resistor, Size 0603, 1%
4.99kꢀ Resistor, Size 0603, 1%
2.87kꢀ Resistor, Size 0603, 1%
2.00kꢀ Resistor, Size 0603, 1%
1.18kꢀ Resistor, Size 0603, 1%
806ꢀ Resistor, Size 0603, 1%
475ꢀ Resistor, Size 0603, 1%
R2
CRCW06032R00FKEA
CRCW060319K6FKEA
CRCW06032K49FKEA
CRCW060320K0FKEA
CRCW060310K0FKEA
CRCW06034K99FKEA
CRCW06032K87FKEA
CRCW06032K006FKEA
CRCW06031K18FKEA
CRCW0603806RFKEA
CRCW0603475RFKEA
R3
R4
R5
R6, R14, R17
R7
R8
R9
R10
R11
R12
25
M9999-071311-A
July 2011
Micrel, Inc.
MIC26603
Bill of Materials (Continued)
Item
Part Number
Manufacturer
Vishay Dale(7)
Vishay Dale(7)
Vishay Dale(7)
Description
Qty.
R13
CRCW06030000FKEA
CRCW060349R9FKEA
CRCW06031R21FKEA
Open
0ꢀ Resistor, Size 0603, 5%
49.9ꢀ Resistor, Size 0603, 1%
1.21ꢀ Resistor, Size 0603, 1%
1
1
2
R15
R16, R18
R20
28V, 6A Hyper Light Load™ Synchronous
DC/DC Buck Regulator
U1
MIC26603YJL
Micrel. Inc.(9)
1
Notes:
1. AVX: www.avx.com.
2. Murata: www.murata.com.
3. TDK: www.tdk.com.
4. EPCOS: www.epcos.com.
5. SANYO: www.sanyo.com.
6. Diode Inc.: www.diodes.com.
7. Vishay: www.vishay.com.
8. Cooper Bussmann: www.cooperbussmann.com.
9. Micrel, Inc.: www.micrel.com.
26
M9999-071311-A
July 2011
Micrel, Inc.
MIC26603
PCB Layout Recommendations
Figure 10. MIC26603 Evaluation Board Top Layer
Figure 11. MIC26603 Evaluation Board Mid-Layer 1 (Ground Plane)
27
M9999-071311-A
July 2011
Micrel, Inc.
MIC26603
PCB Layout Recommendations (Continued)
Figure 12. MIC26603 Evaluation Board Mid-Layer 2
Figure 13. MIC26603 Evaluation Board Bottom Layer
28
M9999-071311-A
July 2011
Micrel, Inc.
MIC26603
Recommended Land and Solder Stencil Pattern
29
M9999-071311-A
July 2011
Micrel, Inc.
MIC26603
Package Information
28-Pin 5mm x 6mm MLF® (YJL)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
Micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this data sheet. This
information is not intended as a warranty and Micrel does not assume responsibility for its use. Micrel reserves the right to change circuitry,
specifications and descriptions at any time without notice. No license, whether express, implied, arising by estoppel or otherwise, to any intellectual
property rights is granted by this document. Except as provided in Micrel’s terms and conditions of sale for such products, Micrel assumes no liability
whatsoever, and Micrel disclaims any express or implied warranty relating to the sale and/or use of Micrel products including liability or warranties
relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright or other intellectual property right.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant
into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully
indemnify Micrel for any damages resulting from such use or sale.
© 2011 Micrel, Incorporated.
30
M9999-071311-A
July 2011
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