MIC28513 [MICROCHIP]
45V, 4A Synchronous Buck Regulator;
| 型号： | MIC28513 |
| 厂家： | 
MICROCHIP |
| 描述： | 45V, 4A Synchronous Buck Regulator |
| 文件： | 总34页 (文件大小：1514K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
MIC28513
45V, 4A Synchronous Buck Regulator
Features
General Description
• 4.6V to 45V Operating Input Voltage Supply
• Up to 4A Output Current
The MIC28513 is a synchronous step-down switching
regulator with internal power switches capable of
providing up to 4A output current from a wide input
supply range from 4.6V to 45V. The output voltage is
adjustable down to 0.8V with a guaranteed accuracy of
• Integrated High-Side and Low-Side N-Channel
MOSFETs
• HyperLight Load (MIC28513-1) and Hyper Speed
Control (MIC28513-2) Architecture
±1%.
A
constant switching frequency can be
programmed from 200 kHz to 680 kHz. The
MIC28513’s Hyper Speed Control® and HyperLight
• Enable Input and Power Good (PGOOD) Output
Load® architectures allow for high VIN (low VOUT
)
• Programmable Current-Limit and Foldback
“Hiccup” Mode Short-Circuit Protection
operation and ultra-fast transient response while
reducing the required output capacitance. The
MIC28513-1’s HyperLight Load architecture also
provides very good light load efficiency.
• Built-In 5V Regulator for Single-Supply Operation
• Adjustable 200 kHz to 680 kHz Switching
Frequency
The MIC28513 offers a full suite of features to ensure
protection under fault conditions. These include
undervoltage lockout to ensure proper operation under
power sag conditions, internal soft-start to reduce
inrush current, foldback current limit, “hiccup” mode
short-circuit protection, and thermal shutdown.
• Fixed 5 ms Soft-Start
• Internal Compensation and Thermal Shutdown
• Thermally-Enhanced 24-Pin 3 mm x 4 mm FQFN
Package
• –40°C to +125°C Junction Temperature Range
Applications
• Industrial Power Supplies
• Distributed Supply Regulation
• Base Station Power Supplies
• Wall Transformer Regulation
• High-Voltage Single-Board Systems
Package Type
MIC28513
24-Pin 3 mm x 4 mm FQFN (FL)
23
22
21
24
20
1
2
3
4
5
DL
PVDD
19
18
PGND
DH
VDD
ILIM
PVIN
LX
17 VIN
16
EN
6
7
8
BST
PGOOD
15
14
PVIN
FB
AGND
13
PVIN
9
10
11
12
2016 Microchip Technology Inc.
DS20005522A-page 1
MIC28513
Typical Application Circuit
MIC28513
3x4 FQFN
2.2μF
VDD
PVDD
BST
10Ω
0.1μF
6.8μH
MIC28513
2.2kΩ
ILIM
VOUT
VIN
EN
5V (0A to 4A)
5.5V to 45V
SW
LX
PVIN
VIN
100kΩ
10.0kΩ
470pF
100kΩ
100kΩ
0.1μF
47μF x2
FREQ
FB
1.91kΩ
AGND
PGND
Functional Block Diagram
VIN
CIN
DBST
CVDD
VIN
VDD PVDD
PVIN
17
19
20
4, 7, 8, 9, 25
BST
LINEAR
REGULATOR
6
UVLO
RBST
DH
3
THERMAL
SHUTDOWN
M1
CBST
OFF
ON
L
12, SW
21,
16
24
VOUT
R3
R4
27
EN
FIXED T
LX
5
ESTIMATIOONN
ZCD
CONTROL
LOGIC
FREQ
SOFT-START
COUT
1
DL
RLIM
PGND
10,
PVDD
M2
3.3V
RPGOOD
POWER GOOD
COMPARATOR
11,
22,
23,
26
15
CURRENT
LIMIT
DETECTION
PGOOD
RINJ
X90%
2
PGND
ILIM
VREF
0.8V
CINJ
18
R1
13
AGND
14
CFF
FB
R2
DS20005522A-page 2
2016 Microchip Technology Inc.
MIC28513
1.0
ELECTRICAL CHARACTERISTICS
Absolute Maximum Ratings †
PVIN, VIN to PGND..................................................................................................................................... –0.3V to +50V
V
V
DD, PVDD to PGND..................................................................................................................................... –0.3V to +6V
BST to VSW, VLX ......................................................................................................................................... –0.3V to +6V
VBST to PGND......................................................................................................................................–0.3V to (VIN + 6V
VSW, VLX to PGND...........................................................................................................................–0.3V to (VIN + 0.3V)
VFREQ, VILIM, VEN to AGND.............................................................................................................–0.3V to (VIN + 0.3V)
VLX, VFB, VPG, VFREQ, VILIM, VEN to AGND................................................................................... –0.3V to (VDD + 0.3V)
PGND to AGND ........................................................................................................................................ –0.3V to +0.3V
ESD Rating(1) (HBM) ..............................................................................................................................................1.5 kV
ESD Rating(1) (MM) ..................................................................................................................................................150V
Operating Ratings ‡
Supply Voltage (PVIN, VIN)......................................................................................................................... +4.6V to +45V
Enable Input (VEN)..............................................................................................................................................0V to VIN
VSW, VFREQ, VILIM, VEN ......................................................................................................................................0V to VIN
† Notice: Stresses above those listed under “Absolute Maximum Ratings” may cause permanent damage to the device.
This is a stress rating only and functional operation of the device at those or any other conditions above those indicated
in the operational sections of this specification is not intended. Exposure to maximum rating conditions for extended
periods may affect device reliability.
‡ Notice: The device is not guaranteed to function outside its operating ratings.
Note 1: Devices are ESD sensitive. Handling precautions are recommended. Human body model, 1.5 kΩ in series
with 100 pF.
2016 Microchip Technology Inc.
DS20005522A-page 3
MIC28513
TABLE 1-1:
ELECTRICAL CHARACTERISTICS
Electrical Characteristics: VIN = 12V, TA = 25°C, unless noted. Bold values indicate –40°C ≤ TJ ≤ +125°C.
(Note 1).
Parameters
Min.
Typ.
Max.
Units
Conditions
Power Supply Input
Input Voltage Range (PVIN,
VIN)
4.6
—
45
V
—
Quiescent Supply Current
—
—
—
0.4
0.7
0.1
0.75
1.5
10
mA
VFB = 1.5V (MIC28513-1)
VFB = 1.5V (MIC28513-2)
SW unconnected, VEN = 0V
Shutdown Supply Current
VDD Supply
µA
VDD Output Voltage
4.8
3.8
—
5.2
4.2
400
2
5.4
4.6
—
V
V
VIN = 7V to 45V, IVDD = 10 mA
V
DD UVLO Threshold
DD UVLO Hysteresis
VDD rising
V
mV
%
—
—
Load Regulation at 40 mA
Reference
0.6
4.0
Feedback Reference Voltage
0.792
0.784
—
0.8
0.8
5
0.808
0.816
500
V
nA
V
25°C (±1%)
–40°C ≤ TJ ≤ +125°C (±2%)
FB Bias Current
Enable Control
EN Logic Level High
EN Logic Level Low
EN Hysteresis
VFB = 0.8V
1.8
—
—
—
—
—
—
0.6
—
—
—
—
200
5
mV
µA
EN Bias Current
Oscillator
40
V
EN = 12V
Switching Frequency
450
—
680
340
85
800
—
kHz
%
V
FREQ = VIN
VFREQ = 50% VIN
Maximum Duty Cycle
Minimum Duty Cycle
Minimum Off-Time
Internal MOSFET
—
—
—
—
0
—
VFB > 0.8V
—
110
200
270
ns
High-Side NMOS
On-Resistance
—
—
37
20
—
—
mꢀ
—
—
Low-Side NMOS
On-Resistance
Short-Circuit Protection
Current-Limit Threshold
Short-Circuit Threshold
–30
–24
50
–14
–7
0
8
mV
µA
V
FB = 0.79V
VFB = 0V
FB = 0.79V
VFB = 0V
Current-Limit Source Current
Short-Circuit Source Current
70
90
43
V
25
36
Note 1: Specification for packaged product only.
DS20005522A-page 4
2016 Microchip Technology Inc.
MIC28513
TABLE 1-1:
ELECTRICAL CHARACTERISTICS (CONTINUED)
Electrical Characteristics: VIN = 12V, TA = 25°C, unless noted. Bold values indicate –40°C ≤ TJ ≤ +125°C.
(Note 1).
Parameters
Min.
Typ.
Max.
Units
Conditions
Leakage
SW, BST Leakage Current
Power Good (PGOOD)
PGOOD Threshold Voltage
PGOOD Hysteresis
—
—
50
µA
—
85
—
—
—
90
6
95
—
%VOUT Sweep VFB from low to high
Sweep VFB from high to low
PGOOD Delay Time
100
70
—
µs
Sweep VFB from low to high
FB < 90% x VNOM, IPGOOD = 1 mA
PGOOD Low Voltage
Thermal Protection
200
mV
V
Overtemperature Shutdown
—
—
160
15
—
—
°C
°C
TJ Rising
—
Overtemperature Shutdown
Hysteresis
Soft-Start
Soft-Start Time
—
5
—
ms
—
Note 1: Specification for packaged product only.
2016 Microchip Technology Inc.
DS20005522A-page 5
MIC28513
TEMPERATURE SPECIFICATIONS
Parameters
Temperature Ranges
Sym.
Min.
Typ.
Max.
Units
Conditions
Junction Operating Temperature
Storage Temperature Range
Junction Temperature
TJ
TS
TJ
—
–40
–65
—
—
—
—
—
+125
+150
+150
+300
°C
°C
°C
°C
Note 1
—
—
Lead Temperature
—
Soldering, 10s
—
Package Thermal Resistances
Thermal Resistance 3 mm x 4 mm
FQFN-24LD
JA
—
30
—
°C/W
Note 1: The maximum allowable power dissipation is a function of ambient temperature, the maximum allowable
junction temperature and the thermal resistance from junction to air (i.e., TA, TJ, JA). Exceeding the
maximum allowable power dissipation will cause the device operating junction temperature to exceed the
maximum +125°C rating. Sustained junction temperatures above +125°C can impact the device reliability.
DS20005522A-page 6
2016 Microchip Technology Inc.
MIC28513
2.0
TYPICAL PERFORMANCE CURVES
Note: The graphs and tables provided following this note are a statistical summary based on a limited number of
samples and are provided for informational purposes only. The performance characteristics listed herein
are not tested or guaranteed. In some graphs or tables, the data presented may be outside the specified
operating range (e.g., outside specified power supply range) and therefore outside the warranted range.
FIGURE 2-1:
Switching Frequency vs.
FIGURE 2-4:
V
Voltage vs. Input
DD
Output Voltage (MIC28513-1).
Voltage.
FIGURE 2-2:
Feedback Voltage vs.
FIGURE 2-5:
V
UVLO Threshold vs.
DD
Temperature (MIC28513-1).
Temperature (MIC28513-1).
FIGURE 2-3:
Temperature (MIC28513-2).
Feedback Voltage vs.
FIGURE 2-6:
vs. V ).
Line Regulation Error (V
OUT
IN
2016 Microchip Technology Inc.
DS20005522A-page 7
MIC28513
.
FIGURE 2-7:
Enable Threshold vs. Input
FIGURE 2-10:
Output Voltage vs. Output
Voltage.
Current (MIC28513-2).
FIGURE 2-11:
Output Current (MIC28513-2).
Switching Frequency vs.
FIGURE 2-8:
Current vs. Input Voltage (MIC28513-1).
V
Operating Supply
IN
FIGURE 2-12:
Output Peak Current Limit
FIGURE 2-9:
V
Operating Supply
IN
vs. Temperature (MIC28513-1).
Current vs. Input Voltage (MIC28513-2).
DS20005522A-page 8
2016 Microchip Technology Inc.
MIC28513
FIGURE 2-13:
Output Peak Current Limit
FIGURE 2-16:
Efficiency (V = 36V) vs.
IN
vs. Temperature (MIC28513-2).
Output Current (MIC28513-1).
FIGURE 2-14:
Efficiency (V = 12V) vs.
FIGURE 2-17:
IC Power Dissipation vs.
IN
Output Current (MIC28513-1).
Output Current (V = 12V).
IN
FIGURE 2-15:
Efficiency (V = 24V) vs.
FIGURE 2-18:
IC Power Dissipation vs.
IN
Output Current (MIC28513-1).
Output Current (V = 24V).
IN
2016 Microchip Technology Inc.
DS20005522A-page 9
MIC28513
FIGURE 2-19:
IC Power Dissipation vs.
FIGURE 2-22:
36V Input Thermal Derating.
Output Current (V = 36V).
IN
FIGURE 2-20:
12V Input Thermal Derating.
FIGURE 2-23:
Efficiency (V = 12V) vs.
IN
Output Current (MIC28513-2).
FIGURE 2-24:
Output Current (MIC28513-2).
Efficiency (V = 24V) vs.
IN
FIGURE 2-21:
24V Input Thermal Derating.
DS20005522A-page 10
2016 Microchip Technology Inc.
MIC28513
VEN
(10V/div)
VIN = 12V
VOUT = 5V
VOUT
(2V/div)
I
OUT = 4A
IIL
(5A/div)
Time (2ms/div)
FIGURE 2-25:
Efficiency (V = 36V) vs.
FIGURE 2-28:
Enable Turn-On.
IN
Output Current (MIC28513-2).
VEN
(10V/div)
VIN
(10V/div)
VOUT
VIN = 12V
VOUT = 5V
OUT = 4A
(2V/div)
I
VOUT
(2V/div)
VIN = 12V
VOUT = 5V
I
OUT = 4A
V
(5V/diSvW)
IIL
(5A/div)
IIL
(5A/div)
Time (400μs/div)
Time (2ms/div)
FIGURE 2-29:
Enable Turn-Off.
FIGURE 2-26:
Turn-On.
VIN
(10V/div)
VOUT
VIN = 12V
VOUT = 5V
OUT = 0A
VPRE-BIAS = 2V
I
(2V/div)
VIN = 12V
VOUT = 5V
V
I
OUT = 4A
(5V/diSvW)
VOUT
(2V/div)
V
(10V/diSvW)
IIL
(5A/div)
Time (1ms/div)
Time (200μs/div)
FIGURE 2-30:
MIC28513-1 V Start-Up
IN
FIGURE 2-27:
Turn-Off.
with Pre-Biased Output.
2016 Microchip Technology Inc.
DS20005522A-page 11
MIC28513
VIN = 12V
VOUT = 5V
I
OUT = NL
VPRE-BIAS = 2V
VOUT = 3.3V
IOUT = 0.5A
VEN = 5V
VOUT
(2V/div)
VIN
(2V/div)
VOUT
(2V/div)
V
(10V/diSvW)
Time (2ms/div)
Time (20ms/div)
FIGURE 2-31:
MIC28513-2 V Start-Up
FIGURE 2-34:
V
UVLO Thresholds.
IN
IN
with Pre-Biased Output.
VIN = 12V
VOUT = 5V
VIN
I
OUT = 4A
(10V/div)
VPRE-BIAS = 2V
VOUT
(2V/div)
VIN = 12V
VOUT = short
VOUT
(2V/div)
V
(5V/diSvW)
IL
V
(10V/diSvW)
(5A/div)
Time (1ms/div)
Time (1ms/div)
FIGURE 2-35:
Turn-On Into Short-Circuit.
FIGURE 2-32:
MIC28513-1 V Start-Up
IN
with Pre-Biased Output.
VIN = 12V
VOUT = 5V
VIN = 12V
VOUT = 5V
VEN
(2V/div)
I
OUT = 4A
I
OUT = short
VPRE-BIAS = 2V
VOUT
(100mV/div)
VOUT
(2V/div)
V
(10V/diSvW)
IL
V
(10V/diSvW)
(5A/div)
Time (4ms/div)
Time (2ms/div)
FIGURE 2-36:
Enabled Into Short-Circuit.
FIGURE 2-33:
MIC28513-2 V Start-Up
IN
with Pre-Biased Output.
DS20005522A-page 12
2016 Microchip Technology Inc.
MIC28513
VIN = 12V
VOUT = 5V
VOUT
(2V/div)
VOUT
(2V/div)
IOUT
(5A/div)
V
V
(10V/diSvW)
(5V/diSvW)
Time (20μs/div)
Time (4ms/div)
FIGURE 2-37:
Overcurrent Protection.
FIGURE 2-40:
Output Recovery from
Thermal Shutdown.
VIN = 12V
VOUT = 5V
IOUT = 0A
VOUT
(50mV/div)
AC-Coupled
VOUT
(2V/div)
IL
(5A/div)
V
V
(10V/diSvW)
(10V/diSvW)
Time (100μs/div)
Time (200μs/div)
FIGURE 2-38:
Overcurrent Protection
FIGURE 2-41:
MIC28513-1 Switching
Retry.
Waveforms (I
= 0A).
OUT
VIN = 12V
VOUT = 5V
VOUT
(20mV/div)
AC-Coupled
I
OUT = 0A
VOUT
(2V/div)
IL
(5A/div)
V
V
(10V/diSvW)
(10V/diSvW)
Time (4ms/div)
Time (1μs/div)
FIGURE 2-39:
Thermal Shutdown.
Output Recovery from
FIGURE 2-42:
Waveforms (I
MIC28513-2 Switching
= 0A).
OUT
2016 Microchip Technology Inc.
DS20005522A-page 13
MIC28513
VIN = 12V
VOUT = 5V
VOUT
(200mV/div)
AC-Coupled
VOUT
(20mV/div)
AC-Coupled
I
OUT = 4A
VIN = 12V
VOUT = 5V
I
OUT = 0A to 4A
V
IOUT
(2A/div)
(10V/diSvW)
Time (1μs/div)
Time (1ms/div)
FIGURE 2-43:
MIC28513-1 Switching
= 4A).
FIGURE 2-46:
Response (0A to 4A).
MIC28513-2 Transient
Waveforms (I
OUT
VIN = 12V
VOUT = 5V
VOUT
(200mV/div)
AC-Coupled
I
OUT = 4A
VOUT
(20mV/div)
AC-Coupled
V
IL
(10V/diSvW)
(1A/div)
Time (1μs/div)
Time (1ms/div)
FIGURE 2-44:
MIC28513-2 Switching
= 4A).
FIGURE 2-47:
Response (0A to 1.3A).
MIC28513-1 Transient
Waveforms (I
OUT
VOUT
(200mV/div)
AC-Coupled
VOUT
(200mV/div)
AC-Coupled
VIN = 12V
VOUT = 5V
I
OUT = 0A to 1.3A
IL
IOUT
(1A/div)
(2A/div)
Time (1ms/div)
Time (1ms/div)
FIGURE 2-45:
MIC28513-1 Transient
FIGURE 2-48:
MIC28513-2 Transient
Response (0A to 4A).
Response (0A to 1.3A).
DS20005522A-page 14
2016 Microchip Technology Inc.
MIC28513
VOUT
(100mV/div)
AC-Coupled
VOUT
(100mV/div)
AC-Coupled
VIN = 12V
VOUT = 5V
I
OUT = 2.6 to 4A
IL
IOUT
(2A/div)
(1A/div)
Time (1ms/div)
Time (1ms/div)
FIGURE 2-49:
MIC28513-1 Transient
FIGURE 2-52:
MIC28513-2 Transient
Response (1.3A to 2.6A).
Response (2.6A to 4A).
VOUT
(100mV/div)
AC-Coupled
VOUT
(50mV/div)
AC-Coupled
VIN = 12V
VOUT = 5V
VIN = 12V to 60V
VOUT = 5V
I
OUT = 1.3A to 2.6A
I
OUT = 3A
VIN
(10V/div)
IOUT
(1A/div)
V
(20V/diSvW)
Time (1ms/div)
Time (4ms/div)
FIGURE 2-50:
MIC28513-2 Transient
FIGURE 2-53:
Input Voltage Transient
Response (1.3A to 2.6A).
Response.
VOUT
(100mV/div)
AC-Coupled
VOUT
(50mV/div)
AC-Coupled
VIN = 12V to 60V
VOUT = 5V
I
OUT = 3A
VIN
(10V/div)
IL
V
(2A/div)
(20V/diSvW)
Time (1ms/div)
Time (4ms/div)
FIGURE 2-51:
MIC28513-1 Transient
FIGURE 2-54:
Input Voltage Transient
Response (2.6A to 4A).
Response.
2016 Microchip Technology Inc.
DS20005522A-page 15
MIC28513
3.0
PIN DESCRIPTIONS
The descriptions of the pins are listed in Table 3-1.
TABLE 3-1:
Pin Number
PIN FUNCTION TABLE
Symbol
Description
1
2
3
DL
PGND
DH
Low-Side Gate Drive. Internal low-side power MOSFET gate connection. This pin must
be left unconnected or floating.
PGND is the return path for the low-side driver circuit. Connect to the source of low-side
MOSFET (PGND, pins 10, 11 22, 23, and 26) through a low-impedance path.
High-Side Gate Drive. Internal high-side power MOSFET gate connection. This pin
must be left unconnected or floating.
4, 7, 8, 9, 25
(25 is ePad)
PVIN
Power Input Voltage. The PVIN pins supply power to the internal power switch. Connect
all PVIN pins together and bypass locally with ceramic capacitors. The positive terminal
of the input capacitor should be placed as close as possible to the PVIN pins, the
negative terminal of the input capacitor should be placed as close as possible to the
PGND pins 10,11, 22, 23, and 26.
5
6
LX
The LX pin is the return path for the high-side driver circuit. Connect the negative
terminal of the bootstrap capacitor directly to this pin. Also connect this pin to the SW
pins 12, 21, and 27, with a low-impedance path. The controller monitors voltages on this
and PGND for zero current detection.
BST
Bootstrap Pin. This pin provides bootstrap supply for the high-side gate driver circuit.
Connect a 0.1 µF capacitor and an optional resistor in series from the LX (pin 5) to the
BST pin.
10, 11, 22, 23,
26
(26 is ePad)
PGND
Power Ground. These pins are connected to the source of the low-side MOSFET. They
are the return path for the step-down regulator power stage and should be tied together.
The negative terminal of the input decoupling capacitor should be placed as close as
possible to these pins.
12, 21, 27
(27 is ePad)
SW
AGND
FB
Switch Node. The SW pins are the internal power switch outputs. These pins should be
tied together and connected to the output inductor.
13
Analog Ground. The analog ground for VDD and the control circuitry. The analog ground
return path should be separate from the power ground (PGND) return path.
14
Feedback Input. The FB pin sets the regulated output voltage relative to the internal
reference. This pin is connected to a resistor divider from the regulated output such that
the FB pin is at 0.8V when the output is at the desired voltage.
15
16
17
PGOOD The power good output is an open drain output requiring an external pull-up resistor to
external bias. This pin is a high impedance open circuit when the voltage at FB pin is
higher than 90% of the feedback reference voltage (typically 0.8V).
EN
Enable Input. The EN pin enables the regulator. When the pin is pulled below the
threshold, the regulator will shut down to an ultra-low current state. A precise threshold
voltage allows the pin to operate as an accurate UVLO. Do not tie EN to VDD
VIN
Supply voltage for the internal LDO. The VIN operating voltage range is from 4.6V to
45V. A ceramic capacitor from VIN to AGND is required for decoupling. The decoupling
capacitor should be placed as close as possible to the supply pin.
18
19
ILIM
Current Limit Setting. Connect a resistor from this pin to the SW pin node to allow for
accurate current limit sensing programming of the internal low-side power MOSFET.
VDD
Internal +5V Linear Regulator: VDD is the internal supply bus for the IC. Connect to an
external 1 µF bypass capacitor. When VIN is <5.5V, this regulator operates in drop-out
mode. Connect VDD to VIN.
20
24
PVDD
A 5V supply input for the low-side N-channel MOSFET driver circuit, which can be tied
to VDD externally. A 1 μF ceramic capacitor from PVDD to PGND is recommended for
decoupling.
FREQ
Switching Frequency Adjust pin. Connect this pin to VIN to operate at 680 kHz. Place a
resistor divider network from VIN to the FREQ pin to program the switching frequency.
DS20005522A-page 16
2016 Microchip Technology Inc.
MIC28513
EQUATION 4-2:
DMAX = 1 – tOFFMIN fSW
4.0
FUNCTIONAL DESCRIPTION
The MIC28513 is an adaptive on-time synchronous
buck regulator with integrated high-side and low-side
MOSFETs suitable for high-input voltage to low-output
voltage conversion applications. It is designed to
operate over a wide input voltage range, from 4.6V to
45V, which is suitable for automotive and industrial
applications. The output is adjustable with an external
resistive divider. An adaptive on-time control scheme is
employed to produce a constant switching frequency in
continuous-conduction mode and reduced switching
frequency in discontinuous-operation mode, improving
light-load efficiency. Overcurrent protection is
implemented by sensing the low-side MOSFET’s
RDS(ON). The device features internal soft-start, enable,
UVLO, and thermal shutdown.
It is not recommended to use MIC28513 with an
OFF-time close to tOFF(MIN) during steady-state
operation.
The adaptive ON-time control scheme results in a
constant switching frequency in the MIC28513. The
actual ON-time and resulting switching frequency will
vary with the different rising and falling times of the
external MOSFETs. Also, the minimum tON results in a
lower switching frequency in high VIN to VOUT
applications. During load transients, the switching
frequency is changed due to the varying OFF-time.
Figure 4-1 shows the allowable range of the output
voltage versus the input voltage. The minimum output
voltage is 0.8V which is limited by the reference
voltage. The maximum output voltage is 24V which is
limited by the internal circuitry.
4.1
Theory of Operation
As illustrated in the Functional Block Diagram, the
output voltage is sensed by the feedback (FB) pin via
voltage dividers R1 and R2, and compared to a 0.8V
reference voltage VREF at the error comparator through
a low-gain transconductance (gM) amplifier. If the
feedback voltage decreases and the amplifier output is
below 0.8V, then the error comparator will trigger the
control logic and generate an ON-time period. The
ON-time period length is predetermined by the fixed
tON estimator circuitry:
30
25
fSW = 600kHz
20
fSW = 400kHz
fSW = 200kHz
15
10
5
ALLOWABLE RANGE
0.8V (MINIMUM)
EQUATION 4-1:
VOUT
tONESTIMATED = -----------------------
VIN fSW
0
5
45
15
35
55
25
INPUT VOLTAGE (V)
Where:
FIGURE 4-1:
Range vs. Input Voltage.
Allowable Output Voltage
VOUT
VIN
Output Voltage
Power Stage Input Voltage
Switching Frequency
To illustrate the control loop operation, both the
steady-state and load transient scenarios will be
analyzed.
fSW
At the end of the ON-time period, the internal high-side
driver turns off the high-side MOSFET and the low-side
driver turns on the low-side MOSFET. The OFF-time
period length depends upon the feedback voltage in
most cases. When the feedback voltage decreases
and the output of the gM amplifier is below 0.8V, then
the ON-time period is triggered and the OFF-time
period ends. If the OFF-time period determined by the
feedback voltage is less than the minimum OFF-time
Figure 4-2 shows the MIC28513 control loop timing
during steady-state operation. During steady-state, the
gM amplifier senses the feedback voltage ripple, which
is proportional to the output voltage ripple and the
inductor current ripple, to trigger the ON-time period.
The ON-time is predetermined by the tON estimator.
The termination of the OFF-time is controlled by the
feedback voltage. At the valley of the feedback voltage
ripple, which occurs when VFB falls below VREF, the
OFF period ends and the next ON-time period is
triggered through the control logic circuitry.
tOFF(MIN)
, which is about 200 ns (typical), the
MIC28513 control logic will apply the tOFF(MIN) instead.
The tOFF(MIN) is required to maintain enough energy in
the boost capacitor (CBST) to drive the high-side
MOSFET.
The maximum duty cycle is obtained from
Equation 4-2.
2016 Microchip Technology Inc.
DS20005522A-page 17
MIC28513
current ripple if the ESR of the output capacitor is large
enough. The MIC28513 control loop has the advantage
of eliminating the need for slope compensation.
IL
IOUT
¨IL(PP)
In order to meet the stability requirements, the
MIC28513 feedback voltage ripple should be in phase
with the inductor current ripple and large enough to be
sensed by the gM amplifier and the error comparator.
The recommended feedback voltage ripple is 20 mV ~
100 mV.
VOUT
¨VOUT(PP) = ESRCOUT× ¨IL(PP)
VFB
R2
¨VFB(PP) = ¨VOUT(PP)
×
VREF
R1 + R2
If a low-ESR output capacitor is selected, then the
feedback voltage ripple may be too small to be sensed
by the gM amplifier and the error comparator. Also, if
the ESR of the output capacitor is very low, the output
voltage ripple and the feedback voltage ripple are not
necessarily in phase with the inductor current ripple. In
these cases, ripple injection is required to ensure
proper operation. Please refer to the Ripple Injection
subsection for more details about the ripple injection
technique.
HSD
TRIGGER ON-TIME IF VFB IS BELOW VREF
ESTIMATED ON-TIME
FIGURE 4-2:
Timing.
MIC28513 Control Loop
Figure 4-3 shows the operation of the MIC28513 during
a load transient. The output voltage drops due to the
sudden load increase, which causes the VFB to be less
than VREF. This will cause the error comparator to
trigger an ON-time period. At the end of the ON-time
period, a minimum OFF-time tOFF(MIN) is generated to
charge CBST because the feedback voltage is still
below VREF. Then, the next ON-time period is triggered
due to the low feedback voltage. Therefore, the
switching frequency changes during the load transient,
but returns to the nominal fixed frequency once the
output has stabilized at the new load current level. With
the varying duty cycle and switching frequency, the
output recovery time is fast and the output voltage
deviation is small in MIC28513 converter.
4.2
Discontinuous Mode (MIC28513-1
Only)
In continuous mode, the inductor current is always
greater than zero; however, at light loads the
MIC28513-1 is able to force the inductor current to
operate in discontinuous mode. Discontinuous mode
occurs when the inductor current falls to zero, as
indicated by trace (IL) shown in Figure 4-4. During this
period, the efficiency is optimized by shutting down all
the non-essential circuits and minimizing the supply
current. The MIC28513-1 wakes up and turns on the
high-side MOSFET when the feedback voltage VFB
drops below 0.8V.
FULL LOAD
IOUT
The MIC28513-1 has a zero crossing comparator that
monitors the inductor current by sensing the voltage
drop across the low-side MOSFET during its ON-time.
If the VFB > 0.8V and the inductor current goes slightly
negative, then the MIC28513-1 automatically powers
down most of the IC circuitry and goes into a low-power
mode.
NO LOAD
VOUT
Once the MIC28513-1 goes into discontinuous mode,
both DH and DL are low, which turns off the high-side
and low-side MOSFETs. The load current is supplied
by the output capacitors and VOUT drops. If the drop of
VOUT causes VFB to go below VREF, then all the circuits
will wake up into normal continuous mode. First, the
bias currents of most circuits reduced during the
discontinuous mode are restored, and then a tON pulse
is triggered before the drivers are turned on to avoid
any possible glitches. Finally, the high-side driver is
turned on. Figure 4-4 shows the control loop timing in
discontinuous mode.
VFB
VREF
HSD
TOFF(MIN)
FIGURE 4-3:
Response.
MIC28513 Load Transient
Unlike true current-mode control, the MIC28513 uses
the output voltage ripple to trigger an ON-time period.
The output voltage ripple is proportional to the inductor
DS20005522A-page 18
2016 Microchip Technology Inc.
MIC28513
4.5
Current Limit
IL CROSSES 0 AND VFB > 0.8.
DISCONTINUOUS MODE STARTS
The MIC28513 uses the RDS(ON) of the internal
low-side power MOSFET to sense overcurrent
conditions. In each switching cycle, the inductor current
is sensed by monitoring the low-side MOSFET during
its ON period. The sensed voltage, V(ILIM), is compared
with the power ground (PGND) after a blanking time of
150 ns.
V
< 0.8. WAKEUP FROM
IL
DFISB CONTINUOUS MODE.
0
VFB
The voltage drop of the resistor RILIM is compared with
the low-side MOSFET voltage drop to set the
overcurrent trip level. The small capacitor connected
from the ILIM pin to PGND can be added to filter the
switching node ringing, allowing a better short limit
measurement. The time constant created by RILIM and
the filter capacitor should be much less than the
minimum off time.
VREF
ZC
VHSD
The overcurrent limit can be programmed by using
Equation 4-3:
ESTIMATED ON-TIME
EQUATION 4-3:
VLSD
ICLIM – 0.5 ILPP RDSON + VCL
RILIM = ----------------------------------------------------------------------------------------------------
ICL
Where:
ICLIM
Desired Current Limit
FIGURE 4-4:
MIC28513-1 Control Loop
Timing (Discontinuous Mode).
∆IL(PP)
Inductor Current Peak-to-Peak
Use Equation 4-4 to calculate the
inductor ripple current
During discontinuous mode, the bias current of most
circuits are reduced. As a result, the total power supply
current during discontinuous mode is only about
450 μA, allowing the MIC28513-1 to achieve high
efficiency in light load applications.
RDS(ON)
VCL
On-Resistance of Low-Side MOSFET
Current-limit threshold.
14 mV (typical absolute value).
ICL
Current-limit source current.
80 µA (typical).
4.3
VDD Regulator
The MIC28513 provides a 5V regulated VDD to bias
internal circuitry for VIN ranging from 5.5V to 45V.
When VIN is less than 5.5V, VDD should be tied to VIN
pins to bypass the internal linear regulator.
The peak-to-peak inductor current ripple is calculated
with Equation 4-4.
EQUATION 4-4:
4.4
Soft-Start
VOUT VINMAX – VOUT
ILPP = -------------------------------------------------------------------
INMAX fSW L
V
Soft-start reduces the power supply inrush current at
startup by controlling the output voltage rise time while
the output capacitor charges.
The MOSFET RDS(ON) varies 30% to 40% with
temperature; therefore, it is recommended to use the
RDS(ON) at maximum junction temperature with a 20%
margin to calculate RILIM in Equation 4-3.
The MIC28513 implements an internal digital soft-start
by ramping up the 0.8V reference voltage (VREF) from
0 to 100% in about 5 ms with 9.7 mV steps. This
controls the output voltage rate of rise at turn on,
minimizing inrush current and eliminating output
voltage overshoot. Once the soft-start cycle ends, the
related circuitry is disabled to reduce current
consumption.
In case of hard short, the current-limit threshold is
folded down to allow an indefinite hard short on the
output without any destructive effect. It is mandatory to
make sure that the inductor current used to charge the
output capacitor during soft-start is under the folded
short limit; otherwise the supply will go into hiccup
mode and may not be finishing the soft-start
successfully.
2016 Microchip Technology Inc.
DS20005522A-page 19
MIC28513
4.6
Power Good (PGOOD)
The power good (PGOOD) pin is an open-drain output
that indicates logic-high when the output is nominally
90% of its steady-state voltage.
4.7
MOSFET Gate Drive
The Functional Block Diagram shows a bootstrap
circuit, consisting of DBST, CBST, and RBST. This circuit
supplies energy to the high-side drive circuit. Capacitor
CBST is charged, while the low-side MOSFET is on,
and the voltage on the SW pin is approximately 0V.
When the high-side MOSFET driver is turned on,
energy from CBST is used to turn the MOSFET on. As
the high-side MOSFET turns on, the voltage on the SW
pin increases to approximately VIN. Diode DBST is
reverse-biased and CBST floats high while continuing to
bias the high-side gate driver. The bias current of the
high-side driver is less than 10 mA, so a 0.1 μF to 1 μF
capacitor is sufficient to hold the gate voltage with
minimal droop for the power stroke (high-side
switching) cycle, i.e. ∆BST = 10 mA x 1.25 μs/0.1 μF =
125 mV. When the low-side MOSFET is turned back
on, CBST is then recharged through the boost diode. A
30ꢀ resistor RBST, which is in series with the BST pin,
is required to slow down the turn-on time of the
high-side N-channel MOSFET.
DS20005522A-page 20
2016 Microchip Technology Inc.
MIC28513
5.2
Setting the Switching Frequency
5.0
5.1
APPLICATION INFORMATION
The MIC28513 switching frequency can be adjusted by
changing the resistor divider network from VIN.
Output Voltage Setting
Components
The MIC28513 requires two resistors to set the output
voltage as shown in Figure 5-1.
R1
FB
gM AMP
R2
VREF
FIGURE 5-2:
Switching Frequency
Adjustment.
FIGURE 5-1:
Configuration.
Voltage Divider
Equation 5-3 gives the estimated switching frequency.
The output voltage is determined by Equation 5-1.
EQUATION 5-3:
EQUATION 5-1:
R17
R17 + R19
-------------------------
fSW = f0
R1
R2
VOUT = VFB 1 + ------
Where:
f0
Where:
VFB
Switching frequency when R17 is open;
typically 600 kHz.
0.8V
Figure 5-3 shows the switching frequency versus the
resistor R17 when R19 = 100 kꢀ.
A typical value of R1 used on the standard evaluation
board is 10 kꢀ. If R1 is too large, it may allow noise to
be introduced into the voltage feedback loop. If R1 is
too small in value, it will decrease the efficiency of the
power supply, especially at light loads. Once R1 is
selected, R2 can be calculated using Equation 5-2:
EQUATION 5-2:
VFB R1
R2 = -----------------------------
VOUT – VFB
FIGURE 5-3:
Switching Frequency vs.
R17.
5.3
Inductor Selection
Values for inductance, peak, and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine
the peak-to-peak inductor ripple current. Generally,
higher inductance values are used with higher input
voltages. Larger peak-to-peak ripple currents will
increase the power dissipation in the inductor and
2016 Microchip Technology Inc.
DS20005522A-page 21
MIC28513
MOSFETs. Larger output ripple currents will also
require more output capacitance to smooth out the
larger ripple current. Smaller peak-to-peak ripple
EQUATION 5-7:
PLCU = ILRMS2 DCR
currents require
a larger inductance value and
therefore a larger and more expensive inductor. A good
compromise between size, loss and cost is to set the
inductor ripple current to be equal to 20% of the
maximum output current. The inductance value is
calculated by:
The resistance of the copper wire, DCR, increases with
the temperature. The value of the winding resistance
used should be at the operating temperature.
EQUATION 5-8:
EQUATION 5-4:
DCRHT = DCR20C 1 + 0.0042 TH – T20C
VOUT VINMAX – VOUT
L = -------------------------------------------------------------------
V
INMAX ILPP fSW
Where:
Where:
fSW
∆IL(PP)
TH
Temperature of wire under full load
Ambient temperature
T20C
Switching Frequency
DCR(20C)
Room temperature winding resistance
(usually specified by the manufacturer)
The peak-to-peak inductor current
ripple; typically 20% of the maximum
output current
5.4
Output Capacitor Selection
In continuous conduction mode, the peak inductor
current is equal to the average output current plus one
half of the peak-to-peak inductor current ripple.
The type of the output capacitor is usually determined
by its equivalent series resistance (ESR). Voltage and
RMS current capability are also important factors in
selecting an output capacitor. Recommended capacitor
types are ceramic, tantalum, low-ESR aluminum
electrolytic, OS-CON and POSCAP. For high ESR
electrolytic capacitors, ESR is the main cause of the
output ripple. The output capacitor ESR also affects the
control loop from a stability point of view. For a low ESR
ceramic output capacitor, ripple is dominated by the
reactive impedance. The maximum value of ESR is
calculated by Equation 5-9.
EQUATION 5-5:
ILPK = IOUT + 0.5 ILPP
The RMS inductor current is used to calculate the I2R
losses in the inductor.
EQUATION 5-6:
2
ILPP
2
EQUATION 5-9:
ILRMS
=
IOUTMAX + --------------------
I2
VOUTPP
---------------------------
ESRC
OUT
ILPP
Maximizing efficiency requires the proper selection of
core material and minimizing the winding resistance.
The high frequency operation of the MIC28513
requires the use of ferrite materials for all but the most
cost sensitive applications. Lower cost iron powder
cores may be used but the increase in core loss will
reduce the efficiency of the power supply. This is
especially noticeable at low output power. The winding
resistance decreases efficiency at the higher output
current levels.
Where:
∆VOUT(PP) Peak-to-Peak Output Voltage Ripple
∆IL(PP) Peak-to-Peak Inductor Current Ripple
The total output ripple is a combination of the ESR and
output capacitance. The total ripple is calculated by
Equation 5-10.
EQUATION 5-10:
The winding resistance must be minimized although
this usually comes at the expense of a larger inductor.
The power dissipated in the inductor is equal to the sum
of the core and copper losses. At higher output loads,
the core losses are usually insignificant and can be
ignored. At lower output currents, the core losses can
be a significant contributor. Core loss information is
usually available from the magnetics vendor. Copper
loss in the inductor is calculated by Equation 5-7:
VOUTPP
=
2 ILPP
+ ILPP ESRCOUT2
-------------------------------------
COUT fSW 8
Where:
D
Duty Cycle
COUT
fSW
Output Capacitance Value
Switching Frequency
DS20005522A-page 22
2016 Microchip Technology Inc.
MIC28513
As described in the Theory of Operation subsection of
the Functional Description, the MIC28513 requires at
least 20 mV peak-to-peak ripple at the FB pin for the gM
amplifier and the error comparator to operate properly.
Also, the ripple on FB pin should be in phase with the
inductor current. Therefore, the output voltage ripple
caused by the output capacitors value should be much
smaller than the ripple caused by the output capacitor
ESR. If low-ESR capacitors, such as ceramic
capacitors, are selected as the output capacitors, a
ripple injection method should be applied to provide the
enough feedback voltage ripple. Refer to the Ripple
Injection subsection for details.
EQUATION 5-14:
CINRMS IOUTMAX D 1 – D
I
The power dissipated in the input capacitor is:
EQUATION 5-15:
PDISSCIN = ICINRMS2 ESRCIN
5.6
Ripple Injection
The voltage rating of the capacitor should be twice the
output voltage for a tantalum and 20% greater for
aluminum electrolytic or OS-CON. The output capacitor
RMS current is calculated in Equation 5-11.
The VFB ripple required for proper operation of the
MIC28513’s gM amplifier and error comparator is
20 mV to 100 mV. However, the output voltage ripple is
generally designed as 1% to 2% of the output voltage.
If the feedback voltage ripple is so small that the gM
amplifier and error comparator can’t sense it, then the
MIC28513 will lose control and the output voltage is not
regulated. In order to have some amount of VFB ripple,
a ripple injection method is applied for low output
voltage ripple applications.
EQUATION 5-11:
ILPP
IC
= ------------------
12
OUTRMS
The power dissipated in the output capacitor is:
EQUATION 5-12:
The applications are divided into three situations
according to the amount of the feedback voltage ripple:
• Enough ripple at the feedback voltage due to the
large ESR of the output capacitors (Figure 5-4).
The converter is stable without any ripple
injection.
PDISSCOUT = ICOUTRMS2 ESRCOUT
5.5
Input Capacitor Selection
L
SW
The input capacitor for the power stage input VIN
should be selected for ripple current rating and voltage
rating. Tantalum input capacitors may fail when
subjected to high inrush currents, caused by turning the
input supply on. A tantalum input capacitor’s voltage
rating should be at least two times the maximum input
voltage to maximize reliability. Aluminum electrolytic,
OS-CON, and multilayer polymer film capacitors can
handle the higher inrush currents without voltage
de-rating. The input voltage ripple will primarily depend
on the input capacitor’s ESR. The peak input current is
equal to the peak inductor current, so:
COUT
R1
MIC28513
FB
R2
ESR
FIGURE 5-4:
Enough Ripple at FB.
The feedback voltage ripple is:
EQUATION 5-16:
R2
R1 + R2
-------------------
VFBPP
Where:
∆IL(PP)
=
ESR
ILPP
COUT
EQUATION 5-13:
VIN = ILPK ESRCIN
Peak-to-Peak Value of the Inductor
Current Ripple
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming
the peak-to-peak inductor current ripple is low:
• Inadequate ripple at the feedback voltage due to
the small ESR of the output capacitors.
The output voltage ripple is fed into the FB pin
through a feed-forward capacitor, CFF in this
situation, as shown in Figure 5-5. The typical CFF
value is selected by using Equation 5-17.
2016 Microchip Technology Inc.
DS20005522A-page 23
MIC28513
In Equation 5-19 and Equation 5-20, it is assumed that
the time constant associated with CFF must be much
greater than the switching period:
EQUATION 5-17:
10
fSW
-------
R1 CFF
EQUATION 5-21:
With the feed-forward capacitor, the feedback
voltage ripple is very close to the output voltage
ripple.
1
T
---------------- = -- « 1
fSW
If the voltage divider resistors R1 and R2 are in the kꢀ
range, a CFF of 1 nF to 100 nF can easily satisfy the
large time constant requirements. Also, a 100 nF
injection capacitor CINJ is used in order to be
EQUATION 5-18:
VFBPP ESR ILPP
considered as short for
frequencies.
a wide range of the
L
SW
The process of sizing the ripple injection resistor and
capacitors is as follows.
COUT
MIC28513
R1
R2
FB
CFF
• Select CFF to feed all output ripples into the
feedback pin and make sure the large time
constant assumption is satisfied. Typical choice of
CFF is 1 nF to 100 nF if R1 and R2 are in the kꢀ
range.
ESR
FIGURE 5-5:
Inadequate Ripple at FB.
• Select RINJ according to the expected feedback
voltage ripple using Equation 5-22:
• Virtually no ripple at the FB pin voltage due to the
very low ESR of the output capacitors.
In this situation, the output voltage ripple is less than
20 mV. Therefore, additional ripple is injected into
the FB pin from the switching node SW via a resistor
RINJ and a capacitor CINJ, as shown in Figure 5-6.
EQUATION 5-22:
VFBPP
fSW
D 1 – D
----------------------- ----------------------------
Kdiv
=
VIN
L
The value of RINJ is calculated using Equation 5-23.
SW
CINJ
COUT
ESR
MIC28513
R1
R2
RINJ
CFF
EQUATION 5-23:
FB
1
Kdiv
RINJ = R1//R2 ---------- – 1
• Select CINJ as 100 nF, which could be considered
as short for a wide range of the frequencies.
FIGURE 5-6:
Invisible Ripple at FB.
The injected ripple is calculated via:
EQUATION 5-19:
VFBPP = VIN Kdiv D 1 – D
Where:
1
----------------
fSW
VIN
D
Power stage input voltage
Duty cycle
fSW
τ
Switching frequency
(R1//R2//RINJ) x CFF
EQUATION 5-20:
R1//R2
Kdiv = ----------------------------------
RINJ + R1//R2
DS20005522A-page 24
2016 Microchip Technology Inc.
MIC28513
operating voltage must be derated by 50%.
6.0
PCB LAYOUT GUIDELINES
• In “Hot-Plug” applications, a Tantalum or
Electrolytic bypass capacitor must be used to limit
the overvoltage spike seen on the input supply
with power is suddenly applied.
PCB Layout is critical to achieve reliable, stable and
efficient performance. A ground plane is required to
control EMI and minimize the inductance in power,
signal and return paths.
Figure 6-1 is optimized from a small form-factor point of
view and shows the top and bottom layers of a
four-layer PCB. It is recommended to use Mid-Layer 1
as a continuous ground plane.
6.3
SW Node
• Do not route any digital lines underneath or close
to the SW node.
• Keep the switch node (SW) away from the
feedback (FB) pin.
6.4
Output Capacitor
• Use a copper island to connect the output
capacitor ground terminal to the input capacitor
ground terminal.
• Phase margin will change as the output capacitor
value and ESR changes. Contact the factory if the
output capacitor is different from what is shown in
the BOM.
• The feedback trace should be separate from the
power trace and connected as close as possible
to the output capacitor. Sensing a long
high-current load trace can degrade the DC load
regulation.
FIGURE 6-1:
Four-Layer Board.
Top and Bottom Layers of a
The following guidelines should be followed to ensure
proper operation of the MIC28513 converter.
6.5
Thermal Measurements
Measuring the IC’s case temperature is recommended
to ensure it is within its operating limits. Although this
might seem like a very elementary task, it is easy to get
erroneous results. The most common mistake is to use
the standard thermal couple that comes with a thermal
meter. This thermal couple wire gauge is large, typically
22 gauge, and behaves like a heatsink, resulting in a
lower case measurement.
6.1
IC
• The analog ground pin (AGND) must be
connected directly to the ground planes. Do not
route the AGND pin to the PGND pin on the top
layer.
• Place the IC close to the point-of-load (POL).
• Use copper planes to route the input and output
power lines.
Two methods of temperature measurement are using a
smaller thermal couple wire or an infrared
thermometer. If a thermal couple wire is used, it must
be constructed of 36 gauge wire or higher then (smaller
wire size) to minimize the wire heat-sinking effect. In
addition, the thermal couple tip must be covered in
either thermal grease or thermal glue to make sure that
the thermal couple junction is making good contact with
the case of the IC. Omega brand thermal couple
(5SC-TT-K-36-36) is adequate for most applications.
• Analog and power grounds should be kept
separate and connected at only one location.
6.2
Input Capacitor
• Place the input capacitors on the same side of the
board and as close to the PVIN and PGND pins as
possible.
• Place several vias to the ground plane close to
the input capacitor ground terminal.
Wherever possible, an infrared thermometer is
recommended. The measurement spot size of most
infrared thermometers is too large for an accurate
reading on a small form factor ICs. However, a IR
thermometer from Optris has a 1 mm spot size, which
makes it a good choice for measuring the hottest point
on the case. An optional stand makes it easy to hold the
beam on the IC for long periods of time.
• Use either X7R or X5R dielectric input capacitors.
Do not use Y5V or Z5U type capacitors.
• Do not replace the ceramic input capacitor with
any other type of capacitor. Any type of capacitor
can be placed in parallel with the input capacitor.
• If a Tantalum input capacitor is placed in parallel
with the input capacitor, it must be recommended
for switching regulator applications and the
For more information about the Evaluation board layout, please contact Microchip sales.
2016 Microchip Technology Inc.
DS20005522A-page 25
MIC28513
7.0
PACKAGING INFORMATION
24-Lead FQFN 3 mm x 4 mm Package Outline and Recommended Land Pattern
Note:
For the most current package drawings, please see the Microchip Packaging Specification located at
http://www.microchip.com/packaging
DS20005522A-page 26
2016 Microchip Technology Inc.
MIC28513
Note:
For the most current package drawings, please see the Microchip Packaging Specification located at
http://www.microchip.com/packaging
2016 Microchip Technology Inc.
DS20005522A-page 27
MIC28513
NOTES:
DS20005522A-page 28
2016 Microchip Technology Inc.
MIC28513
APPENDIX A: REVISION HISTORY
Revision A (May 2016)
• Converted Micrel document MIC28513 to Micro-
chip data sheet template DS20005522A.
• Minor text changes throughout.
2015 Microchip Technology Inc.
DS20005522A-page 29
MIC28513
NOTES:
DS20005522A-page 30
2015 Microchip Technology Inc.
MIC28513
PRODUCT IDENTIFICATION SYSTEM
To order or obtain information, e.g., on pricing or delivery, contact your local Microchip representative or sales office.
Examples:
XX
PART NO.
Device
X
X
a)
MIC28513-1YFL:
45V, 4A Synchronous Buck
Regulator, HyperLight
Load, –40°C to +125°C
Junction Temperature
Range, 24LD FQFN
45V, 4A Synchronous Buck
Regulator, Hyper Speed
Control, –40°C to +125°C
Junction
Package
Temperature
Architecture
Device:
MIC28513:
45V, 4A Synchronous Buck Regulator
b)
MIC28513-2YFL:
Architecture:
1
2
=
=
HyperLight Load
Hyper Speed Control
Temperature Range,
24LD FQFN
Temperature:
Package:
Y
=
=
–40°C to +125°C
Note 1: FQFN is a lead-free package. Pb-Free lead
finish is Matte Tin.
FL
24-Pin 3 mm x 4 mm FQFN; Note 1
2015 Microchip Technology Inc.
DS20005522A-page 31
MIC28513
NOTES:
DS20005522A-page 32
2015 Microchip Technology Inc.
Note the following details of the code protection feature on Microchip devices:
•
Microchip products meet the specification contained in their particular Microchip Data Sheet.
•
Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the
intended manner and under normal conditions.
•
There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our
knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip’s Data
Sheets. Most likely, the person doing so is engaged in theft of intellectual property.
•
•
Microchip is willing to work with the customer who is concerned about the integrity of their code.
Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not
mean that we are guaranteeing the product as “unbreakable.”
Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of our
products. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such acts
allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act.
Information contained in this publication regarding device
applications and the like is provided only for your convenience
and may be superseded by updates. It is your responsibility to
ensure that your application meets with your specifications.
MICROCHIP MAKES NO REPRESENTATIONS OR
WARRANTIES OF ANY KIND WHETHER EXPRESS OR
IMPLIED, WRITTEN OR ORAL, STATUTORY OR
OTHERWISE, RELATED TO THE INFORMATION,
INCLUDING BUT NOT LIMITED TO ITS CONDITION,
QUALITY, PERFORMANCE, MERCHANTABILITY OR
FITNESS FOR PURPOSE. Microchip disclaims all liability
arising from this information and its use. Use of Microchip
devices in life support and/or safety applications is entirely at
the buyer’s risk, and the buyer agrees to defend, indemnify and
hold harmless Microchip from any and all damages, claims,
suits, or expenses resulting from such use. No licenses are
conveyed, implicitly or otherwise, under any Microchip
intellectual property rights unless otherwise stated.
Trademarks
The Microchip name and logo, the Microchip logo, AnyRate,
dsPIC, FlashFlex, flexPWR, Heldo, JukeBlox, KeeLoq,
KeeLoq logo, Kleer, LANCheck, LINK MD, MediaLB, MOST,
MOST logo, MPLAB, OptoLyzer, PIC, PICSTART, PIC32 logo,
RightTouch, SpyNIC, SST, SST Logo, SuperFlash and UNI/O
are registered trademarks of Microchip Technology
Incorporated in the U.S.A. and other countries.
ClockWorks, The Embedded Control Solutions Company,
ETHERSYNCH, Hyper Speed Control, HyperLight Load,
IntelliMOS, mTouch, Precision Edge, and QUIET-WIRE are
registered trademarks of Microchip Technology Incorporated
in the U.S.A.
Analog-for-the-Digital Age, Any Capacitor, AnyIn, AnyOut,
BodyCom, chipKIT, chipKIT logo, CodeGuard, dsPICDEM,
dsPICDEM.net, Dynamic Average Matching, DAM, ECAN,
EtherGREEN, In-Circuit Serial Programming, ICSP, Inter-Chip
Connectivity, JitterBlocker, KleerNet, KleerNet logo, MiWi,
motorBench, MPASM, MPF, MPLAB Certified logo, MPLIB,
MPLINK, MultiTRAK, NetDetach, Omniscient Code
Generation, PICDEM, PICDEM.net, PICkit, PICtail,
PureSilicon, RightTouch logo, REAL ICE, Ripple Blocker,
Serial Quad I/O, SQI, SuperSwitcher, SuperSwitcher II, Total
Endurance, TSHARC, USBCheck, VariSense, ViewSpan,
WiperLock, Wireless DNA, and ZENA are trademarks of
Microchip Technology Incorporated in the U.S.A. and other
countries.
SQTP is a service mark of Microchip Technology Incorporated
in the U.S.A.
Microchip received ISO/TS-16949:2009 certification for its worldwide
headquarters, design and wafer fabrication facilities in Chandler and
Tempe, Arizona; Gresham, Oregon and design centers in California
and India. The Company’s quality system processes and procedures
are for its PIC® MCUs and dsPIC® DSCs, KEELOQ® code hopping
devices, Serial EEPROMs, microperipherals, nonvolatile memory and
analog products. In addition, Microchip’s quality system for the design
and manufacture of development systems is ISO 9001:2000 certified.
Silicon Storage Technology is a registered trademark of
Microchip Technology Inc. in other countries.
GestIC is a registered trademarks of Microchip Technology
Germany II GmbH & Co. KG, a subsidiary of Microchip
Technology Inc., in other countries.
All other trademarks mentioned herein are property of their
respective companies.
QUALITYꢀMANAGEMENTꢀꢀSYSTEMꢀ
CERTIFIEDꢀBYꢀDNVꢀ
© 2016, Microchip Technology Incorporated, Printed in the
U.S.A., All Rights Reserved.
ISBN: 978-1-5224-0542-9
== ISO/TSꢀ16949ꢀ==ꢀ
2016 Microchip Technology Inc.
DS20005522A-page 33
Worldwide Sales and Service
AMERICAS
ASIA/PACIFIC
ASIA/PACIFIC
EUROPE
Corporate Office
2355 West Chandler Blvd.
Chandler, AZ 85224-6199
Tel: 480-792-7200
Fax: 480-792-7277
Technical Support:
http://www.microchip.com/
support
Asia Pacific Office
China - Xiamen
Tel: 86-592-2388138
Fax: 86-592-2388130
Austria - Wels
Tel: 43-7242-2244-39
Fax: 43-7242-2244-393
Suites 3707-14, 37th Floor
Tower 6, The Gateway
Harbour City, Kowloon
China - Zhuhai
Tel: 86-756-3210040
Fax: 86-756-3210049
Denmark - Copenhagen
Tel: 45-4450-2828
Fax: 45-4485-2829
Hong Kong
Tel: 852-2943-5100
Fax: 852-2401-3431
India - Bangalore
Tel: 91-80-3090-4444
Fax: 91-80-3090-4123
France - Paris
Tel: 33-1-69-53-63-20
Fax: 33-1-69-30-90-79
Australia - Sydney
Tel: 61-2-9868-6733
Fax: 61-2-9868-6755
Web Address:
www.microchip.com
India - New Delhi
Tel: 91-11-4160-8631
Fax: 91-11-4160-8632
Germany - Dusseldorf
Tel: 49-2129-3766400
Atlanta
Duluth, GA
Tel: 678-957-9614
Fax: 678-957-1455
China - Beijing
Tel: 86-10-8569-7000
Fax: 86-10-8528-2104
Germany - Karlsruhe
Tel: 49-721-625370
India - Pune
Tel: 91-20-3019-1500
China - Chengdu
Tel: 86-28-8665-5511
Fax: 86-28-8665-7889
Germany - Munich
Tel: 49-89-627-144-0
Fax: 49-89-627-144-44
Austin, TX
Tel: 512-257-3370
Japan - Osaka
Tel: 81-6-6152-7160
Fax: 81-6-6152-9310
Boston
China - Chongqing
Tel: 86-23-8980-9588
Fax: 86-23-8980-9500
Italy - Milan
Tel: 39-0331-742611
Fax: 39-0331-466781
Westborough, MA
Tel: 774-760-0087
Fax: 774-760-0088
Japan - Tokyo
Tel: 81-3-6880- 3770
Fax: 81-3-6880-3771
China - Dongguan
Tel: 86-769-8702-9880
Italy - Venice
Tel: 39-049-7625286
Chicago
Itasca, IL
Tel: 630-285-0071
Fax: 630-285-0075
Korea - Daegu
Tel: 82-53-744-4301
Fax: 82-53-744-4302
China - Hangzhou
Tel: 86-571-8792-8115
Fax: 86-571-8792-8116
Netherlands - Drunen
Tel: 31-416-690399
Fax: 31-416-690340
Korea - Seoul
Cleveland
Tel: 82-2-554-7200
Fax: 82-2-558-5932 or
82-2-558-5934
China - Hong Kong SAR
Tel: 852-2943-5100
Fax: 852-2401-3431
Poland - Warsaw
Tel: 48-22-3325737
Independence, OH
Tel: 216-447-0464
Fax: 216-447-0643
Spain - Madrid
Tel: 34-91-708-08-90
Fax: 34-91-708-08-91
China - Nanjing
Tel: 86-25-8473-2460
Fax: 86-25-8473-2470
Malaysia - Kuala Lumpur
Tel: 60-3-6201-9857
Fax: 60-3-6201-9859
Dallas
Addison, TX
Tel: 972-818-7423
Fax: 972-818-2924
Sweden - Stockholm
Tel: 46-8-5090-4654
China - Qingdao
Tel: 86-532-8502-7355
Fax: 86-532-8502-7205
Malaysia - Penang
Tel: 60-4-227-8870
Fax: 60-4-227-4068
Detroit
Novi, MI
UK - Wokingham
Tel: 44-118-921-5800
China - Shanghai
Tel: 86-21-5407-5533
Fax: 86-21-5407-5066
Philippines - Manila
Tel: 63-2-634-9065
Fax: 63-2-634-9069
Tel: 248-848-4000
Fax: 44-118-921-5820
Houston, TX
Tel: 281-894-5983
China - Shenyang
Tel: 86-24-2334-2829
Fax: 86-24-2334-2393
Singapore
Tel: 65-6334-8870
Fax: 65-6334-8850
Indianapolis
Noblesville, IN
Tel: 317-773-8323
Fax: 317-773-5453
China - Shenzhen
Tel: 86-755-8864-2200
Fax: 86-755-8203-1760
Taiwan - Hsin Chu
Tel: 886-3-5778-366
Fax: 886-3-5770-955
Los Angeles
Mission Viejo, CA
Tel: 949-462-9523
Fax: 949-462-9608
China - Wuhan
Tel: 86-27-5980-5300
Fax: 86-27-5980-5118
Taiwan - Kaohsiung
Tel: 886-7-213-7828
Taiwan - Taipei
Tel: 886-2-2508-8600
Fax: 886-2-2508-0102
New York, NY
Tel: 631-435-6000
China - Xian
Tel: 86-29-8833-7252
Fax: 86-29-8833-7256
San Jose, CA
Tel: 408-735-9110
Thailand - Bangkok
Tel: 66-2-694-1351
Fax: 66-2-694-1350
Canada - Toronto
Tel: 905-673-0699
Fax: 905-673-6509
07/14/15
DS20005522A-page 34
2015 Microchip Technology Inc.
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