MIC3223YTSE-TR_技术文档

MIC3223

High Power Boost LED Driver with

Integrated FET

General Description

Features

The MIC3223 is a constant current boost LED driver

capable of driving a series string of high power LEDs. The

MIC3223 can be used in general lighting, bulb replacement,

garden pathway lighting and other solid state illumination

applications.

• 4.5V to 20V supply voltage

• 200mV feedback voltage with an accuracy of ±5%

• Step-up output voltage (boost) conversion up to 37V

• 1MHz switching frequency

• 100mΩ/3.5A internal power FET switch

• LEDs can be dimmed using a PWM signal

• User settable LED current (through external resistor)

• Externally programmable soft-start

The MIC3223 is a peak current mode control PWM boost

regulator and the 4.5V and 20V operating input voltage

range allows multiple applications from a 5V or a 12V bus.

The MIC3223 implements a fixed internal 1MHz switching

frequency to allow for a reduction in the design footprint

size. Power consumption has been minimized through the

implementation of a 200mV feedback voltage that provides

an accuracy of ±5%. The MIC3223 can be dimmed through

the use of a PWM signal and features an enable pin for a

low power shutdown state.

• Protection features that include:

–

Output over-voltage protection (OVP)

Under-voltage lockout (UVLO)

Over temperature protection

• Junction temperature range: -40°C to +125°C

• Available in a exposed pad 16-pin TSSOP package

The MIC3223 is a very robust LED driver and offers the

following protection features: over voltage protection (OVP),

thermal shutdown, switch current limiting and under voltage

lockout (UVLO).

Applications

• Architectural lighting

• Industrial lighting

• Signage

The MIC3223 is offered in a low profile exposed pad 16-pin

TSSOP package.

• Landscape lighting (garden/pathway)

• Under cabinet lighting

• MR-16 bulbs

Data sheets and support documentation can be found on

Micrel’s web site at: www.micrel.com.

_______________________________________________________________________________________________________________________

Typical Application

Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com

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MIC3223

Ordering Information

Part Number

Junction Temp. Range

Package

Lead Finish

MIC3223YTSE

–40° to +125°C

16-pin ePad TSSOP

PB- free

Pin Configuration

16-Pin ePad TSSOP (TSE)

Pin Description

Pin Number

Pin Name

Pin Function

Enable (Input): Logic high enables and logic low disables operation.

1

2

EN

SS

Soft Start (Input resistance of 30k). Connect a capacitor to GND for soft-start. Clamp the

pin to a known voltage to control the internal reference voltage and hence the output

current.

3

4

5

COMP

FB

Compensation Pin (Input): Add external R and C-to-GND to stabilize the converter.

Negative Input to Error Amp

OVP

Connect to the centre tap of an external resistor divider, the top of which is tied to Vout

and bottom-to-ground.

6

7,8,9,10

11

PGND

SW

Power Ground

Switch Node (Input): Internal NMOS switch Drain Pin

Input Supply

VIN

12

DRVVDD

For 4.5V < VIN < 6V, connect DRVVDD to VIN. DRVVDD is the input voltage supply for

the converter’s internal power FET gate driver. For VIN > 6V, connect this pin to VDD.

13

VDD

For 4.5V < VIN < 6V, this pin becomes the input voltage supply for the converter’s internal

circuit. For VIN > 6V, this pin is an output of the internal 5.5V regulator that supplies

internal circuits. User must add 10µF decoupling capacitor from VDD-to-AGND.

14

15

16

17

DIM_IN

DIM_OUT

AGND

EP

PWM input to control LED dimming.

Output driver to drive external FET for LED dimming.

Analog Ground

Connect to Power Ground

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MIC3223

Absolute Maximum Ratings⁽¹⁾

Operating Ratings⁽²⁾

Supply Voltage (V_IN).....................................................+22V

Switch Voltage (V_SW)..................................... -0.3V to +42V

Regulated Voltage (V_DD) ...............................-0.3V to +6.5V

Dimming In Voltage (V_{DIM_IN})...............-0.3V to (V_DD+ 0.3V)

Dimming Out Voltage (V_{DIM_OUT})..........-0.3V to (V_DD+ 0.3V)

Soft-Start Voltage (V_SS).......................-0.3V to (V_DD+ 0.3V)

Enable Voltage (V_EN)............................-0.3V to (V_IN+ 0.3V)

Feedback Voltage (V_FB) ......................-0.3V to (V_DD+ 0.3V)

Switch Current (I_SW) ..................................Internally Limited

Comp Voltage (V_COMP).......................-0.3V to (+V_DD+ 0.3V)

FET Driver Supply (V_DRVVDD).........................-0.3V to +6.5V

PGND to AGND ............................................-0.3V to +0.3V

Over Voltage Protection (V_OVP) ...........-0.3V to (V_DD+ 0.3V)

Peak Reflow Temperature (soldering, 10-20sec.) ..... 260°C

Storage Temperature (T_S)..........................-65°C to +150°C

ESD Rating⁽³⁾................................................................+2kV

Supply Voltage (V_IN)...................................... +4.5V to +20V

Switch Voltage (V_SW)....................................................+37V

Junction Temperature (T_J) .........................-40°C to +125°C

Junction Thermal Resistance

ePad TSSOP-16L (θ_JA)...................................36.5°C/W

Electrical Characteristics⁽⁴⁾

V_IN= V_EN= 12V; L = 22µH, C_IN=4.7µF, C_OUT=2x4.7µF; T_A= 25°C, BOLD values indicate –40°C≤ T_J≤ +125°C, unless otherwise noted.

Symbol

V_IN

Parameter

Voltage Supply Range

Under Voltage Lockout

Over Voltage Protection

Quiescent Current

Condition

Min

4.5

Typ

Max

20

Units

V

V_UVLO

V_OVP

I_VIN

Monitoring for V_DD

3

3.7

1.28

2.1

4.4

V

1.216

1.344

5

V

V_FB=250mV

V_EN=0V

mA

µA

mV

nA

V

I_SD

Shutdown Current

10

Room Temperature

Over Temperature

V_FB=200mV

190

200

210

216

V_FB

Feedback Voltage

184

I_FB

Feedback Input Current

Internal Voltage Regulator

Maximum Duty Cycle

V_DDLine Regulation

-450

5.3

90

V_DD

D_MAX

85

95

10.5

10

%

V_LED=18V, V_IN=8V to 16V, I_LED=350mA

0.5

9

%

I_SW

Switch Current Limit

3.5

A

R_SW

I_SW

Switch R_DSONplus R_CS

Switch Leakage Current

100

0.01

mꢀ

µA

V

V_EN=0, V_SW=37V

Turn On

1.5

V_EN

Enable Threshold

Turn Off

0.4

40

V

I_EN

Enable Pin Current

20

µA

V

V_{DIM_TH_H}DIM_IN Threshold High

Logic High

Logic Low

V_{DIM_TH_L}

Hys

DIM_IN Threshold Low

DIM_IN Hysteresis

DIM_IN Pin Current

Dim Delay (Rising)

Dim Delay (Falling

0.4

1

V

500

mV

µA

ns

I_{DIM_IN}

T_DR

V_{DIM_IN}= 5V

DIM_IN Rising

DIM_IN Falling

40

30

T_DF

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Symbol

MIC3223

Parameter

Condition

Min

Typ

Max

Units

DIM_IN =1µs C_{DIM_OUT}= 1.25nF

0.7

1.3

µs

DIM MIN

Minimum Dimming Pulse

DIM_OUT measured from 4V rising to 2.5

falling

0.5

1.5

µs

DIM_OUT pull up resistance

I_{DIM_OUT}= +2mA

R_DO

DIM_OUT Resistance High

DIM_OUT Resistance Low

70

40

ꢀ

Dim Out pull down resistance

I_{DIM_OUT}= -2mA

ꢀ

F_SW

R_SS

Oscillator Frequency

Soft Start Resistance

0.7

30

1

1.3

62

MHz

kꢀ

46

Temperature rising

Hysteresis

165

10

°C

Over Temperature Threshold

Shutdown

T_SD

°C

Notes

1. Exceeding the absolute maximum rating may damage the device.

2. The device is not guaranteed to function outside its operating rating.

3. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5kꢀ in series with 100pF.

4. Specification for packaged product only.

Test Circuit

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Typical Characteristics

Efficiency

VDD Voltage

Current Limit

vs. Input Voltage

98

5.50

5.45

5.40

5.35

5.30

5.25

5.20

5.15

5.10

5.05

5.00

9.5

9.0

8.5

8.0

7.5

7.0

T = 25°C

96

94

92

90

88

86

84

T = 25°C

V_OUT= 25V

I_OUT= 0.5A

V_OUT= 25V

I_OUT= 0.5A

82

80

V_IN= 4.5V to 6V

14 19

4

9

5

10

15

20

5

10

15

20

INPUT VOLTAGE (V)

Feedback Voltage

vs. Input Voltage

Switching Frequency

vs. Input Voltage

Feedback Voltage

vs. Temperature

0.210

0.208

0.206

0.204

0.202

0.200

0.198

0.196

0.194

0.192

0.190

1.2

0.220

0.218

0.216

0.214

0.212

0.210

0.208

0.206

0.204

0.202

0.200

T = 25°C

1.1

1.0

0.9

V_DD= V_IN

V_IN= 12V

V_IN= 4.5V to 6V

V_OUT= 30V

I_OUT= 0.36A

V_OUT= 26V

V_OUT= 30V

I_OUT=0.36A

I_OUT= 0.36A

-40 -20

0

20 40 60 80 100 120

4

9

14

19

4

-40 -20

0

9

14

19

TEMPERATURE (°C)

INPUT VOLTAGE (V)

Current Limit

vs. Temperature

R_{SW _NODE}vs.

Temperature

Switching Frequency

vs. Temperature

0.18

11.0

10.5

10.0

9.5

1.20

1.15

1.10

1.05

1.00

0.95

0.90

0.85

0.80

0.17

0.16

0.15

0.14

0.13

0.12

0.11

0.10

9.0

8.5

8.0

7.5

V_IN= 12V

7.0

V_OUT= 36V

I_SW= 1.3A

V_OUT= 26V

I_OUT= 0.36A

6.5

V_IN= 12V

6.0

-40 -20

0

20

40

60

80 100 120

0

20

40

60

80 100 120

-40 -20

0

20

40

60

80 100 120

TEMPERATURE (°C)

Efficiency

vs. Output Current

Efficiency

vs. Output Current

Efficiency

vs. Output Current

98

96

94

92

90

88

86

84

82

80

96

94

92

90

88

86

84

82

80

20V

16V

12V

96

94

92

90

88

86

84

82

80

14V

18V

8V

10V

V_OUT= 25V

0.5

1

1.5

0

0.5

1

1.5

0

0.5

1

1.5

OUTPUT CURRENT (A)

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MIC3223

Typical Characteristics (continued)

Efficiency

vs. Output Current

96

94

92

90

88

86

84

82

80

94

92

10V

12V

90

88

86

84

82

V_OUT= 25V

80

0

0.5

1

1.5

0

0.5

1

1.5

OUTPUT CURRENT (A)

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MIC3223

Functional Characteristics

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Functional Characteristics (continued)

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MIC3223

Functional Diagram

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MIC3223

current is regulated. If V_FBdrops, V_EAincreases and

therefore the power FET remains on longer so that V_CS

can increase to the level of V_EA. The reverse occurs

when V_FBincreases.

Functional Description

A constant current output converter is the preferred

method for driving LEDs. Small variations in current

have a minimal effect on the light output, whereas small

variations in voltage have a significant impact on light

output. The MIC3223 LED driver is specifically designed

to operate as a constant current LED Driver.

PWM Dimming

This control process just described occurs during each

DIM_IN pulse and when ever DIM_IN is high. When

DIM_IN is low, the boost converter will no longer switch

and the output voltage will drop. For high dimming ratios

use an external PWM Dimming switch as shown in the

Typical Application. When the dim pulse is on the

external switch is on and circuit operates in the closed

loop control mode as described. When the DIM_IN is low

the boost converter does not switch and the external

switch is open and no LED current can flow and the

output voltage does not droop. When DIM_IN goes high

the external switch is driven on and LED current flows.

The output voltage remains the same (about the same)

during each on and off DIM_IN pulse.

The MIC3223 is designed to operate as a boost

converter, where the output voltage is greater than the

input voltage. This configuration allows for the design of

driving multiple LEDs in series to help maintain color and

brightness. The MIC3223 can also be configured as a

SEPIC converter, where the output voltage can be either

above or below the input voltage.

The MIC3223 has an input voltage range, from 4.5V and

20V, to address a diverse range of applications. In

addition, the LED current can be programmed to a wide

range of values through the use of an external resistor.

This provides design flexibility in adjusting the current for

a particular application need.

PWM Dimming can also be used in the Test Circuit in

applications that do not require high dimming ratios. In

the Test Circuit, the load is not removed from the output

voltage between DIM_IN pulses and will therefore drain

the output capacitors. The voltage that the output will

discharge to is determined by the sum of the V_F(forward

voltage drops of the LEDs). When V_OUTcan no longer

forward bias the LEDs, then the LED current will stop

and the output capacitors will stop discharging. During

the next DIM_IN pulse V_OUThas to charge back up

before the full LED current will flow. For applications that

do not require high dimming ratios.

The MIC3223 features a low impedance gate driver

capable of switching large MOSFETs. This low

impedance provides higher operating efficiency.

The MIC3223 can control the brightness of the LEDs via

its PWM dimming capability. Applying a PWM signal (up

to 20kHz) to the DIM_IN pin allows for control of the

brightness of the LEDs.

The MIC3223 boost converter employs peak current

mode control. Peak current mode control offers

advantages over voltage mode control in the following

manner. Current mode control can achieve a superior

line transient performance compared to voltage mode

control and is easier to compensate than voltage mode

control, thus allowing for a less complex control loop

stability design. Page 9 of this datasheet shows the

functional block diagram.

Boost Converter operation

The boost converter is a peak current mode pulse width

modulation (PWM) converter and operates as follows. A

flip-flop (FF) is set on the leading edge of the clock

cycle. When the FF is set, a gate driver drives the power

FET on. Current flows from V_INthrough the inductor (L)

and through the power switch and also through the

current sense resistor to PGND. The voltage across the

current sense resistor is added to a slope compensation

ramp (needed for stability). The sum of the current sense

voltage and the slope compensation voltages (called

V_CS) is fed into the positive terminal of the PWM

comparator. The other input to the PWM comparator is

the error amp output (called V_EA). The error amp’s

negative input is the feedback voltage (V_FB). V_FBis the

voltage across R_ADJ(R5). In this way the output LED

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MIC3223

Output Over Voltage Protection (OVP)

Application Information

The MIC3223 provides an OVP circuitry in order to

protect the system from an overvoltage fault condition.

This OVP threshold can be programmed through the

use of external resistors (R3 and R4 in the Typical

Application). A reference value of 1.245V is used for the

OVP. Equation 3 can be used to calculate the resistor

value for R9 to set the OVP point. Normally use 100k for

R3.

Constant Output Current Converter

The MIC3223 is a peak current mode boost converter

designed to drive high power LEDs with a constant

current output. The MIC3223 operates with an input

voltage range from 4.5V to 20V. In the boost

configuration, the output can be set from V_INup to 37V.

The peak current mode control architecture of the

MIC3223 provides the advantages of superior line

transient response as well as an easier to design

compensation.

R3

Eq. (3)

R4 =

(V_OVP/1.245) −1

VDD

The MIC3223 LED driver features a built-in soft start

circuitry in order to prevent start-up surges. Other

protection features include:

An internal linear regulator is used to provide the

necessary internal bias voltages. When V_INis 6V or

below connect the V_DDpin to V_IN. Use a 10µF ceramic

bypass capacitor.

• Current Limit (I_LIMIT) – Current sensing for over

current and overload protection

DRVVDD

• Over Voltage Protection (OVP) – output over

voltage protection to prevent operation above a

safe upper limit

An internal linear regulator is used to provide the

necessary internal bias voltages to the gate driver that

drives the external FET. When V_INis above 6V connect

DRVVDD to VDD.

• Under Voltage Lockout (UVLO) – UVLO designed

to prevent operation below a safe lower limit

When V_INis 6V or below connect the DRVVDD pin to

VIN. Use a bypass capacitor, 10µF ceramic capacitor.

Setting the LED Current

The current through the LED string is set via the value

chosen for the current sense resistor R_ADJwhich is R5 in

the schematic of the Typical Application. This value can

be calculated using Equation 1:

UVLO

Internal under voltage lock out (UVLO) prevents the part

from being used below a safe V_INvoltage. The UVLO is

3.7V. Operation below 4.5V is not recommended.

0.2V

Eq. (1)

ILED =

R_ADJ

Soft Start

Soft start is employed to lessen the inrush currents

during turn on. At turn on the following occurs;

Another important parameter to be aware of in the boost

converter design is the ripple current. The amount of

ripple current through the LED string is equal to the

output ripple voltage divided by the LED AC resistance

(R_LED– provided by the LED manufacturer) plus the

current sense resistor R_ADJ. The amount of allowable

ripple through the LED string is dependent upon the

application and is left to the designer’s discretion. The

equation is shown in Equation 2.

1. After about 1.5ms C_SSwill start to rise in a

exponential manner according to;

−t

⎛

⎜

⎞

⎟

V_SS= 0.2 1− e^(37kꢀ×C

)

SS

⎜

⎟

⎠

⎝

2. According to the block diagram, V_SSis the ref

node of the error amp. PWM switching start

when V_SSbegins to rise.

V

OUT_RIPPLE

Eq. (2)

ΔI_LED≈

(R_LED+R_ADJ

)

3. When the C_SSis fully charged, 0.2V will be at the

error amp reference and steady state operation

begins.

I_LED×D

C_OUT×F_SW

Where

V_OUT

=

RIPPLE

4. Design for soft-start time using the above

equation.

Reference Voltage

The voltage feedback loop the MIC3223 uses an internal

voltage of 200mV with an accuracy of ±5%. The

feedback voltage is the voltage drop across the current

sense resistor as shown in the Typical Application.

When in regulation the voltage at V_FBwill equal 200mV.

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MIC3223

If high dimming ratios are required, a lower Dimming

frequency is required. During each DIM_IN pulse the

inductor current has to ramp up to it steady state value in

order for the programmed LED current to flow. The

smaller the inductance value the faster this time is and a

narrower DIM_IN pulse can be achieved. But smaller

inductance means higher ripple current.

Figure 1. Soft start

LED Dimming

The MIC3223 LED driver can control the brightness of

the LED string via the use of pulse width modulated

(PWM) dimming. An input signal from DC up to 20kHz

can be applied to the DIM_IN pin (see Typical

Application) to pulse the LED string ON and OFF. It is

recommended to use PWM dimming signals above

120Hz to avoid any recognizable flicker by the human

eye. PWM dimming is the preferred way to dim an LED

in order to prevent color/wavelength shifting. Color

wavelength shifting will occur with analog dimming. By

employing PWM Dimming the output current level

remains constant during each DIM_IN pulse. The boost

converter switches only when DIM_IN is high. Between

DIM_IN pulses the output capacitors will slowly

discharge. The higher the DIM_IN frequency the less the

output capacitors will discharge.

Figure 3. PWM Dimming 20%

Figure 3 shows that switching occurs only during DIM_IN

on pulses. When DIM_IN is low the boost converter

stops switching and the external LED is turned off. The

LED current flows only when DIM_IN is high. Figure 3

shows that the compensation pin (V_COMP) does not

discharge between DIM_IN pulses. Therefore, when the

DIM_IN pulse starts again the converter resumes

operation at the same V_COMPvoltage. This eliminates the

need for the comp pin to charge up during each DIM_IN

pulse and allows for high Dimming ratios.

PWM Dimming Limits

The minimum pulse width of the DIM_IN is determined

by the DIM_IN frequency and the L and C used in the

boost stage output filter. At low DIM_IN frequencies

lower dimming ratios can be achieved.

LED_ON_TIME

Dim_ratio =

PERIOD_PWMD

Figure 4. PWM Dimming 10% and I_LED100Hz

Figure 2. DIM_IN Dimming Ratio

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MIC3223

Figure 5. PWM Dimming 20% and I_LED1kHz

Figure 7. 5µs DIM_IN Pulse

In Figure 4 is at 100Hz dimming frequency and Figure 5

is 1kHz dimming frequency. The time it takes for the

LED current to reach it full value is longer with a lower

Dimming frequency. The reason is the output capacitors

slowly discharge between dimming pulses.

Figure 7 shows the minimum DIM_IN pulse at these

operating conditions before the I_LEDcurrent starts to drop

due to low V_OUT. The converter is ON (switching) only

during a DIM_IN pulse.

Figure 7 shows that at this DIM_IN pulse width the

converter is ON (switching) long enough to generate the

necessary V_OUTto forward bias the LED string at the

programmed current level. Therefore this condition will

result in the desired I_LED

.

Figure 6. PWM Dimming 20% and I_LED1kHz

Figure 6 shows the output voltage V_OUTdischarge

between DIM_IN pulses. The amount of discharge is

dependent on the time between DIM_IN pulses.

Figure 8. 2.5µs DIM_IN Pulse

Figure 8 shows that at this DIM_IN pulse width the

converter in not ON (switching) long enough to generate

the necessary V_OUTto forward bias the LED string at the

programmed current level. As a result the LED current

drops. Therefore, this condition will not result in the

desired I_LED

.

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MIC3223

Design Procedure for a LED Driver

Symbol

Input

V_IN

Parameter

Min

Nom

Max

Units

Input Voltage

8

5

12

14

2

V

A

I_IN

Input Current

Output

LEDs

V_F

Number of LEDs

Forward Voltage of LED

Output Voltage

6

7

3.2

16

3.5

21

4.0

28

V

A

V_OUT

I_LED

LED Current

0.33

0

0.35

40

0.37

I_PP

Required I Ripple

Output Power

mA

W

%

Pout

DIM_IN

OVP

System

F_SW

10.36

100

PWM Dimming

Output Over Voltage Protection

30

V

Switching Frequency

Efficiency

1

MHz

%

eff

80

0.5

V_DIODE

Forward drop of schottky diode

V

Table 1. Design example parameters

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MIC3223

Design Example

Using Equation 5, the following values have been

calculated:

In this example, we will be designing a boost LED driver

operating off a 12V input. This design has been created to

drive 6 LEDs at 350mA with a ripple of about 20%. We are

designing for 80% efficiency at a switching frequency of

1MHz.

V_OUT(max)×I_OUT(max)

I_{IN_RMS(max)}

I_{IN_RMS(nom)}

I_{IN_RMS(min)}

=

= 1.54A_(RMS)

= 0.74A_(RMS)

= 0.46A_(RMS)

eff × V

IN(min)

V_OUT(nom)×I_OUT(nom)

Eq (5)

Select R_ADJ

eff × V

IN(nom)

Having chosen the LED drive current to be 350mA in this

example, the current can be set by choosing the R_ADJ

resistor from Equation 1:

V_OUT(min)×I_OUT(min)

=

eff × V

IN(max)

0.2V

I_OUTis the same as I_LED

.

R_ADJ

=

= 0.57ꢀ

0.35A

Selecting the inductor current (peak-to-peak), I_{L_PP}, to be

between 20% to 50% of I_{IN_RMS(nom)}, in this case 40%, we

obtain:

Use the next lowest standard value 0.56ꢀ.

I_LED= 0.36A

I_{IN_PP(nom)}= 0.4 × I_{IN_RMS(nom)}= 0.4 × 0.74 = 0.30A_P-P

The power dissipation in this resistor is:

It can be difficult to find large inductor values with high

saturation currents in a surface mount package. Due to

this, the percentage of the ripple current may be limited by

the available inductor. It is recommended to operate in the

continuous conduction mode. The selection of L described

here is for continuous conduction mode.

P_RADJ= ILED²× R_ADJ= 71mW

Use a resistor rated at quarter watt or higher.

Operating Duty Cycle

The operating duty cycle can be calculated using Equation

four provided below:

V

×D

×F

IN

Eq. (6)

L =

(

V_OUT− V_IN+ V_DIODE

V_OUT+ V_DIODE

)

I

Eq. (4)

D =

IN_PP SW

Using the nominal values, we get:

12V ×0.44

V_DIODEis the V_fof the output diode D1 in the Typical

Application. It is recommended to use a schottky diode

because it has a lower V_fthan a junction diode.

L =

= 18μH

0.3A ×1MHz

These can be calculated for the nominal (typical) operating

conditions, but should also be understood for the minimum

and maximum system conditions as listed below.

Select the next higher standard inductor value of 22µH.

Going back and calculating the actual ripple current gives:

V_IN(min)× D_max

L × F_SW

8V × 0.72

22μH×1MHz

(

V_OUT(nom)− V

+ V_DIODE

)

IN(nom)

I_{IN_PP(max)}

=

= 0.26A_PP

Dnom =

Dmax =

Dmin =

Dnom =

V_OUT(nom)+ V_DIODE

The average input current is different than the RMS input

current because of the ripple current. If the ripple current is

low, then the average input current nearly equals the RMS

input current. In the case where the average input current

is different than the RMS, equation 7 shows the following:

(

V_OUT(max)− V

+ V_DIODE

)

IN(min)

V_OUT(max)+ V_DIODE

(

V_OUT(min)− V_IN(max)+ V_DIODE

)

V_OUT(min)+ V_DIODE

2

(I_{IN_PP}

12

)

2

Eq. (7)

I_{IN_AVE(max)}

=

(

I_{IN_RMS(max)}

)

−

(

21− 12 − 0.5

)

= 0.44

21+ 0.5

21- 12 + 0.5

21+ 0.5

(0.24)²

12

(1.54)²−

≈ 1.54A

(

)

= 0.44

I_{IN_AVE(max)}

=

D

=

nom

The Maximum Peak input current I_{L_PK}can found using

Equation 8:

Therefore D_nom= 44%, D_max= 72% and D_min= 15%.

Inductor Selection

Eq. (8)

I_{L_PK(max)}= I_{IN_AVE(max)}+ 0.5 ×I_{L_PP(max)}= 1.67A

First calculate the RMS input current (nominal, min and

max) for the system given the operating conditions listed in

the design example table. The minimum value of the RMS

input current is necessary to ensure proper operation.

The saturation current (I_SAT) at the highest operating

temperature of the inductor must be rated higher than this.

The power dissipated in the inductor is:

M9999-011510-A

January 2010

15

Micrel, Inc.

Eq. (9)

MIC3223

P_INDUCTOR= I_{IN_RMS(max)}²× DCR

I_{IN_PP}

(0.3A)

C_IN

=

= 0.75μF

A Coilcraft # MSS1260-223ML is used in this example. Its

V_IN(ripple)× F_SW

8 × 50mV ×1MHz

DCR is 52mꢀ, I_SAT=2.7A

_INDUCTOR= 1.54²× 52 mꢀ = 0.123W

This is the minimum value that should be used. To protect

the IC from inductive spikes or any overshoot, a larger

value of input capacitance may be required.

P

Output Capacitor

Use 2.2µF or higher as a good safe min.

In this LED driver application, the I_LEDripple current is a

more important factor when compared to that of the output

ripple voltage (although the two are directly related). To

find the C_OUTfor a required I_LEDripple use the following

calculation:

Rectifier Diode Selection

A schottky diode is best used here because of the lower

forward voltage and the low reverse recovery time. The

voltage stress on the diode is the max V_OUTand therefore

a diode with a higher rating than max V_OUTshould be used.

An 80% de-rating is recommended here as well.

For an output ripple I_LED(ripple)= 20ma

I_LED(nom)× D_nom

Eq. (10)

C_OUT=

I_LED(ripple)× (R_ADJ+ R_{LED_total})× F_SW

Eq. (14)

Since I_{IN_AVE(max)}occurs when D is at a maximum.

Eq. (15) _DIODE(max)≈ V_DIODE× I_{DIODE_(max)}

I_DIODE(max)= I_OUT(max)= 0.36A

Find the equivalent ac resistance R_{LED_ac}from the

datasheet of the LED. This is the inverse slope of the I_LED

vs. V_fcurve i.e.:

P

A SK35B is used in this example, it’s V_DIODEis 0.5V

ΔV_f

P_DIODE(max)≈ 0.5V × 0.36A = 0.18W

Eq. (11)

R_{LED_ac}=

ΔLED

MIC3223 Power Losses

In this example use R_{LED_ac}= 0.6ꢀ for each LED.

To find the power losses in the MIC3223:

There is about 6mA input from V_INinto the V_DDpin.

If the LEDs are connected in series, multiply R_{LED_ac}= 0.6ꢀ

by the total number of LEDs. In this example of six LEDs,

we obtain the following:

The internal power switch has an RDS_ONof about 170mꢀ

at.

R

_{LED_total}≡ R_dynamic= 6 × 0.6ꢀ = 3.6ꢀ

P

_MIC3223= V_IN× 6mA + PwrFET

Eq. (12)

Eq. (16)

PwrFET = I_{FET_RMS(max)}²× R_{ds_on_@100°}

I_LED(nom)× D_nom

I_LED(ripple)× (R_ADJ+ R_{LED_total})× F_SW

Use 2.2µF or higher.

C_OUT

=

= 1.9μF

+ V_OUT(max)× I_{IN_AVE(max)}× tsw × Fsw

R

_{ds_on_@100°}≈ 160mꢀ

tsw ≈ 30ns is the internal Power FET ON an OFF

There is a trade off between the output ripple and the

rising edge of the DIM_IN pulse. This is because between

PWM dimming pulses, the converter stops pulsing and

C_OUTwill start to discharge. The amount that C_OUTwill

discharge depends on the time between PWM Dimming

pluses. At the next DIM_IN pulse, C_OUThas to be charged

up to the full output voltage V_OUTbefore the desired LED

current flows.

transition time.

2

⎛

⎞

⎟

I_{L_PP}

12

2

⎜

I_SW

=

D I_{IN_AVE(max)}

+

= 1.3A

RMS(max)

⎜

⎟

⎝

⎠

PwrFET = 1.3A²× 160mꢀ + 28V × 1.54A × 30ns

× 1MHz = 1.6W

P_MIC3223= 8 × 6mA + 1.77W = 1.66W

Input Capacitor

Snubber

The input capacitor is shown in the Typical Application.

For superior performance, ceramic capacitors should be

used because of their low equivalent series resistance

(ESR). The input capacitor C_INripple current is equal to the

ripple in the inductor. The ripple voltage across the input

capacitor, C_INis the ESR of C_INtimes the inductor ripple.

The input capacitor will also bypass the EMI generated by

the converter as well as any voltage spikes generated by

A snubber is a damping resistor in series with a DC

blocking capacitor in parallel with the power switch (same

as across the flyback diode because V_OUTis an ac

ground). When the power switch turns off, the drain to

source capacitance and parasitic inductance will cause a

high frequency ringing at the switch node. A snubber

circuit as shown in the application schematic may be

required if ringing is present at the switch node. A critically

damped circuit at the switch node is where R equals the

characteristic impedance of the switch node.

the inductance of the input line. For a required V_IN(ripple)

:

Eq. (13)

M9999-011510-A

January 2010

16

Micrel, Inc.

Eq.(17)

MIC3223

L

parisitic

C_ds

R

=

snubber

The explanation of the method to find the best R snubber

is beyond the scope of this data sheet.

Use R_snubber= 2ꢀ, ½ watt and C_snubber= 470pf to 1000pf.

Figure 10. Simplified Control Loop

The power dissipation in the R_snubberis:

_snubber= C_snubber× V_OUT²× F_SW

Eq. (19)

Where

T(s) = G_ea(s) × G_vc(s) × H(s)

R

P_snubber= 470pF × 28V²× 1MHz = 0.4W

R_ADJ

For a LED driver H(s) =

and

R_ADJ+ R_dynamic

Power Loss in the L

Power Loss in the sckottky diode

P_snubber

0.123 W

0.2 W

0.4 W

1.66 W

2.4W

⎛

⎞

⎟

⎛

⎜

⎝

⎞

⎟

⎠

1

⎜

G_ea(s) = g_mZ_O|| R_comp

+

⎜

⎟

sC_comp

⎝

⎠

MIC3223 Power Loss

Total Losses

Eq. (20)

Efficiency

80%

V_OUT(s)

G_VC(s) =

Table 2. Major Power Losses

V_CONTROL(s)

⎛

⎞

⎟

sL

Table 2 showing the Power losses in the Design Example.

⎜

1−

(

1+ sC_OUTR_ESR

)

D'²R_dynamic

⎜

⎟

1

D'R

2

⎛

⎜

⎞⎛

⎟⎜

⎞

⎟

⎝

⎠

OP

OVP - Over Voltage Protection

=

sR_dynamicC_OUT

Ri

⎛

⎜

⎝

⎞

⎟

⎠

⎝

⎠⎝

⎠

Set OVP higher than the maximum output voltage by at

least one Volt. To find the resistor divider values for OVP

1+

2

use equation 18 and set the OVP = 30V and R_{OVP_H}

=

Where

100kꢀ:

V_OUT

R_OP

=

Is the DC operating point of the converter.

100kꢀ×1.245

30 −1.245

Eq. (18)

R_{OVP_L}

=

= 4.33kꢀ

I_LED

R_dymanicis the ac load the converter sees. When the load

on the converter is a string of LEDs, R_dymanicis the series

sum of the R_LED(ac)of each LED.

Compensation

R_{LED_total}is usually between 0.1ꢀ to 1ꢀ per LED. It can be

calculated from the slope of I_LEDvs. V_fplot of the LED.

Ri = Ai × Rcs = 0.86ꢀ

Ai = 114 and Rcs ≡ 7.5mꢀ; are internal to the ic.

The equation for G_vc(s) is theoretical and should give a

good idea of where the poles and zeros are located.

Figure 9. Current Mode Loop Diagram

D'²R_dynamic

Eq.(20) shows that s =

→ f_RHPZ=

Current mode control simplifies the compensation. In

current mode, the complex poles created by the output L

and C are reduced to a single pole. The explanation for

this is beyond the scope of this datasheet, but it’s

generally thought to be because the inductor becomes a

constant current source and can’t act to change phase.

L

2πL

is a RHP Zero. The loop bandwidth should be about 1/5 to

1/10 of the frequency of R_HPZto ensure stability. From

Equation (20) it is shown that there is only the single pole.

1

s =

→ f_pole

=

and a Zero

R_dynamicC_OUT

2πR_dynamicC_OUT

From the small signal block diagram the loop transfer

function is:

due to the ESR of the output capacitor.

1

s =

→ f_ESR=

R_ESRC_OUT

2πR_ESRC_OUT

M9999-011510-A

January 2010

17

Micrel, Inc.

MIC3223

The error amp is a gm type and the gain G_ea(s) is

This greatly simplifies the compensation.

One needs only to get a bode plot of the transfer function

of the control to output G_vc(s) with a network analyzer

and/or calculate it. From the bode plot find what the gain of

⎛

⎞

⎛

⎞

⎟

⎠

1

⎜

⎟

⎜

Eq. (21)

G_ea(s) = g_mZ_O|| R_comp

+

⎜

⎝

⎟

⎠

sC_comp

⎝

R_HPZ

0.8mA

V

G_vc(s) is at f =

. Next design the error amp gain

g_m

=

and Z_o= 1.2Mꢀ.

10

G_ea(s) so the loop gain at the cross over frequency T(f_co) is

1

f_coR_HPZ

R_HPZ

The zero is f_zero

=

.

0 db where fco =

or less.

2R_compC_comp10

100

10

Error Amp

Error Amp Gain and Phase

60

40

Gain

20

0

-20

-40

-60

-80

Phase

1.E+02

1.E+03

1.E+04

1.E+05

1.E+06

FREQUENCY (Hz)

Set the fco at the mid band where G_ea(f_co) = gm × R_comp. At

f_zero× 10 the phase boost is near its maximum.

Figure 11. Internal Error Amp and External Compensation

Figure 12. Error Amp Transfer Function

M9999-011510-A

January 2010

18

Micrel, Inc.

MIC3223

Other Applications

Figure 13. MIC3223 Typical Application without External PWM Dimming Switch

Audio noise

2. Even though the RRC is very short (tens of

nanoseconds) the peak currents are high (multiple

amperes). These fast current spikes generate EMI

(electromagnetic interference). The amount of RRC

is related to the die size and internal capacitance of

the diode. It is important not to oversize (i.e. not

more than the usual de rating) the diode because

the RRC will be needlessly higher. Example: If a 2A

diode is needed do not use a higher current rated

diode because the RRC will be needlessly higher. If

a 25V diode is needed do not use a 100V etc.

3. The high RRC causes a voltage drop on the ground

trace of the PCB and if the converter control IC is

referenced to this voltage drop, the output regulation

will suffer.

Audio noise from the output capacitors may exits in a

standard boost LED converter. The physical dimensions

of ceramic capacitors change with the voltage applied to

them. During PWM Dimming, the output capacitors in

standard converters are subjected to fast voltage and

current transients that may cause the output capacitors

to oscillate at the PWM Dimming frequency. This is one

reason users may want PWM dimming frequencies

above the audio range.

PCB Layout

1. All typologies of DC-to-DC converters have a

Reverse Recovery Current (RRC) of the flyback

or (freewheeling) diode. Even a Schottky diode,

which is advertised as having zero RRC, it really

is not zero. The RRC of the freewheeling diode

in a boost converter is even greater than in the

Buck converter. This is because the output

voltage is higher than the input voltage and the

diode has to charge up to –V_OUTduring each on-

time pulse and then discharge to V_fduring the

off-time.

4. For good output regulation, it is important to connect

the IC’s reference to the same point as the output

capacitors to avoid the voltage drop caused by RRC.

This is also called a star connection or single point

grounding.

5. Feedback trace: The high impedance traces of the

FB should be short.

M9999-011510-A

January 2010

19

Micrel, Inc.

MIC3223

Evaluation Board Schematic

37V Max 1A LED Driver

M9999-011510-A

January 2010

20

Micrel, Inc.

MIC3223

Bill of Materials

Item

Part Number

Manufacturer

muRata⁽¹⁾

TDK⁽²⁾

Description

Qty

1

GRM319R61E475KA12D

C3216X7R1E475M

C1

Ceramic Capacitor, 4.7µF, 25V, Size 1206, X7R

Ceramic Capacitor, 0.027µF, 6.3V, Size 0603, X7R

Ceramic Capacitor, 10µF, 6.3V, Size 0603, X7R

12063D475KAT2A

AVX⁽³⁾

C2

GRM188R71C273KA01D

GRM188R60J106ME47D

C1608X5R0J106K

muRata

TDK

1

C3, C7

2

08056D106MAT2A

AVX

12105C475KAZ2A

AVX

C4, C6

C5

Ceramic Capacitor, 4.7µF, 50V, Size 1210, X7R

Ceramic Capacitor, 0.047µF, 6.3V, Size 0603, X7R

2

1

GRM32ER71H475KA88L

GRM188R71C473KA01D

0603YC473K4T2A

muRata

AVX

C8

GRM188R72A102KA37D

SK35B

muRata

MCC⁽⁴⁾

Coilcraft⁽⁶⁾

Vishay Dale⁽⁴⁾

Vishay Dale

Ceramic Capacitor, 1000pF, 100V Size 0603, X7R

Schottky Diode, 3A, 50V (SMB)

Inductor, 22µH, 5A

D1

1

2

1

L1

MSD1260-223ML-LD

CRCW0603100KFKEA

CRCW0603549RFKEA

CRCW06033K24FKEA

R1, R3

R2

Resistor, 100k, 1%, Size 0603

Resistor, 549ꢀ, 1%, Size 0603

Resistor, 3.24k, 1%, Size 0603

R4

Resistor, 0.56ꢀ, 1%, 1/2W, Size 1206

(for .35A LED current Change for different ILED)

R5

R6

CRCW1206R560FKEA

RMC 1/4 2 1% R

Vishay Dale

1

Stackpole Electronics,

Inc.⁽⁷⁾

Resistor, 2ꢀ, 1%, 1/2W, Size 1210

Si2318DS

AM2340N

MIC3223

Vishay Siliconix⁽⁴⁾

Analog Power⁽⁸⁾

Micrel, Inc.⁽⁹⁾

Q1

N-Channel 40V MOSFET

1

U1

High Power Boost LED Driver with Integrated FET

1

Notes:

1. Murata: www.murata.com.

2. TDK: www.tdk.com.

3. AVX: www.avx.com.

4. Vishay: www.vishay.com.

5. Internacional Rectifier: www.ift.com.

6. Coilcraft: www.coilcraft.com

7. Stackpole Electronics, Inc.: www.

8. Analog Power: www.analogpowerinc.com

8. Micrel, Inc.: www.micrel.com.

M9999-011510-A

January 2010

21

Micrel, Inc.

MIC3223

PCB Layout Recommendations

Top Layer

Bottom Layer

M9999-011510-A

January 2010

22

Micrel, Inc.

MIC3223

Package Information

16-Pin ePad TSSOP (TSE)

M9999-011510-A

January 2010

23

Micrel, Inc.

MIC3223

Recommended Land Pattern

MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA

TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com

The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its

use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.

Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product

can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant

into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A

Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully

indemnify Micrel for any damages resulting from such use or sale.

M9999-011510-A

January 2010

24


型号：	MIC3223YTSE-TR
厂家：	MICROCHIP
描述：	LED DISPLAY DRIVER 驱动光电二极管接口集成电路
文件：	总24页 (文件大小：1280K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

MIC3223YTSE-TR [MICROCHIP]

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