MC13142D [MOTOROLA]
LOW POWER DC - 1.8 GHz LNA, MIXER and VCO; 低功耗DC - 1.8 GHz的低噪声放大器，混频器和VCO
| 型号： | MC13142D |
| 厂家： | 
MOTOROLA |
| 描述： | LOW POWER DC - 1.8 GHz LNA, MIXER and VCO |
| 文件： | 总16页 (文件大小：269K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
Order this document By MC13142/D
LOW POWER DC – 1.8 GHz
LNA, MIXER and VCO
The MC13142 is intended to be used as a first amplifier, voltage controlled
oscillator and down converter for RF applications. It features wide band
operation, low noise, high gain and high linearity while maintaining low
current consumption. The circuit consists of a Low Noise Amplifier (LNA), a
Voltage Controlled Oscillator (VCO), a buffered oscillator output, a mixer, an
SEMICONDUCTOR
TECHNICAL DATA
Intermediate Frequency amplifier (IF
) and a dc control section. The wide
amp
mixer IF bandwidth allows this part also to be used as an up converter and
exciter amplifier.
• Wide RF Bandwidth: DC–1.8 GHz
• Wide LO Bandwidth: DC–1.8 GHz
• Wide IF Bandwidth: DC–1.8 GHz
• Low Power: 13 mA @ V
= 2.7 – 6.5 V
CC
16
• High Mixer Linearity: P
= 3.0 dBm
i1.0 dB
1
• Linearity Adjustment Increases IP
• Single–Ended 50 Ω Mixer Input
Up to 20 dBm
3in
• Double Balanced Mixer Operation
• Open Collector Mixer Output
D SUFFIX
PLASTIC PACKAGE
CASE 751B
• Single Transistor Oscillator with Collector, Base and Emitter Pinned Out
• Buffered Oscillator Output
(SO–16)
PIN CONNECTIONS
SO–16
EN
RF
V
1
2
3
4
5
6
7
16
15
14
13
12
11
10
out
RF
in
CC
Mx Lin
Cont
V
Mix Lin Cont
EE
Osc E
Osc B
RF
m
V
EE
Osc C, V
IF+
IF–
CC
Buff
ORDERING INFORMATION
Operating
V
8
9
V
CC
EE
Temperature Range
Device
Package
This device contains 176 active transistors.
MC13142D
T
A
= –40° to +85°C
SO–16
This document contains information on a new product. Specifications and information herein
Motorola, Inc. 1998
Rev 1
are subject to change without notice.
MC13142
MAXIMUM RATINGS (T = 25°C, unless otherwise noted.)
A
Rating
Power Supply Voltage
Symbol
Value
Unit
Vdc
Vdc
V
7.0
CC(max)
Operating Supply Voltage Range
V
CC
2.7 to 6.5
NOTE: ESD data available upon request.
ELECTRICAL CHARACTERISTICS (V
= 3.0 V, T = 25°C, LO = –10 dBm @ 950 MHz, IF @ 50 MHz.)
in
CC
A
Characteristic
Symbol
Min
Typ
Max
Unit
Supply Current (Disable)
Pin 15 with Pin 1 @ 0 V
Pin 10 and 11 with Pin 1 @ 0 V
Pin 6 with Pin 1 @ 0 V
I
–230
–110
–20
–
–
–
–
230
110
20
CC_Total
I
CC_15
µA
I
CC_Mix
I
–100
100
CC_6
Supply Current (Enable)
Pin 15 with Pin 1 @ 3.0 V
Pin 10 with Pin 1 @ 3.0 V
Pin 6 with Pin 1 @ 3.0 V
I
8.25
1.0
1.25
6.0
13.5
26
4.5
7.5
14
CC_Total
I
–
–
–
CC_15
mA
I
CC_Mix
I
CC_6
Amplifier Gain (50 Ω Insertion Gain)
Amplifier Reverse Isolation
S
6.5
12
–33
–10
–15
–15
–5.0
1.8
13
–
dB
dB
21
12
S
–
Amplifier Input Match
Γ
–
–
dB
in amp
Amplifier Output Match
Γ
–
–
dB
out amp
Amplifier 1.0 dB Gain Compression
Amplifier Input Third Order Intercept
Amplifier Noise Figure (Application Circuit)
Amplifier Gain @ N.F.
Pin
–18
–8.0
–
dBm
dBm
dB
–1.0 dB
IP3
in
NF
–
1.0
4.0
–
G
–
17
dB
NF
Mixer Voltage Conversion Gain (R = R = 800 Ω)
VG
PG
–
9.0
–
dB
P
L
C
C
Mixer Power Conversion Gain (R = R = 800 Ω)
–7.0
–3.0
–20
12
–2.0
–
dB
P
L
Mixer Input Match
Γ
–
dB
in M
Mixer SSB Noise Figure
NF
–
–
dB
SSBM
Mixer 1.0 dB Gain Compression
Mixer Input Third Order Intercept
Oscillator Buffer Drive (50 Ω)
Pin
–1.0 dBM
–
3.0
–
dBm
dBm
dBm
dBc/Hz
dB
IP3
–
–1.0
–16
–90
–35
–35
–35
–35
–35
–25
–25
–
InM
P
VCO
–19.5
–12
–
Oscillator Phase Noise @ 25 kHz Offset
N
–
–
–
–
–
–
–
–
Φ
RF Feedthrough to RF
P
–
in
m
RFin–RFm
RF
Feedthrough to RF
P
–
dB
out
m
RFout–RFm
LO Feedthrough to IF
LO Feedthrough to RF
LO Feedthrough to RF
P
–
dBm
dBm
dBm
dB
LO–IF
P
–
in
LO–RFin
P
–
m
LO–RFm
Mixer RF Feedthrough to IF
Mixer RF Feedthrough to RF
P
–
RFm–IF
P
–
dB
in
RFm–RFin
2
MOTOROLA RF/IF DEVICE DATA
MC13142
CIRCUIT DESCRIPTION
General
The MC13142 is a low power LNA, double–balanced
(MSG) yields noise performance within a few tenths of a dB
of the minimum noise figure.
Mixer, and VCO. This device is designated for use as the
frontend section in analog and digital FM systems such as
Digital European Cordless Telephone (DECT), PHS, PCS,
Cellular, UHF and 800 MHz Special Mobile Radio (SMR),
UHF Family Radio Services and 902 to 928 MHz cordless
telephones. It features a mixer linearity control to preset or
auto program the mixer dynamic range, an enable function
and a wideband IF so the IC may be used either as a down
converter or an up converter. Further details are covered in
the Pin by Pin Description which shows the equivalent
internal circuit and external circuit requirements.
Mixer
The mixer is a double–balanced four quadrant multiplier
biased class AB allowing for programmable linearity control
via an external current source. An input third order intercept
point of 20 dBm may be achieved. All 3 ports of the mixer are
designed to work up to 1.8 GHz. The mixer has a 50 Ω
single–ended RF input and open collector differential IF
outputs. An on–board Local Oscillator transistor has the
emitter, base and collector pinned out to implement a low
phase noise VCO in various configurations. Additionally, a
buffered LO output is provided for operation with a frequency
synthesizer. The linear gain of the mixer is approximately
0 dB with a SSB noise figure of 12 dB in the IF output circuit
configuration shown in the application example.
Current Regulation/Enable
Temperature compensating voltage independent current
regulators are controlled by the enable function in which
“high” powers up the IC.
Local Oscillator
Low Noise Amplifier (LNA)
The on–chip transistor operates with coaxial transmission
line or LC resonant elements to over 2.0 GHz. Biasing is
done with a temperature compensated current source in the
emitter and a collector to base internal resistor of 7.6 kΩ;
The LNA is internally biased at low supply current
(approximately 2.0 mA emitter current) for optimal noise
figure and gain. The LNA output is biased internally with a
however, an RFC from V
application circuit shows a voltage controlled Clapp oscillator
operating at center frequency of 975 MHz.
to base is recommended. The
600 Ω resistor to V . Input and output matching may be
achieved at various frequencies using few external
components. Matching the LNA for Maximum stable gain
CC
CC
3
MOTOROLA RF/IF DEVICE DATA
MC13142
PIN FUNCTION DESCRIPTION
Pin
16 Pin
SOIC
Equivalent Internal Circuit
(20 Pin LQFP)
Symbol
EN
Description
1
Enable, E Osc
V
CC
In SO–16, both enables, (for the Oscillator/LO Buffer and
LNA/Mixer) are bonded to Pin 1. Enable by pulling up to
1
40 k
2.0 V
V
CC
or to greater than 2.0 V
.
BE
EN
BE
V
CC
70 k
2.0 V
BE
2
3
RF
RF Input
V
in
CC
The input is the base of an NPN low noise amplifier.
Minimum external matching is required to optimize the
input return loss and gain.
600
16
RF
V
ref2
out
V
EE
V
– Negative Supply
pin is taken to an ample dc ground plane through a
EE
V
EE
low impedance path. The path should be kept as short
as possible. A two sided PCB is implemented so that
ground returns can be easily made through via holes.
V
EE
V
ref3
2
RF
in
3
16
RF
out
RF Output
2.0 mA
The output is from the collector of the LNA; it is internally
biased with a 600 Ω resistor to V . As shown in the 926
MHz application receiver the output is conjugately
matched with a shunt L, and series L and C network.
V
CC
EE
4
5
6
Osc E
Osc B
Osc C
On–Board VCO Transistor
The transistor has the emitter, base and collector + V
pins available. Internal biasing which is compensated for
stability over temperature is provided. It is recommended
4
CC
Osc E
10
that the base pin is pulled up to V
through an RFC
5
CC
chosen for the particular oscillator center frequency. The
application circuit shows a modified Colpitts or Clapp
oscillator configuration and its design is discussed in
detail in the application section.
1.5
mA
Osc B
7.6 k
6
V
EE
Osc C
6
8
V
CC
V
CC
Supply Voltage (V
)
CC
pins are provided for the Local Oscillator and
Two V
V
V
CC
LO Buffer Amplifier. The operating supply voltage range
EE
6
is from 2.7 Vdc to 6.5 Vdc. In the PCB layout, the V
trace must be kept as wide as feasible to minimize
CC
V
CC
inductive reactances along the trace. V
should be
CC
at the IC pin as shown in the
component placement view.
CC
decoupled to V
EE
7
LO Buf
7
LO Buff
Local Oscillator Buffer
1.0
mA
This is a buffered output providing –16 dBm
8
(50 Ω termination) to drive the f pin of a PLL
in
synthesizer. Impedance matching to the synthesizer may
be necessary to deliver the optimal signal and to
improve the phase noise performance of the VCO.
V
CC
V
EE
4
MOTOROLA RF/IF DEVICE DATA
MC13142
PIN FUNCTION DESCRIPTION (continued)
Pin
16 Pin
SOIC
Equivalent Internal Circuit
(20 Pin LQFP)
Symbol
Description
9, 12
V
EE
V
, Negative Supply
10
EE
These pins are V
supply for the mixer IF output. In the
EE
IF–
9
application PC board these pins are tied to a common
trace with other V pins.
V
V
EE
CC
EE
V
V
CC
EE
10, 11
IF–, IF+
IF Output
The IF is a differential open collector configuration which
designed to use over a wide frequency range for up
conversion as well as down conversion. Differential to
single–ended circuit configuration and matching options
are discussed in the application section. 6.0 dB of
additional Mixer gain can be achieved by conjugately
matching at the desired IF frequency.
11
IF+
12
V
EE
13
RF
Mixer RF Input
m
V
The mixer input impedance is broadband 50 Ω for
applications up to 1.8 GHz. It easily interfaces with a RF
ceramic filter as shown in the application schematic.
CC
V
ref1
33
33
V
13
EE
RF
14
Mix Lin
Cont
Mixer Linearity Control
m
The mixer linearity control circuit accepts approximately
0 to 2.3 mA control current to set the dynamic range of
the mixer. An Input Third Order Intercept Point, IIP3 of
20 dBm may be achieved at 2.3 mA of control current
(approximately 7.0 mA of additional supply current).
14
Mix Lin
Cont
400 µA
15
V
CC
V
CC
, Power Supply
V
CC
15
V
CC
5
MOTOROLA RF/IF DEVICE DATA
MC13142
APPLICATIONS INFORMATION
Evaluation PC Board
Component Selection
The evaluation PCB is very versatile and is intended to be
used across the entire useful frequency range of this device.
The PC board accommodates all SMT components on the
circuit side (see Circuit Side Component Placement View).
This evaluation board will be discussed and referenced in
this section.
The evaluation PC board is designed to accommodate
specific components, while also being versatile enough to
use components from various manufacturers. The circuit
side placement view is illustrated for the components
specified in the application circuit. The application circuit
schematic specifies particular components that were used to
achieve the results given and specified in the tables but
alternate components of the same Q and value should give
equivalent results.
Figure 1. Application Circuit
(926.5 MHz)
V
CC
PC Rotary SW
51
47 p
RF
Input
Enable
3.0 p
39 nH
100 p
18 nH
*Z
50
=
O
SMA
1
2
3
4
5
6
16
15
14
13
12
11
10
9
Ω
DC Bias
LNA
6.8 nH
V
Toko RF
Filter
CC
Z
= 50 Ω
O
3.9 p
V
Control
Mixer Linearity
Control
3.6 p
2.3 mA
Max
120 k
2.4 p
100 n
VCO
2.4 p
Mixer
47 p
1.0
µ
2.55 nH
MMBV809
33 k
5.6 p
390 nH
Z Transformer
16:1
L
IF
V
CC
Output
IF
Outputs
C
7
8
SMA
100 n
100 p
LO Buffer
LO Buffer
Output
100 p
SMA
NOTE: *50 Ω Microstrip Transmission Line; length shown in Figure 2.
6
MOTOROLA RF/IF DEVICE DATA
MC13142
Figure 2. 900 MHz Circuit Side Component Placement View
MC13142D
Rev A
Mix In
Mix Lin Cont
Toko
926A10
Dielectric
Filter
C
L
IF
Out
16:1
Impedance
Transformer
LNA
Output
100 p
18 nH
PC
Rotary
51
MC13142D
Switch
2.4 p
2.4 p
47 p
6.8 nH
100 p
390 nH
LNA
Input
LO
Buf
Out
5.6 p
33 k
MMBV809
120 k
V–Cont
V
CC
NOTES: The PCB is laidout for the 4DFA (2 pole SMD type) and 4DFB (3 pole SMD type) filters which are available for applications in
cellular and GSM,GPS (1.2–1.5 GHz), DECT, PHS and PCS (1.8–2.0 GHz) and ISM Bands (902–928 MHz and 2.4–2.5 GHz).
In the component placement shown above, the 926.5 MHz dielectric type image filter is used (Toko Part # 4DFA–926A10).
The PCB also accommodates a surface mount SAW filter in an eight or six pin ceramic package for the cellular base and
handset frequencies. Recommended manufacturers are Siemens and Murata.
Traces are provided on the PCB to evaluate the LNA and mixer separately. The component placement view shows external
circuit components used for the 926.5 MHz application circuit. Note: some traces must be cut to accommodate placement of
components; likewise some traces must be shorted. The voltage controlled oscillator is shown with the varactor referenced to
V
ground. The PCB is modified as shown to do this.
EE
16:1 broadband impedance transformer is mini circuits part #TX16–R3T; it is in the leadless surface mount “TX” package.
Components L and C comprise a low pass filter used to provide narrowband matching at a given IF frequency. For example at
49 MHz C = 36 p and L = 330 nH.
The microstrip trace on the ground side of the PCB is intended for a microstrip resonator; it is cut free when using a lump
inductor as done above.
7
MOTOROLA RF/IF DEVICE DATA
MC13142
Input Matching/Components
represents the application circuit. In the cascaded noise
analysis the system noise equation is:
Fsystem = F1 + [(F2 –1)/G1] + [(F3–1)]/[(G1)(G2)]
where:
It is desirable to use a RF ceramic or SAW filter before the
mixer to provide image frequency rejection. The filter is
selected based on cost, size and performance tradeoffs.
Typical RF filters have 3.0 to 5.0 dB insertion loss. The PC
board layout accommodates both ceramic and SAW RF
filters which are offered by various suppliers such as
Siemens, Toko and Murata.
Interface matching between the LNA, RF filter and the
mixer will be required. The interface matching networks
shown in the application circuit are designed for 50 Ω
interfaces.
F1 = the Noise Factor of the MC13142 LNA
G1 = the Gain of the LNA
F2 = the Noise factor of the RF Ceramic Filter
G2 = the Gain of the Ceramic Filter
F3 = the Noise factor of the Mixer
Note: the above terms are defined as linear relationships and
are related to the log form for gain and noise figure by the
following:
In the application circuit, the LNA is conjugately matched
–1
–1
F = Log
G = Log
[(NF in dB)/10] and similarly
[(Gain in dB)/10].
to 50 Ω input and output for 3.0 to 5.0 Vdc V . 17 dB gain
CC
and 1.8 dB noise figure is typical at 926 MHz. The mixer
measures 0 dB gain and 12 dB noise figure as shown in the
application circuit. Typical insertion loss of the Toko ceramic
filter is 3.0 dB. Thus, the overall gain of the frontend receiver
is 14 dB with a 3.3 dB noise figure.
Calculating in terms of gain and noise factor yields the
following:
F1 = 1.51; G1 = 50.11
F2 = 1.99; G2 = 0.5
F3 = 15.85
System Noise Considerations
Thus, substituting in the equation for system noise factor:
Fsystem = 2.12; NFsystem = 3.3 dB
The block diagram shows the cascaded noise stages of
the MC13142 in the frontend receiver subsystem; it
Figure 3. Frontend Subsystem Block Diagram for Noise Analysis
V
CC
f
= 926.5 MHz
LNA
RF
Z Transformer
16:1
330 nH
Mixer
NF Meter
Toko Ceramic
Filter
Noise Source
IF Output
36 p
f
= 49.05 MHz
IF
G2 = –3.0 dB
NF2 = 3.0 dB
G1 = 17 dB
NF1 = 1.8 dB
Local Oscillator
= 975.55 MHz
G3 = 0 dB
NF3 = 12 dB
f
LO
G
= 14 dB
sys
NF
= 3.3 dB
sys
8
MOTOROLA RF/IF DEVICE DATA
MC13142
Figure 4. Circuit Side View
MC13142D
Rev A
Mix In
Mix Lin
Cont
IF
Out
LNA
Output
LNA
Input
V
LO Buf
Out
CC
V–Cont
NOTES: Critical dimensions are 50 mil centers lead to lead in SO–16 footprint.
V
CC
Also line widths to labeled ports excluding V
FR4 PCB, 1/32 inch.
are 50 mil (0.050 inch).
CC
9
MOTOROLA RF/IF DEVICE DATA
MC13142
Figure 5. Ground Side View
V–Cont
V
CC
LNA
Input
LO
Buf Out
LNA
Output
IF
Out
Mix Lin
Cont
Mix In
MC13142D
Rev A
NOTES: FR4 PCB, 1/32 inch.
1.9 GHz FRONT–END FOR WIRELESS SYSTEMS
This application is applicable to both Analog and Digital
systems. With the correct VCO tuning and the appropriate
filter, it will do the front–end for DECT, PHS or PCS. The
MC13142D is available in a SOIC 16 pin package. The part
requires minimal external components, leading to a low cost
system. A circuit board layout with a circuit diagram to
evaluate the IC is shown. Except for the PLL control, all the
wireless systems front–ends will look the same and have the
same basic performance characteristic as the test circuit.
centering the tuning range, adjustment is accomplished by
moving the loop toward or away from some conductive
surface such as a ground plane.
The dielectric filter is referenced to the dc supply which
lessen the parts count and adds distributive capacity for high
frequency bypassing. DC feed to the LNA is through a low
value resistor (220 to 330 Ω) tapped at the filter input, so as not
to load the circuit unnecessarily. There is a small voltage drop
across the resistor, as well as some signal loss. The signal
loss is about 0.73 dB for a 220 Ω resistor and less for larger
values. If one can not afford the voltage drop, an inductor
could replace the resistor at a somewhat increased cost.
Circuit Operation:
LNA Input/Output
An LC filter is incorporated before the LNA to provide
some selectivity. In addition to selectivity, its other function is
to match the antenna impedance (50 Ω) to the LNA input for
best gain and sensitivity (low noise figure). The network
reflects about a 200 Ω source impedance to the device.
The output circuit is a pie network consisting of; the LNA
output capacity, the inductance (the bond wire, package pin
and L2), and the input capacity of the dielectric filter, along
with some added shunt. A 2.4 pF with Toko 4DFA 2 pole filter.
The 2.4 pF is for matching the in–band filter impedance to the
LNA output and has little effect on tuning.
Mixer
Looking from the dielectric filter’s output, the Mixer input is
50 Ω in series with an inductor. This inductor consists of the
printed circuit run, the package pin and bond wire, all in
series. It is modified, to some extent, by the package pin
distributive capacity, but overall at the bandpass frequency
remains inductive. Matching the filter impedance to the Mixer
input only requires a capacitor with a value that, when placed
in series, will resonate with this inductor at the filter bandpass
frequency.
The single–ended input signal is converted internally into
balanced current signals. The two signals drive the two low
impedance inputs (emitters) of a Gilbert Cell. They appear as
Both networks are tuned to band center by adjusting L1
and L2. L1 and L2, as well as L3, are short length of wire
formed in a half loop. Once the correct length is determined in
10
MOTOROLA RF/IF DEVICE DATA
MC13142
current sources to the Cell and can be programmed (via
In this application, the VCO is not required to cover a large
tuning range. Limiting the tuning range to no more than is
required to cover the band (making allowance for
temperature and aging effects) will result in a VCO less
susceptible to on board noise sources. To assure oscillation
while controlling the tuning range the varactor (plus series
capacitor) minimum capacity is chosen to be about equal to
the capacity from Pin 5 (transistor base) to RF ground. The
maximum tuning ratio could be no greater than 1.41 because
the circuit capacity could only double whatever the upper
value capacity the varactor attained. An upper limit on the
varactor capacity along with the effects of the series
capacitor reduces the VCO tuning range to about 1.2 times.
The varactors chosen for the test fixtures were Loral KV2111.
The VCO buffer, as most emitter follower circuits, has the
potential of generating a parasitic oscillation. When a
collector is RF bypassed, a tuned LC circuit is formed
consisting of the bypass capacitor, bond wire plus package
pin inductance and the device effective output capacity. If the
base is low impedance, there is normally enough distributive
collector to emitter capacity for the device to oscillate in the
common base mode. A simple fix without affecting the buffer
otherwise, is to place a small value series resistor in the
collector lead. This will lower the Q of the circuit where it
cannot sustain oscillations. Without the series resistor at Pin 8
or some other damping element, the buffer will oscillate.
Pin 15) for more current. The current is often adjusted for
minimum third order response. In this Fixture it is fixed biased
for most conversion gain.
The Mixer circuit is balanced where both oscillator and RF
are suppressed. This provides IF signals at Pins 9 and 10
which are equal in amplitude and 180 degrees out of phase.
To realize a positive gain one needs to reflect a higher
impedance from the load impedance (50 Ω for this fixture) to
the Mixer output or outputs. Maximum signal transfer would
require a balance to unbalance network. Center tapped
tuned transformers can perform this function but are quite
expensive. If one can afford 3.0 dB less signal, a simple LC
circuit at one of the outputs will work well. The other output is
unused and bypassed to ground.
The most gain is realized when no shunt capacity is added
and L4 is selected to resonate with the terminal capacity.
Adding shunt capacity will lower the gain and increase the
circuit’s bandwidth. A small value series capacitor C4 to the
50 Ω output will control the reflected impedance and
complete the circuit. L4 and C4 will vary in value depending
on the IF frequency.
VCO
The base of the device is the source for driving both the
Gilbert cell and prescaler buffer stages. Because of this, the
oscillator device will operate and drive the Mixer only in the
grounded collector configuration. Additional dc bias is added
through a 1.3 kΩ resistor (tapped for minimum VCO loading)
to reduce the off–set between base and supply.
The external circuit is a modified Colpitts where the
capacitance between base and emitter (Pins 4 and 5), along
with a capacitor from emitter to ac ground, forms the circuit
capacity and the feedback that sustains oscillations. The
effective circuit inductance (looking from the top of the circuit,
the transistor base) consist of L3 in series with varactor diode
D1 and a blocking capacitor. This circuit must appear
inductive for the VCO to operate properly. If the capacity is
too small, the feedback ratio is reduced and the VCO can
cease oscillating. When it becomes to large, it will not vary
the frequency due to the limiting effect of the series loop
capacitance.
PLL
A phase lock loop is added to the test board to evaluate
the VCO. The MC12179 multiplies the crystal reference
frequency by 256 to obtain lock. In a frequency agile system,
the MC12210 would control the VCO and its reference
derived from a crystal. The crystal frequency would be
selected to coincide with the required VCO frequencies and
channels spacing requirements.
Expected Performance
As stated earlier, the MC13142 performance in any of the
systems should mirror the performance obtained in the test
fixture. Fixture power gains of 15 dBm and noise figures of
5.5 dB are typical. The Mixer current can be varied to
enhances battery life as well as alter its output characteristic
for peak performance of a desired or undesired response.
11
MOTOROLA RF/IF DEVICE DATA
MC13142
Figure 6. 1.9 GHz Circuit Component Placement View
Dielectric
Filter
BH 8/97
V
CC
IF Output
1.3 k
2.4 p
330
18
2.4 p
270 p
L4
3.9 k
L2
C4
1.0 nF
RF Input
68
68
0.7 p
0.5 p
0.7 p
0.5 p
L1
C
L3
CV1
0.03
2.4 p
0.006
82 p
MC13142 Test
Circuit
1.3 k
1.3 k
C
0.1
µ
68
PLL Loop Filter
MC12179
Rev 3
12
MOTOROLA RF/IF DEVICE DATA
MC13142
Figure 7. 1.9 GHz Application Circuit
Dielectric
Filter
V
1.3 k
CC
4.7 µF
2.4 p
270 p
18
330
2.4 p
L2
10 nH
L4
C4
1.0 nF
3.9 k
IF
MC13142
0.1 µF
2.4 p
68
18
MC12179
68
0.5 p
0.7 p
50 k
0.5 p
V
CC
0.1
µF
0.7 p
RF
L1
4.0 nH
KV2111
1.3 k
0.1 µF
VCO Control
L3
8.0 nH
2.4 p
0.03
820
10 k
82
0.0062
1.3 k
VCO Loop Filter
13
MOTOROLA RF/IF DEVICE DATA
MC13142
OUTLINE DIMENSIONS
D SUFFIX
PLASTIC PACKAGE
CASE 751B–05
(SO–16)
ISSUE J
–A–
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: MILLIMETER.
3. DIMENSIONS A AND B DO NOT INCLUDE
MOLD PROTRUSION.
16
1
9
8
–B–
P 8 PL
4. MAXIMUM MOLD PROTRUSION 0.15 (0.006)
PER SIDE.
M
S
0.25 (0.010)
B
5. DIMENSION D DOES NOT INCLUDE DAMBAR
PROTRUSION. ALLOWABLE DAMBAR
PROTRUSION SHALL BE 0.127 (0.005) TOTAL
IN EXCESS OF THE D DIMENSION AT
MAXIMUM MATERIAL CONDITION.
G
MILLIMETERS
INCHES
DIM
A
B
C
D
MIN
9.80
3.80
1.35
0.35
0.40
MAX
10.00
4.00
1.75
0.49
1.25
MIN
MAX
0.393
0.157
0.068
0.019
0.049
F
0.386
0.150
0.054
0.014
0.016
R X 45
K
C
F
G
J
K
M
P
R
1.27 BSC
0.050 BSC
–T–
SEATING
PLANE
0.19
0.10
0
0.25
0.25
7
0.008
0.004
0
0.009
0.009
7
J
M
D
16 PL
5.80
0.25
6.20
0.50
0.229
0.010
0.244
0.019
M
S
S
0.25 (0.010)
T
B
A
14
MOTOROLA RF/IF DEVICE DATA
MC13142
Motorola reserves the right to make changes without further notice to any products herein. Motorola makes no warranty, representation or guarantee regarding
the suitability of its products for any particular purpose, nor does Motorola assume any liability arising out of the application or use of any product or circuit, and
specificallydisclaims any and all liability, including without limitation consequential or incidental damages. “Typical” parameters which may be provided in Motorola
datasheetsand/orspecificationscananddovaryindifferentapplicationsandactualperformancemayvaryovertime. Alloperatingparameters,including“Typicals”
must be validated for each customer application by customer’s technical experts. Motorola does not convey any license under its patent rights nor the rights of
others. Motorola products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other
applicationsintended to support or sustain life, or for any other application in which the failure of the Motorola product could create a situation where personal injury
ordeathmayoccur. ShouldBuyerpurchaseoruseMotorolaproductsforanysuchunintendedorunauthorizedapplication,BuyershallindemnifyandholdMotorola
and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees
arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that
Motorola was negligent regarding the design or manufacture of the part. Motorola and
Opportunity/Affirmative Action Employer.
are registered trademarks of Motorola, Inc. Motorola, Inc. is an Equal
15
MOTOROLA RF/IF DEVICE DATA
MC13142
Mfax is a trademark of Motorola, Inc.
JAPAN: Nippon Motorola Ltd.: SPD, Strategic Planning Office, 141,
How to reach us:
USA/EUROPE/Locations Not Listed: Motorola Literature Distribution;
P.O. Box 5405, Denver, Colorado 80217. 1–303–675–2140 or 1–800–441–2447 4–32–1 Nishi–Gotanda, Shagawa–ku, Tokyo, Japan. 03–5487–8488
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Mfax : RMFAX0@email.sps.mot.com – TOUCHTONE 1–602–244–6609
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– http://sps.motorola.com/mfax/
HOME PAGE: http://motorola.com/sps/
MC13142/D
◊
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