LM359M [NSC]
Dual, High Speed, Programmable, Current Mode Norton Amplifiers; 双通道，高速，可编程，电流模式诺顿放大器
| 型号： | LM359M |
| 厂家： | 
National Semiconductor |
| 描述： | Dual, High Speed, Programmable, Current Mode Norton Amplifiers |
| 文件： | 总24页 (文件大小：744K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
October 1998
LM359
Dual, High Speed, Programmable, Current Mode (Norton)
Amplifiers
General Description
Features
n User programmable gain bandwidth product, slew rate,
input bias current, output stage biasing current and total
device power dissipation
The LM359 consists of two current differencing (Norton) in-
put amplifiers. Design emphasis has been placed on obtain-
ing high frequency performance and providing user program-
mable amplifier operating characteristics. Each amplifier is
broadbanded to provide a high gain bandwidth product, fast
slew rate and stable operation for an inverting closed loop
gain of 10 or greater. Pins for additional external frequency
compensation are provided. The amplifiers are designed to
operate from a single supply and can accommodate input
common-mode voltages greater than the supply.
=
n High gain bandwidth product (ISET 0.5 mA)
400 MHz for AV 10 to 100
30 MHz for AV
n High slew rate (ISET 0.5 mA)
60 V/µs for AV 10 to 100
30 V/µs for AV
n Current differencing inputs allow high common-mode
input voltages
=
=
1
=
=
=
1
Applications
n General purpose video amplifiers
n High frequency, high Q active filters
n Photo-diode amplifiers
n Operates from a single 5V to 22V supply
n Large inverting amplifier output swing, 2 mV to VCC
2V
−
>
for f 1 kHz
n Low spot noise,
n Wide frequency range waveform generation circuits
n All LM3900 AC applications work to much higher
frequencies
Typical Application
Connection Diagram
Dual-In-Line Package
DS007788-2
DS007788-1
Top View
=
Order Number LM359J, LM359M or LM359N
See NS Package Number J14A, M14A or N14A
•
•
•
•
AV 20 dB
=
−3 dB bandwidth 2.5 Hz to 25 MHz
<
Differential phase error 1˚ at 3.58 MHz
<
Differential gain error 0.5
% at 3.58 MHz
© 1999 National Semiconductor Corporation
DS007788
www.national.com
Absolute Maximum Ratings (Note 1)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
Input Currents, IIN(+) or IIN(−)
Set Currents, ISET(IN) or ISET(OUT)
Operating Temperature Range
LM359
10 mADC
2 mADC
0˚C to +70˚C
Storage Temperature Range
Lead Temperature
−65˚C to +150˚C
Supply Voltage
22 VDC
±
or 11 VDC
(Soldering, 10 sec.)
260˚C
260˚C
Power Dissipation (Note 2)
J Package
Soldering Information
Dual-In-Line Package
Soldering (10 sec.)
1W
N Package
750 mW
Maximum TJ
Small Outline Package
Vapor Phase (60 sec.)
Infrared (15 sec.)
J Package
+150˚C
+125˚C
215˚C
220˚C
N Package
Thermal Resistance
J Package
See AN-450 “Surface Mounting Methods and Their Effect
on Product Reliability” for other methods of soldering
surface mount devices.
θjA 147˚C/W still air
110˚C/W with 400 linear feet/min air flow
ESD rating to be determined.
N Package
θjA 100˚C/W still air
75˚C/W with 400 linear feet/min air flow
Electrical Characteristics
=
=
=
=
ISET(IN) ISET(OUT) 0.5 mA, Vsupply 12V, TA 25˚C unless otherwise noted
Parameter
Conditions
LM359
Typ
72
Units
Min
Max
=
=
=
Open Loop Voltage
Gain
Vsupply 12V, RL 1k, f 100 Hz
62
dB
dB
=
TA 125˚C
68
=
=
Bandwidth
RIN 1 kΩ, Ccomp 10 pF
15
30
MHz
Unity Gain
=
Gain Bandwidth Product
Gain of 10 to 100
Slew Rate
RIN 50Ω to 200Ω
200
400
MHz
=
=
Unity Gain
RIN 1 kΩ, Ccomp 10 pF
30
60
V/µs
V/µs
dB
<
RIN 200Ω
Gain of 10 to 100
Amplifier to Amplifier
Coupling
=
=
f
100 Hz to 100 kHz, RL 1k
−80
=
=
Mirror Gain
at 2 mA IIN(+), ISET 5 µA, TA 25˚C
0.9
0.9
1.0
1.0
1.1
1.1
µA/µA
µA/µA
=
(Note 3)
at 0.2 mA IIN(+), ISET 5 µA
Over Temp.
=
at 20 µA IIN(+), ISET 5 µA
0.9
1.0
3
1.1
5
µA/µA
%
Over Temp.
∆Mirror Gain
(Note 3)
at 20 µA to 0.2 mA IIN(+)
=
Over Temp, ISET 5 µA
=
Input Bias Current
Inverting Input, TA 25˚C
8
15
30
µA
µA
kΩ
Ω
Over Temp.
Input Resistance (βre)
Output Resistance
Output Voltage Swing
VOUT High
Inverting Input
2.5
3.5
=
=
IOUT 15 mA rms, f 1 MHz
=
RL 600Ω
IIN(−) and IIN(+) Grounded
9.5
10.3
2
V
=
=
0
VOUT Low
IIN(−) 100 µA, IIN(+)
50
mV
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2
Electrical Characteristics (Continued)
=
=
=
=
ISET(IN) ISET(OUT) 0.5 mA, Vsupply 12V, TA 25˚C unless otherwise noted
Parameter Conditions
LM359
Typ
Units
Min
16
Max
Output Currents
=
Source
IIN(−) and IIN(+) Grounded, RL 100Ω
40
4.7
3
mA
mA
mA
=
=
=
Sink (Linear Region)
Sink (Overdriven)
V
comp−0.5V VOUT 1V, IIN(+) 0
=
=
I
IN(−) 100 µA, IIN(+) 0,
1.5
=
VOUT Force 1V
Supply Current
Non-Inverting Input
18.5
50
22
mA
dB
= ∞
Grounded, RL
=
Power Supply Rejection
(Note 4)
f
120 Hz, IIN(+) Grounded
40
Note 1: “Absolute Maximum Ratings” indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is
functional, but do not guarantee specific performance limits.
Note 2: See Maximum Power Dissipation graph.
Note 3: Mirror gain is the current gain of the current mirror which is used as the non-inverting input.
∆Mirror Gain is the % change in A for two different mirror currents at any given temperature.
I
Note 4: See Supply Rejection graphs.
3
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Schematic Diagram
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4
Typical Performance Characteristics
Open Loop Gain
Open Loop Gain
Open Loop Gain
DS007788-40
DS007788-39
DS007788-41
Note: Shaded area refers to LM359
Gain Bandwidth Product
Slew Rate
Gain and Phase
=
Feedback Gain − 100
DS007788-43
DS007788-42
DS007788-44
Inverting Input Bias Current
Inverting Input Bias Current
Mirror Gain
DS007788-47
DS007788-46
DS007788-45
Note: Shaded area refers to LM359
5
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Typical Performance Characteristics (Continued)
Mirror Gain
Mirror Gain
Mirror Current
DS007788-48
DS007788-50
DS007788-49
Note: Shaded area refers to LM359
Note: Shaded area refers to LM359
Supply Current
Supply Rejection
Supply Rejection
DS007788-53
DS007788-52
DS007788-51
Output Sink Current
Output Swing
Output Impedance
DS007788-54
DS007788-55
DS007788-56
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6
Typical Performance Characteristics (Continued)
Amplifier to Amplifier
Noise Voltage
Maximum Power Dissipation
Coupling (Input Referred)
DS007788-58
DS007788-59
DS007788-57
Note: Shaded area refers to LM359J/LM359N
Application Hints
The LM359 consists of two wide bandwidth, decompensated
current differencing (Norton) amplifiers. Although similar in
operation to the original LM3900, design emphasis for these
amplifiers has been placed on obtaining much higher fre-
quency performance as illustrated in Figure 1.
amount of current to flow into the inverting input . The mirror
gain (AI) specification is the measure of how closely these
two currents match. For more details see National Applica-
tion Note AN-72.
DC biasing of the output is accomplished by establishing a
reference DC current into the (+) input, IIN(+), and requiring
the output to provide the (−) input current. This forces the
output DC level to be whatever value necessary (within the
output voltage swing of the amplifier) to provide this DC ref-
erence current, Figure 2.
This significant improvement in frequency response is the
result of using a common-emitter/common-base (cascode)
gain stage which is typical in many discrete and integrated
video and RF circuit designs. Another versatile aspect of
these amplifiers is the ability to externally program many in-
ternal amplifier parameters to suit the requirements of a wide
variety of applications in which this type of amplifier can be
used.
DS007788-7
DS007788-6
FIGURE 1.
FIGURE 2.
DC BIASING
The DC input voltage at each input is a transistor VBE
The LM359 is intended for single supply voltage operation
which requires DC biasing of the output. The current mirror
circuitry which provides the non-inverting input for the ampli-
fier also facilitates DC biasing the output. The basic opera-
tion of this current mirror is that the current (both DC and AC)
flowing into the non-inverting input will force an equal
(
0.6 VDC) and must be considered for DC biasing. For
most applications, the supply voltage, V+, is suitable and
convenient for establishing IIN(+). The inverting input bias
current, Ib(−), is a direct function of the programmable input
stage current (see current programmability section) and to
obtain predictable output DC biasing set IIN(+) ≥ 10Ib(−).
7
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The nVBE biasing configuration is most useful for low noise
applications where a reduced input impedance can be ac-
commodated (see typical applications section).
Application Hints (Continued)
The following figures illustrate typical biasing schemes for
AC amplifiers using the LM359:
OPERATING CURRENT PROGRAMMABILITY (ISET
)
The input bias current, slew rate, gain bandwidth product,
output drive capability and total device power consumption
of both amplifiers can be simultaneously controlled and opti-
mized via the two programming pins ISET(OUT) and ISET(IN)
.
ISET(OUT)
The output set current (ISET(OUT)) is equal to the amount of
current sourced from pin 1 and establishes the class A bias-
ing current for the Darlington emitter follower output stage.
Using a single resistor from pin 1 to ground, as shown in Fig-
ure 6, this current is equal to:
DS007788-8
FIGURE 3. Biasing an Inverting AC Amplifier
DS007788-11
FIGURE 6. Establishing the Output Set Current
The output set current can be adjusted to optimize the
amount of current the output of the amplifier can sink to drive
load capacitance and for loads connected to V+. The maxi-
mum output sinking current is approximately 10 times
ISET(OUT) . This set current is best used to reduce the total
device supply current if the amplifiers are not required to
drive small load impedances.
DS007788-9
ISET(IN)
The input set current ISET(IN) is equal to the current flowing
into pin 8. A resistor from pin 8 to V+ sets this current to be:
FIGURE 4. Biasing a Non-Inverting AC Amplifier
DS007788-12
DS007788-10
FIGURE 7. Establishing the Input Set Current
ISET(IN) is most significant in controlling the AC characteris-
tics of the LM359 as it directly sets the total input stage cur-
rent of the amplifiers which determines the maximum slew
rate, the frequency of the open loop dominant pole, the input
resistance of the (−) input and the biasing current Ib(−). All of
FIGURE 5. nVBE Biasing
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8
One method to avoid this is to use an adjustable current
source which has voltage compliance to generate the set
current as shown in Figure 9.
Application Hints (Continued)
these parameters are significant in wide band amplifier de-
sign. The input stage current is approximately
SET(IN) and by using this relationship the following first order
approximations for these AC parameters are:
3 times
I
DS007788-14
where Ccomp is the total capacitance from the compensation
pin (pin 3 or pin 13) to ground, AVOL is the low frequency
open loop voltage gain in V/V and an ambient temperature of
FIGURE 9. Current Source Programming of ISET
=
=
25˚C is assumed (KT/q
26 mV and βtyp
150). ISET(IN)
also controls the DC input bias current by the expression:
This circuit allows ISET to remain constant over the entire
supply voltage range of the LM359 which also improves
power supply ripple rejection as illustrated in the Typical Per-
formance Characteristics. It should be noted, however, that
the current through the LM334 as shown will change linearly
with temperature but this can be compensated for (see
LM334 data sheet).
which is important for DC biasing considerations.
The total device supply current (for both amplifiers) is also a
direct function of the set currents and can be approximated
by:
Pin 1 must never be shorted to ground or pin 8 never shorted
to V+ without limiting the current to 2 mA or less to prevent
catastrophic device failure.
Isupply 27 x ISET(OUT) + 11 x ISET(IN)
with each set current programmed by individual resistors.
CONSIDERATIONS FOR HIGH FREQUENCY
OPERATION
PROGRAMMING WITH A SINGLE RESISTOR
The LM359 is intended for use in relatively high frequency
applications and many factors external to the amplifier itself
must be considered. Minimization of stray capacitances and
their effect on circuit operation are the primary requirements.
The following list contains some general guidelines to help
accomplish this end:
Operating current programming may also be accomplished
using only one resistor by letting ISET(IN) equal ISET(OUT). The
programming current is now referred to as ISET and it is cre-
ated by connecting a resistor from pin 1 to pin 8 (Figure 8).
1. Keep the leads of all external components as short as
possible.
2. Place components conducting signal current from the
output of an amplifier away from that amplifier’s
non-inverting input.
3. Use reasonably low value resistances for gain setting
and biasing.
4. Use of a ground plane is helpful in providing a shielding
effect between the inputs and from input to output. Avoid
using vector boards.
DS007788-13
5. Use a single-point ground and single-point supply distri-
bution to minimize crosstalk. Always connect the two
grounds (one from each amplifier) together.
=
=
I
SET
I
I
SET(IN)
SET(OUT)
FIGURE 8. Single Resistor Programming of ISET
>
6. Avoid use of long wires ( 2") but if necessary, use
This configuration does not affect any of the internal set cur-
rent dependent parameters differently than previously dis-
cussed except the total supply current which is now equal to:
shielded wire.
7. Bypass the supply close to the device with a low induc-
tance, low value capacitor (typically a 0.01 µF ceramic)
to create a good high frequency ground. If long supply
Isupply 37 x ISET
z
leads are unavoidable, a small resistor ( 10Ω) in series
Care must be taken when using resistors to program the set
current to prevent significantly increasing the supply voltage
above the value used to determine the set current. This
would cause an increase in total supply current due to the re-
sulting increase in set current and the maximum device
power dissipation could be exceeded. The set resistor val-
ue(s) should be adjusted for the new supply voltage.
with the bypass capacitor may be needed and using
shielded wire for the supply leads is also recommended.
COMPENSATION
The LM359 is internally compensated for stability with closed
loop inverting gains of 10 or more. For an inverting gain of
less than 10 and all non-inverting amplifiers (the amplifier al-
ways has 100% negative current feedback regardless of the
9
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cuits which is the effective series inductance (ESL) of the
coupling capacitor which creates an increase in the imped-
ance of the capacitor at high frequencies and can cause an
unexpected gain reduction. Low ESL capacitors like solid
tantalum for large values of C and ceramic for smaller values
are recommended. A parallel combination of the two types is
even better for gain accuracy over a wide frequency range.
Application Hints (Continued)
gain in the non-inverting configuration) some external fre-
quency compensation is required because the stray capaci-
tance to ground from the (−) input and the feedback resistor
add additional lagging phase within the feedback loop. The
value of the input capacitance will typically be in the range of
6 pF to 10 pF for a reasonably constructed circuit board.
When using a feedback resistance of 30 kΩ or less, the best
method of compensation, without sacrificing slew rate, is to
add a lead capacitor in parallel with the feedback resistor
with a value on the order of 1 pF to 5 pF as shown in Figure
10 .
AMPLIFIER DESIGN EXAMPLES
The ability of the LM359 to provide gain at frequencies
higher than most monolithic amplifiers can provide makes it
most useful as a basic broadband amplification stage. The
design of standard inverting and non-inverting amplifiers,
though different than standard op amp design due to the cur-
rent differencing inputs, also entail subtle design differences
between the two types of amplifiers. These differences will
be best illustrated by design examples. For these examples
a practical video amplifier with a passband of 8 Hz to 10 MHz
and a gain of 20 dB will be used. It will be assumed that the
input will come from a 75Ω source and proper signal termi-
nation will be considered. The supply voltage is 12 VDC and
single resistor programming of the operating current, ISET
will be used for simplicity.
,
AN INVERTING VIDEO AMPLIFIER
1. Basic circuit configuration:
DS007788-15
=
C
f
1 pF to 5 pF for stability
FIGURE 10. Best Method of Compensation
Another method of compensation is to increase the effective
value of the internal compensation capacitor by adding ca-
pacitance from the COMP pin of an amplifier to ground. An
external 20 pF capacitor will generally compensate for all
gain settings but will also reduce the gain bandwidth product
and the slew rate. These same results can also be obtained
by reducing ISET(IN) if the full capabilities of the amplifier are
not required. This method is termed over-compensation.
DS007788-17
Another area of concern from a stability standpoint is that of
capacitive loading. The amplifier will generally drive capaci-
tive loads up to 100 pF without oscillation problems. Any
larger C loads can be isolated from the output as shown in
Figure 11. Over-compensation of the amplifier can also be
used if the corresponding reduction of the GBW product can
be afforded.
2. Determine the required ISET from the characteristic
curves for gain bandwidth product.
=
=
GBWMIN 10 x 10 MHz 100 MHz
For a flat response to 10 MHz a closed loop response to two
octaves above 10 MHz (40 MHz) will be sufficient.
=
=
Actual GBW 10 x 40 MHz 400 MHz
=
ISET required 0.5 mA
DS007788-16
FIGURE 11. Isolating Large Capacitive Loads
In most applications using the LM359, the input signal will be
AC coupled so as not to affect the DC biasing of the ampli-
fier. This gives rise to another subtlety of high frequency cir-
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10
Application Hints (Continued)
Final Circuit Using Standard 5%
Tolerance Resistor Values:
3. Determine maximum value for Rf to provide stable DC
biasing
Optimum output DC level for maximum symmetrical swing
without clipping is:
Rf(MAX) can now be found:
DS007788-18
Circuit Performance:
This value should not be exceeded for predictable DC bias-
ing.
4. Select Rs to be large enough so as not to appreciably
load the input termination resistance:
=
Rs ≥ 750Ω; Let Rs 750Ω
5. Select Rf for appropriate gain:
7.5 kΩ is less than the calculated Rf(MAX) so DC predictability
is insured.
=
6. Since Rf 7.5k, for the output to be biased to 5.1 VDC
,
the reference current IIN(+) must be:
DS007788-19
=
V
o(DC)
5.1V
<
Differential phase error 1˚ for 3.58 MHz f
IN
% for 3.58 MHz f
Now Rb can be found by:
<
Differential gain error 0.5
IN
=
low 2.5 Hz
f
−3 dB
A NON-INVERTING VIDEO AMPLIFIER
For this case several design considerations must be dealt
with.
7. Select Ci to provide the proper gain for the 8 Hz mini-
mum input frequency:
•
The output voltage (AC and DC) is strictly a function of
the size of the feedback resistor and the sum of AC and
DC “mirror current” flowing into the (+) input.
•
The amplifier always has 100
% current feedback so ex-
ternal compensation is required. Add a small (1 pF–5 pF)
feedback capacitance to leave the amplifier’s open loop
response and slew rate unaffected.
A larger value of Ci will allow a flat frequency response down
to 8 Hz and a 0.01 µF ceramic capacitor in parallel with Ci
will maintain high frequency gain accuracy.
•
•
To prevent saturating the mirror stage the total AC and
DC current flowing into the amplifier’s (+) input should be
less than 2 mA.
8. Test for peaking of the frequency response and add a
feedback “lead” capacitor to compensate if necessary.
The output’s maximum negative swing is one diode
above ground due to the VBE diode clamp at the (−) input.
11
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Application Hints (Continued)
DESIGN EXAMPLE:
=
=
eIN
50 mV (MAX), fIN
10 MHz (MAX), desired circuit
=
=
=
BW
20 MHz, AV
=
20 dB, driving source impedance
Also, for a closed loop gain of +10, Rf must be 10 times Rs
+ re where re is the mirror diode resistance.
75Ω, V+ 12V.
1. Basic circuit configuration:
4. So as not to appreciably load the 75Ω input termination
resistance the value of (Rs + re) is set to 750Ω.
=
5. For Av 10; Rf is set to 7.5 kΩ.
6. The optimum output DC level for symmetrical AC swing
is:
7. The DC feedback current must be:
DC biasing predictability will be insured because 640 µA is
greater than the minimum of ISET/5 or 100 µA.
For gain accuracy the total AC and DC mirror current should
be less than 2 mA. For this example the maximum AC mirror
current will be:
DS007788-20
2. Select ISET to provide adequate amplifier bandwidth so
that the closed loop bandwidth will be determined by Rf
and Cf. To do this, the set current should program an
amplifier open loop gain of at least 20 dB at the desired
closed loop bandwidth of the circuit. For this example,
an ISET of 0.5 mA will provide 26 dB of open loop gain at
20 MHz which will be sufficient. Using single resistor pro-
therefore the total mirror current range will be 574 µA to 706
µA which will insure gain accuracy.
8. Rb can now be found:
gramming for ISET
:
9. Since Rs + re will be 750Ω and re is fixed by the DC mir-
3. Since the closed loop bandwidth will be determined by
ror current to be:
to obtain a 20 MHz bandwidth, both Rf and Cf should be kept
small. It can be assumed that Cf can be in the range of 1 pF
to 5 pF for carefully constructed circuit boards to insure sta-
bility and allow a flat frequency response. This will limit the
value of Rf to be within the range of:
Rs must be 750Ω–40Ω or 710Ω which can be a 680Ω resis-
tor in series with a 30Ω resistor which are standard 5% toler-
ance resistor values.
10. As a final design step, Ci must be selected to pass the
lower passband frequency corner of 8 Hz for this ex-
ample.
A larger value may be used and a 0.01 µF ceramic capacitor
in parallel with Ci will maintain high frequency gain accuracy.
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12
Application Hints (Continued)
Final Circuit Using Standard 5% Tolerance Resistor Values
DS007788-21
Circuit Performance
DS007788-22
=
V
o(DC)
5.4V
<
Differential phase error 0.5˚
<
Differential gain error 2%
=
low 2.5 Hz
f
−3 dB
13
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The total device power dissipation must always be kept in
mind when selecting an operating supply voltage, the pro-
gramming current, ISET, and the load resistance, particularly
when DC coupling the output to a succeeding stage. To pre-
vent damaging the current mirror input diode, the mirror cur-
rent should always be limited to 10 mA, or less, which is im-
portant if the input is susceptible to high voltage transients.
The voltage at any of the inputs must not be forced more
negative than −0.7V without limiting the current to 10 mA.
Application Hints (Continued)
GENERAL PRECAUTIONS
The LM359 is designed primarily for single supply operation
but split supplies may be used if the negative supply voltage
is well regulated as the amplifiers have no negative supply
rejection.
The supply voltage must never be reversed to the device;
however, plugging the device into a socket backwards would
then connect the positive supply voltage to the pin that has
no internal connection (pin 5) which may prevent inadvertent
device failure.
Typical Applications
DC Coupled Inputs
Inverting
DS007788-23
Non-Inverting
DS007788-24
•
•
Eliminates the need for an input coupling capacitor
Input DC level must be stable and can exceed the supply voltage of the LM359 provided that maximum input currents are not
exceeded.
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14
Typical Applications (Continued)
Noise Reduction using nVBE Biasing
nVBE Biasing with a Negative Supply
DS007788-25
DS007788-26
•
R1 and C2 provide additional filtering of the negative bi-
asing supply
Typical Input Referred Noise Performance
Adding a JFET Input Stage
DS007788-27
DS007788-28
•
•
•
•
•
FET input voltage mode op amp
=
=
=
=
For AV +1; BW 40 MHz, Sr 60 V/µs; CC 51 pF
=
=
=
=
For AV +11; BW 24 MHz, Sr 130 V/µs; CC 5 pF
=
=
=
=
For AV +100; BW 4.5 MHz, Sr 150 V/µs; CC 2 pF
<
VOS is typically 25 mV; 100Ω potentiometer allows a
±
VOS adjust range of ≈ 200 mV
•
Inputs must be DC biased for single supply operation
15
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Typical Applications (Continued)
Photo Diode Amplifier
DS007788-29
z
D1
RCA N-Type Silicon P-I-N Photodiode
•
Frequency response of greater than 10 MHz
•
If slow rise and fall times can be tolerated the gate on the output can be removed. In this case the rise and the fall time of the
LM359 is 40 ns.
2
=
=
•
TPDL 45 ns, TPDH 50 ns − T L output
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Typical Applications (Continued)
Balanced Line Driver
DS007788-30
•
•
•
•
1 MHz−3 dB bandwidth with gain of 10 and 0 dbm into 600Ω
% distortion at full bandwidth; reduced to 0.05% with bandwidth of 10 kHz
0.3
=
=
±
Will drive CL 1500 pF with no additional compensation, 0.01 µF with Ccomp 180 pF
70 dB signal to noise ratio at 0 dbm into 600Ω, 10 kHz bandwidth
17
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Typical Applications (Continued)
Difference Amplifier
Voltage Controlled Oscillator
DS007788-31
DS007788-32
•
CMRR is adjusted for max at expected CM input signal
•
•
5 MHz operation
T2L output
•
•
•
Wide bandwidth
70 dB CMRR typ
Wide CM input voltage range
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18
Typical Applications (Continued)
Phase Locked Loop
DS007788-33
•
•
Up to 5 MHz operation
T2L compatible input
=
All diodes 1N914
19
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Typical Applications (Continued)
Squarewave Generator
DS007788-34
=
f
1 MHz
Output is TTL compatible
!
Frequency is adjusted by R1 & C (R1
R2)
Pulse Generator
DS007788-36
Output is TTL compatible
Duty cycle is adjusted by R1
Frequency is adjusted by C
=
f
1 MHz
=
Duty cycle 20%
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20
Typical Applications (Continued)
Crystal Controlled Sinewave Oscillator
DS007788-37
=
V
f
500 mVp-p
o
=
9.1 MHz
<
THD 2.5%
High Performance 2 Amplifier Biquad Filter(s)
DS007788-35
•
•
•
The high speed of the LM359 allows the center frequency Qo product of the filter to be: fox Qo ≤ 5 MHz
The above filter(s) maintain performance over wide temperature range
One half of LM359 acts as a true non-inverting integrator so only 2 amplifiers (instead of 3 or 4) are needed for the biquad filter
structure
21
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Typical Applications (Continued)
DC Biasing Equations for V01(DC) V02(DC) V+/2
Type I
Type II
Type III
Analysis and Design Equations
Type
VO1
BP
VO2
LP
BP
—
Ci
O
Ri2
Ri2
∞
Ri1
∞
fo
Qo
fZ(notch)
Ho(LP)
R/Ri2
—
Ho(BP)
RQ/Ri2
Ho(HP)
—
Ho(BR)
—
I
II
RQ/R
RQ/R
RQ/R
—
—
∞
HP
Ci
Ci
RQCi/RC
Ci/C
—
∞
III
Notch/
BR
Ri1
—
—
—
Triangle Waveform Generator
DS007788-38
V2 output is TTL compatible
R2 adjusts for symmetry of the triangle waveform
Frequency is adjusted with R5 and C
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22
Physical Dimensions inches (millimeters) unless otherwise noted
Ceramic Dual-In-Line Package (J)
Order Number LM359J
NS Package Number J14A
S.O. Package (M)
Order Number LM359M
NS Package Number M14A
23
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Physical Dimensions inches (millimeters) unless otherwise noted (Continued)
Molded Dual-In-Line Package (N)
Order Number LM359N
NS Package Number N14A
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