SA57250-20GW-G [NXP]
IC 0.3 A SWITCHING REGULATOR, 57.5 kHz SWITCHING FREQ-MAX, PDSO5, 1.50 MM, PLASTIC, MO-178, SOT-25, SOT-23, SOP-5, Switching Regulator or Controller;
| 型号： | SA57250-20GW-G |
| 厂家： | 
NXP |
| 描述： | IC 0.3 A SWITCHING REGULATOR, 57.5 kHz SWITCHING FREQ-MAX, PDSO5, 1.50 MM, PLASTIC, MO-178, SOT-25, SOT-23, SOP-5, Switching Regulator or Controller 开关 光电二极管 |
| 文件： | 总18页 (文件大小：276K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
INTEGRATED CIRCUITS
SA57250-XX
CMOS switching regulator
(PWM controlled)
Product data
2003 Nov 12
Supersedes data of 2001 Aug 01
Philips
Semiconductors
Philips Semiconductors
Product data
CMOS switching regulator (PWM controlled)
SA57250-XX
GENERAL DESCRIPTION
The SA57250-XX is a fully integrated DC/DC converter circuit built
with CMOS technology. It uses a pulse width modulation (PWM)
controlled switching regulator circuit for low ripple and high efficiency
(typically 83%). The regulator has a high precision output with
±2.4% accuracy. Few external components are required.
The SA57250 has an ON/OFF control that may be used to turn off
the converter and conserve power in battery powered applications.
Power off supply current is no more than 0.5 µA. A built-in, soft-start
circuit reduces current inrush and voltage overshoot during start up.
A low resistance CMOS power FET, with low leakage current and
low parasitic capacitance, is on-chip.
FEATURES
• Operates from 0.7 to 9 V
APPLICATIONS
• Mobile and portable phones
DC
• Ultra low operating supply current—typically 17 µA
• ON/OFF power control
• Instrumentation and industrial products
• Other portable, battery-operated equipment
• Built-in power FET
• High efficiency—typically 83%
• High precision output—typically ±2.4%
• Operating temperature range of –40 to +85 °C
• Available output voltages: 2.0, 2.5, 2.8, 3.0, 3.3, 3.6, 5.0 V
DC
• Available in a 5-lead small outline surface mount package
(SOP003)
SIMPLIFIED SYSTEM DIAGRAM
V
OUT
SW
FB
2
5
V
BATT
V
DD
V
SA57250-XX
REF
R
R
PWM CONTROL
GND
4
SOFT START
1
ON/OFF
SA57250-XX as a boost (step-up) converter.
SL01461
Figure 1. Simplified system diagram.
2
2003 Nov 12
Philips Semiconductors
Product data
CMOS switching regulator (PWM controlled)
SA57250-XX
ORDERING INFORMATION
PACKAGE
TEMPERATURE
RANGE
TYPE NUMBER
NAME
DESCRIPTION
VERSION
SOT23-5,
SOT25, SO5
SA57250-XXGW
Plastic small outline package; 5 leads; body width 1.6 mm
SOP003
–40 to +85 °C
NOTE:
Part number marking
The device has seven voltage output options, indicated by the XX
on the Type Number.
Each device is marked with a four letter code. The first three letters
designate the product. The fourth letter, represented by ‘x’, is a date
tracking code.
XX
20
25
28
30
33
36
50
VOLTAGE (Typical)
Part number
Marking
A D V x
A D W x
A D X x
A D Y x
A D Z x
A E A x
A E B x
2.0 V
2.5 V
2.8 V
3.0 V
3.3 V
3.6 V
5.0 V
SA57250-20GW
SA57250-25GW
SA57250-28GW
SA57250-30GW
SA57250-33GW
SA57250-36GW
SA57250-50GW
PIN CONFIGURATION
PIN DESCRIPTION
PIN SYMBOL
DESCRIPTION
1
2
ON/OFF
FB
ON/OFF control pin. Connect to V if not used.
IN
ON/OFF
FB
1
2
3
5
SW
Feedback from the output voltage to the PWM
control.
3
4
5
NC
No connection.
GND
SW
Ground.
N/C
4
GND
Switching voltage to inductor.
SL01455
Figure 2. Pin configuration.
MAXIMUM RATINGS
SYMBOL
PARAMETER
MIN.
–0.3
–0.3
–0.3
–0.3
–
MAX.
11
UNIT
V
V
V
V
V
Power supply voltage
FB pin voltage
DD
FB
11
V
ON/OFF pin voltage
SW pin voltage
11
V
ON/OFF
SW
11
V
I
SW pin current
300
+85
+125
150
mA
°C
°C
mW
SW
T
Operating temperature
Storage temperature
Power dissipation
–40
–40
–
oper
T
stg
P
D
3
2003 Nov 12
Philips Semiconductors
Product data
CMOS switching regulator (PWM controlled)
SA57250-XX
ELECTRICAL CHARACTERISTICS
T
amb
= 25 °C, unless otherwise specified.
SYMBOL
PARAMETER
input voltage
CONDITIONS
Part #
–
MIN.
–
TYP.
–
MAX.
9.0
UNIT
V
V
V
V
V
IN
operating start voltage
oscillation start voltage
operation hold voltage
consumption current 1
I
I
= 1.0 mA
= 1.0 mA
–
–
–
0.9
V
ST1
ST2
HLD
OUT
–
0.7
0.7
–
–
0.8
V
–
–
–
V
OUT
-20
-25
-28
-30
-33
-36
-50
-20
-25
-28
-30
-33
-36
-50
-20
-25
-28
-30
-33
-36
-50
–
11.6
14.3
16.1
17.2
19.1
22.4
38.5
3.1
3.2
3.2
3.2
3.3
3.3
3.5
5.6
4.1
4.1
3.2
3.2
3.2
2.2
–
19.4
23.9
26.8
28.7
31.8
37.3
64.1
6.2
µA
µA
µA
µA
µA
µA
µA
µA
µA
µA
µA
µA
µA
µA
Ω
I
V
= output voltage × 0.95
(ON/OFF is ON)
SS1
OUT
–
–
–
–
–
–
–
I
consumption current 2
V
OUT
= output voltage + 0.5 V
(ON/OFF is OFF)
SS2
–
6.3
–
6.4
–
6.4
–
6.5
–
6.5
–
6.9
–
8.9
R
internal switch-on resistance
V
= 0.4 V
DS(ON)
SW
–
6.5
Ω
–
6.5
Ω
–
5.1
Ω
–
5.1
Ω
–
5.1
Ω
–
3.5
Ω
I
switching transistor leakage
current
V
= V = 9 V
SW
–
1.0
µA
SWO
OUT
∆V
∆V
load ripple voltage
I
= 10 mA ≈ I (following) × 1.25
OUT
–
–
–
–
30
60
–
mV
OUT2
OUT
/∆T
OUT
output voltage temperature
coefficient
–40 °C ≤ T ≤ +85 °C
amb
±50
ppm/°C
amb
f
oscillation frequency
maximum duty ratio
soft start time
V
V
= output voltage × 0.95
= output voltage × 0.95
–
–
–
–
42.5
75
50
83
6.0
–
57.5
90
kHz
%
OSC
OUT
MaxDuty
OUT
t
I
I
= 1.0 mA
3.0
–
12
ms
µA
SS
OUT
power OFF consumption
current
ON/OFF pin = 0 V
0.5
SS(OFF)
V
V
E
ON/OFF pin voltage
ON/OFF pin voltage
efficiency
V
V
= output voltage × 0.95
= output voltage × 0.95
–
0.75
–
–
–
0.3
–
V
V
SH
OUT
–
–
SL
OUT
-20
-25
-28
-30
-33
-36
-50
–
75
79
79
83
83
83
87
%
%
%
%
%
%
%
FFI
–
–
–
–
–
–
–
–
–
–
–
–
4
2003 Nov 12
Philips Semiconductors
Product data
CMOS switching regulator (PWM controlled)
SA57250-XX
TYPICAL PERFORMANCE CURVES
50
5
4
3
2
1
0
V
= OUTPUT VOLTAGE × 0.95
V
= OUTPUT VOLTAGE + 0.5 V
OUT
OUT
40
30
20
10
0
I
SS2
I
SS1
(µA)
(µA)
–40
–20
0
20
40
°C)
60
80
100
–40
–20
0
20
40
°C)
60
80
100
T
amb
T
amb
SL01456
SL01457
Figure 3. Supply current 1 versus temperature.
Figure 4. Supply current 2 versus temperature.
70
50
V
= OUTPUT VOLTAGE × 0.95
T
amb
= 25 °C
OUT
65
60
55
50
45
40
35
40
30
20
10
f
I
SS1, 2
(µA)
OSC
(kHz)
0
30
–40
–20
0
20
40
°C)
60
80
100
0
2
4
6
8
10
T
amb
V
(V)
OUT
SL01458
SL01459
Figure 5. Oscillator frequency versus temperature.
Figure 6. Supply current 1, 2 versus V
.
OUT
70
250
T
V
= 25 °C
T
amb
= 25 °C
amb
= 0.4 V
CONT
200
150
100
50
60
f
(kHz)
OSC
I
SW
50
(mA)
40
0
30
0
1
2
3
4
5
0
1
2
3
4
5
V
(V)
V
(V)
OUT
OUT
SL01462
SL01471
Figure 7. Typical switch current versus V
.
Figure 8. Oscillator frequency versus V
.
OUT
OUT
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2003 Nov 12
Philips Semiconductors
Product data
CMOS switching regulator (PWM controlled)
SA57250-XX
V
= 1.8 V
V
= 1.8 V
IN
IN
V
OUT
V
OUT
OUTPUT VOLTAGE
(20 mV/div)
OUTPUT VOLTAGE
(20 mV/div)
V
SW
V
SW
SW VOLTAGE
(1 V/div)
SW VOLTAGE
(1 V/div)
t (10 µs/div)
t (10 µs/div)
SL01472
= 1.8 V
SL01473
= 60 mA
Figure 9. Ripple voltage at I
= 200 µA.
Figure 10. Ripple voltage at I
= 10 mA.
OUT
OUT
I
V
OUT
IN
V
V
OUT
IN
OUTPUT VOLTAGE
(20 mV/div)
INPUT VOLTAGE
(1 V/div)
V
V
SW
OUT
SW VOLTAGE
(1 V/div)
OUTPUT VOLTAGE
(1 V/div)
t (10 µs/div)
t (1 ms/div)
SL01474
SL01475
Figure 11. Ripple voltage at I
= 10 mA.
Figure 12. Start-up characteristic V : 0 V → 1.8 V.
IN
OUT
I : 100 µA → 50 mA; V = 1.8 V
OUT IN
I
= 60 mA; V = 1.8 V
IN
OUT
V
I
OUT
LOAD CURRENT
(20 mA/div)
ON/OFF
INPUT VOLTAGE
(1 V/div)
V
V
OUT
OUTPUT VOLTAGE
(50 mV/div)
OUT
OUTPUT VOLTAGE
(1 V/div)
t (1 ms/div)
t (200 µs/div)
SL01476
SL01477
Figure 13. Start-up characteristic V
: 0 V → 3.0 V.
ON/OFF
Figure 14. Output load regulation, increasing current.
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2003 Nov 12
Philips Semiconductors
Product data
CMOS switching regulator (PWM controlled)
SA57250-XX
I
: 50 mA → 100 µA; V = 1.8 V
V
: 1.8 V → 2.4 V; I
= 50 mA
OUT
OUT
IN
IN
I
OUT
LOAD CURRENT
(20 mA/div)
V
IN
INPUT VOLTAGE
(500 mV/div)
V
OUT
V
OUT
OUTPUT VOLTAGE
(50 mV/div)
OUTPUT VOLTAGE
(50 mV/div)
t (5 ms/div)
t (100 µs/div)
SL01478
SL01479
Figure 15. Output load regulation, decreasing current.
Figure 16. Input line regulation, increasing current.
1.0
0.9
0.8
0.7
0.6
0.5
0.4
0.3
V
: 2.4 V → 1.8 V; I
= 50 mA
OUT
IN
V
IN
INPUT VOLTAGE
(500 mV/div)
V
OUT
V
V
ST1
DO
0.2
0.1
0.0
OUTPUT VOLTAGE
(50 mV/div)
0
1
2
3
4
5
6
7
8
9
10
t (200 µs/div)
I , OUTPUT CURRENT (mA)
OUT
SL01480
SL01481
Figure 17. Input line regulation, decreasing current.
Figure 18. Output current versus starting voltage.
1.0
0.9
0.8
0.7
0.6
0.5
0.4
0.3
500
450
400
350
300
250
200
150
100
50
V
= 2 V
OUT
V
V
= 3 V
= 5 V
OUT
OUT
V
V
ST1
DO
0.2
0.1
0.0
0
0
1
2
3
4
5
6
7
8
9
10
0.0
0.5
1.0
1.5
2.0
V , INPUT VOLTAGE (V)
IN
2.5
3.0
3.5
4.0
4.5 5.0
I
, OUTPUT CURRENT (mA)
OUT
SL01482
SL01483
Figure 19. Input voltage versus output current.
Figure 20. Input voltage versus supply current.
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2003 Nov 12
Philips Semiconductors
Product data
CMOS switching regulator (PWM controlled)
SA57250-XX
5.10
5.08
5.06
5.04
5.02
5.00
4.98
100
95
90
85
80
75
70
65
60
55
50
4.96
4.94
4.92
4.90
V
V
V
V
= 0.9 V
= 1.8 V
= 3.0 V
= 4.0 V
V
V
V
V
= 0.9 V
= 1.8 V
= 3.0 V
= 4.0 V
IN
IN
IN
IN
IN
IN
IN
IN
0.01
0.1
1
10
100
1000
0.01
0.1
1
10
100
1000
I , OUTPUT CURRENT (mA)
OUT
I , OUTPUT CURRENT (mA)
OUT
SL01484
SL01485
Figure 21. Output current versus voltage (100 µH inductor).
Figure 22. Output current versus efficiency (100 µH inductor).
100
90
80
70
60
50
40
30
20
10
0
V
= 0.9 V
IN
V
V
= 1.8 V
= 2.4 V
IN
IN
0.01
0.1
1
10
100
1000
I , OUTPUT CURRENT (mA)
OUT
SL01486
Figure 23. Output current versus ripple voltage
(L = 100 µH; C = 33 µF).
0
OUT
8
2003 Nov 12
Philips Semiconductors
Product data
CMOS switching regulator (PWM controlled)
SA57250-XX
TECHNICAL DISCUSSION
voltage error amplifier for maintaining a fixed output voltage, the
oscillator, and a PWM generator are all in the package.
General discussion
The SA57250-XX is a highly integrated boost-mode switching power
supply integrated circuit. Each device is set to provide a fixed output
voltage by having a fully compensated internal voltage feedback
loop. The SA57250-XX operates at a fixed frequency of 50 kHz and
can operate from a single alkaline cell (0.9 V) or up to 9 V.
The SA57250-XX can be turned off via an ON/OFF pin, thus
allowing the system to unpower portions of the circuitry for added
power savings.
The SA57250-XX has an internal common-source power switching
MOSFET that provides the PWM signal to the boost inductor. The
SW
FB
2
5
V
DD
V
SA57250-XX
REF
R
R
PWM CONTROL
GND
4
SOFT START
1
ON/OFF
SL01463
Figure 24. Functional diagram.
9
2003 Nov 12
Philips Semiconductors
Product data
CMOS switching regulator (PWM controlled)
SA57250-XX
APPLICATION INFORMATION
PASSIVE SNUBBER
(OPTIONAL)
The SA57250-XX can be used for a simple boost (step-up)
converter or the less commonly used flyback converter (isolated
boost). The major operating restriction of the simple boost converter
L
0
is that its output voltage must always be above the highest
V
V
IN
D
OUT
expected value of the input voltage. The flyback converter circuit
requires more parts, but the output voltage is not restricted by the
input voltage.
SW
FB
C
C
OUT
IN
Boost converter fundamentals
The boost or step-up converter is a non-dielectrically isolated
switching power supply topology (arrangement of power parts). That
is, the input power source is directly connected to the output load
(ground and signals). A typical boost converter with optional passive
snubber can be seen in Figure 25.
SA57250-XX
ON/OFF
To understand the boost converter’s operation, examine its three
periods of operation. These periods are: the power switch on-time
(period 1); the inductor discharge period (period 2); and the inductor
empty state (period 3). These periods and their associated currents
can be seen in Figure 26.
GND
SL01466
Figure 25. Boost converter.
SPIKE
ENERGY BEING TRANSFERRED TO OUTPUT
+V
OUT
ENERGY BEING
STORED IN INDUCTOR
CORE EMPTY,
PARASITIC CIRCULATING ENERGY
+V
IN
PERIOD 2
PERIOD 1
PERIOD 3
I
× R
DS(ON)
SW
0
I
peak
V
IN
(V – V
)
IN
OUT
L
0
L
0
SL01464
Figure 26. Boost converter waveforms (discontinuous mode).
10
2003 Nov 12
Philips Semiconductors
Product data
CMOS switching regulator (PWM controlled)
SA57250-XX
Period 1: power switch on-time
Period 3: inductor empty state
During this period, a simple circuit loop is formed when the power
switch is on. The input voltage source is connected directly across
DISCONTINUOUS MODE—This period as displayed in Figure 26 occurs
in the discontinuous–mode of operation of a boost converter. It is
identified by a period of “ringing” following the output period
(period 2). The inductor has been completely emptied of its stored
energy and the switched node returns to the level of the input
voltage. Ringing is seen at this node because a resonant circuit is
the boost inductor (L ). A current ramp is exhibited whose slope is
0
described by:
VIN
Eqn. (1)
IL(on)
+
L0
formed by the inductance of L and any parasitic inductances and
0
Energy is then stored within the core material of the inductor and is
described by:
capacitances connected to that node. This ringing has very little
energy and can easily be eliminated by a small passive snubber.
2
CONTINUOUS MODE—If the inductor is not completely emptied of its
stored energy before the power switch turns on again, the converter
is operating in the continuous mode. A small amount of residual flux
(energy) remains in the inductor core and the current waveform
jumps to an initial value when the power switch is again turned-on.
This mode offers some advantages over the discontinuous-mode,
because the peak current seen by the power switch is lower. In low
voltage applications, the inductor can store more energy with lower
peak currents.
Eqn. (2)
Esto + 0.5L0 Ipeak
This current ramp continues until the controller turns off the power
switch.
Period 2: inductor discharge period
The instant the power switch turns off, the current flowing through
the inductor forces the voltage at its output node (switched node) to
rise quickly above the input voltage (spike). This voltage is then
clamped when it exceeds the device’s output voltage and the output
rectifier becomes forward biased. The inductor empties its stored
energy in the form of a linearly decreasing current ramp whose
slope is dictated by:
The continuous mode waveforms can be seen in Figure 27.
VIN * VOUT
Eqn. (3)
IL(off)
[
L0
The stored energy is transferred to the output capacitor. This output
current continues until the magnetic core is completely emptied of its
stored energy or the power switch turns back on.
SPIKE
+V
OUT
+V
IN
ENERGY BEING
STORED IN
INDUCTOR
ENERGY BEING
TRANSFERRED
TO OUTPUT
0
V
IN
L
I
0
peak
(V – V
)
IN
OUT
L
0
RESIDUAL FLUX
SL01465
Figure 27. Continuous mode waveforms.
11
2003 Nov 12
Philips Semiconductors
Product data
CMOS switching regulator (PWM controlled)
SA57250-XX
Determining the value of the boost inductor
The minimum value of the output capacitor can be estimated by
Equation (7).
The precise value of the boost inductor is not critical to the operation
of the SA57250-XX. The value of the boost inductor should be
calculated to provide continuous-mode operation over most of its
operating range. The converter may enter the discontinuous-mode
when the output load current falls to less than about 20 percent of
the full-load current.
(IOUT(max)) (Toff
)
Eqn. (7)
COUT
Where:
u
Vripple(p*p)
I
is the average value of the output load current (A).
OUT
At low input voltages, the time required to store the needed energy
lengthens, but the time needed to empty the inductor’s core of its
energy shrinks. Conversely, at high input voltages, the time needed
to store the energy shrinks while the time needed to empty the core
increases. See Equations (1) and (3). At the extremes of these
conditions, the converter will fall out of regulation, that is the output
voltage will begin to fall, because the time needed for either storing
or emptying the stored inductor energy is too short to support the
output load current.
T
V
is the nominal off–time of the power switch (sec) [ 10 µs].
off
is the desired amount of ripple voltage (V ).
p–p
ripple
Finding the value of the input capacitor is done by Equation (8).
(Ipeak) (Ton
Vdrop
)
Eqn. (8)
CIN
Where:
u
I
is the expected maximum peak current of the switch (A).
peak
T
V
is the on-time of the switch (sec) [ 10 µs].
on
Equation (4) determines the nominal value of the inductance.
is the desired amount of voltage drop across the capacitor
drop
VIN(min) Ton
(V ).
p–p
Eqn. (4)
L0 ^
Ipeak
These calculations should produce a good estimate of the needed
values of the input and output capacitors to yield the desired ripple
voltages.
Where:
V
is the lowest expected input operating voltage (V).
IN(min)
T
is about 10 µs or one-half the switching period (s).
is the maximum peak current for the SA57250-XX (0.3 A).
on
Selecting the output rectifier
I
peak
The output rectifier (D) is critical to the efficiency and low-noise
operation of the boost converter. The majority of the loss within the
supply will be caused by the output rectifier. Three parameters are
important in the rectifier’s operation within a boost-mode supply.
These are defined below.
This is an estimated inductor value and you can select an
inductance value slightly higher or lower with little effect on the
converter’s operation. If the design falls out of regulation within the
desired operating range, reduce the inductance value, but by no
more than 30 percent.
Forward voltage drop (V )—This is the voltage across the rectifier
f
when a forward current is flowing through the rectifier. A P-N
ultra-fast diode exhibits a 0.7 – 1.4 volt drop, and this drop is
relatively fixed over the range of forward currents. A Schottky diode
exhibits a 0.3 – 0.6 volt drop and appears more resistive during the
forward conduction periods. That is, its forward voltage drop
increases with increasing currents. You can gain an advantage by
purposely over-rating the current rating of a Schottky rectifier.
Determining the minimum value of the capacitors
The input and output capacitors experience the current waveforms
seen in Figures 26 and 27. The peak currents can be typically
between 3 to 6 times the average currents flowing into the input and
from the output. This makes the choice of capacitor an issue of how
much ripple voltage can be tolerated on the capacitor’s terminals
and how much heating the capacitor can tolerate. At the power
levels produced by the SA57250-XX heating is not a major issue.
Reverse recovery time (T )—This is an issue when the boost
rr
supply is operating in the continuous-mode. T is the amount of time
rr
The Equivalent Series Resistance (ESR) of the capacitor, the
resistance that appears between its terminals, and the actual
capacitance causes heat to be generated within the case whenever
there is current entering or exiting the capacitor. ESR also adds to
the apparent voltage drop across the capacitor. The heat that is
generated can be approximated by Equation (5).
required for the rectifier to assume an open circuit when a forward
current is flowing and a reverse voltage is then placed across its
terminals. P-N ultra-fast rectifiers typically have a 25–40 ns reverse
recovery time. Schottky rectifiers have a very short or no reverse
recovery time.
Forward recovery time (T )—This is the amount of time before a
fr
Eqn. (5)
PD(in watts) ^ (1.8Iav) 2(RESR
)
rectifier begins conducting forward current after a forward voltage is
placed across its terminals. This parameter is not always well
specified by the rectifier manufacturers. It causes a spike to appear
when the power switch turns off. This particular point in its operation
causes the most radiated noise. Several rectifiers may have to be
evaluated for the prototype. After the final output rectifier selection is
made, if the spike is still causing a problem a small passive snubber
can be placed across the rectifier.
ESR’s effect on the capacitor voltage is given by Equation (6).
DVC ^ Ipeak(RESR
)
(expressed as V
)
Eqn. (6)
p-p
A ceramic capacitor would typically be used in this application if the
required value is less than 1 – 10 µF, or a tantalum capacitor for
required values of 10 µF and above. Lower cost aluminum
electrolytic capacitors can be used, but you should confirm that the
higher ESRs typically exhibited by these capacitors does not cause
a problem.
For this boost application, the best choice of output rectifier is a low
forward drop, 0.5 – 1 ampere, 20 volt Schottky rectifier such as the
Philips part number BAT120A.
12
2003 Nov 12
Philips Semiconductors
Product data
CMOS switching regulator (PWM controlled)
SA57250-XX
Flyback converter
The SA57250-XX can also be used to create a flyback converter,
COILTRONICS
CTX100-1P
V
V
OUT
BATT
also known as an isolated boost converter. The advantage of a
flyback converter is that the input voltage can go higher or lower
than the output voltage without affecting the operation of the
converter. The only restrictions are the peak current flowing into the
switch pin (SW) and the breakdown voltage of the SW and feedback
3.3 V
@ 0.1 A
47 µF
@ 10 V
47 µF
@ 10 V
(V ) pins.
FB
SW
One transformer can accommodate a variety of output voltages in
different applications, because the circuit will change the on and
off–times to provide the desired output voltage.
FB
SA57250-33
The output voltage of the flyback can be changed by using a
SA57250-XX with the desired output voltage, with no other changes
to the circuit.
ON/OFF
GND
Selecting the components
SL01467
It is best to operate the transformer in the continuous-mode where
the highest expected peak primary current is below the maximum
current rating of the SA57250-XX switch.
Figure 28. Flyback converter circuit.
Because the SA57250-XX is a peak current-limited IC, begin with a
peak current equal to or less than the maximum current rating of the
part (0.3 A). A reasonable value of the primary inductance can be
found in Equation (9).
V
+ V
OUT
IN
(1:1 TRANSFORMER)
Ton
Ipeak
Eqn. (9)
Lpri t 5VIN(min)
V
IN
Where:
I
is 0.3 A or less.
peak
T
on
is the maximum expected on-time of the switch (≈10 µs).
V
is the lowest expected input voltage (V)
IN(min)
Then select an off-the-shelf transformer such as the Coiltronics
CTX100–1P, a 1:1 turns ratio transformer that has a primary
inductance of 100 µH. It does not reach saturation until the primary
current reaches 440 mA, which is above the expected peak current
of the flyback converter. The 1:1 turns ratio should work for output
voltages from 0.8 to 2 times the highest input voltage, and produce
the output voltage set by the SA57250-XX. The only other restriction
is that the input voltage plus the output voltage must be less than
the breakdown voltage of the SA57250-XX (9 V).
+V
OUT
GROUND (0 V)
–V
IN
Use Equation (8) to determine the minimum value for the input
capacitor. A 0.1 V drop is desired across this capacitor.
(0.3A) (10ms)
CIN
u
+ 30mF
0.1V
I
SW(peak)
A 47 µF at 6 V tantalum capacitor would be suitable.
For the design example, the output voltage will be +3.3 V with a
maximum output current of 50 mA. The input voltage can vary
between +1.8 V and 4.0 V. The design can be seen in Figure 28,
and the expected waveforms can be seen in Figure 29.
V
IN
I
= I
diode
peak
I
diode(peak)
(1:1 TRANSFORMER)
SL01468
Figure 29. Flyback converter waveforms.
13
2003 Nov 12
Philips Semiconductors
Product data
CMOS switching regulator (PWM controlled)
SA57250-XX
Designing a passive snubber
If the switching power supply is generating too much radio
PASSIVE SNUBBER
V
V
OUT
BATT
frequency interference (RFI) a passive snubber can be added.
A passive snubber is a series resistor and capacitor placed across
any component that exhibits a resonant “ringing”. This series R-L-C
loop creates a lossy or damped tank circuit that dissipates the
ringing energy. The design is critical, because it introduces another
loss within the converter.
R
SW
Designing a snubber is an empirical process, mainly because it
involves undefined parasitic capacitances and inductances
contributed by the PCB layout, leakage inductance, and device
capacitances. The snubber should be placed across the major
source of the spike or ringing which is the output rectifier for a boost
converter (see Figure 25) and the primary windings of the
transformer in a flyback transformer (see Figure 30).
FB
SA57250-XX
ON/OFF
GND
The usual design process is:
SL01469
1. Measure the period of the undesired ringing (T ).
0
2. Place a very small ceramic capacitor (about 10 pF) across the
output rectifier or primary winding.
Figure 30. Passive snubber in flyback converter.
3. Re-measure the period of the undesired ringing. The new period
should be about 3 times that of T . If it is less than this, place a
0
slightly larger value of capacitor across the output rectifier or
primary winding.
4. Once the desired increase in the ringing period is achieved with
a capacitance (C ), place a resistor in series with the capacitor
0
whose value is approximately:
T0
Eqn. (10)
Rsnubber
^
2pC0
This should produce a snubber that does not load the circuit and
introduces a very small loss.
14
2003 Nov 12
Philips Semiconductors
Product data
CMOS switching regulator (PWM controlled)
SA57250-XX
Designing the PCB for effective heat dissipation
Laying out the printed circuit board
The maximum junction temperature is +125 °C which should not be
exceeded under any operating conditions. Designing a PCB that
includes a heatsink system under the device is the key to cooler
operation of the circuit, and the long–term reliable operation of the
converter.
The design of the printed circuit board (PCB) is critical to the proper
operation of all switching power supplies. Its design affects the
supply stability, radio frequency interference behavior and the
reliability of the converter.
Never use the autoroute feature of any PCB design program
because this will always produce traces that are too long and too
thin.
The major sources of heat within the converter are the power switch
inside the SA57250-XX, the resistive losses within the inductor, and
losses associated with the output rectifier. These losses can be
estimated by the following equations:
The input and output capacitors are the only source or sink of the
high frequency currents found in a switching power supply. All
connections to the switching power supply from the outside circuits
should be made to the input or output capacitor terminals (+ and –).
Internally, the layout should adhere to a “one-point” grounding
system, as shown in Figure 31.
Power switch:
2
Eqn. (11)
PD(sw) ^ TON IPK RDS(ON) fSW
Inductor:
2
Eqn. (12)
PD(L0) ^ Ipeak Rwinding
Output rectifier:
L0
V
V
IN
OUT
Eqn. (13)
PD(rect) ^ IOUT(Vfwd)
SW
FB
The thermal resistance (R
) of the SA57250-XX is approximately
th(j-a)
220 °C/W, assuming the device is soldered to a 2 oz. copper FR4
fiberglass circuit board, and that the minimum footprint was used
(copper just under the leads). A rule of thumb in PCB design is that
the thermal resistance can be reduced by 30% for each doubling of
the copper area close to the device. This effect diminishes for areas
greater than five times the minimum PCB footprint. If you take
advantage of this rule, thermal resistance can be reduced by using
wide copper lands when connecting to the leads of the major
power-producing parts. These PCB traces should almost fill the
areas surrounding the converter parts to conduct heat away from
the device. For demanding applications, additional heat dissipation
area can be created by placing a copper island on the opposite side
of the PCB from each wide trace and connecting it to the trace with
vias (plated thru holes).
C
C
IN
OUT
SA57250-XX
INPUT
GROUND
TO ONE POINT
OUTPUT
GROUND
TO ONE POINT
GND
SL01470
Figure 31. Grounding trace for converter.
The traces between the input and output capacitors and the
inductor, power switch and rectifier(s) should be as short and wide
as possible. This reduces the series resistance and inductance that
can be introduced by traces.
The junction temperature can be estimated by Equation (14).
Eqn. (14)
Tj ^ (PD Rth(j-a)Ȁ) ) Tamb(max)
Where:
The guidelines for a PCB layout can be summarized as:
P
D
is the power dissipation (W).
• The traces between the input and output capacitor to the inductor,
R
′ is the effective thermal resistance with the additional
copper (°C/W).
is the highest local expected ambient temperature (°C).
th(j-a)
power switch and the rectifier should be made as short and as
T
amb
wide as possible.
• Strictly adhere to the one–point wiring practices shown in
Figure 31.
• On a 2-sided board, do not run sensitive signals traces under the
AC voltage node.
• The IC (control) ground is terminated at the output capacitor’s
negative terminal.
15
2003 Nov 12
Philips Semiconductors
Product data
CMOS switching regulator (PWM controlled)
SA57250-XX
PACKING METHOD
The SA57250-XX is packed in reels, as shown in Figure 32.
GUARD
BAND
TAPE
TAPE DETAIL
REEL
ASSEMBLY
COVER TAPE
CARRIER TAPE
BARCODE
LABEL
BOX
SL01305
Figure 32. Tape and reel packing method
16
2003 Nov 12
Philips Semiconductors
Product data
CMOS switching regulator (PWM controlled)
SA57250-XX
Plastic small outline package; 5 leads; body width 1.6 mm
SOP003
17
2003 Nov 12
Philips Semiconductors
Product data
CMOS switching regulator (PWM controlled)
SA57250-XX
REVISION HISTORY
Rev
Date
Description
_2
20031112
Product data (9397 750 12323). ECN 853-2270 30329 of 09 September 2003.
Supersedes data of 2001 Aug 01 (9397 750 08687).
Modifications:
• Change package outline version to SOP003 in Ordering information table and Package outline sections.
_1
20010801
Product data (9397 750 08687). ECN 853-2270 26807 of 01 August 2001.
Data sheet status
Product
status
Definitions
[1]
Level
Data sheet status
[2] [3]
I
Objective data
Development
This data sheet contains data from the objective specification for product development.
Philips Semiconductors reserves the right to change the specification in any manner without notice.
II
Preliminary data
Qualification
Production
This data sheet contains data from the preliminary specification. Supplementary data will be published
at a later date. Philips Semiconductors reserves the right to change the specification without notice, in
order to improve the design and supply the best possible product.
III
Product data
This data sheet contains data from the product specification. Philips Semiconductors reserves the
right to make changes at any time in order to improve the design, manufacturing and supply. Relevant
changes will be communicated via a Customer Product/Process Change Notification (CPCN).
[1] Please consult the most recently issued data sheet before initiating or completing a design.
[2] The product status of the device(s) described in this data sheet may have changed since this data sheet was published. The latest information is available on the Internet at URL
http://www.semiconductors.philips.com.
[3] For data sheets describing multiple type numbers, the highest-level product status determines the data sheet status.
Definitions
Short-form specification — The data in a short-form specification is extracted from a full data sheet with the same type number and title. For detailed information see
the relevant data sheet or data handbook.
Limitingvaluesdefinition— Limiting values given are in accordance with the Absolute Maximum Rating System (IEC 60134). Stress above one or more of the limiting
values may cause permanent damage to the device. These are stress ratings only and operation of the device at these or at any other conditions above those given
in the Characteristics sections of the specification is not implied. Exposure to limiting values for extended periods may affect device reliability.
Application information — Applications that are described herein for any of these products are for illustrative purposes only. Philips Semiconductors make no
representation or warranty that such applications will be suitable for the specified use without further testing or modification.
Disclaimers
Life support — These products are not designed for use in life support appliances, devices, or systems where malfunction of these products can reasonably be
expected to result in personal injury. Philips Semiconductors customers using or selling these products for use in such applications do so at their own risk and agree
to fully indemnify Philips Semiconductors for any damages resulting from such application.
Right to make changes — Philips Semiconductors reserves the right to make changes in the products—including circuits, standard cells, and/or software—described
or contained herein in order to improve design and/or performance. When the product is in full production (status ‘Production’), relevant changes will be communicated
viaaCustomerProduct/ProcessChangeNotification(CPCN).PhilipsSemiconductorsassumesnoresponsibilityorliabilityfortheuseofanyoftheseproducts,conveys
nolicenseortitleunderanypatent, copyright, ormaskworkrighttotheseproducts, andmakesnorepresentationsorwarrantiesthattheseproductsarefreefrompatent,
copyright, or mask work right infringement, unless otherwise specified.
Koninklijke Philips Electronics N.V. 2003
Contact information
All rights reserved. Printed in U.S.A.
For additional information please visit
http://www.semiconductors.philips.com.
Fax: +31 40 27 24825
Date of release: 11-03
9397 750 12323
For sales offices addresses send e-mail to:
sales.addresses@www.semiconductors.philips.com.
Document order number:
Philips
Semiconductors
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