MC34023P [ONSEMI]
High Speed Single-Ended PWM Controller; 高速单端PWM控制器
| 型号： | MC34023P |
| 厂家： | 
ONSEMI |
| 描述： | High Speed Single-Ended PWM Controller |
| 文件： | 总19页 (文件大小：473K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
Order this document by MC34023/D
The MC34023 series are high speed, fixed frequency, single–ended pulse
width modulator controllers optimized for high frequency operation. They are
specifically designed for Off–Line and DC–to–DC converter applications
offering the designer a cost–effective solution with minimal external
components. These integrated circuits feature an oscillator, a temperature
compensated reference, a wide bandwidth error amplifier, a high speed
current sensing comparator, and a high current totem pole output ideally
suited for driving a power MOSFET.
16
1
P SUFFIX
PLASTIC PACKAGE
CASE 648
Also included are protective features consisting of input and reference
undervoltage lockouts each with hysteresis, cycle–by–cycle current limiting,
and a latch for single pulse metering.
The flexibility of this series allows it to be easily configured for either
current mode or voltage mode control.
• 50 ns Propagation Delay to Output
16
• High Current Totem Pole Output
1
• Wide Bandwidth Error Amplifier
DW SUFFIX
PLASTIC PACKAGE
CASE 751G
• Fully–Latched Logic with Double Pulse Suppression
• Latching PWM for Cycle–By–Cycle Current Limiting
• Soft–Start Control with Latched Overcurrent Reset
• Input Undervoltage Lockout with Hysteresis
• Low Start–Up Current (500 µA Typ)
(SO–16L)
• Internally Trimmed Reference with Undervoltage Lockout
• 90% Maximum Duty Cycle (Externally Adjustable)
• Precision Trimmed Oscillator
PIN CONNECTIONS
• Voltage or Current Mode Operation to 1.0 MHz
• Functionally Similar to the UC3823
Error Amp
Inverting Input
Error Amp
1
2
16
15
14
V
ref
V
CC
Noninverting Input
Simplified Application
Output
3
4
5
6
Error Amp Output
13
V
C
Clock
16
4
15
12 Power Ground
R
T
5.1V
Reference
V
ref
Current Limit
11
V
CC
C
T
Clock
Reference
Ground
10
9
Ramp
7
8
5
6
UVLO
R
T
Current Limit/
Shutdown
Soft–Start
Oscillator
C
T
(Top View)
13
14
12
V
C
7
3
Ramp
Output
Error Amp
Output
Noninverting
Input
Latching
PWM
Power
Ground
Error
Amp
2
ORDERING INFORMATION
Operating
Inverting
Input
11
9
1
8
Current
Limit Ref
Current
Limit/
Shutdown
Temperature Range
Device
Package
Plastic DIP
SO–16L
Soft–Start
Soft–Start
MC33023P
MC33023DW
MC34023P
T
= –40° to +105°C
A
Ground
10
T
A
= 0° to +70°C
Plastic DIP
This device contains 176 active transistors.
Motorola, Inc. 1996
Rev 2
MC34023 MC33023
MAXIMUM RATINGS
Rating
Symbol
Value
Unit
Power Supply Voltage
V
CC
30
V
Output Driver Supply Voltage
V
20
V
A
C
O
Output Current, Source or Sink (Note 1)
DC
Pulsed (0.5 µs)
I
0.5
2.0
Current Sense, Soft–Start, Ramp, and Error Amp Inputs
Error Amp Output and Soft–Start Sink Current
V
–0.3 to +7.0
V
in
I
O
10
mA
mA
Clock and R Output Current
T
I
5.0
CO
Power Dissipation and Thermal Characteristics
SO–16L Package (Case 751G)
Maximum Power Dissipation @ T = +25°C
Thermal Resistance, Junction–to–Air
DIP Package (Case 648)
P
862
145
mW
°C/W
A
D
R
θJA
1.25
100
W
°C/W
Maximum Power Dissipation @ T = +25°C
P
D
A
Thermal Resistance, Junction–to–Air
R
θJA
Operating Junction Temperature
T
J
+150
°C
°C
Operating Ambient Temperature (Note 2)
MC34023
MC33023
T
A
0 to +70
–40 to +105
Storage Temperature Range
T
stg
–55 to +150
°C
ELECTRICAL CHARACTERISTICS (V
CC
= 15 V, R = 3.65 kΩ, C = 1.0 nF, for typical values T = +25°C, for min/max values T is
the operating ambient temperature range that applies [Note 2], unless otherwise noted.)
T
T
A
A
Characteristic
Symbol
Min
Typ
Max
Unit
REFERENCE SECTION
Reference Output Voltage (I = 1.0 mA, T = +25°C)
V
ref
5.05
–
5.1
2.0
2.0
0.2
–
5.15
15
V
mV
mV
mV/°C
V
O
J
Line Regulation (V
CC
= 10 V to 30 V)
Reg
line
load
S
Load Regulation (I = 1.0 mA to 10 mA)
O
Reg
T
–
15
Temperature Stability
–
–
Total Output Variation over Line, Load, and Temperature
Output Noise Voltage (f = 10 Hz to 10 kHz, T = +25°C)
V
ref
4.95
–
5.25
–
V
n
50
µV
J
Long Term Stability (T = +125°C for 1000 Hours)
S
–
5.0
– 65
–
mV
mA
A
Output Short Circuit Current
OSCILLATOR SECTION
Frequency
I
– 30
–100
SC
kHz
T = +25°C
f
380
370
400
400
420
430
J
osc
Line (V
= 10 V to 30 V) and Temperature (T = T
to T
)
high
CC
Frequency Change with Voltage (V
A
low
= 10 V to 30 V)
∆f
∆f
/∆V
/∆T
–
0.2
2.0
2.8
1.0
1.0
–
%
%
V
CC
osc
Frequency Change with Temperature (T = T
A
to T
)
high
–
low
osc
Sawtooth Peak Voltage
Sawtooth Valley Voltage
V
V
2.6
0.7
3.0
1.25
OSC(P)
V
OSC(V)
Clock Output Voltage
High State
Low State
V
V
V
3.9
–
4.5
2.3
–
2.9
OH
OL
NOTES: 1. Maximum package power dissipation limits must be observed.
2. Low duty cycle pulse techniques are used during test to maintain junction temperature as close to ambient as possible.
T
T
= 0°C for MC34023
= –40°C for MC33023
T
T
= +70°C for MC34023
= +105°C for MC33023
low
low
high
high
2
MOTOROLA ANALOG IC DEVICE DATA
MC34023 MC33023
ELECTRICAL CHARACTERISTICS (V
CC
= 15 V, R = 3.65 kΩ, C = 1.0 nF, for typical values T = +25°C, for min/max values T is
the operating ambient temperature range that applies [Note 2], unless otherwise noted.)
T
T
A
A
Characteristic Symbol
Min
Typ
Max
Unit
ERROR AMPLIFIER SECTION
Input Offset Voltage
V
–
–
–
15
3.0
1.0
–
mV
µA
IO
Input Bias Current
I
IB
0.6
0.1
95
Input Offset Current
I
IO
–
µA
Open–Loop Voltage Gain (V = 1.0 V to 4.0 V)
A
VOL
60
4.0
75
85
dB
O
Gain Bandwidth Product (T = +25°C)
GBW
8.3
95
–
MHz
dB
J
Common Mode Rejection Ratio (V
= 1.5 V to 5.5 V)
CM
= 10 V to 30 V)
CMRR
PSRR
–
Power Supply Rejection Ratio (V
110
–
dB
CC
Output Current, Source (V = 4.0 V)
I
0.5
1.0
3.0
3.6
–
–
mA
O
Source
I
Sink
Output Current, Sink (V = 1.0 V)
O
Output Voltage Swing, High State (I = –0.5 mA)
Output Voltage Swing, Low State (I = 1 mA)
O
V
V
4.5
0
4.75
0.4
5.0
1.0
V
O
OH
OL
Slew Rate
SR
6.0
12
–
V/µs
PWM COMPARATOR SECTION
Ramp Input Bias Current
I
IB
–
–0.5
–5.0
µA
Duty Cycle, Maximum
Duty Cycle, Minimum
DC
80
–
90
–
–
0
%
(max)
DC
(min)
Zero Duty Cycle Threshold Voltage Pin 3(4) (Pin 7(9) = 0 V)
V
1.1
–
1.25
60
1.4
V
th
Propagation Delay (Ramp Input to Output, T = +25°C)
t
100
ns
J
PLH(in/out)
SOFT–START SECTION
Charge Current (V
= 0.5 V)
I
3.0
1.0
9.0
4.0
20
–
µA
Soft–Start
Discharge Current (V
chg
= 1.5 V)
I
mA
Soft–Start
CURRENT SENSE SECTION
dischg
Input Bias Current (Pin 9(12) = 0 V to 4.0 V)
I
–
–
–
–
15
45
µA
mV
V
IB
Current Limit Comparator Input Offset Voltage (Pin 11(14) = 1.1 V)
Current Limit Reference Input Common Mode Range (Pin 11(14))
Shutdown Comparator Threshold
V
IO
V
1.0
1.25
–
–
1.25
1.55
80
CMR
V
1.40
50
V
th
Propagation Delay (Current Limit/Shutdown to Output, T = +25°C)
t
ns
J
PLH(in/out)
OUTPUT SECTION
Output Voltage
V
Low State (I
= 20 mA)
= 200 mA)
= 20 mA)
= 200 mA)
V
–
–
13
12
0.25
1.2
13.5
13
0.4
2.2
–
Sink
OL
(I
High State (I
(I
Sink
Source
Source
V
OH
–
Output Voltage with UVLO Activated (V
= 6.0 V, I
= 0.5 mA)
V
–
–
–
–
0.25
100
30
1.0
500
60
V
CC
Sink
OL(UVLO)
Output Leakage Current (V = 20 V)
I
L
µA
ns
ns
C
Output Voltage Rise Time (C = 1.0 nF, T = +25°C)
t
r
L
J
Output Voltage Fall Time (C = 1.0 nF, T = +25°C)
t
f
30
60
L
J
UNDERVOLTAGE LOCKOUT SECTION
Start–Up Threshold (V
Increasing)
V
8.8
0.4
9.2
0.8
9.6
1.2
V
V
CC
th(on)
UVLO Hysteresis Voltage (V
Decreasing After Turn–On)
V
H
CC
TOTAL DEVICE
Power Supply Current
Start–Up (VCC = 8.0 V)
Operating
I
mA
CC
–
–
0.5
20
1.2
30
NOTES: 1. Maximum package power dissipation limits must be observed.
2. Low duty cycle pulse techniques are used during test to maintain junction temperature as close to ambient as possible.
T
T
= 0°C for MC34023
= –40°C for MC33023
T
T
= +70°C for MC34023
= +105°C for MC33023
low
low
high
high
3
MOTOROLA ANALOG IC DEVICE DATA
MC34023 MC33023
Figure 1. Timing Resistor versus
Oscillator Frequency
Figure 2. Oscillator Frequency versus Temperature
100 k
10 k
1200
1000
800
R
C
= 1.2 k
= 1.0 nF
V
T
= 15 V
T
T
CC
= +25
9
1
3
5
7
°C
1.0 MHz
A
2
4
6
8
CT =
1. 100 nF
2. 47 nF
3. 22 nF
4. 10 nF
5. 4.7 nF
6. 2.2 nF
7. 1.0 nF
8. 470 pF
9. 220 pF
V
= 15 V
CC
600
R
C
= 3.6 k
= 1.0 nF
T
T
400 kHz
50 kHz
400
200
0
R
= 36 k
= 1.0 nF
1.0 k
470
T
C
T
4
5
6
7
10
100
1000
10
10
10
– 55
– 25
0
25
50
75
100
125
f
, OSCILLATOR FREQUENCY (Hz)
T , AMBIENT TEMPERATURE (°C)
osc
A
Figure 3. Error Amp Open Loop Gain and
Phase versus Frequency
Figure 4. PWM Comparator Zero Duty Cycle
Threshold Voltage versus Temperature
120
100
80
0
1.30
1.28
1.26
1.24
V
= 15 V
45
CC
Pin 7(9) = 0 V
Gain
60
Phase
40
90
20
1.22
1.20
135
0
– 20
10
100
1.0 k
10 k
100 k
1.0 M
10 M
– 55
– 25
0
25
50
75
C)
100
125
f, FREQUENCY (Hz)
T , AMBIENT TEMPERATURE (
°
A
Figure 5. Error Amp Small Signal
Transient Response
Figure 6. Error Amp Large Signal
Transient Response
3.0 V
2.5 V
2.0 V
2.55 V
2.5 V
2.45 V
0.1
µs/DIV
0.1 µs/DIV
4
MOTOROLA ANALOG IC DEVICE DATA
MC34023 MC33023
Figure 7. Reference Voltage Change
versus Source Current
Figure 8. Reference Short Circuit Current
versus Temperature
0
66
65.6
65.2
64.8
64.4
64
V
= 15 V
– 5.0
CC
T
= – 55°C
A
V
= 15 V
CC
– 10
– 15
T
= +25°C
T
= +125°C
A
A
– 20
– 25
– 30
10
20
30
40
50
– 55
– 25
0
25
50
75
C)
100
125
0
I
, SOURCE CURRENT (mA)
T , AMBIENT TEMPERATURE (
°
Source
A
Figure 9. Reference Line Regulation
Figure 10. Reference Load Regulation
V
LINE REGULATION 10 V to 24 V
(2.0 ms/DIV)
V
LOAD REGULATION 1.0 mA to 10 mA
(2.0 ms/DIV)
ref
ref
Figure 11. Current Limit Comparator Input
Offset Voltage versus Temperature
Figure 12. Shutdown Comparator Threshold
Voltage versus Temperature
100
60
1.50
1.46
V
= 15 V
CC
V
= 15 V
CC
Pin 11(14) = 1.1 V
20
1.42
1.38
– 20
– 60
1.34
1.30
– 100
– 55
– 25
0
25
50
75
C)
100
125
– 55
– 25
0
25
50
75
C)
100
125
T , AMBIENT TEMPERATURE (
°
T , AMBIENT TEMPERATURE (
°
A
A
5
MOTOROLA ANALOG IC DEVICE DATA
MC34023 MC33023
Figure 13. Soft–Start Charge Current
versus Temperature
Figure 14. Output Saturation Voltage
versus Load Current
0
10
9.5
9.0
Source Saturation
V
V
= 15 V
CC
CC
(Load to Ground)
– 1.0
V
80
= 15 V
s Pulsed Load
CC
µ
120 Hz Rate
= 25
– 2.0
T
°C
A
8.5
8.0
7.5
7.0
2.0
1.0
0
Sink Saturation
(Load to V
Ground
)
CC
– 55
– 25
0
25
50
75
C)
100
125
0
0.2
0.4
I , OUTPUT LOAD CURRENT (A)
O
0.6
0.8
1.0
T , AMBIENT TEMPERATURE (
°
A
Figure 15. Drive Output Rise and Fall Time
Figure 16. Drive Output Rise and Fall Time
OUTPUT RISE & FALL TIME 1.0 nF LOAD
50 ns/DIV
OUTPUT RISE & FALL TIME 10 nF LOAD
50 ns/DIV
Figure 17. Supply Voltage versus Supply Current
30
25
R
C
= 3.65 kΩ
= 1.0 nF
T
T
20
15
V
Increasing
CC
V
Decreasing
CC
10
5.0
0
0
4.0
8.0
12
16
20
V
, SUPPLY VOLTAGE (V)
CC
6
MOTOROLA ANALOG IC DEVICE DATA
MC34023 MC33023
Figure 18. Representative Block Diagram
V
V
in
CC
16
15
Reference
Regulator
V
ref
V
V
CC
CC
4
5
UVLO
Clock
9.2 V
13
V
V
4.2 V
ref
UVLO
Oscillator
C
R
T
14
6
7
C
T
Output
12
R
PWM
Comparator
Q
Power Ground
1.25 V
Ramp
S
PWM Latch
Error Amp Output
3
2
Current
Limit
Error
Amp
+
11
Noninverting Input
Inverting Input
Current Limit Reference
9.0
µA
1
8
9
Current Limit/Shutdown
Soft–Start
0.5 V
Soft–Start Latch
R
S
1.4 V
C
SS
Q
Shutdown
10
Ground
Figure 19. Current Limit Operating Waveforms
C
T
Clock
Soft–Start
Error Amp Output
Ramp
PWM
Comparator
Output
7
MOTOROLA ANALOG IC DEVICE DATA
MC34023 MC33023
OPERATING DESCRIPTION
The MC33023 and MC34023 series are high speed, fixed
limiting the duty cycle. The time it takes for a capacitor to
reach full charge is given by:
frequency, single–ended pulse width modulator controllers
optimized for high frequency operation. They are specifically
designed for Off–Line and DC–to–DC converter applications
offering the designer a cost effective solution with minimal
external components. A representative block diagram is
shown in Figure 18.
5
t
(4.5 • 10 ) C
Soft-Start
A Soft–Start latch is incorporated to prevent erratic
operation of this circuitry. Two conditions can cause the
Soft–Start circuit to latch so that the Soft–Start capacitor
stays discharged. The first condition is activation of an
Oscillator
undervoltage lockout of either V
condition is when current sense input exceeds 1.4 V. Since
this latch is “set dominant”, it cannot be reset until either of
or V . The second
CC
ref
The oscillator frequency is programmed by the values
selected for the timing components R and C . The R pin is
T
T
T
set to a temperature compensated 3.0 V. By selecting the
value of R , the charge current is set through a current mirror
thesesignalsisremovedand,thevoltageatC
than 0.5 V.
isless
Soft–Start
T
for the timing capacitor C . This charge current runs
T
continuously through C . The discharge current is ratioed to
T
PWM Comparator and Latch
be 10 times the charge current, which yields the maximum
A PWM circuit typically compares an error voltage with a
ramp signal. The outcome of this comparison determines the
state of the output. In voltage mode operation the ramp signal
is the voltage ramp of the timing capacitor. In current mode
operation the ramp signal is the voltage ramp induced in a
current sensing element. The ramp input of the PWM
comparator is pinned out so that the user can decide which
mode of operation best suits the application requirements.
The ramp input has a 1.25 V offset such that whenever the
voltage at this pin exceeds the error amplifier output voltage
minus 1.25 V, the PWM comparator will cause the PWM latch
to set, disabling the outputs. Once the PWM latch is set, only
a blanking pulse by the oscillator can reset it, thus initiating
the next cycle.
duty cycle of 90%. C is charged to 2.8 V and discharged to
1.0 V. During the discharge of C , the oscillator generates an
T
internal blanking pulse that resets the PWM Latch and,
inhibits the outputs. The threshold voltage on the oscillator
comparator is trimmed to guarantee an oscillator accuracy of
5.0% at 25°C.
T
Additional dead time can be added by externally
increasing the charge current to C as shown in Figure 23.
T
This changes the charge to discharge ratio of C which is set
T
internally to I
ratio will be:
/10 I
charge
. The new charge to discharge
charge
I
I
l
additiona
charge
)
% Deadtime
10 (I
charge
Current Limiting and Shutdown
A bidirectional clock pin is provided for synchronization or
for master/slave operation. As a master, the clock pin
A pin is provided to perform current limiting and shutdown
operations. Two comparators are connected to the input of
this pin. The reference voltage for the current limit
comparator is not set internally. A pin is provided so the user
can set the voltage. When the voltage at the current limit
input pin exceeds the externally set voltage, the PWM latch is
set, disabling the output. In this way cycle–by–cycle current
limiting is accomplished. If a current limit resistor is used in
series with the power devices, the value of the resistor is
found by:
provides a positive output pulse during the discharge of C .
T
As a slave, the clock pin is an input that resets the PWM latch
and blanks the drive output, but does not discharge C .
T
Therefore, the oscillator is not synchronized by driving the
clock pin alone. Figures 27, 28 and 29 provide suggested
synchronization.
Error Amplifier
A fully compensated Error Amplifier is provided. It features
a typical DC voltage gain of 95 dB and a gain bandwidth
product of 8.3 MHz with 75 degrees of phase margin
(Figure 3). Typical application circuits will have the
noninverting input tied to the reference. The inverting input
will typically be connected to a feedback voltage generated
from the output of the switching power supply. Both inputs
I
Limit Reference Voltage
R
Sense
I
pk (switch)
If the voltage at this pin exceeds 1.4 V, the second
comparator is activated. This comparator sets a latch which,
in turn, causes the soft start capacitor to be discharged. In
this way a “hiccup” mode of recovery is possible in the case
of output short circuits. If a current limit resistor is used in
series with the output devices, the peak current at which the
controller will enter a “hiccup” mode is given by:
have a common mode voltage (V
5.5 V. The Error Amplifier Output is provided for external loop
compensation.
) input range of 1.5 V to
CM
Soft–Start Latch
Soft–Start is accomplished in conjunction with an external
capacitor. The Soft–Start capacitor is charged by an internal
9.0 µA current source. This capacitor clamps the output of
the error amplifier to less than its normal output voltage, thus
1.4 V
I
shutdown
R
Sense
8
MOTOROLA ANALOG IC DEVICE DATA
MC34023 MC33023
Undervoltage Lockout
current feedback loop. It has been shown that the instability is
caused by a double pole at half the switching frequency. If an
external ramp (S ) is added to the on–time ramp (S ) of the
current–sense waveform, stability can be achieved.
One must be careful not to add too much ramp
compensation. If too much is added the system will start to
perform like a voltage mode regulator. All benefits of current
mode control will be lost. Figure 25 is an example of one way
in which external ramp compensation can be implemented.
There are two undervoltage lockout circuits within the IC.
The first senses V
and the second V . During power–up,
e
n
CC
ref
V
must exceed 9.2 V and V must exceed 4.2 V before
ref
CC
the outputs can be enabled and the Soft–Start latch released.
If V falls below 8.4 V or V falls below 3.6 V, the outputs
CC
ref
are disabled and the Soft–Start latch is activated. When the
UVLO is active, the part is in a low current standby mode
allowing the IC to have an off–line bootstrap start–up circuit.
Typical start–up current is 500 µA.
Figure 20. Ramp Compensation
Output
The MC34023 has a high current totem pole output
specifically designed for direct drive of power MOSFETs. It is
capable of up to ± 2.0 A peak drive current with a typical rise
and fall time of 30 ns driving a 1.0 nF load.
Ramp Compensation
Ramp Input
1.25 V
Separate pins for V and Power Ground are provided.
Ramp
Compensation S
C
With proper implementation, a significant reduction of
switching transient noise imposed on the control circuitry is
e
possible. The separate V supply input also allows the
designer added flexibility in tailoring the drive voltage
C
Current
Signal S
n
independent of V
.
CC
A simple equation can be used to calculate the amount of
external ramp slope necessary to add that will achieve
stability in the current loop. For the following equations, the
calculated values for the application circuit in Figure 34 are
also shown.
Reference
A 5.1 V bandgap reference is pinned out and is trimmed to
an initial accuracy of ±1.0% at 25°C. This reference has short
circuit protection and can source in excess of 10 mA for
powering additional control system circuitry.
Design Considerations
V
N
N
O
L
S
P
Do not attempt to construct the converter on
wire–wrap or plug–in prototype boards. With high
frequency, high power, switching power supplies it is
imperative to have separate current loops for the signal paths
and for the power paths. The printed circuit layout should
contain a ground plane with low current signal and high
current switch and output grounds returning on separate
paths back to the input filter capacitor. Shown in Figure 35 is
a printed circuit layout of the application circuit. Note how the
power and ground traces are run. All bypass capacitors and
snubbers should be connected as close as possible to the
specific part in question. The PC board lead lengths must be
less than 0.5 inches for effective bypassing for snubbing.
S
(R )Ai
S
e
where:
V
= DC output voltage
N , N = number of power transformer primary
O
S
P
= or secondary turns
A = gain of the current sense network
= (see Figures 23 and 24)
L = output inductor
= current sense resistance
i
R
S
5
2
8
(
)( )
0.3 0.55
S
For the application circuit:
e
Instabilities
1.8 µ
In current mode control, an instability can be encountered
at any given duty cycle. The instability is caused by the
= 0.115 V/ms
9
MOTOROLA ANALOG IC DEVICE DATA
MC34023 MC33023
PIN FUNCTION DESCRIPTION
Pin
DIP/SOIC
1
Function
Description
Error Amp
Inverting
Input
This pin is usually used for feedback from the output of the power supply.
2
3
Error Amp
Noninverting
Input
This pin is used to provide a reference in which an error signal can be produced on the output of the
error amp. Usually this is connected to V , however an external reference can also be used.
ref
Error Amp
Output
This pin is provided for compensating the error amp for poles and zeros encountered in the power
supply system, mostly the output LC filter.
4
5
6
7
Clock
This is a bidirectional pin used for synchronization.
R
The value of R sets the charge current through timing Capacitor, C .
T T
T
T
C
In conjunction with R , the timing Capacitor sets the switching frequency.
T
Ramp Input
For voltage mode operation this pin is connected to C . For current mode operation this pin is
T
connected through a filter to the current sensing element.
8
9
Soft–Start
A capacitor at this pin sets the Soft–Start time.
Current Limit/
Shutdown
This pin has two functions. First, it provides cycle–by–cycle current limiting. Second, if the current is
excessive, this pin will reinitiate a Soft–Start cycle.
10
11
Ground
This pin is the ground for the control circuitry.
Current Limit
Reference
Input
This pin voltage sets the threshold for cycle–by–cycle current limiting.
12
13
Power Ground
This is a separate power ground return that is connected back to the power source. It is used to reduce
the effects of switching transient noise on the control circuitry.
V
C
This is a separate power source connection for the outputs that is connected back to the power source
input. With a separate power source connection, it can reduce the effects of switching transient noise
on the control circuitry.
14
15
16
Output
This is a high current totem pole output.
V
CC
This pin is the positive supply of the control IC.
V
ref
This is a 5.1 V reference. It is usually connected to the noninverting input of the error amplifier.
Figure 21. Voltage Mode Operation
Figure 22. Current Mode Operation
4
4
5
5
Oscillator
Oscillator
6
6
C
T
C
T
From Current
Sense Element
1.25 V
7
1.25 V
7
3
1
3
1
Output Voltage
Feedback Input
Output Voltage
Feedback Input
2
V
ref
2
V
ref
Incurrentmodecontrol, anRCfiltershouldbeplacedattherampinput
to filter the leading edge spike caused by turn–on of a power MOSFET.
In voltage mode operation, the control range on the output of the Error
Amplifier from 0% to 90% duty cycle is from 2.25 V to 4.05 V.
10
MOTOROLA ANALOG IC DEVICE DATA
MC34023 MC33023
Figure 23. Resistive Current Sensing
Figure 24. Primary Side Current Sensing
9
R
w
I
9
Sense
I
Sense
The addition of an RC filter will eliminate instability caused by the
leading edge spike on the current waveform. This sense signal can also
be used at the ramp input pin for current mode control. For ramp
compensation it is necessary to know the gain of the current feedback
loop. The gain can be calculated by:
The addition of an RC filter will eliminate instability caused by the
leading edge spike on the current waveform. This sense signal can also
be used at the ramp input pin for current mode control. For ramp
compensation it is necessary to know the gain of the current feedback
loop. If a transformer is used, the gain can be calculated by:
R
w
R
A
Sense
i
turns ratio
A
i
turns ratio
Figure 25A. Slope Compensation (Noise Sensitive)
4
5
Oscillator
6
C
C
T
1
1.25 V
R
2
1
Current Sense
Information
7
3
R
This method of slope compensation is easy to implement, however, it
is noise sensitive. Capacitor C provides AC coupling. The oscillator
1
signal is added to the current signal by a voltage divider consisting of
resistors R and R .
1
2
Figure 25B. Slope Compensation (Noise Immune)
Output
Current Sense
Transformer
Ramp
Input
R
w
R
Ramp
Input
M
1.25 V
7
3
R
1.25 V
f
7
Output
C
M
R
Current Sense
C
M
R
f
f
C
C
M
f
Resistor
3
When only one output is used, this method of slope compensation can be used and it is relatively noise immune. Resistor R and capacitor C provide the added
M
M
slope necessary. By choosing R and C with a larger time constant than the switching frequency, you can assume that its charge is linear. First choose C , then
M
M
M
R
canbeadjustedtoachievetherequiredslope. Thediodeprovidesaresetpulseattherampinputattheendofeverycycle. ThechargecurrentI canbecalculated
M
M
by I = C S . Then R can be calculated by R = V /I
M
M e CC M.
M
M
11
MOTOROLA ANALOG IC DEVICE DATA
MC34023 MC33023
Figure 27. External Clock Synchronization
Figure 26. Dead Time Addition
V
ref
5.0 V
0 V
4
5
6
4
5
6
R
DT
Oscillator
Oscillator
R
T
C
T
R
T
C
T
Additional dead time can be added by the addition of a dead time
resistor from V
information.
to C . See text on Oscillator section for more
ref
T
The sync pulse fed into the clock pin must be at least 3.9 V. R and C
T
T
need to be set 10% slower than the sync frequency. This circuit is also
used in Voltage Mode operation for master/slave operation. The clock
signal would be coming from the master which is set at the desired
operating frequency, while the slave is set 10% slower.
Figure 28. Current Mode Master/Slave Operation Over Short Distances
4
4
5
6
V
5
6
ref
Master
Oscillator
Slave
Oscillator
R
T
C
T
Figure 29. Synchronization Over Long Distances
20
16
Reference
MMBT3906
MMBD0914
1.0 k
4.7 k
4
NC
4
2200
5
6
1.15 R
T
Slave
Oscillator
5
6
430
T
Master
Oscillator
MMBT3904
C
T
R
C
T
12
MOTOROLA ANALOG IC DEVICE DATA
MC34023 MC33023
Figure 30. Buffered Maximum Clamp Level
Figure 31. Bipolar Transistor Drive
I
B
1
+
0
–
V
2
V
C
in
+
Base Charge
Removal
V
ref
8
15
R
R
1
C
SS
14
12
2
R
To Current
Sense Input
S
In voltage mode operation, the maximum duty cycle can be clamped. By the
addition of a PNP transistor to buffer the clamp voltage, the Soft–Start current
The totem pole output can furnish negative base current for enhanced
transistor turn–off, with the addition of the capacitor in series with the base.
is not affected by R .
1
V
0.6
clamp
9.0 µA
The new equation for Soft–Start is
t
(C
)
SS
In current mode operation, this circuit will limit the maximum voltage allowed
at the ramp input to end a cycle.
Figure 32. MOSFET Parasitic Oscillations
Figure 33. Isolated MOSFET Drive
V
V
C
V
in
C
15
14
15
14
12
12
R
To Current
Sense Input
S
A series gate resistor may be needed to dampen high frequency parasitic
oscillation caused by the MOSFET’s input capacitance and any series wiring
inductance in the gate–source circuit. The series resistor will also decrease the
MOSFET switching speed. A Schottky diode can reduce the driver’s power
dissipation due to excessive ringing, by preventing the output pin from being
drivenbelowground. TheSchottkydiodealsopreventssubstrateinjectionwhen
the output pin is driven below ground.
The totem pole output can easily drive pulse transformers. A Schottky diode
is recommended when driving inductive loads at high frequencies. The diode
canreducethedriver’spowerdissipationduetoexcessiveringing,bypreventing
the output pin from being driven below ground.
13
MOTOROLA ANALOG IC DEVICE DATA
MC34023 MC33023
+
14
MOTOROLA ANALOG IC DEVICE DATA
MC34023 MC33023
Figure 35. PC Board With Components
1500 pF
100 pF
4.0
″
1N5819
1000 pF
0.01
0.01
1500 pF
6.5
″
(Top View)
15
MOTOROLA ANALOG IC DEVICE DATA
MC34023 MC33023
Figure 36. PC Board Without Components
(Top View)
4.0
″
6.5
″
(Bottom View)
16
MOTOROLA ANALOG IC DEVICE DATA
MC34023 MC33023
OUTLINE DIMENSIONS
P SUFFIX
PLASTIC PACKAGE
CASE 648–08
NOTES:
–A–
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: INCH.
3. DIMENSION L TO CENTER OF LEADS WHEN
FORMED PARALLEL.
4. DIMENSION B DOES NOT INCLUDE MOLD FLASH.
5. ROUNDED CORNERS OPTIONAL.
16
1
9
8
B
INCHES
MILLIMETERS
DIM
A
B
C
D
F
G
H
J
K
L
MIN
MAX
0.770
0.270
0.175
0.021
0.070
MIN
18.80
6.35
3.69
0.39
1.02
2.54 BSC
1.27 BSC
0.21
MAX
19.55
6.85
4.44
0.53
1.77
F
C
L
0.740
0.250
0.145
0.015
0.040
0.100 BSC
0.050 BSC
0.008
S
SEATING
–T–
PLANE
M
K
0.015
0.130
0.305
0.38
3.30
7.74
H
J
0.110
0.295
2.80
7.50
G
D 16 PL
0.25 (0.010)
M
S
0°
10°
0°
10°
M
M
T
A
0.020
0.040
0.51
1.01
DW SUFFIX
PLASTIC PACKAGE
CASE 751G–02
(SO–16L)
–A–
NOTES:
1. DIMENSIONING AND TOLERANCING PER
ANSI Y14.5M, 1982.
16
1
9
8
2. CONTROLLING DIMENSION: MILLIMETER.
3. DIMENSIONS A AND B DO NOT INCLUDE
MOLD PROTRUSION.
4. MAXIMUM MOLD PROTRUSION 0.15 (0.006)
PER SIDE.
–B– P 8 PL
M
M
0.25 (0.010)
B
5. DIMENSION D DOES NOT INCLUDE
DAMBAR PROTRUSION. ALLOWABLE
DAMBAR PROTRUSION SHALL BE 0.13
(0.005) TOTAL IN EXCESS OF D DIMENSION
AT MAXIMUM MATERIAL CONDITION.
G 14 PL
J
F
INCHES
MIN MAX
0.411 10.15
MILLIMETERS
DIM
A
MIN
MAX
10.45
0.400
0.292
0.093
0.014
0.020
B
C
D
F
0.299
0.104
0.019
0.035
7.40
2.35
0.35
0.50
7.60
2.65
0.49
0.90
R X 45°
G
J
K
M
P
R
0.050 BSC
1.27 BSC
C
0.010
0.004
0.012
0.009
0.25
0.10
0.32
0.25
–T–
SEATING
PLANE
0°
7°
0°
7°
M
K
0.395
0.010
0.415 10.05
0.029 0.25
10.55
0.75
D 16 PL
0.25 (0.010)
M
S
S
T
A
B
17
MOTOROLA ANALOG IC DEVICE DATA
MC34023 MC33023
OUTLINE DIMENSIONS
FN SUFFIX
PLASTIC PACKAGE
CASE 775–02
(PLCC)
M
S
S
S
B
0.007 (0.180)
T
L –M
N
Y BRK
–M–
–N–
D
D
M
S
U
0.007 (0.180)
T
L –M
N
–L–
Z
W
20
1
S
S
S
G1
V
0.010 (0.250)
T
L –M
N
X
VIEW D–D
A
M
M
S
S
S
S
0.007 (0.180)
0.007 (0.180)
T
T
L –M
L –M
N
N
Z
R
M
S
S
H
0.007 (0.180)
T L
–M
N
C
K1
E
K
0.004 (0.100)
SEATING
G
–T–
J
PLANE
M
S
S
F
0.007 (0.180)
T
L –M
N
VIEW S
G1
VIEW S
S
S
S
0.010 (0.250)
T
L –M
N
NOTES:
1. DATUMS –L–, –M–, AND –N– DETERMINED WHERE
TOP OF LEAD SHOULDER EXITS PLASTIC BODY
AT MOLD PARTING LINE.
2. DIM G1, TRUE POSITION TO BE MEASURED AT
DATUM –T–, SEATING PLANE.
3. DIM R AND U DO NOT INCLUDE MOLD FLASH.
ALLOWABLE MOLD FLASH IS 0.010 (0.250) PER
SIDE.
4. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
5. CONTROLLING DIMENSION: INCH.
6. THE PACKAGE TOP MAY BE SMALLER THAN THE
PACKAGE BOTTOM BY UP TO 0.012 (0.300).
DIMENSIONS R AND U ARE DETERMINED AT THE
OUTERMOST EXTREMES OF THE PLASTIC BODY
EXCLUSIVE OF MOLD FLASH, TIE BAR BURRS,
GATE BURRS AND INTERLEAD FLASH, BUT
INCLUDING ANY MISMATCH BETWEEN THE TOP
AND BOTTOM OF THE PLASTIC BODY.
7. DIMENSION H DOES NOT INCLUDE DAMBAR
PROTRUSION OR INTRUSION. THE DAMBAR
PROTRUSION(S) SHALL NOT CAUSE THE H
DIMENSION TO BE GREATER THAN 0.037 (0.940).
THE DAMBAR INTRUSION(S) SHALL NOT CAUSE
THE H DIMENSION TO BE SMALLER THAN 0.025
(0.635).
INCHES
MILLIMETERS
DIM
A
B
C
E
F
G
H
J
K
R
U
V
W
X
Y
Z
G1
K1
MIN
MAX
0.395
0.395
0.180
0.110
0.019
MIN
9.78
9.78
4.20
2.29
0.33
MAX
10.03
10.03
4.57
2.79
0.48
0.385
0.385
0.165
0.090
0.013
0.050 BSC
1.27 BSC
0.026
0.032
–
0.66
0.51
0.64
8.89
8.89
1.07
1.07
1.07
–
0.81
–
–
9.04
9.04
1.21
1.21
1.42
0.50
0.020
0.025
0.350
0.350
0.042
0.042
0.042
–
–
0.356
0.356
0.048
0.048
0.056
0.020
2°
10
°
2°
10°
0.310
0.040
0.330
–
7.88
1.02
8.38
–
18
MOTOROLA ANALOG IC DEVICE DATA
MC34023 MC33023
Motorola reserves the right to make changes without further notice to any products herein. Motorola makes no warranty, representation or guarantee regarding
the suitability of its products for any particular purpose, nor does Motorola assume any liability arising out of the application or use of any product or circuit, and
specificallydisclaims any and all liability, including without limitation consequential or incidental damages. “Typical” parameters which may be provided in Motorola
datasheetsand/orspecificationscananddovaryindifferentapplicationsandactualperformancemayvaryovertime. Alloperatingparameters,including“Typicals”
must be validated for each customer application by customer’s technical experts. Motorola does not convey any license under its patent rights nor the rights of
others. Motorola products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other
applicationsintended to support or sustain life, or for any other application in which the failure of the Motorola product could create a situation where personal injury
ordeathmayoccur. ShouldBuyerpurchaseoruseMotorolaproductsforanysuchunintendedorunauthorizedapplication,BuyershallindemnifyandholdMotorola
and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees
arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that
Motorola was negligent regarding the design or manufacture of the part. Motorola and
Opportunity/Affirmative Action Employer.
are registered trademarks of Motorola, Inc. Motorola, Inc. is an Equal
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