DAC5682Z [TI]
16-BIT, 1.0 GSPS 2x-4x INTERPOLATING DUAL-CHANNEL DIGITAL-TO-ANALOG CONVERTER (DAC); 16位， 1.0 GSPS 2倍， 4倍内插双通道数位类比转换器（ DAC ）
| 型号： | DAC5682Z |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | 16-BIT, 1.0 GSPS 2x-4x INTERPOLATING DUAL-CHANNEL DIGITAL-TO-ANALOG CONVERTER (DAC) |
| 文件： | 总56页 (文件大小：2428K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
DAC5682Z
www.ti.com
SLLS853A–AUGUST 2007–REVISED NOVEMBER 2007
16-BIT, 1.0 GSPS 2x-4x INTERPOLATING DUAL-CHANNEL
DIGITAL-TO-ANALOG CONVERTER (DAC)
1
FEATURES
DESCRIPTION
•
16-Bit Digital-to-Analog Converter (DAC)
•
•
1.0 GSPS Update Rate
The DAC5682Z is a dual-channel 16-bit 1.0 GSPS
digital-to-analog converter (DAC) with wideband
LVDS data input, integrated 2x/4x interpolation filters,
on-board clock multiplier and internal voltage
reference. The DAC5682Z offers superior linearity,
noise, crosstalk and PLL phase noise performance.
16-Bit Wideband Input LVDS Data Bus
–
–
8 Sample Input FIFO
Interleaved I/Q data for Dual-DAC Mode
•
High Performance
73 dBc ACLR WCDMA TM1 at 180 MHz
–
The DAC5682Z integrates a wideband LVDS port
with on-chip termination. Full-rate input data can be
transferred to a single DAC channel, or half-rate and
1/4-rate input data can be interpolated by on-board
2x or 4x FIR filters. Each interpolation FIR is
configurable in either Low-Pass or High-Pass mode,
allowing selection of a higher order output spectral
image. An on-chip delay lock loop (DLL) simplifies
LVDS interfacing by providing skew control for the
LVDS input data clock.
•
•
2x-32x Clock Multiplying PLL/VCO
2x or 4x Interpolation Filters
–
–
Stopband Transition 0.4–0.6 Fdata
Filters Configurable in Either Low-Pass or
High-Pass Mode
–
Allows Selection of Higher Order Image
•
•
•
•
Fs/4 Coarse Mixer
On Chip 1.2 V Reference
The DAC5682Z allows both complex or real output.
An optional Fs/4 coarse mixer in complex mode
provides coarse frequency upconversion and the dual
DAC output produces a complex Hilbert Transform
pair. An external RF quadrature modulator then
performs the final single sideband up-conversion.
Differential Scalable Output: 2 to 20 mA
Package: 64-Pin 9 × 9 mm QFN
APPLICATIONS
•
•
•
•
•
Cellular Base Stations
Broadband Wireless Access (BWA)
WiMAX 802.16
Fixed Wireless Backhaul
Cable Modem Termination System (CMTS)
The DAC5682Z is characterized for operation over
the industrial temperature range of –40°C to 85°C
and is available in a 64-pin QFN package. Other
single-channel members of the family include the
interpolating DAC5681Z
DAC5681.
and non-interpolating
ORDERING INFORMATION
TA
ORDER CODE
DAC5682ZIRGCT
DAC5682ZIRGCR
PACKAGE DRAWING/TYPE(1)(2)(3)
TRANSPORT MEDIA
Small Tape and Reel
Large Tape and Reel
QUANTITY
250
RGC / 64QFN Quad Flatpack
No-Lead
–40°C to 85°C
2000
(1) Thermal Pad Size: 7,4 mm × 7,4 mm
(2) MSL Peak Temperature: Level-3-260C-168 HR
(3) For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI
website at www.ti.com.
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Copyright © 2007, Texas Instruments Incorporated
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
DAC5682Z
www.ti.com
SLLS853A–AUGUST 2007–REVISED NOVEMBER 2007
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
FUNCTIONAL BLOCK DIAGRAM
PLL Bypass
CLKIN
FDAC
EXTIO
1.2V
Reference
Clock Multiplying
PLL 2x-32x
Clock
Distribution
FDAC/2
FDAC/4
EXTLO
BIASJ
CLKINC
DCLKP
PLL Control
DLL Control
PLL Enable
(x2 Bypass)
Sync Disable
Delay Lock
Loop (DLL)
DACA_gain
Mode Control
(x1 Bypass)
DCLKN
4
A-Offset
13
A
B
D15P
D15N
IOUTA1
IOUTA2
16bit
DAC
x2
47t 76dB HBF
x2
47t 76dB HBF
16
16
FIR0
FIR1
D0P
D0N
IOUTB1
IOUTB2
16bit
DAC
x2
47t 76dB HBF
x2
47t 76dB HBF
16
16
2
13
2
2
e
e
SYNC=’0->1'
(transition)
e
l
e
e
u
b
l
4
d
d
SYNCP
SYNCN
a
a
t
o
o
n
e
V
E
M
M
s
f
y
DACB_gain
f
1
0
1
a
l
TXEnable=’1'
O
R
R
M
M
e
-
I
I
F
C
F
C
B
D
SW_Sync
FIFO Sync Disable
Sync & Control
2
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SLLS853A–AUGUST 2007–REVISED NOVEMBER 2007
DAC5682Z
RGC PACKAGE
(TOP VIEW)
1
48
47
46
45
44
43
42
41
40
39
38
37
36
35
34
33
SDENB
SCLK
SDIO
SDO
VFUSE
D0N
CLKVDD
CLKIN
2
3
CLKINC
GND
4
5
SYNCP
6
SYNCN
D15P
7
D0P
8
D15N
D1N
DAC5682Z
9
IOVDD
D1P
10
11
12
13
14
15
16
DVDD
D2N
DVDD
D14P
D14N
D13P
D13N
D12P
D12N
D2P
D3N
D3P
D4N
D4P
TERMINAL FUNCTIONS
TERMINAL
I/O
DESCRIPTION
NAME
AVDD
BIASJ
NO.
51, 54, 55,
59, 62
Analog supply voltage. (3.3V)
I
57
O
Full-scale output current bias. For full-scale output current, connect a 960 Ω resistor to GND.
Positive external clock input with a self-bias of approximately CLKVDD/2. With the clock multiplier PLL
enabled, CLKIN provides lower frequency reference clock. If the PLL is disabled, CLKIN directly provides
clock for DAC up to 1GHz.
CLKIN
2
I
CLKINC
3
1
I
I
Complementary external clock input. (See the CLKIN description)
Internal clock buffer supply voltage. (1.8 V)
CLKVDD
LVDS positive input data bits 0 through 15. Each positive/negative LVDS pair has an internal 100 Ω
termination resistor. Order of bus can be reversed via rev_bus bit in CONFIG5 register. Data format relative
to DCLKP/N clock is Double Data Rate (DDR) with two data samples input per DCLKP/N clock. In
dual-channel mode, data for the A-channel is input while DCLKP is high.
7, 11, 13,
15, 17, 19,
21, 23, 27,
29, 31, 33,
35, 37, 40,
42
D[15..0]P
I
D15P is most significant data bit (MSB) – pin 7
D0P is least significant data bit (LSB) – pin 42
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SLLS853A–AUGUST 2007–REVISED NOVEMBER 2007
TERMINAL FUNCTIONS (continued)
TERMINAL
I/O
DESCRIPTION
NAME
NO.
8, 12, 14,
16, 18, 20,
22, 24, 28,
30, 32, 34,
36, 38, 41,
43
LVDS negative input data bits 0 through 15. (See D[15:0]P description above)
D15N is most significant data bit (MSB) – pin 8
D0N is least significant data bit (LSB) – pin 43
D[15..0]N
I
LVDS positive input clock. Unlike the other LVDS inputs, the DCLKP/N pair is self-biased to approximately
DVDD/2 and does not have an internal termination resistor in order to optimize operation of the DLL circuit.
See the “DLL Operation” section. For proper external termination, connect a 100 Ω resistor across LVDS
clock source lines followed by series 0.01 µF capacitors connected to each of DCLKP and DCLKN pins (see
Figure 26). For best performance, the resistor and capacitors should be placed as close as possible to these
pins.
DCLKP
25
26
I
DCLKN
DVDD
I
I
LVDS negative input clock. (See the DCLKP description)
Digital supply voltage. (1.8 V)
10, 39, 50,
63
Used as external reference input when internal reference is disabled (i.e., EXTLO connected to AVDD).
EXTIO
56
58
I/O Used as 1.2V internal reference output when EXTLO = GND, requires a 0.1 µF decoupling capacitor to
AGND when used as reference output.
EXTLO
GND
O
I
Internal reference ground. Connect to AVDD to disable the internal reference.
4, Thermal
Pad
Pin 4 and the Thermal Pad located on the bottom of the QFN package is ground for AVDD, DVDD and
IOVDD supplies.
A-Channel DAC current output. An offset binary data pattern of 0x0000 at the DAC input results in a full
scale current sink and the least positive voltage on the IOUTA1 pin. Similarly, a 0xFFFF data input results in
a 0 mA current sink and the most positive voltage on the IOUTA1 pin. In single DAC mode, outputs appear
on the IOUTA1/A2 pair only.
IOUTA1
52
O
A-Channel DAC complementary current output. The IOUTA2 has the opposite behavior of the IOUTA1
described above. An input data value of 0x0000 results in a 0mA sink and the most positive voltage on the
IOUTA2 pin.
IOUTA2
53
O
IOUTB1
IOUTB2
IOVDD
61
60
9
O
O
I
B-Channel DAC current output. See the IOUTA1 description above.
B-Channel DAC complementary current output. See the IOUTA2 description above.
Digital I/O supply voltage (3.3V) for pins RESETB, SCLK, SDENB, SDIO, SDO.
PLL loop filter connection. If not using the clock multiplying PLL, the LPF pin may be left open. Set both
PLL_bypass and PLL_sleep control bits for reduced power dissipation.
LPF
64
I
RESETB
SCLK
49
47
48
46
I
I
I
Resets the chip when low. Internal pull-up.
Serial interface clock. Internal pull-down.
SDENB
SDIO
Active low serial data enable, always an input to the DAC5682Z. Internal pull-up.
I/O Serial interface data, bi-directional. Default setting sets SDIO as an input. Internal pull-down.
Serial interface data, uni-directional data output, if SDIO is an input. SDO is 3-stated when the 3 pin interface
mode is selected (register 0x08 bit 1). Internal pull-down.
SDO
45
O
LVDS SYNC positive input data. The SYNCP/N LVDS pair has an internal 100 Ω termination resistor. By
SYNCP
5
I
default, the SYNCP/N input must be logic ‘1’ to enable a DAC analog output. See the LVDS SYNCP/N
Operation paragraph for a detailed description.
SYNCN
VFUSE
6
I
I
LVDS SYNC negative input data.
Digital supply voltage. (1.8V) Connect to DVDD pins for normal operation. This supply pin is also used for
factory fuse programming.
44
4
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SLLS853A–AUGUST 2007–REVISED NOVEMBER 2007
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature range (unless otherwise noted)
(1)
VALUE
–0.5 to 2.3
UNIT
V
DVDD(2)
CLKVDD(2)
AVDD(2)
–0.5 to 2.3
V
Supply voltage range
–0.5 to 4
V
IOVDD(2)
Voltage between GND, and CLKGND
AVDD to DVDD
–0.5 to 4
V
–0.5 to 0.5
V
–2 to 2.6
V
CLKVDD to DVDD
–0.5 to 0.5
V
IOVDD to AVDD
–0.5 to 0.5
V
(2)
D[15..0]P ,D[15..0]N, SYNCP, SYNCN
DCLKP, DCLKN(2)
–0.5 to DVDD + 0.5
–0.3 to 2.1
V
Supply voltage range
V
CLKIN, CLKINC(2)
–0.5 to CLKVDD + 0.5
–0.5 to IOVDD + 0.5
–0.5 to AVDD + 0.5
–0.5 to AVDD + 0.5
20
V
(2)
SDO, SDIO, SCLK, SDENB, RESETB
V
(2)
IOUTA1/B1, IOUTA2/B2
LPF, EXTIO, EXTLO, BIASJ(2)
V
V
Peak input current (any input)
mA
mA
°C
°C
Peak total input current (all inputs)
–30
Operating free-air temperature range, TA: DAC5682ZI
Storage temperature range
–40 To 85
–65 To 150
(1) Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of these or any other conditions beyond those indicated under recommended operating conditions is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
(2) Measured with respect to GND.
THERMAL CHARACTERISTICS
over operating free-air temperature range (unless otherwise noted)
THERMAL CONDUCTIVITY
64ld QFN
UNIT
(1)
TJ
Maximum junction temperature
125
20
16
7
°C
Theta junction-to-ambient (still air)
Theta junction-to-ambient (150 lfm)
Theta junction-to-case
θJA
°C/W
θJC
θJP
°C/W
°C/W
Theta junction-to-pad
0.2
(1) Air flow or heat sinking reduces θJA and may be required for sustained operation at 85° under maximum operating conditions.
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SLLS853A–AUGUST 2007–REVISED NOVEMBER 2007
ELECTRICAL CHARACTERISTICS — DC SPECIFICATION
over operating free-air temperature range , AVDD = 3.3 V, CLKVDD = 1.8 V, IOVDD = 3.3 V, DVDD = 1.8 V, IoutFS = 20 mA
(unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
Resolution
DC ACCURACY(1)
16
Bits
INL
Integral nonlinearity
Differential nonlinearity
1 LSB = IOUTFS/216
±4
±2
LSB
DNL
ANALOG OUTPUT
Course gain linearity
±0.04
0.01
1
LSB
Offset error
Mid code offset
%FSR
%FSR
%FSR
%FSR
Gain error
Without internal reference
With internal reference
Gain error
0.7
Gain mismatch
With internal reference, dual DAC, SSB mode
–2
2
Minimum full scale output current(2)
Maximum full scale output current(2)
2
mA
V
20
AVDD
–0.5V
AVDD
+ 0.5V
Output Compliance range(3)
IOUTFS = 20 mA
Output resistance
Output capacitance
300
5
kΩ
pF
REFERENCE OUTPUT
Vref
Reference voltage
Reference output current(4)
1.14
0.1
1.2
1.26
1.25
V
100
nA
REFERENCE INPUT
VEXTIO Input voltage range
V
Input resistance
1
95
MΩ
CONFIG6: BiasLPF_A and BiasLPF = 1
CONFIG6: BiasLPF_A and BiasLPF = 0
Small signal bandwidth
Input capacitance
kHz
pF
472
100
TEMPERATURE COEFFICIENTS
ppm of
FSR/°C
Offset drift
±1
Without internal reference
With internal reference
±15
±30
±8
ppm of
FSR/°C
Gain drift
Reference voltage drift
POWER SUPPLY
ppm/°C
Analog supply voltage, AVDD
Digital supply voltage, DVDD
Clock supply voltage, CLKVDD
I/O supply voltage, IOVDD
3.0
1.71
1.71
3.0
3.3
1.8
1.8
3.3
133
455
45
3.6
2.15
2.15
3.6
V
V
V
V
I(AVDD)
I(DVDD)
Analog supply current
Digital supply current
mA
mA
mA
mA
Mode 4 (below)
I(CLKVDD) Clock supply current
I(IOVDD) IO supply current
12
(1) Measured differential across IOUTA1 and IOUTA2 or IOUTB1 and IOUTB2 with 25 Ω each to AVDD.
(2) Nominal full-scale current, IoutFS, equals 16 × IBIAS current.
(3) The lower limit of the output compliance is determined by the CMOS process. Exceeding this limit may result in transistor breakdown,
resulting in reduced reliability of the DAC5682Z device. The upper limit of the output compliance is determined by the load resistors and
full-scale output current. Exceeding the upper limit adversely affects distortion performance and integral nonlinearity.
(4) Use an external buffer amplifier with high impedance input to drive any external load.
6
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SLLS853A–AUGUST 2007–REVISED NOVEMBER 2007
ELECTRICAL CHARACTERISTICS — DC SPECIFICATION (continued)
over operating free-air temperature range , AVDD = 3.3 V, CLKVDD = 1.8 V, IOVDD = 3.3 V, DVDD = 1.8 V, IoutFS = 20 mA
(unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
1.0
MAX
UNIT
mA
I(AVDD)
I(DVDD)
Sleep mode, AVDD supply current
Sleep mode, DVDD supply current
1.5
mA
Mode 6 (below)
Sleep mode, CLKVDD supply
current
I(CLKVDD)
I(IOVDD)
2.5
mA
Sleep mode, IOVDD supply current
AVDD + IOVDD current, 3.3V
DVDD + CLKVDD current, 1.8V
Power Dissipation
2.0
135
450
1255
145
485
1350
135
480
1310
145
505
1400
5
mA
mA
mA
mW
mA
mA
mW
mA
mA
mW
mA
mA
mW
mA
mA
mW
mA
mA
mW
Mode 1: 2X2, PLL = OFF, CLKIN = 983.04 MHz
FDAC = 983.04MHz, IF = 184.32 MHz
DACA and DACB ON, 4 carrier WCDMA
AVDD + IOVDD current, 3.3V
DVDD + CLKVDD current, 1.8V
Power Dissipation
Mode 2: 2X2, PLL = ON (8X), CLKIN = 122.88
MHz
FDAC = 983.04MHz, IF = 184.32 MHz
DACA and DACB ON, 4 carrier WCDMA
AVDD + IOVDD current, 3.3V
DVDD + CLKVDD current, 1.8V
Power Dissipation
Mode 3: 2X4, CMIX0 = Fs/4, PLL = OFF, CLKIN =
983.04 MHz
FDAC = 983.04MHz, IF = 215.04 MHz
DACA and DACB ON, 4 carrier WCDMA
P
AVDD + IOVDD current, 3.3V
DVDD + CLKVDD current, 1.8V
Power Dissipation
Mode 4: 2X4, CMIX0 = Fs/4, PLL = ON (8X),
CLKIN = 122.88 MHz
FDAC = 983.04MHz, IF = 215.04 MHz
DACA and DACB ON, 4 carrier WCDMA
1600
AVDD + IOVDD current, 3.3V
DVDD + CLKVDD current, 1.8V
Power Dissipation
Mode 5: 2X2, CMIX0 = Fs/4, PLL = OFF, CLKIN =
983.04 MHz
FDAC = 983.04MHz, Digital Logic Disabled
DACA and DACB SLEEP, Static Data Pattern
185
350
3.0
AVDD + IOVDD current, 3.3V
DVDD + CLKVDD current, 1.8V
Power Dissipation
Mode 6: 2X4, PLL = OFF, CLKIN = OFF
FDAC = OFF, Digital Logic Disabled
DACA and DACB = SLEEP, Static Data Pattern
4.0
17.0
30.0
0.2 %FSR/V
85 °C
PSSR
T
Power supply rejection ratio
Operating range
–2
–40
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SLLS853A–AUGUST 2007–REVISED NOVEMBER 2007
ELECTRICAL CHARACTERISTICS — AC SPECIFICATION(1)
Over recommended operating free-air temperature range, AVDD, IOVDD = 3.3 V, CLKVDD, DVDD = 1.8 V, IOUTFS = 20 mA,
4:1 transformer output termination, 50Ω doubly terminated load (unless otherwise noted)
PARAMETER
ANALOG OUTPUT
TEST CONDITIONS
MIN
TYP MAX UNIT
Maximum output update
rate
fCLK
1000
MSPS
Output settling time to
0.1%
Transition: Code 0x0000 to 0xFFFF
ts(DAC)
tpd
10.4
3
ns
ns
ns
Output propagation delay
Output rise time 10% to
90%
tr(IOUT)
2
Output fall time 90% to
10%
tf(IOUT)
2
ns
AC PERFORMANCE
1X1, PLL off, CLKIN = 500 MHz, DACA on, IF = 5.1 MHz,
First Nyquist Zone < fDATA/2
81
80
77
73
69
63
60
60
73
81
72
64
64
83
73
66
63
93
85
Spurious free dynamic
range
2X2, PLL off, CLKIN = 1000 MHz, DACA and DACB on,
IF = 5.1 MHz, First Nyquist Zone < fDATA/2
SFDR
dBc
2X2, PLL off, CLKIN = 1000 MHz, DACA and DACB on,
IF = 20.1 MHz, First Nyquist Zone < fDATA/2
2X2, PLL off, CLKIN = 500 MHZ, DACA and DACB on, Single tone,
0 dBFS, IF = 20.1 MHz
2X2, PLL off, CLKIN = 1000 MHZ, DACA and DACB on, Single
tone, 0 dBFS, IF = 20.1 MHz
2X2, PLL off, CLKIN = 1000 MHZ, DACA and DACB on, Single
tone, 0 dBFS, IF = 70.1 MHz
SNR
Signal-to-noise ratio
dBc
2X4, PLL off, CLKIN = 1000 MHZ, DACA and DACB on, Single
tone, 0 dBFS, IF = 180 MHz
2X2 CMIX, PLL off, CLKIN = 1000 MHZ, DACA and DACB on,
Single tone, 0 dBFS, IF = 300.2 MHz
2X2, PLL off, CLKIN = 1000 MHZ, DACA and DACB on, Four tone,
each -12 dBFS, IF = 24.7, 24.9, 25.1 and 25.3 MHz
2X2, PLL off, CLKIN = 1000 MHZ, DACA and DACB on,
IF = 20.1 and 21.1 MHz
Third-order two-tone
intermodulation (each
tone at –6 dBFS)
2X2, PLL off, CLKIN = 1000 MHZ, DACA and DACB on,
IF = 70.1 and 71.1 MHz
IMD3
IMD
dBc
dBc
2X2 CMIX, PLL off, CLKIN = 1000 MHZ, DACA and DACB on,
IF = 150.1 and 151.1 MHz
Four-tone intermodulation 2X2 CMIX, PLL off, CLKIN = 1000 MHz, DACA and DACB on,
(each tone at –12 dBFS)
fOUT = 298.4, 299.2, 300.8 and 301.6 MHz
Single carrier, baseband, 2X2, PLL off, CLKIN = 983.04 MHz,
DACA and DACB on(3)
80
Single carrier, IF = 180 MHz, 2X2, PLL off,
CLKIN = 983.04 MHz, DACA and DACB on
Adjacent channel leakage
ratio
ACLR(2)
dBc
dBc
Four carrier, IF = 180 MHz, 2X2 CMIX, PLL off,
CLKIN = 983.04 MHz, DACA and DACB on
Four carrier, IF = 275 MHz, 2X2 CMIX, PLL off,
CLKIN = 983.04 MHz, DACA and DACB on
50-MHz offset, 1-MHz BW, Single Carrier, baseband, 2X2, PLL off,
CLKIN = 983.04
Noise floor(4)
50-MHz offset, 1-MHz BW, Four Carrier, baseband, 2X2, PLL off,
CLKIN = 983.04.
(1) Measured single-ended into 50 Ω load.
(2) W-CDMA with 3.84 MHz BW, 5-MHz spacing, centered at IF. TESTMODEL 1, 10 ms
(3) Valid over 25°C to 85°C
(4) Carrier power measured in 3.84 MHz BW.
8
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SLLS853A–AUGUST 2007–REVISED NOVEMBER 2007
ELECTRICAL CHARACTERISTICS (DIGITAL SPECIFICATIONS)
over recommended operating free-air temperature range, AVDD, IOVDD = 3.3V, CLKVDD, DVDD = 1.8V.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX UNIT
LVDS INTERFACE: D[15:0]P, D[15:0]N, SYNCP/N, DCLKP/N
Positive-going
175
VITH+
differential input
voltage threshold
mV
mV
Negative-going
differential input
voltage threshold
–175
1.2
VITH–
Input Common
Mode
SYNCP/N, D[15:0]P and D[15:0]N only
DCLKP/N only
VCOM1
V
V
Input Common
Mode
DVDD/
2
VCOM2
ZT
Internal termination SYNCP/N, D[15:0]P and D[15:0]N only
100
110
120
Ω
LVDS Input
capacitance
CL
2
pF
DCLKP/N: 0 to 125MHz (see Figure 33)
DCLK to Data
Setup_min
Hold_min
Positive
Negative
Positive
Negative
Positive
Negative
Positive
Negative
Positive
Negative
Positive
Negative
Positive
Negative
Positive
1100
–600
1000
–1800
800
tS, tH
CONFIG5 DLL_bypass = 1, CONFIG10 = '00000000'
DCLKP/N = 150MHz
CONFIG5 DLL_bypass = 0,
DCLKP/N = 200MHz
CONFIG5 DLL_bypass = 0
–1300
600
DCLKP/N = 250MHz
CONFIG5 DLL_bypass = 0
–1000
450
DCLK to Data
Skew(1)
DCLKP/N = 300MHz
CONFIG5 DLL_bypass = 0
ps
–800
400
tSKEW(A), [Please contact
DCLKP/N = 350 MHz
tSKEW(B)
factory for
CONFIG5 DLL_bypass = 0
recommended DLL
settings]
–700
300
DCLKP/N = 400 MHz CONFIG5 DLL_bypass = 0
Refer to supported data rate [fDATA
]
–600
300
DCLKP/N = 450 MHz CONFIG5 DLL_bypass = 0
Refer to supported data rate [fDATA
]
–500
350
DCLKP/N = 500 MHz, T = 25 °C to 85 °C CONFIG5
DLL_bypass = 0
Negative
–300
Refer to supported data rate [fDATA
]
DLL Enabled, T = 25 °C to 85 °C
DDR format, DCLKP frequency: 125 to 500 MHz
250
250
1000
Input data rate
supported
DLL Enabled, T = –40 °C
DDR format, DCLK frequency: 125 to 375 MHz
fDATA
750 MSPS
250
DLL Disabled, T = –40 °C to 85 °C
DDR format, DCLKP frequency: <125 MHz
CMOS INTERFACE: SDO, SDIO, SCLK, SDENB, RESETB
High-level input
voltage
2
0
3
0
VIH
V
Low-level input
voltage
0.8
V
VIL
High-level input
current
–40
–40
40
IIH
µA
Low-level input
current
40
IIL
µA
CMOS Input
capacitance
5
CI
pF
(1) Positive skew: Clock ahead of data.
Negative skew: Data ahead of clock.
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ELECTRICAL CHARACTERISTICS (DIGITAL SPECIFICATIONS) (continued)
over recommended operating free-air temperature range, AVDD, IOVDD = 3.3V, CLKVDD, DVDD = 1.8V.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX UNIT
IOVDD
– 0.2
Iload = –100 µA
V
VOH
SDO, SDIO
0.8 x
IOVDD
Iload = –2mA
Iload = 100 µA
Iload = 2 mA
V
0.2
V
V
VOL
SDO, SDIO
Setup time,
0.22 x
IOVDD
ts(SDENB) SDENB to rising
edge of SCLK
20
10
5
ns
ns
ns
Setup time, SDIO
ts(SDIO)
valid to rising edge
of SCLK
Hold time, SDIO
valid to rising edge
of SCLK
th(SDIO)
t(SCLK)
t(SCLKH)
t(SCLK)
Period of SCLK
100
40
ns
ns
ns
High time of SCLK
Low time of SCLK
40
Data output delay
after falling edge of
SCLK
td(Data)
10
25
ns
ns
Minimum RESETB
pulse width
tRESET
CLOCK INPUT (CLKIN/CLKINC)
Duty cycle
50%
1
Differential voltage
0.5
V
V
CLKIN/CLKINC
input common
mode
CLKVD
D/2
PHASE LOCKED LOOP
DAC output at 600 kHz offset, 100 MHz, 0-dBFS tone,
2X4, fDATA = 250 MSPS, CLKIN/C = 250 MHz,
PLL_m = '00111', PLL_n = '001', VCO_div2 = 0,
PLL_range = '1111', PLL_gain = '00'
–125
–146
Phase noise
dBc/ Hz
DAC output at 6 MHz offset, 100 MHz, 0-dBFS tone,
2X4, fDATA = 250 MSPS, CLKIN/C = 250 MHz,
PLL_m = '00111', PLL_n = '001', VCO_div2 = 0,
PLL_range = '1111', PLL_gain = '00'
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ELECTRICAL CHARACTERISTICS (DIGITAL SPECIFICATIONS) (continued)
over recommended operating free-air temperature range, AVDD, IOVDD = 3.3V, CLKVDD, DVDD = 1.8V.
PARAMETER
TEST CONDITIONS
MIN
TYP
220
300
260
240
210
270
250
240
220
210
250
230
220
210
200
MAX UNIT
PLL_gain = '00', PLL_range = '0000' (0)
160
290
MHz
MHz/V
MHz
PLL_gain = '01', PLL_range = '0001' (1)
PLL_gain = '01', PLL_range = '0010' (2)
PLL_gain = '01', PLL_range = '0011' (3)
PLL_gain = '01', PLL_range = '0100' (4)
PLL_gain = '10', PLL_range = '0101' (5)
PLL_gain = '10', PLL_range = '0110' (6)
PLL_gain = '10', PLL_range = '0111' (7)
PLL_gain = '10', PLL_range = '1000' (8)
PLL_gain = '10', PLL_range = '1001' (9)
PLL_gain = '11', PLL_range = '1010' (A)
PLL_gain = '11', PLL_range = '1011' (B)
PLL_gain = '11', PLL_range = '1100' (C)
PLL_gain = '11', PLL_range = '1101' (D)
PLL_gain = '11', PLL_range = '1110' (E)
PLL_gain = '11', PLL_range = '1111' (F)
290
400
480
560
620
690
740
790
840
880
920
960
1000
1030
1060
460
MHz/V
MHz
520
MHz/V
MHz
570
MHz/V
MHz
620
MHz/V
MHz
740
MHz/V
MHz
780
MHz/V
MHz
820
PLL/VCO
Operating
Frequency,
Typical VCO Gain
MHz/V
MHz
850
MHz/V
MHz
880
MHz/V
MHz
940
MHz/V
MHz
990
MHz/V
MHz
1020
1040
1070
1090
MHz/V
MHz
MHz/V
MHz
MHz/V
MHz
190
160
MHz/V
PFD Maximum
Frequency
MHz
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TYPICAL CHARACTERISTICS
10
10
8
8
6
6
4
4
2
0
2
0
-2
-4
-6
-2
-4
-6
-8
-8
-10
-10
0
10000 20000 30000 40000 50000 60000
0
10000 20000 30000 40000 50000 60000
Code
Code
Figure 1. Integral Nonlinearity
Figure 2. Differential Nonlinearity
10
10
0
F
F
I
= 250 MSPS,
F
F
I
= 250 MSPS,
data
data
0
-10
-20
-30
-40
-50
= 20 MHz Complex,
= 20 MHz Complex,
IN
IN
= 270 MHz,
= 20 MHz,
-10
-20
-30
-40
-50
F
F
x4 Interpolation
PLL Off
x4 Interpolation
CMIX F /4
S
PLL Off
-60
-70
-60
-70
-80
-90
-80
-90
0
50 100 150 200 250 300 350 400 450 500
f - Frequency - MHz
0
50 100 150 200 250 300 350 400 450 500
f - Frequency - MHz
Figure 3. Single-Tone Spectral Plot
Figure 4. Single-Tone Spectral Plot
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TYPICAL CHARACTERISTICS (continued)
95
90
85
80
75
70
10
F
F
= 250 MSPS,
F
= 250 MSPS,
data
data
x4 Interpolation,
PLL Off
0
= -80 MHz Complex,
IN
(-80+250=170)
= 170 MHz, CMIX, F /4
-10
I
F
S
x4 Interpolation
PLL Off
-20
-30
-6 dBFS
-12 dBFS
-40
-50
-60
-70
-80
0 dBFS
65
60
0
10
20
30
40
50
0
50 100 150 200 250 300 350 400 450 500
f - Frequency - MHz
IF - Intermediate Frequency - MHz
Figure 5. Single-Tone Spectral Plot
Figure 6. In-Band SFDR vs IF
90
85
90
F
= 250 MSPS,
F
= 250 MSPS,
data
x4 Interpolation,
PLL Off
data
x4 Interpolation,
PLL Off
85
80
75
70
65
60
55
50
80
75
70
65
60
55
50
45
40
0
40
80 120 160 200 240 280 320
IF - Intermediate Frequency - MHz
0
50 100 150 200 250 300 350 400 450 500
IF - Intermediate Frequency - MHz
Figure 7. Out-Of-Band SFDR vs IF
Figure 8. Two Tone IMD vs Output Frequency
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TYPICAL CHARACTERISTICS (continued)
0
85
80
89.5 and 90.5 MHz
(CMIX Off)
F
F
I
= 500 MSPS,
data
-10
= 40 0.5 MHz Real,
IN
= 40 MHz,
F
-20
-30
-40
-50
-60
-70
-80
x2 Interpolation
PLL Off
75
70
65
Shift to 340 MHz
(F /8 On)
S
Shift to 215 MHz
(F /4 On)
60
55
S
F
F
= 250 MSPS
data
= 90 MHz 0.5 MHz Compleꢀ
in
50
45
ꢀ4 Interpolation, PLL Off
Three modes: CMIX, F /8, and F /4
-90
S
S
-100
35 36 37 38 39 40 41 42 43 44 45
f - Frequency - MHz
-30
-25
-20
Amplitude - dBFS
Figure 9. Two Tone IMD vs Amplitude
-15
-10
-5
0
Figure 10. Two-Tone IMD Spectral Plot
10
0
85
80
Fdata = 500 MSPS,
= 0 0.5 MHz,
Fdata = 491.52 MSPS,
= I
F
IN
F
IN
F
I
= 250 MHz (Fs/4)
F
x2 Interpolation
PLL Off
-10
-20
-30
-40
-50
-60
-70
-80
-90
PLL Off
75
PLL On
70
65
0
61.44
122.88
184.32
245.76
248.5 249.0 249.5 250.0 250.5 251.0 251.5 252.0 252.5
f - Frequency - MHz
f - Frequency - MHz
Figure 11. Two-Tone IMD Spectral Plot
Figure 12. Single Carrier W-CDMA Test Model 1
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TYPICAL CHARACTERISTICS (continued)
-20
-20
-30
-40
-50
Carrier Power: -7.60 dBm,
Carrier Power: -7.60 dBm,
ACLR (5 MHz): 77.49 dB,
ACLR (10 MHz): 82.45 dB,
Fdata = 245.76 MSPS,
ACLR (5 MHz): 80.66 dB,
ACLR (10 MHz): 82.61 dB,
Fdata = 245.76 MSPS,
IF = 61.44 MHz,
-30
-40
-50
-60
-70
-80
-90
-100
IF = 61.44 MHz,
x4 Interpolation
PLL Off
x4 Interpolation
PLL On
-60
-70
-80
-90
-100
-110
-120
-110
-120
48.9
53.9
58.9 63.9
f - Frequency - MHz
68.9
73.9
48.9
53.9
58.9
63.9
68.9
73.9
f - Frequency - MHz
Figure 13. Single Carrier W-CDMA Test Model 1
Figure 14. Single Carrier W-CDMA Test Model 1
-20
-30
-20
-30
Carrier Power: -8.66 dBm,
ACLR (5 MHz): 73.19 dB,
ACLR (10 MHz): 80.07 dB,
Fdata = 491.52 MSPS,
IF = 184.32 MHz,
Carrier Power: -8.66 dBm,
ACLR (5 MHz): 68.61 dB,
ACLR (10 MHz): 75.91 dB,
Fdata = 491.52 MSPS,
IF = 184.32 MHz,
-40
-50
-40
x2 Interpolation
PLL Off
x2 Interpolation
PLL On
-50
-60
-60
-70
-70
-80
-80
-90
-90
-100
-100
-110
-120
-110
-120
172
177
182 187
f - Frequency - MHz
192
197
172
177
182
187
192
197
f - Frequency - MHz
Figure 15. Single Carrier W-CDMA Test Model 1
Figure 16. Single Carrier W-CDMA Test Model 1
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TYPICAL CHARACTERISTICS (continued)
-20
-30
-40
-50
-60
-20
-30
-40
-50
-60
-70
-80
-90
-100
Carrier Power: -8.99 dBm,
ACLR (5 MHz): 68.22 dB,
ACLR (10 MHz): 74.15 dB,
Fdata = 245.76 MSPS,
IF = Baseband,
Carrier Power: -8.99 dBm,
ACLR (5 MHz): 64.23 dB,
ACLR (10 MHz): 71.27 dB,
Fdata = 245.76 MSPS,
IF = Baseband,
x4 Interpolation
CMIX PLL Off
x4 Interpolation
CMIX PLL On
-70
-80
-90
-100
-110
-120
-110
-120
233
238
243
248
253
258
233
238
243
248
253
258
f - Frequency - MHz
f - Frequency - MHz
Figure 17. Single Carrier W-CDMA Test Model 1
Figure 18. Single Carrier W-CDMA Test Model 1
-20
-30
-20
-30
Carrier Power: -11.98 dBm,
ACLR (5 MHz): 69.74 dB,
ACLR (10 MHz): 75.41 dB,
Fdata = 491.52 MSPS,
IF = 184.32 MHz,
Carrier Power: -11.98 dBm,
ACLR (5 MHz): 66.16 dB,
ACLR (10 MHz): 72.84 dB,
Fdata = 491.52 MSPS,
IF = 184.32 MHz,
-40
-50
-60
-40
x2 Interpolation
PLL Off
x2 Interpolation
PLL On
-50
-60
-70
-80
-70
-80
-90
-90
-100
-110
-120
-100
-110
-120
160 165 170 175 180 185 190 195 200 205 210
f - Frequency - MHz
160 165 170 175 180 185 190 195 200 205 210
f - Frequency - MHz
Figure 19. Two Carrier W-CDMA Test Model 1
Figure 20. Two Carrier W-CDMA Test Model 1
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TYPICAL CHARACTERISTICS (continued)
-20
-20
Carrier Power: -15.85 dBm,
Carrier Power: -15.85 dBm,
ACLR (5 MHz): 65.85 dB,
ACLR (10 MHz): 69.60 dB,
Fdata = 491.52 MSPS,
IF = 184.32 MHz,
ACLR (5 MHz): 69.66 dB,
ACLR (10 MHz): 70.65 dB,
Fdata = 491.52 MSPS,
IF = 184.32 MHz,
-30
-40
-50
-30
-40
x2 Interpolation
PLL Off
x2 Interpolation
PLL On
-50
-60
-70
-80
-90
-60
-70
-80
-90
-100
-110
-120
-100
-110
-120
160 165 170 175 180 185 190 195 200 205 210
f - Frequency - MHz
160 165 170 175 180 185 190 195 200 205 210
f - Frequency - MHz
Figure 21. Four Carrier W-CDMA Test Model 1
Figure 22. Four Carrier W-CDMA Test Model 1
-20
-20
Carrier Power: -15.20 dBm,
ACLR (5 MHz): 71.18 dB,
ACLR (10 MHz): 72.26 dB,
Fdata = 491.52 MSPS,
IF = 184.32 MHz,
Carrier Power: -15.20 dBm,
ACLR (5 MHz): 66.53 dB,
-30 ACLR (10 MHz): 69.68 dB,
Fdata = 491.52 MSPS,
-30
IF = 184.32 MHz,
x2 Interpolation
PLL On
-40
-50
-40
x2 Interpolation
PLL Off
-50
-60
-60
-70
-80
-90
-70
-80
-90
-100
-100
-110
-120
-110
-120
160 165 170 175 180 185 190 195 200 205 210
f - Frequency - MHz
160 165 170 175 180 185 190 195 200 205 210
f - Frequency - MHz
Figure 23. Three Carrier W-CDMA Test Model 1 with Gap
Figure 24. Three Carrier W-CDMA Test Model 1 with Gap
TEST METHODOLOGY
Typical AC specifications were characterized with the DAC5682ZEVM using the test configuration shown in
Figure 25. A sinusoidal master clock frequency is generated by an HP8665B signal generator and into a splitter.
One output drives an Agilent 8133A pulse generator, and the other drives the CDCM7005 clock driver. The
8133A converts the sinusoidal frequency into a square wave output clock and drives an Agilent ParBERT
81250A pattern-generator clock. On the EVM, the DAC5682Z CLKIN/C input clock is driven by an CDCM7005
clock distribution chip that is configured to simply buffer the external 8665B clock or divide it down for PLL test
configurations.
The DAC5682Z output is characterized with a Rohde and Schwarz FSU spectrum analyzer. For WCDMA signal
characterization, it is important to use a spectrum analyzer with high IP3 and noise subtraction capability so that
the spectrum analyzer does not limit the ACPR measurement. For all specifications, both DACA and DACB are
measured and the lowest value used as the specification.
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DAC5682ZEVM SMA Adapter Board
DAC5682Z DAC
100
Agilent 81205A
ParBERT
3.3 V
100
D15
P
N
3.3 V
3.3 V
Rohde &
Schwartz
FSU
Spectrum
Analyzer
DAC-A
DAC-B
D0
P
N
3.3 V
100
100
100
100
3.3 V
Pattern
Memory
SYNC
DCLK
P
N
Swap Cable
For DAC-B
measurements
100
P
N
3.3 V
100
DLL
opt.
PLL
36 each
SMA-SMA cables
Loop
Filter
CDCM7005
Opt.
Clock
Divider
100
Optional
Divider
DAC5682ZEVM
HP8665B
Synthesized
Signal
Agilent 8133A
Pulse Generator
Generator
Figure 25. DAC5682Z Test Configuration for Normal Clock Mode
DEFINITION OF SPECIFICATIONS
Adjacent Carrier Leakage Ratio (ACLR): Defined for a 3.84Mcps 3GPP W-CDMA input signal measured in a
3.84MHz bandwidth at a 5MHz offset from the carrier with a 12dB peak-to-average ratio.
Analog and Digital Power Supply Rejection Ratio (APSSR, DPSSR): Defined as the percentage error in the
ratio of the delta IOUT and delta supply voltage normalized with respect to the ideal IOUT current.
Differential Nonlinearity (DNL): Defined as the variation in analog output associated with an ideal 1 LSB
change in the digital input code.
Gain Drift: Defined as the maximum change in gain, in terms of ppm of full-scale range (FSR) per °C, from the
value at ambient (25°C) to values over the full operating temperature range.
Gain Error: Defined as the percentage error (in FSR%) for the ratio between the measured full-scale output
current and the ideal full-scale output current.
Integral Nonlinearity (INL): Defined as the maximum deviation of the actual analog output from the ideal output,
determined by a straight line drawn from zero scale to full scale.
Intermodulation Distortion (IMD3, IMD): The two-tone IMD3 or four-tone IMD is defined as the ratio (in dBc) of
the worst 3rd-order (or higher) intermodulation distortion product to either fundamental output tone.
Offset Drift: Defined as the maximum change in DC offset, in terms of ppm of full-scale range (FSR) per °C,
from the value at ambient (25°C) to values over the full operating temperature range.
Offset Error: Defined as the percentage error (in FSR%) for the ratio of the differential output current
(IOUT1–IOUT2) and the mid-scale output current.
Output Compliance Range: Defined as the minimum and maximum allowable voltage at the output of the
current-output DAC. Exceeding this limit may result reduced reliability of the device or adversely affecting
distortion performance.
Reference Voltage Drift: Defined as the maximum change of the reference voltage in ppm per degree Celsius
from value at ambient (25°C) to values over the full operating temperature range.
Spurious Free Dynamic Range (SFDR): Defined as the difference (in dBc) between the peak amplitude of the
output signal and the peak spurious signal.
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Signal to Noise Ratio (SNR): Defined as the ratio of the RMS value of the fundamental output signal to the
RMS sum of all other spectral components below the Nyquist frequency, including noise, but excluding the first
six harmonics and dc.
TYPICAL APPLICATION SCHEMATIC
(1) Power supply decoupling capacitors not shown.
(2) Internal Reference configuration shown.
Figure 26. Schematic
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DETAILED DESCRIPTION
The primary modes of operation, listed in Table 1, are selected by registers CONFIG1, CONFIG2 and CONFIG3.
Table 1. DAC5682Z Modes of Operation
LVDS
Input
Data
Max Total
Input Bus
Rate
No. of
FIR0,
FIR1,
Max DCLK
Max CLKIN Freq [DDR]
Max Input Data
Rate Per Chan
(#Ch @ MSPS)
Max Signal
Mode
Name
DACs Interp. CMIX0 CMIX1
Out
Device
Config.
BW Per DAC
(2)
Factor Mode
Mode
Mode
Freq (MHz)(1)
(MHz)
(MSPS)
(MHz)
1X1
1
X1
–
–
Single Real
A
1000
500
1000
1 at 1000
500
(Bypass)
1X2
1
1
1
1
1
1
2
2
2
2
X2
X2
X4
X4
X4
X4
X1
X2
X2
X2
–
–
LP
HP
LP
HP
LP
HP
–
Single Real
Single Real
Single Real
Single Real
Single Real
Single Real
Dual Real
Dual Real
Dual Real
Complex
A
A
1000
1000
1000
1000
1000
1000
500
250
250
125
125
125
125
500
500
500
500
500
500
1 at 500
1 at 500
1 at 250
1 at 250
1 at 250
1 at 250
2 at 500
2 at 500
2 at 500
2 at 500
200
200
100
100
50
1X2 HP
1X4
LP
LP
HP
HP
–
A
250
1X4 LP/HP
1X4 HP/LP
1X4 HP/HP
2X1
A
250
A
250
A
250
50
A/B
A/B
A/B
A/B
1000
1000
1000
1000
250
200
200
200
2X2
–
LP
HP
1000
1000
1000
2X2 HP
2X2 CMIX
–
–
LP,
Fs/4
2X4
2
2
2
X4
X4
X4
LP
LP
LP
LP
Dual Real
Dual Real
Complex
A/B
A/B
A/B
1000
1000
1000
250
250
250
500
500
500
2 at 250
2 at 250
2 at 250
100
100
100
2X4 LP/HP
2X4 CMIX
HP
LP,
Fs/4
2X4 HP/LP
2X4 HP/HP
2
2
X4
X4
HP
HP
LP
Dual Real
Dual Real
A/B
A/B
1000
1000
250
250
500
500
2 at 250
2 at 250
50
50
HP
(1) Also the final DAC sample rate in MSPS.
(2) Assumes a 40% passband for FIR0 and/or FIR1 filters in all modes except 1X1 and 2X1 where simple Nyquist frequency is listed.
Slightly wider bandwidths may be achievable depending on filtering requirements. Refer to FIR Filters section for more detail on filter
characteristics. Also refer to Table 7 for IF placement and upconversion considerations.
Table 2. Register Map
(MSB)
Bit 7
(LSB)
Bit 0
Name
Address
Default
Bit 6
Bit 5
Bit 4
Bit 3
Bit 2
Bit 1
STATUS0
CONFIG1
CONFIG2
0x00
0x01
0x02
0x03
0x10
0xC0
PLL_lock
DLL_lock
Unused
Unused
FIR2x4x
device_ID(2:0)
SLFTST _ena
version(1:0)
FIFO_offset(2:0)
CMIX0_mode(1:0)
DAC_delay(1:0)
fir_ena
Twos_ comp
DAC_offset _ena
Unused
dual_DAC
Unused
CMIX1_mode(1:0)
SLFTST_err
_mask
Pattern_err
_mask
CONFIG3
STATUS4
CONFIG5
0x03
0x04
0x05
0x70
0x00
0x00
FIFO_err_ mask
FIFO_err
SwapAB_ out
Unused
B_equals _A
Unused
SW_sync
SW_sync _sel
Unused
SLFTST_err
rev_bus
Pattern_ err
Unused
FIFO_
sync_dis
PLL_
bypass
SIF4
clkdiv_ sync_dis
Sleep_B
Reserved
BiasLPF_A
DLL_ bypass
BiasLPF_B
Reserved
CONFIG6
CONFIG7
CONFIG8
CONFIG9
CONFIG10
CONFIG11
CONFIG12
CONFIG13
CONFIG14
CONFIG15
0x06
0x07
0x08
0x09
0x0A
0x0B
0x0C
0x0D
0x0E
0x0F
0x0C
0xFF
0x00
0x00
0x00
0x00
0x00
0x00
0x00
0x00
Hold_sync _dis
Unused
Sleep_A
PLL_ sleep
DLL_ sleep
DACA_gain(3:0)
DACB_gain(3:0)
DLL_ restart
Reserved
Reserved
PLL_m(4:0)
PLL_n(2:0)
DLL_delay(3:0)
VCO_div2
DLL_invclk
DLL_ifixed(2:0)
PLL_LPF _reset
PLL_gain(1:0)
Offset_sync
PLL_range(3:0)
Reserved(1:0)
OffsetA(12:8)
OffsetA(7:0)
SDO_func_sel(2:0)
OffsetB(12:8)
OffsetB(7:0)
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Register name: STATUS0 – Address: 0x00, Default = 0x03
Bit 7
Bit 6
Bit 5
Bit 4
Bit 3
Bit 2
0
Bit 1
1
Bit 0
1
PLL_lock
0
DLL_lock
0
Unused
0
device_ID(2:0)
0
version(1:0)
0
PLL_lock:
DLL_lock:
Asserted when the internal PLL is locked. (Read Only)
Asserted when the internal DLL is locked. Once the DLL is locked, this bit should remain a
‘1’ unless the DCLK input clock is removed or abruptly changes frequency causing the DLL
to fall out of lock. (Read Only)
device_ID(2:0): Returns ‘000’ for DAC5682Z Device_ID code. (ReadOnly)
version(1:0): A hardwired register that contains the register set version of the chip. (ReadOnly)
version (1:0)
Identification
‘01’
‘10’
‘11'
PG1.0 Initial Register Set
PG1.1 Register Set
Production Register Set
Register name: CONFIG1 – Address: 0x01, Default = 0x10
Bit 7
Bit 6
Bit 5
Bit 4
Bit 3
Bit 2
0
Bit 1
Bit 0
0
DAC_delay(1:0)
Unused
0
FIR_ena
1
SLFTST_ena
0
FIFO_offset(2:0)
0
0
0
DAC_delay(1:0): DAC data delay adjustment. (0–3 periods of the DAC clock) This can be used to adjust
system level output timing. The same delay is applied to both DACA and DACB data paths.
FIR_ena:
When set, the interpolation filters are enabled.
SLFTST_ena:
When set, a Digital Self Test (SLFTST) of the core logic is enabled. Refer to Digital Self
Test Mode section for details on SLFTST operation.
FIFO_offset(2:0): Programs the FIFO’s output pointer location, allowing the input pointer to be shifted –4 to
+3 positions upon SYNC. Default offset is 0 and is updated upon each sync event – unless
disabled via FIFO_sync_dis in CONFIG5 register.
FIFO_offset(2:0)
Offset
+3
011
010
001
000
111
110
101
100
+2
+1
0
–1
–2
–3
–4
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Register name: CONFIG2 – Address: 0x02, Default = 0xC0
Bit 7
Bit 6
Bit 5
Bit 4
Bit 3
Bit 2
0
Bit 1
0
Bit 0
Twos_comp
1
dual_DAC
1
FIR2x4x
0
Unused
0
CMIX1_mode(1:0)
CMIX0_mode(1:0)
0
0
Twos_comp:
dual_DAC:
When set (default) the input data format is expected to be 2’s complement, otherwise
offset binary format is expected.
Selects between dual DAC mode (default) and single DAC mode. This bit is also used
to select input interleaved data.
FIR2x4x:
When set, 4X interpolation of the input data is performed, otherwise 2X interpolation.
CMIX1_mode(1:0):
Determines the mode of FIR1 and final CMIX1 blocks. Settings apply to both A and B
channels. Refer to Table 9 for a detailed description of CMIX1 modes.
Mode
CMIX1_mode(1) CMIX1_mode(0)
Normal (Low
Pass)
0
0
High Pass
+FDAC /4
–FDAC/4
0
1
1
1
0
1
CMIX0_mode(1:0): Determines the mode of FIR0 and CMIX0 blocks. Since CMIX0 is located between FIR0
and FIR1, its output is half-rate. Refer to Table 8 for a detailed description of CMIX0
modes. The table below shows the effective Fs/4 or ±Fs/8 mixing with respect to the final
DAC sample rate. Settings apply to both A and B channels.
Mode
CMIX1_mode(1) CMIX1_mode(0)
Normal (Low
Pass)
0
0
High Pass
+FDAC /8
–FDAC/8
0
1
1
1
0
1
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Register name: CONFIG3 – Address: 0x03, Default = 0x70
Bit 7
Bit 6
Bit 5
Bit 4
Bit 3
Bit 2
Bit 1
Bit 0
DAC_offset
_ena
SLFTST_err
_mask
FIFO_err_
mask
Pattern_err_
mask
SwapAB_out
B_equals_A
SW_sync
SW_sync_sel
0
1
1
1
0
0
0
0
DAC_offset_ena:
When set, the values of OffsetA(12:0) and OffsetB(12:0) in CONFIG12 through
CONFIG15 registers are summed into the DAC-A and DAC-B data paths. This provides
a system-level offset adjustment capability that is independent of the input data.
SLFTST_err_mask: When set, masks out the SLFTST_err bit in STATUS4 register. Refer to Digital Self
Test Mode section for details on SLFTST operation.
FIFO_err_mask:
Pattern_err_mask:
SwapAB_out:
When set, masks out the FIFO_err bit in STATUS4 register.
When set, masks out the Pattern err bit in STATUS4 register.
When set, the A/B data paths are swapped prior to routing to the DAC-A and DAC-B
outputs.
B_equals_A:
When set, the data routed to DAC-A is also routed to DAC-B. This allows wire OR’ing of
the two DAC outputs together at the board level to create a 2X drive strength single
DAC output.
SW_sync:
This bit can be used as a substitute for the LVDS external SYNC input pins for both
synchronization and transmit enable control.
SW_sync_sel:
When set, the SW_sync bit is used as the only synchronization input and the LVDS
external SYNC input pins are ignored.
Register name: STATUS4 – Address: 0x04, Default = 0x00
Bit 7
Bit 6
Bit 5
Bit 4
Bit 3
Bit 2
Bit 1
Bit 0
Unused
0
SLFTST_err
0
FIFO_err
0
Pattern_err
0
Unused
0
Unused
0
Unused
0
Unused
0
SLFTST_err:
Asserted when the Digital Self Test (SLFTST) fails. To clear the error, write a ‘0’ to this
register bit. This bit is also output on the SDO pin when the Self Test is enabled via
SLFTST_ena control bit in CONFIG1. Refer to Digital Self Test Mode section for details on
SLFTST operation.
FIFO_err:
Asserted when the FIFO pointers over run each other causing a sample to be missed. To
clear the error, write a ‘0’ to this register bit.
Pattern_err:
A digital checkerboard pattern compare function is provided for board level confidence
testing and DLL limit checks. If the Pattern_err_mask bit via CONFIG3 is cleared, logic is
enabled to continuously monitor input FIFO data. Any received data pattern other than
0xAAAA or 0x5555 causes this bit to be set. To clear the error, flush out the previous
pattern error by inputting at least 8 samples of the 0xAAAA and/or 0x5555, then write a ‘0’
to this register bit.
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Register name: CONFIG5 – Address: 0x05, Default = 0x00
Bit 7
SIF4
Bit 6
Bit 5
Bit 4
Bit 3
Bit 2
Bit 1
Bit 0
rev_bus
clkdiv_sync
_dis
FIFO_sync _dis
Reserved
DLL_bypass
PLL_bypass
Reserved
0
0
0
0
0
0
0
0
SIF4:
When set, the serial interface is in 4 pin mode, otherwise it is in 3 pin mode. Refer to
SDO_func_sel(2:0) bits in CONFIG14 register for options available to output status
indicator data on the SDO pin.
rev_bus:
Reverses the LVDS input data bus so that the MSB to LSB order is swapped. This
function is provided to ease board level layout and avoid wire crossovers in case the
LVDS data source output bus is mirrored with respect to the DAC’s input data bus.
clkdiv_sync_dis:
FIFO_sync_dis:
Disables the clock divider sync when this bit is set.
Disables the FIFO offset sync when this bit is set. See FIFO_offset(2:0) bits in CONFIG1
register
Reserved (Bit 3):
DLL_bypass:
Set to 0 for proper operation.
When set, the DLL is bypassed and the LVDS data source is responsible for providing
correct setup and hold timing.
PLL_bypass:
When set, the PLL is bypassed.
Set to 0 for proper operation.
Reserved (Bit 0):
Register name: CONFIG6 – Address: 0x06, Default = 0x0C
Bit 7
Bit 6
Bit 5
Bit 4
Bit 3
Bit 2
Bit 1
Bit 0
Hold_sync _dis
0
Unused
0
Sleep_B
0
Sleep_A
0
BiasLPF_A
1
BiasLPF_B
1
PLL_sleep
0
DLL_sleep
0
Hold_sync_dis: When set, disables the sync to the FIFO output HOLD block. Typically this bit should be
cleared to ‘0’ for normal operation and also follow the same value as the FIFO_sync_dis
control bit in CONFIG5.
Sleep_B:
When set, DACB is put into sleep mode. DACB is not automatically set into sleep mode
when configured for single DAC mode via dual_DAC bit in CONFIG2. Set this Sleep_B bit
for the lowest power configuration in single DAC mode since output is on DACA only.
Sleep_A:
When set, DACA is put into sleep mode.
BiasLPF_A:
Enables a 95 kHz low pass filter corner on the DACA current source bias when set. If this bit
is cleared, a 472 Hz filter corner is used.
BiasLPF_B:
Enables a 95 kHz low pass filter corner on the DACB current source bias when set. If this bit
is cleared, a 472 Hz filter corner is used.
PLL_sleep:
DLL_sleep:
When set, the PLL is put into sleep mode.
When set, the DLL is put into sleep mode.
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Register name: CONFIG7 – Address: 0x07, Default = 0xFF
Bit 7
Bit 6
Bit 5
Bit 4
Bit 3
1
Bit 2
Bit 1
1
Bit 0
1
DACA_gain(3:0)
DACB_gain(3:0)
1
1
1
1
1
DACA_gain(3:0):
Scales DACA output current in 16 equal steps.
VEXTIO
x DACA_gain + 1
(
)
R
bias
DACB_gain(3:0):
Same as above except for DACB.
Register name: CONFIG8 – Address: 0x08, Default = 0x00
Bit 7
Bit 6
Bit 5
Bit 4
Bit 3
0
Bit 2
Bit 1
0
Bit 0
0
Reserved
0
DLL_restart
0
Reserved
0
0
0
Reserved (7:3): Set to ‘00000’ for proper operation.
DLL_restart:
This bit is used to restart the DLL. When this bit is set, the internal DLL loop filter is reset to
zero volts, and the DLL delay line is held at the center of its bias range. When cleared, the
DLL will acquire lock to the DCLK signal. A DLL restart is accomplished by setting this bit
with a serial interface write, and then clearing this bit with another serial interface write. Any
interruption in the DCLK signal or changes to the DLL programming in the CONFIG10
register must be followed by this DLL restart sequence. Also, when this bit is set, the
DLL_lock indicator in the STATUS0 register is cleared.
Reserved (1:0): Set to ‘00’ for proper operation
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Register name: CONFIG9 – Address: 0x09, Default = 0x00
Bit 7
Bit 6
Bit 5
Bit 4
Bit 3
0
Bit 2
0
Bit 1
Bit 0
0
PLL_m(4:0)
0
PLL_n(2:0)
0
0
0
0
PLL_m: M portion of the M/N divider of the PLL thermometer encoded:
PLL_m(4:0)
00000
M value
1
00001
2
4
00011
00111
8
01111
16
11111
32
All other values
Invalid
PLL_n: N portion of the M/N divider of the PLL thermometer encoded. If supplying a high rate CLKIN
frequency, the PLL_n value should be used to divide down the input CLKIN to maintain a maximum
PFD operating of 160 MHz.
PLL_n(2:0)
N value
000
001
1
2
011
4
8
111
All other values
Invalid
PLL Function:
é
ù
M
ê( )ú
f
=
x f
ref
vco
N
êë ( )úû
where ƒref is the frequency of the external DAC clock input on the CLKIN/CLKINC pins.
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Register name: CONFIG10 – Address: 0x0A, Default = 0x00
Bit 7
Bit 6
Bit 5
Bit 4
Bit 3
Bit 2
0
Bit 1
Bit 0
0
DLL_delay(3:0)
DLL_invclk
0
DLL_ifixed(2:0)
0
0
0
0
0
DLL_delay(3:0):
The DCLKP/N LVDS input data clock has a DLL to automatically skew the clock to LVDS
data timing relationship, providing proper setup and hold times. DLL_delay(3:0) is used to
manually adjust the DLL delay ± from the fixed delay set by DLL_ifixed(2:0). Adjustment
amounts are approximate.
DLL_delay(3:0)
1000
Delay Adjust (degrees)
50°
55°
1001
1010
60°
1011
65°
1100
70°
1101
75°
1110
80°
1111
85°
0000
90° (Default)
95°
0001
0010
100°
105°
110°
115°
120°
125°
0011
0100
0101
0110
0111
DLL_invclk:
When set, used to invert an internal DLL clock to force convergence to a different solution.
This can be used in the case where the DLL delay adjustment has exceeded the limits of
its range.
DLL_ifixed(2:0):
Adjusts the DLL delay line bias current. Refer to the Electrical Characteristics table. Used
in conjunction with the DLL_invclk bit to select appropriate delay range for a given DCLK
frequency:
'011' – maximum bias current and minimum delay range
'000' – mid scale bias current
'101' – minimum bias current and maximum delay range
'100' – do not use.
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Register name: CONFIG11 – Address: 0x0B, Default = 0x00
Bit 7
Bit 6
Bit 5
Bit 4
Bit 3
Bit 2
Bit 1
0
Bit 0
0
PLL_LPF_ reset
0
VCO_div2
0
PLL_gain(1:0)
PLL_range(3:0)
0
0
0
0
PLL_LPF_reset: When a logic high, the PLL loop filter (LPF) is pulled down to 0V. Toggle from ‘1’ to ‘0’ to
restart the PLL if an over-speed lock-up occurs. Over-speed can happen when the process
is fast, the supplies are higher than nominal, etc., resulting in the feedback dividers missing
a clock.
VCO_div2:
When set, the PLL CLOCK output is 1/2 the PLL VCO frequency. Used to run the VCO at
2X the needed clock frequency to reduce phase noise for lower input clock rates.
PLL_gain(1:0):
Used to adjust the PLL’s Voltage Controlled Oscillator (VCO) gain, KVCO. Refer to the
Electrical Characteristics table. By increasing the PLL_gain, the VCO can cover a broader
range of frequencies; however, the higher gain also increases the phase noise of the PLL.
In general, lower PLL_gain settings result in lower phase noise. The KVCO of the VCO can
also affect the PLL stability and is used to determine the loop filter components. See
section on determining the PLL filter components for more detail.
PLL_range(3:0): Programs the PLL VCO fixed bias current. Refer to the Electrical Characteristics table. This
setting, in conjunction with the PLL_gain(1:0), sets the achievable frequency range of the
PLL VCO:
'000' – minimum bias current and lowest VCO frequency range
'111' – maximum bias current and highest VCO frequency range
Register name: CONFIG12 – Address: 0x0C, Default = 0x00
Bit 7
Reserved(1:0)
Bit 6
Bit 5
Bit 4
Bit 3
Bit 2
Bit 1
0
Bit 0
0
Offset_sync
0
OffsetA(12:8)
0
0
0
0
0
Reserved(1:0):
Offset_sync:
Set to ‘00’ for proper operation.
On a change from ‘0’ to ‘1’ the values of the OffsetA(12:0) and OffsetB(12:0) control
registers are transferred to the registers used in the DAC-A and DAC-B offset calculations.
This double buffering allows complete control by the user as to when the change in the
offset value occurs. This bit does not auto-clear. Prior to updating new offset values, it is
recommended that the user clear this bit.
OffsetA(12:8):
Upper 5 bits of the offset adjustment value for the A data path. (SYNCED via Offset_sync)
Register name: CONFIG13 – Address: 0x0D, Default = 0x00
Bit 7
Bit 6
Bit 5
Bit 4
Bit 3
Bit 2
0
Bit 1
0
Bit 0
0
OffsetA(7:0)
0
0
0
0
0
OffsetA(7:0):
Lower 8 bits of the offset adjustment value for the A data path. (SYNCED via Offset_sync)
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Register name: CONFIG14 – Address: 0x0E, Default = 0x00
Bit 7
Bit 6
Bit 5
Bit 4
Bit 3
Bit 2
Bit 1
0
Bit 0
0
SDO_func_sel(2:0)
0
OffsetB(12:8)
0
0
0
0
0
SDO_func_sel(2:0): Selects the signal for output on the SDO pin. When using the 3 pin serial interface
mode, this allows the user to multiplex several status indicators onto the SDO pin. In 4
pin serial interface mode, programming this register to view one of the 5 available status
indicators will override normal SDO serial interface operation.
SDO_func_sel
(2:0)
Output to SDO
000, 110, 111
Normal SDO function
PLL_lock
001
010
011
100
101
DLL_lock
Pattern_err
FIFO_err
SLFTST_err
OffsetB(12:8):
Upper 5 bits of the offset adjustment value for the B data path. (SYNCED via Offset_sync)
Register name: CONFIG15 – Address: 0x0F, Default = 0x00
Bit 7
0
Bit 6
0
Bit 5
0
Bit 4
0
Bit 3
0
Bit 2
0
Bit 1
0
Bit 0
0
OffsetB(7:0)
OffsetB(7:0): Lower 8 bits of the offset adjustment value for the B data path. (SYNCED via Offset_sync)
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SERIAL INTERFACE
The serial port of the DAC5682Z is a flexible serial interface which communicates with industry standard
microprocessors and microcontrollers. The interface provides read/write access to all registers used to define the
operating modes of DAC5682Z. It is compatible with most synchronous transfer formats and can be configured
as a 3 or 4 pin interface by SIF4 in register CONFIG5. In both configurations, SCLK is the serial interface input
clock and SDENB is serial interface enable. For 3 pin configuration, SDIO is a bidirectional pin for both data in
and data out. For 4 pin configuration, SDIO is data in only and SDO is data out only.
Each read/write operation is framed by signal SDENB (Serial Data Enable Bar) asserted low for 2 to 5 bytes,
depending on the data length to be transferred (1–4 bytes). The first frame byte is the instruction cycle which
identifies the following data transfer cycle as read or write, how many bytes to transfer, and what address to
transfer the data. Table 3 indicates the function of each bit in the instruction cycle and is followed by a detailed
description of each bit. Frame bytes 2 to 5 comprise the data transfer cycle.
Table 3. Instruction Byte of the Serial Interface
MSB
LSB
Bit
7
6
5
4
3
2
1
0
Description
R/W
N1
N0
A4
A3
A2
A1
A0
R/W
[N1 : N0]
Identifies the following data transfer cycle as a read or write operation. A high indicates a read
operation from DAC5682Z and a low indicates a write operation to DAC5682Z.
Identifies the number of data bytes to be transferred per Table 5 below. Data is transferred MSB
first.
Table 4. Number of Transferred Bytes Within One
Communication Frame
N1
0
N0
0
Description
Transfer 1 Byte
Transfer 2 Bytes
Transfer 3 Bytes
Transfer 4 Bytes
0
1
1
0
1
1
[A4 : A0]
Identifies the address of the register to be accessed during the read or write operation. For
multi-byte transfers, this address is the starting address. Note that the address is written to the
DAC5682Z MSB first and counts down for each byte.
Figure 27 shows the serial interface timing diagram for a DAC5682Z write operation. SCLK is the serial interface
clock input to DAC5682Z. Serial data enable SDENB is an active low input to DAC5682Z. SDIO is serial data in.
Input data to DAC5682Z is clocked on the rising edges of SCLK.
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Instruction Cycle
Data Transfer Cycle (s)
SDENB
SCLK
SDIO
r/w N1 N0 A4 A3 A2 A1 A0 D7 D6 D5 D4 D3 D2 D1 D0
t
(SDENB)
t
S
SCLK
SDENB
SCLK
SDIO
t
SCLKL
t
(SDIO)
h
t
SCLKH
t
(SDIO)
S
Figure 27. Serial Interface Write Timing Diagram
Figure 28 shows the serial interface timing diagram for a DAC5682Z read operation. SCLK is the serial interface
clock input to DAC5682Z. Serial data enable SDENB is an active low input to DAC5682Z. SDIO is serial data in
during the instruction cycle. In 3 pin configuration, SDIO is data out from DAC5682Z during the data transfer
cycle(s), while SDO is in a high-impedance state. In 4 pin configuration, SDO is data out from DAC5682Z during
the data transfer cycle(s). At the end of the data transfer, SDO will output low on the final falling edge of SCLK
until the rising edge of SDENB when it will 3-state.
Instruction Cycle
Data Transfer Cycle(s)
SDENB
SCLK
SDIO
SDO
r/w N1 N0
-
A3 A2 A1 A0 D7 D6 D5 D4 D3 D2 D1 D0
D7 D6 D5 D4 D3 D2 D1 D0
0
0
3 pin Configuration Output
4 pin Configuration Output
SDENB
SCLK
SDIO
SDO
Data n
Data n-1
t
(Data)
d
Figure 28. Serial Interface Read Timing Diagram
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FIR FILTERS
Figure 29 shows the magnitude spectrum response for the identical 47-tap FIR0 and FIR1 filters. The transition
band is from 0.4 to 0.6 × FIN (the input data rate for the FIR filter) with <0.002 dB of pass-band ripple and
approximately 76dB of stop-band attenuation. Figure 30 shows the region from 0.35 to 0.45 × FIN – up to 0.44x
FIN there is less than 0.4 dB attenuation. The composite spectrum for x4 interpolation mode, the cascaded
response of FIR0 and FIR1, is shown in Figure 31. The filter taps for both FIR0 and FIR1 are listed in Table 5.
Figure 29. Magnitude Spectrum for FIR0 and FIR1
Figure 30. FIR0 and FIR1 Transition Band
Figure 31. Magnitude Composite Spectrum for 4x Interpolation Mode
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Table 5. FIR0 and FIR1 Digital Filter Taps
TAP#
1, 47
COEFF
–5
TAP#
2, 46
4, 44
6, 42
8, 40
10, 38
12, 36
14, 34
16, 32
18, 30
20, 28
22, 26
—
COEFF
0
0
3, 45
18
5, 43
–42
0
7, 41
85
0
9, 39
–158
272
0
11, 37
13, 35
15, 33
17, 31
19, 29
21, 27
23, 25
24
0
–444
704
0
0
–1106
1795
–3295
10368
16384
0
0
0
—
—
—
DUAL-CHANNEL REAL UPCONVERSION
The DAC5682Z can be used in a dual channel mode with real upconversion by mixing with a 1, –1, … sequence
in the signal chain to invert the spectrum. This mixing mode maintains isolation of the A and B channels. The two
points of mixing, CMIX0 and CMIX1, follow each FIR filter. The mixing modes for each CMIX block is control by
CMIX0_mode(1:0) and CMIX1(1:0) in register CONFIG2. The wide bandwidths of both FIR0 and FIR1 (40%
passband) provide options for setting four different frequency ranges, listed in Table 6. With the High Pass/Low
Pass (2X2 HP/LP mode) and Low Pass/High Pass (2X2 LP/HP mode) settings, the unconverted signal is
spectrally inverted.
Table 6. Dual-Channel Real Upconversion Options
FIR0,
CMIX0
MODE
FIR1,
CMIX1
MODE
INTERP.
FACTOR
INPUT
OUTPUT
SIGNAL
SPECTRUM
INVERTED?
MODE NAME
FREQUENCY(1)
FREQUENCY(1)
BANDWIDTH(1)
2X4
X4
X4
X4
X4
LP
HP
HP
LP
LP
LP
HP
HP
0.0 to 0.4 × fDATA
0.0 to 0.4 × fDATA
0.0 to 0.4 × fDATA
0.0 to 0.4 × fDATA
0.0 to 0.4 × fDATA
0.6 to 0.8 × fDATA
1.2 to 1.4 × fDATA
1.6 to 2.0 × fDATA
0.4 × fDATA
0.2 × fDATA
0.2 × fDATA
0.4 × fDATA
No
Yes
No
2X4 HP/LP
2X4 HP/HP
2X4 LP/HP
Yes
(1) fDATAis the input data rate of each channel after de-interleaving.
COARSE MIXERS: CMIX0 AND CMIX1
The DAC5682Z has two coarse mixer (CMIX) blocks: CMIX0 follows FIR0 and CMIX1 follows FIR1. (See
Figure 32) Each CMIX block provides mixing capability of fixed frequencies Fs/2 (real) or ±Fs/4 (complex) with
respect to the output frequency of the preceding FIR block. Since FIR0 and CMIX0 are only used in x4
interpolation modes, the output is half-rate relative to the DAC output frequency. Therefore, an ±Fs/4 mixing
sequence results in ±FDAC/8 frequency shift at the DAC output.
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Table 7. CMIX0 Mixer Sequences
CMIX0_mode(0)
Mode
CMIX0_mode(1)
MIXING SEQUENCE
Normal
(Low Pass, No Mixing)
0
0
1
1
0
1
0
1
FIR0A = { +A, +A , +A, +A }
FIR0B = { +B, +B , +B, +B }
High Pass
FIR0A = { +A, –A , +A, –A }
FIR0B = { +B, –B , +B, –B }
+FDAC /8 (+Fs/4)
–FDAC /8 (–Fs/4)
FIR0A = { +A, –B , –A, +B }
FIR0B = { +B, +A , –B, –A }
FIR0A = { +A, +B , –A, –B }
FIR0B = { +B, –A , –B, +A }
Table 8. CMIX1 Mixer Sequences
Mode
CMIX1_mode(1)
CMIX1_mode(0)
MIXING SEQUENCE
Normal
(Low Pass, No Mixing)
0
0
1
1
0
DACA = { +A, +A , +A, +A }
DACB = { +B, +B , +B, +B }
High Pass (Fs/2)
1
0
1
DACA = { +A, –A , +A, –A }
DACB = { +B, –B , +B, –B }
+FDAC /4
DACA = { +A, –B , –A, +B }
DACB = { +B, +A , –B, –A }
–FDAC /4
DACA = { +A, +B , –A, –B }
DACB = { +B, –A , –B, +A }
A Data In
x2
x2
A Data Out
FIR1
FIR0
B Data In
x2
x2
B Data Out
Block Diagram
(same for each)
0
1
A Mix In
A Mix Out
0
1
1
-1
1
B Mix Out
B Mix In
0
0
1
1
-1
CMIXx_mode(1:0)
Mix Sequencer
Figure 32. CMIX0 and CMIX1 Coarse Mixers Block Diagram
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CLOCK AND DATA MODES
There are two modes of operation to drive the internal clocks on the DAC5682Z. Timing diagrams for both
modes are shown in Figure 33. EXTERNAL CLOCK MODE accepts an external full-rate clock input on the
CLKIN/CLKINC pins to drive the DACs and final logic stages while distributing an internally divided down clock
for lower speed logic such as the interpolating FIRs. PLL CLOCK MODE uses an internal clock multiplying PLL
to derive the full-rate clock from an external lower rate reference frequency on the CLKIN/CLKINC pins. In both
modes, an LVDS half-rate data clock (DCLKP/DCLKN) is provided by the user and is typically generated by a
toggling data bit to maintain LVDS data to DCLK timing alignment. LVDS data relative to DCLK is input using
Double Data Rate (DDR) switching using both rising and falling edges as shown in the both figures below. The
CONFIG10 register contains user controlled settings for the DLL to adjust for the DCLK input frequency and
various tSKEW timing offsets between the LVDS data and DCLK. The CDCM7005 from Texas Instruments is
recommended for providing phase aligned clocks at different frequencies for device-to-device clock distribution
and multiple DAC synchronization.
CLKINC
PLL = 4X
CLKIN
Two Clock Mode Shown: PLL = 4X and EXTERNAL (PLL = OFF)
CLKIN
EXTERNAL
CLKINC
DACCLK
(Internal)
DCLKN
DCLKP
t
t
SKEW(B)
SKEW(A)
t
S
t
Valid Data (A)
Valid Data (B)
H
SYNCN
Transmit Enable / Synchronization Event
SYNCP
D[15:0]N
D[15:0]P
A
B
A
B
A
A
A
A
A
A
N+1
Single DAC Mode (1X1)
Dual DAC Mode (2X2)
1
3
0
2
N
A
B
N-2
0
1
0
1
N-2
Figure 33. Clock and Data Timing Diagram
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PLL CLOCK MODE
In PLL Clock Mode, the user provides an external reference clock to the CLKIN/C input pins. Refer to Figure 34.
An internal clock multiplying PLL uses the lower-rate reference clock to generate a high-rate clock for the DAC.
This function is very useful when a high-rate clock is not already available at the system level; however, the
internal VCO phase noise in PLL Clock Mode may degrade the quality of the DAC output signal when compared
to an external low jitter clock source.
The internal PLL has a type four phase-frequency detector (PFD) comparing the CLKIN/C reference clock with a
feedback clock to drive a charge pump controlling the VCO operating voltage and maintaining synchronization
between the two clocks. An external low-pass filter is required to control the loop response of the PLL. See the
Low-Pass Filter section for the filter setting calculations. This is the only mode where the LPF filter applies.
The input reference clock N-Divider is selected by CONFIG9 PLL_n(2:0) for values of ÷1, ÷2, ÷4 or ÷8. The VCO
feedback clock M-Divider is selected by CONFIG9 PLL_m(4:0) for values of ÷1, ÷2, ÷4, ÷8, ÷16 or ÷32. The
combination of M-Divider and N-Divider form the clock multiplying ratio of M/N. If the reference clock frequency is
greater than 160 MHz, use a N-Divider of ÷2, ÷4 or ÷8 to avoid exceeding the maximum PFD operating
frequency.
For DAC sample rates less than 500MHz, the phase noise of DAC clock signal can be improved by programming
the PLL for twice the desired DAC clock frequency, and setting the CONFIG11 VCO_div2 bit. If not using the
PLL, set CONFIG5 PLL_bypass and CONFIG6 PLL_sleep to reduce power consumption. In some cases, it
may be useful to reset the VCO control voltage by toggling CONFIG11 PLL_LPF_reset.
External
Loop
Filter
PLL Bypass
Clock Multiplying PLL
To internal
DAC clock
distribution
CLKIN
FREF/N
FVCO/M
N–Divider
(1, 2, 4, 8)
FVCO
VCO
FREF
Charge
Pump
PFD
CLKINC
FPLL
FVCO/2
M-Divider
( 1,2,4,8,16,32)
÷2
FVCO
PLL Sleep
PLL_sleep
(CONFIG6)
PLL_m(4:0)
(CONFIG9)
VCO_div2 PLL_bypass
(CONFIG11) via CONFIG5
PLL_gain(1:0),
PLL_range(3:0)
(CONFIG11)
PLL_n(2:0)
(CONFIG9)
PLL_LPF_reset
(CONFIG11)
Figure 34. Functional Block Diagram for PLL
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CLOCK INPUTS
Figure 35 shows an equivalent circuit for the LVDS data input clock (DCLKP/N).
27 kW
DVDD
DCLKP
Note: Input and output common mode
level self-biases to approximately DVDD/2,
or 0.9 V normal.
DVDD
GND
GND
DCLKN
27 kW
Figure 35. DCLKP/N Equivalent Input Circuit
Figure 36 shows an equivalent circuit for the DAC input clock (CLKIN/C).
6 kW
CLKVDD
CLKIN
Note: Input and output common mode
level self-biases to approximately CLKVDD/2,
or 0.9 V normal.
CLKVDD
GND
GND
CLKINC
6 kW
Figure 36. CLKIN/C Equivalent Input Circuit
Figure 37 shows the preferred configuration for driving the CLKIN/CLKINC input clock with a differential
ECL/PECL source.
0.01 mF
CLKIN
+
Differential
ECL
or
C
100 W
AC
(LV)PECL
Source
-
CLKINC
82.5 W
0.01 mF
R
R
130 W
T
T
130 W
82.5 W
V
TT
Figure 37. Preferred Clock Input configuration With a Differential ECL/PECL Clock Source
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LVDS DATA INTERFACING
Interfacing very high-speed LVDS data and clocks presents a big challenge to system designers as they have
unique constraints and are often implemented with specialized circuits to increase bandwidth. One such
specialized LVDS circuit used in many FPGAs and ASICs is a SERializer-DESerializer (SERDES) block. For
interfacing to the DAC5682Z, only the SERializer functionality of the SERDES block is required. SERDES drivers
accept lower rate parallel input data and output a serial stream using a shift register at a frequency multiple of
the data bit width. For example, a 4-bit SERDES block can accept parallel 4-bit input data at 250 MSPS and
output serial data 1000 MSPS.
External clock distribution for FPGA and ASIC SERDES drivers often have a chip-to-chip system constraint of a
limited input clock frequency compared to the desired LVDS data rate. In this case, an internal clock multiplying
PLL is often used in the FPGA or ASIC to drive the high-rate SERDES outputs. Due to this possible system
clocking constraint, the DAC5682Z accommodates a scheme where a toggling LVDS SERDES data bit can
provide a “data driven” half-rate clock (DCLK) from the data source. A DLL on-board the DAC is used to shift the
DCLK edges relative to LVDS data to maintain internal setup and hold timing.
To increase bandwidth of a single 16-bit input bus, the DAC5682Z assumes Double Data Rate (DDR) style
interfacing of data relative to the half-rate DCLK. Refer to Figure 38 and Figure 39 providing an example
implementation using FPGA-based LVDS data and clock interfaces to drive the DAC5682Z. In this example, an
assumed system constraint is that the FPGA can only receive a 250 MHz maximum input clock while the desired
DAC clock is 1000 MHz. A clock distribution chip such as the CDCM7005 is useful in this case to provide
frequency and phase locked clocks at 250 MHz and 1000 MHz.
FPGA / ASIC
DAC5682Z DAC
DAC
TRF3703 AQM
5V
Term
I-Signal
LPF
Antenna
100
SERDES
D15
1.0 GBPS
(DDR)
I
100
100
SERDES
SERDES
PA
5V
Term
D0
Q-Signal
LPF
Q
DAC
SYNC
To TX
Feedback
1.0 GHz
DLL
90
100
SERDES
Control
DCLK
To RX
Path
opt.
PLL
0
500 MHz
Toggling
Data Bit
4x Clock
Multiplier
Loop
Filter
~ 2.1 GHz
TRF3761-X PLL/VCO
DAC5682Z
Control
Div
1/2/4
Freq/Phase Locked
250 MHz
1.0 GHz
VCO
VCTRL_IN
N-
Divider
10 MHz
REF
OSC
100
DLL
Term
÷1
Loop
Filter
CDCM7005
PFD
Charge
Pump
REF_IN
R-
Div
÷4
CPOUT
PLL
Synth
Loop
Filter
Clock Divider /
Distribution
Status & Control
VCXO_STATUS
REF_STATUS
Status &
Control
VCXO
CDCM7005
Control
PLL_LOCK
1000 MHz
TRF3761-X
Control
Figure 38. Example Direct Conversion System Diagram
From the example provided by Figure 39, driving LVDS data into the DAC using SERDES blocks requires a
parallel load of 4 consecutive data samples to shift registers. Color is used in the figure to indicate how data and
clocks flow from the FPGA to the DAC5682Z. The figure also shows the use of the SYNCP/N input, which along
with DCLK, requires 18 individual SERDES data blocks to drive the DAC’s input data FIFO that provides an
elastic buffer to the DAC5682Z digital processing chain.
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Using common “data driven”
SERDES blocks, relative
delays from CLK, SYNC and
DATA are matched. (200pS)
FPGA
M
1000
250MHz
÷1
DCLK
Lock Loop
4
4
Ref CLK
Gen &
Sync
CLKA
(500MHz)
CLKB
(500MHz)
System
SYNC
C
250MHz
(bit 15)
4
4
D15N
16
To DAC
1000MSPS DDR
(2 bits/CLKIN cycle)
16
16
16
D0P
4b SERDES
(bit 0)
LVDS
D0N
250 MHz (FPGA)
DDR C
0
CLKA F
CLKB F
DLL Phase Offset control
determines CLKA/B skew.
SYNC input combines TXENABLE
function (normally “1”) and SYNChronizer
function (“0” to “1” transition)
SYNC
1
SYNCP/N
SERDES
S3[15:0]
S4[15:0]
Sam
Sam
Sample “S3”
Sample “S4”
15] S1[15]
Bit 0 Data Nibble
S4[0] S3[0] S2[0] S1[0]
D0P/N
SERDES
Figure 39. Example FPGA-Based LVDS Data Flow to DAC
LVDS Inputs (D[15:0]P/N and SYNCP/N)
To Adjacent
LVDS Input
D[15:0]P,
SYNCP
100 pF
Total
LVDS
Receiver
Ref Note (1)
D[15:0]N,
SYNCN
Note (1):
R
node common
CENTER
To Adjacent
LVDS Input
to all D[15:0]P/N and SYNCP/N
receiver inputs
Figure 40. D[15:0]P/N and SYNCP/N LVDS Input Configuration
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Example
V
1.40 V
1.00 V
A
DAC5682Z
D[15:0]P,
SYNCP
V
B
LVDS
V
V
A
400 mV
0 V
A,B
Receiver
V
A
V
=
COM1
(V +V )/2
D[15:0]N,
SYNCN
A
B
V
B
-400 mV
V
B
GND
1
0
Logical Bit
Equivalent
Figure 41. LVDS Data (DxP/N, SYNCP/N Pairs) Input Levels
Example LVDS Data Input Levels
APPLIED VOLTAGES
RESULTING
DEFERENTIAL
VOLTAGE
RESULTING
COMMON-MODE
VOLTAGE
LOGICAL BIT BINARY
EQUIVALENT
VA
VB
VA,B
VCOM1
1.4 V
1.0 V
1.2 V
0.8 V
1.0 V
1.4 V
0.8 V
1.2 V
400 mV
–400 mV
400 mV
–400 mV
1.2 V
1
0
1
0
1.0 V
Note: AC Coupled
DAC5682Z
)
0.01 mF
Self-bias (VBIAS
DCLKP
DCLKN
VA,B
DLL
Circuit
VA
0.01 mF
VB
GND
VCOM2 =~ DVDD/2
Figure 42. LVDS Clock (DCLKP/N) Input Levels
LVDS SYNCP/N Operation
The SYNCP/N LVDS input control functions as a combination of Transmit Enable (TXENABLE) and
Synchronization trigger. If SYNCP is low, the transmit chain is disabled so input data from the FIFO is ignored
while zeros are inserted into the data path. If SYNCP is raised from low to high, a synchronization event occurs
with behavior defined by individual control bits in registers CONFIG1, CONFIG5 and CONFIG6. The SYNCP/N
control is sampled and input into the FIFO along with the other LVDS data to maintain timing alignment with the
data bus. Refer to Figure 39.
The software_sync_sel and software_sync controls in CONFIG3 provide a substitute for external SYNCP/N
control; however, since the serial interface is used no timing control is provided with respect to the DAC clock.
DLL OPERATION
The DAC5682Z provides a digital Delay Lock Loop (DLL) to skew the LVDS data clock (DCLK) relative to the
data bits, D[15:0] and SYNC, in order to maintain proper setup and hold timing. Since the DLL operates
closed-loop, it requires a stable DCLK to maintain delay lock. Refer to the description of DLL_ifixed(2:0) and
DLL_delay(3:0) control bits in the CONFIG10 register. Prior to initializing the DLL, the DLL_ifixed value should
be programmed to match the expected DCLK frequency range. To initialize the DLL, refer to the DLL_Restart
programming bit in the CONFIG8 register. After initialization, the status of the DLL can be verified by reading the
DLL_Lock bit from STATUS0. See Startup Sequence below.
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RECOMMENDED STARTUP SEQUENCE
The following startup sequence is recommend to initialization the DAC5682Z:
1. Supply all 1.8V (CLKVDD, DVDD, VFUSE) and 3.3V (AVDD and IOVDD) voltages.
2. Provide stable CLKIN/C clock.
3. Toggle RESETB pin for a minimum 25 nSec active low pulse width.
4. Program all desired SIF registers. Set DLL_Restart bit during this write cycle. Ensure that DLL_ifixed and
DLL_phase bits to correspond with the DCLKP/N frequency and delay.
5. Provide stable DCLKP/N clock. (This can also be provided earlier in the sequence)
6. Clear the DLL_Restart bit when the DCLKP/N clock is expected to be stable.
7. Verify the status of DLL_Lock and repeat until set to ‘1’. DLL_Lock can be monitored by reading the
STATUS0 register or by monitoring the SDO pin in 3-wire SIF mode. (See description for CONFIG14
SDO_func_sel.)
8. Enable transmit of data by asserting the LVDS SYNCP/N input or setting CONFIG3 SW_sync bit. (See
description for CONFIG3 SW_sync and SW_sync_sel) The SYNC source must be held at a logic ‘1’ to
enable data flow through the DAC.
9. Provide data flow to LVDS D[15:0]P/N pins. If using the LVDS SYNCP/N input, data can be input
simultaneous with the logic ‘1’ transition of SYNCP/N.
CMOS DIGITAL INPUTS
Figure 43 shows a schematic of the equivalent CMOS digital inputs of the DAC5682Z. SDIO and SCLK have
pull-down resistors while RESETB and SDENB have pull-up resistors internal the DAC5682Z. See the
specification table for logic thresholds. The pull-up and pull-down circuitry is approximately equivalent to 100kΩ.
IOVDD
IOVDD
internal
digital in
RESETB
SDENB
internal
digital in
SDIO
SCLK
IOGND
IOGND
Figure 43. CMOS/TTL Digital Equivalent Input
DIGITAL SELF TEST MODE
The DAC5682Z has a Digital Self Test (SLFTST) mode to designed to enable board level testing without
requiring specific input data test patterns. The SLFTST mode is enabled via the CONFIG1 SLFTST_ena bit and
results are only valid when CONFIG3 SLFTST_err_mask bit is cleared. An internal Linear Feedback Shift
Register (LFSR) is used to generate the input test patterns for the full test cycle while a checksum result is
computed on the digital signal chain outputs. The LVDS input data bus is ignored in SLFTST mode. After the test
cycle completes, if the checksum result does not match a hardwired comparison value, the STATUS4
SLFTST_err bit is set and will remain set until cleared by writing a ‘0’ to the SLFTST_err bit. A full self test cycle
requires no more than 400,000 CLKIN/C clock cycles to complete and will automatically repeat until the
SLFTEST_ena bit is cleared.
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To initiate a the Digital Self Test:
1. Provide a normal CLKIN/C input clock. (The PLL is not used in SLFTST mode)
2. Provide a RESETB pulse to perform a hardware reset on device.
3. Program the registers with the values shown in Table 9. These register values contain the settings to
properly configure the SLFTST including SLFTST_ena and SLFTST_err_mask bits
4. Provide a ‘1’ on the SYNCP/N input to initiate TXENABLE.
5. Wait at a minimum of 400,000 CLKIN/C cycles for the SLFTST to complete. Example: If CLKIN = 1GHz, then
the wait period is 400,000 × 1 / 1GHz = 400 µSec.
6. Read STATUS4 SLFTST_err bit. If set, a self test error has occurred. The SLFTST_err status may
optionally be programmed to output on the SDO pin if using the 3-bit SIF interface. See Table 9 Note (1).
7. (Optional) The SLFTST function automatically repeats until SLFTST_ena bit is cleared. To the loop the test,
write a ‘0’ to STATUS4 SLFTST_err to clear previous errors and continue at step 5 above.
8. To continue normal operating mode, provide another RESETB pulse and reprogram registers to the desired
normal settings.
Table 9. Digital Self Test (SLFTST) Register Values
REGISTER
CONFIG1
CONFIG2
CONFIG3
STATUS4
CONFIG5
CONFIG6
CONFIG12
CONFIG13
CONFIG14(1)
CONFIG15
All others
ADDRESS (hex)
VALUE (Binary)
00011000
11101010
10110000
00000000
00000110
00001111
00001010
01010101
00001010
10101010
Default
VALUE (Hex)
01
02
03
04
05
06
0C
0D
0E
0F
–
18
EA
B0
00
06
0F
0A
55
0A
AA
Default
(1) If using a 3-bit SIF interface, the SDO pin can be programmed to report SLFTST_err status via the SDO_fun_sel(2:0) bits. In this case,
set CONFIG14 = ‘10101010’ or AA hex.
REFERENCE OPERATION
The DAC5682Z uses a bandgap reference and control amplifier for biasing the full-scale output current. The
full-scale output current is set by applying an external resistor RBIAS to pin BIASJ. The bias current IBIAS through
resistor RBIAS is defined by the on-chip bandgap reference voltage and control amplifier. The default full-scale
output current equals 16 times this bias current and can thus be expressed as:
IOUTFS = 16 × IBIAS = 16 × VEXTIO / RBIAS
Each DAC has a 4-bit independent coarse gain control via DACA_gain(3:0) and DACB_gain(3:0) in the
CONFIG7 register. Using gain control, the IOUTFS can be expressed as:
IOUTAFS = (DACA_gain + 1) × IBIAS = (DACA_gain + 1) × VEXTIO / RBIAS
IOUTBFS = (DACB_gain + 1) × IBIAS = (DACB_gain + 1) × VEXTIO / RBIAS
where VEXTIO is the voltage at terminal EXTIO. The bandgap reference voltage delivers an accurate voltage of
1.2 V. This reference is active when terminal EXTLO is connected to AGND. An external decoupling capacitor
CEXT of 0.1 µF should be connected externally to terminal EXTIO for compensation. The bandgap reference can
additionally be used for external reference operation. In that case, an external buffer with high impedance input
should be applied in order to limit the bandgap load current to a maximum of 100 nA. The internal reference can
be disabled and overridden by an external reference by connecting EXTLO to AVDD. Capacitor CEXT may
hence be omitted. Terminal EXTIO thus serves as either input or output node.
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The full-scale output current can be adjusted from 20 mA down to 2 mA by varying resistor RBIAS or changing the
externally applied reference voltage. The internal control amplifier has a wide input range, supporting the
full-scale output current range of 20 dB.
DAC TRANSFER FUNCTION
The CMOS DAC’s consist of a segmented array of NMOS current sinks, capable of sinking a full-scale output
current up to 20 mA. Differential current switches direct the current to either one of the complementary output
nodes IOUT1 or IOUT2. (DACA = IOUTA1 or IOUTA2 and DACB = IOUTB1 or IOUTB2.) Complementary output
currents enable differential operation, thus canceling out common mode noise sources (digital feed-through,
on-chip and PCB noise), dc offsets, even order distortion components, and increasing signal output power by a
factor of two.
The full-scale output current is set using external resistor RBIAS in combination with an on-chip bandgap voltage
reference source (+1.2 V) and control amplifier. Current IBIAS through resistor RBIAS is mirrored internally to
provide a maximum full-scale output current equal to 16 times IBIAS
.
The relation between IOUT1 and IOUT2 can be expressed as:
IOUT1 = – IOUTFS – IOUT2
We will denote current flowing into a node as – current and current flowing out of a node as + current. Since the
output stage is a current sink the current can only flow from AVDD into the IOUT1 and IOUT2 pins. The output
current flow in each pin driving a resistive load can be expressed as:
IOUT1 = IOUTFS × (65536 – CODE) / 65536
IOUT2 = IOUTFS × CODE / 65536
where CODE is the decimal representation of the DAC data input word.
For the case where IOUT1 and IOUT2 drive resistor loads RL directly, this translates into single ended voltages
at IOUT1 and IOUT2:
VOUT1 = AVDD – | IOUT1 | × RL
VOUT2 = AVDD – | IOUT2 | × RL
Assuming that the data is full scale (65536 in offset binary notation) and the RL is 25 Ω, the differential voltage
between pins IOUT1 and IOUT2 can be expressed as:
VOUT1 = AVDD – | –0 mA | × 25 Ω = 3.3 V
VOUT2 = AVDD – | –20 mA | × 25 Ω = 2.8 V
VDIFF = VOUT1 – VOUT2 = 0.5 V
Note that care should be taken not to exceed the compliance voltages at node IOUT1 and IOUT2, which would
lead to increased signal distortion.
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DAC OUTPUT SINC RESPONSE
Due to sampled nature of a high-speed DAC’s, the well known sin(x)/x (or SINC) response can significantly
attenuate higher frequency output signals. See the Figure 44 which shows the unitized SINC attenuation roll-off
with respect to the final DAC sample rate in 4 Nyquist zones. For example, if the final DAC sample rate FS = 1.0
GSPS, then a tone at 440MHz is attenuated by 3.0dB. Although the SINC response can create challenges in
frequency planning, one side benefit is the natural attenuation of Nyquist images. The increased over-sampling
ratio of the input data provided by the DAC5682Z’s 2x and 4x digital interpolation modes improve the SINC
roll-off (droop) within the original signal’s band of interest
Figure 44. Unitized DAC sin(x)/x (SINC) Response
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ANALOG CURRENT OUTPUTS
Figure 45 shows a simplified schematic of the current source array output with corresponding switches.
Differential switches direct the current of each individual NMOS current source to either the positive output node
IOUT1 or its complementary negative output node IOUT2. The output impedance is determined by the stack of
the current sources and differential switches, and is typically >300 kΩ in parallel with an output capacitance of 5
pF.
The external output resistors are referred to an external ground. The minimum output compliance at nodes
IOUT1 and IOUT2 is limited to AVDD – 0.5 V, determined by the CMOS process. Beyond this value, transistor
breakdown may occur resulting in reduced reliability of the DAC5682Z device. The maximum output compliance
voltage at nodes IOUT1 and IOUT2 equals AVDD + 0.5 V. Exceeding the minimum output compliance voltage
adversely affects distortion performance and integral non-linearity. The optimum distortion performance for a
single-ended or differential output is achieved when the maximum full-scale signal at IOUT1 and IOUT2 does not
exceed 0.5 V.
AVDD
R
R
LOAD
LOAD
IOUT1
IOUT2
S(1)
S(N)
S(2)
S(2)C
S(N)C
S(1)C
...
Figure 45. Equivalent Analog Current Output
The DAC5682Z can be easily configured to drive a doubly terminated 50Ω cable using a properly selected RF
transformer. Figure 46 and Figure 47 show the 50Ω doubly terminated transformer configuration with 1:1 and 4:1
impedance ratio, respectively. Note that the center tap of the primary input of the transformer has to be
connected to AVDD to enable a dc current flow. Applying a 20 mA full-scale output current would lead to a 0.5
VPP for a 1:1 transformer and a 1 VPP output for a 4:1 transformer. The low dc-impedance between IOUT1 or
IOUT2 and the transformer center tap sets the center of the ac-signal at AVDD, so the 1 VPP output for the 4:1
transformer results in an output between AVDD + 0.5 V and AVDD – 0.5 V.
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AVDD
(3.3 V)
50 W
1 : 1
IOUT1
IOUT2
R
LOAD
50 W
100 W
50 W
AVDD (3.3 V)
Figure 46. Driving a Doubly Terminated 50 Ω Cable Using a 1:1 Impedance Ratio Transformer
AVDD (3.3 V)
100 W
4 : 1
IOUT1
R
LOAD
50 W
IOUT2
100 W
AVDD (3.3 V)
Figure 47. Driving a Doubly Terminated 50 Ω Cable Using a 4:1 Impedance Ratio Transformer
DESIGNING THE PLL LOOP FILTER
To minimize phase noise given for a given fDAC and M/N, the values of PLL_gain and PLL_range are selected
so that GVCO is minimized and within the MIN and MAX frequency for a given setting.
The external loop filter components C1, C2, and R1 are set by the GVCO, M/N, the loop phase margin φd and the
loop bandwidth ωd. Except for applications where abrupt clock frequency changes require a fast PLL lock time, it
is suggested that φd be set to at least 80 degrees for stable locking and suppression of the phase noise side
lobes. Phase margins of 60 degrees or less can be sensitive to board layout and decoupling details.
See Figure 48, the recommend external loop filter topology. C1, C2, and R1 are calculated by the following
equations:
t2
t3
t1 - t2
t3
t32
æ
ö
C1 = t1 1 -
C2 =
R1 =
ç
÷
t1 t3 - t2
(
)
è
ø
(1)
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where,
t1 =
K K
d
tan Φ + sec Φ
d
1
VCO
d
tan Φ + sec Φ
(
d
t2 =
t3 =
)
d
2
w
tan Φ + sec Φ
d
w
d
(
)
d
d
w
d
(2)
charge pump current: iqp = 1 mA
vco gain: KVCO = 2π × GVCO rad/V
PFD Frequency: ωd ≤160 MHz
phase detector gain: Kd = iqp ÷ (2 × π × M) A/rad
An Excel spreadsheet is available from Texas Instruments for automatically calculating the values for C1, R1 and
C2.
DAC5682Z PLL
LPF
R1
(Pin 64)
PLL
External
Loop
Filter
C2
C1
Figure 48. Recommended External Loop Filter Topology
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APPLICATIONS EXAMPLES
DIGITAL INTERFACE AND CLOCKING CONSIDERATIONS FOR APPLICATION EXAMPLES
The DAC5682Z’s LVDS digital input bus can be driven by an FPGA or digital ASIC. This input signal can be
generated directly by the FPGA, or fed by a Texas Instruments Digital Up Converter (DUC) such as the GC5016
or GC5316. Optionally, a GC1115 Crest Factor Reduction (CFR) or Digital Pre-Distortion (DPD) processor may
be inserted in the digital signal chain for improving the efficiency of high-power RF amplifiers. For the details on
the DAC’s high-rate digital interface, refer to the LVDS Data Interfacing section.
A low phase noise clock for the DAC at the final sample rate can be generated by a VCXO and a Clock
Synchronizer/PLL such as the Texas Instruments CDCM7005, which can also provide other system clocks at the
VCXO frequency divided by /1, /2, /3, /4, /6, /8 or/16. An optional system clocking solution can use the DAC in
clock multiplying PLL mode in order to avoid distributing a high-frequency clock at the DAC sample rate;
however, the internal VCO phase noise of the DAC in PLL mode may degrade the quality of the DAC output
signal.
SINGLE COMPLEX INPUT, REAL IF OUTPUT RADIO
Refer to Figure 49 for an example Single Complex Input, Real IF Output Radio. The DAC5682Z receives an
interleaved complex I/Q baseband input data stream and increases the sample rate through interpolation by a
factor of 2 or 4. By performing digital interpolation on the input data, undesired images of the original signal can
be push out of the band of interest and more easily suppressed with analog filters. Complex mixing is available at
each stage of interpolation using the CMIX0 and CMIX1 blocks to up-convert the signal to a frequency placement
at a multiples ±Fdac/8 or ±Fdac/4. Only the real portion of the digital signal is converted by DAC-A while DAC-B
can be programmed to sleep mode for reduced power consumption. The DAC output signal would typically be
terminated with a transformer (see the Analog Current Output section). An IF filter, either LC or SAW, is used to
suppress the DAC Nyquist zone images and other spurious signals before being mixed to RF with a mixer. The
TRF3671 Frequency Synthesizer, with integrated VCO, may be used to drive the LO input of the mixer for
frequencies between 375 and 2380 MHz.
Interleaved
I/Q Data
3.3V
FPGA
DAC5682Z DAC
3.3V
100
D15P/N
D0P/N
RF
Processing
DAC-A
I
100
100
3.3V
DAC-B
Sleep
Q
SYNCP/N
DCLKP/N
100
DLL
opt.
PLL
PLL/
DLL
375 MHz Min to 2380 MHz Max
(Depends on divider and
“dash #” of TRF3761)
1000 MHz
Loop
Filter
100
250 MHz
Div
1/2/4
VCXO
÷4
÷1
VCO
VCTRL_IN
N-
Divider
Loop
Filter
Loop
Filter
Clock Divider /
Distribution
PLL
PFD
R-
Div
CDCM7005
CPOUT
10 MHz
OSC
TRF3761-X PLL/VCO
Note: For clarity, only signal paths are shown.
Figure 49. System Diagram of a Complex Input, Real IF Output Radio
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APPLICATIONS EXAMPLES (continued)
DUAL CHANNEL REAL IF OUTPUT RADIO
Refer to Figure 50 for an example Dual Channel Real IF Output Radio. The DAC5682Z receives an interleaved
A/B input data stream and increases the sample rate through interpolation by a factor of 2 or 4. By performing
digital interpolation on the input data, undesired images of the original signal can be push out of the band of
interest and more easily suppressed with analog filters. Real mixing is available at each stage of interpolation
using the CMIX0 and CMIX1 blocks to up-convert the signal. (See Dual-Channel Real Upconversion section)
Both DAC output signals would typically be terminated with a transformer (see the Analog Current Output
section). An IF filter, either LC or SAW, is used to suppress the DAC Nyquist zone images and other spurious
signals before being mixed to RF with a mixer. The TRF3671 Frequency Synthesizer, with integrated VCO, may
be used to drive a common LO input of the mixers for frequencies between 375 and 2380 MHz. Alternatively, two
separate TRF3761 synthesizers could be used for independent final RF frequency placement.
Interleaved
3.3V
FPGA
A/B Data
DAC5682Z DAC
3.3V
100
D15P/N
D0P/N
RF
Processing
DAC-A
DAC-B
I
3.3V
100
100
3.3V
3.3V
RF
Processing
Q
SYNCP/N
DCLKP/N
3.3V
100
DLL
opt.
PLL
PLL/
DLL
375 MHz Min to 2380 MHz Max
(Depends on divider and
“dash #” of TRF3761)
1000 MHz
Loop
Filter
100
250 MHz
Div
1/2/4
VCXO
÷4
÷1
VCO
VCTRL_IN
N-
Divider
Loop
Filter
Loop
Filter
Clock Divider /
Distribution
PLL
PFD
R-
Div
CDCM7005
CPOUT
10 MHz
OSC
TRF3761-X PLL/VCO
Note: For clarity,only signal paths are shown.
Figure 50. System Diagram of a Dual Channel Real IF Output Radio
DIRECT CONVERSION RADIO
Refer to Figure 51 for an example Direct Conversion Radio. The DAC5682Z receives an interleaved complex I/Q
baseband input data stream and increases the sample rate through interpolation by a factor of 2 or 4. By
performing digital interpolation on the input data, undesired images of the original signal can be push out of the
band of interest and more easily suppressed with analog filters.
For a Zero IF (ZIF) frequency plan, complex mixing of the baseband signal is not required. Alternatively, for a
Complex IF frequency plan the input data can be placed at an pre-placed at an IF within the bandwidth
limitations of the interpolation filters. In addition, complex mixing is available at each stage of interpolation using
the CMIX0 and CMIX1 blocks to up-convert the signal to a frequency placement at a multiples ±Fdac/8 or
±Fdac/4. The output of both DAC channels is used to produce a Hilbert transform pair and can be expressed as:
A(t) = I(t)cos(ωct) – Q(t)sin(ωct) m(t)
A(t) = I(t)cos(ωct) – Q(t)sin(ωct) mh(t)
(3)
(4)
where m(t) and mh(t) connote a Hilbert transform pair and ωc is the sum of the CMIX0 and CMIX1 frequencies.
The complex output is input to an analog quadrature modulator (AQM) such as the Texas Instruments
TRF3703-33 for a single side-band (SSB) up conversion to RF. A passive (resistor only) interface to the AQM is
recommended, with an optional LC filter network. The TRF3671 Frequency Synthesizer with integrated VCO may
be used to drive the LO input of the TRF3703-33 for frequencies between 375 and 2380 MHz. Upper
single-sideband upconversion is achieved at the output of the analog quadrature modulator, whose output is
expressed as:
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APPLICATIONS EXAMPLES (continued)
RF(t) = I(t)cos(ωc + ωLO)t – Q(t)sin(ωc + ωLO)t
(5)
Flexibility is provided to the user by allowing for the selection of negative CMIX mixing sequences to produce a
lower-sideband upconversion. Note that the process of complex mixing translates the signal frequency from 0 Hz
means that the analog quadrature modulator IQ imbalance produces a sideband that falls outside the signal of
interest. DC offset error in DAC and AQM signal path may produce LO feed-through at the RF output which may
fall in the band of interest. To suppress the LO feed-through, the DAC5682Z provides a digital offset correction
capability for both DAC-A and DAC-B paths. (See DAC_offset_ena bit in CONFIG3.)
The complex IF architecture has several advantages over the real IF architecture:
•
•
•
•
Uncalibrated side-band suppression ~ 35 dBc compared to 0 dBc for real IF architecture.
Direct DAC to AQM interface – no amplifiers required
Nonharmonic clock-related spurious signals fall out-of-band
DAC 2nd Nyquist zone image is offset fDAC compared with fDAC– 2 × IF for a real IF architecture, reducing the
need for filtering at the DAC output.
•
Uncalibrated LO feed through for AQM is ~ 35 dBc and calibration can reduce or completely remove the LO
feed through.
5V
Interleaved
FPGA
I/Q Data
DAC5682Z DAC
100
D15P/N
DAC-A
I
D0P/N
100
RF OUT
DAC-B
100
Q
SYNCP/N
DCLKP/N
90
100
DLL
opt.
PLL
0
TRF3703-33 AQM
PLL/
DLL
375 MHz Min to 2380 MHz Max
(Depends on divider and
“dash #” of TRF3761)
1000 MHz
Loop
Filter
100
250 MHz
Div
1/2/4
VCXO
÷4
÷1
VCO
VCTRL_IN
N-
Divider
Loop
Filter
Loop
Filter
Clock Divider /
Distribution
PLL
PFD
R-
Div
CDCM7005
CPOUT
10 MHz
OSC
TRF3761-X PLL/VCO
Note: For clarity,only signal paths are shown .
Figure 51. System Diagram of Direct Conversion Radio
CMTS/VOD TRANSMITTER
The exceptional SNR of the DAC5682Z enables a dual-cable modem termination system (CMTS) or video on
demand (VOD) QAM transmitter in excess of the stringent DOCSIS specification, with >74 dBc and 75 dBc in the
adjacent and alternate channels.
Refer to Figure 50 for an example Dual Channel Real IF Output Radio – this signal chain is nearly identical to a
typical system using the DAC5682Z for a cost optimized dual channel two QAM transmitter. A GC5016 would
take four separate symbol rate inputs and provide pulse shaping and interpolation to ~ 128 MSPS. The four QAM
carriers would be combined into two groups of two QAM carriers with intermediate frequencies of approximately
30 MHz to 40 MHz. The GC5016 would output two real data streams to one DAC5682Z through an FPGA for
CMOS to LVDS translation. The DAC5682Z would function as a dual-channel device and provide 2x or 4x
interpolation to increase the frequency of the 2nd Nyquist zone image. The two signals are then output through
the two DAC outputs, through a transformer and to an RF upconverter.
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APPLICATIONS EXAMPLES (continued)
HIGH-SPEED ARBITRARY WAVEFORM GENERATOR
The 1GSPS bandwidth input data bus combined with the 16-bit DAC resolution of the DAC5682Z allows
wideband signal generation for test and measurement applications. In this case, interpolation is not desired by
the FPGA-based waveform generator as it can make use of the full Nyquist bandwidth of up to 500MHz.
FPGA
DAC5682Z DAC
100
100
D15P/N
D0P/N
DAC-A
DAC-B
Sleep
100
SYNCP/N
DCLKP/N
100
DLL
Figure 52. System Diagram of Arbitrary Waveform Generator
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PACKAGE OPTION ADDENDUM
www.ti.com
8-Nov-2007
PACKAGING INFORMATION
Orderable Device
Status (1)
Package Package
Pins Package Eco Plan (2) Lead/Ball Finish MSL Peak Temp (3)
Qty
Type
QFN
QFN
Drawing
DAC5682ZIRGCR
DAC5682ZIRGCT
ACTIVE
ACTIVE
RGC
64
64
2000
250
TBD
TBD
Call TI
Call TI
Call TI
Call TI
RGC
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check
http://www.ti.com/productcontent for the latest availability information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements
for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered
at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and
package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS
compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame
retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder
temperature.
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