INA326EA/2K5 [TI]
Precision, Rail-to-Rail I/O INSTRUMENTATION AMPLIFIER; 精密，轨到轨输入/输出仪表放大器
| 型号： | INA326EA/2K5 |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | Precision, Rail-to-Rail I/O INSTRUMENTATION AMPLIFIER |
| 文件： | 总21页 (文件大小：458K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
INA326
INA327
I
N
A
I
N
A
3
2
6
3
2
7
SBOS222D – NOVEMBER 2001 – REVISED NOVEMBER 2004
Precision, Rail-to-Rail I/O
INSTRUMENTATION AMPLIFIER
FEATURES
DESCRIPTION
The INA326 and INA327 (with shutdown) are high-perfor-
● PRECISION
mance, low-cost, precision instrumentation amplifiers with
rail-to-rail input and output. They are true single-supply
instrumentation amplifiers with very low DC errors and input
common-mode ranges that extends beyond the positive and
negative rails. These features make them suitable for appli-
cations ranging from general-purpose to high-accuracy.
LOW OFFSET: 100µV (max)
LOW OFFSET DRIFT: 0.4µV/°C (max)
EXCELLENT LONG-TERM STABILITY
VERY-LOW 1/f NOISE
● TRUE RAIL-TO-RAIL I/O
Excellent long-term stability and very low 1/f noise assure
low offset voltage and drift throughout the life of the product.
INPUT COMMON-MODE RANGE:
20mV Below Negative Rail to 100mV Above
Positive Rail
The INA326 (without shutdown) comes in the MSOP-8 pack-
age. The INA327 (with shutdown) is offered in an MSOP-10.
Both are specified over the industrial temperature range,
–40°C to +85°C, with operation from –40°C to +125°C.
WIDE OUTPUT SWING: Within 10mV of Rails
SUPPLY RANGE: Single +2.7V to +5.5V
● SMALL SIZE
microPACKAGE: MSOP-8, MSOP-10
INA326 AND INA327 RELATED PRODUCTS
● LOW COST
PRODUCT FEATURES
INA337
INA114
INA118
INA122
INA128
INA321
Precision, 0.4µV/°C Drift, Specified –40°C to +125°C
50µV VOS, 0.5nA IB, 115dB CMR, 3mA IQ, 0.25µV/°C Drift
50µV VOS, 1nA IB, 120dB CMR, 385µA IQ, 0.5µV/°C Drift
250µV VOS, –10nA IB, 85µA IQ, Rail-to-Rail Output, 3µV/°C Drift
50µV VOS, 2nA IB, 125dB CMR, 750µA IQ, 0.5µV/°C Drift
500µV VOS, 0.5pA IB, 94dB CMRR, 60µA IQ, Rail-to-Rail Output
APPLICATIONS
● LOW-LEVEL TRANSDUCER AMPLIFIER FOR
BRIDGES, LOAD CELLS, THERMOCOUPLES
● WIDE DYNAMIC RANGE SENSOR
MEASUREMENTS
V+
V−
● HIGH-RESOLUTION TEST SYSTEMS
● WEIGH SCALES
7
2
1
VIN−
4
● MULTI-CHANNEL DATA ACQUISITION
6
VO
SYSTEMS
R1
INA326
8
3
● MEDICAL INSTRUMENTATION
● GENERAL-PURPOSE
5
G = 2(R2/R1)
VIN+
R2
C2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Copyright © 2001-2004, Texas Instruments Incorporated
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
www.ti.com
PACKAGE/ORDERING INFORMATION(1)
SPECIFIED
TEMPERATURE
RANGE
PACKAGE
DESIGNATOR
PACKAGE
MARKING
ORDERING
NUMBER
TRANSPORT
MEDIA, QUANTITY
PRODUCT
PACKAGE-LEAD
INA326
MSOP-8
DGK
–40°C to +85°C
B26
INA326EA/250
INA326EA/2K5
Tape and Reel, 250
Tape and Reel, 2500
"
"
"
"
"
INA327
MSOP-10
DGS
–40°C to +85°C
B27
INA327EA/250
INA327EA/2K5
Tape and Reel, 250
Tape and Reel, 2500
"
"
"
"
"
NOTE: (1) For the most current package and ordering information, download the latest version of this data sheet and see the Package Option Addendum located
at the end of the data sheet.
ABSOLUTE MAXIMUM RATINGS(1)
ELECTROSTATIC
DISCHARGE SENSITIVITY
Supply Voltage .................................................................................. +5.5V
Signal Input Terminals: Voltage(2) ..............................–0.5V to (V+) + 0.5V
Current(2) ................................................... ±10mA
This integrated circuit can be damaged by ESD. Texas
Instruments recommends that all integrated circuits be handled
with appropriate precautions. Failure to observe proper han-
dling and installation procedures can cause damage.
Output Short-Circuit ................................................................. Continuous
Operating Temperature Range ....................................... –40°C to +125°C
Storage Temperature Range .......................................... –65°C to +150°C
Junction Temperature .................................................................... +150°C
Lead Temperature (soldering, 10s) ............................................... +300°C
ESD damage can range from subtle performance degrada-
tion to complete device failure. Precision integrated circuits
may be more susceptible to damage because very small
parametric changes could cause the device not to meet its
published specifications.
NOTES: (1) Stresses above these ratings may cause permanent damage.
Exposure to absolute maximum conditions for extended periods may degrade
device reliability. These are stress ratings only, and functional operation of the
device at these or any other conditions beyond those specified is not implied.
(2)Inputterminalsarediodeclampedtothepower-supplyrails. Inputsignalsthat
can swing more than 0.5V beyond the supply rails should be current limited to
10mA or less.
PIN CONFIGURATION
Top View
INA326
INA327
R1
V+
VO
R2
R1
VIN−
VIN+
V−
1
2
3
4
8
7
6
5
R1
R1
1
2
3
4
5
10
9
V+
VIN−
VIN+
VO
8
R2
V−
7
Enable
(Connect to V+)
6
MSOP- 8
MSOP- 10
INA326, INA327
2
SBOS222D
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ELECTRICAL CHARACTERISTICS: VS = +2.7V to +5.5V
BOLDFACE limits apply over the specified temperature range, TA = –40°C to +85°C
At TA = +25°C, RL = 10kΩ, G = 100 (R1 = 2kΩ, R2 = 100kΩ), external gain set resistors, and IACOMMON = VS /2, with external equivalent filter corner of 1kHz, unless
otherwise noted.
INA326EA, INA327EA
PARAMETER
CONDITION
MIN
TYP
MAX
UNITS
INPUT
Offset Voltage, RTI
Over Temperature
vs Temperature
vs Power Supply
Long-Term Stability
Input Impedance, Differential
Common-Mode
Input Voltage Range
Safe Input Voltage
VOS
VS = +5V, VCM = VS /2
±20
±100
±124
±0.4
µV
µV
µV/°C
µV/V
dVOS/dT
±0.1
±3
See Note (1)
1010 || 2
1010 || 14
PSR
VS = +2.7V to +5.5V, VCM = VS /2
±20
Ω || pF
Ω || pF
V
V
dB
(V–) – 0.02
(V–) – 0.5
100
(V+) + 0.1
(V+) + 0.5
Common-Mode Rejection
Over Temperature
CMR VS = +5V, VCM = (V–) – 0.02V to (V+) + 0.1V
114
94
dB
INPUT BIAS CURRENT
Bias Current
vs Temperature
Offset Current
VCM = VS /2
VS = +5V
IB
±0.2
±2
nA
nA
See Typical Characteristics
IOS
VS = +5V
±0.2
±2
NOISE
Voltage Noise, RTI
f = 10Hz
RS = 0Ω, G = 100, R1 = 2kΩ, R2 = 100kΩ
RS = 0Ω, G = 10, R1 = 20kΩ, R2 = 100kΩ
√Hz
√Hz
√Hz
33
33
33
0.8
nV/
nV/
nV/
f = 100Hz
f = 1kHz
f = 0.01Hz to 10Hz
Voltage Noise, RTI
f = 10Hz
µVp-p
√Hz
√Hz
√Hz
120
97
97
4
nV/
nV/
nV/
f = 100Hz
f = 1kHz
f = 0.01Hz to 10Hz
Current Noise, RTI
f = 1kHz
f = 0.01Hz to 10Hz
Output Ripple, VO Filtered(2)
µVp-p
√Hz
0.15
4.2
pA/
pAp-p
See Applications Information
GAIN
Gain Equation
Range of Gain
Gain Error(3)
vs Temperature
Nonlinearity
G = 2(R2/R1)
< 0.1
> 10000
±0.2
±25
V/V
%
ppm/°C
% of FS
G = 10, 100, VS = +5V, VO = 0.075V to 4.925V
G = 10, 100, VS = +5V, VO = 0.075V to 4.925V
G = 10, 100, VS = +5V, VO = 0.075V to 4.925V
±0.08
±6
±0.004
±0.01
OUTPUT
Voltage Output Swing from Rail
RL = 100kΩ
RL = 10kΩ, VS = +5V
5
10
mV
mV
mV
pF
75
75
Over Temperature
Capacitive Load Drive
Short-Circuit Current
500
±25
ISC
mA
INTERNAL OSCILLATOR
Frequency of Auto-Correction
Accuracy
90
±20
kHz
%
FREQUENCY RESPONSE
Bandwidth(4), –3dB
Slew Rate(4)
BW
SR
G = 1 to 1k
VS = +5V, All Gains, CL = 100pF
1
kHz
Filter Limited
Settling Time(4), 0.1%
0.01%
0.1%
0.01%
Overload Recovery(4)
tS 1kHz Filter, G = 1 to 1k, VO = 2V step, CL = 100pF
0.95
1.3
130
160
30
ms
ms
µs
µs
µs
µs
10kHz Filter, G = 1 to 1k, VO = 2V step, CL = 100pF
1kHz Filter, 50% Output Overload, G = 1 to 1k
10kHz Filter, 50% Output Overload, G = 1 to 1k
5
INA326, INA327
3
SBOS222D
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ELECTRICAL CHARACTERISTICS: VS = +2.7V to +5.5V (Cont.)
BOLDFACE limits apply over the specified temperature range, TA = –40°C to +85°C
At TA = +25°C, RL = 10kΩ, G = 100 (R1 = 2kΩ, R2 = 100kΩ), external gain set resistors, and IACOMMON = VS /2, with external equivalent filter corner of 1kHz, unless
otherwise noted.
INA326EA, INA327EA
PARAMETER
CONDITION
MIN
TYP
MAX
UNITS
POWER SUPPLY
Specified Voltage Range
Quiescent Current
Over Temperature
+2.7
+5.5
3.4
3.7
V
mA
mA
IQ
IO = 0, Diff VIN = 0V, VS = +5V
2.4
SHUTDOWN
Disable (Logic Low Threshold)
Enable (Logic High Threshold)
Enable Time(5)
Disable Time
Shutdown Current and Enable Pin Current
0.25
V
V
µs
µs
µA
1.6
75
100
2
VS = +5V, Disabled
5
TEMPERATURE RANGE
Specified Range
Operating Range
Storage Range
–40
–40
–65
+85
+125
+150
°C
°C
°C
Thermal Resistance
θJA
MSOP-8, MSOP-10 Surface-Mount
150
°C/W
NOTES: (1) 1000-hour life test at 150°C demonstrated randomly distributed variation in the range of measurement limits—approximately 10µV. (2) See Applications
Information section, and Figures 1 and 3. (3) Does not include error and TCR of external gain-setting resistors. (4) Dynamic response is limited by filtering. Higher
bandwidths can be achieved by adjusting the filter. (5) See Typical Characteristics, “Input Offset Voltage vs Warm-Up Time”.
INA326, INA327
4
SBOS222D
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TYPICAL CHARACTERISTICS
At TA = 25°C, VS = +5V, Gain = 100, and RL = 10kΩ with external equivalent filter corner of 1kHz, unless otherwise noted.
GAIN vs FREQUENCY
1kHz FILTER
GAIN vs FREQUENCY
10kHz FILTER
80
60
80
60
G = 1k
G = 1k
40
40
G = 100
G = 10
G = 1
G = 100
G = 10
G = 1
20
20
0
0
−20
−40
−20
−40
10
100
1k
10k
100k
1M
10
10
1
100
1k
10k
100k
1M
1M
10k
Frequency (Hz)
Frequency (Hz)
COMMON- MODE REJECTION vs FREQUENCY
1kHz FILTER
COMMON- MODE REJECTION vs FREQUENCY
10kHz FILTER
160
140
120
100
80
160
140
120
100
80
G = 1k
G = 100
G = 1k
G = 10
G = 1
G = 100
60
60
G = 1
G = 10
40
40
20
20
10
100
1k
10k
100k
1M
100
1k
10k
100k
Frequency (Hz)
Frequency (Hz)
INPUT- REFERRED VOLTAGE NOISE AND
INPUT BIAS CURRENT NOISE vs FREQUENCY
10kHz FILTER
POWER- SUPPLY REJECTION vs FREQUENCY
G = 100, 1k
120
100
80
60
40
20
0
10k
1k
1
Current Noise
(all gains)
G = 10
G = 1
0.1
G = 1
G = 10
100
10
0.01
0.001
Filter Frequency
10kHz
G = 100
G = 1000
1kHz
10
100
1k
10k
100k
10
100
1k
Frequency (Hz)
Frequency (Hz)
INA326, INA327
5
SBOS222D
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TYPICAL CHARACTERISTICS (Cont.)
At TA = 25°C, VS = +5V, Gain = 100, and RL = 10kΩ with external equivalent filter corner of 1kHz, unless otherwise noted.
INPUT OFFSET VOLTAGE vs TURN- ON TIME
1kHz FILTER, G = 100
INPUT OFFSET VOLTAGE vs WARM- UP TIME
10kHz FILTER, G = 100
Filter
Settling
Time
Device
Turn- On
Time
Filter
Settling
Time
Device
Turn- On
Time
(75µs)
0
1
2
0
0.1
0.2
0.3
0.4
Turn- On Time (ms)
Warm- Up Time (ms)
SMALL- SIGNAL RESPONSE
G = 1, 10, AND 100
SMALL- SIGNAL STEP RESPONSE
G = 1000
1kHz Filter
10kHz Filter
500µs/div
500µs/div
LARGE- SIGNAL RESPONSE
G = 1 TO 1000
0.01Hz TO 10Hz VOLTAGE NOISE
1kHz Filter
10kHz Filter
10s/div
500µs/div
INA326, INA327
6
SBOS222D
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TYPICAL CHARACTERISTICS (Cont.)
At TA = 25°C, VS = +5V, Gain = 100, and RL = 10kΩ with external equivalent filter corner of 1kHz, unless otherwise noted.
OFFSET VOLTAGE DRIFT PRODUCTION DISTRIBUTION
G = 1
OFFSET VOLTAGE PRODUCTION DISTRIBUTION
G = 1
Offset Voltage Drift (µV/°C)
Offset Voltage (µV)
OFFSET VOLTAGE DRIFT PRODUCTION DISTRIBUTION
G = 10
OFFSET VOLTAGE PRODUCTION DISTRIBUTION
G = 10
Offset Voltage Drift (µV/°C)
Offset Voltage (µV)
OFFSET VOLTAGE PRODUCTION DISTRIBUTION
G = 100, 1000
OFFSET VOLTAGE DRIFT PRODUCTION DISTRIBUTION
G = 100, 1000
Offset Voltage (µV)
Offset Voltage Drift (µV/°C)
INA326, INA327
7
SBOS222D
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TYPICAL CHARACTERISTICS (Cont.)
At TA = 25°C, VS = +5V, Gain = 100, and RL = 10kΩ with external equivalent filter corner of 1kHz, unless otherwise noted.
GAIN ERROR PRODUCTION DISTRIBUTION
G = 100
INPUT- REFERRED RIPPLE SPECTRUM
G = 100
−100
−110
−120
−130
−140
−150
−160
−170
−180
100.000
31.600
1.000
0.316
0.100
0.030
0.010
0.003
0.001
0
200k
400k
600k
800k
1M
Frequency (Hz)
Gain Error (m%)
QUIESCENT CURRENT vs TEMPERATURE
VS = +5V
INPUT BIAS CURRENT vs TEMPERATURE
3.0
2.5
2.0
1.5
1.0
0.5
0
2.0
1.5
1.0
IB−
0.5
VS = +2.7V
0
−0.5
−1.0
−1.5
−2.0
IB+
25
Temperature (°C)
25
Temperature (°C)
−50
−25
0
50
75
100
125
−50
−25
0
50
75
100
125
INA326, INA327
8
SBOS222D
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SETTING THE GAIN
APPLICATIONS INFORMATION
The INA326 is a 2-stage amplifier with each stage gain set
by R1 and R2, respectively (see Figure 5, “Inside the INA326”,
for details). Overall gain is described by the equation:
Figure 1 shows the basic connections required for operation of
the INA326. A 0.1µF capacitor, placed close to and across the
power-supply pins is strongly recommended for highest accu-
racy. RoCo is an output filter that minimizes auto-correction
circuitry noise. This output filter may also serve as an anti-
aliasing filter ahead of an Analog-to-Digital (A/D) converter. It
is also optional based on desired precision.
R2
G = 2
(1)
R1
The stability and temperature drift of the external gain-setting
resistors will affect gain by an amount that can be directly
inferred from the gain equation (1).
The output reference terminal is taken at the low side of R2
(IACOMMON).
Resistor values for commonly used gains are shown in
Figure 1. Gain-set resistor values for best performance are
different for +5V single-supply and for ±2.5V dual-supply
operation. Optimum value for R1 can be calculated by:
The INA326 uses a unique internal topology to achieve excel-
lent Common-Mode Rejection (CMR). Unlike conventional
instrumentation amplifiers, CMR is not affected by resistance
in the reference connections or sockets. See “Inside the
INA326” for further detail. To achieve best high-frequency
CMR, minimize capacitance on pins 1 and 8.
R1 = VIN, MAX/12.5µA
(2)
where R1 must be no less than 2kΩ.
+2.5V
7
−2.5V
DESIRED
GAIN
R1
(Ω)
R2 || C2
(Ω || nF)
0.1µF
0.1
0.2
0.5
1
400k
400k
400k
200k
100k
40k
20k
10k
4k
20k || 5
40k || 2.5
100k || 1
100k || 1
100k || 1
100k || 1
100k || 1
100k || 1
100k || 1
100k || 1
200k || 0.5
500k || 0.2
1M || 0.1
2M || 0.05
5M || 0.02
10M || 0.01
2
1
VIN−
RO
100Ω
4
5
VO
6
VO Filtered
R1
INA326
2
(1)
8
3
CO
1µF
5
G = 2(R2/R1)
10
VIN+
f
O = 1kHz
20
50
(1)
R2
C2
100
200
500
1000
2000
5000
10000
2k
2k
(2)
2k
IACOMMON
2k
2k
(1) C2 and CO combine to form a 2-pole response that is −3dB at 1kHz.
Each individual pole is at 1.5kHz.
(2) Output voltage is referenced to IACOMMON (see text).
2k
2k
Single-supply operation may require
R2 > 100kΩ for full output swing.
This may produce higher input referred
V+
DESIRED
GAIN
R1
(Ω)
R2 || C2
(Ω || nF)
0.1µF
offset voltage. See Offset Voltage,
Drift, and Circuit Values for detail.
0.1
0.2
0.5
1
400k
400k
400k
400k
200k
80k
40k
20k
8k
20k || 5
7
2
1
VIN−
40k || 2.5
100k || 1
RO
4
VO
100Ω
200k || 0.5
200k || 0.5
200k || 0.5
200k || 0.5
200k || 0.5
200k || 0.5
200k || 0.5
200k || 0.5
500k || 0.2
1M || 0.1
6
VO Filtered
R1
INA326
2
(1)
8
3
CO
1µF
5
5
G = 2(R2/R1)
O = 1kHz
10
VIN+
f
(3)
20
50
(1)
R2
C2
100
200
500
1000
2000
5000
10000
4k
2k
(2)
IACOMMON
2k
2k
2k
2M || 0.05
5M || 0.02
10M || 0.01
(1) C2 and CO combine to form a 2-pole response that is −3dB at 1kHz.
Each individual pole is at 1.5kHz.
(2) Output voltage is referenced to IACOMMON (see text).
(3) Output offset voltage required for measurement near zero (see Figure 28).
2k
2k
NOTES: (1) C2 and CO combine to form a 2-pole response that is –3dB at 1kHz. Each individual pole is at 1.5kHz. (2) Output voltage is referenced to
IACOMMON (see text). (3) Output offset voltage required for measurement near zero (see Figure 6).
FIGURE 1. Basic Connections. NOTE: Connections for INA327 differ—see Pin Configuration for detail.
INA326, INA327
9
SBOS222D
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Following this design procedure for R1 produces the maximum
possible input stage gain for best accuracy and lowest noise.
The enable time following shutdown is 75µs plus the settling
time due to filters (see Typical Characteristics, “Input Offset
Voltage vs Warm-up Time”). Disable time is 100µs. This
allows the INA327 to be operated as a “gated” amplifier, or
to have its output multiplexed onto a common output bus.
When disabled, the output assumes a high-impedance state.
Circuit layout and supply bypassing can affect performance.
Minimize the stray capacitance on pins 1 and 8. Use recom-
mended supply bypassing, including a capacitor directly from
pin 7 to pin 4 (V+ to V–), even with dual (split) power supplies
(see Figure 1).
INA327 PIN 5
OFFSET VOLTAGE, DRIFT, AND CIRCUIT VALUES
Pin 5 of the INA327 should be connected to V+ to ensure
proper operation.
As with other multi-stage instrumentation amplifiers, input-
referred offset voltage depends on gain and circuit values. The
specified offset and drift performance is rated at R1 = 2kΩ,
R2 = 100kΩ, and VS = ±2.5V. Offset voltage and drift for other
circuit values can be estimated from the following equations:
DYNAMIC PERFORMANCE
The typical characteristic “Gain vs Frequency” shows that the
INA326 has nearly constant bandwidth regardless of gain.
This results from the bandwidth limiting from the recom-
mended filters.
VOS = 10µV + (50nA)(R2)/G
(3)
(4)
dVOS/dT = 0.12µV/°C + (0.16nA/°C)(R2)/G
These equations might imply that offset and drift can be
minimized by making the value of R2 much lower than the
values indicated in Figure 1. These values, however, have
been chosen to assure that the output current into R2 is kept
less than or equal to ±25µA, while maintaining R1’s value
greater than or equal to 2kΩ. Some applications with limited
output voltage swing or low power-supply voltage may allow
lower values for R2, thus providing lower input-referred offset
voltage and offset voltage drift.
NOISE PERFORMANCE
Internal auto-correction circuitry eliminates virtually all 1/f
noise (noise that increases at low frequency) in gains of 100
or greater. Noise performance is affected by gain-setting
resistor values. Follow recommendations in the “Setting
Gain” section for best performance.
Total noise is a combination of input stage noise and output
stage noise. When referred to the input, the total mid-band
noise is:
Conversely, single-supply operation with R2 grounded re-
quires that R2 values be made larger to assure that current
remains under 25µA. This will increase the input-referred
offset voltage and offset voltage drift.
800nV / Hz
VN = 33nV / Hz +
(5)
G
The output noise has some 1/f components that affect
performance in gains less than 10. See typical characteristic
“Input-Referred Voltage Noise vs Frequency.”
Circuit conditions that cause more than 25µA to flow in R2 will
not cause damage, but may produce more nonlinearity.
High-frequency noise is created by internal auto-correction
circuitry and is highly dependent on the filter characteristics
chosen. This may be the dominant source of noise visible
when viewing the output on an oscilloscope. Low cutoff
frequency filters will provide lowest noise. Figure 3 shows the
typical noise performance as a function of cutoff frequency.
INA327 ENABLE FUNCTION
The INA327 adds an enable/shutdown function to the INA326.
Its pinout differs from the INA326—see the Pin Configuration
for detail.
The INA327 can be enabled by applying a logic HIGH
voltage level to the Enable pin. Conversely, a logic LOW
voltage level will disable the amplifier, reducing its supply
current from 2.4mA to typically 2µA. For battery-operated
applications, this feature may be used to greatly reduce the
average current and extend battery life. This pin should be
connected to a valid high or low voltage or driven, not left
open circuit. The Enable pin can be modeled as a CMOS
input gate as in Figure 2.
1k
G = 1000
100
G = 100
10
V+
G = 10
G = 1
10
Required Filter Cutoff Frequency (Hz)
1
2µA
1
100
1k
10k
Enable
6
FIGURE 3. Total Output Noise vs Required Filter Cutoff
Frequency.
FIGURE 2. Enable Pin Model.
10
INA326, INA327
SBOS222D
www.ti.com
Applications sensitive to the spectral characteristics of high-
frequency noise may require consideration of the spurious
frequencies generated by internal clocking circuitry. “Spurs”
occur at approximately 90kHz and its harmonics (see typical
characteristic “Input-Referred Ripple Spectrum”) which may
be reduced by additional filtering below 1kHz.
Thermocouple
INA326
5
Insufficient filtering at pin 5 can cause nonlinearity with large
output voltage swings (very near the supply rails). Noise
must be sufficiently filtered at pin 5 so that noise peaks do not
“hit the rail” and change the average value of the signal.
Figure 3 shows guidelines for filter cutoff frequency.
FIGURE 4. Providing Input Bias Current Return Path.
HIGH-FREQUENCY NOISE
INPUT PROTECTION
C2 and CO form filters to reduce internally generated auto-
correction circuitry noise. Filter frequencies can be chosen to
optimize the trade-off between noise and frequency re-
sponse of the application, as shown in Figure 3. The cutoff
frequencies of the filters are generally set to the same
frequency. Figure 3 shows the typical output noise for four
gains as a function of the –3dB cutoff frequency of each filter
response. Small signals may exhibit the addition of internally
generated auto-correction circuitry noise at the output. This
noise, combined with broadband noise, becomes most evi-
dent in higher gains with filters of wider bandwidth.
The inputs of the INA326 are protected with internal diodes
connected to the power-supply rails. These diodes will clamp
the applied signal to prevent it from damaging the input
circuitry. If the input signal voltage can exceed the power
supplies by more than 0.5V, the input signal current should
be limited to less than 10mA to protect the internal clamp
diodes. This can generally be done with a series input
resistor. Some signal sources are inherently current-limited
and do not require limiting resistors.
FILTERING
INPUT BIAS CURRENT RETURN PATH
Filtering can be adjusted through selection of R2C2 and
ROCO for the desired trade-off of noise and bandwidth.
Adjustment of these components will result in more or less
ripple due to auto-correction circuitry noise and will also
affect broadband noise. Filtering limits slew rate, settling
time, and output overload recovery time.
The input impedance of the INA326 is extremely high—
approximately 1010Ω. However, a path must be provided for
the input bias current of both inputs. This input bias current is
approximately ±0.2nA. High input impedance means that this
input bias current changes very little with varying input voltage.
It is generally desirable to keep the resistance of RO relatively
low to avoid DC gain error created by the subsequent stage
loading. This may result in relatively high values for CO to
produce the desired filter response. The impedance of ROCO
can be scaled higher to produce smaller capacitor values if
the load impedance is very high.
Input circuitry must provide a path for this input bias current
for proper operation. Figure 4 shows provision for an input
bias current path in a thermocouple application. Without a
bias current path, the inputs will float to an undefined poten-
tial and the output voltage may not be valid.
Certain capacitor types greater than 0.1µF may have dielec-
tric absorption effects that can significantly increase settling
time in high-accuracy applications (settling to 0.01%). Polypro-
pylene, polystyrene, and polycarbonate types are generally
good. Certain “high-K” ceramic types may produce slow
settling “tails.” Settling time to 0.1% is not generally affected
by high-K ceramic capacitors. Electrolytic types are not
recommended for C2 and CO.
INPUT COMMON-MODE RANGE
Common instrumentation amplifiers do not respond linearly with
common-mode signals near the power-supply rails, even if “rail-
to-rail” op amps are used. The INA326 uses a unique topology
to achieve true rail-to-rail input behavior (see Figure 5, “Inside
the INA326”). The linear input voltage range of each input
terminal extends to 20mV below the negative rail, and 100mV
above the positive rail.
INA326, INA327
11
SBOS222D
www.ti.com
INSIDE THE INA326
The INA326 uses a new, unique internal circuit topology
A1 and A2’s output stages. A2 combines the current in R1
with a mirrored replica of the current from A1. The result-
ing current in A2’s output and associated current mirror is
two times the current in R1. This current flows in (or out)
of pin 5 into R2. The resulting gain equation is:
that provides true rail-to-rail input. Unlike other instrumen-
tation amplifiers, it can linearly process inputs up to 20mV
below the negative power-supply rail, and 100mV above
the positive power-supply rail. Conventional instrumenta-
tion amplifier circuits cannot deliver such performance,
even if rail-to-rail op amps are used.
R2
G = 2
R1
The ability to reject common-mode signals is derived in
most instrumentation amplifiers through a combination of
amplifier CMR and accurately matched resistor ratios.
The INA326 converts the input voltage to a current.
Current-mode signal processing provides rejection of com-
mon-mode input voltage and power-supply variation with-
out accurately matched resistors.
Amplifiers A1, A2, and their associated mirrors are pow-
ered from internal charge-pumps that provide voltage
supplies that are beyond the positive and negative supply
rails. As a result, the voltage developed on R2 can actually
swing 20mV below the negative power-supply rail, and
100mV above the positive supply rail. A3 provides a
buffered output of the voltage on R2. A3’s input stage is
also operated from the charge-pumped power supplies for
true rail-to-rail operation.
A simplified diagram shows the basic circuit function. The
differential input voltage, (VIN+) – (VIN–) is applied across
R1. The signal-generated current through R1 comes from
V+
V−
0.1µF
7
4
Current Mirror
INA326
IR1
IR1
2
1
VIN−
A1
Current Mirror
IR1
R1
Current Mirror
IR1
8
3
2IR1
2IR1
A2
VIN+
VO
6
A3
2IR1
2IR1
2IR1
Current Mirror
5
R2
C2
IACOMMON
FIGURE 5. Simplified Circuit Diagram.
INA326, INA327
12
SBOS222D
www.ti.com
APPLICATION CIRCUITS
2
R0
1
6
VO
R1
INA326
5
8
VREF
C0
3
R′2
R2 and R′2 are chosen to
R2
C2
create a small output offset
voltage (e.g., 100mV).
Gain is determined by
the parallel combination
of R2 and R′2.
G = 2 (R2 || R′2)/R1
FIGURE 6. Generating Output Offset Voltage.
VREF
2
1
RO
100Ω
A/D
Converter
6
INA326
2kΩ
8
3
5
CO
1µF
200kΩ
200kΩ
C2
G = 2(200kΩ || 200kΩ)/2kΩ = 100
FIGURE 7. Output Referenced to VREF/2.
+5V
RS must be chosen
so that the input voltage
does not exceed 100mV
beyond the rail.
RS
IL
2
1
7
RO
RL
100Ω
2kΩ
6
R1
VO
INA326
8
3
5
CO
1µF
NOTE: Connection point
of V+ will include ( ) or
exclude ( ) quiescent
R2
C2
current in the measurement
as desired. Output offset
required for measurements
near zero (see Figure 6).
R2
VO = 2(IL × RS)
R1
FIGURE 8. High-Side Current Shunt Measurement.
INA326, INA327
13
SBOS222D
www.ti.com
+5V
2
1
RO
100Ω
R2
R1
7
VO = 2(IL × RS)
VO
6
R1
INA326
5
8
3
CO
1µF
RL
R2
C2
2kΩ
IL
RS must be chosen so that
the input voltage does not
exceed 20mV beyond the rail.
RS
NOTE: Connection point of V− will include (
) or
exclude ( ) quiescent current in the measurement
as desired. Output offset required for measurements
near zero (see Figure 6).
FIGURE 9. Low-Side Current Shunt Measurement.
1nF
R
F = 100kΩ
NOTE: 0.2% accuracy. Current shunt
+5V
monitor circuit can be designed for −250V supply
with appropriate selection of high- voltage FET.
2
3
7
RF
RI
6
V
O = 2(IL × RS)
OPA336PA
4
RSTART
100kΩ
RPULL- DOWN
200kΩ
ZVN4525G
8.45kΩ
(zetex)
RL
(High- Voltage
n- Channel
2
1
IL
7
FET)
VCC
+
ZMM5231BDICT
5.1V
6
RI = 2kΩ
0.1µF
RS
VS
=
0mV
INA326
GND
4
5
8
3
to 50mV max
−
−48V
FIGURE 10. Low-Side –48V Current Shunt Monitor.
+48V
+
7
3
VCC
VSHUNT = 0mV
to 50mV
8
ZMM5231BDICT
5.1V
RI
2kΩ
6
RSHUNT
0.1µF
INA326
5
1
2
GND
ZVP4525
(zetex)
−
4
Load
(High- Voltage
p- Channel FET)
8.45kΩ
+5V
3
2
7
6
VO = 0.1V to 4.9V
OPA336PA
1nF
75kΩ
4
49.9kΩ
165kΩ
FIGURE 11. High-Side +48V Current Shunt Monitor.
14
INA326, INA327
SBOS222D
www.ti.com
2
1
+
VO = VIN (100) + VDAC
6
VIN
2kΩ
INA326
8
3
5
−
+5V
1nF
100kΩ
2
1
7
INA326
4
+15V
7
DAC
VDAC = 0.075V
to 4.925V
6
NC(1)
R1
VD
2
3
5
8
3
6
VO
OPA277
FIGURE 12. Output Offset Adjustment.
4
(2)
VCM
−15V
+1.8V to +5V
R2
C2
+5V
9
Logic
NOTES: (1) NC denotes No Connection.
(2) Typical swing capability −20mV to (+5V + 100mV).
2
1
6
Enable
8
R1
INA327
7
10
3
4
(1)
R2
1nF
FIGURE 14. Output from Pin 5 to Allow Swing Beyond the Rail.
+5V
9
2
1
6
Enable
8
R3
VO
INA327
7
10
3
+5V
4
0V < VDAC < +5V
(1)
R4
1nF
((+VREF) − (VDAC))
2
1
IOUT
=
± 50nA
7
INA326
4
DAC
R1
R1
200kΩ
6
8
3
5
+5V
9
VREF = +2.5V
RF = 10k
CF
2
1
6
Enable
8
R5
INA327
NOTE: Output resistance is typically 800MΩ.
Resolution < 5nA. Recommended values of CF = 1nF to 1µF.
7
10
3
4
(1)
R6
1nF
NOTE: (1) R2, R4, and R6 could be a
single, shared resistor to save board space.
FIGURE 13. Multiplexed Output.
FIGURE 15. Programmable ±25µA Current Source with High
Output Resistance.
INA326, INA327
15
SBOS222D
www.ti.com
VREF = +2.5V
+2.5V
2
1
7
INA326
4
DAC
6
RI = 200kΩ
5
8
3
49.9Ω
10kΩ
0.1µF
−2.5V
IO
V
REF − VDAC
10kΩ
49.9Ω
IOUT = 2
1 +
RL
200kΩ
IO = ±5mA with
0.1µA stability.
FIGURE 16. Programmable ±5mA Current Source.
RI = 1kΩ
RF = 100kΩ
VI
+30V
7
20kΩ
2
3
6
VO = –27V
OPA551
+5V
4
VOS = –100µV at 200mA
IB
20kΩ
2
1
−30V
RF
7
G = −
= −100V/V
RI
6
2kΩ
INA326
5
8
3
Offset of the high- voltage op amp
is controlled by the INA326.
4
Internal charge pump in the INA326 allows
this node to swing 20mV below ground
without a negative supply.
1MΩ
10nF
NOTES: (1) The OPA551 is a 60V op amp. (2) The INA326 does not require a
negative supply to correct for negative VOS values from the high-voltage op amp.
(3) Voltage offset contribution of IB (OPA551) is 100pA • 2kΩ = 0.2µV.
FIGURE 17. ±27V Output at 200mA Amplifier with 100µV Offset.
INA326, INA327
16
SBOS222D
www.ti.com
FIGURE 18. Single-Supply PID Temperature Control Loop.
INA326, INA327
17
SBOS222D
www.ti.com
PACKAGE OPTION ADDENDUM
www.ti.com
3-Nov-2004
PACKAGING INFORMATION
ORDERABLE DEVICE
STATUS(1)
PACKAGE TYPE
PACKAGE DRAWING
PINS
PACKAGE QTY
INA326EA/250
INA326EA/2K5
INA327EA/250
INA327EA/2K5
ACTIVE
ACTIVE
ACTIVE
ACTIVE
VSSOP
VSSOP
VSSOP
VSSOP
DGK
DGK
DGS
DGS
8
8
250
2500
250
10
10
2500
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
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TI warrants performance of its hardware products to the specifications applicable at the time of sale in
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