LM21212-1 [TI]
具有频率同步功能的 2.95-5.5V、12A、电压模式同步降压稳压器;
| 型号： | LM21212-1 |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | 具有频率同步功能的 2.95-5.5V、12A、电压模式同步降压稳压器 开关 控制器 开关式稳压器 开关式控制器 电源电路 开关式稳压器或控制器 |
| 文件： | 总33页 (文件大小：1074K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LM21212-1
www.ti.com
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
12A High Efficiency Synchronous Point of Load Buck Regulator with Frequency
Synchronization
Check for Samples: LM21212-1
1
FEATURES
DESCRIPTION
The LM21212-1 is a monolithic synchronous point of
load buck regulator that is capable of delivering up to
12A of continuous output current while producing an
output voltage down to 0.6V with outstanding
efficiency. The device is optimized to work over an
input voltage range of 2.95V to 5.5V, making it suited
for a wide variety of low voltage systems. The voltage
mode control loop provides high noise immunity,
narrow duty cycle capability and can be compensated
to be stable with any type of output capacitance,
providing maximum flexibility and ease of use.
2
•
Integrated 7.0 mΩ High Side and 4.3 mΩ Low
Side FET Switches
•
•
300 kHz to 1.5 MHz Frequency SYNC pin
Adjustable Output Voltage From 0.6V to VIN
(100% duty cycle capable), ±1% Reference
•
•
•
•
•
•
•
Input Voltage Range 2.95V to 5.5V
Startup Into Pre-Biased Loads
Output Voltage Tracking Capability
Wide Bandwidth Voltage Loop Error Amplifier
Adjustable Soft-Start With External Capacitor
Precision Enable Pin With Hysteresis
The LM21212-1 features internal over voltage
protection (OVP) and over-current protection (OCP)
for increased system reliability. A precision enable pin
and integrated UVLO allow turn-on of the device to
be tightly controlled and sequenced. Startup inrush
currents are limited by both an internally fixed and
externally adjustable soft-start circuit. Fault detection
and supply sequencing are possible with the
integrated power good circuit.
Integrated OVP, OCP, OTP, UVLO and Power-
Good
•
Thermally Enhanced HTSSOP-20 Exposed Pad
Package
APPLICATIONS
The LM21212-1 is designed to work well in multi-rail
power supply architectures. The output voltage of the
device can be configured to track an external voltage
rail using the SS/TRK pin. The switching frequency
can be synchronized to the falling edge of a clock
between frequencies of 300kHz to 1.5MHz.
•
•
•
Broadband, Networking and Wireless
Communications
High-Performance FPGAs, ASICs and
Microprocessors
Simple to Design, High Efficiency Point of
Load Regulation From a 5V or 3.3V Bus
If the output is pre-biased at startup, it will not sink
current, allowing the output to smoothly rise past the
pre-biased voltage. The regulator is offered in a 20-
pin HTSSOP package with an exposed pad that can
be soldered to the PCB, eliminating the need for
bulky heatsinks.
SIMPLIFIED APPLICATION CIRCUIT
HTSSOP-20
L
OUT
C
5,6,7
11-16
V
PVIN
AVIN
SW
V
OUT
IN
C
IN
R
F
OUT
4
3
C
C3
R
R
FB1
C
F
R
C2
LM21212-1
19
18
FB
EN
C
C1
R
C1
optional
optional
COMP
FB2
2
1
SS/
TRK
C
C
C2
SS
17
PGOOD
SYNC
PGND AGND
8,9,10
20
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
2
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2011–2013, Texas Instruments Incorporated
LM21212-1
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
www.ti.com
CONNECTION DIAGRAM
Top View
1
2
3
4
5
6
7
8
9
20 AGND
19 FB
SYNC
SS/TRK
EN
18 COMP
17 PGOOD
16 SW
AVIN
PVIN
PVIN
PVIN
PGND
PGND
EP
15 SW
14 SW
13 SW
12 SW
PGND 10
11 SW
Figure 1. Top View
HTSSOP-20 Package
PIN DESCRIPTIONS
Pins
Name
Description
1
SYNC
Frequency Synchronization input pin. Applying a clock signal to this pin will force the device to
switch at the clock frequency. If left unconnected, the frequency will default to 1 MHz.
2
SS/TRK
Soft-start control pin. An internal 2 µA current source charges an external capacitor connected
between this pin and AGND to set the output voltage ramp rate during startup. This pin can also be
used to configure the tracking feature.
3
4
EN
Active high enable input for the device. If not used, the EN pin can be left open, which will go high
due to an internal current source.
AVIN
Analog input voltage supply that generates the internal bias. It is recommended to connect PVIN to
AVIN through a low pass RC filter to minimize the influence of input rail ripple and noise on the
analog control circuitry.
5,6,7
PVIN
Input voltage to the power switches inside the device. These pins should be connected together at
the device. A low ESR input capacitance should be located as close as possible to these pins.
8,9,10
11-16
17
PGND
SW
Power ground pins for the internal power switches.
Switch node pins. These pins should be tied together locally and connected to the filter inductor.
Open-drain power good indicator.
PGOOD
COMP
FB
18
Compensation pin is connected to the output of the voltage loop error amplifier.
Feedback pin is connected to the inverting input of the voltage loop error amplifier.
Quiet analog ground for the internal reference and bias circuitry.
19
20
AGND
EP
Exposed Pad
Exposed metal pad on the underside of the package with an electrical and thermal connection to
PGND. It is recommended to connect this pad to the PC board ground plane in order to improve
thermal dissipation.
2
Submit Documentation Feedback
Copyright © 2011–2013, Texas Instruments Incorporated
Product Folder Links: LM21212-1
LM21212-1
www.ti.com
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
ABSOLUTE MAXIMUM RATINGS(1)(2)
PVIN(3), AVIN to GND
SW(4), EN, FB, COMP, PGOOD, SS/TRK to GND
−0.3V to +6V
−0.3V to PVIN + 0.3V
−65°C to 150°C
260°C
Storage Temperature
Lead Temperature (Soldering, 10 sec.)
(5)
ESD Rating, Human Body Model
2kV
(1) Absolute Maximum Ratings indicate limits beyond witch damage to the device may occur. Operating Ratings indicate conditions for
which the device is intended to be functional, but do not ensure specific performance limits. For ensured specifications and test
conditions, see the Electrical Characteristics.
(2) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and
specifications.
(3) The PVIN pin can tolerate transient voltages up to 6.5 V for a period of up to 6ns. These transients can occur during the normal
operation of the device.
(4) The SW pin can tolerate transient voltages up to 9.0 V for a period of up to 6ns, and -1.0V for a duration of 4ns. These transients can
occur during the normal operation of the device.
(5) The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor to each pin.
OPERATING RATINGS(1)
PVIN, AVIN to GND
+2.95V to +5.5V
−40°C to +125°C
24°C/W
Junction Temperature
(2)
θJA
(1) Absolute Maximum Ratings indicate limits beyond witch damage to the device may occur. Operating Ratings indicate conditions for
which the device is intended to be functional, but do not ensure specific performance limits. For ensured specifications and test
conditions, see the Electrical Characteristics.
(2) Thermal measurements were performed on a 2x2 inch, 4 layer, 2 oz. copper outer layer, 1 oz.copper inner layer board with twelve 8 mil.
vias underneith the EP of the device and an additional sixteen 8 mil. vias under the unexposed package.
ELECTRICAL CHARACTERISTICS
Unless otherwise stated, the following conditions apply: VPVIN, AVIN = 5V. Limits in standard type are for TJ = 25°C only, limits
in bold face type apply over the junction temperature (TJ) range of −40°C to +125°C. Minimum and maximum limits are
specified through test, design, or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25°C,
and are provided for reference purposes only.
Symbol
SYSTEM
VFB
Parameter
Conditions
Min
Typ
Max
Units
Feedback pin voltage
VIN = 2.95V to 5.5V
-1%
0.6
1%
V
ΔVOUT/ΔIOUT Load Regulation
0.02
%VOUT
A
/
/
ΔVOUT/ΔVIN Line Regulation
0.1
%VOUT
V
RDSON HS
RDSON LS
ICLR
High Side Switch On Resistance
ISW = 12A
ISW = 12A
7.0
4.3
17
9.0
6.0
19
mΩ
mΩ
A
Low Side Switch On Resistance
HS Rising Switch Current Limit
LS Falling Switch Current Limit
Zero Cross Voltage
15
ICLF
12
A
VZX
-8
3
12
3.0
70
mV
mA
µA
V
IQ
Operating Quiescent Current
Shutdown Quiescent Current
AVIN Under Voltage Lockout
AVIN Under Voltage Lockout Hysteresis
1.5
50
ISD
VEN = 0V
VUVLO
VUVLOHYS
VTRACKOS
ISS
AVIN Rising
2.45
140
-10
2.70
200
6
2.95
280
20
mV
mV
µA
µs
SS/TRACK PIN accuracy (VSS - VFB
Soft-Start Pin Source Current
)
0 < VTRACK < 0.55V
1.3
1.9
500
2.5
675
tINTSS
Internal Soft-Start Ramp to Vref
CSS = 0
350
Copyright © 2011–2013, Texas Instruments Incorporated
Submit Documentation Feedback
3
Product Folder Links: LM21212-1
LM21212-1
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
www.ti.com
ELECTRICAL CHARACTERISTICS (continued)
Unless otherwise stated, the following conditions apply: VPVIN, AVIN = 5V. Limits in standard type are for TJ = 25°C only, limits
in bold face type apply over the junction temperature (TJ) range of −40°C to +125°C. Minimum and maximum limits are
specified through test, design, or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25°C,
and are provided for reference purposes only.
Symbol
SYSTEM
tRESETSS
OSCILLATOR
fSYNCR
Parameter
Conditions
Min
Typ
Max
Units
DeviceReset to Soft-Start Ramp
50
110
200
µs
SYNC Frequency Range
300
950
1500
1050
kHz
kHz
ns
ns
ns
ns
ns
ns
V
fDEFAULT
tSY_SW
Default (no SYNC signal) Frequency
Time from SYNC falling to VSW Rising
Minimum SYNC pin pulse width, high or low
HS OCP Blanking Time
1000
200
100
55
tSY_MIN
tHSBLANK
tLSBLANK
tZXBLANK
tMINON
Rising edge of SW to ICLR comparison
Falling edge of SW to ICLF comparison
Falling edge of SW to VZX comparison
LS OCP Blanking Time
400
120
140
0.8
Zero Cross Blanking Time
Minimum HS on-time
ΔVramp
PWM Ramp p-p Voltage
ERROR AMPLIFIER
VOL
GBW
Error Amplifier Open Loop Voltage Gain
ICOMP = -65µA to 1mA
VFB = 0.6V
95
11
1
dBV/V
MHz
nA
Error Amplifier Gain-Bandwidth Product
Feedback Pin Bias Current
IFB
ICOMPSRC
ICOMPSINK
POWERGOOD
VOVP
COMP Output Source Current
COMP Output Sink Current
1
mA
65
µA
Over Voltage Protection Rising Threshold
Over Voltage Protection Hysteresis
VFB Rising
VFB Falling
VFB Rising
VFB Falling
105
82
112.5
2
120
97
%VFB
%VFB
%VFB
%VFB
µs
VOVPHYS
VUVP
VUVPHYS
tPGDGL
Under Voltage Protection Rising Threshold
Under Voltage Protection Hysteresis
90
2.5
15
PGOOD Deglitch Low (OVP/UVP Condition
Duration to PGOOD Falling)
tPGDGH
PGOOD Deglitch High (minimum low pulse)
PGOOD Pull-down Resistance
12
20
1
µs
Ω
RPGOOD
10
40
IPGOODLEAK PGOOD Leakage Current
VPGOOD = 5V
nA
LOGIC
VIHSYNC
VILSYNC
VIHENR
VENHYS
IEN
SYNC Pin Logic High
SYNC Pin Logic Low
EN Pin Rising Threshold
EN Pin Hysteresis
2.0
V
V
0.8
1.45
180
VEN Rising
VEN = 0V
1.20
50
1.35
110
2
V
mV
µA
EN Pin Pullup Current
THERMAL SHUTDOWN
TTHERMSD
Thermal Shutdown
165
10
°C
°C
TTHERMSDHYS Thermal Shutdown Hysteresis
4
Submit Documentation Feedback
Copyright © 2011–2013, Texas Instruments Incorporated
Product Folder Links: LM21212-1
LM21212-1
www.ti.com
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
TYPICAL PERFORMANCE CHARACTERISTICS
Unless otherwise specified: VVIN = 5V, VOUT = 1.2V, L= 0.56µH (1.8mΩ RDCR), CSS = 33nF, fSW = 1 MHz, TA = 25°C for
efficiency curves, loop gain plots and waveforms, and TJ = 25°C for all others.
Efficiency
Efficiency
96
94
92
90
88
86
84
82
80
100
98
96
94
92
90
88
86
84
82
80
FSW = 500kHz
VOUT = 3.3
VOUT = 1.2
FSW = 1MHz
FSW = 1.5MHz
0
2
4
6
8
10
12
0
2
4
6
8
10
12
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Figure 2.
Figure 3.
Efficiency
(VOUT = 2.5 V, fSW= 300 kHz , Inductor P/N SER2010-
102MLD)
Load Regulation
100
98
96
94
92
90
88
86
0.04
0.03
0.02
0.01
0.00
-0.01
-0.02
-0.03
-0.04
VIN = 3.3V
VIN = 4.0V
VIN = 5.0V
VIN = 5.5V
84
82
80
VIN = 3.3V
VIN = 5.0V
0
2
4
6
8
10
12
0
2
4
6
8
10
12
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Figure 4.
Figure 5.
Copyright © 2011–2013, Texas Instruments Incorporated
Submit Documentation Feedback
5
Product Folder Links: LM21212-1
LM21212-1
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
www.ti.com
TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Unless otherwise specified: VVIN = 5V, VOUT = 1.2V, L= 0.56µH (1.8mΩ RDCR), CSS = 33nF, fSW = 1 MHz, TA = 25°C for
efficiency curves, loop gain plots and waveforms, and TJ = 25°C for all others.
Line Regulation
Non-Switching IQTOTAL vs. VIN
0.10
0.08
0.06
0.04
0.02
0.00
-0.02
-0.04
-0.06
-0.08
-0.10
1.5
1.4
1.3
1.2
1.1
1.0
IOUT = 0A
IOUT = 12A
3.0
3.5
4.0
4.5
5.0
5.5
3.0
3.5
4.0
4.5
5.0
5.5
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Figure 6.
Figure 7.
Non-Switching IAVIN and IPVIN vs. Temperature
VFB vs. Temperature
1.20
1.17
1.14
1.11
1.08
1.05
1.02
0.99
0.96
0.93
0.90
0.180
0.172
0.164
0.156
0.148
0.140
0.132
0.124
0.116
0.108
0.100
0.602
0.601
0.600
0.599
0.598
IAVIN
IPVIN
-40 -20
0
20 40 60 80 100 120
-40 -20
0
20 40 60 80 100 120
JUNCTION TEMPERATURE (°C)
JUNCTION TEMPERATURE (°C)
Figure 8.
Figure 9.
Enable Threshold and Hysteresis vs. Temperature
UVLO Threshold and Hysteresis vs. Temperature
160
2.80
2.78
2.76
2.74
2.72
2.70
2.68
2.66
2.64
2.62
2.60
300
285
270
255
240
225
210
195
180
165
150
V IHENR
V ENHYS
V UVLO
V UVLOHYS
1.37
1.36
1.35
1.34
1.33
1.32
1.31
1.30
1.29
1.28
152
144
136
128
120
112
104
96
88
80
-40 -20
0
20 40 60 80 100 120
-40 -20
0
20 40 60 80 100 120
JUNCTION TEMPERATURE (°C)
JUNCTION TEMPERATURE (°C)
Figure 10.
Figure 11.
6
Submit Documentation Feedback
Copyright © 2011–2013, Texas Instruments Incorporated
Product Folder Links: LM21212-1
LM21212-1
www.ti.com
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Unless otherwise specified: VVIN = 5V, VOUT = 1.2V, L= 0.56µH (1.8mΩ RDCR), CSS = 33nF, fSW = 1 MHz, TA = 25°C for
efficiency curves, loop gain plots and waveforms, and TJ = 25°C for all others.
Enable Low Current vs. Temperature
OVP/UVP Threshold vs. Temperature
60
58
56
54
52
50
48
46
44
42
40
0.68
VUVP
VOVP
0.66
0.64
0.62
0.60
0.58
0.57
0.54
0.52
0.50
-40 -20
0
20 40 60 80 100 120
-40 -20
0
20 40 60 80 100 120
JUNCTION TEMPERATURE (°C)
JUNCTION TEMPERATURE (°C)
Figure 12.
Figure 13.
Minimum On-Time vs. Temperature
FET Resistance vs. Temperature
160
156
152
148
144
140
136
132
128
124
120
10
9
LOW SIDE
HIGH SIDE
8
7
6
5
4
3
2
-40 -20
0
20 40 60 80 100 120
-40 -20
0
20 40 60 80 100 120
JUNCTION TEMPERATURE (°C)
JUNCTION TEMPERATURE (°C)
Figure 14.
Figure 15.
Peak Current Limit vs. Temperature
17.5
17.4
17.3
17.2
17.1
17.0
16.9
16.8
16.7
16.6
16.5
-40 -20
0
20 40 60 80 100 120
AMBIENT TEMPERATURE (°C)
Figure 16.
Copyright © 2011–2013, Texas Instruments Incorporated
Submit Documentation Feedback
7
Product Folder Links: LM21212-1
LM21212-1
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
www.ti.com
TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Unless otherwise specified: VVIN = 5V, VOUT = 1.2V, L= 0.56µH (1.8mΩ RDCR), CSS = 33nF, fSW = 1 MHz, TA = 25°C for
efficiency curves, loop gain plots and waveforms, and TJ = 25°C for all others.
SYNC Signal Lost
SYNC Signal Acquired
VOUT (200mV/Div)
VSYNC (2V/Div)
VOUT (200 mV/Div)
VSYNC (2V/Div)
VSWITCH (2V/Div)
VSWITCH (2V/Div)
4 µs/DIV
4 µs/DIV
Figure 17.
Figure 18.
Load Transient Response
Output Voltage Ripple
VOUT (50 mV/Div)
VOUT (10 mV/Div)
IOUT (5A/Div)
100 µs/DIV
Figure 19.
1 µs/DIV
Figure 20.
Startup with Prebiased Output
Startup with SS/TRK Open Circuit
VOUT (500 mV/Div)
VOUT (500 mV/Div)
VPGOOD (5V/Div)
VENABLE (5V/Div)
VPGOOD (5V/Div)
VENABLE (5V/Div)
IOUT (10A/Div)
200 µs/DIV
Figure 22.
2 ms/DIV
Figure 21.
8
Submit Documentation Feedback
Copyright © 2011–2013, Texas Instruments Incorporated
Product Folder Links: LM21212-1
LM21212-1
www.ti.com
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Unless otherwise specified: VVIN = 5V, VOUT = 1.2V, L= 0.56µH (1.8mΩ RDCR), CSS = 33nF, fSW = 1 MHz, TA = 25°C for
efficiency curves, loop gain plots and waveforms, and TJ = 25°C for all others.
Startup with applied Track Signal
Output Over-Current Condition
V
(5V/Div)
PGOOD
VOUT (500 mV/Div)
V
OUT
(1V/Div)
VTRACK (500 mV/Div)
VPGOOD (5V/Div)
IOUT (10A/Div)
I
(10A/Div)
L
200 ms/DIV
Figure 23.
100 µs/DIV
Figure 24.
Copyright © 2011–2013, Texas Instruments Incorporated
Submit Documentation Feedback
9
Product Folder Links: LM21212-1
LM21212-1
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
www.ti.com
BLOCK DIAGRAM
PLL
SYNC
Ilimit high
PVIN
Over
temp
VREF
+
-
PVIN
Driver
AVIN
UVLO
2.7V
SD
+
-
OR
Precision
enable
AVIN
EN
1.35V
+
-
Control
Logic
PWM
Zero-cross
comparator
AVIN
+
-
OSC
RAMP
+
-
SW
PWM
INT
SS
PVIN
Driver
+
0.6V
SS/TRK
FB
EA
-
OVP
COMP
+
-
Ilimit low
0.68V
0.54V
Powerbad
OR
+
-
PGND
UVP
PGOOD
OR
AGND
10
Submit Documentation Feedback
Copyright © 2011–2013, Texas Instruments Incorporated
Product Folder Links: LM21212-1
LM21212-1
www.ti.com
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
OPERATION DESCRIPTION
GENERAL
The LM21212-1 switching regulator features all of the functions necessary to implement an efficient low voltage
buck regulator using a minimum number of external components. This easy to use regulator features two
integrated switches and is capable of supplying up to 12A of continuous output current. The regulator utilizes
voltage mode control with trailing edge modulation to optimize stability and transient response over the entire
output voltage range. The device can operate at high switching frequency allowing use of a small inductor while
still achieving high efficiency. The precision internal voltage reference allows the output to be set as low as 0.6V.
Fault protection features include: current limiting, thermal shutdown, over voltage protection, and shutdown
capability. The device is available in the HTSSOP-20 package featuring an exposed pad to aid thermal
dissipation. The LM21212-1 can be used in numerous applications to efficiently step-down from a 5V or 3.3V
bus.
FREQUENCY SYNCHRONIZATION
The sync (SYNC) pin allows the LM21212-1 to be switched at an external clock frequency. When a clock signal
is present on the SYNC pin within the allowable frequency range, 300 kHz to 1.5 MHz, the device will
synchronize the turn-on of the high side FET (switch rising) to the negative edge of the clock signal, as seen in
Figure 25 . If no clock signal is present, the LM21212-1 will default to a switching frequency of 1 MHz. The clock
signal can be present on the SYNC pin before the device is powered on with no loading on the clock signal.
Alternatively, if no clock is present while the device is powered up, it will begin switching at the default frequency
of 1 MHz. Once the clock signal is present, the device will begin synchronizing to the clock frequency. The length
of time necessary for the synchronization depends on the clock frequency.
V
SYNC
V
IH_SYNC
V
IL_SYNC
Time
V
SW
V
IN
Time
t
SY_SW
Figure 25. Frequency synchronization
PRECISION ENABLE
The enable (EN) pin allows the output of the device to be enabled or disabled with an external control signal.
This pin is a precision analog input that enables the device when the voltage exceeds 1.35V (typical). The EN pin
has 110 mV of hysteresis and will disable the output when the enable voltage falls below 1.24V (typical). If the
EN pin is not used, it can be left open, and will be pulled high by an internal 2 µA current source. Since the
enable pin has a precise turn-on threshold it can be used along with an external resistor divider network from VIN
to configure the device to turn-on at a precise input voltage.
Copyright © 2011–2013, Texas Instruments Incorporated
Submit Documentation Feedback
11
Product Folder Links: LM21212-1
LM21212-1
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
www.ti.com
UVLO
The LM21212-1 has a built-in under-voltage lockout protection circuit that keeps the device from switching until
the input voltage reaches 2.7V (typical). The UVLO threshold has 200 mV of hysteresis that keeps the device
from responding to power-on glitches during start up. If desired the turn-on point of the supply can be changed
by using the precision enable pin and a resistor divider network connected to VIN as shown in Figure 30 in the
design guide.
CURRENT LIMIT
The LM21212-1 has current limit protection to avoid dangerous current levels on the power FETs and inductor. A
current limit condition is met when the current through the high side FET exceeds the rising current limit level
(ICLR). The control circuitry will respond to this event by turning off the high side FET and turning on the low side
FET. This forces a negative voltage on the inductor, thereby causing the inductor current to decrease. The high
side FET will not conduct again until the lower current limit level (ICLF) is sensed on the low side FET. At this
point, the device will resume normal switching.
A current limit condition will cause the internal soft-start voltage to ramp downward. After the internal soft-start
ramps below the Feedback (FB) pin voltage, (nominally 0.6 V), FB will begin to ramp downward, as well. This
voltage foldback will limit the power consumption in the device, thereby protecting the device from continuously
supplying power to the load under a condition that does not fall within the device SOA. After the current limit
condition is cleared, the internal soft-start voltage will ramp up again. Figure 26 shows current limit behavior with
VSS, VFB, VOUT and VSW
.
SHORT-CIRCUIT PROTECTION
In the unfortunate event that the output is shorted with a low impedance to ground, the LM21212-1 will limit the
current into the short by resetting the device. A short-circuit condition is sensed by a current-limit condition
coinciding with a voltage on the FB pin that is lower than 100 mV. When this condition occurs, the device will
begin its reset sequence, turning off both power FETs and discharging the soft-start capacitor after tRESETSS
(nominally 110 µs). The device will then attempt to restart. If the short-circuit condition still exists, it will reset
again, and repeat until the short-circuit is cleared. The reset prevents excess current flowing through the FETs in
a highly inefficient manner, potentially causing thermal damage to the device or the bus supply.
12
Submit Documentation Feedback
Copyright © 2011–2013, Texas Instruments Incorporated
Product Folder Links: LM21212-1
LM21212-1
www.ti.com
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
I
clr
I
clf
I
L
V
SS
V
FB
100 mV
V
OUT
V
SW
SHORT-CIRCUIT
REMOVED
CURRENT LIMIT
SHORT-CIRCUIT
Figure 26. Current Limit Conditions
THERMAL PROTECTION
Internal thermal shutdown circuitry is provided to protect the integrated circuit in the event that the maximum
junction temperature is exceeded. When activated, typically at 165°C, the LM21212-1 tri-states the power FETs
and resets soft start. After the junction cools to approximately 155°C, the device starts up using the normal start
up routine. This feature is provided to prevent catastrophic failures from accidental device overheating. Note that
thermal limit will not stop the die from operating above the specified maximum operating temperature,125°C. The
die should be kept under 125°C to ensure correct operation.
POWERGOOD FLAG
The PGOOD pin provides the user with a way to monitor the status of the LM21212-1. In order to use the
PGOOD pin, the application must provide a pull-up resistor to a desired DC voltage (i.e. Vin). PGOOD will
respond to a fault condition by pulling the PGOOD pin low with the open-drain output. PGOOD will pull low on
the following conditions – 1) VFB moves above or below the VOVP or VUVP, respectively 2) The enable pin is
brought below the enable threshold 3) The device enters a pre-biased output condition (VFB>VSS).
Figure 27 shows the conditions that will cause PGOOD to fall.
Copyright © 2011–2013, Texas Instruments Incorporated
Submit Documentation Feedback
13
Product Folder Links: LM21212-1
LM21212-1
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
www.ti.com
t
RESETSS
0.6V
V
ss
V
V
ovp
V
V
OVPHYS
V
V
FB
uvp
UVPHYS
EN
V
PGOOD
V
SW
PRE-BIASED
STARTUP
OVP
UVP
DISABLE
t
PGDGL
t
PGDGH
Figure 27. PGOOD Conditions
LIGHT LOAD OPERATION
The LM21212-1 offers increased efficiency when operating at light loads. Whenever the load current is reduced
to a point where the peak to peak inductor ripple current is greater than two times the load current, the device will
enter the diode emulation mode preventing significant negative inductor current. The point at which this occurs is
the critical conduction boundary and can be calculated by the following equation:
(VIN œ VOUT) x D
IBOUNDARY
=
2 x L x fSW
(1)
Several diagrams are shown in Figure 28 illustrating continuous conduction mode (CCM), discontinuous
conduction mode (DCM), and the boundary condition.
It can be seen that in diode emulation mode, whenever the inductor current reaches zero the SW node will
become high impedance. Ringing will occur on this pin as a result of the LC tank circuit formed by the inductor
and the parasitic capacitance at the node. If this ringing is of concern an additional RC snubber circuit can be
added from the switch node to ground.
At very light loads, usually below 100mA, several pulses may be skipped in between switching cycles, effectively
reducing the switching frequency and further improving light-load efficiency.
14
Submit Documentation Feedback
Copyright © 2011–2013, Texas Instruments Incorporated
Product Folder Links: LM21212-1
LM21212-1
www.ti.com
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
Continuous Conduction Mode (CCM)
V
IN
Time (s)
Continuous Conduction Mode (CCM)
I
AVERAGE
Time (s)
DCM - CCM Boundary
I
AVERAGE
Time (s)
Discontinuous Conduction Mode (DCM)
V
IN
Time (s)
Discontinuous Conduction Mode (DCM)
I
Peak
Time (s)
Figure 28. Modes of Operation for LM21212-1
Copyright © 2011–2013, Texas Instruments Incorporated
Submit Documentation Feedback
15
Product Folder Links: LM21212-1
LM21212-1
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
www.ti.com
DESIGN GUIDE
OUTPUT VOLTAGE
The first step in designing the LM21212-1 application is setting the output voltage. This is done by using a
voltage divider between VOUT and AGND, with the middle node connected to VFB. When operating under steady-
state conditions, the LM21212-1 will force VOUT such that VFB is driven to 0.6 V.
V
OUT
LM21212-1
R
R
FB1
FB
0.6V
FB2
Figure 29. Setting VOUT
A good starting point for the lower feedback resistor, RFB2, is 10kΩ. RFB1 can then be calculated the following
equation:
RFB1 + RFB2
0.6V
VOUT
=
RFB2
(2)
PRECISION ENABLE
The enable (EN) pin of the LM21212-1 allows the output to be toggled on and off. This pin is a precision analog
input. When the voltage exceeds 1.35V, the controller will try to regulate the output voltage as long as the input
voltage has exceeded the UVLO voltage of 2.70V. There is an internal current source connected to EN so if
enable is not used, the device will turn on automatically. If EN is not toggled directly the device can be
preprogrammed to turn on at a certain input voltage higher than the UVLO voltage. This can be done with an
external resistor divider from AVIN to EN and EN to AGND as shown below in Figure 30.
Input Power
Supply
AVIN
LM21212-1
EN
R
R
A
V
OUT
B
Figure 30. Enable Startup Through Vin
The resistor values of RA and RB can be relatively sized to allow EN to reach the enable threshold voltage
depending on the input supply voltage. With the enable current source accounted for, the equation solving for RA
is shown below:
RB
V
PVIN - 1.35V
RA =
1.35V - IENRB
(3)
In the above equation, RA is the resistor from VIN to enable, RB is the resistor from enable to ground, IEN is the
internal enable pull-up current (2µA) and 1.35V is the fixed precision enable threshold voltage. Typical values for
RB range from 10kΩ to 100kΩ.
16
Submit Documentation Feedback
Copyright © 2011–2013, Texas Instruments Incorporated
Product Folder Links: LM21212-1
LM21212-1
www.ti.com
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
SOFT START
When EN has exceeded 1.35V, and both PVIN and AVIN have exceeded the UVLO threshold, the LM21212-1
will begin charging the output linearly to the voltage level dictated by the feedback resistor network. The
LM21212-1 employs a user adjustable soft start circuit to lengthen the charging time of the output set by a
capacitor from the soft start pin to ground. After enable exceeds 1.35V, an internal 2 µA current source begins to
charge the soft start capacitor. This allows the user to limit inrush currents due to a high output capacitance and
not cause an over current condition. Adding a soft-start capacitor can also reduce the stress on the input rail.
Larger capacitor values will result in longer start up times. Use the equation below to approximate the size of the
soft-start capacitor:
tSS x ISS
= CSS
0.6V
(4)
where ISSis nominally 2 µA and tSS is the desired startup time. If VIN is higher than the UVLO level and enable is
toggled high the soft start sequence will begin. There is a small delay between enable transitioning high and the
beginning of the soft start sequence. This delay allows the LM21212-1 to initialize its internal circuitry. Once the
output has charged to 90% of the nominal output voltage the power good flag will transition high. This behavior is
illustrated in Figure 31.
90% V
(V
OUT
)
V
OUT
UVP
Enable
Delay
RESETSS
(t
)
0V
V
EN
V
PGOOD
Time
Soft Start Time (t
ss
)
Figure 31. Soft Start Timing
As shown above, the size of the capacitor is influenced by the nominal feedback voltage level 0.6V, the soft-start
charging current ISS (2 µA), and the desired soft start time. If no soft-start capacitor is used then the LM21212-1
defaults to a minimum startup time of 500 µs. The LM21212-1 will not startup faster than 500 µs. When enable is
cycled or the device enters UVLO, the charge developed on the soft-start capacitor is discharged to reset the
startup process. This also happens when the device enters short circuit mode from an over-current event.
INDUCTOR SELECTION
The inductor (L) used in the application will influence the ripple current and the efficiency of the system. The first
selection criteria is to define a ripple current, ΔIL. In a buck converter, it is typically selected to run between 20%
to 30% of the maximum output current. Figure 32 shows the ripple current in a standard buck converter operating
in continuous conduction mode. Larger ripple current will result in a smaller inductance value, which will lead to a
lower series resistance in the inductor, and improved efficiency. However, larger ripple current will also cause the
device to operate in discontinuous conduction mode at a higher average output current.
Copyright © 2011–2013, Texas Instruments Incorporated
Submit Documentation Feedback
17
Product Folder Links: LM21212-1
LM21212-1
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
www.ti.com
V
SW
V
IN
Time
I
L
I
= I
OUT
DI
L
L AVG
Time
Figure 32. Switch and Inductor Current Waveforms
Once the ripple current has been determined, the appropriate inductor size can be calculated using the following
equation:
(VIN œ VOUT
)
D
L =
fSW
üIL
(5)
OUTPUT CAPACITOR SELECTION
The output capacitor, COUT, filters the inductor ripple current and provides a source of charge for transient load
conditions. A wide range of output capacitors may be used with the LM21212-1 that provide various advantages.
The best performance is typically obtained using ceramic, SP or OSCON type chemistries. Typical trade-offs are
that the ceramic capacitor provides extremely low ESR to reduce the output ripple voltage and noise spikes,
while the SP and OSCON capacitors provide a large bulk capacitance in a small volume for transient loading
conditions.
When selecting the value for the output capacitor, the two performance characteristics to consider are the output
voltage ripple and transient response. The output voltage ripple can be approximated by using the following
formula:
1
DVOUT ꢀ DIL x
RESR
+
8 x fSW x COUT
(6)
where ΔVOUT (V) is the amount of peak to peak voltage ripple at the power supply output, RESR (Ω) is the series
resistance of the output capacitor, fSW (Hz) is the switching frequency, and COUT (F) is the output capacitance
used in the design. The amount of output ripple that can be tolerated is application specific; however a general
recommendation is to keep the output ripple less than 1% of the rated output voltage. Keep in mind ceramic
capacitors are sometimes preferred because they have very low ESR; however, depending on package and
voltage rating of the capacitor the value of the capacitance can drop significantly with applied voltage. The output
capacitor selection will also affect the output voltage droop during a load transient. The peak droop on the output
voltage during a load transient is dependent on many factors; however, an approximation of the transient droop
ignoring loop bandwidth can be obtained using the following equation:
2
L x DIOUTSTEP
VDROOP = DIOUTSTEP x RESR
+
COUT x (VIN - VOUT
)
(7)
where, COUT (F) is the minimum required output capacitance, L (H) is the value of the inductor, VDROOP (V) is the
output voltage drop ignoring loop bandwidth considerations, ΔIOUTSTEP (A) is the load step change, RESR (Ω) is
the output capacitor ESR, VIN (V) is the input voltage, and VOUT (V) is the set regulator output voltage. Both the
tolerance and voltage coefficient of the capacitor should be examined when designing for a specific output ripple
or transient droop target.
18
Submit Documentation Feedback
Copyright © 2011–2013, Texas Instruments Incorporated
Product Folder Links: LM21212-1
LM21212-1
www.ti.com
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
INPUT CAPACITOR SELECTION
Quality input capacitors are necessary to limit the ripple voltage at the PVIN pin while supplying most of the
switch current during the on-time. Additionally, they help minimize input voltage droop in an output current
transient condition. In general, it is recommended to use a ceramic capacitor for the input as it provides both a
low impedance and small footprint. Use of a high grade dielectric for the ceramic capacitor, such as X5R or X7R,
will provide improved over-temperature performance and also minimize the DC voltage derating that occurs with
Y5V capacitors. The input capacitors CIN1 and CIN2 should be placed as close as possible to the PVIN and PGND
pins.
Non-ceramic input capacitors should be selected for RMS current rating and minimum ripple voltage. A good
approximation for the required ripple current rating is given by the relationship:
IIN-RMS = IOUT D(1 - D)
(8)
As indicated by the RMS ripple current equation, highest requirement for RMS current rating occurs at 50% duty
cycle. For this case, the RMS ripple current rating of the input capacitor should be greater than half the output
current. For best performance, low ESR ceramic capacitors should be placed in parallel with higher capacitance
capacitors to provide the best input filtering for the device.
When operating at low input voltages (3.3V or lower), additional capacitance may be necessary to protect from
triggering an under-voltage condition on an output current transient. This will depend on the impedance between
the input voltage supply and the LM21212-1, as well as the magnitude and slew rate of the output transient.
The AVIN pin requires a 1 µF ceramic capacitor to AGND and a 1Ω resistor to PVIN. This RC network will filter
inherent noise on PVIN from the sensitive analog circuitry connected to AVIN.
CONTROL LOOP COMPENSATION
The LM21212-1 incorporates a high bandwidth amplifier between the FB and COMP pins to allow the user to
design a compensation network that matches the application. This section will walk through the various steps in
obtaining the open loop transfer function.
There are three main blocks of a voltage mode buck switcher that the power supply designer must consider
when designing the control system; the power train, modulator, and the compensated error amplifier. A closed
loop diagram is shown in Figure 33.
PWM Modulator
Power Train
V
IN
R
DCR
L
OUT
V
OUT
DRIVER
SW
R
C
ESR
R
O
-
PWM
+
OUT
Error Amplifier and Compensation
0.6V
FB
+
COMP
R
FB1
EA
-
C
R
C1
C1
R
C2
C
C3
R
FB2
C
C2
Figure 33. Loop Diagram
Copyright © 2011–2013, Texas Instruments Incorporated
Submit Documentation Feedback
19
Product Folder Links: LM21212-1
LM21212-1
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
www.ti.com
The power train consists of the output inductor (L) with DCR (DC resistance RDCR), output capacitor (C0) with
ESR (effective series resistance RESR), and load resistance (Ro). The error amplifier (EA) constantly forces FB to
0.6V. The passive compensation components around the error amplifier help maintain system stability. The
modulator creates the duty cycle by comparing the error amplifier signal with an internally generated ramp set at
the switching frequency.
There are three transfer functions that must be taken into consideration when obtaining the total open loop
transfer function; COMP to SW (Modulator) , SW to VOUT (Power Train), and VOUT to COMP (Error Amplifier).
The COMP to SW transfer function is simply the gain of the PWM modulator.
Vin
GPWM
=
üVramp
(9)
where ΔVRAMP is the oscillator peak-to-peak ramp voltage (nominally 0.8 V). The SW to COMP transfer function
includes the output inductor, output capacitor, and output load resistance. The inductor and capacitor create two
complex poles at a frequency described by:
RO + RDCR
1
fLC
=
2p
LOUTCOUT(RO + RESR
)
(10)
In addition to two complex poles, a left half plane zero is created by the output capacitor ESR located at a
frequency described by:
1
fesr
=
2pCoResr
(11)
A Bode plot showing the power train response can be seen below.
60
40
20
0
0
-40
-80
-120
-160
-200
-240
-280
-320
-360
-20
-40
-60
GAIN
PHASE
-80
100
1k
10k 100k
1M
10M
FREQUENCY (HZ)
Figure 34. Power Train Bode Plot
The complex poles created by the output inductor and capacitor cause a 180° phase shift at the resonant
frequency as seen in Figure 34. The phase is boosted back up to -90° due to the output capacitor ESR zero. The
180° phase shift must be compensated out and phase boosted through the error amplifier to stabilize the closed
loop response. The compensation network shown around the error amplifier in Figure 33 creates two poles, two
zeros and a pole at the origin. Placing these poles and zeros at the correct frequencies will stabilize the closed
loop response. The Compensated Error Amplifier transfer function is:
s
s
+1
+1
+1
+1
2pfZ1
2pfZ2
GEA = Km
s
s
s
2pfP1
2pfP2
(12)
20
Submit Documentation Feedback
Copyright © 2011–2013, Texas Instruments Incorporated
Product Folder Links: LM21212-1
LM21212-1
www.ti.com
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
The pole located at the origin gives high open loop gain at DC, translating into improved load regulation
accuracy. This pole occurs at a very low frequency due to the limited gain of the error amplifier, however, it can
be approximated at DC for the purposes of compensation. The other two poles and two zeros can be located
accordingly to stabilize the voltage mode loop depending on the power stage complex poles and Q. Figure 35 is
an illustration of what the Error Amplifier Compensation transfer function will look like.
100
80
60
40
20
0
90
45
GAIN
PHASE
0
-45
-90
-135
-180
-20
100
1k
10k 100k 1M
10M
FREQUENCY (Hz)
Figure 35. Type 3 Compensation Network Bode Plot
As seen in Figure 35, the two zeros (fLC/2, fLC) in the comensation network give a phase boost. This will cancel
out the effects of the phase loss from the output filter. The compensation network also adds two poles to the
system. One pole should be located at the zero caused by the output capacitor ESR (fESR) and the other pole
should be at half the switching frequency (fSW/2) to roll off the high frequency response. The dependancy of the
pole and zero locations on the compensation components is described below.
fLC
1
fZ1
=
=
=
2pRC1CC1
2
1
fZ2 = fLC
2p(RC1 + RFB1)CC3
1
fP1 = fESR
=
=
2pRC2CC3
fsw
CC1 + CC2
fP2
=
2pRC1 CC1CC2
2
(13)
An example of the step-by-step procedure to generate compensation component values using the typical
application setup (see Figure 40) is given. The parameters needed for the compensation values are given in the
table below.
Parameter
VIN
Value
5.0V
VOUT
IOUT
fCROSSOVER
L
1.2V
12A
100 kHz
0.56 µH
1.8 mΩ
150 µF
1.0 mΩ
0.8V
RDCR
CO
RESR
ΔVRAMP
fSW
500 kHz
Copyright © 2011–2013, Texas Instruments Incorporated
Submit Documentation Feedback
21
Product Folder Links: LM21212-1
LM21212-1
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
www.ti.com
where ΔVRAMP is the oscillator peak-to-peak ramp voltage (nominally 0.8V), and fCROSSOVER is the frequency at
which the open-loop gain is a magnitude of 1. It is recommended that the fcrossover not exceed one-fifth of the
switching frequency. The output capacitance, CO, depends on capacitor chemistry and bias voltage. For Multi-
Layer Ceramic Capacitors (MLCC), the total capacitance will degrade as the DC bias voltage is increased.
Measuring the actual capacitance value for the output capacitors at the output voltage is recommended to
accurately calculate the compensation network. The example given here is the total output capacitance using the
three MLCC output capacitors biased at 1.2V, as seen in the typical application schematic, Figure 40. Note that it
is more conservative, from a stability standpoint, to err on the side of a smaller output capacitance value in the
compensation calculations rather than a larger, as this will result in a lower bandwidth but increased phase
margin.
First, a the value of RFB1 should be chosen. A typical value is 10kΩ. From this, the value of RC1 can be
calculated to set the mid-band gain so that the desired crossover frequency is achieved:
DVRAMP
fcrossover
RFB1
RC1
=
VIN
fLC
100 kHz 0.8 V
10 kW
=
=
5.0 V
17.4 kHz
9.2 kW
(14)
(15)
Next, the value of CC1 can be calculated by placing a zero at half of the LC double pole frequency (fLC):
1
CC1
=
pfLCRC1
= 1.99 nF
Now the value of CC2 can be calculated to place a pole at half of the switching frequency (fSW):
CC1
CC2
=
pfSWRC1 CC1 -1
= 71 pF
(16)
(17)
RC2 can then be calculated to set the second zero at the LC double pole frequency:
R
FB1 fLC
fESR - fLC
RC2
=
= 166W
Last, CC3 can be calculated to place a pole at the same frequency as the zero created by the output capacitor
ESR:
1
CC3
=
2pfESRRC2
= 898 pF
(18)
An illustration of the total loop response can be seen in Figure 36.
22
Submit Documentation Feedback
Copyright © 2011–2013, Texas Instruments Incorporated
Product Folder Links: LM21212-1
LM21212-1
www.ti.com
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
200
150
100
50
160
140
120
100
80
GAIN
PHASE
60
40
20
0
0
-20
-40
-50
10
100
1k
10k 100k
1M
FREQUENCY (Hz)
Figure 36. Loop Response
It is important to verify the stability by either observing the load transient response or by using a network
analyzer. A phase margin between 45° and 70° is usually desired for voltage mode systems. Excessive phase
margin can cause slow system response to load transients and low phase margin may cause an oscillatory load
transient response. If the load step response peak deviation is larger than desired, increasing fCROSSOVER and
recalculating the compensation components may help but usually at the expense of phase margin.
THERMAL CONSIDERATIONS
The thermal characteristics of the LM21212-1 are specified using the parameter θJA, which relates the junction
temperature to the ambient temperature. Although the value of θJA is dependant on many variables, it still can be
used to approximate the operating junction temperature of the device.
To obtain an estimate of the device junction temperature, one may use the following relationship:
TJ = PD
q
JA + TA
(19)
(20)
and
2
PD = PIN (1 - Efficiency) - IOUT RDCR
Where:
TJ is the junction temperature in °C, PIN is the input power in Watts (PIN = VIN x IIN), θJA is the junction to ambient
thermal resistance for the LM21212-1, TA is the ambient temperature in °C, and IOUT is the output load current.
It is important to always keep the operating junction temperature (TJ) below 125°C for reliable operation. If the
junction temperature exceeds 165°C the device will cycle in and out of thermal shutdown. If thermal shutdown
occurs it is a sign of inadequate heatsinking or excessive power dissipation in the device.
Figure 37, shown below, provides a better approximation of the θJA for a given PCB copper area. The PCB used
in this test consisted of 4 layers: 1oz. copper was used for the internal layers while the external layers were
plated to 2oz. copper weight. To provide an optimal thermal connection, a 3 x 4 array of 8 mil. vias under the
thermal pad were used, and an additional sixteen 8 mil. vias under the rest of the device were used to connect
the 4 layers.
Copyright © 2011–2013, Texas Instruments Incorporated
Submit Documentation Feedback
23
Product Folder Links: LM21212-1
LM21212-1
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
www.ti.com
30
28
26
24
22
20
18
16
14
12
10
2
3
4
5
6
7
8
9
10
2
BOARD AREA (in )
Figure 37. Thermal Resistance vs PCB Area (4 Layer Board)
Figure 38 shows a plot of the maximum ambient temperature vs. output current for the typical application circuit
shown in Figure 40, assuming a θJA value of 24 °C/W.
125
120
115
110
105
100
95
90
85
80
0
2
4
6
8
10
12
IOUT (A)
Figure 38. Maximum Ambient Temperature vs. Output Current (0 LFM)
PCB LAYOUT CONSIDERATIONS
PC board layout is an important part of DC-DC converter design. Poor board layout can disrupt the performance
of a DC-DC converter and surrounding circuitry by contributing to EMI, ground bounce, and resistive voltage loss
in the traces. These can send erroneous signals to the DC-DC converter resulting in poor regulation or instability.
Good layout can be implemented by following a few simple design rules.
1. Minimize area of switched current loops. In a buck regulator there are two loops where currents are switched
at high slew rates. The first loop starts from the input capacitor, to the regulator PVIN pin, to the regulator
SW pin, to the inductor then out to the output capacitor and load. The second loop starts from the output
capacitor ground, to the regulator GND pins, to the inductor and then out to the load (see Figure 39). To
minimize both loop areas, the input capacitor should be placed as close as possible to the VIN pin.
Grounding for both the input and output capacitor should be close. Ideally, a ground plane should be placed
on the top layer that connects the PGND pins, the exposed pad (EP) of the device, and the ground
connections of the input and output capacitors in a small area near pin 10 and 11 of the device. The inductor
should be placed as close as possible to the SW pin and output capacitor.
2. Minimize the copper area of the switch node. The six SW pins should be routed on a single top plane to the
pad of the inductor. The inductor should be placed as close as possible to the switch pins of the device with
24
Submit Documentation Feedback
Copyright © 2011–2013, Texas Instruments Incorporated
Product Folder Links: LM21212-1
LM21212-1
www.ti.com
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
a wide trace to minimize conductive losses. The inductor can be placed on the bottom side of the PCB
relative to the LM21212-1, but care must be taken to not allow any coupling of the magnetic field of the
inductor into the sensitive feedback or compensation traces.
3. Have a solid ground plane between PGND, the EP and the input and output cap. ground connections. The
ground connections for the AGND, compensation, feedback, and soft-start components should be physically
isolated (located near pin 1 and 20) from the power ground plane but a separate ground connection is not
necessary. If not properly handled, poor grounding can result in degraded load regulation or erratic switching
behavior.
4. Carefully route the connection from the VOUT signal to the compensation network. This node is high
impedance and can be susceptible to noise coupling. The trace should be routed away from the SW pin and
inductor to avoid contaminating the feedback signal with switch noise.
5. Make input and output bus connections as wide as possible. This reduces any voltage drops on the input or
output of the converter and can improve efficiency. Voltage accuracy at the load is important so make sure
feedback voltage sense is made at the load. Doing so will correct for voltage drops at the load and provide
the best output accuracy.
6. Provide adequate device heatsinking. For most 12A designs a four layer board is recommended. Use as
many vias as is possible to connect the EP to the power plane heatsink. The vias located underneith the EP
will wick solder into them if they are not filled. Complete solder coverage of the EP to the board is required to
achieve the θJA values described in the previous section. Either an adequate amount of solder must be
applied to the EP pad to fill the vias, or the vias must be filled during manufacturing. See the THERMAL
CONSIDERATIONS section to ensure enough copper heatsinking area is used to keep the junction
temperature below 125°C.
L
LM21212-1
V
OUT
SW
PVIN
V
IN
C
IN
C
OUT
PGND
LOOP2
LOOP1
Figure 39. Schematic of LM21212-1 Highlighting Layout Sensitive Nodes
Copyright © 2011–2013, Texas Instruments Incorporated
Submit Documentation Feedback
25
Product Folder Links: LM21212-1
LM21212-1
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
www.ti.com
HTSSOP-20
LO
5,6,7
11-16
VIN
PVIN
EN
SW
VOUT
3
RF
CC3
RC2
RFB1
4
AVIN
CIN1 CIN2 CIN3
CF
CO1 CO2 CO3
LM21212-1
19
18
FB
CC1
RC1
2
SS /
TRK
COMP
RFB2
CSS
CC2
VIN
RPGOOD
500 kHz
1
17
SYNC
PGOOD
PGND AGND
20
8,9,10
Figure 40. Typical Application Schematic 1
Table 1. Bill of Materials (VIN = 3.3 - 5.5V, VOUT = 1.2V, IOUT = 12A, fSW = 500kHz)
ID
DESCRIPTION
VENDOR
PART NUMBER
QUANTITY
CF
CAP, CERM, 1uF, 10V, +/-10%,
X7R, 0603
MuRata
GRM188R71A105KA61D
1
CIN1, CIN2, CIN3
CO1, CO2, CO3
,
CAP, CERM, 100uF, 6.3V, +/-20%,
X5R, 1206
MuRata
TDK
GRM31CR60J107ME39L
C1608C0G1H182J
6
1
1
1
1
1
CC1
CC2
CC3
CSS
LO
CAP, CERM, 1800pF, 50V, +/-5%,
C0G/NP0, 0603
CAP, CERM, 68pF, 50V, +/-5%,
C0G/NP0, 0603
TDK
C1608C0G1H680J
CAP, CERM, 820pF, 50V, +/-5%,
C0G/NP0, 0603
TDK
C1608C0G1H821J
CAP, CERM, 0.033uF, 16V, +/-10%,
X7R, 0603
MuRata
Vishay-Dale
GRM188R71C333KA01D
IHLP4040DZERR56M01
Inductor, Shielded Drum Core,
Powdered Iron, 560nH, 27.5A,
0.0018 ohm, SMD
RF
RES, 1.0 ohm, 5%, 0.1W, 0603
RES, 9.31k ohm, 1%, 0.1W, 0603
RES, 165 ohm, 1%, 0.1W, 0603
RES, 10k ohm, 1%, 0.1W, 0603
Vishay-Dale
Vishay-Dale
Vishay-Dale
Vishay-Dale
CRCW06031R00JNEA
CRCW06039K31FKEA
CRCW0603165RFKEA
CRCW060310K0FKEA
1
1
1
3
RC1
RC2
RFB1, RFB2, RPGOOD
26
Submit Documentation Feedback
Copyright © 2011–2013, Texas Instruments Incorporated
Product Folder Links: LM21212-1
LM21212-1
www.ti.com
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
HTSSOP-20
LO
5,6,7
11-16
VIN
PVIN
EN
SW
VOUT
REN1
REN2
CSS
3
4
RF
CF
CIN1
CC3
RFB1
AVIN
RC2
CO1 CO2
LM21212-1
19
18
FB
CC1
RC1
2
1
SS/
TRK
RFB2
COMP
CC2
VIN
RPGOOD
17
SYNC
OPEN
PGOOD
PGND AGND
20
8,9,10
Figure 41. Typical Application Schematic 2
Table 2. Bill of Materials (VIN = 4.0 - 5.5V, VOUT = 0.9V, IOUT = 8A, fSW = 1MHz)
ID
CF
DESCRIPTION
VENDOR
PART NUMBER
QUANTITY
CAP, CERM, 1uF, 10V, +/-10%,
X7R, 0603
MuRata
GRM188R71A105KA61D
1
CIN1, CO1, CO2
CAP, CERM, 100uF, 6.3V, +/-20%,
X5R, 1206
MuRata
MuRata
TDK
GRM31CR60J107ME39L
GRM1885C1H182JA01D
C1608C0G1H680J
3
1
1
1
1
1
CC1
CC2
CC3
CSS
LO
CAP, CERM, 1800pF, 50V, +/-5%,
C0G/NP0, 0603
CAP, CERM, 68pF, 50V, +/-5%,
C0G/NP0, 0603
CAP, CERM, 470pF, 50V, +/-5%,
C0G/NP0, 0603
TDK
C1608C0G1H471J
CAP, CERM, 0.033uF, 16V, +/-10%,
X7R, 0603
MuRata
GRM188R71C333KA01D
744314024
Inductor, Shielded Drum Core,
Superflux, 240nH, 20A, 0.001 ohm,
SMD
Wurth Elektronik
eiSos
RF
RES, 1.0 ohm, 5%, 0.1W, 0603
RES, 4.87k ohm, 1%, 0.1W, 0603
RES, 210 ohm, 1%, 0.1W, 0603
RES, 10k ohm, 1%, 0.1W, 0603
RES, 19.6k ohm, 1%, 0.1W, 0603
RES, 20.0k ohm, 1%, 0.1W, 0603
Vishay-Dale
Vishay-Dale
Vishay-Dale
Vishay-Dale
Vishay-Dale
Vishay-Dale
CRCW06031R00JNEA
CRCW06034K87FKEA
CRCW0603210RFKEA
CRCW060310K0FKEA
CRCW060319K6FKEA
CRCW060320K0FKEA
1
1
1
3
1
1
RC1
RC2
REN1, RFB1, RPGOOD
REN2
RFB2
Copyright © 2011–2013, Texas Instruments Incorporated
Submit Documentation Feedback
27
Product Folder Links: LM21212-1
LM21212-1
SNVS671E –FEBRUARY 2011–REVISED MARCH 2013
www.ti.com
REVISION HISTORY
Changes from Revision D (March 2013) to Revision E
Page
•
Changed layout of National Data Sheet to TI format .......................................................................................................... 27
28
Submit Documentation Feedback
Copyright © 2011–2013, Texas Instruments Incorporated
Product Folder Links: LM21212-1
PACKAGE OPTION ADDENDUM
www.ti.com
11-Apr-2013
PACKAGING INFORMATION
Orderable Device
LM21212MH-1/NOPB
LM21212MHE-1/NOPB
LM21212MHX-1/NOPB
Status Package Type Package Pins Package
Eco Plan Lead/Ball Finish
MSL Peak Temp
Op Temp (°C)
Top-Side Markings
Samples
Drawing
Qty
(1)
(2)
(3)
(4)
ACTIVE
HTSSOP
HTSSOP
HTSSOP
PWP
20
20
20
73
Green (RoHS
& no Sb/Br)
CU NIPDAU
CU NIPDAU
CU NIPDAU
Level-1-260C-UNLIM
Level-1-260C-UNLIM
Level-1-260C-UNLIM
-40 to 125
LM21212
MH-1
ACTIVE
ACTIVE
PWP
PWP
250
Green (RoHS
& no Sb/Br)
LM21212
MH-1
2500
Green (RoHS
& no Sb/Br)
-40 to 125
LM21212
MH-1
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4)
Multiple Top-Side Markings will be inside parentheses. Only one Top-Side Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a
continuation of the previous line and the two combined represent the entire Top-Side Marking for that device.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
11-Oct-2013
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
B0
K0
P1
W
Pin1
Diameter Width (mm) (mm) (mm) (mm) (mm) Quadrant
(mm) W1 (mm)
LM21212MHE-1/NOPB HTSSOP PWP
LM21212MHX-1/NOPB HTSSOP PWP
20
20
250
178.0
330.0
16.4
16.4
6.95
6.95
7.1
7.1
1.6
1.6
8.0
8.0
16.0
16.0
Q1
Q1
2500
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
11-Oct-2013
*All dimensions are nominal
Device
Package Type Package Drawing Pins
SPQ
Length (mm) Width (mm) Height (mm)
LM21212MHE-1/NOPB
LM21212MHX-1/NOPB
HTSSOP
HTSSOP
PWP
PWP
20
20
250
210.0
367.0
185.0
367.0
35.0
35.0
2500
Pack Materials-Page 2
MECHANICAL DATA
PWP0020AA
MYB20XX (REV E)
4214875/A
02/2013
A. All linear dimensions are in millimeters. Dimensioning and tolerancing per ASME Y14.5M-1994.
NOTES:
B. This drawing is subject to change without notice.
C. Reference JEDEC Registration MO-153, Variation ACT.
www.ti.com
IMPORTANT NOTICE
Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, enhancements, improvements and other
changes to its semiconductor products and services per JESD46, latest issue, and to discontinue any product or service per JESD48, latest
issue. Buyers should obtain the latest relevant information before placing orders and should verify that such information is current and
complete. All semiconductor products (also referred to herein as “components”) are sold subject to TI’s terms and conditions of sale
supplied at the time of order acknowledgment.
TI warrants performance of its components to the specifications applicable at the time of sale, in accordance with the warranty in TI’s terms
and conditions of sale of semiconductor products. Testing and other quality control techniques are used to the extent TI deems necessary
to support this warranty. Except where mandated by applicable law, testing of all parameters of each component is not necessarily
performed.
TI assumes no liability for applications assistance or the design of Buyers’ products. Buyers are responsible for their products and
applications using TI components. To minimize the risks associated with Buyers’ products and applications, Buyers should provide
adequate design and operating safeguards.
TI does not warrant or represent that any license, either express or implied, is granted under any patent right, copyright, mask work right, or
other intellectual property right relating to any combination, machine, or process in which TI components or services are used. Information
published by TI regarding third-party products or services does not constitute a license to use such products or services or a warranty or
endorsement thereof. Use of such information may require a license from a third party under the patents or other intellectual property of the
third party, or a license from TI under the patents or other intellectual property of TI.
Reproduction of significant portions of TI information in TI data books or data sheets is permissible only if reproduction is without alteration
and is accompanied by all associated warranties, conditions, limitations, and notices. TI is not responsible or liable for such altered
documentation. Information of third parties may be subject to additional restrictions.
Resale of TI components or services with statements different from or beyond the parameters stated by TI for that component or service
voids all express and any implied warranties for the associated TI component or service and is an unfair and deceptive business practice.
TI is not responsible or liable for any such statements.
Buyer acknowledges and agrees that it is solely responsible for compliance with all legal, regulatory and safety-related requirements
concerning its products, and any use of TI components in its applications, notwithstanding any applications-related information or support
that may be provided by TI. Buyer represents and agrees that it has all the necessary expertise to create and implement safeguards which
anticipate dangerous consequences of failures, monitor failures and their consequences, lessen the likelihood of failures that might cause
harm and take appropriate remedial actions. Buyer will fully indemnify TI and its representatives against any damages arising out of the use
of any TI components in safety-critical applications.
In some cases, TI components may be promoted specifically to facilitate safety-related applications. With such components, TI’s goal is to
help enable customers to design and create their own end-product solutions that meet applicable functional safety standards and
requirements. Nonetheless, such components are subject to these terms.
No TI components are authorized for use in FDA Class III (or similar life-critical medical equipment) unless authorized officers of the parties
have executed a special agreement specifically governing such use.
Only those TI components which TI has specifically designated as military grade or “enhanced plastic” are designed and intended for use in
military/aerospace applications or environments. Buyer acknowledges and agrees that any military or aerospace use of TI components
which have not been so designated is solely at the Buyer's risk, and that Buyer is solely responsible for compliance with all legal and
regulatory requirements in connection with such use.
TI has specifically designated certain components as meeting ISO/TS16949 requirements, mainly for automotive use. In any case of use of
non-designated products, TI will not be responsible for any failure to meet ISO/TS16949.
Products
Applications
Audio
www.ti.com/audio
amplifier.ti.com
dataconverter.ti.com
www.dlp.com
Automotive and Transportation www.ti.com/automotive
Communications and Telecom www.ti.com/communications
Amplifiers
Data Converters
DLP® Products
DSP
Computers and Peripherals
Consumer Electronics
Energy and Lighting
Industrial
www.ti.com/computers
www.ti.com/consumer-apps
www.ti.com/energy
dsp.ti.com
Clocks and Timers
Interface
www.ti.com/clocks
interface.ti.com
logic.ti.com
www.ti.com/industrial
www.ti.com/medical
Medical
Logic
Security
www.ti.com/security
Power Mgmt
Microcontrollers
RFID
power.ti.com
Space, Avionics and Defense
Video and Imaging
www.ti.com/space-avionics-defense
www.ti.com/video
microcontroller.ti.com
www.ti-rfid.com
www.ti.com/omap
OMAP Applications Processors
Wireless Connectivity
TI E2E Community
e2e.ti.com
www.ti.com/wirelessconnectivity
Mailing Address: Texas Instruments, Post Office Box 655303, Dallas, Texas 75265
Copyright © 2013, Texas Instruments Incorporated
相关型号：
SI9130DB
5- and 3.3-V Step-Down Synchronous ConvertersWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9135LG-T1
SMBus Multi-Output Power-Supply ControllerWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9135LG-T1-E3
SMBus Multi-Output Power-Supply ControllerWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9135_11
SMBus Multi-Output Power-Supply ControllerWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9136_11
Multi-Output Power-Supply ControllerWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9130CG-T1-E3
Pin-Programmable Dual Controller - Portable PCsWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9130LG-T1-E3
Pin-Programmable Dual Controller - Portable PCsWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9130_11
Pin-Programmable Dual Controller - Portable PCsWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9137
Multi-Output, Sequence Selectable Power-Supply Controller for Mobile ApplicationsWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9137DB
Multi-Output, Sequence Selectable Power-Supply Controller for Mobile ApplicationsWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9137LG
Multi-Output, Sequence Selectable Power-Supply Controller for Mobile ApplicationsWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9122E
500-kHz Half-Bridge DC/DC Controller with Integrated Secondary Synchronous Rectification DriversWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
©2020 ICPDF网 联系我们和版权申明