LM2672N-5.0 [TI]
Power Converter High Efficiency 1A Step-Down Voltage Regulator with Features; 电源转换器高效率1A降压型稳压器与特点
| 型号： | LM2672N-5.0 |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | Power Converter High Efficiency 1A Step-Down Voltage Regulator with Features |
| 文件： | 总16页 (文件大小：667K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
TI Confidential - NDA Restrictions
LM2731
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SNVS217F –MAY 2004–REVISED NOVEMBER 2012
LM2731 0.6/1.6 MHz Boost Converters With 22V Internal FET Switch in SOT-23
Check for Samples: LM2731
1
FEATURES
DESCRIPTION
The LM2731 switching regulators are current-mode
boost converters operating at fixed frequencies of 1.6
MHz (“X” option) and 600 kHz (“Y” option).
2
•
22V DMOS FET Switch
•
1.6 MHz (“X”), 0.6 MHz (“Y”) Switching
Frequency
The use of SOT-23 package, made possible by the
minimal power loss of the internal 1.8A switch, and
use of small inductors and capacitors result in the
industry's highest power density. The 22V internal
switch makes these solutions perfect for boosting to
voltages up to 20V.
•
•
•
•
•
•
•
•
Low RDS(ON) DMOS FET
Switch Current up to 1.8A
Wide Input Voltage Range (2.7V–14V)
Low Shutdown Current (<1 µA)
5-Lead SOT-23 Package
These parts have a logic-level shutdown pin that can
be used to reduce quiescent current and extend
battery life.
Uses Tiny Capacitors and Inductors
Cycle-by-Cycle Current Limiting
Internally Compensated
Protection is provided through cycle-by-cycle current
limiting and thermal shutdown. Internal compensation
simplifies design and reduces component count.
APPLICATIONS
•
•
•
•
•
White LED Current Source
PDA’s and Palm-Top Computers
Digital Cameras
Table 1. Switch Frequency
X
Y
1.6 MHz
0.6 MHz
Portable Phones and Games
Local Boost Regulator
Typical Application Circuit
Efficiency vs Load Current
D1
L1/10mH
100
90
MBR0520
5 VIN
5 - 12V Boost
^ó_ ëꢀŒ•]}v
U1
VIN
SW
12V
OUT
500mA
(TYP)
R3
51K
R1/117K
LM2731 —X“
FB
SHDN
SHDN
GND
80
70
C1
2.2mF
CF
R2
13.3K
C2
220pF
GND
4.7mF
0
500
100 200 300 400
LOAD CURRENT (mA)
Efficiency vs Load Current
D1
MBR0520
L1/6.8mH
100
90
3.3 VIN
SHDN
3.3 -5V Boost
^ò_ ëꢀŒ•]}v
U1
VIN
SW
5V
OUT
R3
51K
R1/40.5K
LM2731 —Y“
FB
700mA
(TYP)
SHDN
GND
80
70
C1
2.2mF
CF
470pF
R2
13.3K
C2
22mF
GND
0
400 600
800
200
LOAD CURRENT (mA)
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
2
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
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LM2731
SNVS217F –MAY 2004–REVISED NOVEMBER 2012
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Efficiency vs Load Current
D1
100
90
80
70
60
50
40
30
20
10
0
L1/6.8mH
U1
MBR0520
3.3 VIN
SHDN
VIN
SW
12V
OUT
230mA
(TYP)
R3
51K
R1/117K
LM2731 —Y“
FB
3.3 -12V Boost
^ò_ ëꢀŒ•]}v
SHDN
GND
C1
2.2mF
CF
270pF
R2
13.3K
C2
10mF
GND
0
50 100
250
150 200
LOAD (mA)
D1
MBR0520
Efficiency vs Load Current
9V OUT
240mA (typ)
L1/10mH
U1
3.3 VIN
SHDN
100
90
80
70
60
50
40
30
20
10
0
VIN
SW
R3
51K
R1/84K
LM2731 —X“
FB
D4
D5
D2
D3
3.3 -9V
^ó_ ëꢀŒ•]}v
SHDN
GND
C1
2.2mF
CF
330pF
R2
13.3K
C2
4.7mF
R5
R4
GND
0
50
150
200 250 300
100
LOAD (mA)
B1
LI-ION
3.3 - 4.2V
L1 / 1.5 mH
D1
MBR0520
-
+
VIN
SW
R3
51K
LM2731"Y"
FB
SHDN
GND
WHITE
LED's
FLASH
ENABLE
0
C2
4.7mF
C1
4.7mF
R2
120
Figure 1. White LED Flash Application
Connection Diagram
Top View
Figure 2. 5-Lead SOT-23 Package
See Package Number DBV0005A
2
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SNVS217F –MAY 2004–REVISED NOVEMBER 2012
PIN DESCRIPTIONS
Pin
1
Name
SW
Function
Drain of the internal FET switch.
2
GND
FB
Analog and power ground.
3
Feedback point that connects to external resistive divider.
4
SHDN
VIN
Shutdown control input. Connect to Vin if the feature is not used.
Analog and power input.
5
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
Absolute Maximum Ratings(1)
Storage Temperature Range
Operating Junction Temperature Range
Lead Temp. (Soldering, 5 sec.)
Power Dissipation(2)
FB Pin Voltage
−65°C to +150°C
−40°C to +125°C
300°C
Internally Limited
−0.4V to +6V
−0.4V to +22V
−0.4V to +14.5V
−0.4V to VIN + 0.3V
265°C/W
SW Pin Voltage
Input Supply Voltage
SHDN Pin Voltage
θJA (SOT23-5)
ESD Rating(3)
Human Body Model
2 kV
(1) Absolute Maximum Ratings indicate limits beyond which damage to the component may occur. Electrical specifications do not apply
when operating the device outside of the limits set forth under the operating ratings which specify the intended range of operating
conditions.
(2) The maximum power dissipation which can be safely dissipated for any application is a function of the maximum junction temperature,
TJ(MAX) = 125°C, the junction-to-ambient thermal resistance for the SOT-23 package, θJ-A = 265°C/W, and the ambient temperature,
TA. The maximum allowable power dissipation at any ambient temperature for designs using this device can be calculated using the
formula:
. If power dissipation exceeds the maximum specified above, the internal thermal protection
circuitry will protect the device by reducing the output voltage as required to maintain a safe junction temperature.
(3) The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin.
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Electrical Characteristics
Limits in standard typeface are for TJ = 25°C, and limits in boldface type apply over the full operating temperature range
(−40°C ≤ TJ ≤ +125°C). Unless otherwise specified: VIN = 5V, VSHDN = 5V, IL = 0A.
Parameter
Test Conditions
Min(1)
2.7
5.4
8
Typ(2)
Max(1)
Units
VIN
Input Voltage
14
V
V
VOUT (MIN)
Minimum Output Voltage
Under Load
RL = 43Ω
VIN = 2.7V
VIN = 3.3V
VIN = 5V
7
10
16
7.5
11
15
5
X Option(3)
RL = 43Ω
VIN = 2.7V
VIN = 3.3V
VIN = 5V
6
Y Option(3)
8.75
RL = 15Ω
VIN = 2.7V
VIN = 3.3V
VIN = 5V
3.75
5
X Option(3)
6.5
10
5
RL = 15Ω
VIN = 2.7V
VIN = 3.3V
VIN = 5V
4
Y Option(3)
5.5
7
10
2
ISW
Switch Current Limit
See(4)
1.8
A
1.4
RDS(ON)
Switch ON Resistance
ISW = 100 mA
Vin = 5V
260
300
400
500
mΩ
ISW = 100 mA
Vin = 3.3V
450
550
SHDNTH
ISHDN
VFB
Shutdown Threshold
Device ON
1.5
V
Device OFF
0.50
Shutdown Pin Bias Current VSHDN = 0
VSHDN = 5V
0
0
µA
2
Feedback Pin Reference
Voltage
VIN = 3V
1.205
1.230
1.255
V
IFB
IQ
Feedback Pin Bias Current VFB = 1.23V
60
2
500
3.0
2
nA
Quiescent Current
VSHDN = 5V, Switching "X"
VSHDN = 5V, Switching "Y"
mA
1.0
VSHDN = 5V, Not Switching
VSHDN = 0
400
0.024
500
1
µA
ΔVFB/ΔVIN
FB Voltage Line
Regulation
2.7V ≤ VIN ≤ 14V
0.02
%/V
MHz
FSW
Switching Frequency(5)
Maximum Duty Cycle(5)
Switch Leakage
“X” Option
1
0.40
78
1.6
0.60
86
1.85
0.8
“Y” Option
DMAX
“X” Option
%
“Y” Option
88
93
IL
Not Switching VSW = 5V
1
µA
(1) Limits are guaranteed by testing, statistical correlation, or design.
(2) Typical values are derived from the mean value of a large quantity of samples tested during characterization and represent the most
likely expected value of the parameter at room temperature.
(3) L = 10 µH, COUT = 4.7 µF, duty cycle = maximum
(4) Switch current limit is dependent on duty cycle (see Typical Performance Characteristics).
(5) Guaranteed limits are the same for Vin = 3.3V input.
4
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Typical Performance Characteristics
Unless otherwise specified: VIN = 5V, SHDN pin tied to VIN.
Iq Vin (Active)
Iq Vin (Active)
vs
Temperature - "Y"
vs
Temperature - "X"
2.2
2.15
2.1
1.25
1.2
1.15
1.1
2.05
2
1.05
1
1.95
1.9
0.95
1.85
0.9
1.8
50
100
75
150
125
0
25
-50 -25
-50 -25
0
25 50 75 100 125 150
TEMPERATURE (oC)
TEMPERATURE (oC)
Figure 3.
Figure 4.
Oscillator Frequency
vs
Temperature - "X"
Oscillator Frequency
vs
Temperature - "Y"
1.58
1.56
1.54
1.52
1.5
0.6
0.58
0.56
0.54
0.52
0.5
VIN = 5V
VIN = 5V
VIN = 3.3V
VIN = 3.3V
1.48
1.46
1.44
1.42
1.4
0.48
150
75 100 125
-25
0
25
-50
50
50
100
75
150
0
25
125
-50 -25
TEMPERATURE (oC)
TEMPERATURE (oC)
Figure 5.
Figure 6.
Max. Duty Cycle
vs
Temperature - "X"
Max. Duty Cycle
vs
Temperature - "Y"
96.8
96.7
96.6
96.5
96.4
96.3
96.2
96.1
96
93
92.9
92.8
92.7
92.6
92.5
92.4
92.3
92.2
92.1
VIN = 3.3V
VIN = 5V
VIN = 5V
VIN = 3.3V
95.9
50
100
75
150
125
0
25
-50 -25
50
100
75
150
0
25
125
-50 -25
TEMPERATURE (oC)
TEMPERATURE (oC)
Figure 7.
Figure 8.
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Typical Performance Characteristics (continued)
Unless otherwise specified: VIN = 5V, SHDN pin tied to VIN.
Iq Vin (Idle)
Feedback Bias Current
vs
vs
Temperature
Temperature
380
375
370
365
360
355
350
345
340
0.09
0.08
0.07
0.06
0.05
0.04
0.03
0.02
0.01
0
50
100
75
150
125
0
25
-50 -25
50
100
75
150
125
0
25
-50 -25
TEMPERATURE (oC)
TEMPERATURE (oC)
Figure 9.
Figure 10.
Feedback Voltage
vs
Temperature
RDS(ON)
vs
Temperature
1.231
1.23
0.5
0.45
0.4
1.229
1.228
1.227
1.226
1.225
1.224
1.223
1.222
Vin = 3.3V
0.35
0.3
Vin = 5V
0.25
0.2
0.15
0.1
0.05
0
-40 -25
0
25 50 75 100 125
-40 -25
0
25 50 75 100 125
TEMPERATURE (oC)
TEMPERATURE (oC)
Figure 11.
Figure 12.
Current Limit
vs
Temperature
RDS(ON)
vs
VIN
350
300
250
200
150
100
50
2.6
2.5
2.4
2.3
2.2
2.1
2
0
-40 -25
0
25 50 75 100 125
2.5 3.5
4.5
5.5
6.5
7.5 8.5
9.5
TEMPERATURE (oC)
VIN (V)
Figure 13.
Figure 14.
6
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Typical Performance Characteristics (continued)
Unless otherwise specified: VIN = 5V, SHDN pin tied to VIN.
Efficiency
Efficiency
vs
vs
Load Current - "X"
VIN = 2.7V, VOUT = 5V
100
Load Current - "X"
VIN = 3.3V, VOUT = 5V
100
90
80
70
60
50
40
30
20
10
0
90
80
70
60
50
40
30
20
10
0
70
0
0
50
100
150
200
LOAD (mA)
250 300
0
100 200
400
300
600
500
LOAD (mA)
Figure 15.
Figure 16.
Efficiency
vs
Efficiency
vs
Load Current - "X"
Load Current - "X"
VIN = 4.2V, VOUT = 5V
VIN = 2.7V, VOUT = 12V
100
90
80
70
60
50
40
30
20
10
0
80
70
60
50
40
30
20
10
0
0
200 400
800
1000
1400
1200
600
0
10
20
30
40
50
LOAD (mA)
LOAD (mA)
Figure 17.
Figure 18.
Efficiency
vs
Efficiency
vs
Load Current - "X"
VIN = 5V, VOUT = 12V
Load Current - "X"
VIN = 3.3V, VOUT = 12V
100
90
80
70
60
50
40
30
20
10
0
80
70
60
50
40
30
20
10
0
0
100
200
400
LOAD (mA)
300
500 600
0
20 40 60 80 100 120 140 160
LOAD (mA)
Figure 19.
Figure 20.
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Typical Performance Characteristics (continued)
Unless otherwise specified: VIN = 5V, SHDN pin tied to VIN.
Efficiency
Efficiency
vs
Load Current - "Y"
VIN = 2.7V, VOUT = 5V
vs
Load Current - "X"
VIN = 5V, VOUT = 18V
100
90
100
90
80
70
60
50
40
30
20
10
0
80
70
60
50
40
30
20
10
0
0
50
100
200
350
150
300
100
350 400
250 300
250
0
50
150
200
LOAD (mA)
LOAD (mA)
Figure 21.
Figure 22.
Efficiency
vs
Efficiency
vs
Load Current - "Y"
Load Current - "Y"
VIN = 3.3V, VOUT = 5V
VIN = 4.2V, VOUT = 5V
100
100
90
80
70
60
50
40
30
20
10
0
90
80
70
60
50
40
30
20
10
0
0
100
200 300 400
500 600 700
800
1200
1000
600 800
1400
0
200 400
LOAD (mA)
LOAD (mA)
Figure 23.
Figure 24.
Efficiency
vs
Efficiency
vs
Load Current - "Y"
Load Current - "Y"
VIN = 2.7V, VOUT = 12V
VIN = 3.3V, VOUT = 12V
100
90
80
70
60
50
40
30
20
10
0
100
90
80
70
60
50
40
30
20
10
0
0
20
40
60
80
0
100
50
150
200
250
LOAD (mA)
LOAD (mA)
Figure 25.
Figure 26.
8
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Typical Performance Characteristics (continued)
Unless otherwise specified: VIN = 5V, SHDN pin tied to VIN.
Efficiency
vs
Load Current - "Y"
VIN = 5V, VOUT = 12V
100
90
80
70
60
50
40
30
20
10
0
0
100
200
400
300
500 600
LOAD (mA)
Figure 27.
BLOCK DIAGRAM
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THEORY OF OPERATION
The LM2731 is a switching converter IC that operates at a fixed frequency (0.6 or 1.6 MHz) for fast transient
response over a wide input voltage range and incorporates pulse-by-pulse current limiting protection. Because
this is current mode control, a 33 mΩ sense resistor in series with the switch FET is used to provide a voltage
(which is proportional to the FET current) to both the input of the pulse width modulation (PWM) comparator and
the current limit amplifier.
At the beginning of each cycle, the S-R latch turns on the FET. As the current through the FET increases, a
voltage (proportional to this current) is summed with the ramp coming from the ramp generator and then fed into
the input of the PWM comparator. When this voltage exceeds the voltage on the other input (coming from the
Gm amplifier), the latch resets and turns the FET off. Since the signal coming from the Gm amplifier is derived
from the feedback (which samples the voltage at the output), the action of the PWM comparator constantly sets
the correct peak current through the FET to keep the output voltage in regulation.
Q1 and Q2 along with R3 - R6 form a bandgap voltage reference used by the IC to hold the output in regulation.
The currents flowing through Q1 and Q2 will be equal, and the feedback loop will adjust the regulated output to
maintain this. Because of this, the regulated output is always maintained at a voltage level equal to the voltage at
the FB node "multiplied up" by the ratio of the output resistive divider.
The current limit comparator feeds directly into the flip-flop that drives the switch FET. If the FET current reaches
the limit threshold, the FET is turned off and the cycle terminated until the next clock pulse. The current limit
input terminates the pulse regardless of the status of the output of the PWM comparator.
Application Hints
SELECTING THE EXTERNAL CAPACITORS
The best capacitors for use with the LM2731 are multi-layer ceramic capacitors. They have the lowest ESR
(equivalent series resistance) and highest resonance frequency which makes them optimum for use with high
frequency switching converters.
When selecting a ceramic capacitor, only X5R and X7R dielectric types should be used. Other types such as
Z5U and Y5F have such severe loss of capacitance due to effects of temperature variation and applied voltage,
they may provide as little as 20% of rated capacitance in many typical applications. Always consult capacitor
manufacturer’s data curves before selecting a capacitor. High-quality ceramic capacitors can be obtained from
Taiyo-Yuden, AVX, and Murata.
SELECTING THE OUTPUT CAPACITOR
A single ceramic capacitor of value 4.7 µF to 10 µF will provide sufficient output capacitance for most
applications. If larger amounts of capacitance are desired for improved line support and transient response,
tantalum capacitors can be used. Aluminum electrolytics with ultra low ESR such as Sanyo Oscon can be used,
but are usually prohibitively expensive. Typical AI electrolytic capacitors are not suitable for switching frequencies
above 500 kHz due to significant ringing and temperature rise due to self-heating from ripple current. An output
capacitor with excessive ESR can also reduce phase margin and cause instability.
In general, if electrolytics are used, it is recommended that they be paralleled with ceramic capacitors to reduce
ringing, switching losses, and output voltage ripple.
SELECTING THE INPUT CAPACITOR
An input capacitor is required to serve as an energy reservoir for the current which must flow into the coil each
time the switch turns ON. This capacitor must have extremely low ESR, so ceramic is the best choice. We
recommend a nominal value of 2.2 µF, but larger values can be used. Since this capacitor reduces the amount of
voltage ripple seen at the input pin, it also reduces the amount of EMI passed back along that line to other
circuitry.
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FEED-FORWARD COMPENSATION
Although internally compensated, the feed-forward capacitor Cf is required for stability (see Figure 29). Adding
this capacitor puts a zero in the loop response of the converter. The recommended frequency for the zero fz
should be approximately 6 kHz. Cf can be calculated using the formula:
Cf = 1 / (2 X π X R1 X fz)
(1)
SELECTING DIODES
The external diode used in the typical application should be a Schottky diode. A 20V diode such as the
MBR0520 is recommended.
The MBR05XX series of diodes are designed to handle a maximum average current of 0.5A. For applications
exceeding 0.5A average but less than 1A, a Microsemi UPS5817 can be used.
LAYOUT HINTS
High frequency switching regulators require very careful layout of components in order to get stable operation
and low noise. All components must be as close as possible to the LM2731 device. It is recommended that a 4-
layer PCB be used so that internal ground planes are available.
As an example, a recommended layout of components is shown:
Figure 28. Recommended PCB Component Layout
Some additional guidelines to be observed:
1. Keep the path between L1, D1, and C2 extremely short. Parasitic trace inductance in series with D1 and C2
will increase noise and ringing.
2. The feedback components R1, R2 and CF must be kept close to the FB pin of U1 to prevent noise injection
on the FB pin trace.
3. If internal ground planes are available (recommended) use vias to connect directly to ground at pin 2 of U1,
as well as the negative sides of capacitors C1 and C2.
SETTING THE OUTPUT VOLTAGE
The output voltage is set using the external resistors R1 and R2 (see Figure 29). A minimum value of 13.3 kΩ is
recommended for R2 to establish a divider current of approximately 92 µA. R1 is calculated using the formula:
R1 = R2 X (VOUT/1.23 − 1)
(2)
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SWITCHING FREQUENCY
The LM2731 is provided with two switching frequencies: the “X” version is typically 1.6 MHz, while the “Y” version
is typically 600 kHz. The best frequency for a specific application must be determined based on the trade-offs
involved:
Higher switching frequency means the inductors and capacitors can be made smaller and cheaper for a given
output voltage and current. The down side is that efficiency is slightly lower because the fixed switching losses
occur more frequently and become a larger percentage of total power loss. EMI is typically worse at higher
switching frequencies because more EMI energy will be seen in the higher frequency spectrum where most
circuits are more sensitive to such interference.
Figure 29. Basic Application Circuit
DUTY CYCLE
The maximum duty cycle of the switching regulator determines the maximum boost ratio of output-to-input
voltage that the converter can attain in continuous mode of operation. The duty cycle for a given boost
application is defined as:
VOUT + VDIODE - VIN
Duty Cycle =
VOUT + VDIODE - VSW
(3)
This applies for continuous mode operation.
INDUCTANCE VALUE
The first question we are usually asked is: “How small can I make the inductor?” (because they are the largest
sized component and usually the most costly). The answer is not simple and involves trade-offs in performance.
Larger inductors mean less inductor ripple current, which typically means less output voltage ripple (for a given
size of output capacitor). Larger inductors also mean more load power can be delivered because the energy
stored during each switching cycle is:
E = L/2 X (lp)2
(4)
Where “lp” is the peak inductor current. An important point to observe is that the LM2731 will limit its switch
current based on peak current. This means that since lp(max) is fixed, increasing L will increase the maximum
amount of power available to the load. Conversely, using too little inductance may limit the amount of load
current which can be drawn from the output.
Best performance is usually obtained when the converter is operated in “continuous” mode at the load current
range of interest, typically giving better load regulation and less output ripple. Continuous operation is defined as
not allowing the inductor current to drop to zero during the cycle. It should be noted that all boost converters shift
over to discontinuous operation as the output load is reduced far enough, but a larger inductor stays “continuous”
over a wider load current range.
To better understand these trade-offs, a typical application circuit (5V to 12V boost with a 10 µH inductor) will be
analyzed. We will assume:
VIN = 5V, VOUT = 12V, VDIODE = 0.5V, VSW = 0.5V
(5)
Since the frequency is 1.6 MHz (nominal), the period is approximately 0.625 µs. The duty cycle will be 62.5%,
which means the ON time of the switch is 0.390 µs. It should be noted that when the switch is ON, the voltage
across the inductor is approximately 4.5V.
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Using the equation:
V = L (di/dt)
(6)
We can then calculate the di/dt rate of the inductor which is found to be 0.45 A/µs during the ON time. Using
these facts, we can then show what the inductor current will look like during operation:
Figure 30. 10 µH Inductor Current, 5V–12V Boost (LM2731X)
During the 0.390 µs ON time, the inductor current ramps up 0.176A and ramps down an equal amount during the
OFF time. This is defined as the inductor “ripple current”. It can also be seen that if the load current drops to
about 33 mA, the inductor current will begin touching the zero axis which means it will be in discontinuous mode.
A similar analysis can be performed on any boost converter, to make sure the ripple current is reasonable and
continuous operation will be maintained at the typical load current values.
MAXIMUM SWITCH CURRENT
The maximum FET switch current available before the current limiter cuts in is dependent on duty cycle of the
application. This is illustrated in the graphs below which show typical values of switch current for both the "X" and
"Y" versions as a function of effective (actual) duty cycle:
3000
2500
VIN = 5V
2000
VIN = 3.3V
1500
1000
VIN = 2.7V
500
VIN = 3V
0
20 30 40 50 60 70 80 90 100
DUTY CYCLE (%) = [1 - EFF*(VIN / VOUT)]
Figure 31. Switch Current Limit vs Duty Cycle - "X"
3000
2500
VIN = 5V
VIN = 3.3V
2000
1500
1000
500
0
VIN = 3V
VIN = 2.7V
20 30 40 50 60 70 80 90 100
DUTY CYCLE (%) = [1 - EFF*(VIN / VOUT)]
Figure 32. Switch Current Limit vs Duty Cycle - "Y"
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CALCULATING LOAD CURRENT
As shown in the figure which depicts inductor current, the load current is related to the average inductor current
by the relation:
ILOAD = IIND(AVG) x (1 - DC)
(7)
(8)
(9)
Where "DC" is the duty cycle of the application. The switch current can be found by:
ISW = IIND(AVG) + ½ (IRIPPLE
)
Inductor ripple current is dependent on inductance, duty cycle, input voltage and frequency:
IRIPPLE = DC x (VIN-VSW) / (f x L)
combining all terms, we can develop an expression which allows the maximum available load current to be
calculated:
ILOAD(max) = (1 - DC) x (ISW(max) - DC (VIN - VSW))
2fL
(10)
The equation shown to calculate maximum load current takes into account the losses in the inductor or turn-OFF
switching losses of the FET and diode. For actual load current in typical applications, we took bench data for
various input and output voltages for both the "X" and "Y" versions of the LM2731 and displayed the maximum
load current available for a typical device in graph form:
1200
1000
800
VOUT = 5V
600
VOUT = 8V
400
VOUT = 10V
VOUT = 12V
200
VOUT = 18V
0
2
3
4
5
6
7
8
9
10 11
VIN (V)
Figure 33. Max. Load Current (typ) vs VIN - "X"
1200
1000
800
VOUT = 5V
600
VOUT = 8V
400
VOUT = 10V
VOUT = 12V
200
0
2
3
4
5
6
7
8
VIN (V)
Figure 34. Max. Load Current (typ) vs VIN - "Y"
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DESIGN PARAMETERS VSW AND ISW
The value of the FET "ON" voltage (referred to as VSW in the equations) is dependent on load current. A good
approximation can be obtained by multiplying the "ON Resistance" of the FET times the average inductor
current.
FET on resistance increases at VIN values below 5V, since the internal N-FET has less gate voltage in this input
voltage range (see Typical Performance Characteristics curves). Above VIN = 5V, the FET gate voltage is
internally clamped to 5V.
The maximum peak switch current the device can deliver is dependent on duty cycle. For higher duty cycles, see
Typical Performance Characteristics curves.
THERMAL CONSIDERATIONS
At higher duty cycles, the increased ON time of the FET means the maximum output current will be determined
by power dissipation within the LM2731 FET switch. The switch power dissipation from ON-state conduction is
calculated by:
P(SW) = DC x IIND(AVE)2 x RDS(ON)
(11)
There will be some switching losses as well, so some derating needs to be applied when calculating IC power
dissipation.
INDUCTOR SUPPLIERS
Recommended suppliers of inductors for this product include, but are not limited to Sumida, Coilcraft, Panasonic,
TDK and Murata. When selecting an inductor, make certain that the continuous current rating is high enough to
avoid saturation at peak currents. A suitable core type must be used to minimize core (switching) losses, and
wire power losses must be considered when selecting the current rating.
SHUTDOWN PIN OPERATION
The device is turned off by pulling the shutdown pin low. If this function is not going to be used, the pin should be
tied directly to VIN. If the SHDN function will be needed, a pull-up resistor must be used to VIN (approximately
50k-100kΩ recommended). The SHDN pin must not be left unterminated.
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