LM2735YQMF/NOPB_技术文档

LM2735

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LM2735/LM2735-Q1 520kHz/1.6MHz – Space-Efficient Boost and SEPIC DC-DC Regulator

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1

FEATURES

DESCRIPTION

The LM2735 is an easy-to-use, space-efficient 2.1A

low-side switch regulator ideal for Boost and SEPIC

DC-DC regulation. It provides all the active functions

to provide local DC/DC conversion with fast-transient

response and accurate regulation in the smallest PCB

area. Switching frequency is internally set to either

520kHz or 1.6MHz, allowing the use of extremely

small surface mount inductor and chip capacitors

while providing efficiencies up to 90%. Current-mode

control and internal compensation provide ease-of-

use, minimal component count, and high-

2

•

Input Voltage Range 2.7V to 5.5V

Output Voltage Range 3V to 24V

•

2.1A Switch Current over Full Temperature

Range

•

Current-Mode Control

Logic High Enable Pin

Ultra Low Standby Current of 80 nA in

Shutdown

•

170 mΩ NMOS Switch

performance regulation over

a wide range of

±2% Feedback Voltage Accuracy

Ease-of-Use, Small Total Solution Size

operating conditions. External shutdown features an

ultra-low standby current of 80 nA ideal for portable

applications. Tiny SOT-23, WSON, and MSOP-

–

Internal Soft-Start

PowerPAD

packages

provide

space-savings.

Internal Compensation

Two Switching Frequencies

520 kHz (LM2735-Y)

1.6 MHz (LM2735-X)

Additional features include internal soft-start, circuitry

to reduce inrush current, pulse-by-pulse current limit,

and thermal shutdown.

Uses Small Surface Mount Inductors and

Chip Capacitors

–

Tiny SOT-23, WSON, and MSOP-PowerPAD

Packages

•

LM2735-Q1 is AEC-Q100 Grade 1 Qualified and

is Manufactured on an Automotive Grade Flow

APPLICATIONS

•

LCD Display Backlighting For Portable

Applications

•

OLED Panel Power Supply

USB Powered Devices

Digital Still and Video Cameras

White LED Current Source

Automotive

1

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of

Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

2

All trademarks are the property of their respective owners.

PRODUCTION DATA information is current as of publication date.

Products conform to specifications per the terms of the Texas

Instruments standard warranty. Production processing does not

necessarily include testing of all parameters.

LM2735

SNVS485F –JUNE 2007–REVISED APRIL 2013

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Typical Boost Application Circuit

V

IN

2.7V-5.5V

12V

L

D

1

R

2

C

3

R

3

4

5

3

2

1

C

2

R

1

C

1

GND

Figure 1.

Figure 2. Efficiency vs Load Current V_O= 12V

Connection Diagrams

1

SW

GND

FB

5

VIN

1

2

6

5

PGND

VIN

SW

2

3

4

AGND

EN

3

4

FB

Figure 3. 5-Pin SOT-23 (Top View)

See Package Number DBV

Figure 4. 6-Pin WSON (Top View)

See Package Number NGG

NC

PGND

VIN

NC

1

2

3

4

8

7

6

5

SW

AGND

FB

EN

Figure 5. 8-Pin MSOP-PowerPAD (Top View)

See Package Number DGN

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PIN DESCRIPTIONS - 5-PIN SOT-23

1

Name

SW

Function

Output switch. Connect to the inductor, output diode.

2

GND

FB

Signal and power ground pin. Place the bottom resistor of the feedback network as close as possible to this pin.

Feedback pin. Connect FB to external resistor divider to set output voltage.

3

4

EN

Shutdown control input. Logic high enables operation. Do not allow this pin to float or be greater than VIN + 0.3V.

Supply voltage for power stage, and input supply voltage.

5

VIN

PIN DESCRIPTIONS - 6-PIN WSON

Name

PGND

VIN

Function

1

2

Power ground pin. Place PGND and output capacitor GND close together.

Supply voltage for power stage, and input supply voltage.

3

EN

Shutdown control input. Logic high enables operation. Do not allow this pin to float or be greater than VIN + 0.3V.

Feedback pin. Connect FB to external resistor divider to set output voltage.

Signal ground pin. Place the bottom resistor of the feedback network as close as possible to this pin & pin 4.

Output switch. Connect to the inductor, output diode.

4

FB

5

AGND

SW

6

DAP

GND

Signal & Power ground. Connect to pin 1 & pin 5 on top layer. Place 4-6 vias from DAP to bottom layer GND plane.

PIN DESCRIPTIONS - 8-PIN MSOP-PowerPAD

Name

Function

1

2

No Connect

PGND

VIN

Power ground pin. Place PGND and output capacitor GND close together.

Supply voltage for power stage, and input supply voltage.

Shutdown control input. Logic high enables operation. Do not allow this pin to float or be greater than VIN + 0.3V.

Feedback pin. Connect FB to external resistor divider to set output voltage.

Signal ground pin. Place the bottom resistor of the feedback network as close as possible to this pin & pin 5

Output switch. Connect to the inductor, output diode.

3

4

EN

5

FB

6

AGND

SW

7

8

No Connect

DAP

GND

Signal & Power ground. Connect to pin 2 & pin 6 on top layer. Place 4-6 vias from DAP to bottom layer GND plane.

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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam

during storage or handling to prevent electrostatic damage to the MOS gates.

Absolute Maximum Ratings⁽¹⁾⁽²⁾

V_IN

-0.5V to 7V

-0.5V to 26.5V

-0.5V to 3.0V

-0.5V to 7.0V

2kV

SW Voltage

FB Voltage

EN Voltage

ESD Susceptibility⁽³⁾

Junction Temperature⁽⁴⁾

Storage Temp. Range

Soldering Information

150°C

-65°C to 150°C

220°C

Infrared/Convection Reflow (15sec)

(1) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for

which the device is intended to be functional, but specific performance is not ensured. For ensured specifications and the test

conditions, see Electrical Characteristics.

(2) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and

specifications.

(3) The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin.

(4) Thermal shutdown will occur if the junction temperature exceeds the maximum junction temperature of the device

Operating Ratings⁽¹⁾

V_IN

2.7V to 5.5V

3V to 24V

V_SW

(2)

V_EN

0V to V_IN

Junction Temperature Range

Power Dissipation

−40°C to +125°C

400 mW

(Internal) SOT-23

(1) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for

which the device is intended to be functional, but specific performance is not ensured. For ensured specifications and the test

conditions, see Electrical Characteristics.

(2) Do not allow this pin to float or be greater than V_IN+0.3V.

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Electrical Characteristics

Limits in standard type are for T_J= 25°C only; limits in boldface type apply over the junction temperature range of (T_J= -

40°C to 125°C). Minimum and Maximum limits are specified through test, design, or statistical correlation. Typical values

represent the most likely parametric norm at T_J= 25°C, and are provided for reference purposes only. V_IN= 5V unless

otherwise indicated under the Conditions column.

Symbol

Parameter

Conditions

Min

Typ

Max

Units

−40°C ≤ to T_J≤ +125°C (SOT-23)

0°C ≤ to T_J≤ +125°C (SOT-23)

−40°C ≤ to T_J≤ +125°C (WSON)

−0°C ≤ to T_J≤ +125°C (WSON)

1.230 1.255 1.280

1.236 1.255 1.274

1.225 1.255 1.285

1.229 1.255 1.281

1.220 1.255 1.290

V_FB

Feedback Voltage

V

−40°C ≤ to T_J≤ +125°C

(MSOP-PowerPAD)

0°C ≤ to T_J≤ +125°C (MSOP-PowerPAD) 1.230 1.255 1.280

ΔV_FB/V_IN

Feedback Voltage Line Regulation

Feedback Input Bias Current

V_IN= 2.7V to 5.5V

0.06

0.1

1600

520

96

%/V

µA

I_FB

1

LM2735-X

1200

360

88

2000

680

kHz

F_SW

Switching Frequency

Maximum Duty Cycle

Minimum Duty Cycle

Switch On Resistance

LM2735-Y

LM2735-X

D_MAX

%

LM2735-Y

91

99

LM2735-X

5

D_MIN

LM2735-Y

2

SOT-23 and MSOP-PowerPAD

WSON

170

190

3

330

350

R_DS(ON)

mΩ

I_CL

SS

Switch Current Limit

Soft Start

2.1

A

4

ms

mA

LM2735-X

7.0

3.4

80

11

7

Quiescent Current (switching)

I_Q

LM2735-Y

Quiescent Current (shutdown)

Undervoltage Lockout

All Options V_EN= 0V

nA

V

VIN Rising

2.3

1.9

2.65

0.4

UVLO

VIN Falling

See⁽¹⁾

1.7

1.8

Shutdown Threshold Voltage

Enable Threshold Voltage

Switch Leakage

V_{EN_TH}

V

I_-SW

I_-EN

V_SW= 24V

1.0

100

80

µA

nA

Enable Pin Current

Sink/Source

WSON and MSOP-PowerPAD Package

SOT-23 Package

Junction to Ambient

0 LFPM Air Flow⁽²⁾

θ_JA

°C/W

118

18

WSON and MSOP-PowerPAD Package

SOT-23 Package

θ_JC

Junction to Case⁽²⁾

°C/W

°C

60

T_SD

Thermal Shutdown Temperature⁽³⁾

Thermal Shutdown Hysteresis

160

10

(1) Do not allow this pin to float or be greater than V_IN+0.3V.

(2) Applies for packages soldered directly onto a 3” x 3” PC board with 2oz. copper on 4 layers in still air.

(3) Thermal shutdown will occur if the junction temperature exceeds the maximum junction temperature of the device

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Typical Performance Characteristics

Current Limit vs Temperature

FB Pin Voltage vs Temperature

Figure 6.

Figure 7.

Oscillator Frequency vs Temperature - "X"

Oscillator Frequency vs Temperature - "Y"

Figure 8.

Figure 9.

Typical Maximum Output Current vs V_IN

R_DSONvs Temperature

Figure 10.

Figure 11.

6

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Typical Performance Characteristics (continued)

LM2735X Efficiency vs Load Current, Vo = 20V

LM2735Y Efficiency vs Load Current, Vo = 20V

Figure 12.

Figure 13.

LM2735X Efficiency vs Load Current, Vo = 12V

LM2735Y Efficiency vs Load Current, Vo = 12V

Figure 14.

Figure 15.

Output Voltage Load Regulation

Output Voltage Line Regulation

Figure 16.

Figure 17.

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Simplified Internal Block Diagram

EN

V

IN

Thermal

SHDN

Control Logic

-

UVLO = 2.3V

+

Oscillator

Corrective - Ramp

SW

S

R

Q

cv

1.6MHz

+

-

R

FB

-

+

V

= 1.255V

NMOS

REF

Internal

Compensation

I

LIMIT

I

SENSE-AMP

Soft-Start

+

-

GND

Figure 18. Simplified Block Diagram

APPLICATION INFORMATION

THEORY OF OPERATION

The LM2735 is a constant frequency PWM boost regulator IC that delivers a minimum of 2.1A peak switch

current. The regulator has a preset switching frequency of either 520 kHz or 1.60 MHz. This high frequency

allows the LM2735 to operate with small surface mount capacitors and inductors, resulting in a DC/DC converter

that requires a minimum amount of board space. The LM2735 is internally compensated, so it is simple to use,

and requires few external components. The LM2735 uses current-mode control to regulate the output voltage.

The following operating description of the LM2735 will refer to the Simplified Internal Block Diagram (Figure 18)

the simplified schematic (Figure 19), and its associated waveforms (Figure 20). The LM2735 supplies a regulated

output voltage by switching the internal NMOS control switch at constant frequency and variable duty cycle. A

switching cycle begins at the falling edge of the reset pulse generated by the internal oscillator. When this pulse

goes low, the output control logic turns on the internal NMOS control switch. During this on-time, the SW pin

voltage (V_SW) decreases to approximately GND, and the inductor current (I_L) increases with a linear slope. I_Lis

measured by the current sense amplifier, which generates an output proportional to the switch current. The

sensed signal is summed with the regulator’s corrective ramp and compared to the error amplifier’s output, which

is proportional to the difference between the feedback voltage and V_REF. When the PWM comparator output goes

high, the output switch turns off until the next switching cycle begins. During the switch off-time, inductor current

discharges through diode D1, which forces the SW pin to swing to the output voltage plus the forward voltage

(V_D) of the diode. The regulator loop adjusts the duty cycle (D) to maintain a constant output voltage .

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V

( )

L t

-

+

D

1

I

( )

L t

L

1

I

( )

C t

Q

1

V

IN

V

Control

( )

t

SW

V

( )

O t

C

1

-

Figure 19. Simplified Schematic

V

+ V

D

O

V

( )

t

sw

t

V

IN

V

( )

L t

t

V

-V

OUT D

IN

I

i

( )

t

L

t

I

( )

t

DIODE

t

i

- i

-

OUT

(

L

)

I

( )

t

Capacitor

t

- i

OUT

Dv

V

( )

t

OUT

DT

T

S

Figure 20. Typical Waveforms

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CURRENT LIMIT

The LM2735 uses cycle-by-cycle current limiting to protect the internal NMOS switch. It is important to note that

this current limit will not protect the output from excessive current during an output short circuit. The input supply

is connected to the output by the series connection of an inductor and a diode. If a short circuit is placed on the

output, excessive current can damage both the inductor and diode.

Design Guide

ENABLE PIN / SHUTDOWN MODE

The LM2735 has a shutdown mode that is controlled by the Enable pin (EN). When a logic low voltage is applied

to EN, the part is in shutdown mode and its quiescent current drops to typically 80 nA. Switch leakage adds up to

another 1 µA from the input supply. The voltage at this pin should never exceed V_IN+ 0.3V.

THERMAL SHUTDOWN

Thermal shutdown limits total power dissipation by turning off the output switch when the IC junction temperature

exceeds 160°C. After thermal shutdown occurs, the output switch doesn’t turn on until the junction temperature

drops to approximately 150°C.

SOFT-START

This function forces V_OUTto increase at a controlled rate during start up. During soft-start, the error amplifier’s

reference voltage ramps to its nominal value of 1.255V in approximately 4.0ms. This forces the regulator output

to ramp up in a more linear and controlled fashion, which helps reduce inrush current.

INDUCTOR SELECTION

The Duty Cycle (D) can be approximated quickly using the ratio of output voltage (V_O) to input voltage (V_IN):

V_OUT

V_IN

1

≈

’

÷

=

∆

Å

-

1 D

D

«

◊

(1)

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Therefore:

D =

SNVS485F –JUNE 2007–REVISED APRIL 2013

V_OUT- V

IN

V_OUT

(2)

Power losses due to the diode (D1) forward voltage drop, the voltage drop across the internal NMOS switch, the

voltage drop across the inductor resistance (R_DCR) and switching losses must be included to calculate a more

accurate duty cycle (See Calculating Efficiency, and Junction Temperature for a detailed explanation). A more

accurate formula for calculating the conversion ratio is:

V_OUT

V_IN

h

D

=

Å

where

•

Where η equals the efficiency of the LM2735 application.

(3)

The inductor value determines the input ripple current. Lower inductor values decrease the size of the inductor,

but increase the input ripple current. An increase in the inductor value will decrease the input ripple current.

Di

I

( )

L t

L

i

L

V

-V

OUT

IN

V

IN

L

DT

T

S

t

S

Figure 21. Inductor Current

2DiL

DT_S

V

≈

’

IN

= ∆

÷

∆

L

«

◊

V

≈

’

∆

^IN÷ x DT_S

Âi =

÷

L

2L

«

◊

(4)

A good design practice is to design the inductor to produce 10% to 30% ripple of maximum load. From the

previous equations, the inductor value is then obtained.

V_IN

2 x Di_L

≈

’

x DT

÷

∆

L =

S

«

◊

where

•

1/T_S= F_SW= switching frequency

(5)

One must also ensure that the minimum current limit (2.1A) is not exceeded, so the peak current in the inductor

must be calculated. The peak current (I_LPK) in the inductor is calculated by:

IL_pk= I_IN+ ΔI_L

(6)

or

IL_pk= I_OUT/ D' + ΔI_L

(7)

When selecting an inductor, make sure that it is capable of supporting the peak input current without saturating.

Inductor saturation will result in a sudden reduction in inductance and prevent the regulator from operating

correctly. Because of the speed of the internal current limit, the peak current of the inductor need only be

specified for the required maximum input current. For example, if the designed maximum input current is 1.5A

and the peak current is 1.75A, then the inductor should be specified with a saturation current limit of >1.75A.

There is no need to specify the saturation or peak current of the inductor at the 3A typical switch current limit.

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Because of the operating frequency of the LM2735, ferrite based inductors are preferred to minimize core losses.

This presents little restriction since the variety of ferrite-based inductors is huge. Lastly, inductors with lower

series resistance (DCR) will provide better operating efficiency. For recommended inductors see Example

Circuits.

INPUT CAPACITOR

An input capacitor is necessary to ensure that V_INdoes not drop excessively during switching transients. The

primary specifications of the input capacitor are capacitance, voltage, RMS current rating, and ESL (Equivalent

Series Inductance). The recommended input capacitance is 10 µF to 44 µF depending on the application. The

capacitor manufacturer specifically states the input voltage rating. Make sure to check any recommended

deratings and also verify if there is any significant change in capacitance at the operating input voltage and the

operating temperature. The ESL of an input capacitor is usually determined by the effective cross sectional area

of the current path. At the operating frequencies of the LM2735, certain capacitors may have an ESL so large

that the resulting impedance (2πfL) will be higher than that required to provide stable operation. As a result,

surface mount capacitors are strongly recommended. Multilayer ceramic capacitors (MLCC) are good choices for

both input and output capacitors and have very low ESL. For MLCCs it is recommended to use X7R or X5R

dielectrics. Consult capacitor manufacturer datasheet to see how rated capacitance varies over operating

conditions.

OUTPUT CAPACITOR

The LM2735 operates at frequencies allowing the use of ceramic output capacitors without compromising

transient response. Ceramic capacitors allow higher inductor ripple without significantly increasing output ripple.

The output capacitor is selected based upon the desired output ripple and transient response. The initial current

of a load transient is provided mainly by the output capacitor. The output impedance will therefore determine the

maximum voltage perturbation. The output ripple of the converter is a function of the capacitor’s reactance and

its equivalent series resistance (ESR):

≈

∆

«

’

V_OUTx D

÷

◊

ÂV_OUTÂI x R

+

ESR

=

L

2x F_SWx R_Loadx C_OUT

(8)

When using MLCCs, the ESR is typically so low that the capacitive ripple may dominate. When this occurs, the

output ripple will be approximately sinusoidal and 90° phase shifted from the switching action .

Given the availability and quality of MLCCs and the expected output voltage of designs using the LM2735, there

is really no need to review any other capacitor technologies. Another benefit of ceramic capacitors is their ability

to bypass high frequency noise. A certain amount of switching edge noise will couple through parasitic

capacitances in the inductor to the output. A ceramic capacitor will bypass this noise while a tantalum will not.

Since the output capacitor is one of the two external components that control the stability of the regulator control

loop, most applications will require a minimum at 4.7 µF of output capacitance. Like the input capacitor,

recommended multilayer ceramic capacitors are X7R or X5R. Again, verify actual capacitance at the desired

operating voltage and temperature.

SETTING THE OUTPUT VOLTAGE

The output voltage is set using the following equation where R1 is connected between the FB pin and GND, and

R2 is connected between V_OUTand the FB pin.

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V_O

C₃

R₂

R₁

V_FB

R_LOAD

Figure 22. Setting Vout

A good value for R1 is 10kΩ.

≈

∆

«

’

÷

◊

V_OUT

V_REF

÷

-1 x

R₂=

R₁

(9)

COMPENSATION

The LM2735 uses constant frequency peak current mode control. This mode of control allows for a simple

external compensation scheme that can be optimized for each application. A complicated mathematical analysis

can be completed to fully explain the LM2735’s internal & external compensation, but for simplicity, a graphical

approach with simple equations will be used. Below is a Gain & Phase plot of a LM2735 that produces a 12V

output from a 5V input voltage. The Bode plot shows the total loop Gain & Phase without external compensation.

80

60

40

20

0

180

gm-Pole

RC-Pole

90

Vi = 5V

Vo = 12V

Io = 500 mA

Co = 10 mF

Lo = 5 mH

0

-20

-40

-60

-80

gm-Zero

-90

RHP-Zero

-180

1M

10

100

1k

10k

100k

FREQUENCY

Figure 23. LM2735 Without External Compensation

One can see that the Crossover frequency is fine, but the phase margin at 0dB is very low (22°). A zero can be

placed just above the crossover frequency so that the phase margin will be bumped up to a minimum of 45°.

Below is the same application with a zero added at 8 kHz.

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80

60

40

20

0

180

90

gm-Pole

RC-Pole

Vi = 5V

Vo = 12V

Io = 500 mA

Co = 10 mF

Lo = 5 mH

D = 0.625

Cf = 220 pF

Fz-cf = 8 kHz

gm-zero

0

RHP-Zero = 107 kHz

Fp-cf = 77 kHz

Fp-rc = 660 Hz

-20

-40

-60

-80

-90

-180

Ext (Cf)

-Zero

Ext (Cf)-Pole

RHP-Zero

100k 1M

10

100

1k

10k

FREQUENCY

Figure 24. LM2735 With External Compensation

The simplest method to determine the compensation component value is as follows.

Set the output voltage with the following equation.

≈

∆

«

’

÷

◊

V_OUT

V_REF

÷

-1 x

R₂=

R₁

where

•

R1 is the bottom resistor and R2 is the resistor tied to the output voltage.

(10)

The next step is to calculate the value of C3. The internal compensation has been designed so that when a zero

is added between 5 kHz & 10 kHz the converter will have good transient response with plenty of phase margin

for all input & output voltage combinations.

1

F_{ZERO- CF}

ç

5 kHz 10 kHz

=

2p R xC

(

)

f

2

(11)

Lower output voltages will have the zero set closer to 10 kHz, and higher output voltages will usually have the

zero set closer to 5 kHz. It is always recommended to obtain a Gain/Phase plot for your actual application. One

could refer to the Typical applications section to obtain examples of working applications and the associated

component values.

Pole @ origin due to internal gm amplifier:

F_P-ORIGIN

(12)

Pole due to output load and capacitor:

1

F_P-RC

=

2p(R_LoadC_OUT

)

(13)

This equation only determines the frequency of the pole for perfect current mode control (CMC). I.e, it doesn’t

take into account the additional internal artificial ramp that is added to the current signal for stability reasons. By

adding artificial ramp, you begin to move away from CMC to voltage mode control (VMC). The artifact is that the

pole due to the output load and output capacitor will actually be slightly higher in frequency than calculated. In

this example it is calculated at 650 Hz, but in reality it is around 1 kHz.

The zero created with capacitor C3 & resistor R2:

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V_O

C₃

R₂

R₁

V_FB

R_LOAD

Figure 25. Setting External Pole-Zero

1

F_{ZERO- CF}

=

2p R xC

(

)

3

2

(14)

There is an associated pole with the zero that was created in the above equation.

1

F_{POLE CF}

=

-

2p((R₁R₂) x C₃)

(15)

It is always higher in frequency than the zero.

A right-half plane zero (RHPZ) is inherent to all boost converters. One must remember that the gain associated

with a right-half plane zero increases at 20dB per decade, but the phase decreases by 45° per decade. For most

applications there is little concern with the RHPZ due to the fact that the frequency at which it shows up is well

beyond crossover, and has little to no effect on loop stability. One must be concerned with this condition for large

inductor values and high output currents.

2

'

( )

2p x L

D R_Load

RHP_ZERO

=

(16)

There are miscellaneous poles and zeros associated with parasitics internal to the LM2735, external

components, and the PCB. They are located well over the crossover frequency, and for simplicity are not

discussed.

PCB Layout Considerations

When planning layout there are a few things to consider when trying to achieve a clean, regulated output. The

most important consideration when completing a Boost Converter layout is the close coupling of the GND

connections of the C_OUTcapacitor and the LM2735 PGND pin. The GND ends should be close to one another

and be connected to the GND plane with at least two through-holes. There should be a continuous ground plane

on the bottom layer of a two-layer board except under the switching node island. The FB pin is a high impedance

node and care should be taken to make the FB trace short to avoid noise pickup and inaccurate regulation. The

feedback resistors should be placed as close as possible to the IC, with the AGND of R1 placed as close as

possible to the GND (pin 5 for the WSON) of the IC. The V_OUTtrace to R2 should be routed away from the

inductor and any other traces that are switching. High AC currents flow through the V_IN, SW and V_OUTtraces, so

they should be as short and wide as possible. However, making the traces wide increases radiated noise, so the

designer must make this trade-off. Radiated noise can be decreased by choosing a shielded inductor. The

remaining components should also be placed as close as possible to the IC. Please see Application Note AN-

1229 SNVA054 for further considerations and the LM2735 demo board as an example of a four-layer layout.

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Below is an example of a good thermal & electrical PCB design. This is very similar to our LM2735

demonstration boards that are obtainable via the Texas Instruments website. The demonstration board consists

of a two layer PCB with a common input and output voltage application. Most of the routing is on the top layer,

with the bottom layer consisting of a large ground plane. The placement of the external components satisfies the

electrical considerations, and the thermal performance has been improved by adding thermal vias and a top layer

“Dog-Bone”.

C_IN

PCB

V_IN

P_GND

FB

EN

4

3

L₁

A_GND

V_IN

C_IN

5

2

P_GND

1

C_OUT

SW

6

D₁

V_O

Figure 26. Example of Proper PCB Layout

Thermal Design

When designing for thermal performance, one must consider many variables:

Ambient Temperature: The surrounding maximum air temperature is fairly explanatory. As the temperature

increases, the junction temperature will increase. This may not be linear though. As the surrounding air

temperature increases, resistances of semiconductors, wires and traces increase. This will decrease the

efficiency of the application, and more power will be converted into heat, and will increase the silicon junction

temperatures further.

Forced Airflow: Forced air can drastically reduce the device junction temperature. Air flow reduces the hot spots

within a design. Warm airflow is often much better than a lower ambient temperature with no airflow.

External Components: Choose components that are efficient, and you can reduce the mutual heating between

devices.

PCB design with thermal performance in mind:

The PCB design is a very important step in the thermal design procedure. The LM2735 is available in three

package options (5 pin SOT-23, 8 pin MSOP-PowerPAD & 6 pin WSON). The options are electrically the same,

but difference between the packages is size and thermal performance. The WSON and MSOP-PowerPAD have

thermal Die Attach Pads (DAP) attached to the bottom of the packages, and are therefore capable of dissipating

more heat than the SOT-23 package. It is important that the customer choose the correct package for the

application. A detailed thermal design procedure has been included in this data sheet. This procedure will help

determine which package is correct, and common applications will be analyzed.

There is one significant thermal PCB layout design consideration that contradicts a proper electrical PCB layout

design consideration. This contradiction is the placement of external components that dissipate heat. The

greatest external heat contributor is the external Schottky diode. It would be nice if you were able to separate by

distance the LM2735 from the Schottky diode, and thereby reducing the mutual heating effect. This will however

create electrical performance issues. It is important to keep the LM2735, the output capacitor, and Schottky

diode physically close to each other (see Figure 26). The electrical design considerations outweigh the thermal

considerations. Other factors that influence thermal performance are thermal vias, copper weight, and number of

board layers.

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Definitions

Heat energy is transferred from regions of high temperature to regions of low temperature via three basic

mechanisms: radiation, conduction and convection.

Radiation Electromagnetic transfer of heat between masses at different temperatures.

Conduction Transfer of heat through a solid medium.

Convection Transfer of heat through the medium of a fluid; typically air.

Conduction & Convection will be the dominant heat transfer mechanism in most applications.

R_θJCThermal impedance from silicon junction to device case temperature.

R_θJAThermal impedance from silicon junction to ambient air temperature.

C_θJCThermal Delay from silicon junction to device case temperature.

C_θCAThermal Delay from device case to ambient air temperature.

R

_θJA& R_θJCThese two symbols represent thermal impedances, and most data sheets contain associated values

for these two symbols. The units of measurement are °C/Watt.

R

_θJAis the sum of smaller thermal impedances (see Figure 27). The capacitors represent delays that are present

from the time that power and its associated heat is increased or decreased from steady state in one medium until

the time that the heat increase or decrease reaches steady state on the another medium.

R

qC -A

C

qC -A

T

A

C

R

qJ-CASE

C

qJ-CASE

T

J

Internal- P

DISS

Figure 27. Simplified Thermal Impedance Model

The datasheet values for these symbols are given so that one might compare the thermal performance of one

package against another. In order to achieve a comparison between packages, all other variables must be held

constant in the comparison (PCB size, copper weight, thermal vias, power dissipation, V_IN, V_OUT, Load Current

etc). This does shed light on the package performance, but it would be a mistake to use these values to calculate

the actual junction temperature in your application.

T - T_A

J

R_qJA

=

P_Dissipation

(17)

We will talk more about calculating the variables of this equation later, and how to eventually calculate a proper

junction temperature with relative certainty. For now we need to define the process of calculating the junction

temperature and clarify some common misconceptions.

R_θJA[Variables]:

•

Input Voltage, Output Voltage, Output Current, RDSon.

Ambient temperature & air flow.

•

Internal & External components power dissipation.

Package thermal limitations.

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•

PCB variables (copper weight, thermal via’s, layers component placement).

It would be wrong to assume that the top case temperature is the proper temperature when calculating R_θJC

value. The R_θJCvalue represents the thermal impedance of all six sides of a package, not just the top side. This

document will refer to a thermal impedance called R_ψJC. R_ψJCrepresents a thermal impedance associated with

just the top case temperature. This will allow one to calculate the junction temperature with a thermal sensor

connected to the top case.

LM2735 Thermal Models

Heat is dissipated from the LM2735 and other devices. The external loss elements include the Schottky diode,

inductor, and loads. All loss elements will mutually increase the heat on the PCB, and therefore increase each

other’s temperatures.

L

1

I

D

( )

L t

1

V

( )

t

OUT

Q

1

V

IN

C

1

Figure 28. Thermal Schematic

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R

qCASE-AMB

T

CASE

C

qCASE-AMB

R

qJ-CASE

C

qJ-CASE

INTERNAL

SMALL

LARGE

P

DISS

P

DISS-TOP

T

AMBIENT

P

DISS-PCB

T

JUNCTION

C

qJ-PCB

R

qJ-PCB

DEVICE

EXTERNAL

P

DISS

R

qPCB-AMB

T

PCB

C

qPCB-AMB

PCB

Figure 29. Associated Thermal Model

Calculating Efficiency, and Junction Temperature

The complete LM2735 DC/DC converter efficiency (η) can be calculated in the following manner.

P_OUT

h =

P_IN

or

P_OUT

h =

P_OUT+ P_LOSS

(18)

Power loss (P_LOSS) is the sum of two types of losses in the converter, switching and conduction. Conduction

losses usually dominate at higher output loads, where as switching losses remain relatively fixed and dominate at

lower output loads.

Losses in the LM2735 Device: P_LOSS= P_COND+ P_SW+ P_Q

Conversion ratio of the Boost Converter with conduction loss elements inserted:

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≈

∆

’

÷

Å

≈

’

÷

◊

V_OUT

D x V_D

1

∆

÷

1

Å

D

∆

1-

=

∆

V

R_DCR+ D x R

V

(

)

IN

DSON

IN

«

∆

«

÷

◊

1+

Å²

R _OUT

D

(19)

(20)

(21)

One can see that if the loss elements are reduced to zero, the conversion ratio simplifies to:

V_OUT

1

Å

D

=

V

IN

And we know:

h

Å

D

V_OUT

=

V

IN

Therefore:

Å

≈

∆

D x V_D

’

÷

1-

V

V_OUT

∆

÷

IN

Å

h =

=

D

V

R_DCR+ D x R

(

)

IN

DSON

∆

«

÷

◊

1+

Å²

R_OUT

D

(22)

Calculations for determining the most significant power losses are discussed below. Other losses totaling less

than 2% are not discussed.

A simple efficiency calculation that takes into account the conduction losses is shown below:

Å

≈

∆

D x V_D

’

÷

1-

V

∆

÷

IN

h ö

R_DCR+ D x R

(

)

DSON

∆

«

÷

◊

1+

Å²

R_OUT

D

(23)

The diode, NMOS switch, and inductor DCR losses are included in this calculation. Setting any loss element to

zero will simplify the equation.

V_Dis the forward voltage drop across the Schottky diode. It can be obtained from the manufacturer’s Electrical

Characteristics section of the data sheet.

The conduction losses in the diode are calculated as follows:

P_DIODE= V_Dx I_O

(24)

Depending on the duty cycle, this can be the single most significant power loss in the circuit. Care should be

taken to choose a diode that has a low forward voltage drop. Another concern with diode selection is reverse

leakage current. Depending on the ambient temperature and the reverse voltage across the diode, the current

being drawn from the output to the NMOS switch during time D could be significant, this may increase losses

internal to the LM2735 and reduce the overall efficiency of the application. Refer to Schottky diode

manufacturer’s data sheets for reverse leakage specifications, and typical applications within this data sheet for

diode selections.

Another significant external power loss is the conduction loss in the input inductor. The power loss within the

inductor can be simplified to:

2

P_IND= I_INR_DCR

(25)

2

≈

’

÷

◊

I_OR_DCR

∆

«

P

=

IND

'

D

(26)

The LM2735 conduction loss is mainly associated with the internal NFET:

P_COND-NFET= I²_SW-rmsx R_DSONx D

(27)

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Di

I

sw(t)

t

Figure 30. LM2735 Switch Current

2

Di

I_IND

1

3

D x

D

Isw-rms = I_IND

1 +

I

IND

ö

P_IND= I_IN²x R_IND-DCR

(small ripple approximation)

P_COND-NFET= I_IN²x R_DSONx D

(28)

(29)

≈I_O’²

P

x R_DSONx D

=

∆

«

÷

COND- NFET

'

D

◊

(30)

The value for should be equal to the resistance at the junction temperature you wish to analyze. As an example,

at 125°C and V_IN= 5V, R_DSON= 250 mΩ (See typical graphs for value).

Switching losses are also associated with the internal NMOS switch. They occur during the switch on and off

transition periods, where voltages and currents overlap resulting in power loss.

The simplest means to determine this loss is to empirically measuring the rise and fall times (10% to 90%) of the

switch at the switch node:

P_SWR= 1/2(V_OUTx I_INx F_SWx T_RISE

)

(31)

(32)

(33)

P_SWF= 1/2(V_OUTx I_INx F_SWx T_FALL

)

P_SW= P_SWR+ P_SWF

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Table 1. Typical Switch-Node Rise and Fall Times

V_IN

3V

5V

3V

5V

V_OUT

5V

T_RISE

6nS

7nS

T_FALL

4nS

5nS

12V

18V

Quiescent Power Losses

I_Qis the quiescent operating current, and is typically around 4mA.

P_Q= I_Qx V_IN

(34)

Example Efficiency Calculation:

Table 2. Operating Conditions

V_IN

V_OUT

I_OUT

V_D

5V

12V

500mA

0.4V

F_SW

I_Q

1.60MHz

4mA

T_RISE

T_FALL

R_DSon

R_DCR

D

6nS

5nS

250mΩ

50mΩ

0.64

I_IN

1.4A

ΣP_COND+ P_SW+ P_DIODE+ P_IND+ P_Q= P_LOSS

(35)

(36)

Quiescent Power Losses

P_Q= I_Qx V_IN= 20 mW

Switching Power Losses

P_SWR= 1/2(V_OUTx I_INx F_SWx T_RISE) ≊ 6 ns ≊ 80 mW

P_SWF= 1/2(V_OUTx I_INx F_SWx T_FALL) ≊ 5 ns ≊ 70 mW

P_SW= P_SWR+ P_SWF= 150 mW

(37)

(38)

(39)

Internal NFET Power Losses

R_DSON= 250 mΩ

(40)

(41)

P_CONDUCTION= I_IN²x D x R_DSONx 305 mW

Diode Losses

V_D= 0.45V

(42)

(43)

P_DIODE= V_Dx I_IN(1-D) = 236 mW

Inductor Power Losses

R_DCR= 75 mΩ

(44)

(45)

P_IND= I_IN²x R_DCR= 145 mW

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Total Power Losses are:

Table 3. Power Loss Tabulation

V_IN

5V

V_OUT

12V

I_OUT

500mA

0.4V

POUT

6W

V_D

PDIODE

236mW

F_SW

1.6MHz

6nS

T_RISE

PSWR

PSWF

PQ

80mW

70mW

20mW

305mW

145mW

T_FALL

5nS

I_Q

4mA

R_DSon

250mΩ

75mΩ

0.623

86%

PCOND

PIND

R_DCR

D

η

PLOSS

856mW

(46)

P_INTERNAL= P_COND+ P_SW= 475 mW

R_qJA

R_YJC

Calculating

and

T - T_A

J

R_qJA

=

P_Dissipation

and

T - T_CASE

J

R_YJC

=

P_Dissipation

(47)

We now know the internal power dissipation, and we are trying to keep the junction temperature at or below

125°C. The next step is to calculate the value for R_θJAand/or R_ψJC. This is actually very simple to accomplish,

and necessary if you think you may be marginal with regards to thermals or determining what package option is

correct.

The LM2735 has a thermal shutdown comparator. When the silicon reaches a temperature of 160°C, the device

shuts down until the temperature reduces to 150°C. Knowing this, one can calculate the R_θJAor the R_ψJCof a

specific application. Because the junction to top case thermal impedance is much lower than the thermal

impedance of junction to ambient air, the error in calculating R_ψJCis lower than for R_θJA. However, you will need

to attach a small thermocouple onto the top case of the LM2735 to obtain the R_ψJCvalue.

Knowing the temperature of the silicon when the device shuts down allows us to know three of the four variables.

Once we calculate the thermal impedance, we then can work backwards with the junction temperature set to

125°C to see what maximum ambient air temperature keeps the silicon below the 125°C temperature.

Procedure:

Place your application into a thermal chamber. You will need to dissipate enough power in the device so you can

obtain a good thermal impedance value.

Raise the ambient air temperature until the device goes into thermal shutdown. Record the temperatures of the

ambient air and/or the top case temperature of the LM2735. Calculate the thermal impedances.

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Example from previous calculations:

Pdiss = 475 mW

Ta @ Shutdown = 139°C

Tc @ Shutdown = 155°C

T - T_A

T - T_Case-Top

J

:

R_YJC=

R_qJA

=

P_Dissipation

(48)

R

_θJAWSON = 55°C/W

_ψJCWSON = 21°C/W

R

WSON & MSOP-PowerPAD typical applications will produce R_θJAnumbers in the range of 50°C/W to 65°C/W,

and R_ψJCwill vary between 18°C/W and 28°C/W. These values are for PCB’s with two and four layer boards with

0.5 oz copper, and four to six thermal vias to bottom side ground plane under the DAP.

For 5-pin SOT-23 package typical applications, R_θJAnumbers will range from 80°C/W to 110°C/W, and R_ψJCwill

vary between 50°C/W and 65°C/W. These values are for PCB’s with two & four layer boards with 0.5 oz copper,

with two to four thermal vias from GND pin to bottom layer.

Here is a good rule of thumb for typical thermal impedances, and an ambient temperature maximum of 75°C: If

your design requires that you dissipate more than 400mW internal to the LM2735, or there is 750mW of total

power loss in the application, it is recommended that you use the 6 pin WSON or the 8 pin MSOP-PowerPAD

package.

NOTE

NOTE: To use these procedures it is important to dissipate an amount of power within the

device that will indicate a true thermal impedance value. If one uses a very small internal

dissipated value, one can see that the thermal impedance calculated is abnormally high,

and subject to error. The graph below shows the nonlinear relationship of internal power

dissipation vs R_θJA

.

500

450

400

350

300

250

200

150

100

50

0

0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8

P_DISS

Figure 31. R_θJAvs Internal Dissipation for the WSON

and MSOP-PowerPAD Package

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SEPIC Converter

The LM2735 can easily be converted into a SEPIC converter. A SEPIC converter has the ability to regulate an

output voltage that is either larger or smaller in magnitude than the input voltage. Other converters have this

ability as well (CUK and Buck-Boost), but usually create an output voltage that is opposite in polarity to the input

voltage. This topology is a perfect fit for Lithium Ion battery applications where the input voltage for a single cell

Li-Ion battery will vary between 3V & 4.5V and the output voltage is somewhere in between. Most of the analysis

of the LM2735 Boost Converter is applicable to the LM2735 SEPIC Converter.

SEPIC Design Guide:

SEPIC Conversion ratio without loss elements:

V_o

D

=

V_IN

D'

(49)

(50)

Therefore:

V_O

D =

V_O+ V_IN

Small ripple approximation:

In a well-designed SEPIC converter, the output voltage, and input voltage ripple, the inductor ripple and is small

in comparison to the DC magnitude. Therefore it is a safe approximation to assume a DC value for these

components. The main objective of the Steady State Analysis is to determine the steady state duty-cycle, voltage

and current stresses on all components, and proper values for all components.

In a steady-state converter, the net volt-seconds across an inductor after one cycle will equal zero. Also, the

charge into a capacitor will equal the charge out of a capacitor in one cycle.

Therefore:

≈

∆

«

’

÷

◊

D

D^'

I_{L2 =}

x I_L1

and

V_O’

D

≈ ’

≈

∆

«

I_L1

x

=

÷

∆

^'÷

R

« ^D◊

◊

(51)

(52)

Substituting I_L1into I_L2

V_O

=

I_L2

R

The average inductor current of L2 is the average output load.

V

( )

AREA

1

L t

t

(s)

AREA

2

DT

T

S

Figure 32. Inductor Volt-Sec Balance Waveform

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Applying Charge balance on C1:

'

(

)

D V

o

V_C1

=

D

(53)

Since there are no DC voltages across either inductor, and capacitor C6 is connected to Vin through L1 at one

end, or to ground through L2 on the other end, we can say that

V_C1= V_IN

(54)

Therefore:

'

(

)

D V

o

V_IN

=

D

(55)

This verifies the original conversion ratio equation.

It is important to remember that the internal switch current is equal to I_L1and I_L2. During the D interval. Design

the converter so that the minimum specified peak switch current limit (2.1A) is not exceeded.

V

IN

V_O

L

1

D

1

C

6

LM2735

L

2

1

2

3

6

5

4

R₂

C

5

C

3

C

4

R

3

R

1

C

1

C

2

Figure 33. SEPIC CONVERTER Schematic

Steady State Analysis with Loss Elements

v

( )

C1 t

+

( )

L1 t

-

+

i

C1(t)

i

L1(t)

D1(t)

R

v

L1

( )

D1 t

i

C2(t)

L2(t)

sw

-

V

IN

v

( )

L2 t

v

-

( )

O t

v

( )

C2 t

-

R

on

R

L2

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Using inductor volt-second balance & capacitor charge balance, the following equations are derived:

V

≈

∆

«

’

÷

◊

O

I_{L2 =}

R

and

V

’

÷

◊

≈

∆

«

’ ≈

◊

D

D^'

O

I_L1

x

=

∆

«

÷

R

(56)

(57)

≈

’

÷

∆

÷

V_o

1

D

≈

’

= ∆

÷

∆

'

2

≈

∆

’

÷

≈

∆

’

÷

V_IN

D

≈

V_D

’

÷

◊

R

L1

«

◊

R_L2

R

D

≈

∆

’

÷

≈

∆

’

÷

ON

∆

1+

+

∆

2

∆

«

÷

∆

«

÷

◊

∆

«

÷

◊

'

V_O

R

«

◊

«

◊

«

D

◊

Therefore:

≈

’

÷

∆

÷

1

∆

h =

2

∆

≈

’

÷

≈

∆

’

≈

V_D

V_O

’

R

R_L2

R

D

≈

∆

’

÷

D

≈

∆

’

÷

ON

L1

∆

÷

∆

÷

◊

1+

+

∆

2

∆

«

÷

∆

«

÷

◊

∆

«

÷

◊

'

R

«

◊

«

◊

«

D

◊

(58)

One can see that all variables are known except for the duty cycle (D). A quadratic equation is needed to solve

for D. A less accurate method of determining the duty cycle is to assume efficiency, and calculate the duty cycle.

V_O

≈

’

÷

D

x h

=

∆

V_IN

1 - D

«

◊

(59)

(60)

V_O

’

÷

◊

≈

∆

D

=

(V x h) +V_O

IN

«

Vin

Vo

2.7V

3.1V

Vin

Vo

3.3V

3.1V

Vin

Vo

5V

3.1V

Iin 770 mA

Iin 600 mA

Iin 375 mA

Io

h

Io

h

Io

h

500 mA

75%

500 mA

80%

500 mA

83%

Figure 34. Efficiencies for Typical SEPIC Application

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Product Folder Links: LM2735

LM2735

SNVS485F –JUNE 2007–REVISED APRIL 2013

www.ti.com

SEPIC Converter PCB Layout

The layout guidelines described for the LM2735 Boost-Converter are applicable to the SEPIC Converter. Below

is a proper PCB layout for a SEPIC Converter.

C_IN

PCB

V_IN

P_GND

FB

EN

4

3

L₁

V_IN

A_GND

C_IN

5

2

P_GND

1

C_OUT

SW

6

D₁

L₂

V_O

C₆

Figure 35. SEPIC PCB Layout

28

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LM2735

www.ti.com

SNVS485F –JUNE 2007–REVISED APRIL 2013

WSON Package

The LM2735 packaged in the 6–pin WSON:

Figure 36. Internal WSON Connection

For certain high power applications, the PCB land may be modified to a "dog bone" shape (see Figure 37).

Increasing the size of ground plane, and adding thermal vias can reduce the R_θJAfor the application.

COPPER

1

2

6

5

SW

PGND

Vin

AGND

3

4

FB

EN

COPPER

Figure 37. PCB Dog Bone Layout

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Product Folder Links: LM2735

LM2735

SNVS485F –JUNE 2007–REVISED APRIL 2013

www.ti.com

LM2735X SOT-23 Design Example 1

L

1

3

FB

4

EN

R

3

2

GND

V

12V

IN

1

SW

5

Vin

D

1

C

R

3

1

2

C

2

R

LOAD

R

1

Figure 38. LM2735X (1.6MHz): Vin = 5V, Vout = 12V @ 350mA

Part ID

Part Value

2.1A Boost Regulator

22µF, 6.3V, X5R

10µF, 25V, X5R

330pF

Manufacturer

TI

Part Number

LM2735XMF

U1

C1, Input Cap

TDK

C2012X5R0J226M

C3216X5R1E106M

C1608X5R1H331K

STPS120M

C2 Output Cap

TDK

C3 Comp Cap

TDK

D1, Catch Diode

0.4V_fSchottky 1A, 20V_R

15µH 1.5A

ST

L1

R1

R2

R3

Coilcraft

Vishay

MSS5131-153ML

CRCW06031022F

CRCW06038662F

CRCW06031003F

10.2kΩ, 1%

86.6kΩ, 1%

100kΩ, 1%

30

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Product Folder Links: LM2735

LM2735

www.ti.com

SNVS485F –JUNE 2007–REVISED APRIL 2013

LM2735Y SOT-23 Design Example 2

L

1

3

FB

4

EN

R

3

2

GND

V

12V

C

IN

1

SW

5

Vin

D

1

C

R

3

1

2

C

2

R

LOAD

R

1

Figure 39. LM2735Y (520kHz): Vin = 5V, Vout = 12V @ 350mA

Part ID

Part Value

2.1A Boost Regulator

22µF, 6.3V, X5R

10µF, 25V, X5R

330pF

Manufacturer

TI

Part Number

LM2735YMF

U1

C1, Input Cap

TDK

C2012X5R0J226M

C3216X5R1E106M

C1608X5R1H331K

STPS120M

C2 Output Cap

TDK

C3 Comp Cap

TDK

D1, Catch Diode

0.4V_fSchottky 1A, 20V_R

33µH 1.5A

ST

L1

R1

R2

R3

Coilcraft

Vishay

DS3316P-333ML

CRCW06031022F

CRCW06038662F

CRCW06031003F

10.2kΩ, 1%

86.6kΩ, 1%

100kΩ, 1%

Submit Documentation Feedback

31

Product Folder Links: LM2735

LM2735

SNVS485F –JUNE 2007–REVISED APRIL 2013

www.ti.com

LM2735X WSON Design Example 3

V

IN

L

1

D

LM2735

1

R

3

1

2

6

5

C

R

C

2

C

1

5

R

LOAD

C

3

4

3

C

4

1

Figure 40. LM2735X (1.6MHz): Vin = 3.3V, Vout = 12V @ 350mA

Part ID

U1

Part Value

2.1A Boost Regulator

22µF, 6.3V, X5R

No Load

Manufacturer

Part Number

LM2735XSD

TI

C1 Input Cap

C2 Input Cap

TDK

C2012X5R0J226M

C3 Output Cap

10µF, 25V, X5R

No Load

TDK

C3216X5R1E106M

C4 Output Cap

C5 Comp Cap

330pF

TDK

ST

C1608X5R1H331K

STPS120M

D1, Catch Diode

0.4V_fSchottky 1A, 20V_R

6.8µH 2A

L1

R1

R2

R3

Coilcraft

Vishay

DO1813H-682ML

CRCW06031022F

CRCW06038662F

CRCW06031003F

10.2kΩ, 1%

86.6kΩ, 1%

100kΩ, 1%

32

Submit Documentation Feedback

Product Folder Links: LM2735

LM2735

www.ti.com

SNVS485F –JUNE 2007–REVISED APRIL 2013

LM2735Y WSON Design Example 4

V

IN

L

1

D

LM2735

1

R

3

1

2

6

5

C

R

C

2

C

1

5

R

LOAD

C

3

4

3

C

4

1

Figure 41. LM2735Y (520kHz): Vin = 3.3V, Vout = 12V @ 350mA

Part ID

U1

Part Value

2.1A Boost Regulator

22µF, 6.3V, X5R

No Load

Manufacturer

Part Number

LM2735YSD

TI

C1 Input Cap

C2 Input Cap

TDK

C2012X5R0J226M

C3 Output Cap

10µF, 25V, X5R

No Load

TDK

C3216X5R1E106M

C4 Output Cap

C5 Comp Cap

330pF

TDK

ST

C1608X5R1H331K

STPS120M

D1, Catch Diode

0.4V_fSchottky 1A, 20V_R

15µH 2A

L1

R1

R2

R3

Coilcraft

Vishay

MSS5131-153ML

CRCW06031022F

CRCW06038662F

CRCW06031003F

10.2kΩ, 1%

86.6kΩ, 1%

100kΩ, 1%

Submit Documentation Feedback

33

Product Folder Links: LM2735

LM2735

SNVS485F –JUNE 2007–REVISED APRIL 2013

www.ti.com

LM2735Y MSOP-PowerPAD Design Example 5

V

IN

L₁

D

1

R

3

LM2735

1

2

3

4

NC

8

7

6

5

NC

SW

C

2

1

R

PGND

VIN

EN

2

1

C

5

AGND

FB

R

LOAD

C

4

3

Figure 42. LM2735Y (520kHz): Vin = 3.3V, Vout = 12V @ 350mA

Part ID

Part Value

2.1A Boost Regulator

22µF, 6.3V, X5R

No Load

Manufacturer

Part Number

LM2735YMY

U1

TI

C1 Input Cap

C2 Input Cap

TDK

C2012X5R0J226M

C3 Output Cap

C4 Output Cap

C5 Comp Cap

10µF, 25V, X5R

No Load

TDK

C3216X5R1E106M

330pF

TDK

ST

C1608X5R1H331K

STPS120M

D1, Catch Diode

0.4V_fSchottky 1A, 20V_R

15µH 1.5A

L1

R1

R2

R3

Coilcraft

Vishay

MSS5131-153ML

CRCW06031022F

CRCW06038662F

CRCW06031003F

10.2kΩ, 1%

86.6kΩ, 1%

100kΩ, 1%

34

Submit Documentation Feedback

Product Folder Links: LM2735

LM2735

www.ti.com

SNVS485F –JUNE 2007–REVISED APRIL 2013

LM2735X SOT-23 Design Example 6

L

1

3

FB

4

SHDN

R

3

2

GND

V

5V

C

IN

1

SW

5

Vin

D

1

C

R

3

1

2

C

2

R

LOAD

R

1

Figure 43. LM2735X (1.6MHz): Vin = 3V, Vout = 5V @ 500mA

Part ID

Part Value

2.1A Boost Regulator

10µF, 6.3V, X5R

1000pF

Manufacturer

TI

Part Number

LM2735XMF

U1

C1, Input Cap

TDK

C2012X5R0J106K

C1608X5R1H102K

STPS120M

C2, Output Cap

TDK

C3 Comp Cap

TDK

D1, Catch Diode

0.4V_fSchottky 1A, 20V_R

10µH 1.2A

ST

L1

R1

R2

R3

Coilcraft

Vishay

DO1608C-103ML

CRCW08051002F

CRCW08053012F

CRCW06031003F

10.0kΩ, 1%

30.1kΩ, 1%

100kΩ, 1%

Submit Documentation Feedback

35

Product Folder Links: LM2735

LM2735

SNVS485F –JUNE 2007–REVISED APRIL 2013

www.ti.com

LM2735Y SOT-23 Design Example 7

L

1

3

FB

4

SHDN

R

3

2

GND

V

5V

IN

1

SW

5

Vin

D

1

C

R

3

1

2

C

2

R

LOAD

R

1

Figure 44. LM2735Y (520kHz): Vin = 3V, Vout = 5V @ 750mA

Part ID

Part Value

2.1A Boost Regulator

22µF, 6.3V, X5R

1000pF

Manufacturer

TI

Part Number

U1

LM2735YMF

C1 Input Cap

TDK

C2012X5R0J226M

C1608X5R1H102K

STPS120M

C2 Output Cap

TDK

C3 Comp Cap

TDK

D1, Catch Diode

0.4V_fSchottky 1A, 20V_R

22µH 1.2A

ST

L1

R1

R2

R3

Coilcraft

Vishay

MSS5131-223ML

CRCW08051002F

CRCW08053012F

CRCW06031003F

10.0kΩ, 1%

30.1kΩ, 1%

100kΩ, 1%

36

Submit Documentation Feedback

Product Folder Links: LM2735

LM2735

www.ti.com

SNVS485F –JUNE 2007–REVISED APRIL 2013

LM2735X SOT-23 Design Example 8

L

1

3

FB

4

SHDN

R

3

2

GND

V

20V

C

IN

1

SW

5

Vin

D

1

C

R

3

1

2

C

2

R

LOAD

R

1

Figure 45. LM2735X (1.6MHz): Vin = 3.3V, Vout = 20V @ 100mA

Part ID

Part Value

2.1A Boost Regulator

22µF, 6.3V, X5R

4.7µF, 25V, X5R

470pF

Manufacturer

TI

Part Number

LM2735XMF

U1

C1, Input Cap

TDK

C2012X5R0J226M

C3216X5R1E475K

C1608X5R1H471K

MBR0530

C2, Output Cap

TDK

C3 Comp Cap

TDK

D1, Catch Diode

0.4V_fSchottky 500mA, 30V_R

10µH 1.2A

Vishay

Coilcraft

Vishay

L1

R1

R2

R3

DO1608C-103ML

CRCW06031002F

CRCW06031503F

CRCW06031003F

10.0kΩ, 1%

150kΩ, 1%

100kΩ, 1%

Submit Documentation Feedback

37

Product Folder Links: LM2735

LM2735

SNVS485F –JUNE 2007–REVISED APRIL 2013

www.ti.com

LM2735Y SOT-23 Design Example 9

L

1

3

FB

4

SHDN

R

3

2

GND

V

20V

IN

1

SW

5

Vin

D

1

C

R

3

1

2

C

2

R

LOAD

R

1

Figure 46. LM2735Y (520kHz): Vin = 3.3V, Vout = 20V @ 100mA

Part ID

Part Value

2.1A Boost Regulator

22µF, 6.3V, X5R

10µF, 25V, X5R

470pF

Manufacturer

TI

Part Number

LM2735YMF

U1

C1 Input Cap

TDK

C2012X5R0J226M

C3216X5R1E106M

C1608X5R1H471K

MBR0530

C2 Output Cap

TDK

C3 Comp Cap

TDK

D1, Catch Diode

0.4V_fSchottky 500mA, 30V_R

33µH 1.5A

Vishay

Coilcraft

Vishay

L1

R1

R2

R3

DS3316P-333ML

CRCW06031002F

CRCW06031503F

CRCW06031003F

10.0kΩ, 1%

150.0kΩ, 1%

100kΩ, 1%

38

Submit Documentation Feedback

Product Folder Links: LM2735

LM2735

www.ti.com

SNVS485F –JUNE 2007–REVISED APRIL 2013

LM2735X WSON Design Example 10

V

IN

L

1

D

LM2735

1

R

3

1

2

6

5

C

R

C

2

C

1

5

R

LOAD

C

3

4

3

C

4

1

Figure 47. LM2735X (1.6MHz): Vin = 3.3V, Vout = 20V @ 150mA

Part ID

U1

Part Value

2.1A Boost Regulator

22µF, 6.3V, X5R

10µF, 25V, X5R

No Load

Manufacturer

Part Number

LM2735XSD

TI

C1 Input Cap

C2 Input Cap

TDK

C2012X5R0J226M

C3216X5R1E106M

C3 Output Cap

C4 Output Cap

C5 Comp Cap

470pF

TDK

Vishay

Coilcraft

Vishay

C1608X5R1H471K

MBR0530

D1, Catch Diode

0.4V_fSchottky 500mA, 30V_R

8.2µH 2A

L1

R1

R2

R3

DO1813H-822ML

CRCW06031002F

CRCW06031503F

CRCW06031003F

10.0kΩ, 1%

150kΩ, 1%

100kΩ, 1%

Submit Documentation Feedback

39

Product Folder Links: LM2735

LM2735

SNVS485F –JUNE 2007–REVISED APRIL 2013

www.ti.com

LM2735Y WSON Design Example 11

V

IN

L

1

D

LM2735

1

R

3

1

2

6

5

C

R

C

2

C

1

5

R

LOAD

C

3

4

3

C

4

1

Figure 48. LM2735Y (520kHz): Vin = 3.3V, Vout = 20V @ 150mA

Part ID

U1

Part Value

2.1A Boost Regulator

10µF, 6.3V, X5R

10µF, 25V, X5R

No Load

Manufacturer

Part Number

LM2735YSD

TI

C1 Input Cap

C2 Input Cap

TDK

C2012X5R0J106K

C3216X5R1E106M

C3 Output Cap

C4 Output Cap

C5 Comp Cap

470pF

TDK

Vishay

Coilcraft

Vishay

C1608X5R1H471K

MBR0530

D1, Catch Diode

0.4V_fSchottky 500mA, 30V_R

22µH 1.5A

L1

R1

R2

R3

DS3316P-223ML

CRCW06031002F

CRCW06031503F

CRCW06031003F

10.0kΩ, 1%

150kΩ, 1%

100kΩ, 1%

40

Submit Documentation Feedback

Product Folder Links: LM2735

LM2735

www.ti.com

SNVS485F –JUNE 2007–REVISED APRIL 2013

LM2735X WSON SEPIC Design Example 12

V

IN

V_O

L

1

D

1

C

6

LM2735

L

2

1

2

3

6

5

4

R₂

C

5

C

3

C

4

R

3

R

1

C

1

C

2

Figure 49. LM2735X (1.6MHz): Vin = 2.7V - 5V, Vout = 3.3V @ 500mA

Part ID

Part Value

2.1A Boost Regulator

22µF, 6.3V, X5R

No Load

Manufacturer

Part Number

LM2735XSD

U1

TI

C1 Input Cap

TDK

C2012X5R0J226M

C2 Input Cap

C3 Output Cap

10µF, 25V, X5R

No Load

TDK

C3216X5R1E106M

C4 Output Cap

C5 Comp Cap

2200pF

TDK

C1608X5R1H222K

C2012X5R1C225K

STPS120M

C6

2.2µF 16V

D1, Catch Diode

0.4V_fSchottky 1A, 20V_R

6.8µH

ST

L1

L2

Coilcraft

Vishay

DO1608C-682ML

CRCW06031002F

CRCW06031652F

CRCW06031003F

6.8µH

R1

R2

R3

10.2kΩ, 1%

16.5kΩ, 1%

100kΩ, 1%

Submit Documentation Feedback

41

Product Folder Links: LM2735

LM2735

SNVS485F –JUNE 2007–REVISED APRIL 2013

www.ti.com

LM2735Y MSOP-PowerPAD SEPIC Design Example 13

V

IN

L₁

D

C

1

6

R

3

LM2735

1

2

3

4

NC

8

7

6

5

NC

SW

C

2

1

R

PGND

VIN

EN

2

1

C

5

L

2

AGND

FB

R

LOAD

C

4

3

Figure 50. LM2735Y (520kHz): Vin = 2.7V - 5V, Vout = 3.3V @ 500mA

Part ID

Part Value

2.1A Boost Regulator

22µF, 6.3V, X5R

No Load

Manufacturer

Part Number

LM2735YMY

U1

TI

C1 Input Cap

C2 Input Cap

TDK

C2012X5R0J226M

C3 Output Cap

10µF, 25V, X5R

No Load

TDK

C3216X5R1E106M

C4 Output Cap

C5 Comp Cap

2200pF

TDK

C1608X5R1H222K

C2012X5R1C225K

STPS120M

C6

2.2µF 16V

D1, Catch Diode

0.4V_fSchottky 1A, 20V_R

15µH 1.5A

ST

L1

L2

Coilcraft

Vishay

MSS5131-153ML

CRCW06031002F

CRCW06031652F

CRCW06031003F

15µH 1.5A

R1

R2

R3

10.2kΩ, 1%

16.5kΩ, 1%

100kΩ, 1%

42

Submit Documentation Feedback

Product Folder Links: LM2735

LM2735

www.ti.com

SNVS485F –JUNE 2007–REVISED APRIL 2013

LM2735X SOT-23 LED Design Example 14

L

1

3

FB

2

4

R

SHDN

3

C

2

D

1

Vin

1

5

SW

Vin

C

1

R

1

R

2

DIM-CTRL

Figure 51. LM2735X (1.6MHz): Vin = 2.7V - 5V, Vout = 20V @ 50mA

Part ID

Part Value

2.1A Boost Regulator

22µF, 6.3V, X5R

4.7µF, 25V, X5R

0.4V_fSchottky 500mA, 30V_R

15µH 1.5A

Manufacturer

TI

Part Number

LM2735XMF

U1

C1 Input Cap

TDK

C2012X5R0J226M

C3216JB1E475K

MBR0530

C2 Output Cap

TDK

D1, Catch Diode

Vishay

Coilcraft

Vishay

L1

R1

R2

R3

MSS5131-153ML

CRCW080525R5F

CRCW08051000F

CRCW06031003F

25.5Ω, 1%

100Ω, 1%

100kΩ, 1%

Submit Documentation Feedback

43

Product Folder Links: LM2735

LM2735

SNVS485F –JUNE 2007–REVISED APRIL 2013

www.ti.com

LM2735Y WSON FlyBack Design Example 15

+

12V

D

1

R

2

1

C

f

T

1

R

LOAD

V

IN

C

2

LM2735

R

3

C

1

2

6

5

3

R

LOAD

C

1

D

2

-

12V

3

4

Figure 52. LM2735Y (520kHz): Vin = 5V, Vout = ±12V 150mA

Part ID

Part Value

2.1A Boost Regulator

22µF, 6.3V, X5R

10µF, 25V, X5R

Manufacturer

TI

Part Number

U1

LM2735YSD

C1 Input Cap

TDK

C2012X5R0J226M

C3216X5R1E106M

C1608X5R1H331K

MBR0530

C2 Output Cap

TDK

C3 Output Cap

10µF, 25V, X5R

TDK

Cf Comp Cap

330pF

TDK

D1, D2 Catch Diode

0.4V_fSchottky 500mA, 30V_R

Vishay

T1

R1

R2

R3

10.0kΩ, 1%

86.6kΩ, 1%

100kΩ, 1%

Vishay

CRCW06031002F

CRCW06038662F

CRCW06031003F

44

Submit Documentation Feedback

Product Folder Links: LM2735

LM2735

www.ti.com

SNVS485F –JUNE 2007–REVISED APRIL 2013

LM2735X SOT-23 LED Design Example 16

V_RAIL> 5.5V Application

D

L

1

VPWR

R

2

LM2735

C

EN

4

C

1

R

4

5

3

2

1

R

3

C

2

R

1

D

C

2

3

Figure 53. LM2735X (1.6MHz): V_PWR= 9V, Vout = 12V @ 500mA

Part ID

Part Value

2.1A Boost Regulator

10µF, 6.3V, X5R

10µF, 25V, X5R

0.1µF, 6.3V, X5R

1000pF

Manufacturer

TI

Part Number

LM2735XMF

U1

C1, Input Cap

TDK

C2012X5R0J106K

C3216X5R1E106M

C2012X5R0J104K

C1608X5R1H102K

STPS120M

C2, Output Cap

TDK

C3 V_INCap

TDK

C4 Comp Cap

TDK

D1, Catch Diode

0.4V_fSchottky 1A, 20V_R

3.3V Zener, SOT-23

6.8µH 2A

ST

D2

L1

Diodes Inc

Coilcraft

Vishay

BZX84C3V3

DO1813H-682ML

CRCW08051002F

CRCW08058662F

CRCW06031003F

CRCW06034991F

R1

R2

R3

R4

10.0kΩ, 1%

86.6kΩ, 1%

100kΩ, 1%

499Ω, 1%

Submit Documentation Feedback

45

Product Folder Links: LM2735

LM2735

SNVS485F –JUNE 2007–REVISED APRIL 2013

www.ti.com

LM2735X SOT-23 LED Design Example 17

Two Input Voltage Rail Application

D

L

1

VPWR

C

R

LM2735

EN

4

2

C

1

4

5

3

2

1

R

3

C

2

1

V_IN

C

3

Figure 54. LM2735X (1.6MHz): V_PWR= 9V in = 2.7V - 5.5V, Vout = 12V @ 500mA

Part ID

U1

Part Value

2.1A Boost Regulator

10µF, 6.3V, X5R

10µF, 25V, X5R

0.1µF, 6.3V, X5R

1000pF

Manufacturer

TI

Part Number

LM2735XMF

C2012X5R0J106K

C3216X5R1E106M

C2012X5R0J104K

C1608X5R1H102K

STPS120M

C1, Input Cap

C2, Output Cap

C3 V_INCap

C4 Comp Cap

D1, Catch Diode

L1

TDK

0.4V_fSchottky 1A, 20V_R

6.8µH 2A

ST

Coilcraft

Vishay

DO1813H-682ML

CRCW08051002F

CRCW08058662F

CRCW06031003F

R1

10.0kΩ, 1%

R2

86.6kΩ, 1%

R3

100kΩ, 1%

46

Submit Documentation Feedback

Product Folder Links: LM2735

LM2735

www.ti.com

SNVS485F –JUNE 2007–REVISED APRIL 2013

REVISION HISTORY

Changes from Revision E (April 2013) to Revision F

Page

•

Changed layout of National Data Sheet to TI format .......................................................................................................... 46

Submit Documentation Feedback

47

Product Folder Links: LM2735

PACKAGE MATERIALS INFORMATION

www.ti.com

8-Apr-2013

TAPE AND REEL INFORMATION

*All dimensions are nominal

Device

Package Package Pins

Type Drawing

SPQ

Reel

A0

B0

K0

P1

W

Pin1

Diameter Width (mm) (mm) (mm) (mm) (mm) Quadrant

(mm) W1 (mm)

LM2735XMF/NOPB

LM2735XMFX/NOPB

LM2735XMY/NOPB

SOT-23

DBV

DGN

5

8

1000

3000

1000

178.0

8.4

3.2

5.3

3.2

3.4

1.4

4.0

8.0

Q3

Q1

MSOP-

Power

PAD

12.4

12.0

LM2735XMYX/NOPB

LM2735XQMF/NOPB

MSOP-

Power

PAD

DGN

8

3500

330.0

12.4

5.3

3.4

1.4

8.0

12.0

Q1

SOT-23

DBV

NGG

DBV

DGN

5

6

5

8

1000

3000

1000

4500

1000

3000

1000

178.0

330.0

178.0

8.4

3.2

3.3

3.2

5.3

3.2

3.3

3.2

3.4

1.4

1.0

1.4

4.0

8.0

4.0

8.0

Q3

Q1

Q3

Q1

LM2735XQMFX/NOPB SOT-23

LM2735XSD/NOPB

LM2735XSDX/NOPB

LM2735YMF

WSON

SOT-23

12.4

8.4

12.0

8.0

LM2735YMF/NOPB

LM2735YMFX/NOPB

LM2735YMY/NOPB

8.4

8.0

8.4

8.0

MSOP-

Power

PAD

12.4

12.0

LM2735YMYX/NOPB

MSOP-

Power

DGN

8

3500

330.0

12.4

5.3

3.4

1.4

8.0

12.0

Q1

Pack Materials-Page 1

PACKAGE MATERIALS INFORMATION

www.ti.com

8-Apr-2013

Device

Package Package Pins

Type Drawing

SPQ

Reel

A0

B0

K0

P1

W

Pin1

Diameter Width (mm) (mm) (mm) (mm) (mm) Quadrant

(mm) W1 (mm)

PAD

LM2735YQMF/NOPB

SOT-23

DBV

NGG

5

6

1000

3000

1000

4500

1000

4500

178.0

330.0

178.0

330.0

8.4

3.2

3.3

3.2

3.3

1.4

1.0

4.0

8.0

Q3

Q1

LM2735YQMFX/NOPB SOT-23

LM2735YQSD/NOPB

LM2735YQSDX/NOPB

LM2735YSD/NOPB

LM2735YSDX/NOPB

WSON

12.4

12.0

*All dimensions are nominal

Device

Package Type Package Drawing Pins

SPQ

Length (mm) Width (mm) Height (mm)

LM2735XMF/NOPB

LM2735XMFX/NOPB

LM2735XMY/NOPB

LM2735XMYX/NOPB

LM2735XQMF/NOPB

LM2735XQMFX/NOPB

LM2735XSD/NOPB

LM2735XSDX/NOPB

LM2735YMF

SOT-23

DBV

DGN

DBV

NGG

DBV

5

8

5

6

5

1000

3000

1000

3500

1000

3000

1000

4500

1000

210.0

367.0

210.0

367.0

210.0

185.0

367.0

185.0

367.0

185.0

35.0

MSOP-PowerPAD

SOT-23

WSON

SOT-23

LM2735YMF/NOPB

SOT-23

Pack Materials-Page 2

PACKAGE MATERIALS INFORMATION

www.ti.com

8-Apr-2013

Device

Package Type Package Drawing Pins

SPQ

Length (mm) Width (mm) Height (mm)

LM2735YMFX/NOPB

LM2735YMY/NOPB

LM2735YMYX/NOPB

LM2735YQMF/NOPB

LM2735YQMFX/NOPB

LM2735YQSD/NOPB

LM2735YQSDX/NOPB

LM2735YSD/NOPB

LM2735YSDX/NOPB

SOT-23

MSOP-PowerPAD

SOT-23

DBV

DGN

DBV

NGG

5

8

5

6

3000

1000

3500

1000

3000

1000

4500

1000

4500

210.0

367.0

210.0

367.0

210.0

367.0

185.0

367.0

185.0

367.0

185.0

367.0

35.0

SOT-23

WSON

Pack Materials-Page 3

MECHANICAL DATA

DGN0008A

MUY08A (Rev A)

BOTTOM VIEW

www.ti.com

MECHANICAL DATA

NGG0006A

SDE06A (Rev A)

www.ti.com

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型号：	LM2735YQMF/NOPB
厂家：	TEXAS INSTRUMENTS
描述：	LM2735/LM2735-Q1 520kHz/1.6MHz â Space-Efficient Boost and SEPIC DC-DC Regulator LM2735 / LM2735 -Q1 520kHz / 1.6MHz的â ????节省空间的升压型和SEPIC DC- DC稳压器稳压器
文件：	总58页 (文件大小：1676K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

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