LM3406HVQMHXQ1 [TI]
用于驱动高功率 LED 的 1.5A 汽车类恒流降压稳压器 | PWP | 14 | -40 to 150;
| 型号： | LM3406HVQMHXQ1 |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | 用于驱动高功率 LED 的 1.5A 汽车类恒流降压稳压器 | PWP | 14 | -40 to 150 驱动 光电二极管 接口集成电路 稳压器 |
| 文件： | 总41页 (文件大小：1408K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LM3406, LM3406HV, LM3406HV-Q1
SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014
LM3406 1.5-A, Constant Current, Buck Regulator for Driving High Power LEDs
1 FEATURES
3 DESCRIPTION
The LM3406 family are monolithic switching
regulators designed to deliver constant currents to
high power LEDs. Ideal for automotive, industrial, and
general lighting applications, they contain a high-side
N-channel MOSFET switch with a current limit of
2.0A (typical) for step-down (Buck) regulators.
Controlled on-time with true average current and an
external current sense resistor allow the converter
output voltage to adjust as needed to deliver a
constant current to series and series-parallel
connected LED arrays of varying number and type.
LED dimming via pulse width modulation (PWM) is
achieved using a dedicated logic pin or by PWM of
the power input voltage. The product feature set is
rounded out with low-power shutdown and thermal
shutdown protection.
1
•
LM3406HV-Q1
–
–
–
Automotive Grade Device
AEC-Q100 Grade 1 Qualified
Operating Ambient Temperature: –40°C to
125°C
•
•
•
•
•
•
Integrated 2.0A MOSFET
VIN Range 6V to 42V (LM3406)
VIN Range 6V to 75V (LM3406HV)
VIN Range 6V to 75V (LM3406HV-Q1)
True Average Output Current Control
1.7A Minimum Output Current Limit Over
Temperature
•
•
•
•
•
•
Cycle-by-Cycle Current Limit
The LM3406HV-Q1 is AEC-Q100 grade 1 qualified.
PWM Dimming with Dedicated Logic Input
PWM Dimming with Power Input Voltage
Simple Control Loop Compensation
Low Power Shutdown
Supports All-Ceramic Output Capacitors and
Capacitor-less Outputs
•
•
Thermal Shutdown Protection
TSSOP-14 Package
2 APPLICATIONS
•
•
•
•
•
LED Driver
Constant Current Source
Automotive Lighting
General Illumination
Industrial Lighting
3.1 TYPICAL APPLICATION
C
B
VIN
L1
VIN,VINS
RON
BOOT
SW
R
ON
C
IN
D1
IF
LM3406/06HV
VOUT
DIM
CS
R
SNS
COMP
VCC
GND
C
C
F
C
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LM3406, LM3406HV, LM3406HV-Q1
SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014
www.ti.com
3.1 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
3.1 CONNECTION DIAGRAM
LM3406 family
1
2
3
4
5
6
7
SW
VIN
VIN
14
13
12
11
10
9
SW
BOOT
NC
VINS
VCC
RON
COMP
DIM
DAP
VOUT
CS
GND
8
14-Lead Exposed Pad Plastic TSSOP Package
See Package Number PWP0014A
PIN DESCRIPTIONS
Pin(s)
Name
SW
Description
Switch pin
Application Information
1,2
3
Connect these pins to the output inductor and Schottky diode.
Connect a 22 nF ceramic capacitor from this pin to the SW pins.
No internal connection. Leave this pin unconnected.
BOOT
NC
MOSFET drive bootstrap pin
No Connect
4
5
VOUT
Output voltage sense pin
Connect this pin to the output node where the inductor and the first LED's
anode connect.
6
CS
Current sense feedback pin
Set the current through the LED array by connecting a resistor from this pin
to ground.
7
8
GND
DIM
Ground pin
Connect this pin to system ground.
Input for PWM dimming
Connect a logic-level PWM signal to this pin to enable/disable the power
MOSFET and reduce the average light output of the LED array. Logic high
= output on, logic low - output off.
9
10
COMP
RON
VCC
VINS
VIN
Error amplifier output
On-time control pin
Connect a 0.1 µF ceramic capacitor with X5R or X7R dielectric from this pin
to ground.
A resistor connected from this pin to VIN sets the regulator controlled on-
time.
11
Output of the internal 7V linear
regulator
Bypass this pin to ground with a minimum 0.1 µF ceramic capacitor with
X5R or X7R dielectric.
12
Input voltage PWM dimming
comparator input
Connect this pin to the anode of the input diode to allow dimming by PWM
of the input voltage
13,14
DAP
Input voltage pin
Nominal operating input range for this pin is 6V to 42V (LM3406) or 6V to
75V (LM3406HV, LM3406HV-Q1).
DAP
Thermal Pad
Connect to ground. Place 4-6 vias from DAP to bottom layer ground plane.
2
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SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014
(1)
4 ABSOLUTE MAXIMUM RATINGS
If Military/Aerospace specified devices are required, contact the Texas Instruments Semiconductor Sales Office/
Distributors for availability and specifications.
LM3406
LM3406HV, LM3406HV-Q1
LM3406
–-0.3V to 45V
VIN to GND
–-0.3V to 76V
–-0.3V to 45V
–-0.3V to 76V
–-0.3V to 45V
–-0.3V to 76V
–-0.3V to 59V
–-0.3V to 76V
–1.5V to 45V
–1.5V to 76V
–-0.3V to 45V
–-0.3V to 76V
–-0.3V to 14V
–-0.3V to 14V
–-0.3V to 7V
–-0.3V to 7V
–-0.3V to 7V
–-0.3V to 7V
150°C
VINS to GND
VOUT to GND
BOOT to GND
SW to GND
LM3406HV, LM3406HV-Q1
LM3406
LM3406HV, LM3406HV-Q1
LM3406
LM3406HV, LM3406HV-Q1
LM3406
LM3406HV, LM3406HV-Q1
LM3406
BOOT to VCC
LM3406HV, LM3406HV-Q1
BOOT to SW
VCC to GND
DIM to GND
COMP to GND
CS to GND
RON to GND
Junction Temperature
Storage Temp. Range
-65°C to 125°C
2kV
(2)
ESD Rating
Soldering Information
Lead Temperature (Soldering, 10sec)
Infrared/Convection Reflow (15sec)
260°C
235°C
(1) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur, including inoperability and degradation of
device reliability and/or performance. Functional operation of the device and/or non-degradation at the Absolute Maximum Ratings or
other conditions beyond those indicated in the Operating Ratings is not implied. The recommended Operating Ratings indicate
conditions at which the device is functional and the device should not be operated beyond such conditions.
(2) The human body model is a 100 pF capacitor discharged through a 1.5-kΩ resistor into each pin.
5 RECOMMENDED OPERATING CONDITIONS(1)
LM3406
LM3406HV, LM3406HV-Q1
LM3406, LM3406HV
LM3406HV-Q1
6V to 42V
VIN
6V to 75V
−40°C to +125°C
−40°C to +150°C
−40°C to +125°C
50°C/W
Junction Temperature Range
Ambient Temperature Range
Thermal Resistance θJA (TSSOP-14 Package)(3)
LM3406HV-Q1(2)
(1) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur, including inoperability and degradation of
device reliability and/or performance. Functional operation of the device and/or non-degradation at the Absolute Maximum Ratings or
other conditions beyond those indicated in the Operating Ratings is not implied. The recommended Operating Ratings indicate
conditions at which the device is functional and the device should not be operated beyond such conditions.
(2) The LM3406HV-Q1 can operate at an ambient temperature of up to +125°C as long as the junction temperature maximum of +150°C is
not exceeded.
(3) θJA of 50°C/W with DAP soldered to a minimum of 2 square inches of 1-oz. copper on the top or bottom PCB layer.
Copyright © 2008–2014, Texas Instruments Incorporated
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SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014
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6 ELECTRICAL CHARACTERISTICS LM3406/LM3406HV/LM3406HV-Q1
VIN = 24V unless otherwise indicated. Unless otherwise specified, datasheet typicals and limits apply to LM3406, LM3406HV
and LM3406HV-Q1. Typicals and limits appearing in plain type apply for TA = TJ = +25°C (1). Limits appearing in boldface
type apply over full Operating Temperature Range. Datasheet min/max specification limits are specified by design, test, or
statistical analysis.
Parameter
Test Conditions
Min
Typ
Max
Units
REGULATION COMPARATOR AND ERROR AMPLIFIER
187.5
191.0(2)
200
210
210.0(2)
VREF
CS Regulation Threshold
CS Decreasing, SW turns on
mV
V0V
CS Over-voltage Threshold
CS Bias Current
CS Increasing, SW turns off
CS = 0V
300
0.9
83
mV
µA
µA
µA
µS
ICS
IVOUT
VOUT Bias Current
VOUT = 24V
ICOMP
COMP Pin Current
CS = 0V
25
Gm-CS
SHUTDOWN
Error Amplifier Transconductance
150 mV < CS < 250 mV
145
Shutdown Threshold
RON Increasing
RON Increasing
RON Decreasing
0.3
0.3
0.7
0.7
40
1.05
VSD-TH
V
Shutdown Threshold (LM3406HV-Q1)
Shutdown Hysteresis
1.066
VSD-HYS
mV
ON AND OFF TIMER
tOFF-MIN Minimum Off-time
CS = 0V
230
1300
1300
280
Programmed On-time
VIN = 24V, VO = 12V, RON = 200kΩ
VIN = 24V, VO = 12V, RON = 200kΩ
800
800
1800
1850
tON
ns
Programmed On-time (LM3406HV-Q1)
Minimum On-time
tON-MIN
VINS COMPARATOR
VINS-TH VINS Pin Threshold
IIN-2WD VINS Pin Input Current
INTERNAL REGULATOR
VCC Regulated Output
VINS decreasing
VINS = 24V * 0.7
70
25
%VIN
µA
0 mA < ICC < 5 mA
0 mA < ICC < 5 mA
6.4
6.4
7
7
7.4
7.5
VCC-REG
V
VCC Regulated Output (LM3406HV-Q1)
ICC = 5 mA, 6.0V < VIN < 8.0V,
Non-switching
VIN-DO
VIN - VCC
300
mV
VCC-BP-TH
VCC-LIM
VCC-UV-TH
VCC-UV-HYS
IIN-OP
VCC Bypass Threshold
VIN Increasing
8.8
20
V
VCC Current Limit
VIN = 24V, VCC = 0V
VCC Increasing
4
mA
V
VCC Under-voltage Lock-out Threshold
VCC Under-voltage Lock-out Hysteresis
IIN Operating Current
5.3
150
1.2
240
VCC Decreasing
Non-switching, CS = 0.5V
RON = 0V
mV
mA
µA
IIN-SD
IIN Shutdown Current
350
CURRENT LIMIT
Current Limit Threshold
Current Limit Threshold (LM3406HV-Q1)
DIM COMPARATOR
1.7
2.1
2.1
2.7
ILIM
A
1.65
2.60
VIH
Logic High
DIM Increasing
DIM Decreasing
DIM = 1.5V
2.2
V
V
VIL
Logic Low
0.8
IDIM-PU
DIM Pull-up Current
80
µA
MOSFET AND DRIVER
RDS-ON
Buck Switch On Resistance
ISW = 200 mA, BOOT = 6.3V
BOOT–SW Increasing
0.37
2.9
0.75
4.3
Ω
V
VDR-UVLO
VDR-HYS
BOOT Under-voltage Lock-out Threshold
BOOT Under-voltage Lock-out Hysteresis
1.7
BOOT–SW Decreasing
370
mV
THERMAL SHUTDOWN
TSD
Thermal Shutdown Threshold
Thermal Shutdown Hysteresis
165
25
°C
°C
TSD-HYS
(1) Typical values represent most likely parametric norms at the conditions specified.
(2) Specified with junction temperature from 0°C - 125°C.
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SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014
ELECTRICAL CHARACTERISTICS LM3406/LM3406HV/LM3406HV-Q1 (continued)
VIN = 24V unless otherwise indicated. Unless otherwise specified, datasheet typicals and limits apply to LM3406, LM3406HV
and LM3406HV-Q1. Typicals and limits appearing in plain type apply for TA = TJ = +25°C (1). Limits appearing in boldface
type apply over full Operating Temperature Range. Datasheet min/max specification limits are specified by design, test, or
statistical analysis.
Parameter
Test Conditions
Min
Typ
Max
Units
THERMAL RESISTANCE
θJA Junction to Ambient
(3)
TSSOP-14 Package
50
°C/W
(3) θJA of 50°C/W with DAP soldered to a minimum of 2 square inches of 1-oz. copper on the top or bottom PCB layer.
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SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014
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7 TYPICAL PERFORMANCE CHARACTERISTICS
Figure 1. Efficiency vs. Number of InGaN LEDs in Series
Figure 2. Efficiency Vs. Output Current
(1)
(1)
Figure 3. VREF vs Temperature
Figure 4. VREF vs VIN, LM3406
Figure 5. VREF vs VIN, LM3406HV/LM3406HV-Q1
Figure 6. Current Limit vs Temperature
(1) VIN = 24V, IF = 1A, TA = 25°C, and the load consists of three InGaN LEDs in series unless otherwise noted. See the Bill of Materials
table at the end of the datasheet.
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SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014
TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Figure 7. Current Limit vs VIN, LM3406
Figure 8. Current Limit vs VIN, LM3406HV/LM3406HV-Q1
Figure 9. VCC vs VIN
Figure 10. VO-MAX vs VIN, LM3406
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7.1 BLOCK DIAGRAM
GND
7V BIAS
REGULATOR
VIN
VCC
VIN
SENSE
VCC
UVLO
THERMAL
SHUTDOWN
BYPASS
SWITCH
RON
-
+
0.7V
300 ns MIN
OFF TIMER
Complete
ON TIMER
Complete
On-Time
Current
Generator
VOUT
VINS
BOOT
Start
Start
VIN
SD
GATE DRIVE
UVLO
VIN
x 0.7
+
-
5V
LEVEL
SHIFT
75 éA
DIM
+
-
SW
LOGIC
1.5V
COMP
BUCK
0.2V
SWITCH
CURRENT
SENSE
CURRENT
LIMIT OFF
TIMER
+
-
+
-
+
-
2.0A
CS
8
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SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014
8 APPLICATION INFORMATION
8.1 THEORY OF OPERATION
The LM3406, LM3406HV and LM3406HV-Q1 are buck regulators with a wide input voltage range, low voltage
reference, and a fast output enable/disable function. These features combine to make them ideal for use as a
constant current source for LEDs with forward currents as high as 1.5A. The controlled on-time (COT)
architecture uses a comparator and a one-shot on-timer that varies inversely with input and output voltage
instead of a fixed clock. The LM3406 family also employs an integrator circuit that averages the output current.
When the converter runs in continuous conduction mode (CCM) the controlled on-time maintains a constant
switching frequency over changes in both input and output voltage. These features combine to give the LM3406
family an accurate output current, fast transient response, and constant switching frequency over a wide range of
conditions.
8.2 CONTROLLED ON-TIME OVERVIEW
shows a simplified version of the feedback system used to control the current through an array of LEDs. A
differential voltage signal, VSNS, is created as the LED current flows through the current setting resistor, RSNS
.
VSNS is fed back by the CS pin, where it is integrated and compared against an error amplifier-generated
reference. The error amplifier is a transconductance (Gm) amplifier which adjusts the voltage on COMP to
maintain a 200 mV average at the CS pin. The on-comparator turns on the power MOSFET when VSNS falls
below the reference created by the Gm amp. The power MOSFET conducts for a controlled on-time, tON, set by
an external resistor, RON, the input voltage, VIN and the output voltage, VO. On-time can be estimated by the
following simplified equation (for the most accurate version of this expression see the Appendix):
VO
tON = 1 x 10-11 x RON
x
VIN
(1)
At the conclusion of tON the power MOSFET turns off and must remain off for a minimum of 230 ns. Once this
tOFF-MIN is complete the CS comparator compares the integrated VSNS and reference again, waiting to begin the
next cycle.
VO
LED 1
V
F
IF
LM3406/06HV
LED n
V
F
-
+
CS
VSNS
One-shot
-
+
0.2V
IF
R
SNS
COMP
Comparator and One-Shot
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8.3 SWITCHING FREQUENCY
The LM3406 family does not contain a clock, however the on-time is modulated in proportion to both input
voltage and output voltage in order to maintain a relatively constant frequency. On-time tON, duty cycle D and
switching frequency fSW are related by the following expression:
fSW = D / tON
(2)
(3)
(4)
(5)
D = (VO + VD) / (VIN - VSW + VD)
VD = Schottky diode (typically 0.5V)
VSW = IF x RDSON
The LM3406 family regulators should be operated in continuous conduction mode (CCM), where inductor current
stays positive throughout the switching cycle. During steady-state CCM operation, the converter maintains a
constant switching frequency that can be estimated using the following equation (for the most accurate version,
particularly for applications that will have an input or output voltage of less than approximately 12V, see the
Appendix):
1
fSW
=
1 x 10-11 x RON
(6)
(7)
8.4 SETTING LED CURRENT
LED current is set by the resistor RSNS, which can be determined using the following simple expression due to
the output averaging:
RSNS = 0.2 / IF
(8)
8.5 MAXIMUM NUMBER OF SERIES LEDS
LED driver designers often want to determine the highest number of LEDs that can be driven by their circuits.
The limit on the maximum number of series LEDs is set by the highest output voltage, VO-MAX, that the LED driver
can provide. A buck regulator cannot provide an output voltage that is higher than the minimum input voltage,
and in pratice the maximum output voltage of the LM3406 family is limited by the minimum off-time as well. VO-
determines how many LEDs can be driven in series. Referring to the illustration in , output voltage is
MAX
calculated as:
VO-MAX = VIN-MIN x (1 - fSW x tOFF-MIN
)
(9)
tOFF-MIN = 230 ns
Once VO-MAX has been calculated, the maximum number of series LEDs, nMAX, can be calculated by the following
espression and rounding down:
nMAX = VO-MAX / VF
(10)
VF = forward voltage of each LED
At low switching frequency VO-MAX is higher, allowing the LM3406 family to regulate output voltages that are
nearly equal to input voltage, and this can allow the system to drive more LEDs in series. Low switching
frequencies are not always desireable, however, because they require larger, more expensive components.
8.6 CALCULATING OUTPUT VOLTAGE
Even though output current is the controlled parameter in LED drivers, output voltage must still be calculated in
order to design the complete circuit. Referring to the illustration in , output voltage is calculated as:
VO = n x VF + VSNS
(11)
VSNS = sense voltage of 200 mV, n = number of LEDs in series
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8.7 MINIMUM ON-TIME
The minimum on-time for the LM3406 family is 280 ns (typical). One practical example of reaching the minimum
on-time is when dimming the LED light output with a power FET placed in parallel to the LEDs. When the FET is
on, the output voltage drops to 200 mV. This results in a small duty cycle and in most circuits requires an on-time
that would be less than 280 ns. In such a case the LM3406 family keeps the on-time at 280 ns and increases the
off-time as much as needed, which effectively reduces the switching frequency.
8.8 HIGH VOLTAGE BIAS REGULATOR (VCC)
The LM3406 family contains an internal linear regulator with a 7V output, connected between the VIN and the
VCC pins. The VCC pin should be bypassed to the GND pin with a 0.1 µF ceramic capacitor connected as close
as possible to the pins of the IC. VCC tracks VIN until VIN reaches 8.8V (typical) and then regulates at 7V as
VIN increases. The LM3406 family comes out of UVLO and begins operating when VCC crosses 5.3V. This is
shown graphically in the Typical Performance curves.
Connecting an external supply to VCC to power the gate drivers is not recommended. However, it may be done if
certain precautions are taken. Be sure that the external supply will not violate any absolute maximum conditions
and will at no point exceed the voltage applied to the VIN pins. Under certain conditions, some ringing may be
present on the SW and BOOT pins when VCC is driven with an external supply. It is important to ensure that the
absolute maximum ratings of these pins are not violated during the ringing or else damage to the device may
occur.
8.9 INTERNAL MOSFET AND DRIVER
The LM3406 family features an internal power MOSFET as well as a floating driver connected from the SW pin
to the BOOT pin. Both rise time and fall time are 20 ns each (typical) and the approximate gate charge is 9 nC.
The high-side rail for the driver circuitry uses a bootstrap circuit consisting of an internal high-voltage diode and
an external 22 nF capacitor, CB. VCC charges CB through the internal diode while the power MOSFET is off.
When the MOSFET turns on, the internal diode reverse biases. This creates a floating supply equal to the VCC
voltage minus the diode drop to drive the MOSFET when its source voltage is equal to VIN.
8.10 FAST LOGIC PIN FOR PWM DIMMING
The DIM pin is a TTL compatible input for PWM dimming of the LED. A logic low (below 0.8V) at DIM will disable
the internal MOSFET and shut off the current flow to the LED array. While the DIM pin is in a logic low state the
support circuitry (driver, bandgap, VCC) remains active in order to minimize the time needed to turn the LED
array back on when the DIM pin sees a logic high (above 2.2V). A 75 µA (typical) pull-up current ensures that the
LM3406 family is on when DIM pin is open circuited, eliminating the need for a pull-up resistor. Dimming
frequency, fDIM, and duty cycle, DDIM, are limited by the LED current rise time and fall time and the delay from
activation of the DIM pin to the response of the internal power MOSFET. In general, fDIM should be at least one
order of magnitude lower than the steady state switching frequency in order to prevent aliasing.
8.11 INPUT VOLTAGE COMPARATOR FOR PWM DIMMING
Adding an external input diode and using the internal VINS comparator allows the LM3406 family to sense and
respond to dimming that is done by PWM of the input voltage. This method is also referred to as "Two-Wire
Dimming", and a typical application circuit is shown in . If the VINS pin voltage falls 70% below the VIN pin
voltage, the LM3406 family disables the internal power FET and shuts off the current to the LED array. The
support circuitry (driver, bandgap, VCC) remains active in order to minimize the time needed to the turn the LED
back on when the VINS pin voltage rises and exceeds 70% of VIN. This minimizes the response time needed to
turn the LED array back on.
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INPUT VOLTAGE COMPARATOR FOR PWM DIMMING (continued)
C
B
L1
D1
C
VIN
BOOT
SW
VIN
R
ON
IN
D2
RON
IF
LM3406/06HV
VINS
VOUT
CS
DIM
R
SNS
COMP
VCC
GND
C
C
F
C
Typical Application using Two-Wire Dimming
8.12 PARALLEL MOSFET FOR HIGH-SPEED PWM DIMMING
For applications that require dimming at high frequency or with wide dimming duty cycle range neither the VINS
comparator or the DIM pin are capable of slewing the LED current from 0 to the target level fast enough. For
such applications the LED current slew rate can by increased by shorting the LED current with a N-MOSFET
placed in parallel to the LED or LED array, as shown in . While the parallel FET is on the output current flows
through it, effectively reducing the output voltage to equal the CS pin voltage of 0.2V. This dimming method
maintains a continuous current through the inductor, and therefore eliminates the biggest delay in turning the
LED(s) or and off. The trade-off with parallel FET dimming is that more power is wasted while the FET is on,
although in most cases the power wasted is small compared to the power dissipated in the LEDs. Parallel FET
circuits should use no output capacitance or a bare minimum for noise filtering in order to minimize the slew rate
of output voltage. Dimming FET Q1 can be driven from a ground-referenced source because the source stays at
0.2V along with the CS pin.
C
B
IF
VIN
L1
VIN,VINS
RON
BOOT
SW
R
ON
C
IN
D1
Q1
LM3406/06HV
VOUT
DIM
CS
R
SNS
COMP
VCC
GND
C
C
F
C
Dimming with a Parallel FET
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8.13 PEAK CURRENT LIMIT
The current limit comparator of the LM3406 family will engage whenever the power MOSFET current (equal to
the inductor current while the MOSFET is on) exceeds 2.1A (typical). The power MOSFET is disabled for a cool-
down time that of approximately 100 µs. At the conclusion of this cool-down time the system re-starts. If the
current limit condition persists the cycle of cool-down time and restarting will continue, creating a low-power
hiccup mode, minimizing thermal stress on the LM3406 family and the external circuit components.
8.14 OVER-VOLTAGE/OVER-CURRENT COMPARATOR
The CS pin includes an output over-voltage/over-current comparator that will disable the power MOSFET
whenever VSNS exceeds 300 mV. This threshold provides a hard limit for the output current. Output current
overshoot is limited to 300 mV / RSNS by this comparator during transients. The OVP/OCP comparator limits the
maximum ripple voltage at the CS pin to 200 mVP-P
.
8.15 OUTPUT OPEN-CIRCUIT
The most common failure mode for power LEDs is a broken bond wire, and the result is an output open-circuit.
When this happens the feedback path is disconnected, and the output voltage will attempt to rise. In buck
converters the output voltage can only rise as high as the input voltage, and the minimum off-time requirement
ensures that VO(MAX) is slightly less than VIN. shows a method using a zener diode, Z1, and zener limiting
resistor, RZ, to limit output voltage to the reverse breakdown voltage of Z1 plus 200 mV. The zener diode reverse
breakdown voltage, VZ, must be greater than the maximum combined VF of all LEDs in the array. The maximum
recommended value for RZ is 1 kΩ.
The output stage (SW and VOUT pins) of the LM3406 family is capable of withstanding VO(MAX) indefinitely as
long as the output capacitor is rated to handle the full input voltage. When an LED fails open-circuit and there is
no output capacitor present the surge in output voltage due to the collapsing magnetic field in the output inductor
can exceed VIN and can damage the LM3406 family IC. As an alternative to the zener clamp method described
previously, a diode can be connected from the output to the input of the regulator circuit that will clamp the
inductive surge to one VD above VIN.
Regardless of which protection method is used a resistance in series with the VOUT pin, ROUT, is recommended
to limit the current in the event the VOUT pin is pulled below ground when the LED circuit is reconnected. This
can occur frequently if the lead lengths to the LEDs are long and the inductance is significant. A resistor between
1 kΩ and 10 kΩ is recommended.
D2
C
B
VIN
L1
VIN,VINS
RON
BOOT
SW
R
ON
C
IN
D1
IF
LM3406/6HV
Z1
VOUT
CS
R
Z
R
OUT
DIM
R
SNS
COMP
VCC
GND
C
C
F
C
Two Methods of Output Open Circuit Protection
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8.16 LOW POWER SHUTDOWN
The LM3406 family can be placed into a low power state (IIN-SD = 240 µA) by grounding the RON pin with a
signal-level MOSFET as shown in . Low power MOSFETs like the 2N7000, 2N3904, or equivalent are
recommended devices for putting the LM3406 family into low power shutdown. Logic gates can also be used to
shut down the LM3406 family as long as the logic low voltage is below the over temperature minimum threshold
of 0.3V. Noise filter circuitry on the RON pin can cause a few pulses with longer on-times than normal after RON
is grounded or released. In these cases the OVP/OCP comparator will ensure that the peak inductor or LED
current does not exceed 300 mV / RSNS
.
C
B
VIN
L1
VIN,VINS
RON
BOOT
SW
R
ON
C
IN
D1
IF
LM3406/06HV
ON/OFF
Q1
VOUT
2N7000 or
equivalent
DIM
CS
R
SNS
COMP
VCC
GND
C
C
F
C
Low Power Shutdown
8.17 THERMAL SHUTDOWN
Internal thermal shutdown circuitry is provided to protect the IC in the event that the maximum junction
temperature is exceeded. The threshold for thermal shutdown is 165°C with a 25°C hysteresis (both values
typical). During thermal shutdown the MOSFET and driver are disabled.
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8.1 DESIGN CONSIDERATIONS
8.1.1 SWITCHING FREQUENCY
Switching frequency is selected based on the trade-offs between efficiency (better at low frequency), solution
size/cost (smaller at high frequency), and the range of output voltage that can be regulated (wider at lower
frequency.) Many applications place limits on switching frequency due to EMI sensitivity. The on-time of the
LM3406 family can be programmed for switching frequencies ranging from the 10’s of kHz to over 1 MHz. This
on-time varies in proportion to both VIN and VO in order to maintain first-order control over switching frequency,
however in practice the switching frequency will shift in response to large swings in VIN or VO. The maximum
switching frequency is limited only by the minimum on-time and minimum off-time requirements.
8.1.2 LED RIPPLE CURRENT
Selection of the ripple current, ΔiF, through the LED array is similar to the selection of output ripple voltage in a
standard voltage regulator. Where the output ripple in a voltage regulator is commonly ±1% to ±5% of the DC
output voltage, LED manufacturers generally recommend values for ΔiF ranging from ±5% to ±20% of IF. Higher
LED ripple current allows the use of smaller inductors, smaller output capacitors, or no output capacitors at all.
Lower ripple current requires more output inductance, higher switching frequency, or additional output
capacitance, and may be necessary for applications that are not intended for human eyes, such as machine
vision or industrial inspection.
8.1.3 BUCK CONVERTERS WITHOUT OUTPUT CAPACITORS
The buck converter is unique among non-isolated topologies because of the direct connection of the inductor to
the load during the entire switching cycle. By definition an inductor will control the rate of change of current that
flows through it, and this control over current ripple forms the basis for component selection in both voltage
regulators and current regulators. A current regulator such as the LED driver for which the LM3406 family was
designed focuses on the control of the current through the load, not the voltage across it. A constant current
regulator is free of load current transients, and has no need of output capacitance to supply the load and
maintain output voltage. Referring to the Typical Application circuit on the front page of this datasheet, the
inductor and LED can form a single series chain, sharing the same current. When no output capacitor is used,
the same equations that govern inductor ripple current, ΔiL, also apply to the LED ripple current, ΔiF. For a
controlled on-time converter such as LM3406 family the ripple current is described by the following expression:
VIN - VO
x tON
'iL = 'iF =
L
(12)
The triangle-wave inductor current ripple flows through RSNS and produces a triangle-wave voltage at the CS pin.
To provide good signal to noise ratio (SNR) the amplitude of CS pin ripple voltage, ΔvCS, should be at least 25
mVP-P. ΔvCS is described by the following:
ΔvCS = ΔiF x RSNS
(13)
8.1.4 BUCK CONVERTERS WITH OUTPUT CAPACITORS
A capacitor placed in parallel with the LED(s) can be used to reduce the LED current ripple while keeping the
same average current through both the inductor and the LED array. With an output capacitor the output
inductance can be lowered, making the magnetics smaller and less expensive. Alternatively, the circuit could be
run at lower frequency but keep the same inductor value, improving the power efficiency. Both the peak current
limit and the OVP/OCP comparator still monitor peak inductor current, placing a limit on how large ΔiL can be
even if ΔiF is made very small. Adding a capacitor that reduces ΔiF to well below the target provides headroom
for changes in inductance or VIN that might otherwise push the peak LED ripple current too high.
shows the equivalent impedances presented to the inductor current ripple when an output capacitor, CO, and its
equivalent series resistance (ESR) are placed in parallel with the LED array. Note that ceramic capacitors have
so little ESR that it can be ignored. The entire inductor ripple current still flows through RSNS to provide the
required 25 mV of ripple voltage for proper operation of the CS comparator.
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DESIGN CONSIDERATIONS (continued)
'i
L
C
O
'i
r
D
'i
C
F
ESR
'i
L
R
SNS
LED and CO Ripple Current
To calculate the respective ripple currents the LED array is represented as a dynamic resistance, rD. LED
dynamic resistance is not always specified on the manufacturer’s datasheet, but it can be calculated as the
inverse slope of the LED’s VF vs. IF curve. Note that dividing VF by IF will give an incorrect value that is 5x to 10x
too high. Total dynamic resistance for a string of n LEDs connected in series can be calculated as the rD of one
device multiplied by n. Inductor ripple current is still calculated with the expression from Buck Regulators without
Output Capacitors. The following equations can then be used to estimate ΔiF when using a parallel capacitor:
'iL
'iF =
rD
1 +
ZC
1
ZC = ESR +
2Sꢀx fSW x CO
(14)
The calculation for ZC assumes that the shape of the inductor ripple current is approximately sinusoidal.
Small values of CO that do not significantly reduce ΔiF can also be used to control EMI generated by the
switching action of the LM3406 family. EMI reduction becomes more important as the length of the connections
between the LED and the rest of the circuit increase.
8.1.5 INPUT CAPACITORS
Input capacitors at the VIN pin of the LM3406 family are selected using requirements for minimum capacitance
and rms ripple current. The input capacitors supply pulses of current approximately equal to IF while the power
MOSFET is on, and are charged up by the input voltage while the power MOSFET is off. All switching regulators
have a negative input impedance due to the decrease in input current as input voltage increases. This inverse
proportionality of input current to input voltage can cause oscillations (sometimes called ‘power supply
interaction’) if the magnitude of the negative input impedance is greater the the input filter impedance. Minimum
capacitance can be selected by comparing the input impedance to the converter’s negative resistance; however
this requires accurate calculation of the input voltage source inductance and resistance, quantities which can be
difficult to determine. An alternative method to select the minimum input capacitance, CIN(MIN), is to select the
maximum input voltage ripple which can be tolerated. This value, ΔvIN(MAX), is equal to the change in voltage
across CIN during the converter on-time, when CIN supplies the load current. CIN(MIN) can be selected with the
following:
IF x tON
CIN (MIN)
=
'VIN (MAX)
(15)
A good starting point for selection of CIN is to use an input voltage ripple of 5% to 10% of VIN. A minimum input
capacitance of 2x the CIN(MIN) value is recommended for all LM3406 family circuits. To determine the rms current
rating, the following formula can be used:
IIN(rms) = IF x
D(1 - D)
(16)
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DESIGN CONSIDERATIONS (continued)
Ceramic capacitors are the best choice for the input to the LM3406 family due to their high ripple current rating,
low ESR, low cost, and small size compared to other types. When selecting a ceramic capacitor, special
attention must be paid to the operating conditions of the application. Ceramic capacitors can lose one-half or
more of their capacitance at their rated DC voltage bias and also lose capacitance with extremes in temperature.
A DC voltage rating equal to twice the expected maximum input voltage is recommended. In addition, the
minimum quality dielectric which is suitable for switching power supply inputs is X5R, while X7R or better is
preferred.
8.1.6 RECIRCULATING DIODE
The LM3406 family is a non-synchronous buck regulator that requires a recirculating diode D1 (see the Typical
Application circuit) to carrying the inductor current during the MOSFET off-time. The most efficient choice for D1
is a Schottky diode due to low forward drop and near-zero reverse recovery time. D1 must be rated to handle the
maximum input voltage plus any switching node ringing when the MOSFET is on. In practice all switching
converters have some ringing at the switching node due to the diode parasitic capacitance and the lead
inductance. D1 must also be rated to handle the average current, ID, calculated as:
ID = (1 – D) x IF
(17)
This calculation should be done at the maximum expected input voltage. The overall converter efficiency
becomes more dependent on the selection of D1 at low duty cycles, where the recirculating diode carries the
load current for an increasing percentage of the time. This power dissipation can be calculating by checking the
typical diode forward voltage, VD, from the I-V curve on the product datasheet and then multiplying it by ID. Diode
datasheets will also provide a typical junction-to-ambient thermal resistance, θJA, which can be used to estimate
the operating die temperature of the device. Multiplying the power dissipation (PD = ID x VD) by θJA gives the
temperature rise. The diode case size can then be selected to maintain the Schottky diode temperature below
the operational maximum.
8.2 Transient Protection Considerations
Considerations need to be made when external sources, loads or connections are made to the switching
converter circuit due to the possibility of Electrostatic Discharge (ESD) or Electric Over Stress (EOS) events
occurring and damaging the integrated circuit (IC) device. All IC device pins contain zener based clamping
structures that are meant to clamp ESD. ESD events are very low energy events, typically less than 5µJ
(microjoules). Any event that transfers more energy than this may damage the ESD structure. Damage is
typically represented as a short from the pin to ground as the extreme localized heat of the ESD / EOS event
causes the aluminum metal on the chip to melt, causing the short. This situation is common to all integrated
8.2.1 CS PIN PROTECTION
When hot swapping in a load (e.g. test points, load boards, LED stack), any residual charge on the load will be
immediately transferred through the output capacitor to the CS pin, which is then damaged as shown in below.
The EOS event due to the residual charge from the load is represented as VTRANSIENT
.
From measurements, we know that the 8V ESD structure on the CS pin can typically withstand 25mA of direct
current (DC). Adding a 1kΩ resistor in series with the CS pin, shown in , results in the majority of the transient
energy to pass through the discrete sense resistor rather than the device. The series resistor limits the peak
current that can flow during a transient event, thus protecting the CS pin. With the 1kΩ resistor shown, a 33V,
49A transient on the LED return connector terminal could be absorbed as calculated by:
V = 25mA * 1kΩ + 8V = 33V
I = 33V / 0.67Ω = 49A
(18)
(19)
This is an extremely high energy event, so the protection measures previously described should be adequate to
solve this issue.
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Transient Protection Considerations (continued)
LM3406
SW
Module
Connector
Module
Connector
V
TRANSIENT
CS
8V
~ 0.675
GND
CS Pin, Transient Path
LM3406
SW
Module
Connector
Module
Connector
V
TRANSIENT
CS
1k5
8V
~ 0.675
GND
CS Pin, Transient Path with Protection
Adding a resistor in series with the CS pin causes the observed output LED current to shift very slightly. The
reason for this is twofold: (1) the CS pin has about 20pF of inherent capacitance inside it which causes a slight
delay (20ns for a 1kΩ series resistor), and (2) the comparator that is watching the voltage at the CS pin uses a
pnp bipolar transistor at its input. The base current of this pnp transistor is approximately 100nA which will cause
a 0.1mV change in the 200mV threshold. These are both very minor changes and are well understood. The shift
in current can either be neglected or taken into consideration by changing the current sense resistance slightly.
8.2.2 CS PIN PROTECTION WITH OVP
When designing output overvoltage protection into the switching converter circuit using a zener diode, transient
protection on the CS pin requires additional consideration. As shown in , adding a zener diode from the output to
the CS pin (with the series resistor) for output overvoltage protection will now again allow the transient energy to
be passed through the CS pin’s ESD structure thereby damaging it.
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Transient Protection Considerations (continued)
Adding an additional series resistor to the CS pin as shown in will result in the majority of the transient energy to
pass through the sense resistor thereby protecting the LM340X device.
LM3406
SW
Module
Connector
Module
Connector
V
TRANSIENT
CS
1 k5
8V
~ 0.675
GND
CS Pin with OVP, Transient Path
LM3406
SW
Module
Connector
Module
Connector
V
TRANSIENT
CS
1 k5
5005
8V
~ 0.675
GND
CS Pin with OVP, Transient Path with Protection
8.2.3 VIN PIN PROTECTION
The VIN pin also has an ESD structure from the pin to GND with a breakdown voltage of approximately 80V. Any
transient that exceeds this voltage may damage the device. Although transient absorption is usually present at
the front end of a switching converter circuit, damage to the VIN pin can still occur.
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Transient Protection Considerations (continued)
When VIN is hot swapped in, the current that rushes in to charge CIN up to the VIN value also charges (energizes)
the circuit board trace inductance as shown in . The excited trace inductance then resonates with the input
capacitance (similar to an under-damped LC tank circuit) and causes voltages at the VIN pin to rise well in
excess of both VIN and the voltage at the module input connector as clamped by the input TVS. If the resonating
voltage at the VIN pin exceeds the 80V breakdown voltage of the ESD structure, the ESD structure will activate
and then “snap-back” to a lower voltage due to its inherent design. If this lower snap-back voltage is less than
the applied nominal VIN voltage, then significant current will flow through the ESD structure resulting in the IC
being damaged.
An additional TVS or small zener diode should be placed as close as possible to the VIN pins of each IC on the
board, in parallel with the input capacitor as shown in . A minor amount of series resistance in the input line
would also help, but would lower overall conversion efficiency. For this reason, NTC resistors are often used as
inrush limiters instead.
LM3406
Board Trace
Inductance
VIN
Module
Connector
80V
C
IN
TVS
V
IN
GND
Module
Connector
VIN Pin with Typical Input Protection
LM3406
Board Trace
Inductance
VIN
Module
Connector
80V
TVS
TVS or
smaller
V
C
IN
IN
zener diode
GND
Module
Connector
VIN Pin with Additional Input Protection
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Transient Protection Considerations (continued)
8.2.4 GENERAL COMMENTS REGARDING OTHER PINS
Any pin that goes “off-board” through a connector should have series resistance of at least 1kΩ to 10kΩ in series
with it to protect it from ESD or other transients. These series resistors limit the peak current that can flow (or
cause a voltage drop) during a transient event, thus protecting the pin and the device. Pins that are not used
should not be left floating. They should instead be tied to GND or to an appropriate voltage through resistance.
8.3 Design Example 1
The first example circuit uses the LM3406 to create a flexible LED driver capable of driving anywhere from one to
five white series-connected LEDs at a current of 1.5A ±5% from a regulated DC voltage input of 24V ±10%. In
addition to the ±5% tolerance specified for the average output current, the LED ripple current must be controlled
to 10%P-P of the DC value, or 150 mAP-P. The typical forward voltage of each individual LED at 1.5A is 3.9V,
hence the output voltage ranges from 4.1V to 19.7V, adding in the 0.2V drop for current sensing. A complete bill
of materials can be found in Table 1 at the end of this datasheet.
C
B
IF = 1.5A ±5%
VIN = 24V ±10%
L1
VIN
BOOT
SW
R
ON
C
IN
D1
LED1
LEDn
RON
One to
five
LEDs
C
O
LM3406
VOUT
CS
DIM
R
SNS
COMP
VCC
GND
C
C
C
F
Schematic for Design Example 1
8.3.1 RON and tON
A moderate switching frequency of 500 kHz will balance the requirements of inductor size and overall power
efficiency. The LM3406 will allow some shift in switching frequency when VO changes due to the number of LEDs
in series, so the calculation for RON is done at the mid-point of three LEDs in series, where VO = 11.8V. Note that
the actual RON calculation is done with the high accuracy expression listed in the Appendix.
1
RON
=
fSW x 1 x 10-11
(20)
(21)
RON = 144 kΩ
The closest 1% tolerance resistor is 143 kΩ. The switching frequency and on-time of the circuit should be
checked for one, three and five LEDs using the equations relating RON and tON to fSW. As with the RON
calculation, the actual fSW and tON values have been calculated using the high accuracy expressions listed in the
Appendix.
1
fSW
=
1 x 10-11 x RON
(22)
(23)
(24)
(25)
fSW(1 LED) = 362 kHz
fSW(3 LEDs) = 504 kHz
fSW(5 LEDs) = 555 kHz
VO
tON = 1 x 10-11 x RON
x
VIN
(26)
(27)
tON(1 LED) = 528 ns
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Design Example 1 (continued)
tON(3 LEDs) = 1014 ns
(28)
(29)
tON(5 LEDs) = 1512 ns
8.3.2 OUTPUT INDUCTOR
Since an output capacitor will be used to filter some of the AC ripple current, the inductor ripple current can be
set higher than the LED ripple current. A value of 40%P-P is typical in many buck converters:
ΔiL = 0.4 x 1.5 = 0.6AP-P
(30)
With the target ripple current determined the inductance can be chosen:
VIN - VO
x tON
L =
'iL
(31)
(32)
LMIN = [(24 – 11.8) x 1.01 x 10-6] / (0.6) = 20.5 µH
The closest standard inductor value is 22 µH. The average current rating should be greater than 1.5A to prevent
overheating in the inductor. Inductor current ripple should be calculated for one, three and five LEDs:
ΔiL(1 LED) = [(24 - 4.1) x 5.28 x 10-7] / 22 x 10-6 = 478 mAP-P
ΔiL(3 LEDs) = [(24 - 11.8) x 1.01 x 10-6] / 22 x 10-6 = 560 mAP-P
ΔiL(5 LEDs) = [(24 - 19.7) x 1.51 x 10-6] / 22 x 10-6 = 295 mAP-P
(33)
(34)
(35)
The peak LED/inductor current is then estimated. This calculation uses the worst-case ripple current which
occurs with three LEDs.
IL(PEAK) = IL + 0.5 x ΔiL(MAX)
(36)
(37)
IL(PEAK) = 1.5 + 0.5 x 0.56 = 1.78A
In order to prevent inductor saturation the inductor’s peak current rating must be above 1.8A. A 22 µH off-the
shelf inductor rated to 2.1A (peak) and 1.9A (average) with a DCR of 59 mΩ will be used.
8.3.3 USING AN OUTPUT CAPACITOR
This application does not require high frequency PWM dimming, allowing the use of an output capacitor to
reduce the size and cost of the output inductor while still meeting the 10%P-P target for LED ripple current. To
select the proper output capacitor the equation from Buck Regulators with Output Capacitors is re-arranged to
yield the following:
'iF
x rD
ZC
=
'iL - 'iF
(38)
The dynamic resistance, rD,of one LED can be calculated by taking the tangent line to the VF vs. IF curve in the
LED datasheet. shows an example rD calculation.
ÂIF
ÂVF
Calculating rD from the VF vs. IF Curve
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Design Example 1 (continued)
Extending the tangent line to the ends of the plot yields values for ΔVF and ΔIF of 0.7V and 2000 mA,
respectively. Dynamic resistance is then:
rD = ΔVF / ΔIF = 0.5V / 2A = 0.25Ω
(39)
The most filtering (and therefore the highest output capacitance) is needed when rD is lowest, which is when
there is only one LED. Inductor ripple current with one LED is 478 mAP-P. The required impedance of CO is
calculated:
ZC = [0.15 / (0.478 - 0.15] x 0.35 = 0.114Ω
(40)
A ceramic capacitor will be used and the required capacitance is selected based on the impedance at 362 kHz:
CO = 1/(2 x π x 0.16 x 3.62 x 105) = 3.9 µF
(41)
This calculation assumes that CO will be a ceramic capacitor, and therefore impedance due to the equivalent
series resistance (ESR) and equivalent series inductance (ESL) of of the device is negligible. The closest 10%
tolerance capacitor value is 4.7 µF. The capacitor used should be rated to 25V or more and have an X7R
dielectric. Several manufacturers produce ceramic capacitors with these specifications in the 1206 case size. A
typical value for ESR of 3 mΩ can be read from the curve of impedance vs. frequency in the product datasheet.
8.3.4 RSNS
Using the expression for RSNS
:
RSNS = 0.2 / IF
(42)
(43)
RSNS = 0.2 / 1.5 = 0.133Ω
Sub-1Ω resistors are available in both 1% and 5% tolerance. A 1%, 0.13Ω device is the closest value, and a
0.33W, 1210 size device will handle the power dissipation of 290 mW. With the resistance selected, the average
value of LED current is re-calculated to ensure that current is within the ±5% tolerance requirement. From the
expression for average LED current:
IF = 0.2 / 0.13 = 1.54A, 3% above the target current
(44)
8.3.5 INPUT CAPACITOR
Following the calculations from the Input Capacitor section, ΔvIN(MAX) will be 24V x 2%P-P = 480 mV. The
minimum required capacitance is calculated for the largest tON, corresponding to five LEDs:
CIN(MIN) = (1.5 x 1.5 x 10-6) / 0.48 = 4.7 µF
(45)
As with the output capacitor, this required value is low enough to use a ceramic capacitor, and again the effective
capacitance will be lower than the rated value with 24V across CIN. Reviewing plots of %C vs. DC Bias for
several capacitors reveals that a 4.7 µF, 1812-size capacitor in X7R rated to 50V loses about 40% of its rated
capacitance at 24V, hence two such caps are needed.
Input rms current is high in buck regulators, and the worst-case is when the duty cycle is 50%. Duty cycle in a
buck regulator can be estimated as D = VO / VIN, and when this converter drives three LEDs the duty cycle will
be nearly 50%.
IIN-RMS = 1.5 x Sqrt(0.5 x 0.5) = 750 mA
(46)
Ripple current ratings for 1812 size ceramic capacitors are typically higher than 2A, so two of them in parallel can
tolerate more than enough for this design.
8.3.6 RECIRCULATING DIODE
The input voltage of 24V ±5% requires Schottky diodes with a reverse voltage rating greater than 30V. The next
highest standard voltage rating is 40V. Selecting a 40V rated diode provides a large safety margin for the ringing
of the switch node and also makes cross-referencing of diodes from different vendors easier.
The next parameters to be determined are the forward current rating and case size. The lower the duty cycle the
more thermal stress is placed on the recirculating diode. When driving one LED the duty cycle can be estimated
as:
D = 4.1 / 24 = 0.17
(47)
The estimated average diode current is then:
ID = (1 - 0.17) x 1.54 = 1.28A
(48)
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Design Example 1 (continued)
A 2A-rated diode will be used. To determine the proper case size, the dissipation and temperature rise in D1 can
be calculated as shown in the Design Considerations section. VD for a case size such as SMB in a 40V, 2A
Schottky diode at 1.5A is approximately 0.4V and the θJA is 75°C/W. Power dissipation and temperature rise can
be calculated as:
PD = 1.28 x 0.4 = 512 mW TRISE = 0.51 x 75 = 38°C
(49)
8.3.7 CB, CC AND CF
The bootstrap capacitor CB should always be a 22 nF ceramic capacitors with X7R dielectric. A 25V rating is
appropriate for all application circuits. The COMP pin capacitor CC and the linear regulator filter capacitor CF
should always be 100 nF ceramic capacitors, also with X7R dielectric and a 25V ratings.
8.3.8 EFFICIENCY
To estimate the electrical efficiency of this example the power dissipation in each current carrying element can
be calculated and summed. Electrical efficiency, η, should not be confused with the optical efficacy of the circuit,
which depends upon the LEDs themselves. One calculation will be detailed for three LEDs in series, where VO =
11.8V, and these calculations can be repeated for other numbers of LEDs.
Total output power, PO, is calculated as:
PO = IF x VO = 1.54 x 11.8 = 18.2W
(50)
(51)
(52)
(53)
(54)
(55)
Conduction loss, PC, in the internal MOSFET:
PC = (IF2 x RDSON) x D = (1.542 x 0.75) x 0.5 = 890 mW
Gate charging and VCC loss, PG, in the gate drive and linear regulator:
PG = (IIN-OP + fSW x QG) x VIN PG = (600 x 10-6 + 5 x 105 x 9 x 10-9) x 24 = 122 mW
Switching loss, PS, in the internal MOSFET:
PS = 0.5 x VIN x IF x (tR + tF) x fSW PS = 0.5 x 24 x 1.54 x 40 x 10-9 x 5 x 105 = 370 mW
AC rms current loss, PCIN, in the input capacitor:
PCIN = IIN(rms)2 x ESR = 0.752 0.003 = 2 mW (negligible)
DCR loss, PL, in the inductor
PL = IF2 x DCR = 1.542 x 0.06 = 142 mW
Recirculating diode loss, PD = (1 - 0.5) x 1.54 x 0.4 = 300 mW
Current Sense Resistor Loss, PSNS = 293 mW
Electrical efficiency, η = PO / (PO + Sum of all loss terms) = 18.2 / (18.2 + 2.1) = 89%
Temperature Rise in the LM3406 IC is calculated as:
TLM3406 = (PC + PG + PS) x θJA = (0.89 + 0.122 + 0.37) x 50 = 69°C
(56)
8.4 Design Example 2
The second example circuit uses the LM3406 to drive a single white LED at 1.5A ±10% with a ripple current of
20%P-P in a typical 12V automotive electrical system. The two-wire dimming function will be employed in order to
take advantage of the legacy 'theater dimming' method which dims and brightens the interior lights of
automobiles by chopping the input voltage with a 200Hz PWM signal. As with the previous example, the typical
VF of a white LED is 3.9V, and with the current sense voltage of 0.2V the total output voltage will be 4.1V. The
LED driver must operate to specifications over an input range of 9V to 16V as well as operating without suffering
damage at 28V for two minutes (the 'double battery jump-start' test) and for 300 ms at 40V (the 'load-dump' test).
The LED driver must also be able to operate without suffering damage at inputs as low as 6V to satisfy the 'cold
crank' tests. A complete bill of materials can be found in Table 2 at the end of this datasheet.
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Design Example 2 (continued)
VIN = 6V (cold-crank)
VIN = 9V to 16V (nominal)
VIN = 28V (2 minutes)
C
B
IF = 1.5A
L1
VIN = 40V (300 ms)
D1
VIN
BOOT
SW
R
ON
C
IN
D2
RON
LED1
C
O
LM3406
DIM
VOUT
CS
VINS
R
SNS
COMP
VCC
GND
C
C
C
F
Schematic for Design Example 2
8.4.1 RON and tON
A switching frequency of 450 kHz helps balance the requirements of inductor size and overall power efficiency,
but more importantly keeps the switching frequency below 530 kHz, where the AM radio band begins. This
design will concentrate on meeting the switching frequency and LED current requirements over the nominal input
range of 9V to 16V, and will then check to ensure that the transient conditions do not cause the LM3406 to
overheat. The LM3406 will allow a small shift in switching frequency when VIN changes, so the calculation for
RON is done at the typical expected condition where VIN = 13.8V and VO = 4.1V. The actual RON calculation uses
the high accuracy equation listed in the Appendix.
1
RON
=
fSW x 1 x 10-11
(57)
(58)
RON = 124 kΩ
The closest 1% tolerance resistor is 124 kΩ. The switching frequency and on-time of the circuit should be
checked at VIN-MIN and VIN-MAX which are 9V and 16V, respectively. The actual fSW and tON values have been
calculated with the high accuracy equations in the APPENDIX.
1
fSW
=
1 x 10-11 x RON
(59)
(60)
(61)
fSW(VMIN) = 463 kHz
fSW(VMAX) = 440 kHz
VO
tON = 1 x 10-11 x RON
x
VIN
(62)
(63)
(64)
tON(VMIN) = 1090 ns
tON(VMAX) = 650 ns
8.4.2 OUTPUT INDUCTOR
Since an output capacitor will be used to filter some of the LED ripple current, the inductor ripple current can be
set higher than the LED ripple current. A value of 40%P-P is typical in many buck converters:
ΔiL = 0.4 x 1.5 = 0.6AP-P
(65)
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Design Example 2 (continued)
The minimum inductance required to ensure a ripple current of 600 mAP-P or less is calculated at VIN-MAX
:
VIN - VO
x tON
L =
'iL
(66)
(67)
LMIN = [(16 – 4.1) x 6.5 x 10-7] / (0.6) = 12.9 µH
The closest standard inductor value is 15 µH. The average current rating should be greater than 1.5A to prevent
overheating in the inductor. Inductor current ripple should be calculated for VIN-MIN and VIN-MAX
:
ΔiL(VMIN) = [(9 - 4.1) x 6.5 x 10-7] / 15 x 10-6 = 357 mAP-P
(68)
(69)
ΔiL(VMAX) = [(16 - 4.1) x 1.09 x 10-6] / 15 x 10-6 = 516 mAP-P
The peak LED/inductor current is then estimated. This calculation uses the worst-case ripple current which
occurs at VIN-MAX
.
IL(PEAK) = IL + 0.5 x ΔiL(MAX)
(70)
(71)
IL(PEAK) = 1.5 + 0.5 x 0.516 = 1.76A
In order to prevent inductor saturation the inductor’s peak current rating must be above 1.8A. A 15 µH off-the
shelf inductor rated to 2.4A (peak) and 2.2A (average) with a DCR of 47 mΩ will be used.
8.4.3 USING AN OUTPUT CAPACITOR
This application does not require high frequency PWM dimming, allowing the use of an output capacitor to
reduce the size and cost of the output inductor while still meeting the 20%P-P (300 mA) target for LED ripple
current. To select the proper output capacitor the equation from Buck Regulators with Output Capacitors is re-
arranged to yield the following:
'iF
x rD
ZC
=
'iL - 'iF
(72)
The dynamic resistance, rD,of one LED can be calculated by taking the tangent line to the VF vs. IF curve in the
LED datasheet. shows an example rD calculation.
ÂIF
ÂVF
Calculating rD from the VF vs. IF Curve
Extending the tangent line to the ends of the plot yields values for ΔVF and ΔIF of 0.7V and 2000 mA,
respectively. Dynamic resistance is then:
rD = ΔVF / ΔIF = 0.5V / 2A = 0.25Ω
(73)
The most filtering (and therefore the highest output capacitance) is needed when ΔIL is highest, which occurs at
VIN-MAX. Inductor ripple current with one LED is 516 mAP-P. The required impedance of CO is calculated:
ZC = [0.3 / (0.516 - 0.3] x 0.35 = 0.35Ω
(74)
A ceramic capacitor will be used and the required capacitance is selected based on the impedance at 440 kHz:
CO = 1/(2 x π x 0.49 x 4.4 x 105) = 1.03 µF
(75)
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Design Example 2 (continued)
This calculation assumes that CO will be a ceramic capacitor, and therefore impedance due to the equivalent
series resistance (ESR) and equivalent series inductance (ESL) of of the device is negligible. The closest 10%
tolerance capacitor value is 1.5 µF. The capacitor used should have an X7R dielectric and should be rated to
50V. The high voltage rating ensures that CO will not be damaged if the LED fails open circuit and a load dump
occurs. Several manufacturers produce ceramic capacitors with these specifications in the 1206 case size. With
only 4V of DC bias a 50V rated ceramic capacitor will have better than 90% of it's rated capacitance, which is
more than enough for this design.
8.4.4 RSNS
Using the expression for RSNS
:
RSNS = 0.2 / IF
(76)
(77)
RSNS = 0.2 / 1.5 = 0.133Ω
Sub-1Ω resistors are available in both 1% and 5% tolerance. A 1%, 0.13Ω device is the closest value, and a
0.33W, 1210 size device will handle the power dissipation of 290 mW. With the resistance selected, the average
value of LED current is re-calculated to ensure that current is within the ±5% tolerance requirement. From the
expression for average LED current:
IF = 0.2 / 0.13 = 1.54A, 3% above the target current
(78)
8.4.5 INPUT CAPACITOR
Controlling input ripple current and voltage is critical in automotive applications where stringent conducted
electromagnetic interference tests are required. ΔvIN(MAX) will be limited to 300 mVP-P or less. The minimum
required capacitance is calculated for the largest tON, 1090 ns, which occurs at the minimum input voltage. Using
the equations from the Input Capacitors section:
CIN(MIN) = (1.5 x 1.09 x 10-6) / 0.3 = 5.5 µF
(79)
As with the output capacitor, this required value is low enough to use a ceramic capacitor, and again the effective
capacitance will be lower than the rated value with 16V across CIN. Reviewing plots of %C vs. DC Bias for
several capacitors reveals that a 3.3 µF, 1210-size capacitor in X7R rated to 50V loses about 22% of its rated
capacitance at 16V, hence two such caps are needed.
Input rms current is high in buck regulators, and the worst-case is when the duty cycle is 50%. Duty cycle in a
buck regulator can be estimated as D = VO / VIN, and when VIN drops to 9V the duty cycle will be nearly 50%.
IIN-RMS = 1.5 x Sqrt(0.5 x 0.5) = 750 mA
(80)
Ripple current ratings for 1210 size ceramic capacitors are typically higher than 2A, so two of them in parallel can
tolerate more than enough for this design.
8.4.6 RECIRCULATING DIODE
To survive an input voltage transient of 40V the Schottky diode must be rated to a higher voltage. The next
highest standard voltage rating is 60V. Selecting a 60V rated diode provides a large safety margin for the ringing
of the switch node and also makes cross-referencing of diodes from different vendors easier.
The next parameters to be determined are the forward current rating and case size. The lower the duty cycle the
more thermal stress is placed on the recirculating diode. When driving one LED the duty cycle can be estimated
as:
D = 4.1 / 13.8 = 0.3
(81)
The estimated average diode current is then:
ID = (1 - 0.3) x 1.54 = 1.1A
(82)
A 2A-rated diode will be used. To determine the proper case size, the dissipation and temperature rise in D1 can
be calculated as shown in the Design Considerations section. VD for a case size such as SMB in a 60V, 2A
Schottky diode at 1.5A is approximately 0.4V and the θJA is 75°C/W. Power dissipation and temperature rise can
be calculated as:
PD = 1.1 x 0.4 = 440 mW TRISE = 0.44 x 75 = 33°C
(83)
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Design Example 2 (continued)
8.4.7 CB, CC AND CF
The bootstrap capacitor CB should always be a 22 nF ceramic capacitors with X7R dielectric. A 25V rating is
appropriate for all application circuits. The COMP pin capacitor CC and the linear regulator filter capacitor CF
should always be 100 nF ceramic capacitors, also with X7R dielectric and a 25V ratings.
8.4.8 EFFICIENCY
To estimate the electrical efficiency of this example the power dissipation in each current carrying element can
be calculated and summed. One calculation will be detailed for the nominal input voltage of 13.8V, and these
calculations can be repeated for other numbers of LEDs.
Total output power, PO, is calculated as:
PO = IF x VO = 1.54 x 4.1 = 6.3W
(84)
(85)
(86)
(87)
(88)
(89)
Conduction loss, PC, in the internal MOSFET:
PC = (IF2 x RDSON) x D = (1.542 x 0.75) x 0.3 = 530 mW
Gate charging and VCC loss, PG, in the gate drive and linear regulator:
PG = (IIN-OP + fSW x QG) x VIN PG = (600 x 10-6 + 4.5 x 105 x 9 x 10-9) x 13.8 = 64 mW
Switching loss, PS, in the internal MOSFET:
PS = 0.5 x VIN x IF x (tR + tF) x fSW PS = 0.5 x 13.8 x 1.54 x 40 x 10-9 x 4.5 x 105 = 190 mW
AC rms current loss, PCIN, in the input capacitor:
PCIN = IIN(rms)2 x ESR = 0.752 0.003 = 2 mW (negligible)
DCR loss, PL, in the inductor
PL = IF2 x DCR = 1.542 x 0.05 = 120 mW
Recirculating diode loss, PD = (1 - 0.3) x 1.54 x 0.4 = 430 mW
Current Sense Resistor Loss, PSNS = 293 mW
Electrical efficiency, η = PO / (PO + Sum of all loss terms) = 6.3 / (6.3 + 1.6) = 80%
Temperature Rise in the LM3406 IC is calculated as:
TLM3406 = (PC + PG + PS) x θJA = (0.53 + 0.06 + 0.19) x 50 = 39°C
(90)
8.5 Thermal Considerations During Input Transients
The error amplifier of the LM3406 ensures that average LED current is controlled even at the transient load-
dump voltage of 40V, leaving thermal considerations as a primary design consideration during high voltage
transients. A review of the operating conditions at an input of 40V is still useful to make sure that the LM3406 die
temperature is not exceeded. Switching frequency drops to 325 kHz, the on-time drops to 350 ns, and the duty
cycle drops to 0.12. Repeating the calculations for conduction, gate charging and switching loss leads to a total
internal loss of 731 mW, and hence a die temperature rise of 37°C. The LM3406 should operate properly even if
the ambient temperature is as high a 85°C.
8.6 Layout Considerations
The performance of any switching converter depends as much upon the layout of the PCB as the component
selection. The following guidelines will help the user design a circuit with maximum rejection of outside EMI and
minimum generation of unwanted EMI.
8.6.1 COMPACT LAYOUT
Parasitic inductance can be reduced by keeping the power path components close together and keeping the
area of the loops that high currents travel small. Short, thick traces or copper pours (shapes) are best. In
particular, the switch node (where L1, D1, and the SW pin connect) should be just large enough to connect all
three components without excessive heating from the current it carries. The LM3406 family operates in two
distinct cycles whose high current paths are shown in :
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Layout Considerations (continued)
+
-
Buck Converter Current Loops
The dark grey, inner loop represents the high current path during the MOSFET on-time. The light grey, outer loop
represents the high current path during the off-time.
8.6.2 GROUND PLANE AND SHAPE ROUTING
The diagram of is also useful for analyzing the flow of continuous current vs. the flow of pulsating currents. The
circuit paths with current flow during both the on-time and off-time are considered to be continuous current, while
those that carry current during the on-time or off-time only are pulsating currents. Preference in routing should be
given to the pulsating current paths, as these are the portions of the circuit most likely to emit EMI. The ground
plane of a PCB is a conductor and return path, and it is susceptible to noise injection just as any other circuit
path. The continuous current paths on the ground net can be routed on the system ground plane with less risk of
injecting noise into other circuits. The path between the input source and the input capacitor and the path
between the recirculating diode and the LEDs/current sense resistor are examples of continuous current paths.
In contrast, the path between the recirculating diode and the input capacitor carries a large pulsating current.
This path should be routed with a short, thick shape, preferably on the component side of the PCB. Do not place
any vias near the anode of Schottky diode. Instead, multiple vias in parallel should be used right at the pad of the
input capacitor to connect the component side shapes to the ground plane. A second pulsating current loop that
is often ignored is the gate drive loop formed by the SW and BOOT pins and capacitor CB. To minimize this loop
and the EMI it generates, keep CB close to the SW and BOOT pins.
8.6.3 CURRENT SENSING
The CS pin is a high-impedance input, and the loop created by RSNS, RZ (if used), the CS pin and ground should
be made as small as possible to maximize noise rejection. RSNS should therefore be placed as close as possible
to the CS and GND pins of the IC.
8.6.4 REMOTE LED ARRAYS
In some applications the LED or LED array can be far away (several inches or more) from the LM3406 family, or
on a separate PCB connected by a wiring harness. When an output capacitor is used and the LED array is large
or separated from the rest of the converter, the output capacitor should be placed close to the LEDs to reduce
the effects of parasitic inductance on the AC impedance of the capacitor. The current sense resistor should
remain on the same PCB, close to the LM3406 family.
Remote LED arrays and high speed dimming with a parallel FET must be treated with special care. The parallel
dimming FET should be placed on the same board and/or heatsink as the LEDs to minimize the loop area
between them, as the switching of output current by the parallel FET produces a pulsating current just like the
switching action of the LM3406's internal power FET and the Schottky diode. shows the path that the inductor
current takes through the LED or through the dimming FET. To minimize the EMI from parallel FET dimming the
parasitic inductance of the loop formed by the LED and the dimming FET (where only the dark grey arrows
appear) should be reduced as much as possible. Parasitic inductance of a loop is mostly controlled by the loop
area, hence making this loop as physically small (short) as possible will reduce the inductance.
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Layout Considerations (continued)
Buck Inductor is
Continuous
Current Source
Parallel FET Dimming Current Loops
Table 1. BOM for Design Example 1
ID
U1
Part Number
LM3406
Type
LED Driver
Inductor
Size
eTSSOP-14
10 x 10 x 4.5mm
SMB
Parameters
42V, 2A
Qty
1
Vendor
NSC
TDK
L1
SLF10145T-220M1R-PF
CMSH2-40
22 µH, 1.9A, 59 mΩ
40V, 2A
1
D1
Schottky Diode
Capacitor
Capacitor
Capacitor
1
Central Semi
Vishay
Vishay
TDK
Cc, Cf
Cb
VJ0603Y104KXXAT
VJ0603Y223KXXAT
C4532X7R1H475M
0603
100 nF 10%
22 nF 10%
2
0603
1
Cin1
Cin2
1812
4.7 µF, 50V
2
Co
C2012X7R1E105M
ERJ14RQFR13V
CRCW08051433F
Capacitor
Resistor
Resistor
0805
1210
0805
1.0 µF, 25V
0.13Ω 1%
143 kΩ 1%
1
1
1
TDK
Panasonic
Vishay
Rsns
Ron
Table 2. BOM for Design Example 2
ID
U1
Part Number
LM3406
Type
LED Driver
Inductor
Size
eTSSOP-14
10 x 10 x 4.5mm
SMB
Parameters
42V, 2A
Qty
Vendor
NSC
1
1
1
2
1
2
L1
SLF10145T-150M2R2-P
CMSH2-60
15 µH, 2.2A, 47 mΩ
60V, 2A
TDK
D1
Schottky Diode
Capacitor
Capacitor
Capacitor
Central Semi
Vishay
Vishay
TDK
Cc, Cf
Cb
VJ0603Y104KXXAT
VJ0603Y223KXXAT
C3225X7R1H335M
0603
100 nF 10%
22 nF 10%
0603
Cin1
Cin2
1210
3.3 µF, 50V
Co
Rsns
Ron
Rpd
C3216X7R1H105M
ERJ14RQFR13V
CRCW08051243F
CRCW08051002F
Capacitor
Resistor
Resistor
Resistor
1206
1210
0805
0805
0.15 µF, 50V
0.13Ω 1%
1
1
1
1
TDK
Panasonic
Vishay
124 kΩ 1%
10 kΩ 1%
Vishay
Table 3. Bill of Materials for Efficiency Curves
ID
U1
Q1
D1
L1
Part Number
LM3406
Type
Buck LED Driver
N-MOSFET
Schottky Diode
Inductor
Size
eTSSOP-14
SOT23-6
SMA
Parameters
42V, 1.5A
Qty
Vendor
NSC
1
1
1
1
2
Si3458DV
60V, 2.8A
Vishay
Central Semi
TDK
CMSH2-60M
60V, 2A
VLF10045T-330M2R3
C4532X7R1H685M
10 x 10 x 4.5mm
1812
33 µH, 2.3A, 70 mΩ
6.8 µF, 50V
Cin1
Cin2
Capacitor
TDK
Co
C3216X7R1H474M
VJ0603Y104KXXAT
Capacitor
Capacitor
1206
0603
470 nF, 50V
100 nF 10%
1
2
TDK
Cf ,Cc
Vishay
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Table 3. Bill of Materials for Efficiency Curves (continued)
ID
Cb
Part Number
VJ0603Y223KXXAT
ERJ6RQFR56V
ERJ6RQFR62V
ERJ6RQFR30V
ERJ6RQFR16V
CRCW08051433F
CRCW06031002F
160-1512
Type
Size
0603
0805
0805
0805
0805
0805
0603
0.062"
Parameters
22 nF 10%
0.56Ω 1%
0.62Ω 1%
0.3Ω 1%
Qty
1
Vendor
Vishay
Capacitor
Resistor
Resistor
Resistor
Resistor
Resistor
Resistor
Terminal
R3.5
R.7
1
Panasonic
Panasonic
Panasonic
Panasonic
Vishay
1
R1
1
R1.5
Ron
0.16Ω 1%
143kΩ 1%
10 kΩ 1%
1
1
Rpd Rout
2
Vishay
OFF*
DIM1
DIM2
3
Cambion
VIN GND
CS/LED-
Vo/LED+
160-1026
Terminal
0.094"
2
Cambion
8.7 APPENDIX
The following expressions provide the best accuracy for users who wish to create computer-based simulations or
circuit calculators:
9.92 x 10-12 x (VO + 0.65) x RON
+ 1.75 x 10-7
tON
RON
fSW
=
VIN ± 1.5
(91)
(92)
(93)
(D ± fSW x 1.75 x 10-7) x (VIN ± 1.5)
9.92 x 10-12 x fSW x (VO + 0.65)
=
D x (VIN ± 1.5)
=
9.92 x 10-12 x (VO + 0.65) x R
ON + 1.75 x 10-7 x (VIN ± 1.5)
Copyright © 2008–2014, Texas Instruments Incorporated
Submit Documentation Feedback
31
Product Folder Links: LM3406 LM3406HV LM3406HV-Q1
LM3406, LM3406HV, LM3406HV-Q1
SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014
www.ti.com
9 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision D (April 2013) to Revision E
Page
•
Added availability of LM3406HV-Q1, the automotive grade device throughout the data sheet. ........................................... 1
Changes from Revision C (May 2013) to Revision D
Page
•
Changed layout of National Data Sheet to TI format .......................................................................................................... 31
32
Submit Documentation Feedback
Copyright © 2008–2014, Texas Instruments Incorporated
Product Folder Links: LM3406 LM3406HV LM3406HV-Q1
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status Package Type Package Pins Package
Eco Plan
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
Samples
Drawing
Qty
(1)
(2)
(3)
(4/5)
(6)
LM3406HVMH/NOPB
LM3406HVMHX/NOPB
LM3406HVQMHQ1
LM3406HVQMHXQ1
LM3406MH/NOPB
ACTIVE
HTSSOP
HTSSOP
HTSSOP
HTSSOP
HTSSOP
HTSSOP
PWP
14
14
14
14
14
14
94
RoHS & Green
SN
Level-1-260C-UNLIM
Level-1-260C-UNLIM
Level-1-260C-UNLIM
Level-1-260C-UNLIM
Level-1-260C-UNLIM
Level-1-260C-UNLIM
-40 to 125
-40 to 125
-40 to 150
-40 to 150
-40 to 125
-40 to 125
LM3406
HVMH
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
PWP
2500 RoHS & Green
94 RoHS & Green
2500 RoHS & Green
94 RoHS & Green
2500 RoHS & Green
SN
SN
SN
SN
SN
LM3406
HVMH
PWP
LM3406Q
HVMH
PWP
LM3406Q
HVMH
PWP
LM3406
MH
LM3406MHX/NOPB
PWP
LM3406
MH
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based
flame retardants must also meet the <=1000ppm threshold requirement.
(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
Addendum-Page 1
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
(6)
Lead finish/Ball material - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead finish/Ball material values may wrap to two
lines if the finish value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
OTHER QUALIFIED VERSIONS OF LM3406HV, LM3406HV-Q1 :
Catalog: LM3406HV
•
Automotive: LM3406HV-Q1
•
NOTE: Qualified Version Definitions:
Catalog - TI's standard catalog product
•
Automotive - Q100 devices qualified for high-reliability automotive applications targeting zero defects
•
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
5-Jan-2022
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
B0
K0
P1
W
Pin1
Diameter Width (mm) (mm) (mm) (mm) (mm) Quadrant
(mm) W1 (mm)
LM3406HVMHX/NOPB HTSSOP PWP
14
14
14
2500
2500
2500
330.0
330.0
330.0
12.4
12.4
12.4
6.95
6.95
6.95
5.6
5.6
5.6
1.6
1.6
1.6
8.0
8.0
8.0
12.0
12.0
12.0
Q1
Q1
Q1
LM3406HVQMHXQ1
LM3406MHX/NOPB
HTSSOP PWP
HTSSOP PWP
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
5-Jan-2022
*All dimensions are nominal
Device
Package Type Package Drawing Pins
SPQ
Length (mm) Width (mm) Height (mm)
LM3406HVMHX/NOPB
LM3406HVQMHXQ1
LM3406MHX/NOPB
HTSSOP
HTSSOP
HTSSOP
PWP
PWP
PWP
14
14
14
2500
2500
2500
367.0
367.0
367.0
367.0
367.0
367.0
35.0
35.0
35.0
Pack Materials-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
5-Jan-2022
TUBE
*All dimensions are nominal
Device
Package Name Package Type
Pins
SPQ
L (mm)
W (mm)
T (µm)
B (mm)
LM3406HVMH/NOPB
LM3406HVQMHQ1
LM3406MH/NOPB
PWP
PWP
PWP
HTSSOP
HTSSOP
HTSSOP
14
14
14
94
94
94
495
495
495
8
8
8
2514.6
2514.6
2514.6
4.06
4.06
4.06
Pack Materials-Page 3
PACKAGE OUTLINE
PWP0014A
PowerPADTM TSSOP - 1.2 mm max height
S
C
A
L
E
2
.
4
0
0
PLASTIC SMALL OUTLINE
C
6.6
6.2
TYP
SEATING PLANE
PIN 1 ID
AREA
A
0.1 C
12X 0.65
14
1
2X
5.1
4.9
3.9
NOTE 3
7
8
0.30
14X
0.19
4.5
4.3
B
0.1
C A B
SEE DETAIL A
(0.15) TYP
4X (0.2)
NOTE 5
4X (0.05)
NOTE 5
8
7
THERMAL
PAD
0.25
GAGE PLANE
3.255
3.205
15
1.2 MAX
0.15
0.05
0 - 8
14
1
0.75
0.50
DETAIL A
(1)
TYPICAL
3.155
3.105
4214867/A 09/2016
PowerPAD is a trademark of Texas Instruments.
NOTES:
1. All linear dimensions are in millimeters. Any dimensions in parenthesis are for reference only. Dimensioning and tolerancing
per ASME Y14.5M.
2. This drawing is subject to change without notice.
3. This dimension does not include mold flash, protrusions, or gate burrs. Mold flash, protrusions, or gate burrs shall not
exceed 0.15 mm per side.
4. Reference JEDEC registration MO-153.
5. Features may differ and may not be present.
www.ti.com
EXAMPLE BOARD LAYOUT
PWP0014A
PowerPADTM TSSOP - 1.2 mm max height
PLASTIC SMALL OUTLINE
(3.4)
NOTE 9
(3.155)
SYMM
SOLDER MASK
DEFINED PAD
SEE DETAILS
14X (1.5)
1
14
14X (0.45)
(1.1)
TYP
15
SYMM
(3.255)
(5)
NOTE 9
12X (0.65)
8
7
(
0.2) TYP
VIA
(R0.05) TYP
(1.1) TYP
METAL COVERED
BY SOLDER MASK
(5.8)
LAND PATTERN EXAMPLE
SCALE:10X
METAL UNDER
SOLDER MASK
SOLDER MASK
OPENING
SOLDER MASK
OPENING
METAL
0.05 MIN
ALL AROUND
0.05 MAX
ALL AROUND
SOLDER MASK
DEFINED
NON SOLDER MASK
DEFINED
SOLDER MASK DETAILS
PADS 1-14
4214867/A 09/2016
NOTES: (continued)
6. Publication IPC-7351 may have alternate designs.
7. Solder mask tolerances between and around signal pads can vary based on board fabrication site.
8. This package is designed to be soldered to a thermal pad on the board. For more information, see Texas Instruments literature
numbers SLMA002 (www.ti.com/lit/slma002) and SLMA004 (www.ti.com/lit/slma004).
9. Size of metal pad may vary due to creepage requirement.
www.ti.com
EXAMPLE STENCIL DESIGN
PWP0014A
PowerPADTM TSSOP - 1.2 mm max height
PLASTIC SMALL OUTLINE
(3.155)
BASED ON
0.125 THICK
STENCIL
14X (1.5)
(R0.05) TYP
1
14
14X (0.45)
15
(3.255)
BASED ON
0.125 THICK
STENCIL
SYMM
12X (0.65)
8
7
SEE TABLE FOR
METAL COVERED
BY SOLDER MASK
SYMM
(5.8)
DIFFERENT OPENINGS
FOR OTHER STENCIL
THICKNESSES
SOLDER PASTE EXAMPLE
EXPOSED PAD
100% PRINTED SOLDER COVERAGE BY AREA
SCALE:10X
STENCIL
THICKNESS
SOLDER STENCIL
OPENING
0.1
3.53 X 3.64
3.155 X 3.255 (SHOWN)
2.88 X 2.97
0.125
0.15
0.175
2.67 X 2.75
4214867/A 09/2016
NOTES: (continued)
10. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate
design recommendations.
11. Board assembly site may have different recommendations for stencil design.
www.ti.com
IMPORTANT NOTICE AND DISCLAIMER
TI PROVIDES TECHNICAL AND RELIABILITY DATA (INCLUDING DATA SHEETS), DESIGN RESOURCES (INCLUDING REFERENCE
DESIGNS), APPLICATION OR OTHER DESIGN ADVICE, WEB TOOLS, SAFETY INFORMATION, AND OTHER RESOURCES “AS IS”
AND WITH ALL FAULTS, AND DISCLAIMS ALL WARRANTIES, EXPRESS AND IMPLIED, INCLUDING WITHOUT LIMITATION ANY
IMPLIED WARRANTIES OF MERCHANTABILITY, FITNESS FOR A PARTICULAR PURPOSE OR NON-INFRINGEMENT OF THIRD
PARTY INTELLECTUAL PROPERTY RIGHTS.
These resources are intended for skilled developers designing with TI products. You are solely responsible for (1) selecting the appropriate
TI products for your application, (2) designing, validating and testing your application, and (3) ensuring your application meets applicable
standards, and any other safety, security, regulatory or other requirements.
These resources are subject to change without notice. TI grants you permission to use these resources only for development of an
application that uses the TI products described in the resource. Other reproduction and display of these resources is prohibited. No license
is granted to any other TI intellectual property right or to any third party intellectual property right. TI disclaims responsibility for, and you
will fully indemnify TI and its representatives against, any claims, damages, costs, losses, and liabilities arising out of your use of these
resources.
TI’s products are provided subject to TI’s Terms of Sale or other applicable terms available either on ti.com or provided in conjunction with
such TI products. TI’s provision of these resources does not expand or otherwise alter TI’s applicable warranties or warranty disclaimers for
TI products.
TI objects to and rejects any additional or different terms you may have proposed. IMPORTANT NOTICE
Mailing Address: Texas Instruments, Post Office Box 655303, Dallas, Texas 75265
Copyright © 2022, Texas Instruments Incorporated
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