LM3431AQMH/NOPB_技术文档

LM3431

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SNVS547G –NOVEMBER 2007–REVISED MAY 2013

3-Channel Constant Current LED Driver with Integrated Boost Controller

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1

FEATURES

DESCRIPTION

The LM3431 is a 3-channel linear current controller

combined with a boost switching controller ideal for

driving LED backlight panels in space critical

applications. The LM3431 drives 3 external NPN

transistors or MOSFETs to deliver high accuracy

constant current to 3 LED strings. Output current is

adjustable to drive strings in excess of 200 mA. The

LM3431 can be expanded to drive as many as 6 LED

strings.

2

•

LM3431Q/LM3431AQ are Automotive Grade

Products that are AEC-Q100 Grade 1 Qualified

(–40°C to 125°C operating junction

temperature)

•

3-Channel Programmable LED Current

High Accuracy Linear Current Regulation

Analog and Digital PWM Dimming Control

Up to 25kHz Dimming Frequency

>100:1 Contrast Ratio

The boost controller drives an external NFET switch

for step-up regulation from input voltages between 5V

and 36V. The LM3431 features LED cathode

feedback to minimize regulator headroom and

optimize efficiency.

Integrated Boost Controller

5V-36V Input Voltage Range

Adjustable Switching Frequency up to 1MHz

LED Short and Open Protection

A DIM input pin controls LED brightness from analog

or digital control signals. Dimming frequencies up to

25 kHz are possible with a contrast ratio of 100:1.

Contrast ratios greater than 1000:1 are possible at

lower dimming frequencies.

Selectable Fault Shutdown or Automatic

Restart

•

Programmable Fault Delay

Programmable Cycle by Cycle Current Limit

Output Over Voltage Protection

No Audible Noise

The LM3431 eliminates audible noise problems by

maintaining constant output voltage regulation during

LED dimming. Additional features include LED short

and open protection, fault delay/error flag, cycle by

cycle current limit, and thermal shutdown for both the

IC and LED array. The enhanced LM3431A features

reduced offset voltage for higher accuracy LED

current.

Enable Pin

LED Over-Temperature Shutdown Input

Thermal Shutdown

TSSOP-28 Exposed Pad Package

APPLICATIONS

•

Automotive Infotainment Displays

Small to Medium Format Displays

TYPICAL APPLICATION CIRCUIT

Vin: 5V to 36V

+

VIN

EN

LG

CS

VCC

MODE/F

DIM

LEDOFF

ILIM

PGND

THM

PWM Dim

input

LM3431

AFB

SC

CFB

NDRV1

SNS1

NDRV2

SNS2

NDRV3

SNS3

REF

REFIN

COMP

FF

VCC

RT

SS/SH

DLY

SGND

Ch.2/

Ch.3

Ch.2

Ch.3

1

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of

Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

2

All trademarks are the property of their respective owners.

PRODUCTION DATA information is current as of publication date.

Products conform to specifications per the terms of the Texas

Instruments standard warranty. Production processing does not

necessarily include testing of all parameters.

LM3431

SNVS547G –NOVEMBER 2007–REVISED MAY 2013

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CONNECTION DIAGRAM

VIN

PGND

VCC

1

2

3

28 EN

27 DIM

26 THM

25 NDRV1

24 SNS1

LG

CS

4

5

6

7

23 NDRV2

22 SNS2

ILIM

MODE/F

21 NDRV3

20 SNS3

FF

8

9

RT

REF 10

REFIN 11

COMP 12

SGND 13

AFB 14

19 LEDOFF

18 SC

17 CFB

16 DLY

15 SS/SH

Exposed Pad

Connect to SGND

Figure 1. 28 Lead Plastic Exposed Pad TSSOP

Top View

See Package Number PWP0028A

PIN DESCRIPTIONS

Pin No.

Pin Name

VIN

Description

1

2

3

4

5

6

Power supply input.

PGND

VCC

LG

Power ground pin. Connect to ground.

Internal reference voltage output. Bypass to PGND with a minimum 4.7 µF capacitor.

Boost controller gate drive output. Connect to the NFET gate.

CS

Boost controller current sense pin. Connect to the top side of the boost current sense resistor.

ILIM

Boost controller current limit adjust pin. Connect a resistor from this pin to the Boost current sense resistor to set

the current limit threshold.

7

8

9

MODE/F

FF

Dimming mode selection pin. Pull high for digital PWM control. Or connect to a capacitor to GND to set the

internal dimming frequency.

Feedforward pin. Connect to a resistor to ground to control the output voltage over/undershoot during PWM

dimming.

RT

Frequency adjust pin. Connect a resistor from this pin to ground to set the operating frequency of the boost

controller.

10

11

12

13

14

REF

REFIN

COMP

SGND

AFB

Reference voltage. Use this pin to provide the REFIN voltage.

This pin sets the LED current feedback voltage. Connect to a resistor divider from the REF pin.

Output of the error amplifier. Connect to the compensation network.

Signal ground pin. Connect to ground.

Anode feedback pin. The boost controller voltage feedback during LED off time. Connect this pin to a resistor

divider from the output voltage.

15

16

17

SS/SH

DLY

Soft-start and sample-hold pin. Connect a capacitor from this pin to ground to set the soft-start time.

Fault delay pin. Connect a capacitor from this pin to ground to set the delay time for shutdown.

CFB

Cathode feedback pin. The boost controller voltage feedback. Connect through a diode to the bottom cathode of

each LED string.

18

19

SC

LED short circuit detection pin. Connect through a diode to the bottom cathode of each string.

LEDOFF

A dual function pin. The LEDOFF signal controls external drivers during PWM dimming. Or connect to ground to

enable automatic fault restart.

20

21

22

23

SNS3

NDRV3

SNS2

Current feedback for channel 3. Connect to the top of the channel 3 current sense resistor.

Base drive for the channel 3 current regulator. Connect to the NPN base or NFET gate.

Current feedback for channel 2.

NDRV2

Base drive for the channel 2 current regulator.

2

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PIN DESCRIPTIONS (continued)

Pin No.

24

Pin Name

SNS1

NDRV1

THM

Description

Current feedback for channel 1.

Base drive for the channel 1 current regulator.

25

26

LED thermal monitor input pin. When pulled below 1.2V, device enters standby mode.

PWM dimming input pin. Accepts a digital PWM or analog voltage level input to control LED current duty cycle.

Enable pin. Connect to VIN through a resistor divider to set an external UVLO threshold. Pull low to shutdown.

Exposed pad. Connect to SGND.

27

DIM

28

EN

EP

These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam

during storage or handling to prevent electrostatic damage to the MOS gates.

(1)

ABSOLUTE MAXIMUM RATINGS

If Military/Aerospace specified devices are required, contact the Texas Instruments Semiconductor Sales Office/

Distributors for availability and specifications.

VALUE / UNIT

VIN

–0.3V to 37V

–0.3V to 10V

–0.3V to 7V

-0.3V to 40V

–0.3V to 7V

–65°C to +150°C

20sec, 240°C

75sec, 219°C

2 kV

EN

DIM

MODE/F

REFIN

THM

DLY

Voltages from the indicated pins to SGND:

SNSx

NDRVx

CFB

SC

AFB

CS

VCC

Storage Temperature

Infrared

Soldering Dwell Time, Temperature

ESD Rating Human Body Model⁽²⁾

Vapor Phase

(1) Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which

operation of the device is intended to be functional. For ensured specifications and test conditions, see the ELECTRICAL

CHARACTERISTICS.

(2) The human body model is a 100pF capacitor discharged through a 1.5 kΩ resistor into each pin.

(1)

RECOMMENDED OPERATING CONDITIONS

VALUE / UNIT

VIN

4.5V to 36V

-40°C to +125°C

32°C/W

Junction Temperature Range

Thermal Resistance (θ_JA

Power Dissipation ⁽³⁾, TSSOP-28

)

⁽²⁾, TSSOP-28 (0.5W)

3.1W

(1) Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which

operation of the device is intended to be functional. For specifications and test conditions, see the ELECTRICAL CHARACTERISTICS.

(2) The Thermal Resistance specifications are based on a JEDEC standard 4-layer pcb. θ_JAwill vary with board size and copper area.

(3) The maximum allowable power dissipation is a function of the maximum junction temperature, T_{J_MAX}, the junction-to-ambient thermal

resistance, θ_JA, and the ambient temperature, T_A. The maximum allowable power dissipation at any ambient temperature is calculated

using: P_{D_MAX}= (T_{J_MAX}– T_A)/θ_JA. The maximum power dissipation is determined using T_A= 25°C, and T_{J_MAX}= 125°C.

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SNVS547G –NOVEMBER 2007–REVISED MAY 2013

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ELECTRICAL CHARACTERISTICS

Specifications in standard type are for T_J= 25°C only, and limits in boldface type apply over the junction temperature (T_J)

range of -40°C to +125°C. Unless otherwise stated, V_IN= 12V. Minimum and Maximum limits are specified through test,

design, or statistical correlation. Typical values represent the most likely parametric norm at T_J= 25°C, and are provided for

(1)

reference purposes only.

Parameter

Test Conditions

Min

Typ

Max

4.85

Units

SYSTEM

I_Q

(2)

Operating VIN Current

DIM = 5V

EN = 1V

4.0

3.7

15

mA

µA

V

I_{Q_SB}

Standby mode VIN current

Shutdown mode VIN Current

VCC voltage

I_{Q_SD}

V_CC

EN = 0V, Vin = 36V

23

Iload = 25 mA, Vin = 5.5 to 36V

4.80

5

5.24

VCC_ILIM

UVLO

VCC current limit

72

mA

V

UVLO threshold

VIN rising, measured at VCC

4.36

0.28

0.75

4.50

hysteresis

V

V_{EN_ST}

V_EN

Enable pin Standby threshold

Enable pin On threshold

hysteresis

EN rising

V

1.185 1.230 1.275

V

115

165

mV

LINEAR CURRENT CONTROLLER

V_REF

Reference Voltage

IREF < 300 µA

2.45

2.5

14

2.55

80

V

I_REFIN

REFIN input bias current

REFIN = 300 mV

5.5V < VIN < 36V

nA

ΔV_REF/ ΔV_INLine regulation

0.000

1

%/V

V_NDRV

NDRVx drive voltage capability

INDRVx = 5 mA

3.7

6

V

I_{NDRV_SK}

I_{NDRV_SC}

I_SNS

NDRVx drive sink current

NDRV_X= 0.9V

4

8

20

30

+5

+3

5.5

6

mA

µA

NDRVx drive source current

SNSx input bias current

NDRV_X= 0.9V

10

15

20

SNSx = 300 mV

V_OS

SNSx amp offset voltage

REFIN = 300 mV (LM3431)

REFIN = 300 mV (LM3431A)

REFIN = 300 mV, 25°C (LM3431)

-5

-3

mV

V_OS

SNSx amp offset voltage

(3)

V_{OS_DELTA}

Ch. To Ch. offset voltage mismatch

REFIN = 300 mV, -40°C to +125°C

(LM3431)

(3)

V_{OS_DELTA}

Ch. To Ch. offset voltage mismatch

REFIN = 300 mV, 25°C (LM3431A)

3.5

mV

REFIN = 300 mV, -40°C to +125°C

(LM3431A)

4

bw

SNSx amp bandwidth

LEDOFF voltage

DIM threshold

At unity gain

DIM low

2

5

MHz

V

V_LEDOFF

V_DIM

MODE/F > 4V

1.9

0.8

0.4

100

90

2.3

V

hysteresis

V

(4)

T_DIM

Minimum internal DIM pulse width

DIM to NDRV delay time

MODE/F threshold

µs

ns

V

DIM_{DLY_R}

DIM_{DLY_F}

TH_MODE/F

I_MODE/F

DIM rising

DIM falling

For Digital Dimming control

3.8

40

MODE/F source/sink current

MODE/F minimum voltage

MODE/F peak voltage

uA

V

V_{MODE_L}

V_{MODE_H}

Analog dimming mode

0.37

2.5

V

(1) All room temperature limits are 100% production tested. All limits at temperature extremes are specified through correlation using

standard Statistical Quality Control (SQC) methods. All limits are used to calculate Average Outgoing Quality Level (AOQL).

(2) IQ specifies the current into the VIN pin and applies to non-switching operation.

(3) V_{OS_DELTA}specifies the maximum absolute difference between the offset of any pair of SNS amplifiers.

(4) The minimum DIM pulse width is an internal signal. Any pulse width may be applied to the DIM pin or generated via analog dimming

mode. A pulse width less than 0.4 µs will be internally extended to 0.4 µs.

4

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SNVS547G –NOVEMBER 2007–REVISED MAY 2013

ELECTRICAL CHARACTERISTICS (continued)

Specifications in standard type are for T_J= 25°C only, and limits in boldface type apply over the junction temperature (T_J)

range of -40°C to +125°C. Unless otherwise stated, V_IN= 12V. Minimum and Maximum limits are specified through test,

design, or statistical correlation. Typical values represent the most likely parametric norm at T_J= 25°C, and are provided for

reference purposes only. ⁽¹⁾

Parameter

Test Conditions

Min

Typ

Max

Units

PROTECTION

V_{SC_SHORT}

V_{SC_OPEN}

I_{DLY_SC}

I_{DLY_SK}

V_DLY

SC high threshold

LED short circuit fault, SC rising

LED open circuit fault, SC rising

DLY = 1.0V

5.7

3.16

39

6

6.2

3.87

73

V

SC open clamp voltage

DLY source current

3.50

57

µA

V

DLY sink current

DLY = 1.0V

1.8

2.8

350

1.6

1.23

9.6

40

DLY threshold voltage

DLY reset threshold voltage

DLY blank time

DLY rising

2.40

1.19

3.16

1.27

V_{DLY_reset}

T_{DLY_BLK}

V_THM

DLY falling

mV

µs

V

DIM rising

THM threshold

I_THM

THM hysteresis current

ILIM max source current

AFB overvoltage threshold

AFB undervoltage threshold

Thermal shutdown threshold

THM = 1V

µA

V

I_ILIM

COMP = 2.0V

31

46

V_{AFB_max}

V_{AFB_UVP}

T_SD

1.87

0.73

2.0

0.85

160

2.22

0.98

AFB falling

V

°C

BOOST CONTROLLER

V_CFBCFB voltage

I_CFB

DIM high

1.60

35

1.71

50

1.82

65

V

µA

CFB source current

CFB_TC

CFB temperature coefficient

-2.6

0.001

19

mV/°C

%/V

µA

ΔV_CFB/ ΔV_INCFB Line regulation

5.5V < VIN < 36V

At EN going high

At end of soft-start cycle

R_RT= 34.8 kΩ

I_SS/SH

V_{SS_END}

V_RT

SS/SH source current

SS/SH voltage

13

24

1.80

1.85

1.22

700

200

1000

170

85

1.90

V

RT voltage

V

F_SW

Switching Frequency

Minimum Switching Frequency

Maximum Switching Frequency

Minimum on time

R_RT= 34.8 kΩ

651

180

900

749

220

kHz

R_RT= 130 kΩ

R_RT= 22.6 kΩ

1100

230

T_{on_min}

D_MAX

ns

%

Maximum duty cycle

80

ILIM_gm

Vslope

I_{COMP_SC}

I_{COMP_SK}

EA_gm

ILIM amplifier transconductance

Slope compensation

COMP to ILIM gain

85

umho

mV

µA

umho

Ω

Peak voltage per cycle

75

COMP source current

COMP sink current

VCOMP = 1.2V, AFB = 0.5V

VCOMP =1.2V, AFB = 1.5V

CFB to COMP gain, DIM high

Source Current = 200 mA, VIN = 5.5V

Sink Current = 200 mA

155

150

230

6.4

Error amplifier transconductance

Gate Drive On Resistance

R_LG

2.2

Ω

I_LG

Driver Output Current

Source, LG = 2.5V, VIN = 5.5V

Sink, LG = 2.5V

0.35

0.70

A

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TYPICAL PERFORMANCE CHARACTERISTICS

Unless otherwise specified the following conditions apply: V_IN= 12V, T_J= 25°C.

V_REFvs. Temperature

C_FBVoltage vs. Temperature

2.53

2.52

2.51

2.50

2.49

2.48

2.47

2.46

2.50

2.25

2.00

1.75

1.50

1.25

1.00

-40 -20

0

20 40 60 80 100 120 140

-40 -20

0

20 40 60 80 100 120 140

TEMPERATURE (°C)

Figure 2.

Figure 3.

SNS 1, 2, 3 V_OSvs Temperature (LM3431 or LM3431A)

Delta V_OSMax vs Temperature (LM3431 or LM3431A)

1.5

3.0

1.0

0.5

0

2.5

2.0

1.5

1.0

-0.5

-1.0

-1.5

0.5

0

-40 -20

0

20 40 60 80 100 120 140

-40 -20

0

20 40 60 80 100 120 140

TEMPERATURE (°C)

Figure 4.

Figure 5.

IQ_SB vs Temperature

IQ_SD vs Temperature

16.0

3.80

15.5

15.0

14.5

14.0

13.5

13.0

3.70

3.60

3.50

3.40

3.30

-40 -20

0

20 40 60 80 100 120 140

-40 -20

0

20 40 60 80 100 120 140

TEMPERATURE (°C)

Figure 6.

Figure 7.

6

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TYPICAL PERFORMANCE CHARACTERISTICS (continued)

Unless otherwise specified the following conditions apply: V_IN= 12V, T_J= 25°C.

Efficiency vs. Input Voltage LED Current = 140 mA x 3, LED

Normalized Switching Frequency vs. Temperature (700 kHz)

Vf = 25V

1.04

100

100%

dimming duty

1.02

1.00

0.98

0.96

0.94

90

50%

80

10%

70

60

50

5

9

13

17

21

-40 -20

0

20 40 60 80 100 120 140

TEMPERATURE (°C)

INPUT VOLTAGE (V)

Figure 8.

Figure 9.

Line Transient Response

Dimming Transient Response

Vout

100 mV/Div

ILED

5 mA/Div

Vout

200 mV/Div

Vsw

20V/Div

VIN

5V/Div

ILED

200 mA/Div

400 ms/DIV

200 ms/DIV

Figure 10.

Figure 11.

LED Ripple Current

NDRV Waveforms

ILED

5 mA/Div

ILED

50 mA/Div

Vcathode

100 mV/Div

NDRV (FET)

2V/Div

REFIN

10 mV/Div

NDRV (NPN)

1V/Div

DIM

5V/Div

400 ns/DIV

1 ms/DIV

Figure 12.

Figure 13.

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BLOCK DIAGRAM

SC

DLY

FF

THM

3.8V

60 uA

CFB

LED on

open

short

-

+

LED on

override

FF step scaling

+

-

VIN

blank

time

ovp/uvp

CFB fault

6V

CS

2 uA

vin

Linear Reg

LED

ref

ilimit

vcc

on

ff

+

-

ramp

fault T1, T2

+

-

2.8V

R

S

Q

VCC

LG

I limit

-

latch off/restart

+

and COMP

clamp

set_clk

logic

fault T3

uvlo

PGND

+

-

ILIM

Restart Mode

sd

ff

TSD

EN

RT

+

-

COMP

EA

uvp

ovp

ramp

A

B

+

-

CFB

AFB

+

-

AFB sense

set_clk

4V

A

1.7V

-

+

MODE/F

B

LED on = A

SS/SH

-

+

fault

CFB check

ss

ref

DIM

SS/SH logic

Restart Mode

CFB fault

SGND

LED on

LEDOFF

REFIN

SNS3

NDRV3

SNS2

NDRV2

SNS1

NDRV1

REF

8

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OPERATION DESCRIPTION

The LM3431 combines a boost controller and 3 constant current regulator controllers in one device. To simplify

the description, these two blocks will be described separately as Boost Controller and LED Current Regulator. All

descriptions and component numbers refer to the Figure 14 schematic. The LED bottom cathode nodes (VC1 –

VC4) are referred to simply as the cathode.

BOOST CONTROLLER

The LM3431 is a current-mode, PWM boost controller. Although the LM3431 may be operated in either

continuous or discontinuous conduction mode, the following guidelines are designed for continuous conduction

operation. This mode of operation gives lower output ripple and better LED current regulation.

In continuous conduction mode (when the inductor current never reaches zero), the boost regulator operates in

two cycles. In the first cycle of operation, the NFET is turned on and current ramps up and is storing energy in

the inductor. During this cycle, diode D1 is reverse biased and load current is supplied by the output capacitors -

C8 and C9 in Figure 14.

In the second cycle, the NFET is off and the diode is forward biased. Inductor current is transferred to the load

and output capacitor. The ratio of these two cycles determines the output voltage and is expressed as D or D’:

V_IN

D‘ = 1-D =

V_OUT

(1)

where D is the duty cycle of the switch.

V_IN

D = 1 -

V_OUT

(2)

Maximum duty cycle is limited to 85% typically.

As input voltage approaches the nominal output voltage, duty cycle and switch on time are reduced. When the

on time reaches minimum, pulse skipping will occur. This increases the output ripple voltage and can cause

regulator saturation and poor LED current regulation. If input voltage equals or exceeds the set output voltage,

switching will stop and the output voltage will become unregulated. This will force an increase in the LED

cathode voltage and NPN regulator power dissipation. Although this condition can be tolerated, it is not

recommended.

Therefore, input voltage should be restricted to keep on time above the minimum (see Switching Frequency

section) and at least 1V below the set output voltage.

ENABLE and UVLO

The EN pin is a dual function pin combining both enable and programmable undervoltage lockout (UVLO). The

shutdown threshold is 0.75V. When EN is pulled below this threshold, the LM3431 will shutdown and IQ will be

reduced to 15 µA typically. The typical EN pin UVLO threshold is 1.23V. When the EN voltage is above this

threshold, the LM3431 will begin softstart. Below the UVLO threshold, the LM3431 will remain in standby mode.

A resistor divider, shown as R1 and R2 in Figure 14, can be used to program the UVLO threshold at the EN pin.

This feature is used to shutdown the IC at an input voltage higher than the internal VCC UVLO threshold of 4.4V.

The EN UVLO should be set just below the minimum input voltage for the application.

The internal UVLO is monitored at the VCC pin. When VCC is below the threshold of 4.4V, the LM3431 is in

standby mode (See VCC section).

10

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Soft-Start

The SS/SH pin is a dual function softstart and sample/hold pin. The SH function is described in sections below.

When the EN pin is pulled above the programmable UVLO threshold and VCC rises above the internal UVLO

threshold, the SS/SH pin begins sourcing current. This charges the SS cap (C6) and the SS pin voltage in turn

controls the output voltage ramp-up, sensed via the AFB pin. The softstart capacitor is calculated as shown

below, where 20 µA is the typical softstart source current:

t_ssx 19 mA

1.85V

C6 =

(3)

The LED current regulators are held off until softstart is completed. During softstart, current limit is active and the

CFB pin is monitored for a cathode short fault (See LED Protection section). When the SS/SH voltage reaches

1.85V, the current regulators are activated, LED current begins flowing, and output voltage control is transferred

to the CFB pin. Typical startup is shown below in Figure 15.

ILED

100 mA/Div

Vout

10V/Div

VSS/SH

1V/Div

VEN

5V/Div

4 ms/DIV

Figure 15. Typical Startup Waveforms (From power-on, DIM = high)

Output Voltage, OVP, and SH

The LM3431 boost controls the LED cathode voltage in order to drive the LED strings with sufficient headroom at

optimum efficiency. When the LED strings are on, voltage is regulated to 1.7V (typical) at the CFB pin, which is

one diode Vf above the LED cathode voltage. Therefore, when the LED strings are on, the output voltage (LED

anodes, shown as VA in Figure 14) will vary according to the Vf of the LED string, while the LED cathode voltage

will be regulated via the CFB pin.

The AFB pin is used to regulate the output voltage when the LED strings are off, which is during startup and

dimming off cycles. During LED-off times, the cathode voltage is not regulated.

AFB should set the initial output voltage to at least 1.0V (CFB voltage minus one diode drop) above the

maximum LED string forward voltage. This ensures that there is enough headroom to drive the LED strings at

startup and keeps the SS/SH voltage below its maximum. The AFB pin voltage at the end of softstart is 1.85V

typically, which determines the ratio of the feedback resistors according to the following equation:

V_OUT(MAX)- 1.85V

R19 = R18 x =

1.85V

(4)

The AFB resistors also set the output over-voltage (OVP) threshold. The OVP threshold is monitored during both

LED on and LED off states and protects against any over voltage condition, including all LEDs open (See Open

LED section).The OVP threshold at Vout can be calculated as follows:

2.0 x (R19+ R18)

V_OVP

=

R18

(5)

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Because OVP has a fixed 2.0V threshold sensed at AFB, a larger value for R19 will increase the OVP threshold

of the output voltage. During an open LED fault, the output voltage will increase by 2.6V typically (see LED

Protection section). Therefore, at least this much headroom above the nominal output voltage is required to avoid

a false OVP error. Note that because of the high output voltage setting at the end of softstart, a brief open LED

error may occur during the short time it takes for the cathode voltage to drop to its nominal level. Figure 15

shows a typical startup waveform, where both Vout and the SS/SH voltage reach their peak before the LED

current turns on. Once LED current starts, SS/SH and Vout drop to the nominal operating point.

While the LEDs are on, the AFB voltage is sampled to the SS/SH pin. During LED-off time, this SS/SH voltage is

used as the reference voltage to regulate the output. This allows the output voltage to remain stable between on

and off dimming cycles, even though there may be wide variation in the LED string forward voltage. The SS/SH

pin has a maximum voltage of 1.9V. Therefore, the AFB voltage when the LEDs are on must be below this limit

for proper regulation. This will be ensured by setting the AFB resistors as described above. During LED-off

cycles, there is minimal loading on the output, which forces the boost controller into pulse skipping mode. In this

mode, switching is stopped completely, or for multiple cycles until the AFB feedback voltage falls below the

SS/SH reference level.

Switching Frequency

The switching frequency can be set between 200 kHz and 1 MHz with a resistor from the RT pin to ground. The

frequency setting resistor (R6 in Figure 14) can be determined according to the following empirically derived

equation:

-1.06

R_T= 35403 x f_SW

Where

•

f_SWis in kHz

the R_Tresult is in kohm.

(6)

1200

1000

800

600

400

200

0

20 40 60 80 100 120 140 160

(kW)

R

T

Figure 16. Switching Frequency vs R_T

For a given application, the maximum switching frequency is limited by the minimum on time. When the LM3431

reaches its minimum on-time, pulse skipping will occur and output ripple will increase. To avoid this, set the

operating frequency below the following maximum setting:

D

f_SW(MAX)

=

t_ON(MIN)

(7)

Inductor Selection

Figure 17 shows how the inductor current, I_L, varies during a switching cycle.

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Vout

200 mV/Div

IL

500 mA/Div

Vsw

10V/Div

400 ns/DIV

Figure 17. Inductor Current, SW Voltage, and V_OUT

The important quantities in determining a proper inductance value are I_L(AVE)(the average inductor current) and

Δi_L(the peak to peak inductor current ripple). If Δi_Lis larger than 2 x I_L(AVE), the inductor current will drop to zero

for a portion of the cycle and the converter will operate in discontinuous conduction mode. If Δi_Lis smaller than 2

x I_L, the inductor current will stay above zero and the converter will operate in continuous conduction mode.

To determine the minimum L, first calculate the I_L(AVE)at both minimum and maximum input voltage:

I_OUT

I_L(AVE)

=

D‘

Where

•

I_OUTis the sum of all LED string currents at 100% dimming

(8)

I_L(AVE)will be highest at the minimum input voltage. Then determine the minimum L based on Δi_Lwith the

following equation:

V_IN(MAX)x D_(MIN)

L_(MIN)

=

Di_Lx f_SW

(9)

A good starting point is to set Δi_Lto 150% of the minimum I_L(AVE)and calculate using that value. The maximum

recommended Δi_Lis 200% of I_L(AVE)to maintain continuous current in normal operation. In general a smaller

inductor (higher ripple current) will give a better dimming response due to the higher dI/dt. This is shown

graphically below.

Vcathode

2V/Div

ILED

100 mA/Div

IL

500 mA/Div

1 ms/DIV

Figure 18. Inductor Current During Dimming

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The resulting peak to peak inductor current is:

V_INx D

Di_L=

L x f_SW

(10)

(11)

And the resulting peak inductor current is:

Di_L

2

IL_PEAK= IL_(AVE)

+

Peak inductor current will occur at minimum V_IN.

The inductor must be rated to handle both the average current and peak current, which is the same as the peak

switch current. As switching frequency increases, less inductance is required. However, some minimum

inductance value is required to ensure stability at duty cycles greater than 50%. The minimum inductance

required for stability can be calculated as:

R3 x (V_OUT- 2 x VIN_(MIN)

)

L_(MIN)

=

f_SWx 75 mV x 2

Where

•

R3 is the sense resistor determined in the next section.

(12)

Although the inductor must be large enough to meet both the stability and the Δi_Lrequirements, a value close to

minimum will typically give the best performance.

Current Sensing

Switch current is sensed via the sense resistor, R3, while the switch is on and the inductor is charging. The

sensed current is used to control switching and to monitor current limit. To optimize the control signal, a typical

sense voltage between 50mV and 200mV is recommended. The sense resistor can therefore be calculated by

the following equation:

50 mV

IL_{(AVE_MIN)}

200 mV

^{Ç R3 Ç}IL_{(AVE_MAX)}

(13)

Since I_L(AVE)will vary with input voltage, R3 should be determined based on the full input voltage range, although

the resulting value may extend somewhat outside the recommended range.

Current Limit

Current limit occurs when the voltage across the sense resistor (measured at the CS pin) equals the current limit

threshold voltage. The current limit threshold is set by R4. This value can be calculated as follows:

(IL_(LIM)x R3) + (D x 75 mV)

R4 =

40 mA

(14)

Where 40 µA is the typical ILIM source current in current limit and ILlim is the peak (not average) inductor current

which triggers current limit.

To avoid false triggering, current limit should be set safely above the peak inductor current level. However, the

current limit resistor also has some effect on the control loop as seen in the block diagram. For this reason, R4

should not be set much higher than necessary. When current limit is activated, the NFET will be turned off

immediately until the next cycle. Current limit will typically result in a drop in output and cathode voltage. This will

cause the COMP pin voltage to increase to maximum, which will trigger a fault and start the DLY pin source

current (see LED Protection section). The LM3431 will continue to operate in current limit with reduced on-time

until the DLY pin has reached its threshold. However, the current limit cannot reduce the on-time below the

minimum specification.

In a boost switcher, there is a direct current path between input and output. Therefore, although the LM3431 will

shutdown in a shorted output condition, there are no means to limit the current flowing from input to output.

Note that if the maximum duty cycle of 85% (typical) is reached, the LM3431 will behave as though current limit

has occurred.

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VCC

The VCC pin is the output of the internal voltage regulator. It must be bypassed to PGND with a minimum 4.7 µF

ceramic capacitor. Although VCC is capable of supplying up to 72 mA, external loads will increase the power

dissipation and temperature rise within the LM3431. See the TSD section for more detail. Above 72 mA, the VCC

voltage will drop due to current limit. Since the UVLO threshold is monitored at this pin, UVLO may be enabled

by a VCC over current event.

For input voltages between 4.5V and 5.5V, connect VCC to V_INthrough a 4.7Ω resistor. This will hold VCC

above the UVLO threshold and allow operation at input voltages as low as 4.5V. It may also be necessary to add

additional V_INand VCC capacitance for low V_INoperation.

Diode Selection

The average current through D1 is the average load current (total LED current), and the peak current through the

diode is the peak inductor current. Therefore, the diode should be rated to handle more than the peak inductor

current which was calculated earlier. The diode must also be capable of handling the peak reverse voltage,

which is equal to the output voltage (LED Anode voltage). To improve efficiency, a low Vf Schottky diode is

recommended. Diode power loss is calculated as:

P_DIODE= Vf x I_OUT

(15)

NFET Selection

The drive pin of the LM3431 boost switcher, LG, must be connected to the gate of an external NFET. The NFET

drain is connected to the inductor and the source is connected to the sense resistor. The LG pin will drive the

gate at 5V typically.

The critical parameters for selection of a MOSFET are:

1. Maximum drain current rating, I_D(MAX)

2. Maximum drain to source voltage, V_DS(MAX)

3. On-resistance, R_DS(ON)

4. Total gate charge, Q_g

In the on-state, the switch current is equal to the inductor current. Therefore, the maximum drain current, I_D, must

be rated higher than the current limit setting. The average switch current (_ID(AVE)) is given in the equation below:

I_D(AVE)= I_L(AVE)x D

(16)

The off-state voltage of the NFET is approximately equal to the output voltage plus the diode Vf. Therefore,

V_DS(MAX)of the NFET must be rated higher than the maximum output voltage. The power losses in the NFET can

be separated into conduction losses and switching losses. The conduction loss, Pcond, is the I²R loss across the

NFET. The maximum conduction loss is given by:

2

P_COND= R_DS(ON)x D_MAXx I_L(AVE)

where

•

D_MAXis the maximum duty cycle for the given application

R_DS(ON)is the on resistance at high temperature

(17)

wThe switching losses can be roughly calculated by the following equation:

f_SWx IL_(AVE)x V_OUTx (t_ON+ t_OFF

)

P_SW

=

2

Where

•

t_ONand t_OFFare the NFET turn-on and turn-off times.

(18)

Power is also consumed in the LM3431 in the form of gate charge losses, P_g. These losses can be calculated

using the formula:

P_g= f_SWx Q_gx V_IN

where

•

Q_gis the NFET total gate charge

(19)

15

Pg adds to the total power dissipation of the LM3431 (See TSD section).

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Fast switching FETs can cause noise spikes at the SW node which may affect performance. To reduce these

spikes a drive resistor up to 10Ω can be placed between LG and the NFET gate.

Input Capacitor Selection

Because the inductor is at the input of a boost converter, the input current waveform is continuous and triangular.

The inductor ensures that the input capacitor sees relatively low ripple currents. The rms current in the input

capacitor is given by:

Di_L

12

I_{RMS_IN}

=

(20)

The input capacitor must be capable of handling this rms current. Input ripple voltage increases with increasing

ESR as well as decreasing input capacitance. A typical value of 10 µF will work well for most applications. For

low input voltages, additional input capacitance may be required to prevent tripping the UVLO. Additionally, a

ceramic capacitor of 1 µF or larger should be placed close to the VIN pin to prevent noise from interfering with

normal device operation.

Output Capacitor Selection

The output capacitor in a boost converter provides all the output current when the switch is on and the inductor is

charging. As a result, the output capacitor sees relatively large ripple currents. The output capacitor must be

capable of handling more than the rms current, which can be estimated as:

D

D‘²

DiL²

12

I_OUT²x

D^‘x

=

+

I_{RMS_OUT}

(21)

Additionally, the ESR of the output capacitor affects the output ripple and has an effect on transient response

during dimming. For low output ripple voltage, low ESR ceramic capacitors are recommended. Although not a

critical parameter, excessive output ripple can affect LED current.

The output capacitance requirement is somewhat arbitrary and depends mostly on dimming frequency. Although

a minimum value of 4 µF is recommended, at lower dimming frequencies, the longer LED-off times will typically

require more capacitance to reduce output voltage transients.

When ceramic capacitors are used, audible noise may be generated during LED dimming. Audible noise

increases with the amplitude of output voltage transients. To minimize this noise, use the smallest case sizes and

if possible, use a larger number of capacitors in parallel to reduce the case size of each. Output transients are

also minimized via the FF pin (See Setting FF section). Setting the dimming frequency above 18 kHz or below

500 Hz will also help eliminate the audible effects of output voltage transients.

When selecting an output capacitor, always consider the effective capacitance at the output voltage, which can

be less than 50% of the capacitance specified at 0V. Use this effective capacitance value for the compensation

calculations below.

Compensation

Once the output capacitor is selected, the control loop characteristics and compensation can be determined. The

COMP pin is provided to ensure stable operation and optimum transient performance over a wide range of

applications. The following equations define the control-to-output or power stage of the loop:

KD

f_P1

=

2p x R_Lx C_OUT

1

f_Z1

=

2p x ESR x C_OUT

V_OUTx (D‘)²

RHP_Z=

2p x I_OUTx L

f_SW

fpn =

2

(22)

Where R_Lis the load resistance corresponding to LED current, and K_fis calculated as shown:

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V_OUT

I_OUT

R_L=

D‘³x R_L

D‘²x 75 mV

+

KD = 1+

L x f_SWx 2

I_OUTx R3

(23)

Since the control-to-output response will shift with input voltage, the compensation should be calculated at both

the minimum and maximum input voltage.

The zero created by the ESR of the output capacitor, f_z1, is generally at a very high frequency if the ESR is small.

If low ESR capacitors are used f_z1can be neglected and if high ESR capacitors are used, C_C2can be added (see

below).

A current mode control boost regulator has an inherent right half plane zero, RHPz. This has the effect of a zero

in the gain plot, causing a +20dB/decade increase, but has the effect of a pole in the phase, subtracting 90° in

the phase plot. This can cause instability if the control loop is influenced by this zero. To ensure the RHP zero

does not cause instability, the control loop must be designed to have a bandwidth of less than one third the

frequency of the RHP zero. The regulator also has a double pole, fpn, at one half the switching frequency. The

control loop bandwidth must be lower than 1/5 of fpn. A typical control-to-output gain response is shown in

Figure 19 below.

fpn

fz1

fp1

RHPz

FREQUENCY

Figure 19. Typical Control-to-Output Bode Plot

Once the control-to-output response has been determined, the compensation components are selected. A series

combination of Rc and Cc is recommended for the compensation network, shown as R9 and C4 in the typical

application circuit. The series combination of Rc and Cc introduces a pole-zero pair according to the following

equations:

1

f_ZC

=

2p x R_Cx C_C

1

f_PC

=

2p x R_Ox C_C

where

•

R_Ois the output impedance of the error amplifier, approximately 500 kΩ.

(24)

The initial value of R_Cis determined based on the required crossover frequency from the following equations

using the maximum input voltage:

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B

R_C=

EA_gmx ILIM_gmx R4

f_CROSS

B =

f_P1x A_cm

V_IN

A_cm

=

R3 x I_OUTx KD

Where

•

B is the mid-frequency compensation gain (in v/v)

R4 is the current limit setting resistor

Acm is the control-output DC gain

the gm values are given in the ELECTRICAL CHARACTERISTICS table

(25)

Fcross is the maximum allowable crossover frequency, based on the calculated values of f_pnand RHPz. Any Rc

value lower than the value calculated above can be used and will ensure a low enough crossover frequency. Rc

should set the B value typically between 0.01v/v and 0.1v/v (-20db to -40db). Larger values of R_Cwill give a

higher loop bandwidth.

However, because the dynamic response of the LM3431 is enhanced by the FF pin (See Setting FF section) the

R_Cvalue can be set conservatively. The typical range for R_Cis between 300ohm and 3 kΩ. Next, select a value

for Cc to set the compensation zero, f_zc, to a frequency greater or equal to the maximum calculated value of f_p1

(f_zccancels the power pole, f_p1). Since an fzc value of up to a half decade above fp1 is acceptable, choose a

standard capacitor value smaller than calculated. Confirm that fpc, the dominant low frequency pole in the control

loop, is less than 100 Hz and below f_p1. The typical range for C_Cis between 10 nF and 100 nF. The

compensation zero-pole pair is shown graphically below, along with the total control loop, which is the sum of the

compensation and output-control response. Since the calculated crossover frequency is an approximation,

stability should always be verified on the bench.

COMP

LOOP

0

fpc

fzc

FREQUENCY (kHz)

RHPz

Figure 20. Typical Compensation and Total Loop Bode Plots

When using an output capacitor with a high ESR value, another pole, f_pc2, may be introduced to cancel the zero

created by the ESR. This is accomplished by adding another capacitor, C_C2, shown as C13 in the Figure 14. The

pole should be placed at the same frequency as f_z1. This pole can be calculated as:

1

f_PC2

=

2p x C_C2x (R_C// R_O)

(26)

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To ensure this equation is valid, and that C_C2can be used without negatively impacting the effects of R_Cand C_C,

f_pc2must be at least 10 times greater than f_zc.

LED CURRENT REGULATOR

Setting LED Current

LED current is independently regulated in each of 3 strings by regulating the voltage at the SNS pins. Each SNS

pin is connected to a sense resistor, shown in the typical application schematic as R10 - R13. The sense resistor

value is calculated as follows:

REFIN

R_SNS

=

I_LED+ I_NDRV

Where

•

I_LEDis the current in each LED string

REF_INis the regulated voltage at the REFIN pin

INDRV is the NPN base drive current

(27)

If using NFETs, INDRV can be ignored. A minimum REFIN voltage of 100 mV is required, and 200mV to 300mV

is recommended for most applications. The REFIN voltage is set with a resistor divider connected to the REF

pin, shown as R7 and R8 in the typical application schematic. The resistor values are calculated as follows:

2.5V - REFIN

R7 = R8 x

REFIN

(28)

The sum of R7 and R8 should be approximately 100k to avoid excessive loading on the REF pin.

NDRV

The NDRV pins drive the base of the external NPN or N-channel MOSFET current regulators. Each pin is

capable of driving up to 15 mA of base current typically. Therefore, NPN devices with sufficient gain must be

selected. The required NDRV current can be calculated from the following equation, where β is the NPN

transistor gain.

I_LED

I_NDRV

=

b

(29)

If NFETs are used, the NDRV current can be ignored. NPN transistors should be selected based on speed and

power handling capability. A fast NPN with short rise time will give the best dimming response. However, if the

rise time is too fast, some ringing may occur in the LED current. This ringing can be improved with a resistor in

series with the NDRV pins. The NPNs must be able to handle a power equal to I_LEDx NPN voltage. Note that the

NPN voltage can be as high as approximately 5.5V in a fault condition. The NDRV pins have a limited slew rate

capability which can increase the turn-on delay time when driving NFETs. This delay increases the minimum

dimming on-time and can affect the dimming linearity at high dimming frequencies. Low V_GSthreshold NFETs are

recommended to ensure that they will turn fully on within the required time. At dimming frequencies above 10

kHz, NPN transistors are recommended for the best performance.

CFB and SC Diodes

The bottom of each LED string is connected to the CFB and SC pins through diodes as shown in Figure 14. The

CFB pin receives voltage feedback from the lowest cathode voltage. The other string cathode voltages will vary

above the regulated CFB voltage. The actual cathode voltage on these strings will depend on the LED forward

voltages. This ensures that the lowest cathode voltage (highest Vf) will be regulated with enough headroom for

the NPN regulator. The SC pin monitors for LED fault conditions and limits the maximum cathode voltage (See

LED Protection section). In this way, each LED string’s cathode is maintained within a window between minimum

headroom and fault condition.

Both the CFB and SC diodes must be rated to at least 100 µA, and the CFB diode should have a reverse voltage

rating higher than V_OUT. With these requirements in mind, it is best to use the smallest possible case size in

order to minimize diode capacitance which can slow the LED current rise and fall times.

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Dimming

The LM3431 is compatible with both analog and digital LED dimming signals. The MODE/F pin is used to select

analog or digital mode. When MODE/F is pulled above 3.8V, digital mode is enabled and a PWM signal up to 25

kHz can be applied to the DIM pin. In this mode, the LED current regulators will be active when DIM is above 2V

(typical) and inactive when DIM is pulled below 1.1V (typical). Although any pulse width may be used at the DIM

pin, 0.4 µs is the minimum LED on time (in either digital or analog mode). This limits the minimum dimming duty

cycle at high dimming frequencies. For example, at 20 kHz, the dimming duty cycle is limited to 0.8% minimum.

At lower dimming frequencies, the dimming duty cycle can be much lower and the minimum depends on the

application conditions including the FF setting (see Setting FF section). In analog dimming mode, the MODE/F

pin is used to set the PWM dimming frequency, and duty cycle is controlled by varying the analog voltage level at

the DIM pin. To operate in analog mode, connect a capacitor from MODE/F to ground, shown as C5 in the

typical application (without the pull-up resistor installed). The dimming frequency is set according to the following

equation:

40 mA

2 x f_DIMx 2.13

C5 =

(30)

In analog mode, the MODE/F pin will generate a triangle wave with a peak of 2.5V and minimum of 0.37V. The

DIM pin voltage is compared to the MODE/F voltage to create an internal PWM dimming signal whose duty cycle

is proportional to the DIM voltage. When the DIM voltage is above 2.5V, the duty cycle is 100%. Duty cycle will

vary linearly with DIM voltage as shown in Figure 21. Typical analog dimming waveforms are shown below in

Figure 22.

100

80

60

40

20

0

0.3

0.9

1.5

2.1

2.7

DIM (V)

Figure 21. Analog Mode Dimming Duty Cycle vs. DIM voltage

ILED

100 mA/Ddiv

LEDOFF

5V/Div

MODE/F

1V/Div

DIM

1V/Div

2 ms/DIV

Figure 22. Analog Dimming Mode Waveforms

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In PWM dimming, the average LED current is equal to the set LED current (I_LED) multiplied by the dimming duty

cycle. The average LED current tracks the dimming ratio with exceptional linearity. However, the accuracy of

average LED current depends somewhat on the rise and fall times of the external current regulators. This

becomes more apparent with short on-times. To ensure good linearity, select NPN regulators with short and

similar rise and fall times.

Setting FF

To minimize voltage transients during LED dimming, the output voltage is regulated via the AFB pin during LED

off times. However, because the control loop has a limited response time, voltage transients can never be

completely eliminated. If these transients are large enough, LED current will be affected and ceramic output

capacitors may generate audible noise. The FF pin speeds up the loop response time, and thus minimizes output

voltage transients during dimming.

A resistor connected from FF to ground, Rff, sets the FF current which is injected into the control loop at the

rising and falling edge of the dimming signal. In this way, the FF pin creates a correction signal before the control

loop can respond. A smaller FF resistor will generate a larger correction signal. The minimum recommended Rff

value is 10k.

Since the amount of FF correction required for a given application depends on many factors, it is best to

determine a FF resistor value through bench testing. Use the following procedure to determine an optimal Rff

value:

An Rff value of approximately 20k is a good starting point. A 20 kΩ potentiometer in series with a 10 kΩ resistor

works well for bench testing.

The dimming frequency must be selected before setting Rff. Confirm that boost switching operation is stable at

100% dimming duty cycle.

Adjust Rff until the COMP pin voltage is between 0.8V and 0.9V. Next, monitor the cathode voltage response at

a low dimming duty cycle while adjusting Rff until the overshoot and undershoot is minimal or there is a slight

overshoot.

Check the cathode voltage response at the lowest input voltage and lowest dimming duty cycle and adjust Rff if

necessary. This is typically the worst case condition.

The curves in Figure 23 below show the variation in cathode voltage with different Rff settings. Notice that at the

ideal setting, both the cathode voltage and COMP voltage are flat. For clarity, the 3 cathode voltage curves in

this figure have been offset; all FF settings will result in the cathode voltage settling at 1.2V typically.

COMP

500 mV/Div

RFF

low

RFF

ideal

RFF

high

Vcathode

1V/Div

ILED

100 mA/Div

20 ms/DIV

Figure 23. FF Setting Example

Once an Rff value has been set, check the cathode voltage over the input voltage range and dimming duty

range. Some further adjustment may be necessary.

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In practice the FF pin also has a small effect on the control loop response. As a final step, switching stability at

100% dimming duty should be re-verified once the Rff value has been selected. At the optimal Rff setting, output

voltage transients will be minimized and the cathode voltage will be stable across the range of input voltage and

dimming duty cycle.

The ideal cathode response illustrated in Figure 23 may not be achievable over the entire input voltage range.

However, LED current will not be affected as long as the cathode voltage remains above the regulator saturation

voltage and below the open LED fault threshold (See Open LED section).

A wide input voltage range will cause a wider variation in the feedforward effect, thus making duty cycles less

than 1% more difficult to achieve. For any given application there is a minimum achievable dimming duty cycle.

Below this duty cycle, the cathode voltage will begin to drift higher, eventually appearing as an open LED fault

(See LED Protection section).

During an LED open fault condition, cathode voltage overshoot will tend to increase. If Rff is not set

appropriately, high overshoots may be detected as an LED short fault and lead to shutdown.

LED Protection

Fault Modes and Fault Delay

The LM3431 provides 3 types of protection against several types of potential faults. Table 1 summarizes the fault

protections and groups the fault responses into three types (the auto-restart option is described in the next

section).

Table 1. Fault Mode Summary

Fault

1 LED open

Mechanism

SC > 3.1V

Action

Response

continue to regulate

Shutdown or auto-restart

stand by

Type

DLY charges

No DLY flag

1

1 LED short

SC > 3.1V

All LEDs open

Output over-voltage

multiple LED short

Multiple LED short, VIN<6V

Cathode short

Current limit

AFB > 2.0V

SC > 6.0V

2

AFB < 0.85V

CFB low at startup

COMP at max

VCC or EN low

IC over temperature

THM < 1.2V

UVLO

TSD

stand by

3

THM

stand by

When Type 1 or Type 2 faults occur, the DLY pin begins sourcing current (57 µA typical). A capacitor connected

from DLY to ground (C7) sets the DLY voltage ramp and shutdown delay time. For a Type 1 fault, the LM3431

will continue to regulate, although the DLY pin remains high. In this condition, the DLY pin will charge to a

maximum of 3.6V (typical).

In case of a Type 2 fault, when the DLY voltage reaches 2.8V (typical), the LM3431 will shut down and the DLY

pin will remain at 3.6V.

For evaluation and debugging purposes, Type 2 shutdown can be disabled by grounding the DLY pin. It is not

recommended to leave the DLY pin open.

For any fault other than a cathode short, the DLY pin will discharge (sinking 1.8 µA) when the fault is removed

before shutdown occurs. Since most fault conditions can only be sensed during the LED-on dimming period, the

DLY pin will not charge during LED-off times. When the LEDs are off, DLY is in a high impedance state and its

voltage will remain constant. If a fault is removed during the LED-off period, DLY will begin discharging at the

next LED-on cycle. If the fault is not removed, DLY will continue charging at the next LED-on cycle. Therefore,

the DLY charging time is controlled by both the DLY capacitor and the dimming duty cycle. The time for the DLY

pin to charge to the shutdown threshold can be calculated as shown:

C7 x 2.8V

1

D_DIM

x

t_dly =

57 mA

(31)

22

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Where D_DIMis the dimming duty cycle. Figure 24 below shows the DLY pin charging during dimming due to a

Type 1 fault:

Vcathode

2V/Div

DLY

2V/Div

ILED

100 mA/Div

1 ms/DIV

Figure 24. DLY Charging, 1 LED Open Fault

When the LED string turns on, there is a 1.6 µs typical blanking time for fault detection. This ensures that the

LED cathode voltage will reach its regulation point and faults will not be falsely triggered. However, faults can not

be detected during short dimming cycles of less than 1.6 µs.

When a Type 3 fault occurs, DLY does not charge. The LM3431 will enter standby mode and restart from

softstart when the fault condition is removed.

Fault Shutdown and Automatic Restart

In normal operation, the LM3431 must be powered off or put into standby via the EN pin to restart after a fault

shutdown. However, the LEDOFF pin can be connected to GND to enable the automatic restart feature. During

startup, the LEDOFF voltage is monitored and if grounded, auto-restart mode is enabled.

In auto-restart mode, the DLY pin will be discharged by a 1.8 µA sink current after a Type 2 shutdown. In this

mode, DLY will not reach 3.6V, but will start discharging from the shut down threshold of 2.8V. When the DLY

pin voltage falls to 350 mV (typical) the LM3431 will restart from softstart mode. In this way, the DLY capacitor

controls the restart delay time. If the LEDOFF pin is used to control additional LED strings (see LEDOFF: Adding

Additional channels), then the automatic restart feature cannot be enabled.

In the case of an output over-voltage fault (all LEDs open), DLY will not discharge until the AFB voltage falls

below the OVP threshold. Figure 25 below shows an OVP fault with auto-restart activated. The output voltage

increases when all LEDs are opened, causing DLY to charge. DLY remains at 2.8V until Vout falls below the

OVP threshold. When DLY discharges to 350 mV, softstart begins. In auto-restart mode, the LM3431 will re-start

continually until the fault is removed. In this example, the fault is removed and normal operation continues after

one attempted re-start.

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Vout

10V/Div

SS/SH

1V/Div

DLY

2V/Div

ILED

500 mA/Div

10 ms/DIV

Figure 25. OVP and Auto-Restart

Open LED

If any LED string fails open, the boost regulator will sense a low voltage at the CFB pin. This will cause the

output voltage to increase, causing the other LED string cathode voltages to also increase. When the SC pin

voltage rises to 3.1V, a Type 1 fault will be triggered and the DLY pin will begin sourcing current. In this mode,

the SC voltage will be clamped at 3.5V (typical) and the regulators will continue to operate. At this higher cathode

voltage, power dissipation will increase in the external NPN regulators. Power dissipation will also increase in the

LM3431 since any open string will cause the NDRV pin to source its maximum current. A one LED open fault

condition is shown in Figure 24 above. The open LED causes the cathode voltage to increase, and DLY charges

during each LED on cycle while current continues to be regulated in the other LED strings.

If all LED strings fail open, the same action will cause the output voltage to increase. However, in this case, SC

will be held low and cannot sense the failure. Instead, this failure mode is sensed by AFB. When AFB reaches its

over-voltage threshold of 2.0V (typical), a Type 2 fault will be triggered, the DLY pin will begin sourcing current,

and the LM3431 will shut down.

Unlike the SC and CFB fault detection, the AFB pin is always monitored. Therefore, DLY charging time will not

be affected by the dimming duty cycle and any over-voltage condition will cause DLY to charge.

Shorted LED

If an LED fails short circuit, the SC voltage will increase. When SC reaches 3.1V, the same Type 1 fault as an

open LED will be triggered. Current in the affected string will continue to be regulated, with the cathode clamped

at one diode V_fabove 3.5V. As in the case of 1 LED open, the power dissipation will increase in the external

NPN regulator of the shorted string.

However, if enough LEDs or an entire string are shorted, the SC pin will rise to the short circuit threshold of 6.0V.

This will cause a Type 2 fault, and the LM3431 will shut down when the DLY threshold is reached.

When an LED string is shorted, the LM3431 will attempt to reduce the SC voltage to 3.5V. As a result, switching

will stop, and the cathode voltage will be brought to the minimum level, which is Vin. If Vin is less than

approximately 6V and the DLY time is long enough, SC will fall below the 6V short circuit fault threshold. In this

case, the shorted string fault will be detected as an AFB under-voltage (UVP) fault.

When AFB falls below 0.85V (typical) a Type 2 fault will be triggered. As is the case with OVP detection, the AFB

UVP threshold is monitored during both LED on and LED off cycles. A UVP fault will cause DLY to charge,

unaffected by the dimming duty cycle. Figure 26 below shows the sudden cathode voltage increase due to an

LED string short. DLY begins charging and charges continuously when an AFB under-voltage is detected,

eventually causing a shutdown.

24

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Vcathode

5V/Div

DLY

1V/Div

ILED

200 mA/Div

400 ms/DIV

Figure 26. LED String Short Fault and UVP Detection

Shorted Cathode

At the end of softstart, the CFB voltage is monitored. In normal startup, the LED strings are off and CFB voltage

increases with the output voltage. If the CFB voltage stays below approximately 1.9V, a cathode short to ground

condition is detected and a Type 2 fault is triggered. At the end of soft-start, the DLY pin will begin sourcing

current and it will continue sourcing until the shutdown threshold is reached, even if the short condition is

removed.

When a cathode short occurs, the LEDs in the affected string will be driven on during the soft-start and DLY

periods. Therefore, the DLY and soft-start time should be set short enough for the LED string to withstand the

burst of unregulated current.

Thermal Considerations

To optimize performance under all conditions, the LM3431 controls the temperature coefficients of critical

parameters and provides over-temperature protection for both the IC and LEDs.

THM

The THM pin is designed to monitor for over-temperature conditions at the LED array. This is done with a

negative TC thermistor mounted at the LED panel. The THM circuit is a resistor divider from a reference voltage

to ground, shown in Figure 27 as R17 and Rth. As the thermistor temperature increases, the THM pin voltage will

decrease. When THM drops to 1.23V (typical), a Type 3 fault is triggered and the LM3431 will enter standby until

the thermistor temperature decreases and THM voltage increases. Thermistors are typically specified by their

resistance at 25°C, and by their beta constant which describes the temperature coefficient. The resistance value

at the desired shutdown temperature can be calculated from the beta constant or found in the thermistor

datasheet table. Once the shutdown temperature resistance is known, the R17 value can be calculated as shown

below.

VCC - 1.23V

R17 = Rth @ T x

1.23V

(32)

where Rth@T is the thermistor resistance at the desired shutdown temperature. Although VCC is shown in the

typical application schematic, any regulated voltage source can be used in its place, including V_REF

.

In shutdown, THM sinks 10 µA to create some hysteresis. An R17 value of at least 20 kΩ is recommended to

create sufficient hysteresis. Larger values of R17 (and Rth) will generate larger hysteresis.

If more hysteresis is required, a resistor can be added in series with THM as shown below:

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VCC

R17

PGND

THM

AFB

SC

Rhys

Rth

Figure 27. THM Circuit with Hysteresis

The THM hysteresis can be determined by calculating the restart threshold as shown below. If R_HYSis not

installed, calculate Rth@restart using an R_HYSvalue of 0Ω.

1.23V œ (10 mA x R_HYS

)

Rth @ restart =

VCC - 1.23V œ (10 mA x R_HYS

)

- 10 mA

R17

(33)

Where 10 µA is the THM sink current, and Rth@restart is the thermistor resistance at the restart temperature.

Refer to the manufacturer datasheet to find the restart temperature at the calculated resistance or use the beta

constant to calculate the restart temperature.

During startup (and re-start), the THM monitor is active. Therefore, the thermistor temperature must be below the

restart threshold for the LM3431 to startup.

TSD

If the LM3431 internal junction temperature increases above 160°C TSD is activated. This is a Type 3 fault

condition. Device temperature rise is determined by internal power dissipation primarily in the LG and NDRVx

drivers. The power dissipation can be estimated as follows:

P_D= P_IQ+ P_NDRV+ P_VCC+ P_G

P_IQ= V_INx I_Q

(34)

Where

•

I_Qis 4.0 mA typically

(35)

P_NDRV= (V_IN- REFIN - Vbe) x I_NDRVx D_DIMx #Strings

Where

•

REFIN+Vbe is the NDRV voltage

I_NDRVwas calculated previously in the NDRV section

(36)

For the case of open LEDs, I_NDRVon the open string will be at the maximum of 15 mA. The LM3431 power

dissipation will be highest in open LED conditions at 100% dimming duty. If NFETs are used for regulation,

P_NDRVwill be a function of dimming frequency and can be calculated as:

P_{NDRV_FET}= f dim x Q_gx V_IN

P_VCC= (V_IN- V_CC) x I_VCC

(37)

Where

•

_VCCis any current being drawn from the VCC pin, such as external op-amp power, or THM voltage divider. (38)

The LG power dissipation, PG is given in the NFET Selection section.

Temperature rise can then be calculated as:

T_RISE= P_Dx θ_JA

Where

•

θ_JAis typically 32°C/W and varies with pcb copper area (Refer to the PCB Layout section)

(39)

26

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Although the TSD threshold is 160°C, the LM3431 may not operate within specification at temperatures above

the maximum rating of 125°C. Power dissipation should be limited to ensure that device temperature stays within

this limit.

Temperature Coefficients

Several device specifications are designed to vary with temperature. To maintain optimum headroom control and

minimum NPN power dissipation, CFB regulation has a tempco of -2.6 mV/°C. This is matched to the typical

tempco of the small signal diodes used for the cathode feedback connection. Although the CFB voltage will vary

with temperature, the cathode voltage will remain stable. The SS/SH pin rises to 1.85V typically during soft start.

This voltage has a tempco of approximately -2.2 mV/°C, which is designed to follow the tempco of the LED

strings. At then end of soft start, the anode voltage will be greater than the maximum LED forward voltage,

regardless of operating temperature. To avoid false errors, the AFB overvoltage threshold has a tempco of -1.4

mV/°C. Of course, these temperature monitoring features are most effective with the LM3431 mounted within the

same ambient temperature as the LEDs.

LEDOFF: Adding Additional channels

Although the LM3431 has three internal current controllers, more channels can easily be added. A fourth LED

string is shown in Figure 14 connected to VC4.

For additional channels, the sense resistor should be the same value as the main three channels. During startup

and dimming off time, LEDOFF rises to 5V, which quickly turns off the external driver. While the LED strings are

on, the LEDOFF signal is low, allowing normal regulation. If LEDOFF is used to add additional channels, it

cannot be used to enable auto-restart mode.

All additional channels must also be connected through diodes to the SC and CFB pins as shown in the typical

application schematic. The op-amp used to drive the additional channel current regulator must be fast enough to

drive the regulator fully on within the DLY blanking time. A slew rate of 5V/µsec is typically sufficient. Also, the

op-amp output must be capable of completely turning off the NPN regulator, which requires a drive voltage no

greater than the REFIN voltage. A rail-to-rail type op-amp is recommended.

Finally, the R14 resistor should be large enough to limit V_CCcurrent during the LED-off cycle. A value of at least

1k is recommended. Any additional channels will have a longer turn-on delay time than channels 1-3. An

additional delay time of 250 ns is typical. The added delay can affect dimming linearity at on times less than 1 µs.

LED Current Accuracy

LED string current accuracy is affected by factors both internal and external to the LM3431. For any single string

the maximum deviation from ideal is simply the sum of the sense resistor, offset error, REF voltage, REFIN

resistor divider accuracy, and bipolar gain variation:

Db x 100

2 x b²

5 mV x 100

REFIN

Acc_single% = ê A_R10+ 2% + A_R7+ A_R8

+

Where

•

A_R10is the sense resistor % accuracy

2% is the REF voltage accuracy, A_R7

A_R8are the REFIN setting resistors % accuracy

5 mV is the maximum SNS amp offset voltage (use 3 mV for LM3431A)

β is the gain of NPN transistor

Δβ is the specified range of gain in the NPN

(40)

The string-to-string accuracy is the maximum difference in current between any two strings. It is best calculated

using the RSS method:

2

+

Db x100

2 x b²

6 mV x 100

REFIN

2

Acc_s-s% = ê 2 x A_R10

+

Where

•

6 mV is the maximum SNS amp delta offset voltage (V_{OS_DELTA}over temperature, use 4 mV for LM3431A) and

we are assuming the sense resistors have the same accuracy rating

(41)

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If FETs are used, the β term can be ignored in both equations. The LED current in each string will be within

±Acc_single% of the set current. And the difference between any two strings will be within ±Acc_s-s% of each

other.

PCB Layout

Good PCB layout is critical in all switching regulator designs. A poor layout can cause EMI problems, excess

switching noise, and improper device operation. The following key points should be followed to ensure a quality

layout.

Traces carrying large AC currents should be as wide and short as possible to minimize trace inductance and

associated noise spikes.

These areas, shown hatched in Figure 28, are:

•

The connection between the output capacitor and diode

The PGND area between the output capacitor, R3 sense resistor, and bulk input capacitor

The switch node

The current sensing circuitry in current mode controllers can be easily affected by switching noise. Although the

LM3431 imposes 170ns of blanking time at the beginning of every cycle to ignore this noise, some may remain

after the blanking time. Following the important guidelines below will help minimize switching noise and its effect

on current sensing.

As shown in Figure 28, ground the output capacitor as close as possible to the bottom of the sense resistor. This

connection should be somewhat isolated from the rest of the PGND plane (place no ground plane vias in this

area). The V_OUTside of the output capacitor should be placed close to the diode.

The SW node (the node connecting the diode anode, inductor, and FET drain) should be kept as small as

possible. This node is one of the main sources for radiated EMI. Sensitive traces should not be routed in the

area of the SW node or inductor.

The CS pin is sensitive to noise. Be sure to route this trace away from the inductor and the switch node. The CS,

LG, and ILIM traces should be kept as short as possible. As shown below, R4 must be grounded close to the

ground side of R3.

The VCC capacitor should be placed as close as possible to the IC and grounded close to the PGND pin. Take

care in routing any other VCC traces away from noise sources and use decoupling capacitors when using VCC

as an external voltage supply.

A ceramic input capacitor must be connected as close as possible to the VIN pin and grounded close to the

PGND pin.

An isolated ground area shown as SGND is recommended for small signal ground connections. The SGND

plane should connect to both the exposed pad (EP) and SGND pin. The SGND and PGND ground planes should

be connected to their respective pins and both pins should be connected only through the exposed pad, EP.

Components connecting all of the following pins should be placed close to the device and grounded to the SGND

plane: REF, REFIN, AFB, COMP, RT, FF, MODE/F, and SS/SH. These components and their traces should not

be routed near the switch node or inductor. The LED current sense resistors should be grounded to the SGND

plane for accurate current sensing. This area, shown as LGND in Figure 28, should be somewhat separated from

SGND and must provide enough copper area for the total LED current.

If driving more than 3 channels, the layout of the additional channels should be within a minimal area with short

trace lengths. This will help to reduce ringing and delay times. Connections to the LED array should be as short

as possible. Less than 25 cm is recommended. Longer lead lengths can cause excessive ringing or oscillation.

A large, continuous ground plane should be placed as an inner or bottom layer for thermal dissipation. This plane

should be considered as a PGND area and not used for SGND connections. To optimize thermal performance,

multiple vias should be placed directly below the exposed pad to increase heat flow into the ground plane. The

recommended number of vias is 10-12 with a hole diameter between 0.20 mm and 0.33 mm. See TI Lit Number

SNVA183 for more information.

28

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VIN

SW

PGND

Vout

R3

R4

EP

LGND

SGND

Figure 28. Example PCB Layout

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REVISION HISTORY

Changes from Revision F (May 2013) to Revision G

Page

•

Changed layout of National Data Sheet to TI format .......................................................................................................... 29

30

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PACKAGE MATERIALS INFORMATION

www.ti.com

5-Jan-2022

TAPE AND REEL INFORMATION

*All dimensions are nominal

Device

Package Package Pins

Type Drawing

SPQ

Reel

A0

B0

K0

P1

W

Pin1

Diameter Width (mm) (mm) (mm) (mm) (mm) Quadrant

(mm) W1 (mm)

LM3431AMHX/NOPB HTSSOP PWP

LM3431AQMHX/NOPB HTSSOP PWP

28

2500

330.0

16.4

6.8

10.2

1.6

8.0

16.0

Q1

LM3431MHX/NOPB

HTSSOP PWP

LM3431QMHX/NOPB HTSSOP PWP

Pack Materials-Page 1

PACKAGE MATERIALS INFORMATION

www.ti.com

5-Jan-2022

*All dimensions are nominal

Device

Package Type Package Drawing Pins

SPQ

Length (mm) Width (mm) Height (mm)

LM3431AMHX/NOPB

LM3431AQMHX/NOPB

LM3431MHX/NOPB

LM3431QMHX/NOPB

HTSSOP

PWP

28

2500

367.0

35.0

Pack Materials-Page 2

PACKAGE MATERIALS INFORMATION

www.ti.com

5-Jan-2022

TUBE

*All dimensions are nominal

Device

Package Name Package Type

Pins

SPQ

L (mm)

W (mm)

T (µm)

B (mm)

LM3431AMH/NOPB

LM3431AQMH/NOPB

LM3431MH/NOPB

LM3431QMH/NOPB

PWP

HTSSOP

28

48

495

8

2514.6

4.06

Pack Materials-Page 3

PACKAGE OUTLINE

PWP0028A

PowerPAD^TM- 1.1 mm max height

S

C

A

L

E

1

.

8

0

PLASTIC SMALL OUTLINE

C

6.6

6.2

TYP

SEATING PLANE

A

PIN 1 ID

AREA

0.1 C

26X 0.65

28

1

9.8

9.6

NOTE 3

2X

8.45

14

B

15

0.30

0.19

28X

1.1 MAX

4.5

4.3

0.1

C A

B

NOTE 4

0.20

0.09

TYP

SEE DETAIL A

3.15

2.75

0.25

GAGE PLANE

5.65

5.25

0.10

0.02

THERMAL

PAD

0 - 8

0.7

0.5

DETAIL A

(1)

TYPICAL

4214870/A 10/2014

PowerPAD is a trademark of Texas Instruments.

NOTES:

1. All linear dimensions are in millimeters. Any dimensions in parenthesis are for reference only. Dimensioning and tolerancing

per ASME Y14.5M.

2. This drawing is subject to change without notice.

3. This dimension does not include mold flash, protrusions, or gate burrs. Mold flash, protrusions, or gate burrs shall not

exceed 0.15 mm, per side.

4. This dimension does not include interlead flash. Interlead flash shall not exceed 0.25 mm, per side.

5. Reference JEDEC registration MO-153, variation AET.

www.ti.com

EXAMPLE BOARD LAYOUT

PWP0028A

PowerPAD^TM- 1.1 mm max height

PLASTIC SMALL OUTLINE

(3.4)

NOTE 9

(3)

SOLDER

MASK

OPENING

SOLDER MASK

DEFINED PAD

28X (1.5)

28X (1.3)

28X (0.45)

1

28

26X

(0.65)

SYMM

(5.5)

(9.7)

SOLDER

MASK

OPENING

(1.3) TYP

14

15

(

0.2) TYP

(1.3)

SEE DETAILS

(0.65) TYP

(0.9) TYP

(6.1)

VIA

SYMM

METAL COVERED

BY SOLDER MASK

HV / ISOLATION OPTION

0.9 CLEARANCE CREEPAGE

OTHER DIMENSIONS IDENTICAL TO IPC-7351

(5.8)

IPC-7351 NOMINAL

0.65 CLEARANCE CREEPAGE

LAND PATTERN EXAMPLE

SCALE:6X

SOLDER MASK

OPENING

SOLDER MASK

OPENING

METAL

METAL UNDER

SOLDER MASK

0.05 MAX

ALL AROUND

0.05 MIN

ALL AROUND

SOLDER MASK

DEFINED

NON SOLDER MASK

DEFINED

SOLDER MASK DETAILS

4214870/A 10/2014

NOTES: (continued)

6. Publication IPC-7351 may have alternate designs.

7. Solder mask tolerances between and around signal pads can vary based on board fabrication site.

8. This package is designed to be soldered to a thermal pad on the board. For more information, see Texas Instruments literature

numbers SLMA002 (www.ti.com/lit/slma002) and SLMA004 (www.ti.com/lit/slma004).

9. Size of metal pad may vary due to creepage requirement.

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EXAMPLE STENCIL DESIGN

PWP0028A

PowerPAD^TM- 1.1 mm max height

PLASTIC SMALL OUTLINE

(3)

BASED ON

0.127 THICK

STENCIL

METAL COVERED

BY SOLDER MASK

28X (1.5)

28X (1.3)

28X (0.45)

1

28

26X (0.65)

28X (0.45)

(5.5)

SYMM

BASED ON

0.127 THICK

STENCIL

14

15

SEE TABLE FOR

DIFFERENT OPENINGS

SYMM

(6.1)

FOR OTHER STENCIL

THICKNESSES

(5.8)

HV / ISOLATION OPTION

0.9 CLEARANCE CREEPAGE

OTHER DIMENSIONS IDENTICAL TO IPC-7351

IPC-7351 NOMINAL

0.65 CLEARANCE CREEPAGE

SOLDER PASTE EXAMPLE

EXPOSED PAD

100% PRINTED SOLDER COVERAGE AREA

SCALE:6X

STENCIL

THICKNESS

SOLDER STENCIL

OPENING

0.1

3.55 X 6.37

3.0 X 5.5 (SHOWN)

2.88 X 5.16

0.127

0.152

0.178

2.66 X 4.77

4214870/A 10/2014

NOTES: (continued)

10. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate

design recommendations.

11. Board assembly site may have different recommendations for stencil design.

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IMPORTANT NOTICE AND DISCLAIMER

TI PROVIDES TECHNICAL AND RELIABILITY DATA (INCLUDING DATA SHEETS), DESIGN RESOURCES (INCLUDING REFERENCE

DESIGNS), APPLICATION OR OTHER DESIGN ADVICE, WEB TOOLS, SAFETY INFORMATION, AND OTHER RESOURCES “AS IS”

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IMPLIED WARRANTIES OF MERCHANTABILITY, FITNESS FOR A PARTICULAR PURPOSE OR NON-INFRINGEMENT OF THIRD

PARTY INTELLECTUAL PROPERTY RIGHTS.

These resources are intended for skilled developers designing with TI products. You are solely responsible for (1) selecting the appropriate

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These resources are subject to change without notice. TI grants you permission to use these resources only for development of an

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TI objects to and rejects any additional or different terms you may have proposed. IMPORTANT NOTICE

Mailing Address: Texas Instruments, Post Office Box 655303, Dallas, Texas 75265


型号：	LM3431AQMH/NOPB
厂家：	TEXAS INSTRUMENTS
描述：	具有集成升压控制器的 3 通道汽车类 LED 恒流驱动器 \| PWP \| 28 \| -40 to 125 驱动控制器开关光电二极管驱动器
文件：	总39页 (文件大小：1985K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

LM3431AQMH/NOPB [TI]

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