LM43601-Q1 [TI]
通过汽车级认证的 3.5V 至 36V、1A 同步降压电压转换器;型号: | LM43601-Q1 |
厂家: | TEXAS INSTRUMENTS |
描述: | 通过汽车级认证的 3.5V 至 36V、1A 同步降压电压转换器 转换器 |
文件: | 总53页 (文件大小:2543K) |
中文: | 中文翻译 | 下载: | 下载PDF数据表文档文件 |
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LM43601-Q1
ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
LM43601-Q1 3.5V 至 36V 1A 同步降压转换器
1 特性
3 说明
1
•
•
符合汽车应用 标准
具有符合 AEC-Q100 标准的下列特性:
LM43601-Q1 稳压器是一款易于使用的同步降压直流/
直流转换器,能够驱动高达 1A 的负载电流,输入电压
范围为 3.5V 至 36V(42V 瞬态)。LM43601-Q1 以极
小的解决方案尺寸提供出色的效率、输出精度和压降电
压。扩展系列能够以引脚对引脚兼容封装提供 0.5A、
2A 和 3A 负载电流选项。采用峰值电流模式控制来实
现简单控制环路补偿和逐周期电流限制。可选 功能 包
括可编程开关频率、同步、电源正常标志、精确使能、
内部软启动、可扩展软启动和跟踪,可为各种 应用提
供灵活且易于使用的平台。轻载时的断续传导和自动频
率调制可提升轻载效率。此系列只需要很少的外部组
件,而且引脚排列可实现简单且优化的 PCB 布局。保
护 采用了 包括热关断、VCC 欠压锁定、逐周期电流限
制和输出短路保护。LM43601-Q1 器件采用 HTSSOP
(PWP) 16 引线式封装 (6.6mm × 5.1mm × 1.2mm),
引线间距为 0.65mm。该器件与 LM4360x 和
–
器件温度 1 级:-40℃ 至 +125℃ 的环境运行温
度范围
•
•
•
33µA 稳压静态电流
可在轻负载条件下实现高效率(DCM 和 PFM)
经测试符合 EN55022/CISPR 22 电磁干扰 (EMI)
标准
•
•
集成同步整流
可调频率范围:200kHz 至 2.2MHz(默认值为
500kHz)
•
•
•
与外部时钟频率同步
内部补偿
与几乎任一陶瓷、固态电解、钽和铝质电容器组合
一同工作时保持稳定
•
•
•
•
•
•
•
•
•
电源正常状态标志
软启动至预偏置负载
LM4600x 系列引脚对引脚兼容。LM43601A-Q1 版本
针对 PFM 运行模式进行了优化,推荐用于新设计。
内部软启动:4.1ms
可由外部电容器延长的软启动时间
输出电压跟踪功能
器件信息(1)
器件型号
LM43601-Q1
LM43601A-Q1
封装
封装尺寸(标称值)
精确使能实现系统欠压闭锁 (UVLO)
具有断续模式的输出短路保护
过温保护
使用 LM43601-Q1 并借助 WEBENCH® 电源设计
器创建定制设计
HTSSOP (16)
6.60mm × 5.10mm
(1) 如需了解所有可用封装,请参阅数据表末尾的可订购产品附
录。
空白
2 应用
•
•
•
•
AM 以下波段汽车应用
电信系统
通用宽 VIN 稳压
高效负载点稳压
简化原理图
辐射发射图
VIN = 12V,VOUT = 3.3V,FS= 500kHz,IOUT = 1A
L
VIN
RENT
VOUT
VIN
SW
COUT
CFF
CIN
LM43601-Q1
CBOOT
dBuV
80
RFBT
CBOOT
FB
Vertical Polarization
ENABLE
RENB
70
Horizontal Polarization
60
50
VCC
RFBB
SS/TRK
RT
CVCC
CSS
EN 55022 Class B Limit
40
30
20
BIAS
RT
CBIAS
10
Evaluation Board Emissions
SYNC
AGND
PGOOD
PGND
30
100
Frequency (MHz)
1000
RSYNC
Tie BIAS to PGND
when VOUT < 3.3 V
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
English Data Sheet: SNVSAA0
LM43601-Q1
ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
www.ti.com.cn
目录
7.3 Feature Description................................................. 15
7.4 Device Functional Modes........................................ 23
Applications and Implementation ...................... 24
8.1 Application Information............................................ 24
8.2 Typical Applications ................................................ 24
Power Supply Recommendations...................... 41
1
2
3
4
5
6
特性.......................................................................... 1
应用.......................................................................... 1
说明.......................................................................... 1
修订历史记录 ........................................................... 2
Pin Configuration and Functions......................... 3
Specifications......................................................... 4
6.1 Absolute Maximum Ratings ...................................... 4
6.2 ESD Ratings.............................................................. 4
6.3 Recommended Operating Conditions....................... 4
6.4 Thermal Information.................................................. 5
6.5 Electrical Characteristics........................................... 5
6.6 Timing Requirements................................................ 7
6.7 Switching Characteristics.......................................... 7
6.8 Typical Characteristics.............................................. 8
Detailed Description ............................................ 14
7.1 Overview ................................................................. 14
7.2 Functional Block Diagram ....................................... 14
8
9
10 Layout................................................................... 41
10.1 Layout Guidelines ................................................. 41
10.2 Layout Example .................................................... 44
11 器件和文档支持 ..................................................... 45
11.1 器件支持................................................................ 45
11.2 接收文档更新通知 ................................................. 45
11.3 社区资源................................................................ 45
11.4 商标....................................................................... 45
11.5 静电放电警告......................................................... 45
11.6 Glossary................................................................ 45
12 机械、封装和可订购信息....................................... 45
7
4 修订历史记录
Changes from Revision A (August 2015) to Revision B
Page
•
•
已添加 第 1 页上的 LM43601A-Q1 信息;添加了 Webench 链接.......................................................................................... 1
Added RPGOOD values for LM43601A-Q1................................................................................................................................ 6
Changes from Original (July 2015) to Revision A
Page
•
已更改 从“预览”更改为“生产数据” ........................................................................................................................................... 1
2
Copyright © 2015–2017, Texas Instruments Incorporated
LM43601-Q1
www.ti.com.cn
ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
5 Pin Configuration and Functions
PWP Package
16-Pin HTSSOP
Top View
SW
SW
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
PGND
PGND
VIN
CBOOT
VCC
VIN
PAD
EN
BIAS
SS/TRK
AGND
FB
SYNC
RT
PGOOD
Pin Functions
PIN
DESCRIPTION
(1)
NAME
NUMBER
I/O
Switching output of the regulator. Internally connected to both power MOSFETs. Connect to
power inductor.
SW
1, 2
P
Boot-strap capacitor connection for high-side driver. Connect a high quality 470-nF capacitor from
CBOOT to SW.
CBOOT
VCC
3
4
P
P
Internal bias supply output for bypassing. Connect bypass capacitor from this pin to AGND. Do not
connect external load to this pin. Never short this pin to ground during operation.
Optional internal LDO supply input. To improve efficiency, it is recommended to tie to VOUT when
3.3 V ≤ VOUT ≤ 28 V, or tie to an external 3.3-V or 5-V rail if available. When used, place a bypass
capacitor (1 to 10 µF) from this pin to ground. Tie to ground when not in use. Do not float
BIAS
5
P
Clock input to synchronize switching action to an external clock. Use proper high speed
termination to prevent ringing. Connect to ground if not used. Do not float.
SYNC
RT
6
7
8
A
A
A
Connect a resistor RT from this pin to AGND to program switching frequency. Leave floating for
500 kHz default switching frequency.
Open drain output for power-good flag. Use a 10-kΩ to 100-kΩ pullup resistor to logic rail or other
DC voltage no higher than 12 V.
PGOOD
Feedback sense input pin. Connect to the midpoint of feedback divider to set VOUT. Do not short
this pin to ground during operation.
FB
9
A
G
A
AGND
SS/TRK
10
11
Analog ground pin. Ground reference for internal references and logic. Connect to system ground.
Soft-start control pin. Leave floating for internal soft-start slew rate. Connect to a capacitor to
extend soft start time. Connect to external voltage ramp for tracking.
Enable input to the LM43601-Q1: High = ON and low = OFF. Connect to VIN, or to VIN through
resistor divider or to an external voltage or logic source. Do not float.
EN
12
A
P
Supply input pins to internal LDO and high side power FET. Connect to power supply and bypass
capacitors CIN. Path from VIN pin to high frequency bypass CIN and PGND must be as short as
possible.
VIN
13, 14
Power ground pins, connected internally to the low side power FET. Connect to system ground,
PAD, AGND, ground pins of CIN and COUT. Path to CIN must be as short as possible.
PGND
PAD
15, 16
—
G
G
Low impedance connection to AGND. Connect to PGND on PCB. Major heat dissipation path of
the die. Must be used for heat sinking to ground plane on PCB.
(1) P = Power, G = Ground, A = Analog
Copyright © 2015–2017, Texas Instruments Incorporated
3
LM43601-Q1
ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
www.ti.com.cn
6 Specifications
6.1 Absolute Maximum Ratings
Over operating free-air temperature range (unless otherwise noted)(1)
PARAMETER
VIN to PGND
MIN
–0.3
–0.3
–0.3
–0.3
–0.3
–0.3
–0.3
–0.3
–3.5
–0.3
–0.3
–65
MAX
42
UNIT
EN to PGND
VIN + 0.3
3.6
FB, RT, SS/TRK to AGND
Input voltages
PGOOD to AGND
SYNC to AGND
15
V
5.5
BIAS to AGND
30
AGND to PGND
0.3
SW to PGND
VIN + 0.3
42
SW to PGND less than 10-ns transients
CBOOT to SW
Output voltages
V
5.5
VCC to AGND
3.6
Storage temperature range, Tstg
150
°C
(1) Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under Recommended Operating
Conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
6.2 ESD Ratings
VALUE
±2000
±750
UNIT
Human-body model (HBM), per AEC Q100-002(1)
Charged-device model (CDM), per AEC Q100-011
V(ESD)
Electrostatic discharge
V
(1) AEC Q100-002 indicates that HBM stressing shall be in accordance with the ANSI/ESDA/JEDEC JS-001 specification.
6.3 Recommended Operating Conditions
Over operating free-air temperature range (unless otherwise noted)(1)
PARAMETER
VIN to PGND
MIN
MAX
UNIT
3.5
–0.3
–0.3
–0.3
–0.3
3.3
36
VIN
1.1
12
EN
FB
Input voltages
PGOOD
V
BIAS input not used
BIAS input used
AGND to PGND
VOUT
0.3
VIN or 28(2)
–0.1
1
0.1
28
Output voltage
Output current
V
A
IOUT
0
1
Operating junction temperature range, TJ
–40
125
°C
(1) Operating Ratings indicate conditions for which the device is intended to be functional, but do not ensure specific performance limits. For
ensured specifications, see Electrical Characteristics.
(2) Whichever is lower.
4
Copyright © 2015–2017, Texas Instruments Incorporated
LM43601-Q1
www.ti.com.cn
ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
6.4 Thermal Information
LM43601-Q1
THERMAL METRIC(1)(2)
PWP (HTSSOP)
UNIT
16 PINS
39.9(3)
26.9
RθJA
Junction-to-ambient thermal resistance
°C/W
°C/W
°C/W
°C/W
°C/W
°C/W
RθJC(top)
RθJB
Junction-to-case (top) thermal resistance
Junction-to-board thermal resistance
21.7
ψJT
Junction-to-top characterization parameter
Junction-to-board characterization parameter
Junction-to-case (bottom) thermal resistance
0.8
ψJB
21.5
RθJC(bot)
2.3
(1) For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
(2) The package thermal impedance is calculated in accordance with JESD 51-7 standard with a 4-layer board and 1 W power dissipation.
(3)
RθJA is highly related to PCB layout and heat sinking. See Figure 107 for measured RθJA vs PCB area from a 2-layer board and a 4-
layer board.
6.5 Electrical Characteristics
Limits apply over the recommended operating junction temperature (TJ) range of –40°C to +125°C, unless otherwise stated.
Minimum and Maximum limits are specified through test, design or statistical correlation. Typical values represent the most
likely parametric norm at TJ = 25°C, and are provided for reference purposes only. Unless otherwise stated, the following
conditions apply: VIN = 12 V, VOUT = 3.3 V, FS = 500 kHz.
PARAMETER
CONDITIONS
MIN
TYP
MAX UNIT
SUPPLY VOLTAGE (VIN PINS)
VIN-MIN-ST
ISHDN
Minimum input voltage for start-up
3.8
3.1
V
Shutdown quiescent current
VEN = 0 V
1.1
6
µA
VEN = 3.3 V
VFB = 1.5 V
VBIAS = 3.4 V external
Operating quiescent current (non-
switching) from VIN
IQ-NONSW
11
µA
µA
VEN = 3.3 V
VFB = 1.5 V
VBIAS = 3.4 V external
Operating quiescent current (non-
switching) from external VBIAS
IBIAS-NONSW
85
33
140
VEN = 3.3 V
IOUT = 0 A
RT = open
VBIAS = VOUT = 3.3 V
RFBT = 1 Meg
Operating quiescent current
(switching)
IQ-SW
µA
ENABLE (EN PIN)
Voltage level to enable the internal
LDO output VCC
VEN-VCC-H
VEN-VCC-L
VEN-VOUT-H
VENABLE high level
VENABLE low level
VENABLE high level
1.2
2
V
V
V
Voltage level to disable the internal
LDO output VCC
0.4
Precision enable level for switching
and regulator output: VOUT
2.1
2.42
Hysteresis voltage between VOUT
VEN-VOUT-HYS precision enable and disable
thresholds
VENABLE hysteresis
VEN = 3.3 V
–305
0.8
mV
µA
ILKG-EN
Enable input leakage current
1.75
INTERNAL LDO (VCC PIN AND BIAS PIN)
VCC
Internal LDO output voltage VCC
VIN ≥ 3.8 V
3.3
V
V
VCC rising threshold
3.14
Undervoltage lockout (UVLO)
thresholds for VCC
VCC-UVLO
Hysteresis voltage between rising and
falling thresholds
–567
2.96
–74
mV
V
VBIAS rising threshold
3.2
Internal LDO input change over
threshold to BIAS
VBIAS-ON
Hysteresis voltage between rising and
falling thresholds
mV
Copyright © 2015–2017, Texas Instruments Incorporated
5
LM43601-Q1
ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
www.ti.com.cn
Electrical Characteristics (continued)
Limits apply over the recommended operating junction temperature (TJ) range of –40°C to +125°C, unless otherwise stated.
Minimum and Maximum limits are specified through test, design or statistical correlation. Typical values represent the most
likely parametric norm at TJ = 25°C, and are provided for reference purposes only. Unless otherwise stated, the following
conditions apply: VIN = 12 V, VOUT = 3.3 V, FS = 500 kHz.
PARAMETER
CONDITIONS
MIN
TYP
MAX UNIT
VOLTAGE REFERENCE (FB PIN)
TJ = 25ºC
1.009
0.999
1.016
1.016
0.2
1.023
V
VFB
Feedback voltage
TJ = -40ºC to 125ºC
FB = 1.016 V
1.039
ILKG-FB
Input leakage current at FB pin
65
nA
THERMAL SHUTDOWN
Shutdown threshold
Recovery threshold
160
150
(1)
TSD
Thermal shutdown
ºC
CURRENT LIMIT AND HICCUP
IHS-LIMIT
ILS-LIMIT
SOFT START (SS/TRK PIN)
Peak inductor current limit
2.07
0.94
2.45
1.1
2.71
1.25
A
A
Valley Inductor current limit
ISSC
Soft-start charge current
Soft-start discharge resistance
1.17
2.2
16
2.85
µA
RSSD
UVLO, TSD, OCP, or EN = 0 V
kΩ
POWER GOOD (PGOOD PIN)
Power-good flag overvoltage tripping
threshold
VPGOOD-HIGH
% of FB voltage
% of FB voltage
110%
90%
113%
Power-good flag undervoltage tripping
threshold
VPGOOD-LOW
VPGOOD-HYS
83%
Power-good flag recovery hysteresis
% of FB voltage
6%
40
LM43601-Q1: VEN = 3.3 V
LM43601-Q1: VEN = 0 V
LM43601A-Q1: VEN = 3.3 V
LM43601A-Q1: VEN = 0 V
125
150
150
350
60
PGOOD pin pulldown resistance
when power bad
RPGOOD
Ω
69
150
MOSFETS(2)
IOUT = 1 A
VBIAS = VOUT = 3.3 V
RDS-ON-HS
High-side MOSFET ON-resistance
Low-side MOSFET ON-resistance
419
231
mΩ
mΩ
IOUT = 1 A
VBIAS = VOUT = 3.3 V
RDS-ON-LS
(1) Specified by design
(2) Measured at package pins
6
Copyright © 2015–2017, Texas Instruments Incorporated
LM43601-Q1
www.ti.com.cn
ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
6.6 Timing Requirements
PARAMETER
MIN
TYP
MAX
UNIT
CURRENT LIMIT AND HICCUP
NOC
TOC
Hiccup wait cycles when LS current limit tripped
Hiccup retry delay time
32
Cycles
ms
5.5
SOFT START (SS/TRK PIN)
TSS
Internal soft-start time when SS pin open circuit
3.86
ms
POWER GOOD (PGOOD PIN)
TPGOOD-RISE Power-good flag rising transition deglitch delay
TPGOOD-FALL Power-good flag falling transition deglitch delay
220
220
µs
µs
6.7 Switching Characteristics
Over operating free-air temperature range (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
SW (SW PIN)
Minimum high side MOSFET ON-
time
(1)
TON-MIN
125
200
165
250
ns
ns
Minimum high side MOSFET OFF-
time
(1)
TOFF-MIN
OSCILLATOR (SW PINS AND SYNC PIN)
FOSC-
DEFAULT
Oscillator default frequency
RT pin open circuit
445
500
570
kHz
Minimum adjustable frequency
200
2200
10%
kHz
kHz
FADJ
Maximum adjustable frequency
Frequency adjust accuracy
With 1% resistors at RT pin
VSYNC-HIGH Sync clock high level threshold
VSYNC-LOW Sync clock low level threshold
DSYNC-MAX Sync clock maximum duty cycle
DSYNC-MIN Sync clock minimum duty cycle
2
V
V
0.4
90%
10%
Minimum sync clock ON- and OFF-
TSYNC-MIN
time
80
ns
(1) Specified by design
Copyright © 2015–2017, Texas Instruments Incorporated
7
LM43601-Q1
ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
www.ti.com.cn
6.8 Typical Characteristics
Unless otherwise specified, VIN = 12 V, VOUT = 3.3 V, FS = 500 kHz, L = 18 µH, COUT = 100 µF, CFF = 33 pF. See Application
Performance Curves for Bill of Materials (BOM) for other VOUT and FS combinations.
100
90
80
70
60
50
40
30
20
10
0
100
90
80
70
60
50
40
30
20
10
0
VIN = 8V
VIN = 8V
VIN = 12V
VIN = 18V
VIN = 24V
VIN = 28V
VIN = 36V
VIN = 12V
VIN = 18V
VIN = 24V
VIN = 28V
VIN = 36V
0.001
0.010
0.100
1.000
0.001
0.010
0.100
1.000
Load Current (A)
Load Current (A)
C002
C004
VOUT = 3.3 V
FS = 500 kHz
Figure 1. Efficiency
VOUT = 5 V
FS = 200 kHz
Figure 2. Efficiency
100
90
80
70
60
50
40
30
20
10
100
90
80
70
60
50
40
30
20
10
VIN = 12V
VIN = 18V
VIN = 24V
VIN = 28V
VIN = 36V
VIN = 12V
VIN = 18V
VIN = 24V
VIN = 28V
0
0
0.001
0.010
0.100
1.000
0.001
0.010
0.100
1.000
Load Current (A)
Load Current (A)
C003
C005
VOUT = 5 V
FS = 500 kHz
VOUT = 5 V
FS = 1 MHz
Figure 3. Efficiency
Figure 4. Efficiency
90
80
70
60
50
40
30
20
10
100
90
80
70
60
50
40
30
20
10
VIN = 24V
VIN = 28V
VIN = 36V
VIN = 12V
0
0
0.001
0.010
0.100
1.000
0.001
0.010
0.100
1.000
Load Current (A)
Load Current (A)
C006
C007
VOUT = 5 V
FS = 2.2 MHz
VOUT = 12 V
FS = 500 kHz
Figure 5. Efficiency
Figure 6. Efficiency
8
Copyright © 2015–2017, Texas Instruments Incorporated
LM43601-Q1
www.ti.com.cn
ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
Typical Characteristics (continued)
Unless otherwise specified, VIN = 12 V, VOUT = 3.3 V, FS = 500 kHz, L = 18 µH, COUT = 100 µF, CFF = 33 pF. See Application
Performance Curves for Bill of Materials (BOM) for other VOUT and FS combinations.
3.40
3.38
3.36
3.34
3.32
3.30
3.28
3.26
3.24
3.22
3.20
5.15
5.10
5.05
5.00
4.95
4.90
4.85
4.80
VIN = 8V
VIN = 18V
VIN = 28V
VIN = 12V
VIN = 24V
VIN = 36V
VIN = 8V
VIN = 12V
VIN = 28V
VIN = 18V
VIN = 36V
VIN = 24V
0.001
0.010
0.100
1.000
0.001
0.010
0.100
1.000
Load Current (A)
Load Current (A)
C012
C014
VOUT = 3.3 V
FS = 500 kHz
Figure 7. VOUT Regulation
VOUT = 5 V
FS = 200 kHz
Figure 8. VOUT Regulation
5.20
5.15
5.10
5.05
5.00
4.95
4.90
4.85
5.20
5.15
5.10
5.05
5.00
4.95
4.90
4.85
VIN = 12V
VIN = 18V
VIN = 24V
VIN = 28V
VIN = 36V
VIN = 12V
VIN = 18V
VIN = 24V
VIN = 28V
4.80
4.80
0.001
0.010
0.100
1.000
0.001
0.010
0.100
1.000
Load Current (A)
Load Current (A)
C013
C015
VOUT = 5 V
FS = 500 kHz
Figure 9. VOUT Regulation
VOUT = 5 V
FS = 1 MHz
Figure 10. VOUT Regulation
5.20
5.15
5.10
5.05
5.00
4.95
4.90
4.85
12.5
12.4
12.3
12.2
12.1
12.0
11.9
11.8
11.7
11.6
VIN = 24V
VIN = 28V
VIN = 36V
VIN = 12V
4.80
11.5
0.001
0.010
0.100
1.000
0.001
0.010
0.100
1.000
Load Current (A)
Load Current (A)
C016
C017
VOUT = 5 V
FS = 2.2 MHz
Figure 11. VOUT Regulation
VOUT = 12 V
FS = 500 kHz
Figure 12. VOUT Regulation
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ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
www.ti.com.cn
Typical Characteristics (continued)
Unless otherwise specified, VIN = 12 V, VOUT = 3.3 V, FS = 500 kHz, L = 18 µH, COUT = 100 µF, CFF = 33 pF. See Application
Performance Curves for Bill of Materials (BOM) for other VOUT and FS combinations.
3.50
3.40
3.30
3.20
3.10
3.00
2.90
2.80
2.70
2.60
2.50
5.2
5.0
4.8
4.6
4.4
4.2
4.0
Load = 0.25A
Load = 0.5A
Load = 0.75A
Load = 1A
Load = 0.25A
Load = 0.5A
Load = 0.75A
Load = 1A
3.5
4.0
4.5
5.0
5.0
5.5
6.0
6.5
VIN (V)
VIN (V)
C022
C024
VOUT = 3.3 V
FS = 500 kHz
VOUT = 5 V
FS = 200 kHz
Figure 13. Dropout Curve
Figure 14. Dropout Curve
5.2
5.0
4.8
4.6
4.4
4.2
5.2
5.0
4.8
4.6
4.4
4.2
Load = 0.25A
Load = 0.5A
Load = 0.75A
Load = 1A
Load = 0.25A
Load = 0.5A
Load = 0.75A
Load = 1A
4.0
5.0
4.0
5.0
5.5
6.0
6.5
5.5
6.0
6.5
VIN (V)
VIN (V)
C023
C025
VOUT = 5 V
FS = 500 kHz
VOUT = 5 V
FS = 1 MHz
Figure 15. Dropout Curve
Figure 16. Dropout Curve
5.2
5.0
4.8
4.6
4.4
4.2
12.4
12.2
12.0
11.8
11.6
11.4
11.2
Load = 0.25A
Load = 0.5A
Load = 0.75A
Load = 1A
Load = 0.25A
Load = 0.5A
Load = 0.75A
Load = 1A
4.0
5.0
11.0
12.0
5.5
6.0
6.5
12.5
13.0
13.5
14.0
VIN (V)
VIN (V)
C026
C027
VOUT = 5 V
FS = 2.2 MHz
VOUT = 12 V
FS = 500 kHz
Figure 17. Dropout Curve
Figure 18. Dropout Curve
10
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Typical Characteristics (continued)
Unless otherwise specified, VIN = 12 V, VOUT = 3.3 V, FS = 500 kHz, L = 18 µH, COUT = 100 µF, CFF = 33 pF. See Application
Performance Curves for Bill of Materials (BOM) for other VOUT and FS combinations.
1000000
1000000
100000
100000
Load = 0.01 A
Load = 0.1 A
Load = 0.5 A
Load = 1 A
Load = 0.01 A
Load = 0.1 A
Load = 0.5 A
Load = 1 A
10000
10000
3.4
3.6
3.8
4.0
4.2
4.4
4.6
4.8
5.0
5.0 5.2 5.4 5.6 5.8 6.0 6.2 6.4 6.6 6.8 7.0
VIN (V)
VIN (V)
C001
C001
VOUT = 3.3 V
FS = 500 kHz
VOUT = 5 V
FS = 1 MHz
Figure 19. Switching Frequency vs VIN in Dropout Operation
Figure 20. Switching Frequency vs VIN in Dropout Operation
dBuV
80
dBuV
80
Vertical Polarization
Vertical Polarization
70
70
Horizontal Polarization
Horizontal Polarization
60
50
60
50
EN 55022 Class B Limit
40
EN 55022 Class B Limit
40
30
20
30
20
10
10
Evaluation Board Emissions
Evaluation Board Emissions
30
100
Frequency (MHz)
1000
30
100
Frequency (MHz)
1000
VOUT = 3.3 V
FS = 500 kHz
IOUT = 1 A
VOUT = 5 V
FS = 500 kHz
IOUT = 1 A
Measured on the LM43601QPWPEVM with default BOM. No input
filter used.
Measured on the LM43601QPWPEVM with L = 27 µH, COUT = 66
µF, CFF = 33 pF. No input filter used.
Figure 21. Radiated EMI Curve
Figure 22. Radiated EMI Curve
dBuV
100
dBuV
100
90
80
90
80
70
70
Quasi Peak Limit
Quasi Peak Limit
60
60
Average Limit
Average Limit
50
50
40
30
40
30
20
20
10
10
Measured Peak Emissions
Measured Peak Emissions
10
30
10
30
0.15
1
0.15
1
Frequency (MHz)
Frequency (MHz)
VOUT = 3.3 V
FS = 500 kHz
IOUT = 1 A
VOUT = 5 V
FS = 500 kHz
IOUT = 1 A
Measured on the LM43601QPWPEVM with default BOM. EVM
Input filter: Lin = 1 µH Cd = 47 µF CIN4 = 68 µF
Measured on the LM43601QPWPEVM with L = 18 µH, COUT = 66
µF, CFF = 33 pF. Input filter Lin = 1 µH Cd = 47 µF CIN4 = 68 µF
Figure 23. Conducted EMI Curve
Figure 24. Conducted EMI Curve
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Typical Characteristics (continued)
Unless otherwise specified, VIN = 12 V, VOUT = 3.3 V, FS = 500 kHz, L = 18 µH, COUT = 100 µF, CFF = 33 pF. See Application
Performance Curves for Bill of Materials (BOM) for other VOUT and FS combinations.
4
3.5
3
800
700
600
500
400
300
200
100
0
2.5
2
1.5
1
HS
LS
VIN = 12V
VIN = 24V
0.5
0
-50
0
50
100
150
-50
0
50
100
150
Temperature (°C)
Temperature (°C)
C001
C001
Figure 25. High-Side and Low-Side On-Resistance vs
Junction Temperature
Figure 26. Shutdown Current vs Junction Temperature
2.5
1.4
1.2
1
2
1.5
1
EN-VOUT Rising TH
EN-VOUT Falling TH
EN-VCC Rising TH
EN-VCC Falling TH
0.8
0.6
0.4
0.5
0
0.2
VEN = 3.3V
0
-50
0
50
100
150
-50
0
50
100
150
Temperature (°C)
Temperature (°C)
C001
C001
Figure 27. Enable Threshold vs Junction Temperature
Figure 28. Enable Leakage Current vs
Junction Temperature
120%
1.030
1.025
1.020
1.015
1.010
1.005
1.000
0.995
0.990
115%
110%
105%
100%
95%
90%
OVP Trip Level
85%
VIN = 12V
VIN = 24V
OVP Recover Level
UVP Recover Level
UVP Trip Level
80%
75%
-50
0
50
Temperature (°C)
100
150
-50
0
50
100
150
Temperature (°C)
C001
C001
Figure 29. PGOOD Threshold vs Junction Temperature
Figure 30. Feedback Voltage vs Junction Temperature
12
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ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
Typical Characteristics (continued)
Unless otherwise specified, VIN = 12 V, VOUT = 3.3 V, FS = 500 kHz, L = 18 µH, COUT = 100 µF, CFF = 33 pF. See Application
Performance Curves for Bill of Materials (BOM) for other VOUT and FS combinations.
3.0
2.5
2.0
1.5
1.0
0.5
0.0
70
60
50
40
30
20
10
0
IL Peak Limit
IL Valley Limit
-50
0
50
100
150
0
10
20
30
40
Temperature (°C)
VOUT = 3.3 V
VIN (V)
C001
C001
VIN = 12 V
FS = 500 kHz
VOUT = 3.3 V
FS = 500 kHz
IOUT = 0 A
Figure 31. Peak and Valley Current Limits vs Junction
Temperature
Figure 32. Operating IQ vs VIN With BIAS Connected to VOUT
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7 Detailed Description
7.1 Overview
The LM43601-Q1 regulator is an easy-to-use synchronous step-down DC-DC converter that operates from 3.5 V
to 36 V supply voltage. It is capable of delivering up to 1-A DC load current with exceptional efficiency and
thermal performance in a very small solution size. An extended family is available in 0.5-A , 2-A, and 3-A load
options in pin-to-pin compatible packages.
The LM43601-Q1 employs fixed frequency peak current mode control with discontinuous conduction mode
(DCM) and pulse frequency modulation (PFM) mode at light load to achieve high efficiency across the load
range. The device is internally compensated, which reduces design time, and requires fewer external
components. The switching frequency is programmable from 200 kHz to 2.2 MHz by an external resistor, RT. It
defaults at 500 kHz without RT. The LM43601-Q1 is also capable of synchronization to an external clock within
the 200-kHz to 2.2-MHz frequency range. The wide switching frequency range allows the device to be optimized
to fit small board space at higher frequency, or high efficient power conversion at lower frequency.
Optional features are included for more comprehensive system requirements, including power-good (PGOOD)
flag, precision enable, synchronization to external clock, extendable soft-start time, and output voltage tracking.
These features provide a flexible and easy to use platform for a wide range of applications. Protection features
include over temperature shutdown, VCC undervoltage lockout (UVLO), cycle-by-cycle current limit, and short-
circuit protection with hiccup mode.
The family requires few external components and the pin arrangement was designed for simple, optimum PCB
layout. The LM43601-Q1 device is available in the 16-pin HTSSOP / PWP package (6.6 mm × 5.1 mm × 1.2
mm) with 0.65-mm lead pitch.
7.2 Functional Block Diagram
ENABLE
VCC
BIAS
LDO
VCC
Enable
VIN
Internal
SS
ISSC
Precision
Enable
CBOOT
SS/TRK
HS I Sense
+
EA
+
REF
œ
RC
CC
+ œ
TSD
UVLO
SW
PWM CONTROL LOGIC
PFM
PGood
Detector
PGOOD
OV/UV
Detector
Slope
Comp
FB
HICCUP
Cross Detector
Freq
Foldback
Zero
Oscillator
LS I Sense
AGND
FB
PGood
SYNC
RT
PGND
14
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7.3 Feature Description
7.3.1 Fixed Frequency Peak Current Mode Controlled Step-Down Regulator
The following operating description of the LM43601-Q1 refers to the Functional Block Diagram and to the
waveforms in Figure 33. The LM43601-Q1 is a step-down buck regulator with both high-side (HS) switch and
low-side (LS) switch (synchronous rectifier) integrated. The LM43601-Q1 supplies a regulated output voltage by
turning on the HS and LS NMOS switches with controlled ON time. During the HS switch ON-time, the SW pin
voltage VSW swings up to approximately VIN, and the inductor current IL increases with a linear slope (VIN – VOUT
)
/ L. When the HS switch is turned off by the control logic, the LS switch is turned on after a anti-shoot-through
dead time. Inductor current discharges through the LS switch with a slope of –VOUT / L. The control parameter of
Buck converters are defined as duty cycle D = tON / TSW, where tON is the HS switch ON time and TSW is the
switching period. The regulator control loop maintains a constant output voltage by adjusting the duty cycle D. In
an ideal Buck converter, where losses are ignored, D is proportional to the output voltage and inversely
proportional to the input voltage: D = VOUT / VIN.
V
SW
D = t
ON
/ T
SW
V
IN
t
t
OFF
ON
0
D1
t
-V
T
SW
iL
I
I
LPK
OUT
ûi
L
0
t
Figure 33. SW Node and Inductor Current Waveforms in Continuous Conduction Mode (CCM)
The LM43601-Q1 synchronous buck converter employs peak current mode control topology. A voltage feedback
loop is used to get accurate DC voltage regulation by adjusting the peak current command based on voltage
offset. The peak inductor current is sensed from the HS switch and compared to the peak current to control the
ON-time of the HS switch. The voltage feedback loop is internally compensated, which allows for fewer external
components, makes it easy to design, and provides stable operation with almost any combination of output
capacitors. The regulator operates with fixed switching frequency in continuous conduction mode (CCM) and
discontinuous conduction mode (DCM). At very light load, the LM43601-Q1 will operate in PFM to maintain high
efficiency and the switching frequency will decrease with reduced load current.
7.3.2 Light Load Operation
DCM operation is employed in the LM43601-Q1 when the inductor current valley reaches zero. The LM43601-Q1
is in DCM when load current is less than half of the peak-to-peak inductor current ripple in CCM. In DCM, the LS
switch is turned off when the inductor current reaches zero. Switching loss is reduced by turning off the LS FET
at zero current and the conduction loss is lowered by not allowing negative current conduction. Power conversion
efficiency is higher in DCM than CCM under the same conditions.
In DCM, the HS switch ON time will reduce with lower load current. When either the minimum HS switch ON-time
(TON-MIN) or the minimum peak inductor current (IPEAK-MIN) is reached, the switching frequency will decrease to
maintain regulation. At this point, the LM43601-Q1 operates in PFM. In PFM, switching frequency is decreased
by the control loop when load current reduces to maintain output voltage regulation. Switching loss is further
reduced in PFM operation due to less frequent switching actions. Figure 34 shows an example of switching
frequency decreases with decreased load current.
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Feature Description (continued)
1000000
100000
VIN = 12 V
VIN = 18 V
VIN = 24 V
VIN = 36 V
10000
0.001
0.01
0.1
1
Load (A)
C001
Figure 34. Switching Frequency Decreases With Lower Load Current in PFM Operation
VOUT = 5 V FS = 1 MHz
In PFM operation, a small positive DC offset is required at the output voltage to activate the PFM detector. The
lower the frequency in PFM, the more DC offset is needed at VOUT. See Typical Characteristics for typical DC
offset at very light load. If the DC offset on VOUT is not acceptable for a given application, a static load at output is
recommended to reduce or eliminate the offset. Lowering values of the feedback divider RFBT and RFBB can also
serve as a static load. In conditions with low VIN and/or high frequency, the LM43601-Q1 may not enter PFM
mode if the output voltage cannot be charged up to provide the trigger to activate the PFM detector. Once the
LM43601-Q1 is operating in PFM mode at higher VIN, it will remain in PFM operation when VIN is reduced.
7.3.3 Adjustable Output Voltage
The voltage regulation loop in the LM43601-Q1 regulates output voltage by maintaining the voltage on FB pin
(VFB) to be the same as the internal REF voltage (VREF). A resistor divider pair is needed to program the ratio
from output voltage VOUT to VFB. The resistor divider is connected from the VOUT of the LM43601-Q1 to ground
with the mid-point connecting to the FB pin.
VOUT
RFBT
FB
RFBB
Figure 35. Output Voltage Setting
The voltage reference system produces a precise voltage reference over temperature. The internal REF voltage
is 1.016 V typically. To program the output voltage of the LM43601-Q1 to be a certain value VOUT, RFBB can be
calculated with a selected RFBT by
VFB
RFBB
=
RFBT
VOUT - VFB
(1)
The choice of the RFBT depends on the application. RFBT in the range from 10 kΩ to 100 kΩ is recommended for
most applications. A lower RFBT value can be used if static loading is desired to reduce VOUT offset in PFM
operation. Lower RFBT will reduce efficiency at very light load. Less static current goes through a larger RFBT and
might be more desirable when light load efficiency is critical. But RFBT larger than 1 MΩ is not recommended
because it makes the feedback path more susceptible to noise. Larger RFBT value requires more carefully
designed feedback path on the PCB. The tolerance and temperature variation of the resistor dividers affect the
output voltage regulation. It is recommended to use divider resistors with 1% tolerance or better and temperature
coefficient of 100 ppm or lower.
16
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ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
Feature Description (continued)
If the resistor divider is not connected properly, the output voltage cannot be regulated since the feedback loop is
broken. If the FB pin is shorted to ground, the output voltage will be driven close to VIN, since the regulator sees
very low voltage on the FB pin and tries to regulate it up. The load connected to the output could be damaged
under such a condition. Do not short FB pin to ground when the LM43601-Q1 is enabled. It is important to route
the feedback trace away from the noisy area of the PCB. For more layout recommendations, see the Layout
section.
7.3.4 ENABLE Pin
Voltage on the ENABLE pin (VEN) controls the ON or OFF functionality of the LM43601-Q1. Applying a voltage
less than 0.4 V to the ENABLE input shuts down the operation of the LM43601-Q1. In shutdown mode the
quiescent current drops to typically 1 µA at VIN = 12 V.
The internal LDO output voltage VCC is turned on when VEN is higher than 1.2 V. The switching action and output
regulation are enabled when VEN is greater than 2.1 V (typical). The LM43601-Q1 supplies regulated output
voltage when enabled and output current up to 1 A.
The ENABLE pin is an input and cannot be open circuit or floating. The simplest way to enable the operation of
the LM43601-Q1 is to connect the ENABLE pin to VIN pins directly. This allows self-start-up when VIN is within
the operation range.
Many applications will benefit from the employment of an enable divider RENT and RENB in Figure 36 to establish
a precision system UVLO level for the stage. System UVLO can be used for supplies operating from utility power
as well as battery power. It can be used for sequencing, ensuring reliable operation, or supply protection, such
as a battery discharge voltage level. An external logic signal can also be used to drive EN input for system
sequencing and protection.
VIN
RENT
ENABLE
RENB
Figure 36. System UVLO By Enable Dividers
7.3.5 VCC, UVLO and BIAS
The LM43601-Q1 integrates an internal LDO to generate VCC for control circuitry and MOSFET drivers. The
nominal voltage for VCC is 3.3 V. The VCC pin is the output of the LDO and must be properly bypassed. Place a
high-quality ceramic capacitor with 2.2-µF to 10-µF capacitance and 6.3-V or higher rated voltage as close as
possible to VCC and grounded to the exposed PAD and ground pins. The VCC output pin must not be loaded,
left floating, or shorted to ground during operation. Shorting VCC to ground during operation may cause damage
to the LM43601-Q1.
Undervoltage lockout (UVLO) prevents the LM43601-Q1 from operating until the VCC voltage exceeds 3.14 V
(typical). The VCC UVLO threshold has 567 mV of hysteresis (typically) to prevent undesired shuting down due to
temperary VIN droops.
The internal LDO has two inputs: primary from VIN and secondary from BIAS input. The BIAS input powers the
LDO when VBIAS is higher than the change-over threshold. Power loss of an LDO is calculated by ILDO × (VIN-LDO
-
VOUT-LDO). The higher the difference between the input and output voltages of the LDO, the more power loss
occur to supply the same output current. The BIAS input is designed to reduce the difference of the input and
output voltages of the LDO to reduce power loss and improve LM43601-Q1 efficiency, especially at light load. TI
recommends that the BIAS pin be tied to VOUT when VOUT ≥ 3.3V. Ground the BIAS pin in applications with VOUT
less than 3.3 V. BIAS input can also come from an external voltage source, if available, to reduce power loss.
When used, a 1-µF to 1-0µF high-quality ceramic capacitor is recommended to bypass the BIAS pin to ground.
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Feature Description (continued)
7.3.6 Soft Start and Voltage Tracking (SS/TRK)
The LM43601-Q1 has a flexible and easy-to-use start-up rate control pin: SS/TRK. The soft-start feature is to
prevent inrush current impacting the LM43601-Q1 and its supply when power is first applied. Soft start is
achieved by slowly ramping up the target regulation voltage when the device is first enabled or powered up.
The simplest way to use the device is to leave the SS/TRK pin open circuit. The LM43601-Q1 employs the
internal soft-start control ramp and start-up to the regulated output voltage in 4.1 ms typically.
In applications with a large amount of output capacitors, or higher VOUT, or other special requirements, the soft-
start time can be extended by connecting an external capacitor CSS from SS/TRK pin to AGND. Extended soft-
start time further reduces the supply current needed to charge up output capacitors and supply any output
loading. An internal current source (ISSC = 2.2 µA) charges CSS and generates a ramp from 0 V to VFB to control
the ramp-up rate of the output voltage. For a desired soft start time tSS, the capacitance for CSS can be found by
CSS = ISSC ì tSS
(2)
The soft start capacitor CSS is discharged by an internal FET when VOUT is shut down by hiccup protection due to
excessive load, temperature shutdown due to overheating or ENABLE = logic low. A large CSS capacitor will take
a long time to discharge when ENABLE is toggled low. If ENABLE is toggled high again before the CSS is
completely discharged, then the next resulting soft-start ramp follows the internal soft-start ramp. Only when the
soft-start voltage reaches the leftover voltage on CSS, does the output follow the ramp programmed by CSS. This
behavior looks as if there are two slopes at start-up. If this is not acceptable by a certain application, a R-C low
pass filter can be added to ENABLE to slow down the shutting down of VCC, which allows more time to
discharge CSS
The LM43601-Q1 is capable of start-up into prebiased output conditions. When the inductor current reaches
zero, the LS switch will be turned off to avoid negative current conduction. This operation mode is also called
diode emulation mode. It is built-in by the DCM operation at light loads. With a prebiased output voltage, the
LM43601-Q1 will wait until the soft-start ramp allows regulation above the prebiased voltage. It will then follow
the soft-start ramp to the regulation level.
When an external voltage ramp is applied to the SS/TRK pin, the LM43601-Q1 FB voltage follows the external
ramp if the ramp magnitude is lower than the internal soft-start ramp. A resistor divider pair can be used on the
external control ramp to the SS/TRK pin to program the tracking rate of the output voltage. The final external
ramp voltage applied at the SS/TRK pin should not fall below 1.2 V to avoid abnormal operation.
EXT RAMP
RTRT
SS/TRK
RTRB
Figure 37. Soft-Start Tracking External Ramp
VOUT tracked to an external voltage ramp has the option of ramping up slower or faster than the internal voltage
ramp. VFB always follows the lower potential of the internal voltage ramp and the voltage on the SS/TRK pin.
Figure 38 shows the case when VOUT ramps slower than the internal ramp, while Figure 39 shows when VOUT
ramps faster than the internal ramp. Faster start-up time may result in inductor current tripping current protection
during start-up. Use with special care.
Enable
Internal SS Ramp
Ext Tracking Signal to SS pin
VOUT
Figure 38. Tracking With Longer Start-up Time Than the Internal Ramp
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ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
Feature Description (continued)
Enable
Internal SS Ramp
Ext Tracking Signal to SS pin
VOUT
Figure 39. Tracking With Shorter Start-up Time Than the Internal Ramp
7.3.7 Switching Frequency (RT) and Synchronization (SYNC)
The switching frequency of the LM43601-Q1 can be programmed by the impedance RT from the RT pin to
ground. The frequency is inversely proportional to the RT resistance. The RT pin can be left floating, and the
LM43601-Q1 will operate at 500-kHz default switching frequency. The RT pin is not designed to be shorted to
ground.
For a desired frequency, typical RT resistance can be found by Equation 3.
RT(kΩ) = 40200 / Freq (kHz) – 0.6
(3)
Figure 40 shows RT resistance vs switching frequency FS curve.
250
200
150
100
50
0
0
500
1000
1500
2000
2500
Switching Frequency (kHz)
C008
Figure 40. RT Resistance vs Switching Frequency
Table 1 provides typical RT values for a given FS.
Table 1. Typical Frequency Setting RT Resistance
FS (kHz)
RT (kΩ)
200
200
350
115
500
80.6
53.6
39.2
26.1
19.6
17.8
750
1000
1500
2000
2200
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Feature Description (continued)
The LM43601-Q1 switching action can also be synchronized to an external clock from 200 kHz to 2.2 MHz.
Connect an external clock to the SYNC pin, with proper high speed termination, to avoid ringing. The SYNC pin
should be grounded if not used.
SYNC
EXT CLOCK
RTERM
Figure 41. Frequency Synchronization
The recommendations for the external clock include high level no lower than 2 V, low level no higher than 0.4 V,
duty cycle between 10% and 90%, and both positive and negative pulse width no shorter than 80 ns. When the
external clock fails at logic high or low, the LM43601-Q1 switches at the frequency programmed by the RT
resistor after a time-out period. TI recommends connecting a resistor RT to the RT pin so that the internal
oscillator frequency is the same as the target clock frequency when the LM43601-Q1 is synchronized to an
external clock. This allows the regulator to continue operating at approximately the same switching frequency if
the external clock fails.
The choice of switching frequency is usually a compromise between conversion efficiency and the size of the
circuit. Lower switching frequency implies reduced switching losses (including gate charge losses, switch
transition losses, etc.) and usually results in higher overall efficiency. However, higher switching frequency allows
use of smaller LC output filters and hence a more compact design. Lower inductance also helps transient
response (higher large signal slew rate of inductor current), and reduces the DCR loss. The optimal switching
frequency is usually a trade-off in a given application and thus needs to be determined on a case-by-case basis.
It is related to the input voltage, output voltage, most frequent load current level(s), external component choices,
and circuit size requirement. The choice of switching frequency may also be limited if an operating condition
triggers TON-MIN or TOFF-MIN
.
7.3.8 Minimum ON-Time, Minimum OFF-Time and Frequency Foldback at Drop-Out Conditions
Minimum ON-time, TON-MIN, is the smallest duration of time that the HS switch can be on. TON-MIN is typically 125
ns in the LM43601-Q1. Minimum OFF-time, TOFF-MIN, is the smallest duration that the HS switch can be off. TOFF-
MIN is typically 200 ns in the LM43601-Q1.
In CCM operation, TON-MIN and TOFF-MIN limits the voltage conversion range given a selected switching frequency.
The minimum duty cycle allowed is
DMIN = TON-MIN × FS
(4)
And the maximum duty cycle allowed is
DMAX = 1 – TOFF-MIN × FS
(5)
Given fixed TON-MIN and TOFF-MIN, the higher the switching frequency the narrower the range of the allowed duty
cycle. In the LM43601-Q1, frequency foldback scheme is employed to extend the maximum duty cycle when
TOFF-MIN is reached. The switching frequency decreases once longer duty cycle is needed under low VIN
conditions. The switching frequency can be decreased to approximately 1/10 of the programmed frequency by RT
or the synchronization clock. Such wide range of frequency foldback allows the LM43601-Q1 output voltage to
stay in regulation with a much lower supply voltage VIN. This leads to a lower effective dropout voltage. See
Typical Characteristics for more details.
Given an output voltage, the choice of the switching frequency affects the allowed input voltage range, solution
size and efficiency. The maximum operating supply voltage can be found by
VIN-MAX = VOUT / (FS × TON-MIN
)
(6)
At lower supply voltage, the switching frequency decreases once TOFF-MIN is tripped. The minimum VIN without
frequency foldback can be approximated by
VIN-MIN = VOUT / (1 – FS × TOFF-MIN
)
(7)
Taking considerations of power losses in the system with heavy load operation, VIN-MIN is higher than the result
calculated in Equation 7. With frequency foldback, VIN-MIN is lowered by decreased FS. Figure 42 gives an
example of how FS decreases with decreasing supply voltage VIN at dropout operation.
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Feature Description (continued)
1000000
100000
10000
Load = 0.01 A
Load = 0.1 A
Load = 0.5 A
Load = 1 A
5.0 5.2 5.4 5.6 5.8 6.0 6.2 6.4 6.6 6.8 7.0
VIN (V)
C001
Figure 42. Switching Frequency Decreases in Dropout Operation
VOUT = 5 V, FS = 1 MHz
7.3.9 Internal Compensation and CFF
The LM43601-Q1 is internally compensated with RC = 400 kΩ and CC = 50 pF as shown in Functional Block
Diagram. The internal compensation is designed such that the loop response is stable over the entire operating
frequency and output voltage range. Depending on the output voltage, the compensation loop phase margin can
be low with all ceramic capacitors. TI recommends an external feed-forward capacitor CFF be placed in parallel
with the top resistor divider RFBT for optimum transient performance as shown in Figure 43.
VOUT
RFBT
CFF
FB
RFBB
Figure 43. Feed-Forward Capacitor for Loop Compensation
The feed-forward capacitor CFF in parallel with RFBT places an additional zero before the crossover frequency of
the control loop to boost phase margin. The zero frequency can be found by
fZ-CFF = 1 / ( 2π × RFBT × CFF ).
(8)
An additional pole is also introduced with CFF at the frequency of
fP-CFF = 1 / ( 2π × CFF × ( RFBT // RFBB )).
(9)
The CFF should be selected such that the bandwidth of the control loop without the CFF is centered between fZ-CFF
and fP-CFF. The zero fZ-CFF adds phase boost at the crossover frequency and improves transient response. The
pole fP-CFF helps maintaining proper gain margin at frequency beyond the crossover.
Designs with different combinations of output capacitors need different CFF. Different types of capacitors have
different equivalent series resistance (ESR). Ceramic capacitors have the smallest ESR and need the most CFF.
Electrolytic capacitors have much larger ESR and the ESR zero frequency
fZ-ESR = 1 / ( 2π × ESR × COUT
)
(10)
would be low enough to boost the phase up around the crossover frequency. Designs using mostly electrolytic
capacitors at the output may not need any CFF.
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Feature Description (continued)
The CFF creates a time constant with RFBT that couples in the attenuated output voltage ripple to the FB node. If
the CFF value is too large, it can couple too much ripple to the FB and affect VOUT regulation. It could also couple
too much transient voltage deviation and falsely trip PGOOD thresholds. Therefore, CFF should be calculated
based on output capacitors used in the system. At cold temperatures, the value of CFF might change based on
the tolerance of the chosen component. This may reduce its impedance and ease noise coupling on the FB
node. To avoid this, more capacitance can be added to the output or the value of CFF can be reduced. Please
refer to the Detailed Design Procedure for the calculation of CFF.
7.3.10 Bootstrap Voltage (BOOT)
The driver of the HS switch requires a bias voltage higher than VIN when the HS switch is ON. The capacitor
connected between CBOOT and SW pins works as a charge pump to boost voltage on the CBOOT pin to (VSW
+
VCC). The boot diode is integrated on the LM43601-Q1 die to minimize the bill of material (BOM). A synchronous
switch is also integrated in parallel with the boot diode to reduce voltage drop on CBOOT. A high-quality ceramic
0.47-µF, 6.3-V or higher capacitor is recommended for CCBOOT
.
7.3.11 Power Good (PGOOD)
The LM43601-Q1 has a built-in power-good flag shown on PGOOD pin to indicate whether the output voltage is
within its regulation level. The PGOOD signal can be used for start-up sequencing of multiple rails or fault
protection. The PGOOD pin is an open-drain output that requires a pullup resistor to an appropriate DC voltage.
Voltage detected by the PGOOD pin must never exceed 12 V. A resistor divider pair can be used to divide the
voltage down from a higher potential. A typical range of pullup resistor value is 10 kΩ to 100 kΩ.
When the FB voltage is within the power-good band, +4% above and –7% below the internal reference VREF
typically, the PGOOD switch is turned off and the PGOOD voltage is pulled up to the voltage level defined by the
pullup resistor or divider. When the FB voltage is outside of the tolerance band, +10 % above or –10 % below
VREF typically, the PGOOD switch is turned on and the PGOOD pin voltage is pulled low to indicate power bad.
Both rising and falling edges of the power-good flag have a built-in 220-µs (typical) deglitch delay.
7.3.12 Overcurrent and Short-Circuit Protection
The LM43601-Q1 is protected from overcurrent conditions by cycle-by-cycle current limiting on both peak and
valley of the inductor current. Hiccup mode is activated to prevent over heating if a fault condition persists.
High-side MOSFET over-current protection is implemented by the nature of the peak current mode control. The
HS switch current is sensed when the HS is turned on after a set blanking time. The HS switch current is
compared to the output of the error amplifier (EA) minus slope compensation every switching cycle. See
Functional Block Diagram for more details. The peak current of the HS switch is limited by the maximum EA
output voltage minus the slope compensation at every switching cycle. The slope compensation magnitude at the
peak current is proportional to the duty cycle.
When the LS switch is turned on, the current going through it is also sensed and monitored. The LS switch is not
turned OFF at the end of a switching cycle if its current is above the LS current limit ILS-LIMIT. The LS switch is
kept ON so that inductor current keeps ramping down, until the inductor current ramps below ILS-LIMIT. Then the
LS switch is turned OFF and the HS switch is turned on after a dead time. If the current of the LS switch is higher
than the LS current limit for 32 consecutive cycles and the power-good flag is low, hiccup current protection
mode will be activated. In hiccup mode, the regulator is shut down and kept off for 5.5 ms typically before the
LM43601-Q1 tries to start again. If overcurrent or short-circuit fault condition still exist, hiccup will repeat until the
fault condition is removed. Hiccup mode reduces power dissipation under severe overcurrent conditions,
prevents over heating and potential damage to the device.
Hiccup is only activated when power-good flag is low. Under non-severe overcurrent conditions when VOUT has
not fallen outside of the PGOOD tolerance band, the LM43601-Q1 reduces the switching frequency and keep the
inductor current valley clamped at the LS current limit level. This operation mode allows slight over current
operation during load transients without tripping hiccup. If the power-good flag becomes low, hiccup operation
starts after LS current limit is tripped 32 consecutive cycles.
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Feature Description (continued)
7.3.13 Thermal Shutdown
Thermal shutdown is a built-in self protection to limit junction temperature and prevent damages due to over
heating. Thermal shutdown turns off the device when the junction temperature exceeds 160°C typically to
prevent further power dissipation and temperature rise. Junction temperature will reduce after thermal shutdown.
The LM43601-Q1 attempts to restart when the junction temperature drops to 150°C.
7.4 Device Functional Modes
7.4.1 Shutdown Mode
The EN pin provides electrical ON and OFF control for the LM43601-Q1. When VEN is below 0.4 V, the device is
in shutdown mode. Both the internal LDO and the switching regulator are off. In shutdown mode the quiescent
current drops to 1 µA typically with VIN = 12 V. The LM43601-Q1 also employs under voltage lock out protection.
If VCC voltage is below the UVLO level, the output of the regulator is turned off.
7.4.2 Standby Mode
The internal LDO has a lower enable threshold than the regulator. When VEN is above 1.2 V and below the
precision enable falling threshold (1.8 V typically), the internal LDO regulates the VCC voltage at 3.2 V. The
precision enable circuitry is turned on once VCC is above the UVLO threshold. The switching action and voltage
regulation are not enabled unless VEN rises above the precision enable threshold (2.1 V typically).
7.4.3 Active Mode
The LM43601-Q1 is in active mode when VEN is above the precision enable threshold and VCC is above its UVLO
level. The simplest way to enable the LM43601-Q1 is to connect the EN pin to VIN. This allows self start-up when
the input voltage is in the operation range: 3.5 V to 36 V. See ENABLE Pin and VCC, UVLO and BIAS for details
on setting these operating levels.
In Active Mode, depending on the load current, the LM43601-Q1 is in one of four modes:
1. Continuous conduction mode (CCM) with fixed switching frequency when load current is above half of the
peak-to-peak inductor current ripple;
2. Discontinuous conduction mode (DCM) with fixed switching frequency when load current is lower than half of
the peak-to-peak inductor current ripple in CCM operation;
3. Pulse frequency modulation (PFM) when switching frequency is decreased at very light load;
4. Foldback mode when switching frequency is decreased to maintain output regulation at lower supply voltage
VIN.
7.4.4 CCM Mode
CCM operation is employed in the LM43601-Q1 when the load current is higher than half of the peak-to-peak
inductor current. In CCM peration, the frequency of operation is fixed unless the the minimum HS switch ON-time
(TON_MIN), the minimum HS switch OFF-time (TOFF_MIN) or LS current limit is exceeded. Output voltage ripple is at
a minimum in this mode and the maximum output current of 1 A can be supplied by the LM43601-Q1
7.4.5 Light Load Operation
When the load current is lower than half of the peak-to-peak inductor current in CCM, the LM43601-Q1 operates
in DCM, also known as Diode Emulation Mode (DEM). In DCM operation, the LS FET is turned off when the
inductor current drops to 0 A to improve efficiency. Both switching losses and conduction losses are reduced in
DCM, comparing to forced PWM operation at light load.
At even lighter current loads, PFM is activated to maintain high efficiency operation. When the HS switch ON-
time reduces to TON_MIN or peak inductor current reduces to its minimum IPEAK-MIN, the switching frequency is
reduced to maintain proper regulation. Efficiency is greatly improved by reducing switching and gate drive losses.
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Device Functional Modes (continued)
7.4.6 Self-Bias Mode
For highest efficiency of operation, TI recommends that the BIAS pin be connected directly to VOUT when VOUT
≥
3.3 V. In this self-bias mode of operation, the difference between the input and output voltages of the internal
LDO are reduced and therefore the total efficiency is improved. These efficiency gains are more evident during
light load operation. During this mode of operation, the LM43601-Q1 operates with a minimum quiescent current
of 36 µA (typical). See VCC, UVLO and BIAS for more details.
8 Applications and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The LM43601-Q1 is a step down DC-to-DC regulator. It is typically used to convert a higher DC voltage to a
lower DC voltage with a maximum output current of 1 A. The following design procedure can be used to select
components for the LM43601-Q1. Alternately, the WEBENCH® software may be used to generate complete
designs. When generating a design, the WEBENCH® software utilizes iterative design procedure and accesses
comprehensive databases of components. See 使用 WEBENCH® 工具创建定制设计 for more details.
8.2 Typical Applications
The LM43601-Q1 only requires a few external components to convert from a wide range of supply voltage to
output voltage. Figure 44 shows a basic schematic when BIAS is connected to VOUT. This is recommended for
VOUT ≥ 3.3 V. For VOUT < 3.3 V, connect BIAS to ground, as shown in Figure 45.
L
L
VOUT
VOUT
VIN
VIN
VIN
SW
VIN
SW
COUT
COUT
CIN
LM43601-Q1
CIN
LM43601-Q1
CBOOT
CBOOT
CBOOT
BIAS
CBOOT
BIAS
ENABLE
PGOOD
ENABLE
PGOOD
CBIAS
CFF
CFF
RFBT
RFBT
SS/TRK
RT
SS/TRK
RT
FB
FB
VCC
VCC
SYNC
AGND
SYNC
AGND
RFBB
RFBB
CVCC
CVCC
PGND
PGND
Figure 44. LM43601-Q1 Basic Schematic for
OUT ≥ 3.3 V, Tie BIAS to VOUT
Figure 45. LM43601-Q1 Basic Schematic for
VOUT < 3.3 V, Tie BIAS to Ground
V
The LM43601-Q1 also integrates a full list of optional features to aid system design requirements, such as
precision enable, VCC UVLO, programmable soft start, output voltage tracking, programmable switching
frequency, clock synchronization and power-good indication. Each application can select the features for a more
comprehensive design. A schematic with all features utilized is shown in Figure 46.
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Typical Applications (continued)
L
VIN
VOUT
VIN
SW
COUT
CFF
CIN
LM43601-Q1
CBOOT
RENT
RFBT
CBOOT
FB
ENABLE
RENB
VCC
RFBB
SS/TRK
RT
CVCC
CSS
BIAS
RT
CBIAS
SYNC
AGND
PGOOD
PGND
RSYNC
Tie BIAS to PGND
when VOUT < 3.3 V
Figure 46. LM43601-Q1 Schematic with All Features
The external components must be chosen for the application, but also the stability criteria of the device control
loop. The LM43601-Q1 is optimized to work within a range of external components. The inductance and
capacitance of the LC output filter must considered in conjunction, creating a double pole, responsible for the
corner frequency of the converter. Table 2 can be used to simplify the output filter component selection.
Table 2. L, COUT, and CFF Typical Values
(1)
(2)(3)
(2)(3)
FS (kHz)
VOUT = 1 V
L (µH)
COUT (µF)
CFF (pF)
RT (kΩ)
RFBB (kΩ)
200
500
18
6.8
3.3
1.5
500
330
180
100
none
none
none
none
200
80.6 or open
39.2
100
100
100
100
1000
2200
17.8
VOUT = 3.3 V
200
47
18
10
4.7
220
100
47
44
33
18
12
200
80.6 or open
39.2
442
442
442
442
500
1000
2200
27
17.8
VOUT = 5 V
200
56
27
15
6.8
150
66
68
33
22
18
200
80.6 or open
39.2
255
255
255
255
500
1000
33
2200
22
17.8
VOUT = 12 V
200
(4)
100
47
33
22
15
see note
47
200
80.6 or open
39.2
90.9
90.9
90.9
500
1000
22
33
(1) All the COUT values are after derating. Add more when using ceramics
(2) RFBT = 0 Ω for VOUT = 1 V. RFBT = 1 MΩ for all other VOUT settings.
(3) For designs with RFBT other than 1 MΩ, adjust CFF such that (CFF × RFBT) is unchanged and adjust RFBB such that (RFBT / RFBB) is
unchanged.
(4) High ESR COUT gives enough phase boost, and CFF not needed.
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Typical Applications (continued)
8.2.1 Design Requirements
A detailed design procedure is described based on a design example. For this design example, use the
parameters listed in Table 3 as the input parameters.
Table 3. Design Example Parameters
DESIGN PARAMETER
VALUE
Input voltage VIN
12 V typical, range from 3.8 V to 36 V
Output voltage VOUT
Input ripple voltage
Output ripple voltage
Output current rating
Operating frequency
Soft-start time
3.3 V
400 mV
30 mV
1 A
500 kHz
10 ms
8.2.2 Detailed Design Procedure
8.2.2.1 Custom Design With WEBENCH® Tools
Click here to create a custom design using the LM43601-Q1 device with the WEBENCH® Power Designer.
1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements.
2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.
3. Compare the generated design with other possible solutions from Texas Instruments.
The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time
pricing and component availability.
In most cases, these actions are available:
•
•
•
•
Run electrical simulations to see important waveforms and circuit performance
Run thermal simulations to understand board thermal performance
Export customized schematic and layout into popular CAD formats
Print PDF reports for the design, and share the design with colleagues
Get more information about WEBENCH tools at www.ti.com/WEBENCH.
8.2.2.2 Output Voltage Set-Point
The output voltage of the LM43601-Q1 device is externally adjustable using a resistor divider network. The
divider network is comprised of top feedback resistor RFBT and bottom feedback resistor RFBB. Equation 11 is
used to determine the output voltage of the converter:
VFB
RFBB
=
RFBT
VOUT - VFB
(11)
Choose the value of the RFBT to be 1 MΩ to minimize quiescent current to improve light load efficiency in this
application. With the desired output voltage set to be 3.3 V and the VFB = 1.016 V, the RFBB value can then be
calculated using Equation 11. The formula yields a value of 444.83 kΩ. Choose the closest available value of 442
kΩ for the RFBB. See Adjustable Output Voltage for more details.
8.2.2.3 Switching Frequency
The default switching frequency of the LM43601-Q1 device is set at 500 kHz when RT pin is open circuit. The
switching frequency is selected to be 500 kHz in this application for one less passive components. If other
frequency is desired, use Equation 12 to calculate the required value for RT.
RT(kΩ) = 40200 / Freq (kHz) – 0.6
(12)
For 500 kHz, the calculated RT is 79.8 kΩ and standard value 80.6 kΩ can also be used to set the switching
frequency at 500 kHz.
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8.2.2.4 Input Capacitors
The LM43601-Q1 device requires high frequency input decoupling capacitor(s) and a bulk input capacitor,
depending on the application. The typical recommended value for the high frequency decoupling capacitor is 4.7
µF to 10 µF. A high-quality ceramic type X5R or X7R with sufficiency voltage rating is recommended. The
voltage rating must be greater than the maximum input voltage. To compensate the derating of ceramic
capacitors, a voltage rating of twice the maximum input voltage is recommended. Additionally, some bulk
capacitance can be required, especially if the LM43601-Q1 circuit is not located within approximately 5 cm from
the input voltage source. This capacitor is used to provide damping to the voltage spiking due to the lead
inductance of the cable or trace. The value for this capacitor is not critical but must be rated to handle the
maximum input voltage including ripple. For this design, a 10 µF, X7R dielectric capacitor rated for 100 V is used
for the input decoupling capacitor. The equivalent series resistance (ESR) is approximately 3 mΩ, and the
current rating is 3 A. Include a capacitor with a value of 0.1 µF for high-frequency filtering and place it as close
as possible to the device pins.
NOTE
DC Bias effect: High capacitance ceramic capacitors have a DC Bias effect, which will
have a strong influence on the final effective capacitance. Therefore the right capacitor
value has to be chosen carefully. Package size and voltage rating in combination with
dielectric material are responsible for differences between the rated capacitor value and
the effective capacitance.
8.2.2.5 Inductor Selection
The first criterion for selecting an output inductor is the inductance itself. In most buck converters, this value is
based on the desired peak-to-peak ripple current, ΔiL, that flows in the inductor along with the DC load current.
As with switching frequency, the selection of the inductor is a tradeoff between size and cost. Higher inductance
gives lower ripple current and hence lower output voltage ripple with the same output capacitors. Lower
inductance could result in smaller, less expensive component. An inductance that gives a ripple current of 20% to
40% of the 1 A at the typical supply voltage is a good starting point. ΔiL = (1/5 to 2/5) × IOUT. The peak-to-peak
inductor current ripple can be found by Equation 13 and the range of inductance can be found by Equation 14
with the typical input voltage used as VIN.
(VIN - VOUT )ìD
DiL =
L ìFS
(13)
(VIN - VOUT )ìD
0.4ìFS ìIL-MAX
(VIN - VOUT )ìD
0.2ìFS ìIL-MAX
Ç L Ç
(14)
D is the duty cycle of the converter which in a buck converter it can be approximated as D = VOUT / VIN
,
assuming no loss power conversion. By calculating in terms of amperes, volts, and megahertz, the inductance
value will come out in micro Henries. The inductor ripple current ratio is defined by:
DiL
IOUT
r =
(15)
The second criterion is the inductor saturation current rating. The inductor should be rated to handle the
maximum load current plus the ripple current:
IL-PEAK = ILOAD-MAX + Δ iL
(16)
The LM43601-Q1 has both valley current limit and peak current limit. During an instantaneous short, the peak
inductor current can be high due to a momentary increase in duty cycle. The inductor current rating should be
higher than the HS current limit. It is advised to select an inductor with a larger core saturation margin and
preferably a softer roll off of the inductance value over load current.
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In general, choosing lower inductance in switching power supplies is preferred because it usually corresponds to
faster transient response, smaller DCR, and reduced size for more compact designs. But too low of an
inductance can generate too large of an inductor current ripple such that overcurrent protection at the full load
could be falsely triggered. It also generates more conduction loss, because the RMS current is slightly higher
relative that with lower current ripple at the same DC current. Larger inductor current ripple also implies larger
output voltage ripple with the same output capacitors. With peak current mode control, it is not recommended to
have too small of an inductor current ripple. Enough inductor current ripple improves signal-to-noise ratio on the
current comparator and makes the control loop more immune to noise.
Once the inductance is determined, the type of inductor must be selected. Ferrite designs have very low core
losses and are preferred at high switching frequencies, so design goals can concentrate on copper loss and
preventing saturation. Ferrite core material saturates hard, which means that inductance collapses abruptly when
the peak design current is exceeded. The ‘hard’ saturation results in an abrupt increase in inductor ripple current
and consequent output voltage ripple. Do not allow the core to saturate!
For the design example, a standard 18-μH inductor from Würth, Coiltronics, or Vishay can be used for the 3.3-V
output with plenty of current rating margin.
8.2.2.6 Output Capacitor Selection
The device is designed to be used with a wide variety of LC filters. It is generally desired to use as little output
capacitance as possible to keep cost and size down. The output capacitor (s), COUT, should be chosen with
care since it directly affects the steady state output voltage ripple, loop stability and the voltage over/undershoot
during load current transients.
The output voltage ripple is essentially composed of two parts. One is caused by the inductor current ripple going
through the ESR of the output capacitors:
ΔVOUT-ESR = ΔiL× ESR
(17)
The other is caused by the inductor current ripple charging and discharging the output capacitors:
ΔVOUT-C = ΔiL/ ( 8 × FS × COUT
)
(18)
The two components in the voltage ripple are not in phase, so the actual peak-to-peak ripple is smaller than the
sum of the two peaks.
Output capacitance is usually limited by transient performance specifications if the system requires tight voltage
regulation in the presence of large current steps and fast slew rates. When a fast large load transient happens,
output capacitors provide the required charge before the inductor current can slew to the appropriate level. The
initial output voltage step is equal to the load current step multiplied by the ESR. VOUT continues to droop until
the control loop response increases or decreases the inductor current to supply the load. To maintain a small
over- or under-shoot during a transient, small ESR and large capacitance are desired. But these also come with
higher cost and size. Thus, the motivation is to seek a fast control loop response to reduce the output voltage
deviation.
For a given input and output requirement, the following inequality gives an approximation for an absolute
minimum output cap required:
2
»
…
…
ÿ
Ÿ
≈
’
1
r
Å
Å
COUT
>
ì
ì(1+ D ) + D ì(1+ r)
∆
∆
÷
(
)
÷
(FS ìr ì DVOUT / IOUT
)
12
Ÿ
⁄
«
◊
(19)
Along with this for the same requirement, calculate the maximum ESR with Equation 20:
Å
D
1
ESR <
ì( + 0.5)
FS ìCOUT
r
where
•
•
•
•
•
r = Ripple ratio of the inductor ripple current (ΔIL / IOUT
ΔVOUT = Target output voltage undershoot
D’ = 1 – Duty cycle
)
FS = Switching Frequency
IOUT = Load Current
(20)
28
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ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
A general guideline for COUT range is that COUT should be larger than the minimum required output capacitance
calculated by Equation 19, and smaller than 10 times the minimum required output capacitance or 1 mF. In
applications with VOUT less than 3.3 V, it is critical that low ESR output capacitors are selected. This will limit
potential output voltage overshoots as the input voltage falls below the device normal operating range. To
optimize the transient behavior a feed-forward capacitor could be added in parallel with the upper feedback
resistor. For this design example, two 47-µF, 10-V, X7R ceramic capacitors are used in parallel.
8.2.2.7 Feed-Forward Capacitor
The LM43601-Q1 is internally compensated and the internal R-C values are 400 kΩ and 50 pF, respectively.
Depending on the VOUT and frequency FS, if the output capacitor COUT is dominated by low ESR (ceramic types)
capacitors, it could result in low phase margin. To improve the phase boost an external feedforward capacitor
CFF can be added in parallel with RFBT. CFF is chosen such that phase margin is boosted at the crossover
frequency without CFF. A simple estimation for the crossover frequency without CFF (fx) is shown in Equation 21,
assuming COUT has very small ESR.
2.73
fx =
VOUT ìCOUT
(21)
Equation 22 was tested for CFF:
1
1
CFF
=
ì
2pfx
RFBT ì(RFBT / /RFBB
)
(22)
Equation 22 indicates that the crossover frequency is geometrically centered on the zero and pole frequencies
caused by the CFF capacitor.
For designs with higher ESR, CFF is not needed when COUT has very high ESR and CFF calculated from
Equation 22 should be reduced with medium ESR. Table 2 can be used as a quick starting point.
For the application in this design example, a 33-pF COG capacitor is selected.
8.2.2.8 Bootstrap Capacitors
Every LM43601-Q1 design requires a bootstrap capacitor, CCBOOT. The recommended bootstrap capacitor is
0.47 μF and rated at 6.3 V or higher. The bootstrap capacitor is located between the SW pin and the CBOOT
pin. The bootstrap capacitor must be a high-quality ceramic type with X7R or X5R grade dielectric for
temperature stability.
8.2.2.9 VCC Capacitor
The VCC pin is the output of an internal LDO for LM43601-Q1. The input for this LDO comes from either VIN or
BIAS (see Functional Block Diagram for LM43601-Q1). To insure stability of the part, place a minimum of 2.2-µF,
10-V capacitor from this pin to ground.
8.2.2.10 BIAS Capacitors
For an output voltage of 3.3 V and greater, the BIAS pin can be connected to the output in order to increase light
load efficiency. This pin is an input for the VCC LDO. When BIAS is not connected, the input for the VCC LDO is
internally connected into VIN. Since this is an LDO, the voltage differences between the input and output affects
the efficiency of the LDO. If necessary, a capacitor with a value of 1 μF can be added close to the BIAS pin as
an input capacitor for the LDO.
8.2.2.11 Soft-Start Capacitors
The user can leave the SS/TRK pin floating and the LM43601-Q1 implements a soft-start time of 4.1 ms typically.
In order to use an external soft-start capacitor, the capacitor must be sized such that the soft-start time is longer
than 4.1 ms. Use Equation 23 to calculate the soft-start capacitor value:
CSS = ISSC ì tSS
where
•
•
CSS = soft-start capacitor value (µF)
ISS = soft-start charging current (µA)
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LM43601-Q1
ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
www.ti.com.cn
•
tSS = desired soft start time (s)
(23)
For the desired soft-start time of 10 ms and soft start charging current of 2.2 µA, Equation 23 yields a soft-start
capacitor value of 0.022 µF.
8.2.2.12 Undervoltage Lockout Setpoint
The undervoltage lockout (UVLO) is adjusted using the external voltage divider network of RENT and RENB. RENT
is connected between VIN and the EN pins of the LM43601-Q1 device. RENB is connected between the EN pin
and the GND pin. The UVLO has two thresholds, one for power up when the input voltage is rising and one for
powerdown or brownouts when the input voltage is falling. Equation 24 can be used to determine the VIN
(UVLO) level.
VIN-UVLO-RISING = VENH × (RENB + RENT) / RENB
(24)
The EN rising threshold for LM43601-Q1 is set to be 2.1 V. Choose the value of RENB to be 1 MΩ to minimize
input current going into the converter. If the desired VIN (UVLO) level is at 5 V, then the value of RENT can be
calculated using Equation 25:
RENT = (VIN-UVLO-RISING / VENH × 1) × RENB
(25)
The above equation yields a value of 1.37 MΩ. The resulting falling UVLO threshold can be calculated as
follows:
VIN-UVLO-FALLING = 1.8 × (RENB + RENT) / RENB
(26)
8.2.2.13 PGOOD
A typical pullup resistor value is 10 kΩ to 100 kΩ from the PGOOD pin to a voltage no higher than 12 V. If it is
desired to pull up the PGOOD pin to a voltage higher than 12 V, a resistor can be added from the PGOOD pin to
ground to divide the voltage seen by the PGOOD pin to a value no higher than 12 V.
30
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LM43601-Q1
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ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
8.2.3 Application Performance Curves
See Table 2 for bill of materials for each VOUT and FS combination. Unless otherwise stated, application performance curves
were taken at TA = 25°C.
90
80
VOUT = 1 V FS = 500 kHz
70
60
50
40
30
20
10
0
L=6.8 µH
VOUT
SW
LM43601-Q1
RT
COUT
330 µF
CBOOT
0.47 µF
CBOOT
BIAS
VIN = 3.5V
VIN = 5V
VIN = 8V
VIN = 12V
CBIAS
1 µF
RFBT
1 MΩ
VCC
CVCC
2.2 µF
FB
0.001
0.010
0.100
1.000
Load Current (A)
C001
VOUT = 1 V
FS = 500 kHz
VIN = 12 V
VOUT = 1 V
FS = 500 kHz
Figure 47. BOM for VOUT = 1 V FS = 500 kHz
Figure 48. Efficiency
1.04
1.03
1.02
1.01
1.00
0.99
0.98
VDROP_ON_0.1Ω_LOAD
(100 mV/DIV)
VOUT (50 mV/DIV)
VIN = 3.5V
IL (1 A/DIV)
VIN = 5V
VIN = 8V
VIN = 12V
Time (200 µs/DIV)
0.001
0.010
0.100
1.000
Load Current (A)
C011
VOUT = 1 V
FS = 500 kHz
VOUT = 1 V
FS = 500 kHz
VIN = 12 V
Figure 49. Output Voltage Regulation
Figure 50. Load Transient Between 0.05 A and 1 A
1.2
VDROP_ON_0.1Ω_LOAD
1
0.8
0.6
0.4
0.2
0
(100 mV/DIV)
VOUT (50 mV/DIV)
R,JA = 10 °C/W
R,JA = 20 °C/W
R,JA = 30 °C/W
IL (1 A/DIV)
Time (200 µs/DIV)
50
60
70
80
90
100
110
120
Temperature (°C)
C001
VOUT = 1 V
FS = 500 kHz
VIN = 12 V
VOUT = 1 V
FS = 500 kHz
VIN = 12 V
Figure 51. Load Transient Between 0.1 A and 1 A
Figure 52. Derating Curve
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31
LM43601-Q1
ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
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See Table 2 for bill of materials for each VOUT and FS combination. Unless otherwise stated, application performance curves
were taken at TA = 25°C.
100
90
VOUT = 3.3 V FS = 500 kHz
80
70
L=18 µH
VOUT
SW
60
50
40
30
20
10
0
LM43601-Q1
RT
COUT
100 µF
CBOOT
0.47 µF
CBOOT
BIAS
VIN = 8V
VIN = 12V
VIN = 18V
VIN = 24V
VIN = 28V
VIN = 36V
CBIAS
1 µF
RFBT
1 MΩ
CFF
VCC
CVCC
2.2 µF
33 pF
FB
RFBB
442
kΩ
0.001
0.010
0.100
1.000
Load Current (A)
C002
VOUT = 3.3 V
FS = 500 kHz
VIN = 12 V
VOUT = 3.3 V
FS = 500 kHz
Figure 53. BOM for VOUT = 3.3 V FS = 500 kHz
Figure 54. Efficiency
3.40
3.38
3.36
3.34
3.32
3.30
3.28
3.26
3.24
3.22
3.20
3.50
3.40
3.30
3.20
3.10
3.00
2.90
2.80
2.70
2.60
2.50
Load = 0.25A
Load = 0.5A
Load = 0.75A
Load = 1A
VIN = 8V
VIN = 18V
VIN = 28V
VIN = 12V
VIN = 24V
VIN = 36V
0.001
0.010
0.100
1.000
3.5
4.0
4.5
5.0
Load Current (A)
VIN (V)
C012
C022
VOUT = 3.3 V
FS = 500 kHz
VOUT = 3.3 V
FS = 500 kHz
Figure 55. Output Voltage Regulation
Figure 56. Dropout Curve
1.2
1
VDROP_ON_0.75Ω_LOAD
(750 mV/DIV)
0.8
0.6
0.4
VOUT (200 mV/DIV)
R,JA = 10 °C/W
R,JA = 20 °C/W
R,JA = 30 °C/W
IL (1 A/DIV)
0.2
0
Time (200 µs/DIV)
50
60
70
80
90
100
110
120
Temperature (°C)
C001
VOUT = 3.3 V
FS = 500 kHz
VIN = 12 V
VOUT = 3.3 V
FS = 500 kHz
VIN = 12 V
Figure 57. Load Transient Between 0.1 A and 1 A
Figure 58. Derating Curve
32
Copyright © 2015–2017, Texas Instruments Incorporated
LM43601-Q1
www.ti.com.cn
ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
See Table 2 for bill of materials for each VOUT and FS combination. Unless otherwise stated, application performance curves
were taken at TA = 25°C.
100
90
VOUT = 5 V FS = 500 kHz
80
70
L=27 µH
VOUT
60
50
40
30
20
10
0
SW
LM43601-Q1
RT
COUT
66 µF
CBOOT
0.47 µF
CBOOT
BIAS
VIN = 12V
VIN = 18V
VIN = 24V
VIN = 28V
VIN = 36V
CBIAS
1 µF
RFBT
1 MΩ
CFF
VCC
CVCC
2.2 µF
33 pF
FB
RFBB
255
kΩ
0.001
0.010
0.100
1.000
Load Current (A)
C003
VOUT = 5 V
FS = 500 kHz
VIN = 12 V
VOUT = 5 V
FS = 500 kHz
Figure 59. BOM for VOUT = 5 V FS = 500 kHz
Figure 60. Efficiency
5.20
5.15
5.10
5.05
5.00
4.95
4.90
4.85
4.80
5.2
5.0
4.8
4.6
4.4
4.2
4.0
VIN = 12V
Load = 0.25A
Load = 0.5A
Load = 0.75A
Load = 1A
VIN = 18V
VIN = 24V
VIN = 28V
VIN = 36V
0.001
0.010
0.100
1.000
5.0
5.5
6.0
6.5
Load Current (A)
VIN (V)
C013
C023
VOUT = 5 V
FS = 500 kHz
VOUT = 5 V
FS = 500 kHz
Figure 61. Output Voltage Regulation
Figure 62. Dropout Curve
1.2
VDROP_ON_0.75Ω_LOAD
1
0.8
0.6
0.4
0.2
0
(750 mV/DIV)
VOUT (200 mV/DIV)
R,JA = 10 °C/W
R,JA = 20 °C/W
R,JA = 30 °C/W
IL (1 A/DIV)
Time (200 µs/DIV)
50
60
70
80
90
100
110
120
Temperature (°C)
C001
VOUT = 5 V
FS = 500 kHz
VIN = 12 V
VOUT = 5 V
FS = 500 kHz
VIN = 12 V
Figure 63. Load Transient Between 0.1 A and 1 A
Figure 64. Derating Curve
Copyright © 2015–2017, Texas Instruments Incorporated
33
LM43601-Q1
ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
www.ti.com.cn
See Table 2 for bill of materials for each VOUT and FS combination. Unless otherwise stated, application performance curves
were taken at TA = 25°C.
100
90
VOUT = 5 V FS = 200 kHz
80
70
L=56 µH
VOUT
60
SW
LM43601-Q1
RT
COUT
RT
200
kΩ
CBOOT
50
40
30
20
10
0
CBOOT
BIAS
0.47 µF
150 µF
VIN = 8V
VIN = 12V
VIN = 18V
VIN = 24V
VIN = 28V
VIN = 36V
CBIAS
1 µF
RFBT
1 MΩ
CFF
VCC
CVCC
2.2 µF
68 pF
FB
RFBB
255
kΩ
0.001
0.010
0.100
1.000
Load Current (A)
C004
VOUT = 5 V
FS = 200 kHz
VIN = 12 V
VOUT = 5 V
FS = 200 kHz
Figure 65. BOM for VOUT = 5 V FS = 200 kHz
Figure 66. Efficiency
5.15
5.10
5.05
5.00
4.95
4.90
4.85
4.80
5.2
5.0
4.8
4.6
4.4
4.2
4.0
Load = 0.25A
Load = 0.5A
Load = 0.75A
Load = 1A
VIN = 8V
VIN = 12V
VIN = 28V
VIN = 18V
VIN = 36V
VIN = 24V
0.001
0.010
0.100
1.000
5.0
5.5
6.0
6.5
Load Current (A)
VIN (V)
C014
C024
VOUT = 5 V
FS = 200 kHz
VOUT = 5 V
FS = 200 kHz
Figure 67. Output Voltage Regulation
Figure 68. Dropout Curve
1.2
VDROP_ON_0.75Ω_LOAD
(750 mV/DIV)
1
0.8
0.6
0.4
0.2
0
VOUT (200 mV/DIV)
R,JA = 10 °C/W
R,JA = 20 °C/W
R,JA = 30 °C/W
IL (1 A/DIV)
Time (200 µs/DIV)
50
60
70
80
90
100
110
120
Temperature (°C)
C001
VOUT = 5 V
FS = 200 kHz
VIN = 12 V
VOUT = 5 V
FS = 200 kHz
Figure 69. Load Transient Between 0.1 A and 1 A
Figure 70. Derating Curve
34
Copyright © 2015–2017, Texas Instruments Incorporated
LM43601-Q1
www.ti.com.cn
ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
See Table 2 for bill of materials for each VOUT and FS combination. Unless otherwise stated, application performance curves
were taken at TA = 25°C.
100
90
VOUT = 5 V FS = 1 MHz
80
70
L=15 µH
VOUT
60
LM43601-Q1 SW
RT
COUT
RT
39.2
kΩ
CBOOT
50
40
30
20
10
0
CBOOT
BIAS
0.47 µF
33 µF
CBIAS
1 µF
RFBT
1 MΩ
VIN = 12V
VIN = 18V
VIN = 24V
VIN = 28V
CFF
VCC
CVCC
2.2 µF
22 pF
FB
RFBB
255
kΩ
0.001
0.010
0.100
1.000
Load Current (A)
C005
VOUT = 5 V
FS = 1 MHz
VIN = 12 V
VOUT = 5 V
FS = 1 MHz
VIN = 12 V
Figure 71. BOM for VOUT = 5 V FS = 1 MHz
Figure 72. Efficiency
5.20
5.15
5.10
5.05
5.00
4.95
4.90
4.85
4.80
5.2
5.0
4.8
4.6
4.4
4.2
4.0
Load = 0.25A
Load = 0.5A
Load = 0.75A
Load = 1A
VIN = 12V
VIN = 18V
VIN = 24V
VIN = 28V
0.001
0.010
0.100
1.000
5.0
5.5
6.0
6.5
Load Current (A)
VIN (V)
C015
C025
VOUT = 5 V
FS = 1 MHz
VOUT = 5 V
FS = 1 MHz
Figure 73. Output Voltage Regulation
Figure 74. Dropout Curve
1.2
VDROP_ON_0.75Ω_LOAD
(750 mV/DIV)
1
0.8
0.6
0.4
0.2
0
VOUT (200 mV/DIV)
R,JA = 10 °C/W
R,JA = 20 °C/W
R,JA = 30 °C/W
IL (1 A/DIV)
Time (200 µs/DIV)
50
60
70
80
90
100
110
120
Temperature (°C)
C001
VOUT = 5 V
FS = 1 MHz
VIN = 12 V
VOUT = 5 V
FS = 1 MHz
VIN = 12 V
Figure 75. Load Transient Between 0.1 A and 1 A
Figure 76. Derating Curve
Copyright © 2015–2017, Texas Instruments Incorporated
35
LM43601-Q1
ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
www.ti.com.cn
See Table 2 for bill of materials for each VOUT and FS combination. Unless otherwise stated, application performance curves
were taken at TA = 25°C.
90
80
VOUT = 5 V FS = 2.2 MHz
70
60
L=6.8 µH
VOUT
LM43601-Q1 SW
50
40
30
20
10
0
RT
COUT
22 µF
RT
17.8
kΩ
CBOOT
0.47 µF
CBOOT
BIAS
CBIAS
1 µF
RFBT
1 MΩ
CFF
VCC
CVCC
2.2 µF
18 pF
FB
RFBB
255
kΩ
VIN = 12V
1.000
0.001
0.010
0.100
Load Current (A)
C006
VOUT = 5 V
FS = 1 MHz
VIN = 12 V
VOUT = 5 V
FS = 2.2 MHz
VIN = 12 V
Figure 77. BOM for VOUT = 5 V FS = 2.2 MHz
Figure 78. Efficiency
5.20
5.15
5.10
5.05
5.00
4.95
4.90
4.85
4.80
5.2
5.0
4.8
4.6
4.4
4.2
4.0
Load = 0.25A
Load = 0.5A
Load = 0.75A
Load = 1A
VIN = 12V
1.000
0.001
0.010
0.100
5.0
5.5
6.0
6.5
Load Current (A)
VIN (V)
C016
C026
VOUT = 5 V
FS = 2.2 MHz
VOUT = 5 V
FS = 2.2 MHz
Figure 79. Output Voltage Regulation
Figure 80. Dropout Curve
1.2
VDROP_ON_0.75Ω_LOAD
1
0.8
0.6
0.4
0.2
0
(750 mV/DIV)
VOUT (200 mV/DIV)
R,JA = 10 °C/W
R,JA = 20 °C/W
R,JA = 30 °C/W
IL (1 A/DIV)
Time (200 µs/DIV)
50
60
70
80
90
100
110
120
Temperature (°C)
C001
VOUT = 5 V
FS = 2.2 MHz
VIN = 12 V
VOUT = 5 V
FS = 2.2 MHz
VIN = 12 V
Figure 81. Load Transient Between 0.1 A and 1 A
Figure 82. Derating Curve
36
Copyright © 2015–2017, Texas Instruments Incorporated
LM43601-Q1
www.ti.com.cn
ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
See Table 2 for bill of materials for each VOUT and FS combination. Unless otherwise stated, application performance curves
were taken at TA = 25°C.
100
90
VOUT = 12 V FS = 500 kHz
80
70
L=47 µH
VOUT
SW
60
50
40
30
20
10
0
LM43601-Q1
RT
COUT
CBOOT
CBOOT
BIAS
0.47 µF
22 µF
CBIAS
1 µF
RFBT
1 MΩ
CFF
VCC
VIN = 24V
VIN = 28V
VIN = 36V
CVCC
47 pF
FB
2.2 µF
RFBB
90.9
kΩ
0.001
0.010
0.100
1.000
Load Current (A)
C007
VOUT = 12 V
FS = 500 kHz
VIN = 24 V
VOUT = 12 V
FS = 500 kHz
Figure 83. BOM for VOUT = 12 V FS = 500 kHz
Figure 84. Efficiency
12.5
12.4
12.3
12.2
12.1
12.0
11.9
11.8
11.7
11.6
11.5
12.4
12.2
12.0
11.8
11.6
11.4
11.2
11.0
Load = 0.25A
Load = 0.5A
Load = 0.75A
Load = 1A
VIN = 24V
VIN = 28V
VIN = 36V
0.001
0.010
0.100
1.000
12.0
12.5
13.0
VIN (V)
13.5
14.0
Load Current (A)
C017
C027
VOUT = 12 V
FS = 500 kHz
VOUT = 12 V
FS = 500 kHz
Figure 85. Output Voltage Regulation
Figure 86. Dropout Curve
1.2
1
0.8
0.6
0.4
0.2
0
ILOAD (1 A/DIV)
VOUT (500 mV/DIV)
IL (1 A/DIV)
R,JA = 10 °C/W
R,JA = 20 °C/W
R,JA = 30 °C/W
Time (200 µs/DIV)
50
60
70
80
90
100
110
120
Temperature (°C)
C001
VOUT = 12 V
FS = 500 kHz
VIN = 24 V
VOUT = 12 V
FS = 500 kHz
VIN = 24 V
Figure 87. Load Transient Between 0.1 A and 1 A
Figure 88. Derating Curve
Copyright © 2015–2017, Texas Instruments Incorporated
37
LM43601-Q1
ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
www.ti.com.cn
See Table 2 for bill of materials for each VOUT and FS combination. Unless otherwise stated, application performance curves
were taken at TA = 25°C.
1.2
1.2
1
1
0.8
0.6
0.4
0.2
0
0.8
0.6
0.4
0.2
0
Vin = 12V
Vin = 24V
Vin = 12V
Vin = 24V
50
60
70
80
90
100
110
120
50
60
70
80
90
100
110
120
Temperature (°C)
Temperature (°C)
C001
C001
VOUT = 3.3 V
FS = 500 kHz
RθJA = 20°C/W
VOUT = 5 V
FS = 500 kHz
RθJA = 20°C/W
Figure 89. Derating Curve with RθJA = 20°C/W
Figure 90. Derating Curve with RθJA = 20°C/W
1.2
1
1.2
1
0.8
0.6
0.4
0.2
0
0.8
0.6
0.4
0.2
0
Vin = 12V
Vin = 24V
Vin = 12V
Vin = 24V
50
60
70
80
90
100
110
120
50
60
70
80
90
100
110
120
Temperature (°C)
Temperature (°C)
C001
C001
VOUT = 5 V
FS = 200 kHz
RθJA = 20°C/W
VOUT = 5 V
FS = 1 MHz
RθJA = 20 °C/W
Figure 91. Derating Curve with RθJA = 20 °C/W
Figure 92. Derating Curve with RθJA = 20 °C/W
1000000
100000
10000
1000
1000000
100000
10000
VIN = 8 V
VIN = 12 V
VIN = 18 V
VIN = 24 V
VIN = 36 V
VIN = 12 V
VIN = 18 V
VIN = 24 V
VIN = 36 V
0.001
0.01
0.1
1
0.001
0.01
0.1
1
Load (A)
Load (A)
C001
C001
VOUT = 3.3 V
FS = 500 kHz
VOUT = 5 V
FS = 1 MHz
Figure 93. Switching Frequency vs IOUT in PFM Operation
Figure 94. Switching Frequency vs IOUT in PFM Operation
38
Copyright © 2015–2017, Texas Instruments Incorporated
LM43601-Q1
www.ti.com.cn
ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
See Table 2 for bill of materials for each VOUT and FS combination. Unless otherwise stated, application performance curves
were taken at TA = 25°C.
SW (10 V/DIV)
SW (10 V/DIV)
VOUT (5 mV/DIV)
VOUT (5 mV/DIV)
IL (1 A/DIV)
IL (1 A/DIV)
Time (2 µs/DIV)
Time (2 µs/DIV)
VOUT = 3.3 V
FS = 500 kHz
IOUT = 1A
VOUT = 3.3 V
FS = 500 kHz
IOUT = 40 mA
Figure 95. Switching Waveform in CCM Operation
Figure 96. Switching Waveform in DCM Operation
SW (10 V/DIV)
PGOOD (2 V/DIV)
VOUT (5 mV/DIV)
VOUT (2 V/DIV)
IL (1 A/DIV)
IL (1 A/DIV)
Time (2 µs/DIV)
Time (2 ms/DIV)
VOUT = 3.3 V
FS = 500 kHz
IOUT = 10 mA
VIN = 12V
VOUT = 3.3 V
RLOAD = 3.3 Ω
Figure 97. Switching Waveform in PFM Operation
Figure 98. Start-up Into Full Load with Internal Soft-Start
Rate
PGOOD (2 V/DIV)
PGOOD (2 V/DIV)
VOUT (2 V/DIV)
IL (500 mA/DIV)
VOUT (2 V/DIV)
IL (200 mA/DIV)
Time (2 ms/DIV)
Time (2 ms/DIV)
VIN = 12 V
VOUT = 3.3 V
RLOAD = 6.6 Ω
VIN = 12 V
VOUT = 3.3 V
RLOAD = 33 Ω
Figure 99. Start-up Into Half Load with Internal Soft-Start
Rate
Figure 100. Start-up Into 100 mA with Internal Soft-Start
Rate
Copyright © 2015–2017, Texas Instruments Incorporated
39
LM43601-Q1
ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
www.ti.com.cn
See Table 2 for bill of materials for each VOUT and FS combination. Unless otherwise stated, application performance curves
were taken at TA = 25°C.
PGOOD (2 V/DIV)
PGOOD (10 V/DIV)
VOUT (1 V/DIV)
VOUT (10 V/DIV)
IL (1 A/DIV)
IL (200 mA/DIV)
Time (2 ms/DIV)
Time (5 ms/DIV)
VIN = 12V
VOUT = 3.3 V
RLOAD = Open
VIN = 24 V
VOUT = 12 V
RLOAD = 12 Ω
Figure 101. Start-up Into 1.0 V Pre-biased Voltage
Figure 102. Start-up with External Capacitor CSS = 33 nF
VIN (10 V/DIV)
VOUT (50 mV/DIV)
IL (1 A/DIV)
VIN (10 V/DIV)
VOUT (50 mV/DIV)
IL (500 mA/DIV)
Time (2 ms/DIV)
Time (2 ms/DIV)
VOUT = 3.3 V
FS = 500 kHz
IOUT = 1 A
VOUT = 3.3 V
FS = 500 kHz
IOUT = 0.5 A
Figure 103. Line Transient: VIN Transitions Between 12 V
and 36 V
Figure 104. Line Transient: VIN Transitions Between 12 V
and 36 V
PGOOD (5 V/DIV)
VOUT (5 V/DIV)
IL (1 A/DIV)
Time (10 ms/DIV)
VOUT = 3.3 V
FS = 500 kHz
VIN = 12 V
Figure 105. Short-Circuit Protection and Recover
40
Copyright © 2015–2017, Texas Instruments Incorporated
LM43601-Q1
www.ti.com.cn
ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
9 Power Supply Recommendations
The LM43601-Q1 is designed to operate from an input voltage supply range between 3.5 V and 36 V. This input
supply must be able to withstand the maximum input current and maintain a voltage above 3.5 V. The resistance
of the input supply rail should be low enough that an input current transient does not cause a high enough drop
at the LM43601-Q1 supply voltage that can cause a false UVLO fault triggering and system reset.
If the input supply is located more than a few inches from the LM43601-Q1 additional bulk capacitance may be
required in addition to the ceramic bypass capacitors. The amount of bulk capacitance is not critical, but a 47-µF
or 100-µF electrolytic capacitor is a typical choice.
10 Layout
The performance of any switching converter depends as much upon the layout of the PCB as the component
selection. The following guidelines will help users design a PCB with the best power conversion performance,
thermal performance, and minimized generation of unwanted EMI.
10.1 Layout Guidelines
1. Place ceramic high frequency bypass CIN as close as possible to the LM43601-Q1 VIN and PGND pins.
Grounding for both the input and output capacitors should consist of localized top side planes that connect to
the PGND pins and PAD.
2. Place bypass capacitors for VCC and BIAS close to the pins and ground the bypass capacitors to device
ground.
3. Minimize trace length to the FB pin. Both feedback resistors, RFBT and RFBB must be located close to the FB
pin. Place CFF directly in parallel with RFBT. If VOUT accuracy at the load is important, make sure VOUT sense
is made at the load. Route VOUT sense path away from noisy nodes and preferably through a layer on the
other side of a shielding layer.
4. Use ground plane in one of the middle layers as noise shielding and heat dissipation path.
5. Have a single point ground connection to the plane. The ground connections for the feedback, soft-start, and
enable components should be routed to the ground plane. This prevents any switched or load currents from
flowing in the analog ground traces. If not properly handled, poor grounding can result in degraded load
regulation or erratic output voltage ripple behavior.
6. Make VIN, VOUT and ground bus connections as wide as possible. This reduces any voltage drops on the
input or output paths of the converter and maximizes efficiency.
7. Provide adequate device heat-sinking. Use an array of heat-sinking vias to connect the exposed pad to the
ground plane on the bottom PCB layer. If the PCB has multiple copper layers, these thermal vias can also be
connected to inner layer heat-spreading ground planes. Ensure enough copper area is used for heat-sinking
to keep the junction temperature below 125°C.
10.1.1 Compact Layout for EMI Reduction
Radiated EMI is generated by the high di/dt components in pulsing currents in switching converters. The larger
area covered by the path of a pulsing current, the more electromagnetic emission is generated. The key to
minimize radiated EMI is to identify the pulsing current path and minimize the area of the path. In Buck
converters, the pulsing current path is from the VIN side of the input capacitors to HS switch, to the LS switch,
and then return to the ground of the input capacitors, as shown in Figure 106.
BUCK
CONVERTER
L
VIN
SW
VOUT
COUT
VIN
CIN
PGND
PGND
High di/dt
current
Figure 106. Buck Converter High di / dt Path
Copyright © 2015–2017, Texas Instruments Incorporated
41
LM43601-Q1
ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
www.ti.com.cn
Layout Guidelines (continued)
High frequency ceramic bypass capacitors at the input side provide primary path for the high di/dt components of
the pulsing current. Placing ceramic bypass capacitor(s) as close as possible to the VIN and PGND pins is the
key to EMI reduction.
The SW pin connecting to the inductor should be as short as possible, and just wide enough to carry the load
current without excessive heating. Short, thick traces or copper pours (shapes) should be used for high current
conduction path to minimize parasitic resistance. The output capacitors should be place close to the VOUT end of
the inductor and closely grounded to PGND pin and exposed PAD.
The bypass capacitors on VCC and BIAS pins should be placed as close as possible to the pins respectively and
closely grounded to PGND and the exposed PAD.
10.1.2 Ground Plane and Thermal Considerations
It is recommended to use one of the middle layers as a solid ground plane. Ground plane provides shielding for
sensitive circuits and traces. It also provides a quiet reference potential for the control circuitry. The AGND and
PGND pins should be connected to the ground plane using vias right next to the bypass capacitors. PGND pins
are connected to the source of the internal LS switch. They should be connected directly to the grounds of the
input and output capacitors. The PGND net contains noise at the switching frequency and may bounce due to
load variations. The PGND trace, as well as PVIN and SW traces, should be constrained to one side of the
ground plane. The other side of the ground plane contains much less noise and should be used for sensitive
routes.
It is recommended to provide adequate device heat sinking by utilizing the PAD of the IC as the primary thermal
path. Use a minimum 4 by 4 array of 10 mil thermal vias to connect the PAD to the system ground plane for heat
sinking. The vias should be evenly distributed under the PAD. Use as much copper as possible for system
ground plane on the top and bottom layers for the best heat dissipation. It is recommended to use a four-layer
board with the copper thickness, for the four layers, starting from the top one, 2 oz / 1 oz / 1 oz / 2 oz. Four layer
boards with enough copper thickness and proper layout provides low current conduction impedance, proper
shielding and lower thermal resistance.
The thermal characteristics of the LM43601-Q1 are specified using the parameter RθJA, which characterize the
junction temperature of the silicon to the ambient temperature in a specific system. Although the value of RθJA is
dependant on many variables, it still can be used to approximate the operating junction temperature of the
device. To obtain an estimate of the device junction temperature, one may use the following relationship:
TJ = PD × RθJA + TA
where
•
•
•
•
•
TJ = junction temperature in °C
PD = VIN × IIN × (1 − Efficiency) − 1.1 × IOUT × DCR
DCR = inductor DC parasitic resistance in Ω
RθJA = junction-to-ambient thermal resistance of the device in °C/W
TA = ambient temperature in °C.
(27)
The maximum operating junction temperature of the LM43601-Q1 is 125°C. RθJA is highly related to PCB size
and layout, as well as environmental factors such as heat sinking and air flow. Figure 107 shows measured
results of RθJA with different copper area on a 2-layer board and a 4-layer board.
42
Copyright © 2015–2017, Texas Instruments Incorporated
LM43601-Q1
www.ti.com.cn
ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
Layout Guidelines (continued)
50.0
45.0
40.0
35.0
30.0
25.0
20.0
1W @ 0fpm - 2 layer
2W @ 0fpm - 2 layer
1W @ 0fpm - 4 layer
2W @ 0fpm - 4 layer
20mm x 20mm 30mm x 30mm 40mm x 40mm 50mm x 50mm
Copper Area
C030
Figure 107. Measured RθJA vs PCB Copper Area on a 2-layer Board and a 4-layer Board
10.1.3 Feedback Resistors
To reduce noise sensitivity of the output voltage feedback path, it is important to place the resistor divider and
CFF close to the FB pin, rather than close to the load. The FB pin is the input to the error amplifier, so it is a high
impedance node and very sensitive to noise. Placing the resistor divider and CFF closer to the FB pin reduces the
trace length of FB signal and reduces noise coupling. The output node is a low impedance node, so the trace
from VOUT to the resistor divider can be long if short path is not available.
If voltage accuracy at the load is important, make sure voltage sense is made at the load. Doing so will correct
for voltage drops along the traces and provide the best output accuracy. The voltage sense trace from the load to
the feedback resistor divider should be routed away from the SW node path, the inductor and VIN path to avoid
contaminating the feedback signal with switch noise, while also minimizing the trace length. This is most
important when high value resistors are used to set the output voltage. It is recommended to route the voltage
sense trace on a different layer than the inductor, SW node and VIN path, such that there is a ground plane in
between the feedback trace and inductor / SW node / VIN polygon. This provides further shielding for the voltage
feedback path from switching noises.
版权 © 2015–2017, Texas Instruments Incorporated
43
LM43601-Q1
ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
www.ti.com.cn
10.2 Layout Example
VOUT distribution point
is away from inductor
and past COUT
VOUT sense point is away
from inductor and past COUT
GND
VOUT
GND
COUT
As much copper area as possible,
for better thermal performance
L
Place ceramic bypass caps
close to VIN and PGND pins
1
16
SW
PGND
PGND
Place CBOOT
close to pins
CIN
PAD (17)
SW
15
GND
VIN
2
3
4
5
6
CBOOT
+
Route VOUT
sense trace
away from
SW and VIN
nodes.
VIN
VIN
EN
14
13
CBOOT
VCC
CVCC
Place bypass
caps close to
pins
BIAS
12
11
10
CBIAS
Place RFBB
close to FB
and AGND
Preferably
shielded in an
alternative
layer
SS/TRK
AGND
FB
SYNC
RT
Ground
bypass caps
to DAP
7
8
RFBB
Trace to
FB short
and thin
PGOOD
9
RFBT
CFF
VOUT
sense
GND
As much copper area as possible, for better thermal performance
Preferably use GND Plane as a middle layer for shielding and heat dissipation
Preferably place and route on top layer and use solid copper on bottom layer for heat dissipation
Figure 108. LM43601-Q1 PCB Layout Example
44
版权 © 2015–2017, Texas Instruments Incorporated
LM43601-Q1
www.ti.com.cn
ZHCSE17B –JULY 2015–REVISED NOVEMBER 2017
11 器件和文档支持
11.1 器件支持
11.1.1 使用 WEBENCH® 工具创建定制设计
请单击此处,使用 LM43601-Q1 器件并借助 WEBENCH® 电源设计器创建定制设计。
1. 首先键入输入电压 (VIN)、输出电压 (VOUT) 和输出电流 (IOUT) 要求。
2. 使用优化器拨盘优化关键参数设计,如效率、封装和成本。
3. 将生成的设计与德州仪器 (TI) 的其他解决方案进行比较。
WEBENCH 电源设计器可提供定制原理图以及罗列实时价格和组件供货情况的物料清单。
在多数情况下,可执行以下操作:
•
•
•
•
运行电气仿真,观察重要波形以及电路性能
运行热性能仿真,了解电路板热性能
将定制原理图和布局方案导出至常用 CAD 格式
打印设计方案的 PDF 报告并与同事共享
有关 WEBENCH 工具的详细信息,请访问 www.ti.com/WEBENCH。
11.2 接收文档更新通知
要接收文档更新通知,请导航至 TI.com 上的器件产品文件夹。请单击右上角的提醒我 进行注册,即可每周接收产
品信息更改摘要。有关更改的详细信息,请查看任何已修订文档中包含的修订历史记录。
11.3 社区资源
下列链接提供到 TI 社区资源的连接。链接的内容由各个分销商“按照原样”提供。这些内容并不构成 TI 技术规范,
并且不一定反映 TI 的观点;请参阅 TI 的 《使用条款》。
TI E2E™ 在线社区 TI 的工程师对工程师 (E2E) 社区。此社区的创建目的在于促进工程师之间的协作。在
e2e.ti.com 中,您可以咨询问题、分享知识、拓展思路并与同行工程师一道帮助解决问题。
设计支持
TI 参考设计支持 可帮助您快速查找有帮助的 E2E 论坛、设计支持工具以及技术支持的联系信息。
11.4 商标
E2E is a trademark of Texas Instruments.
WEBENCH is a registered trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
11.5 静电放电警告
这些装置包含有限的内置 ESD 保护。 存储或装卸时,应将导线一起截短或将装置放置于导电泡棉中,以防止 MOS 门极遭受静电损
伤。
11.6 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 机械、封装和可订购信息
以下页面包含机械、封装和可订购信息。这些信息是指定器件的最新可用数据。数据如有变更,恕不另行通知和修
订此文档。如欲获取此数据表的浏览器版本,请参阅左侧的导航。
版权 © 2015–2017, Texas Instruments Incorporated
45
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status Package Type Package Pins Package
Eco Plan
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
Samples
Drawing
Qty
(1)
(2)
(3)
(4/5)
(6)
LM43601AQPWPRQ1
LM43601AQPWPTQ1
ACTIVE
ACTIVE
HTSSOP
HTSSOP
PWP
PWP
16
16
2000 RoHS & Green
250 RoHS & Green
NIPDAU
Level-3-260C-168 HR
Level-3-260C-168 HR
-40 to 125
-40 to 125
43601AQ
NIPDAU
43601AQ
LM43601QPWPRQ1
LM43601QPWPTQ1
NRND
NRND
HTSSOP
HTSSOP
PWP
PWP
16
16
2000 RoHS & Green
250 RoHS & Green
NIPDAU
NIPDAU
Level-3-260C-168 HR
Level-3-260C-168 HR
-40 to 125
-40 to 125
43601Q1
43601Q1
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based
flame retardants must also meet the <=1000ppm threshold requirement.
(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6)
Lead finish/Ball material - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead finish/Ball material values may wrap to two
lines if the finish value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
Addendum-Page 1
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
31-Aug-2021
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
B0
K0
P1
W
Pin1
Diameter Width (mm) (mm) (mm) (mm) (mm) Quadrant
(mm) W1 (mm)
LM43601AQPWPRQ1 HTSSOP PWP
LM43601QPWPRQ1 HTSSOP PWP
16
16
2000
2000
330.0
330.0
12.4
12.4
6.9
6.9
5.6
5.6
1.6
1.6
8.0
8.0
12.0
12.0
Q1
Q1
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
31-Aug-2021
*All dimensions are nominal
Device
Package Type Package Drawing Pins
SPQ
Length (mm) Width (mm) Height (mm)
LM43601AQPWPRQ1
LM43601QPWPRQ1
HTSSOP
HTSSOP
PWP
PWP
16
16
2000
2000
350.0
350.0
350.0
350.0
43.0
43.0
Pack Materials-Page 2
PACKAGE OUTLINE
PWP0016G
PowerPAD TM TSSOP - 1.2 mm max height
S
C
A
L
E
2
.
4
0
0
PLASTIC SMALL OUTLINE
C
6.6
6.2
TYP
SEATING PLANE
PIN 1 ID
AREA
A
0.1 C
14X 0.65
16
1
2X
5.1
4.9
4.55
NOTE 3
8
9
0.30
0.19
4.5
4.3
NOTE 4
16X
B
1.2 MAX
0.1
C A
B
0.18
0.12
TYP
SEE DETAIL A
2X 0.24 MAX
NOTE 6
2X 0.56 MAX
NOTE 6
THERMAL
PAD
0.25
GAGE PLANE
3.29
2.71
0.15
0.05
0 - 8
0.75
0.50
DETAIL A
TYPICAL
(1)
2.41
1.77
4218975/B 01/2016
PowerPAD is a trademark of Texas Instruments.
NOTES:
1. All linear dimensions are in millimeters. Any dimensions in parenthesis are for reference only. Dimensioning and tolerancing
per ASME Y14.5M.
2. This drawing is subject to change without notice.
3. This dimension does not include mold flash, protrusions, or gate burrs. Mold flash, protrusions, or gate burrs shall not
exceed 0.15 mm per side.
4. This dimension does not include interlead flash. Interlead flash shall not exceed 0.25 mm per side.
5. Reference JEDEC registration MO-153.
6. Features may not present.
www.ti.com
EXAMPLE BOARD LAYOUT
PWP0016G
PowerPAD TM TSSOP - 1.2 mm max height
PLASTIC SMALL OUTLINE
(3.4)
NOTE 10
(2.41)
SOLDER MASK
OPENING
SOLDER MASK
DEFINED PAD
SEE DETAILS
16X (1.5)
SYMM
1
16
16X (0.45)
(0.95)
TYP
(5)
SYMM
(3.29)
SOLDER MASK
OPENING
14X (0.65)
9
8
(0.95) TYP
METAL COVERED
BY SOLDER MASK
(
0.2) TYP
VIA
(5.8)
LAND PATTERN EXAMPLE
SCALE:10X
METAL UNDER
SOLDER MASK
SOLDER MASK
OPENING
SOLDER MASK
OPENING
METAL
0.05 MIN
ALL AROUND
0.05 MAX
ALL AROUND
SOLDER MASK
DEFINED
NON SOLDER MASK
DEFINED
SOLDER MASK DETAILS
PADS 1-16
4218975/B 01/2016
NOTES: (continued)
7. Publication IPC-7351 may have alternate designs.
8. Solder mask tolerances between and around signal pads can vary based on board fabrication site.
9. This package is designed to be soldered to a thermal pad on the board. For more information, see Texas Instruments literature
numbers SLMA002 (www.ti.com/lit/slma002) and SLMA004 (www.ti.com/lit/slma004).
10. Size of metal pad may vary due to creepage requirement.
www.ti.com
EXAMPLE STENCIL DESIGN
PWP0016G
PowerPAD TM TSSOP - 1.2 mm max height
PLASTIC SMALL OUTLINE
(2.41)
BASED ON
0.127 THICK
STENCIL
16X (1.5)
1
16
16X (0.45)
(3.29)
SYMM
BASED ON
0.127 THICK
STENCIL
14X (0.65)
(R0.05)
9
8
SYMM
(5.8)
SEE TABLE FOR
METAL COVERED
BY SOLDER MASK
DIFFERENT OPENINGS
FOR OTHER STENCIL
THICKNESSES
SOLDER PASTE EXAMPLE
EXPOSED PAD
100% PRINTED SOLDER COVERAGE BY AREA
SCALE:10X
STENCIL
THICKNESS
SOLDER STENCIL
OPENING
0.1
2.69 X 3.68
2.41 X 3.29 (SHOWN)
2.20 X 3.00
0.127
0.152
0.178
2.04 X 2.78
4218975/B 01/2016
NOTES: (continued)
11. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate
design recommendations.
12. Board assembly site may have different recommendations for stencil design.
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