LM4700TF/NOPB [TI]
30W, 1 CHANNEL, AUDIO AMPLIFIER, PZFM11, ISOLATED, PLASTIC, TO-220, 11 PIN;
| 型号： | LM4700TF/NOPB |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | 30W, 1 CHANNEL, AUDIO AMPLIFIER, PZFM11, ISOLATED, PLASTIC, TO-220, 11 PIN 局域网 放大器 商用集成电路 |
| 文件： | 总20页 (文件大小：701K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
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LM4700
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LM4700 Overture™ Audio Power Amplifier Series
30W Audio Power Amplifier with
Mute and Standby Modes
Check for Samples: LM4700
1
FEATURES
DESCRIPTION
The LM4700 is an audio power amplifier capable of
delivering typically 30W of continuous average output
power into an 8Ω load with less than 0.1% THD+N.
234
•
SPiKe Protection
•
Minimal Amount of External Components
Necessary
The LM4700 has an independent smooth transition
fade-in/out mute and a power conserving standby
mode which can be controlled by external logic.
•
•
•
•
Quiet Fade-in/out Mute Function
Power Conserving Standby-mode
Isolated 11-lead TO-220 Package
Wide Supply Range 20V - 66V
The performance of the LM4700, utilizing its Self
Peak Instantaneous Temperature (°Ke) ( SPiKe™)
protection circuitry, places it in a class above discrete
and hybrid amplifiers by providing an inherently,
dynamically protected Safe Operating Area (SOA).
SPiKe protection means that these parts are
completely safeguarded at the output against
overvoltage, undervoltage, overloads, including
thermal runaway and instantaneous temperature
peaks.
APPLICATIONS
•
•
Component Stereo
Compact Stereo
KEY SPECIFICATIONS
•
THD+N at 1kHz at Continuous Average Output
Power of 25W into 8Ω: 0.1% (max)
•
THD+N from 20Hz to 20kHz at 30W of
Continuous Average Output Power into 8Ω:
0.08% (typ)
•
Standby Current: 2.1mA (typ)
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
2
3
4
Overture is a trademark of Texas Instruments.
SPiKe is a trademark of dcl_owner.
All other trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
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Typical Application
*Optional components dependent upon specific design requirements. Refer to the External Components Description
section for a component functional description.
Figure 1. Typical Audio Amplifier Application Circuit
Connection Diagram
Figure 2. Isolated Plastic Package - Top View
See Package Number NDA0011B for Staggered Lead Isolated Package
See NDJ0011B for Staggered Lead Non-Isolated Package
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
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Absolute Maximum Ratings(1)(2)(3)
Supply Voltage |VCC| + |VEE| (No Signal)
Supply Voltage |VCC| + |VEE| (with Input and Load)
Common Mode Input Voltage
Differential Input Voltage
66V
64V
(VCC or VEE) and |VCC| + |VEE| ≤ 60V
60V
Internally Limited
62.5W
Output Current
Power Dissipation(4)
ESD Susceptibility(5)
2000V
Junction Temperature(6)
150°C
(7)
Thermal Resistance
θJC
θJA
2°C/W
43°C/W
Soldering Information
Storage Temperature
NDA Package (10 sec.)
260°C
−40°C ≤ TA ≤ +150°C
(1) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical
specifications under particular test conditions which guarantee specific performance limits. This assumes that the device is within the
Operating Ratings. Specifications are not guaranteed for parameters where no limit is given, however, the typical value is a good
indication of device performance.
(2) All voltages are measured with respect to the GND (pin 7), unless otherwise specified.
(3) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and
specifications.
(4) For operating at case temperatures above 25°C, the device must be derated based on a 150°C maximum junction temperature and a
thermal resistance of θJC = 2°C/W (junction to case). Refer to the section, DETERMINING THE CORRECT HEAT SINK, in the
Application Information section.
(5) Human body model, 100 pF discharged through a 1.5 kΩ resistor.
(6) The operating junction temperature maximum is 150°C, however, the instantaneous Safe Operating Area temperature is 250°C.
(7) Preliminary engineering evaluation of θJC for the NDA package has been assessed as 2°C/W. This is a preliminary engineering number
and represents the data to this point. Please contact your local TI sales representative for more information.
Operating Ratings(1)(2)
Temperature Range
TMIN ≤ TA ≤ TMAX
−20°C ≤ TA ≤ +85°C
(3)
Supply Voltage |VCC| + |VEE
|
20V to 64V
(1) All voltages are measured with respect to the GND (pin 7), unless otherwise specified.
(2) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical
specifications under particular test conditions which guarantee specific performance limits. This assumes that the device is within the
Operating Ratings. Specifications are not guaranteed for parameters where no limit is given, however, the typical value is a good
indication of device performance.
(3) Operation is guaranteed up to 64V, however, distortion may be introduced from SPiKe Protection Circuitry if proper thermal
considerations are not taken into account. Refer to the Application Information section for a complete explanation.
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Electrical Characteristics
(1)(2)The following specifications are for VCC = +28V, VEE = −28V with RL = 8Ω, unless otherwise specified. Limits apply for TA
= 25°C.
LM4700
Units
Typical(3 Limit(
Symbol
Parameter
Conditions
(Limits)
)
4)
|VCC| + |VEE
|
Power Supply Voltage(5)
GND − VEE ≥ 9V
18
20
64
V (min)
V (max)
(6)
PO
Output Power (Continuous Average) THD + N = 0.1% (max), f = 1 kHz
RL = 8Ω, |VCC| = |VEE| = 28V
30
22
25
15
W/ch (min)
W/ch (min)
%
RL = 4Ω, |VCC| = |VEE| = 20V(7)
THD + N
SR(6)
Total Harmonic Distortion Plus
Noise
30W/ch, RL = 8Ω,
0.08
20 Hz ≤ f ≤ 20 kHz, AV = 26 dB
VIN = 1.414 Vrms, trise = 2 ns
Slew Rate
18
12
40
V/μs (min)
(8)
ITOTAL
Total Quiescent Power Supply
Current
VCM = 0V, VO = 0V, IO = 0 mA
Standby: Off
25
mA (max)
mA
Standby: On
2.1
Standby Pin
VIL
VIH
Standby Low Input Voltage
Standby High Input Voltage
Not in Standby Mode
In Standby Mode
0.8
2.5
V (max)
V (min)
2.0
Mute Pin
VIL
Mute Low Input Voltage
Mute High Input Voltage
Mute Attenuation
Output Not Muted
Output Muted
0.8
2.5
80
V (max)
V (min)
VIH
2.0
115
2.0
AM
VPIN8 = 2.5V
dB (min)
mV (max)
μA (max)
μA (max)
APK (min)
(8)
VOS
Input Offset Voltage
Input Bias Current
VCM = 0V, IO = 0 mA
VCM = 0V, IO = 0 mA
VCM = 0V, IO = 0 mA
15
IB
0.2
0.5
0.2
2.9
IOS
IO
Input Offset Current
Output Current Limit
0.002
3.5
|VCC| = |VEE| = 10V, tON = 10 ms,
VO = 0V
(8)
VOD
Output Dropout Voltage(9)
|VCC − VO|, VCC = 20V, IO = +100 mA
|VO − VEE|, VEE = −20V, IO = −100 mA
1.8
2.5
2.3
3.2
85
V (max)
V (max)
dB (min)
PSRR(8)
CMRR(8)
Power Supply Rejection Ratio
VCC = 30V to 10V, VEE = −30V,
VCM = 0V, IO = 0 mA
115
VCC = 30V, VEE = −30V to −10V
VCM = 0V, IO = 0 mA
110
110
85
80
dB (min)
dB (min)
Common Mode Rejection Ratio
VCC = 35V to 10V, VEE = −10V to −35V,
VCM = 10V to −10V, IO = 0 mA
(8)
AVOL
Open Loop Voltage Gain
Gain-Bandwidth Product
RL = 2 kΩ, ΔVO = 30V
110
7.5
90
5
dB (min)
GBWP
fO = 100 kHz, VIN = 50 mVrms
MHz (min)
(1) All voltages are measured with respect to the GND (pin 7), unless otherwise specified.
(2) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical
specifications under particular test conditions which guarantee specific performance limits. This assumes that the device is within the
Operating Ratings. Specifications are not guaranteed for parameters where no limit is given, however, the typical value is a good
indication of device performance.
(3) Typicals are measured at 25°C and represent the parametric norm.
(4) Limits are guarantees that all parts are tested in production to meet the stated values.
(5) VEE must have at least −9V at its pin with reference to ground in order for the under-voltage protection circuitry to be disabled. In
addition, the voltage differential between VCC and VEE must be greater than 14V.
(6) AC Electrical Test; Refer to Test Circuit #2 .
(7) For a 4Ω load, and with ±20V supplies, the LM4700 can deliver typically 22 Watts of continuous average power per channel with less
than 0.1% (THD+N). With supplies above ±20V, the LM4700 cannot deliver more than 22 watts into 4Ω due to current limiting of the
output transistors. Thus, increasing the power supply above ±20V will only increase the internal power dissipation, not the possible
output power. Increased power dissipation will require a larger heat sink as explained in the Application Information section.
(8) DC Electrical Test; Refer to Test Circuit #1 .
(9) The output dropout voltage, VOD, is the supply voltage minus the clipping voltage. Refer to the Clipping Voltage vs. Supply Voltage
graph in the Typical Performance Characteristics section.
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Electrical Characteristics (continued)
(1) (2) The following specifications are for VCC = +28V, VEE = −28V with RL = 8Ω, unless otherwise specified. Limits apply for TA
= 25°C.
LM4700
Units
Typical(3 Limit(
Symbol
Parameter
Conditions
IHF—A Weighting Filter
(Limits)
)
4)
(6)
eIN
Input Noise
Signal-to-Noise Ratio
2.0
8
μV (max)
RIN = 600Ω (Input Referred)
SNR
PO = 1W, A-Weighted,
Measured at 1 kHz, RS = 25Ω
98
dB
dB
PO = 25W, A-Weighted
108
Measured at 1 kHz, RS = 25Ω
Test Circuit #1
(10)(DC Electrical Test Circuit)
Test Circuit #2
(11)(AC Electrical Test Circuit)
(10) DC Electrical Test; Refer to Test Circuit #1 .
(11) AC Electrical Test; Refer to Test Circuit #2 .
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(11) (AC Electrical Test Circuit)
Bridged Amplifier Application Circuit
Figure 3. Bridged Amplifier Application Circuit
Single Supply Application Circuit
Figure 4. Single Supply Amplifier Application Circuit
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(11) (AC Electrical Test Circuit)
Auxillary Amplifier Application Circuit
Figure 5. Auxillary Amplifier Application Circuit
Equivalent Schematic
(Excluding Active Protection Circuitry)
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(11) (AC Electrical Test Circuit)
External Components Description
Components
Functonal Description
1
2
RB
RI
Prevents currents from entering the amplifier's non-inverting input which may be passed through to the load upon
power down of the system due to the low input impedance of the circuitry when the undervoltage circuitry is off.
This phenomenon occurs when the supply voltages are below 1.5V.
Inverting input resistance to provide AC gain in conjunction with RF. Also creates a highpass filter with CI at fC
=
1/(2πRICI).
3
4
5
RF
CI(1)
Feedback resistance to provide AC gain in conjunction with RI.
Feedback capacitor which ensures unity gain at DC.
CS
Provides power supply filtering and bypassing. Refer to the Supply Bypassing application section for proper
placement and selection of bypass capacitors.
6
7
RV
Acts as a volume control by setting the input voltage level.
(1)
RIN
Sets the amplifier's input terminals DC bias point when CIN is present in the circuit. Also works with CIN to create a
highpass filter at fC = 1/(2πRINCIN). Refer to Figure 5.
(1)
8
CIN
Input capacitor which blocks the input signal's DC offsets from being passed onto the amplifier's inputs.
(1)
9
RSN
Works with CSN to stabilize the output stage by creating a pole that reduces high frequency instabilities. The pole
is set at fC = 1/(2πRSNCSN). Refer to Figure 5.
(1)
10
CSN
Works with RSN to stabilize the output stage by creating a pole that reduces high frequency instabilities.
(2)
(2)
11
12
L
Provides high impedance at high frequencies so that R may decouple a highly capacitive load and reduce the Q of
the series resonant circuit. Also provides a low impedance at low frequencies to short out R and pass audio
signals to the load. Refer to Figure 5.
(2)
R
13
14
15
RA
CA
Provides DC voltage biasing for the transistor Q1 in single supply operation.
Provides bias filtering for single supply operation.
RINP
Limits the voltage difference between the amplifier's inputs for single supply operation. Refer to the CLICKS AND
(2)
POPS application section for a more detailed explanation of the function of RINP
Provides input bias current for single supply operation. Refer to the CLICKS AND POPS application section for a
more detailed explanation of the function of RBI
.
16
17
RBI
RE
.
Establishes a fixed DC current for the transistor Q1 in single supply operation. This resistor stabilizes the half-
supply point along with CA.
(1) Optional components dependent upon specific design requirements.
(2) Optional components dependent upon specific design requirements.
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Typical Performance Characteristics
THD + N
vs
Frequency
THD + N
vs
Frequency
Figure 6.
Figure 7.
THD + N
vs
Frequency
THD + N
vs
Output Power
Figure 8.
Figure 9.
THD + N
vs
Output Power
THD + N
vs
Output Power
Figure 10.
Figure 11.
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Typical Performance Characteristics (continued)
THD + N
vs
Output Power
THD + N
vs
Output Power
Figure 12.
Figure 13.
THD + N
vs
Output Power
Clipping Voltage vs
Supply Voltage
Figure 14.
Figure 15.
Clipping Voltage vs
Supply Voltage
Clipping Voltage vs
Supply Voltage
Figure 16.
Figure 17.
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Typical Performance Characteristics (continued)
Power Dissipation vs
Output Power
Power Dissipation vs
Ouput Power
Figure 18.
Figure 19.
Power Dissipation vs
Output Power
Output Power vs
Load Resistance
Figure 20.
Figure 21.
Output Power vs
Supply Voltage
Output Mute vs
Mute Pin Voltage
Figure 22.
Figure 23.
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Typical Performance Characteristics (continued)
Pulse Response
Large Signal Response
Figure 24.
Figure 25.
Output Mute vs
Mute Pin Voltage
Power Supply
Rejection Ratio
Figure 26.
Figure 27.
Common-Mode
Rejection Ratio
Open Loop
Frequency Response
Figure 28.
Figure 29.
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Typical Performance Characteristics (continued)
Safe Area
Spike Protection Response
Figure 30.
Figure 31.
Supply Current vs
Supply Voltage
Pulse Thermal
Resistance
Figure 32.
Figure 33.
Pulse Thermal
Resistance
Supply Current vs
Output Voltage
Figure 34.
Figure 35.
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Typical Performance Characteristics (continued)
Pulse Power Limit
Pulse Power Limit
Figure 36.
Figure 37.
Supply Current vs
Case Temperature
Standby Current (ICC) vs
Standby Pin Voltage
Figure 38.
Figure 39.
Supply Current (IEE) vs
Standby Pin Voltage
Input Bias Current vs
Case Temperature
Figure 40.
Figure 41.
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APPLICATION INFORMATION
MUTE MODE
By placing a logic-high voltage on the mute pin, the signal going into the amplifiers will be muted. If the mute pin
is left floating or connected to a logic-low level, the amplifier will be in a non-muted state. Refer to the Typical
Performance Characteristics section for curves concerning Mute Attenuation vs Mute Pin Voltage.
STANDBY MODE
The standby mode of the LM4700 allows the user to drastically reduce power consumption when the amplifier is
idle. By placing a logic-high voltage on the standby pin, the amplifier will go into Standby Mode. In this mode, the
current drawn from the VCC supply is typically less than 10 μA total for both amplifiers. The current drawn from
the VEE supply is typically 2.1 mA. Clearly, there is a significant reduction in idle power consumption when using
the standby mode. Refer to the Typical Performance Characteristics section for curves showing Supply Current
vs Standby Pin Voltage for both supplies.
UNDER-VOLTAGE PROTECTION
Upon system power-up, the under-voltage protection circuitry allows the power supplies and their corresponding
capacitors to come up close to their full values before turning on the LM4700 such that no DC output spikes
occur. Upon turn-off, the output of the LM4700 is brought to ground before the power supplies such that no
transients occur at power-down.
OVER-VOLTAGE PROTECTION
The LM4700 contains over-voltage protection circuitry that limits the output current to approximately 3.5 Apk
while also providing voltage clamping, though not through internal clamping diodes. The clamping effect is quite
the same, however, the output transistors are designed to work alternately by sinking large current spikes.
SPiKe PROTECTION
The LM4700 is protected from instantaneous peak-temperature stressing of the power transistor array. The Safe
Operating Area graph in the Typical Performance Characteristics section shows the area of device operation
where SPiKe Protection Circuitry is not enabled. The waveform to the right of the SOA graph exemplifies how
the dynamic protection will cause waveform distortion when enabled.
THERMAL PROTECTION
The LM4700 has a sophisticated thermal protection scheme to prevent long-term thermal stress of the device.
When the temperature on the die reaches 165°C, the LM4700 shuts down. It starts operating again when the die
temperature drops to about 155°C, but if the temperature again begins to rise, shutdown will occur again at
165°C. Therefore, the device is allowed to heat up to a relatively high temperature if the fault condition is
temporary, but a sustained fault will cause the device to cycle in a Schmitt Trigger fashion between the thermal
shutdown temperature limits of 165°C and 155°C. This greatly reduces the stress imposed on the IC by thermal
cycling, which in turn improves its reliability under sustained fault conditions.
Since the die temperature is directly dependent upon the heat sink used, the heat sink should be chosen such
that thermal shutdown will not be reached during normal operation. Using the best heat sink possible within the
cost and space constraints of the system will improve the long-term reliability of any power semiconductor
device, as discussed in the DETERMINING THE CORRECT HEAT SINK Section.
DETERMINING MAXIMUM POWER DISSIPATION
Power dissipation within the integrated circuit package is a very important parameter requiring a thorough
understanding if optimum power output is to be obtained. An incorrect maximum power dissipation calculation
may result in inadequate heat sinking causing thermal shutdown and thus limiting the output power.
Equation (1) exemplifies the theoretical maximum power dissipation point of each amplifier where VCC is the total
supply voltage.
PDMAX = VCC2/2π2RL
(1)
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Thus by knowing the total supply voltage and rated output load, the maximum power dissipation point can be
calculated. Refer to the graphs of Power Dissipation vs Output Power in the Typical Performance Characteristics
section which show the actual full range of power dissipation not just the maximum theoretical point that results
from equation (1).
DETERMINING THE CORRECT HEAT SINK
The choice of a heat sink for a high-power audio amplifier is made entirely to keep the die temperature at a level
such that the thermal protection circuitry does not operate under normal circumstances.
The thermal resistance from the die (junction) to the outside air (ambient) is a combination of three thermal
resistances, θJC, θCS and θSA. The thermal resistance, θJC (junction to case), of the LM4700 is 2°C/W. Using
Thermalloy Thermacote thermal compound, the thermal resistance, θCS (case to sink), is about 0.2°C/W. Since
convection heat flow (power dissipation) is analogous to current flow, thermal resistance is analogous to
electrical resistance, and temperature drops are analogous to voltage drops, the power dissipation out of the
LM4700 is equal to the following:
PDMAX = (TJMAX − TAMB)/θJA
where
•
•
•
TJMAX = 150°C
TAMB is the system ambient temperature
θJA = θJC + θCS + θSA
(2)
Once the maximum package power dissipation has been calculated using equation (1), the maximum thermal
resistance, θSA, (in °C/W) for a heat sink can be calculated. This calculation is made using equation (3) which is
derived by solving for θSA in equation (2).
θSA=[(TJMAX−TAMB)−PDMAX(θJC+θCS)]/PDMAX
(3)
Again it must be noted that the value of θSA is dependent upon the system designer's amplifier requirements. If
the ambient temperature that the audio amplifier is to be working under is higher than 25°C, then the thermal
resistance for the heat sink, given all other things are equal, will need to be smaller.
SUPPLY BYPASSING
The LM4700 has excellent power supply rejection and does not require a regulated supply. However, to improve
system performance as well as eliminate possible oscillations, the LM4700 should have its supply leads
bypassed with low-inductance capacitors having short leads that are located close to the package terminals.
Inadequate power supply bypassing will manifest itself by a low frequency oscillation known as “motorboating” or
by high frequency instabilities. These instabilities can be eliminated through multiple bypassing utilizing a large
tantalum or electrolytic capacitor (10 μF or larger) which is used to absorb low frequency variations and a small
ceramic capacitor (0.1 μF) to prevent any high frequency feedback through the power supply lines.
If adequate bypassing is not provided, the current in the supply leads which is a rectified component of the load
current may be fed back into internal circuitry. This signal causes distortion at high frequencies requiring that the
supplies be bypassed at the package terminals with an electrolytic capacitor of 470 μF or more.
BRIDGED AMPLIFIER APPLICATION
One common power amplifier configuration is shown in Figure 3 and is referred to as “bridged mode” operation.
Bridged mode operation is different from the classical single-ended amplifier configuration where one side of the
output load is connected to ground.
A bridge amplifier design has a distinct advantage over the single-ended configuration, as it provides differential
drive to the load, thus doubling output swing for a specified supply voltage. Consequently, theoretically four times
the output power is possible as compared to a single-ended amplifier under the same conditions. This increase in
attainable output power assumes that the amplifier is not current limited or clipped.
A direct consequence of the increased power delivered to the load by a bridge amplifier is an increase in internal
power dissipation. For each operational amplifier in a bridge configuration, the internal power dissipation will
increase by a factor of two over the single ended dissipation. Since there are two amplifiers used in a bridge
configuration, the maximum system power dissipation point will increase by a factor of four over the figure
obtained by equation (1).
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This value of PDMAX can be used to calculate the correct size heat sink for a bridged amplifier application,
assuming that both IC's are mounted on the same heatsink. Since the internal dissipation for a given power
supply and load is increased by using bridged-mode, the heatsink's θSA will have to decrease accordingly as
shown by equation (3). Refer to the section, DETERMINING THE CORRECT HEAT SINK, for a more detailed
discussion of proper heat sinking for a given application.
SINGLE-SUPPLY AMPLIFIER APPLICATION
The typical application of the LM4700 is a split supply amplifier. But as shown in Figure 4, the LM4700 can also
be used in a single power supply configuration. This involves using some external components to create a half-
supply bias which is used as the reference for the inputs and outputs. Thus, the signal will swing around half-
supply much like it swings around ground in a split-supply application. Along with proper circuit biasing, a few
other considerations must be accounted for to take advantage of all of the LM4700 functions.
The LM4700 possesses a mute and standby function with internal logic gates that are half-supply referenced.
Thus, to enable either the mute or standby function, the voltage at these pins must be a minimum of 2.5V above
half-supply. In single-supply systems, devices such as microprocessors and simple logic circuits used to control
the mute and standby functions, are usually referenced to ground, not half-supply. Thus, to use these devices to
control the logic circuitry of the LM4700, a “level shifter”, like the one shown in Figure 42, must be employed. A
level shifter is not needed in a split-supply configuration since ground is also half-supply.
Figure 42. Level Shift Circuit
When the voltage at the Logic Input node is 0V, the 2N3904 is “off” and thus resistor RC pulls up mute or standby
input to the supply. This enables the mute or standby function. When the Logic Input is 5V, the 2N3904 is “on”
and consequently, the voltage at the collector is essentially 0V. This will disable the mute or standby function,
and thus the amplifier will be in its normal mode of operation. RSHIFT, along with CSHIFT, creates an RC time
constant that reduces transients when the mute or standby functions are enabled or disabled. Additionally, RSHIFT
limits the current supplied by the internal logic gates of the LM4700 which insures device reliability. Refer to the
MUTE MODE and STANDBY MODE sections in the Application Information section for a more detailed
description of these functions.
CLICKS AND POPS
In the typical application of the LM4700 as a split-supply audio power amplifier, the IC exhibits excellent “click”
and “pop” performance when utilizing the mute and standby functions. In addition, the device employs Under-
Voltage Protection, which eliminates unwanted power-up and power-down transients. The basis for these
functions are a stable and constant half-supply potential. In a split-supply application, ground is the stable half-
supply potential. But in a single-supply application, the half-supply needs to charge up just like the supply rail,
VCC
.
This makes the task of attaining a clickless and popless turn-on more challenging. Any uneven charging of the
amplifier inputs will result in output clicks and pops due to the differential input topology of the LM4700.
To achieve a transient free power-up and power-down, the voltage seen at the input terminals should be ideally
the same. Such a signal will be common-mode in nature, and will be rejected by the LM4700. In Figure 4, the
resistor RINP serves to keep the inputs at the same potential by limiting the voltage difference possible between
the two nodes. This should significantly reduce any type of turn-on pop, due to an uneven charging of the
amplifier inputs. This charging is based upon a specific application loading and thus, the system designer may
need to adjust these values for optimum performance.
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As shown in Figure 4, the resistors labeled RBI help bias up the LM4700 off the half-supply node at the emitter of
the 2N3904. But due to the input and output coupling capacitors in the circuit, along with the negative feedback,
there are two different values of RBI, namely 10 kΩ and 200 kΩ. These resistors bring up the inputs at the same
rate resulting in a popless turn-on. Adjusting these resistors values slightly may reduce pops resulting from
power supplies that ramp extremely quick or exhibit overshoot during system turn-on.
AUDIO POWER AMPLlFIER DESIGN
Design a 25W/8Ω Audio Amplifier
Given:
Power Output
Load Impedance
Input Level
25 Wrms
8Ω
1 Vrms(max)
Input Impedance
Bandwidth
47 kΩ
20 Hz to 20 kHz ± 0.25 dB
A designer must first determine the power supply requirements in terms of both voltage and current needed to
obtain the specified output power. VOPEAK can be determined from equation (4) and IOPEAK from equation (5).
(4)
(5)
To determine the maximum supply voltage, the following conditions must be considered. Add the dropout voltage
to the peak output swing VOPEAK, to get the supply rail at a current of IOPEAK. The regulation of the supply
determines the unloaded voltage which is usually about 15% higher. The supply voltage will also rise 10% during
high line conditions. Therefore the maximum supply voltage is obtained from the following equation:
Max Supplies ≈ ± (VOPEAK + VOD) (1 + Regulation) (1.1)
For 25W of output power into an 8Ω load, the required VOPEAK is 20V. A minimum supply rail of ±25V results from
adding VOPEAK and VOD. With regulation, the maximum supplies are ±31.7V and the required IOPEAK is 2.5A from
equation (5). At this point it is a good idea to check the Power Output vs Supply Voltage to ensure that the
required output power is obtainable from the device while maintaining low THD+N. In addition, the designer
should verify that with the required power supply voltage and load impedance, that the required heatsink value
θSA is feasible given system cost and size constraints. Once the heatsink issues have been addressed, the
required gain can be determined from equation (6).
(6)
From equation (6), the minimum AV is AV ≥ 14.14.
By selecting a gain of 21, and with a feedback resistor, RF = 20 kΩ, the value of RI follows from equation (7).
RI = RF (AV − 1)
(7)
Thus with RJ = 1 kΩ a non-inverting gain of 21 will result. Since the desired input impedance was 47 kΩ, a value
of 47 kΩ was selected for RIN. The final design step is to address the bandwidth requirements which must be
stated as a pair of −3 dB frequency points. Five times away from a −3 dB point is 0.17 dB down from passband
response which is better than the required ±0.25 dB specified. This fact results in a low and high frequency pole
of 4 Hz and 100 kHz respectively. As stated in the External Components Description section, RI in conjunction
with CI create a high-pass filter.
CI ≥ 1/(2π * 1 kΩ * 4 Hz) = 39.8 μF; use 39 μF.
(8)
The high frequency pole is determined by the product of the desired high frequency pole, fH, and the gain, AV.
With a AV = 21 and fH = 100 kHz, the resulting GBWP of 2.1 MHz is less than the minimum GBWP of 5 MHz for
the LM4700. This will ensure that the high frequency response of the amplifier will be no worse than 0.17 dB
down at 20 kHz which is well within the bandwidth requirements of the design.
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LM4700
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REVISION HISTORY
Changes from Revision C (April 2013) to Revision D
Page
•
Changed layout of National Data Sheet to TI format .......................................................................................................... 18
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