LM5000-3MTCX [TI]
High Voltage Switch Mode Regulator;
| 型号： | LM5000-3MTCX |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | High Voltage Switch Mode Regulator |
| 文件： | 总24页 (文件大小：1372K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LM5000
www.ti.com
SNVS176D –MAY 2004–REVISED MARCH 2007
LM5000 High Voltage Switch Mode Regulator
Check for Samples: LM5000
1
FEATURES
DESCRIPTION
The LM5000 is
a monolithic integrated circuit
2
•
•
•
80V Internal Switch
specifically designed and optimized for flyback, boost
or forward power converter applications. The internal
power switch is rated for a maximum of 80V, with a
current limit set to 2A. Protecting the power switch
are current limit and thermal shutdown circuits. The
current mode control scheme provides excellent
rejection of line transients and cycle-by-cycle current
limiting. An external compensation pin and the built-in
slope compensation allow the user to optimize the
frequency compensation. Other distinctive features
include softstart to reduce stresses during start-up
and an external shutdown pin for remote ON/OFF
control. There are two operating frequency ranges
available. The LM5000-3 is pin selectable for either
300kHz (FS Grounded) or 700kHz (FS Open). The
LM5000-6 is pin selectable for either 600kHz (FS
Grounded) or 1.3MHz (FS Open). The device is
available in a low profile 16-lead TSSOP package or
a thermally enhanced 16-lead WSON package.
Operating Input Voltage Range of 3.1V to 40V
Pin Selectable Operating Frequency
–
–
300kHz/700kHz (-3)
600kHz/1.3MHz (-6)
•
•
•
•
•
•
•
•
Adjustable Output Voltage
External Compensation
Input Undervoltage Lockout
Softstart
Current Limit
Over Temperature Protection
External Shutdown
Small 16-Lead TSSOP or 16-Lead WSON
Package
APPLICATIONS
•
•
•
•
•
Flyback Regulator
Forward Regulator
Boost Regulator
DSL Modems
Distributed Power Converters
Typical Application Circuit
Figure 1. LM5000 Flyback Converter
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
2
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2004–2007, Texas Instruments Incorporated
LM5000
SNVS176D –MAY 2004–REVISED MARCH 2007
www.ti.com
Connection Diagram
Figure 2. Top View
PIN DESCRIPTIONS
Pin
Name
Function
1
COMP
Compensation network connection. Connected to the output of the voltage error amplifier. The RC
compenstion network should be connected from this pin to AGND. An additional 100pF high frequency
capacitor to AGND is recommended.
2
3
FB
Output voltage feedback input.
SHDN
AGND
PGND
PGND
PGND
PGND
SW
Shutdown control input, Open = enable, Ground = disable.
Analog ground, connect directly to PGND.
Power ground.
4
5
6
Power ground.
7
Power ground.
8
Power ground.
9
Power switch input. Switch connected between SW pins and PGND pins
Power switch input. Switch connected between SW pins and PGND pins
Power switch input. Switch connected between SW pins and PGND pins
Bypass-Decouple Capacitor Connection, 0.1µF ceramic capacitor recommended.
10
11
12
13
SW
SW
BYP
VIN
Analog power input. A small RC filter is recommended, to suppress line glitches. Typical values of 10Ω
and ≥ 0.1µF are recommended.
14
15
16
-
SS
FS
Softstart Input. External capacitor and internal current source sets the softstart time.
Switching frequency select input. Open = Fhigh. Ground = Flow
Factory test pin, connect to ground.
TEST
Exposed Pad
underside of WSON
package
Connect to system ground plane for reduced thermal resistance.
2
Submit Documentation Feedback
Copyright © 2004–2007, Texas Instruments Incorporated
Product Folder Links: LM5000
LM5000
www.ti.com
SNVS176D –MAY 2004–REVISED MARCH 2007
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
Absolute Maximum Ratings(1)(2)
VIN
-0.3V to 40V
-0.3V to 80V
-0.3V to 5V
-0.3V to 3V
-0.3V to 7V
150°C
SW Voltage
FB Voltage
COMP Voltage
All Other Pins
Maximum Junction Temperature
Power Dissipation(3)
Lead Temperature
Infrared (15 sec.)
ESD Susceptibility(4)
Internally Limited
216°C
235°C
Human Body Model
Machine Model
2kV
200V
Storage Temperature
−65°C to +150°C
(1) Absolute maximum ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions for which the
device is intended to be functional, but device parameter specifications may not be ensured. For ensured specifications and test
conditions, see the Electrical Characteristics.
(2) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and
specifications.
(3) The maximum allowable power dissipation is a function of the maximum junction temperature, TJ(MAX), the junction-to-ambient thermal
resistance, θJA, and the ambient temperature, TA. See the Electrical Characteristics table for the thermal resistance of various layouts.
The maximum allowable power dissipation at any ambient temperature is calculated using: PD (MAX) = (TJ(MAX) − TA)/θJA. Exceeding
the maximum allowable power dissipation will cause excessive die temperature, and the regulator will go into thermal shutdown.
(4) The human body model is a 100 pF capacitor discharged through a 1.5kΩ resistor into each pin. The machine model is a 200pF
capacitor discharged directly into each pin.
Operating Conditions
Operating Junction Temperature Range(1)
Supply Voltage(1)
−40°C to +125°C
3.1V to 40V
(1) Supply voltage, bias current product will result in aditional device power dissipation. This power may be significant. The thermal
dissipation design should take this into account.
Electrical Characteristics
Specifications in standard type face are for TJ = 25°C and those with boldface type apply over the full Operating
Temperature Range (TJ = −40°C to +125°C) Unless otherwise specified. VIN = 12V and IL = 0A, unless otherwise specified.
Symbol
Parameter
Quiescent Current
Conditions
Min(1)
Typ(2)
Max(1)
Units
IQ
FB = 2V (Not Switching)
FS = 0V
2.0
2.5
mA
FB = 2V (Not Switching)
FS = Open
2.1
2.5
mA
VSHDN = 0V
18
30
1.2840
2.7
µA
V
VFB
Feedback Voltage
1.2330
1.35
1.259
2.0
ICL
Switch Current Limit
A
%VFB/ΔVIN
Feedback Voltage Line
Regulation
3.1V ≤ VIN ≤ 40V
0.001
0.04
%/V
IB
FB Pin Bias Current(3)
55
200
nA
(1) All limits specified at room temperature (standard typeface) and at temperature extremes (bold typeface). All room temperature limits are
100% production tested. All limits at temperature extremes are specified via correlation using standard Statistical Quality Control (SQC)
methods. All limits are used to calculate Average Outgoing Quality Level (AOQL).
(2) Typical numbers are at 25°C and represent the most likely norm.
(3) Bias current flows into FB pin.
Copyright © 2004–2007, Texas Instruments Incorporated
Submit Documentation Feedback
3
Product Folder Links: LM5000
LM5000
SNVS176D –MAY 2004–REVISED MARCH 2007
www.ti.com
Electrical Characteristics (continued)
Specifications in standard type face are for TJ = 25°C and those with boldface type apply over the full Operating
Temperature Range (TJ = −40°C to +125°C) Unless otherwise specified. VIN = 12V and IL = 0A, unless otherwise specified.
Symbol
BV
Parameter
Conditions
Min(1)
Typ(2)
Max(1)
Units
Output Switch Breakdown
Voltage
TJ = 25°C, ISW = 0.1µA
80
V
TJ = -40°C to + 125°C, ISW
0.5µA
=
76
VIN
gm
Input Voltage Range
3.1
40
V
µmho
V/V
%
Error Amp Transconductance
Error Amp Voltage Gain
ΔI = 5µA
150
410
280
90
750
AV
DMAX
Maximum Duty Cycle
LM5000-3
FS = 0V
FS = 0V
85
85
Maximum Duty Cycle
LM5000-6
90
%
TMIN
fS
Minimum On Time
165
300
700
600
1.3
ns
Switching Frequency LM5000- FS = 0V
3
240
550
360
840
715
1.545
-2
FS = Open
kHz
Switching Frequency LM5000- FS = 0V
6
485
FS = Open
1.055
MHz
µA
µA
mΩ
V
ISHDN
IL
Shutdown Pin Current
Switch Leakage Current
Switch RDSON
VSHDN = 0V
VSW = 80V
ISW = 1A
−1
0.008
160
0.6
5
RDSON
ThSHDN
445
SHDN Threshold
Output High
Output Low
0.9
0.6
0.3
V
UVLO
On Threshold
Off Threshold
VCOMP Trip
2.74
2.60
2.92
2. 77
0.67
11
3.10
2.96
V
V
OVP
ISS
V
Softstart Current
Thermal Resistance
8
14
µA
°C/W
θJA
TSSOP, Package only
WSON, Package only
150
45
4
Submit Documentation Feedback
Copyright © 2004–2007, Texas Instruments Incorporated
Product Folder Links: LM5000
LM5000
www.ti.com
SNVS176D –MAY 2004–REVISED MARCH 2007
Typical Performance Characteristics
Iq (non-switching) vs VIN @ fSW = 300kHz
3.000
Iq (non-switching) vs VIN @ fSW = 700kHz
3.000
2.800
2.600
2.400
2.200
2.000
1.800
1.600
1.400
2.800
2.600
2.400
2.200
2.000
1.800
1.600
1.400
-40oC
-40oC
25oC
25oC
125oC
125oC
1.200
1.000
1.200
1.000
10 15
0
5
20 25
VIN (V)
35 40
30
10 15
0
5
20 25
VIN (V)
35 40
30
Figure 3.
Figure 4.
Iq (switching) vs VIN @ fSW = 300kHz
Iq (switching) vs VIN @ fSW = 700kHz
10
9
10
9
-40oC
8
8
25oC
125oC
7
7
6
6
-40oC
5
5
4
4
125oC
25oC
3
3
2
2
0
5
10 15 20 25 30 35 40
0
5
10 15 20 25 30 35 40
VIN (V)
VIN (V)
Figure 5.
Figure 6.
Vfb vs Temperature
RDS(ON) vs VIN @ ISW =1A
1.2800
1.2700
1.2600
1.2500
1.2400
1.2300
400
350
300
250
200
150
100
50
125oC
25oC
-40oC
0
-40 -20
0
20 40 60 80 100 120
0
5
10 15 20 25 30 35 40
TEMPERATURE (oC)
VIN (V)
Figure 7.
Figure 8.
Copyright © 2004–2007, Texas Instruments Incorporated
Submit Documentation Feedback
5
Product Folder Links: LM5000
LM5000
SNVS176D –MAY 2004–REVISED MARCH 2007
www.ti.com
Typical Performance Characteristics (continued)
Current Limit vs Temperature
Current Limit vs VIN
2
1.95
1.9
2
1.95
1.9
1.85
1.8
1.85
1.8
1.75
1.7
1.75
1.7
1.65
1.6
1.65
1.6
1.55
1.5
1.55
1.5
120
100
40 60
-40
0
20
80
-20
10 15
0
5
20 25
VIN (V)
35 40
30
TEMPERATURE (oC)
Figure 9.
Figure 10.
fSW vs. VIN @ FS = Low (-3)
fSW vs. VIN @ FS = OPEN (-3)
770
750
730
710
690
670
650
630
315
310
305
300
295
290
285
10 15
10 15
0
5
20 25
VIN (V)
35 40
0
5
20 25
VIN (V)
35 40
30
30
Figure 11.
Figure 12.
fSW vs. Temperature @ FS = Low (-3)
fSW vs. Temperature @ FS = OPEN (-3)
770
330
320
310
300
290
280
270
750
730
710
690
670
650
630
-40 -20
0
20 40
80
60
100
120
-40 -20
0
20 40 60 80 100 120
TEMPERATURE (oC)
TEMPERATURE (oC)
Figure 13.
Figure 14.
6
Submit Documentation Feedback
Copyright © 2004–2007, Texas Instruments Incorporated
Product Folder Links: LM5000
LM5000
www.ti.com
SNVS176D –MAY 2004–REVISED MARCH 2007
Typical Performance Characteristics (continued)
fSW vs. Temperature @ FS = Low (-6)
fSW vs. Temperature @ FS = OPEN (-6)
1.38
640
620
600
580
560
540
520
1.34
1.30
1.26
1.22
1.18
1.14
-40 -20
0
20 40 60 80 100 120
-40 -20
0
20 40 60 80 100 120
TEMPERATURE (oC)
TEMPERATURE (oC)
Figure 15.
Figure 16.
Error Amp. Transconductance vs Temp.
600
BYP Pin Voltage vs VIN
8
7
6
5
4
3
2
1
0
125oC
550
500
450
400
350
300
250
-40oC
25oC
-40 -20
0
20 40
80
60
100
120
10 15
0
5
20 25
VIN (V)
35 40
30
TEMPERATURE (oC)
Figure 17.
Figure 18.
Copyright © 2004–2007, Texas Instruments Incorporated
Submit Documentation Feedback
7
Product Folder Links: LM5000
LM5000
SNVS176D –MAY 2004–REVISED MARCH 2007
www.ti.com
Typical Application Diagrams
Figure 19. 300 kHz operation, 48V output
Figure 20. 700 kHz operation, 48V output
8
Submit Documentation Feedback
Copyright © 2004–2007, Texas Instruments Incorporated
Product Folder Links: LM5000
LM5000
www.ti.com
SNVS176D –MAY 2004–REVISED MARCH 2007
Block Diagram
Copyright © 2004–2007, Texas Instruments Incorporated
Submit Documentation Feedback
9
Product Folder Links: LM5000
LM5000
SNVS176D –MAY 2004–REVISED MARCH 2007
www.ti.com
BOOST REGULATOR OPERATION
The LM5000 utilizes a PWM control scheme to regulate the output voltage over all load conditions. The operation
can best be understood referring to the block diagram and Figure 21. At the start of each cycle, the oscillator
sets the driver logic and turns on the NMOS power device conducting current through the inductor, cycle 1 of
Figure 21 (a). During this cycle, the voltage at the COMP pin controls the peak inductor current. The COMP
voltage will increase with larger loads and decrease with smaller. This voltage is compared with the summation
of the SW volatge and the ramp compensation.The ramp compensation is used in PWM architectures to
eliminate the sub-harmonic oscillations that occur during duty cycles greater than 50%. Once the summation of
the ramp compensation and switch voltage equals the COMP voltage, the PWM comparator resets the driver
logic turning off the NMOS power device. The inductor current then flows through the output diode to the load
and output capacitor, cycle 2 of Figure 21 (b). The NMOS power device is then set by the oscillator at the end of
the period and current flows through the inductor once again.
The LM5000 has dedicated protection circuitry running during the normal operation to protect the IC. The
Thermal Shutdown circuitry turns off the NMOS power device when the die temperature reaches excessive
levels. The UVP comparator protects the NMOS power device during supply power startup and shutdown to
prevent operation at voltages less than the minimum input voltage. The OVP comparator is used to prevent the
output voltage from rising at no loads allowing full PWM operation over all load conditions. The LM5000 also
features a shutdown mode. An external capacitor sets the softstart time by limiting the error amp output range,
as the capacitor charges up via an internal 10µA current source.
The LM5000 is available in two operating frequency ranges. The LM5000-3 is pin selectable for either 300kHz
(FS Grounded) or 700kHz (FS Open). The LM5000-6 is pin selectable for either 600kHz (FS Grounded) or
1.3MHz (FS Open)
Operation
Figure 21. Simplified Boost Converter Diagram
(a) First Cycle of Operation (b) Second Cycle Of Operation
CONTINUOUS CONDUCTION MODE
The LM5000 is a current-mode, PWM regulator. When used as a boost regulator the input voltage is stepped up
to a higher output voltage. In continuous conduction mode (when the inductor current never reaches zero at
steady state), the boost regulator operates in two cycles.
In the first cycle of operation, shown in Figure 21 (a), the transistor is closed and the diode is reverse biased.
Energy is collected in the inductor and the load current is supplied by COUT
.
10
Submit Documentation Feedback
Copyright © 2004–2007, Texas Instruments Incorporated
Product Folder Links: LM5000
LM5000
www.ti.com
SNVS176D –MAY 2004–REVISED MARCH 2007
The second cycle is shown in Figure 21 (b). During this cycle, the transistor is open and the diode is forward
biased. The energy stored in the inductor is transferred to the load and output capacitor.
The ratio of these two cycles determines the output voltage. The output voltage is defined approximately as:
VIN
VIN
VOUT
=
, D' = (1-D) =
VOUT
1-D
where
•
•
D is the duty cycle of the switch
D and D′ will be required for design calculations
(1)
SETTING THE OUTPUT VOLTAGE
The output voltage is set using the feedback pin and a resistor divider connected to the output as shown in
Figure 19. The feedback pin is always at 1.259V, so the ratio of the feedback resistors sets the output voltage.
VOUT - 1.259
W
RFB1 = RFB2
x
1.259
(2)
INTRODUCTION TO COMPENSATION
Figure 22. (a) Inductor current. (b) Diode current.
The LM5000 is a current mode PWM regulator. The signal flow of this control scheme has two feedback loops,
one that senses switch current and one that senses output voltage.
Copyright © 2004–2007, Texas Instruments Incorporated
Submit Documentation Feedback
11
Product Folder Links: LM5000
LM5000
SNVS176D –MAY 2004–REVISED MARCH 2007
www.ti.com
To keep a current programmed control converter stable above duty cycles of 50%, the inductor must meet
certain criteria. The inductor, along with input and output voltage, will determine the slope of the current through
the inductor (see Figure 22 (a)). If the slope of the inductor current is too great, the circuit will be unstable above
duty cycles of 50%.
The LM5000 provides a compensation pin (COMP) to customize the voltage loop feedback. It is recommended
that a series combination of RC and CC be used for the compensation network, as shown in Figure 19. The
series combination of RC and CC introduces pole-zero pair according to the following equations:
1
fZC
=
Hz
2pRCCC
(3)
1
fPC
=
Hz
2p(RC + RO)CC
where
•
RO is the output impedance of the error amplifier, 850kΩ
(4)
For most applications, performance can be optimized by choosing values within the range 5kΩ ≤ RC ≤ 20kΩ and
680pF ≤ CC ≤ 4.7nF.
COMPENSATION
This section will present a general design procedure to help insure a stable and operational circuit. The designs
in this datasheet are optimized for particular requirements. If different conversions are required, some of the
components may need to be changed to ensure stability. Below is a set of general guidelines in designing a
stable circuit for continuous conduction operation (loads greater than 100mA), in most all cases this will provide
for stability during discontinuous operation as well. The power components and their effects will be determined
first, then the compensation components will be chosen to produce stability.
INDUCTOR SELECTION
To ensure stability at duty cycles above 50%, the inductor must have some minimum value determined by the
minimum input voltage and the maximum output voltage. This equation is:
2
D
-1
( D')
VINRDSON
(in H)
L >
D
0.144 fs
+1
( D')
where
•
•
•
fs is the switching frequency
D is the duty cycle
RDSON is the ON resistance of the internal switch
(5)
(6)
This equation is only good for duty cycles greater than 50% (D>0.5).
VIND
(in Amps)
DiL =
2Lfs
The inductor ripple current is important for a few reasons. One reason is because the peak switch current will be
the average inductor current (input current) plus ΔiL. Care must be taken to make sure that the switch will not
reach its current limit during normal operation. The inductor must also be sized accordingly. It should have a
saturation current rating higher than the peak inductor current expected. The output voltage ripple is also affected
by the total ripple current.
DC GAIN AND OPEN-LOOP GAIN
Since the control stage of the converter forms a complete feedback loop with the power components, it forms a
closed-loop system that must be stabilized to avoid positive feedback and instability. A value for open-loop DC
gain will be required, from which you can calculate, or place, poles and zeros to determine the crossover
frequency and the phase margin. A high phase margin (greater than 45°) is desired for the best stability and
transient response. For the purpose of stabilizing the LM5000, choosing a crossover point well below where the
right half plane zero is located will ensure sufficient phase margin. A discussion of the right half plane zero and
checking the crossover using the DC gain will follow.
12
Submit Documentation Feedback
Copyright © 2004–2007, Texas Instruments Incorporated
Product Folder Links: LM5000
LM5000
www.ti.com
SNVS176D –MAY 2004–REVISED MARCH 2007
OUTPUT CAPACITOR SELECTION
The choice of output capacitors is somewhat more arbitrary. It is recommended that low ESR (Equivalent Series
Resistance, denoted RESR) capacitors be used such as ceramic, polymer electrolytic, or low ESR tantalum.
Higher ESR capacitors may be used but will require more compensation which will be explained later on in the
section. The ESR is also important because it determines the output voltage ripple according to the approximate
equation:
ΔVOUT ≊ 2ΔiLRESR (in Volts)
(7)
After choosing the output capacitor you can determine a pole-zero pair introduced into the control loop by the
following equations:
1
(in Hz)
fP1
=
2p(RESR + RL)COUT
(8)
1
(in Hz)
fZ1
=
2pRESRCOUT
where
•
RL is the minimum load resistance corresponding to the maximum load current
(9)
The zero created by the ESR of the output capacitor is generally very high frequency if the ESR is small. If low
ESR capacitors are used it can be neglected. If higher ESR capacitors are used see the HIGH OUTPUT
CAPACITOR ESR COMPENSATION section.
RIGHT HALF PLANE ZERO
A current mode control boost regulator has an inherent right half plane zero (RHP zero). This zero has the effect
of a zero in the gain plot, causing an imposed +20dB/decade on the rolloff, but has the effect of a pole in the
phase, subtracting another 90° in the phase plot. This can cause undesirable effects if the control loop is
influenced by this zero. To ensure the RHP zero does not cause instability issues, the control loop should be
designed to have a bandwidth of ½ the frequency of the RHP zero or less. This zero occurs at a frequency of:
VOUT(D')2
(in Hz)
RHPzero =
2pILOADL
where
•
ILOAD is the maximum load current
(10)
SELECTING THE COMPENSATION COMPONENTS
The first step in selecting the compensation components RC and CC is to set a dominant low frequency pole in
the control loop. Simply choose values for RC and CC within the ranges given in the INTRODUCTION TO
COMPENSATION section to set this pole in the area of 10Hz to 100Hz. The frequency of the pole created is
determined by the equation:
1
(in Hz)
fPC
=
2p(RC + RO)CC
where
•
RO is the output impedance of the error amplifier, 850kΩ
(11)
Since RC is generally much less than RO, it does not have much effect on the above equation and can be
neglected until a value is chosen to set the zero fZC. fZC is created to cancel out the pole created by the output
capacitor, fP1. The output capacitor pole will shift with different load currents as shown by the equation, so setting
the zero is not exact. Determine the range of fP1 over the expected loads and then set the zero fZC to a point
approximately in the middle. The frequency of this zero is determined by:
1
(in Hz)
fZC
=
2pCCRC
(12)
Copyright © 2004–2007, Texas Instruments Incorporated
Submit Documentation Feedback
13
Product Folder Links: LM5000
LM5000
SNVS176D –MAY 2004–REVISED MARCH 2007
www.ti.com
Now RC can be chosen with the selected value for CC. Check to make sure that the pole fPC is still in the 10Hz to
100Hz range, change each value slightly if needed to ensure both component values are in the recommended
range. After checking the design at the end of this section, these values can be changed a little more to optimize
performance if desired. This is best done in the lab on a bench, checking the load step response with different
values until the ringing and overshoot on the output voltage at the edge of the load steps is minimal. This should
produce a stable, high performance circuit. For improved transient response, higher values of RC (within the
range of values) should be chosen. This will improve the overall bandwidth which makes the regulator respond
more quickly to transients. If more detail is required, or the most optimal performance is desired, refer to a more
in depth discussion of compensating current mode DC/DC switching regulators.
HIGH OUTPUT CAPACITOR ESR COMPENSATION
When using an output capacitor with a high ESR value, or just to improve the overall phase margin of the control
loop, another pole may be introduced to cancel the zero created by the ESR. This is accomplished by adding
another capacitor, CC2, directly from the compensation pin VC to ground, in parallel with the series combination of
RC and CC. The pole should be placed at the same frequency as fZ1, the ESR zero. The equation for this pole
follows:
1
(in Hz)
fPC2
=
2pCC2(RC //RO)
(13)
To ensure this equation is valid, and that CC2 can be used without negatively impacting the effects of RC and CC,
fPC2 must be greater than 10fPC
.
CHECKING THE DESIGN
The final step is to check the design. This is to ensure a bandwidth of ½ or less of the frequency of the RHP
zero. This is done by calculating the open-loop DC gain, ADC. After this value is known, you can calculate the
crossover visually by placing a −20dB/decade slope at each pole, and a +20dB/decade slope for each zero. The
point at which the gain plot crosses unity gain, or 0dB, is the crossover frequency. If the crossover frequency is
at less than ½ the RHP zero, the phase margin should be high enough for stability. The phase margin can also
be improved some by adding CC2 as discussed earlier in the section. The equation for ADC is given below with
additional equations required for the calculation:
RFB2
gmROD'
{[(wcLeff)// RL]//RL} (in dB)
ADC(DB) = 20log10
(
)
RDSON
RFB1 + RFB2
(14)
(15)
2fs
(in rad/s)
@
wc
nD'
L
Leff =
(D')2
(16)
2mc
m1
(no unit)
n = 1+
(17)
(18)
mc ≊ 0.072fs (in A/s)
VINRDSON
(in V/s)
@
m1
where
•
•
•
RL is the minimum load resistance
VIN is the maximum input voltage
RDSON is the value chosen from the graph "RDSON vs. VIN " in the Typical Performance Characteristics
section
(19)
SWITCH VOLTAGE LIMITS
In a flyback regulator, the maximum steady-state voltage appearing at the switch, when it is off, is set by the
transformer turns ratio, N, the output voltage, VOUT, and the maximum input voltage, VIN (Max):
VSW(OFF) = VIN (Max) + (VOUT +VF)/N
where
•
VF is the forward biased voltage of the output diode, and is typically 0.5V for Schottky diodes and 0.8V for
ultra-fast recovery diodes
(20)
14
Submit Documentation Feedback
Copyright © 2004–2007, Texas Instruments Incorporated
Product Folder Links: LM5000
LM5000
www.ti.com
SNVS176D –MAY 2004–REVISED MARCH 2007
In certain circuits, there exists a voltage spike, VLL, superimposed on top of the steady-state voltage . Usually,
this voltage spike is caused by the transformer leakage inductance and/or the output rectifier recovery time. To
“clamp” the voltage at the switch from exceeding its maximum value, a transient suppressor in series with a
diode is inserted across the transformer primary.
If poor circuit layout techniques are used, negative voltage transients may appear on the Switch pin. Applying a
negative voltage (with respect to the IC's ground) to any monolithic IC pin causes erratic and unpredictable
operation of that IC. This holds true for the LM5000 IC as well. When used in a flyback regulator, the voltage at
the Switch pin can go negative when the switch turns on. The “ringing” voltage at the switch pin is caused by the
output diode capacitance and the transformer leakage inductance forming a resonant circuit at the
secondary(ies). The resonant circuit generates the “ringing” voltage, which gets reflected back through the
transformer to the switch pin. There are two common methods to avoid this problem. One is to add an RC
snubber around the output rectifier(s). The values of the resistor and the capacitor must be chosen so that the
voltage at the Switch pin does not drop below −0.4V. The resistor may range in value between 10Ω and 1 kΩ,
and the capacitor will vary from 0.001 μF to 0.1 μF. Adding a snubber will (slightly) reduce the efficiency of the
overall circuit.
The other method to reduce or eliminate the “ringing” is to insert a Schottky diode clamp between the SW pin
and the PGND pin. The reverse voltage rating of the diode must be greater than the switch off voltage.
OUTPUT VOLTAGE LIMITATIONS
The maximum output voltage of a boost regulator is the maximum switch voltage minus a diode drop. In a
flyback regulator, the maximum output voltage is determined by the turns ratio, N, and the duty cycle, D, by the
equation:
VOUT ≈ N × VIN × D/(1 − D)
(21)
The duty cycle of a flyback regulator is determined by the following equation:
(22)
Theoretically, the maximum output voltage can be as large as desired—just keep increasing the turns ratio of the
transformer. However, there exists some physical limitations that prevent the turns ratio, and thus the output
voltage, from increasing to infinity. The physical limitations are capacitances and inductances in the LM5000
switch, the output diode(s), and the transformer—such as reverse recovery time of the output diode (mentioned
above).
INPUT LINE CONDITIONING
A small, low-pass RC filter should be used at the input pin of the LM5000 if the input voltage has an unusually
large amount of transient noise. Additionally, the RC filter can reduce the dissipation within the device when the
input voltage is high.
Flyback Regulator Operation
The LM5000 is ideally suited for use in the flyback regulator topology. The flyback regulator can produce a single
output voltage, or multiple output voltages.
The operation of a flyback regulator is as follows: When the switch is on, current flows through the primary
winding of the transformer, T1, storing energy in the magnetic field of the transformer. Note that the primary and
secondary windings are out of phase, so no current flows through the secondary when current flows through the
primary. When the switch turns off, the magnetic field collapses, reversing the voltage polarity of the primary and
secondary windings. Now rectifier D5 is forward biased and current flows through it, releasing the energy stored
in the transformer. This produces voltage at the output.
The output voltage is controlled by modulating the peak switch current. This is done by feeding back a portion of
the output voltage to the error amp, which amplifies the difference between the feedback voltage and a 1.259V
reference. The error amp output voltage is compared to a ramp voltage proportional to the switch current (i.e.,
inductor current during the switch on time). The comparator terminates the switch on time when the two voltages
are equal, thereby controlling the peak switch current to maintain a constant output voltage.
Copyright © 2004–2007, Texas Instruments Incorporated
Submit Documentation Feedback
15
Product Folder Links: LM5000
LM5000
SNVS176D –MAY 2004–REVISED MARCH 2007
www.ti.com
Figure 23. LM5000 Flyback Converter
16
Submit Documentation Feedback
Copyright © 2004–2007, Texas Instruments Incorporated
Product Folder Links: LM5000
LM5000
www.ti.com
ITEM
SNVS176D –MAY 2004–REVISED MARCH 2007
PART NUMBER
C4532X7R2A105MT
DESCRIPTION
Capacitor, CER, TDK
VALUE
1µ, 100V
C
C
C
C
C
C
C
C
C
C
C
D
D
D
D
D
T
1
2
3
4
5
6
7
8
9
10
11
1
2
3
4
5
1
1
2
3
4
5
6
7
1
1
C4532X7R2A105MT
C1206C224K5RAC
C1206C104K5RAC
C1206C104K5RAC
C1206C101K1GAC
C1206C104K5RAC
C4532X7S0G686M
C4532X7S0G686M
C1206C221K1GAC
C1206C102K5RAC
BZX84C10-NSA
CMZ5930B-NSA
CMPD914-NSA
Capacitor, CER, TDK
Capacitor, CER, KEMET
Capacitor, CER, KEMET
Capacitor, CER, KEMET
Capacitor, CER, KEMET
Capacitor, CER, KEMET
Capacitor, CER, TDK
Capacitor, CER, TDK
Capacitor, CER, KEMET
Capacitor, CER, KEMET
Central, 10V Zener, SOT-23
Central, 16V Zener, SMA
Central, Switching, SOT-23
Central, Switching, SOT-23
Central, Schottky, SMC
Coilcraft, Transformer
Resistor
1µ, 100V
0.22µ, 50V
0.1µ, 50V
0.1µ, 50V
100p, 100V
0.1µ, 50V
68µ, 4V
68µ, 4V
220p, 100V
1000p, 500V
CMPD914-NSA
CMSH3-40L-NSA
A0009-A
R
R
R
R
R
R
R
Q
U
CRCW12064992F
CRCW12061001F
CRCW12061002F
CRCW12066191F
CRCW120610R0F
CRCW12062003F
CRCW12061002F
CXT5551-NSA
49.9K
1K
Resistor
Resistor
10K
6.19K
10
Resistor
Resistor
Resistor
200K
10K
Resistor
Central, NPN, 180V
Regulator, TI
LM5000-3
Copyright © 2004–2007, Texas Instruments Incorporated
Submit Documentation Feedback
17
Product Folder Links: LM5000
PACKAGE OPTION ADDENDUM
www.ti.com
1-Nov-2013
PACKAGING INFORMATION
Orderable Device
LM5000-3MTC
Status Package Type Package Pins Package
Eco Plan
Lead/Ball Finish
MSL Peak Temp
Op Temp (°C)
-40 to 125
-40 to 125
-40 to 125
-40 to 125
-40 to 125
-40 to 125
-40 to 125
-40 to 125
Device Marking
Samples
Drawing
Qty
(1)
(2)
(6)
(3)
(4/5)
NRND
TSSOP
TSSOP
TSSOP
TSSOP
WSON
WSON
WSON
WSON
PW
16
16
16
16
16
16
16
16
92
TBD
Call TI
CU SN
Call TI
CU SN
CU SN
CU SN
CU SN
CU SN
Call TI
LM5000
3MTC
LM5000-3MTC/NOPB
LM5000-3MTCX
ACTIVE
NRND
PW
PW
92
Green (RoHS
& no Sb/Br)
Level-1-260C-UNLIM
Call TI
LM5000
3MTC
2500
2500
1000
1000
4500
4500
TBD
LM5000
3MTC
LM5000-3MTCX/NOPB
LM5000SD-3/NOPB
LM5000SD-6/NOPB
LM5000SDX-3/NOPB
LM5000SDX-6/NOPB
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
PW
Green (RoHS
& no Sb/Br)
Level-1-260C-UNLIM
Level-1-260C-UNLIM
Level-1-260C-UNLIM
Level-1-260C-UNLIM
Level-1-260C-UNLIM
LM5000
3MTC
NHQ
NHQ
NHQ
NHQ
Green (RoHS
& no Sb/Br)
5000-3
5000-6
5000-3
5000-6
Green (RoHS
& no Sb/Br)
Green (RoHS
& no Sb/Br)
Green (RoHS
& no Sb/Br)
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
Addendum-Page 1
PACKAGE OPTION ADDENDUM
www.ti.com
1-Nov-2013
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6) Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
6-Nov-2015
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
B0
K0
P1
W
Pin1
Diameter Width (mm) (mm) (mm) (mm) (mm) Quadrant
(mm) W1 (mm)
LM5000-3MTCX
TSSOP
PW
PW
16
16
16
16
16
16
2500
2500
1000
1000
4500
4500
330.0
330.0
178.0
178.0
330.0
330.0
12.4
12.4
12.4
12.4
12.4
12.4
6.95
6.95
5.3
5.6
5.6
5.3
5.3
5.3
5.3
1.6
1.6
1.3
1.3
1.3
1.3
8.0
8.0
8.0
8.0
8.0
8.0
12.0
12.0
12.0
12.0
12.0
12.0
Q1
Q1
Q1
Q1
Q1
Q1
LM5000-3MTCX/NOPB TSSOP
LM5000SD-3/NOPB
LM5000SD-6/NOPB
LM5000SDX-3/NOPB
LM5000SDX-6/NOPB
WSON
WSON
WSON
WSON
NHQ
NHQ
NHQ
NHQ
5.3
5.3
5.3
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
6-Nov-2015
*All dimensions are nominal
Device
Package Type Package Drawing Pins
SPQ
Length (mm) Width (mm) Height (mm)
LM5000-3MTCX
LM5000-3MTCX/NOPB
LM5000SD-3/NOPB
LM5000SD-6/NOPB
LM5000SDX-3/NOPB
LM5000SDX-6/NOPB
TSSOP
TSSOP
WSON
WSON
WSON
WSON
PW
PW
16
16
16
16
16
16
2500
2500
1000
1000
4500
4500
367.0
367.0
210.0
210.0
367.0
367.0
367.0
367.0
185.0
185.0
367.0
367.0
35.0
35.0
35.0
35.0
35.0
35.0
NHQ
NHQ
NHQ
NHQ
Pack Materials-Page 2
MECHANICAL DATA
NHQ0016A
SDA16A (Rev A)
www.ti.com
IMPORTANT NOTICE
Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, enhancements, improvements and other
changes to its semiconductor products and services per JESD46, latest issue, and to discontinue any product or service per JESD48, latest
issue. Buyers should obtain the latest relevant information before placing orders and should verify that such information is current and
complete. All semiconductor products (also referred to herein as “components”) are sold subject to TI’s terms and conditions of sale
supplied at the time of order acknowledgment.
TI warrants performance of its components to the specifications applicable at the time of sale, in accordance with the warranty in TI’s terms
and conditions of sale of semiconductor products. Testing and other quality control techniques are used to the extent TI deems necessary
to support this warranty. Except where mandated by applicable law, testing of all parameters of each component is not necessarily
performed.
TI assumes no liability for applications assistance or the design of Buyers’ products. Buyers are responsible for their products and
applications using TI components. To minimize the risks associated with Buyers’ products and applications, Buyers should provide
adequate design and operating safeguards.
TI does not warrant or represent that any license, either express or implied, is granted under any patent right, copyright, mask work right, or
other intellectual property right relating to any combination, machine, or process in which TI components or services are used. Information
published by TI regarding third-party products or services does not constitute a license to use such products or services or a warranty or
endorsement thereof. Use of such information may require a license from a third party under the patents or other intellectual property of the
third party, or a license from TI under the patents or other intellectual property of TI.
Reproduction of significant portions of TI information in TI data books or data sheets is permissible only if reproduction is without alteration
and is accompanied by all associated warranties, conditions, limitations, and notices. TI is not responsible or liable for such altered
documentation. Information of third parties may be subject to additional restrictions.
Resale of TI components or services with statements different from or beyond the parameters stated by TI for that component or service
voids all express and any implied warranties for the associated TI component or service and is an unfair and deceptive business practice.
TI is not responsible or liable for any such statements.
Buyer acknowledges and agrees that it is solely responsible for compliance with all legal, regulatory and safety-related requirements
concerning its products, and any use of TI components in its applications, notwithstanding any applications-related information or support
that may be provided by TI. Buyer represents and agrees that it has all the necessary expertise to create and implement safeguards which
anticipate dangerous consequences of failures, monitor failures and their consequences, lessen the likelihood of failures that might cause
harm and take appropriate remedial actions. Buyer will fully indemnify TI and its representatives against any damages arising out of the use
of any TI components in safety-critical applications.
In some cases, TI components may be promoted specifically to facilitate safety-related applications. With such components, TI’s goal is to
help enable customers to design and create their own end-product solutions that meet applicable functional safety standards and
requirements. Nonetheless, such components are subject to these terms.
No TI components are authorized for use in FDA Class III (or similar life-critical medical equipment) unless authorized officers of the parties
have executed a special agreement specifically governing such use.
Only those TI components which TI has specifically designated as military grade or “enhanced plastic” are designed and intended for use in
military/aerospace applications or environments. Buyer acknowledges and agrees that any military or aerospace use of TI components
which have not been so designated is solely at the Buyer's risk, and that Buyer is solely responsible for compliance with all legal and
regulatory requirements in connection with such use.
TI has specifically designated certain components as meeting ISO/TS16949 requirements, mainly for automotive use. In any case of use of
non-designated products, TI will not be responsible for any failure to meet ISO/TS16949.
Products
Applications
Audio
www.ti.com/audio
amplifier.ti.com
dataconverter.ti.com
www.dlp.com
Automotive and Transportation www.ti.com/automotive
Communications and Telecom www.ti.com/communications
Amplifiers
Data Converters
DLP® Products
DSP
Computers and Peripherals
Consumer Electronics
Energy and Lighting
Industrial
www.ti.com/computers
www.ti.com/consumer-apps
www.ti.com/energy
dsp.ti.com
Clocks and Timers
Interface
www.ti.com/clocks
interface.ti.com
logic.ti.com
www.ti.com/industrial
www.ti.com/medical
Medical
Logic
Security
www.ti.com/security
Power Mgmt
Microcontrollers
RFID
power.ti.com
Space, Avionics and Defense
Video and Imaging
www.ti.com/space-avionics-defense
www.ti.com/video
microcontroller.ti.com
www.ti-rfid.com
www.ti.com/omap
OMAP Applications Processors
Wireless Connectivity
TI E2E Community
e2e.ti.com
www.ti.com/wirelessconnectivity
Mailing Address: Texas Instruments, Post Office Box 655303, Dallas, Texas 75265
Copyright © 2015, Texas Instruments Incorporated
相关型号：
SI9130DB
5- and 3.3-V Step-Down Synchronous ConvertersWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9135LG-T1
SMBus Multi-Output Power-Supply ControllerWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9135LG-T1-E3
SMBus Multi-Output Power-Supply ControllerWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9135_11
SMBus Multi-Output Power-Supply ControllerWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9136_11
Multi-Output Power-Supply ControllerWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9130CG-T1-E3
Pin-Programmable Dual Controller - Portable PCsWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9130LG-T1-E3
Pin-Programmable Dual Controller - Portable PCsWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9130_11
Pin-Programmable Dual Controller - Portable PCsWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9137
Multi-Output, Sequence Selectable Power-Supply Controller for Mobile ApplicationsWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9137DB
Multi-Output, Sequence Selectable Power-Supply Controller for Mobile ApplicationsWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9137LG
Multi-Output, Sequence Selectable Power-Supply Controller for Mobile ApplicationsWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9122E
500-kHz Half-Bridge DC/DC Controller with Integrated Secondary Synchronous Rectification DriversWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
©2020 ICPDF网 联系我们和版权申明