LMK04208NKDR [TI]
具有 6 个可编程输出的超低噪声时钟抖动消除器 | NKD | 64 | -40 to 85;
| 型号： | LMK04208NKDR |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | 具有 6 个可编程输出的超低噪声时钟抖动消除器 | NKD | 64 | -40 to 85 时钟 |
| 文件： | 总137页 (文件大小：2192K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
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LMK04208
ZHCSFH1 –SEPTEMBER 2016
LMK04208 具有双环 PLL 的低噪声时钟抖动消除器
1 特性
3 说明
1
•
超低的均方根值 (RMS) 抖动性能
LMK04208 器件是一款高性能时钟调节器，具备出色
的时钟抖动消除、生成和分配 等高级功能， 能够充分
满足新一代系统要求。双环 PLLatinum™架构利用低
噪声 VCXO 模块能够实现 111fs RMS 抖动（12kHz
至 20MHz）或采用低成本外部晶振及变容二极管实现
低于 200fs 的 RMS 抖动（12kHz 至 20MHz）。
–
–
111fs，RMS 抖动（12kHz 至 20MHz）
123fs，RMS 抖动（100Hz 至 20MHz）
•
•
双环路 PLLatinum™锁相环 (PLL) 架构
PLL1
–
–
集成低噪声晶体振荡器电路
输入时钟丢失时采用保持模式
双环架构由两个高性能锁相环 (PLL)、一个低噪声晶体
振荡器电路以及一个高性能压控振荡器 (VCO) 构成。
第一个 PLL (PLL1) 具有低噪声抖动消除器功能，而第
二个 PLL (PLL2) 执行时钟生成。PLL1 可配置为与外
部 VCXO 模块配合使用，或与具有外部可调晶体和变
容二极管的集成式晶体振荡器配合使用。当应用于很窄
的环路带宽时，PLL1 使用 VCXO 模块或可调晶体的
优异近端相位噪声（偏移低于 50kHz）清理输入时
钟。PLL1 的输出将用作 PLL2 的清理输入参考，以锁
定集成式 VCO。可对 PLL2 的环路带宽进行优化以清
理远端相位噪声（偏移高于 50 kHz），集成式 VCO
优于 VCXO 模块或 PLL1 中使用的可调晶体。
–
自动或手动触发/恢复
•
PLL2
–
–
–
–
标准化锁相环 (PLL) 噪底为 –227dBc/Hz
相位检测器速率最高可达 155MHz
OSCin 倍频器
集成低噪声压控振荡器 (VCO)或外部 VCO 模式
•
两个具有 LOS 的冗余输入时钟
自动和手动切换模式
–
•
•
50% 占空比输出分配，1 至 1045（偶数和奇数）
6 路低电压正射极耦合逻辑 (LVPECL)、低压差分
信令 (LVDS) 或低电压互补金属氧化物半导体
(LVCMOS) 可编程输出
器件信息(1)
•
•
•
数字延迟：固定或可动态调节
器件型号
VCO 频率
时钟输入
模拟延迟控制（步长为 25ps）
LMK04208
2750MHz 至 3072MHz
2
7 路差分输出；最高可达 14 路的单端输出
(1) 要了解所有可用封装，请见数据表末尾的可订购产品附录。
–
多达 6 个 VCXO/晶振缓冲输出
•
•
•
•
•
•
•
时钟速率高达 1536MHz
0 延迟模式
简化电路原理图
Crystal or
VCXO
LMX2582
OSCout
加电时 3 个缺省时钟输出
多模式：双 PLL、单 PLL 和时钟分配
工业温度范围：-40°C 至 +85°C
3.15V 至 3.45V 工作电压
PLL+VCO
Serializer/
Deserializer
Recovered
—dirty“ clocks
or clean clocks
CLKout0
CLKout1
CLKout2
CLKout3
CLKout4
CLKin0
CLKin1
LMK04208
Precision Clock
Conditioner
Backup
Reference
Clock
64 引脚超薄四方扁平无引线 (WQFN) 封装 (9.0mm
× 9.0mm × 0.8mm)
CLKout5
ADC
DAC
Multiple —clean“ clocks at different
frequencies
FPGA
CPLD
2 应用
•
•
•
数据转换器计时
无线基础设施
网络、同步光纤网 (SONET) 或同步数字体系
(SDH)、数字用户线路接入复用器 (DSLAM)
•
•
医疗、视频、军事和航天领域
测试和测量
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
English Data Sheet: SNAS684
LMK04208
ZHCSFH1 –SEPTEMBER 2016
www.ti.com.cn
目录
8.6 Register Maps......................................................... 52
Application and Implementation ........................ 96
9.1 Application Information............................................ 96
9.2 Typical Applications .............................................. 113
9.3 System Examples ................................................. 121
9.4 Do's and Don'ts..................................................... 123
1
2
3
4
5
6
特性.......................................................................... 1
9
应用.......................................................................... 1
说明.......................................................................... 1
修订历史记录 ........................................................... 2
Pin Configuration and Functions......................... 3
Specifications......................................................... 5
6.1 Absolute Maximum Ratings ...................................... 5
6.2 ESD Ratings.............................................................. 5
6.3 Recommended Operating Conditions....................... 5
6.4 Thermal Information.................................................. 6
6.5 Electrical Characteristics........................................... 7
6.6 Timing Requirements.............................................. 13
6.7 Typical Characteristics ........................................... 15
Parameter Measurement Information ................ 16
7.1 Charge Pump Current Specification Definitions...... 16
7.2 Differential Voltage Measurement Terminology...... 17
Detailed Description ............................................ 18
8.1 Overview ................................................................. 18
8.2 Functional Block Diagram ....................................... 22
8.3 Feature Description................................................. 23
8.4 Device Functional Modes........................................ 43
8.5 Programming........................................................... 48
10 Power Supply Recommendations ................... 124
10.1 Pin Connection Recommendations..................... 124
10.2 Current Consumption and Power Dissipation
Calculations............................................................ 126
11 Layout................................................................. 128
11.1 Layout Guidelines ............................................... 128
11.2 Layout Example .................................................. 129
12 器件和文档支持 ................................................... 130
12.1 器件支持.............................................................. 130
12.2 文档支持.............................................................. 130
12.3 接收文档更新通知 ............................................... 130
12.4 社区资源.............................................................. 130
12.5 商标..................................................................... 130
12.6 静电放电警告....................................................... 130
12.7 Glossary.............................................................. 130
13 机械、封装和可订购信息..................................... 130
7
8
4 修订历史记录
日期
修订版本
注释
2016 年 9 月
*
最初发布。
2
Copyright © 2016, Texas Instruments Incorporated
LMK04208
www.ti.com.cn
ZHCSFH1 –SEPTEMBER 2016
5 Pin Configuration and Functions
NKD Package
64-Pin WQFN with Exposed Pad
Top View
64 63 62 61 60 59 58 57 56 55 54 53 52 51 50 49
NC
NC
CLKout3
1
2
48
47
46
45
44
43
42
41
40
39
38
37
36
35
34
33
Vcc10
CLKout0*
CLKout0
NC
DATAuWire
CLKuWire
LEuWire
Vcc9
3
4
5
SYNC
NC
6
CPout2
Vcc8
7
NC
8
Top Down View
NC
OSCout*
OSCout
Vcc7
9
Vcc1
10
11
12
13
14
15
16
LDObyp1
LDObyp2
CLKout1
CLKout1*
NC
OSCin*
OSCin
Vcc6
CPout1
Status_LD
DAP
NC
17 18 19 20 21 22 23 24 25 26 27 28 29 30 31 32
Pin Functions(1)
PIN
I/O
TYPE
DESCRIPTION
NO.
NAME
1, 2
3, 4
5
NC
–
O
–
–
No Connection. These pins must be left floating.
Clock output 0.
CLKout0*, CLKout0
Programmable
NC
–
No Connection. These pins must be left floating.
CLKout Synchronization input or programmable status pin.
No Connection. These pins must be left floating.
Power supply for VCO LDO.
6
SYNC
I/O
–
Programmable
7, 8, 9
10
NC
–
Vcc1
PWR
11
LDObyp1
LDObyp2
CLKout1, CLKout1*
NC
ANLG
ANLG
Programmable
–
LDO Bypass, bypassed to ground with 10-µF capacitor.
LDO Bypass, bypassed to ground with a 0.1-µF capacitor.
Clock output 1.
12
13, 14
15, 16
17
O
–
No Connection. These pins must be left floating.
Power supply for clock output 1.
Vcc2
PWR
(1) See Pin Connection Recommendations.
Copyright © 2016, Texas Instruments Incorporated
3
LMK04208
ZHCSFH1 –SEPTEMBER 2016
www.ti.com.cn
Pin Functions(1) (continued)
PIN
NO.
I/O
TYPE
DESCRIPTION
NAME
18
Vcc3
NC
PWR
Power supply for clock output 2.
19, 20
21, 22
23
–
–
No Connection. These pins must be left floating.
Clock output 2.
CLKout2*, CLKout2
GND
O
Programmable
PWR
Ground.
24
Vcc4
PWR
Power supply for digital.
CLKin1, CLKin1*
Reference Clock Input Port 1 for PLL1. AC or DC Coupled.
Feedback input for external clock feedback input (0-delay
mode). AC or DC Coupled.
FBCLKin, FBCLKin*
Fin/Fin*
25, 26
I
ANLG
External VCO input (External VCO mode). AC or DC
Coupled.
Programmable status pin, default readback output.
Programmable to holdover mode indicator. Other options
available by programming.
27
Status_Holdover
I/O
I
Programmable
ANLG
Reference Clock Input Port 0 for PLL1.
AC or DC Coupled.
28, 29
CLKin0, CLKin0*
30
Vcc5
NC
PWR
–
Power supply for clock inputs.
31, 32
–
I/O
O
No Connection. These pins must be left floating.
Programmable status pin, default lock detect for PLL1 and
PLL2. Other options available by programming.
33
Status_LD
Programmable
34
35
CPout1
Vcc6
ANLG
PWR
Charge pump 1 output.
Power supply for PLL1, charge pump 1.
Feedback to PLL1, Reference input to PLL2.
AC Coupled.
36, 37
OSCin, OSCin*
I
ANLG
38
Vcc7
PWR
Power supply for OSCin, OSCout, and PLL2 circuitry.(2)
Buffered output of OSCin port.(2)
Power supply for PLL2, charge pump 2.
Charge pump 2 output.
39, 40
41
OSCout, OSCout*
Vcc8
O
O
Programmable
PWR
42
CPout2
ANLG
43
Vcc9
PWR
Power supply for PLL2.
44
LEuWire
CLKuWire
DATAuWire
Vcc10
I
I
I
CMOS
CMOS
CMOS
PWR
MICROWIRE Latch Enable Input.
MICROWIRE Clock Input.
45
46
MICROWIRE Data Input.
47
Power supply for clock output 3.
Clock output 3.
48, 49
50, 51
52
CLKout3, CLKout3*
NC
O
–
Programmable
–
No Connection. These pins must be left floating.
Power supply for clock output 4.
Clock output 4.
Vcc11
PWR
53, 54
55, 56
57
CLKout4, CLKout4*
NC
O
–
Programmable
–
No Connection. These pins must be left floating.
Power supply for clock output 5.
Clock output 5.
Vcc12
PWR
58, 59
60, 61
CLKout5, CLKout5*
NC
O
–
Programmable
–
No Connection. These pins must be left floating.
NC. Programmable status pin. Default is input for pin control
of PLL1 reference clock selection. CLKin0 LOS status and
other options available by programming.
62
63
Status_CLKin0
Status_CLKin1
I/O
I/O
Programmable
Programmable
Programmable status pin. Default is input for pin control of
PLL1 reference clock selection. CLKin1 LOS status and
other options available by programming.
64
Vcc13
DAP
PWR
GND
Power supply for clock output 0.
DAP
–
DIE ATTACH PAD, connect to GND.
(2) See Vcc5 (CLKin), Vcc7 (OSCin and OSCout) for information on configuring device for optimum performance.
4
Copyright © 2016, Texas Instruments Incorporated
LMK04208
www.ti.com.cn
ZHCSFH1 –SEPTEMBER 2016
6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted)(1)(2)(1)
MIN
–0.3
–0.3
MAX
3.6
UNIT
V
(3)
VCC
VIN
TL
Supply voltage
Input voltage
VCC + 0.3
260
V
Lead temperature (solder 4 seconds)
Junction temperature
°C
°C
TJ
150
Differential input current (CLKinX/X*,
OSCin/OSCin*, FBCLKin/FBCLKin*, Fin/Fin*)
IIN
± 5
mA
MSL
Tstg
Moisture Sensitivity Level
Storage temperature
3
-65
150
°C
(1) Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
(2) If Military/Aerospace specified devices are required, contact the Texas Instruments Sales Office/Distributors for availability and
specifications.
(3) Never to exceed 3.6 V.
6.2 ESD Ratings
VALUE
UNIT
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001(1)
±2000
V(ESD)
Electrostatic discharge
V
Charged-device model (CDM), per JEDEC specification JESD22-
C101(2)
±750
(1) JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process. Manufacturing with
less than 500-V HBM is possible with the necessary precautions. Pins listed as ±2000 V may actually have higher performance.
(2) JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process. Manufacturing with
less than 250-V CDM is possible with the necessary precautions. Pins listed as ±750 V may actually have higher performance.
6.3 Recommended Operating Conditions
MIN
NOM
MAX
125
85
UNIT
°C
TJ
Junction temperature
Ambient temperature
Supply voltage
TA
VCC = 3.3 V
–40
25
°C
VCC
3.15
3.3
3.45
V
Copyright © 2016, Texas Instruments Incorporated
5
LMK04208
ZHCSFH1 –SEPTEMBER 2016
www.ti.com.cn
6.4 Thermal Information
LMK04208
THERMAL METRIC(1)
NKD (WQFN)
UNIT
64 PINS
25.2
6.9
RθJA
Junction-to-ambient thermal resistance on 4-layer JEDEC PCB(2)(3)
Junction-to-case (top) thermal resistance(4)(5)
Junction-to-board thermal resistance(6)
Junction-to-top characterization parameter(7)
Junction-to-board characterization parameter(8)
Junction-to-case (bottom) thermal resistance(9)
°C/W
°C/W
°C/W
°C/W
°C/W
°C/W
RθJC(top)
RθJB
4.0
ψJT
0.1
ψJB
4.0
RθJC(bot)
0.8
(1) For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
(2) The junction-to-ambient thermal resistance under natural convection is obtained in a simulation on a JEDEC-standard, High-K board, as
specified in JESD51-7, in an environment described in JESD51-2a.
(3) Specification assumes 32 thermal vias connect the die attach pad to the embedded copper plane on the 4-layer JEDEC PCB. These
vias play a key role in improving the thermal performance of the WQFN. Note that the JEDEC PCB is a standard thermal measurement
PCB and does not represent best performance a PCB can achieve. TI recommends that the maximum number of vias be used in the
board layout. R θJA is unique for each PCB.
(4) The junction-to-case(top) thermal resistance is obtained by simulating a cold plate test on the package top. No specific JEDEC standard
test exists, but a close description can be found in the ANSI SEMI standard G30-88.
(5) Case is defined as the DAP (die attach pad)
(6) The junction-to-board thermal resistance is obtained by simulating an environment with a ring cold plate fixture to control the PCB
temperature, as described in JESD51-8.
(7) The junction-to-top characterization parameter, ψJT, estimates the junction temperature of a device in a real system and is extracted
from the simulation data for obtaining RθJA, using a procedure described in JESD51-2a (sections 6 and 7).
(8) The junction-to-board characterization parameter, ψJB, estimates the junction temperature of a device in a real system and is extracted
from the simulation data for obtaining RθJA, using a procedure described in JESD51-2a (sections 6 and 7).
(9) The junction-to-case(bottom) thermal resistance is obtained by simulating a cold plate test on the exposed (power) pad. No specific
JEDEC standard test exists, but a close description can be found in the ANSI SEMI standard G30-88.
6
Copyright © 2016, Texas Instruments Incorporated
LMK04208
www.ti.com.cn
ZHCSFH1 –SEPTEMBER 2016
6.5 Electrical Characteristics
3.15 V ≤ VCC ≤ 3.45 V, –40 °C ≤ TA ≤ 85°C. Typical values represent most likely parametric norms at VCC = 3.3 V, TA = 25°C,
at the Recommended Operating Conditions at the time of product characterization and are not specified.(1)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX UNIT
CURRENT CONSUMPTION
ICC_PD
Power down supply current
1
3
mA
mA
All clock delays disabled,
Supply current with all clocks (CLKoutX) CLKoutX_DIV = 1045,
ICC_CLKS
445
535
and OSCout enabled as LVDS.(2)
EN_SYNC=0
PLL1 and PLL2 locked.
CLKin0/0* and CLKin1/1* INPUT CLOCK SPECIFICATIONS
fCLKin
Clock input frequency(3)
Clock input slew rate(4)
0.001
0.15
0.25
0.5
500
MHz
V/ns
|V|
(1)
SLEWCLKin
VIDCLKin
VSSCLKin
VIDCLKin
VSSCLKin
20% to 80%
0.5
1.55
3.1
AC coupled
CLKinX_BUF_TYPE = 0 (Bipolar)
Clock input
Vpp
|V|
Differential input voltage (see (5) and
Figure 8)
0.25
0.5
1.55
3.1
AC coupled
CLKinX_BUF_TYPE = 1 (MOS)
Vpp
AC coupled to CLKinX; CLKinX* AC
coupled to Ground
CLKinX_BUF_TYPE = 0 (Bipolar)
0.25
0.25
2.4
2.4
Vpp
Vpp
mV
mV
mV
Clock input
VCLKin
Single-ended input voltage(4)
AC coupled to CLKinX; CLKinX* AC
coupled to Ground
CLKinX_BUF_TYPE = 1 (MOS)
DC offset voltage between
CLKin0/CLKin0*
CLKin0* - CLKin0
VCLKin0-offset
VCLKin1-offset
VCLKinX-offset
20
0
Each pin AC coupled
CLKin0_BUF_TYPE = 0 (Bipolar)
DC offset voltage between
CLKin1/CLKin1*
CLKin1* - CLKin1
DC offset voltage between
CLKinX/CLKinX*
Each pin AC coupled
CLKinX_BUF_TYPE = 1 (MOS)
55
CLKinX* - CLKinX
VCLKin- VIH
VCLKin- VIL
High input voltage
Low input voltage
DC coupled to CLKinX; CLKinX* AC
coupled to Ground
CLKinX_BUF_TYPE = 1 (MOS)
2.0
0.0
VCC
0.4
V
V
FBCLKin/FBCLKin* and Fin/Fin* INPUT SPECIFICATIONS
AC coupled
(CLKinX_BUF_TYPE = 0)
MODE = 2 or 8; FEEDBACK_MUX =
6
fFBCLKin
Clock input frequency(4)
Clock input frequency(4)
0.001
0.001
1000
MHz
MHz
AC coupled
(CLKinX_BUF_TYPE = 0)
MODE = 3 or 11
fFin
3100
2.0
Single Ended
AC coupled;
(CLKinX_BUF_TYPE = 0)
VFBCLKin/Fin
0.25
0.15
Vpp
Clock input voltage(4)
AC coupled; 20% to 80%;
(CLKinX_BUF_TYPE = 0)
SLEWFBCLKin/Fin Slew rate on CLKin(4)(1)
0.5
V/ns
(1) In order to meet the jitter performance listed in the subsequent sections of this data sheet, the minimum recommended slew rate for all
input clocks is 0.5 V/ns. This is especially true for single-ended clocks. Phase noise performance begins to degrade as the clock input
slew rate is reduced. However, the device functions at slew rates down to the minimum listed. When compared to single-ended clocks,
differential clocks (LVDS, LVPECL) are less susceptible to degradation in phase noise performance at lower slew rates due to their
common mode noise rejection. However, it is also recommended to use the highest possible slew rate for differential clocks to achieve
optimal phase noise performance at the device outputs.
(2) Load conditions for output clocks: LVDS: 100 Ω differential. See Current Consumption and Power Dissipation Calculations for Icc for
specific part configuration and how to calculate Icc for a specific design.
(3) CLKin0, CLKin1 maximum is specified by characterization, production tested at 200 MHz.
(4) Specified by characterization.
(5) See Differential Voltage Measurement Terminology for definition of VID and VOD voltages.
Copyright © 2016, Texas Instruments Incorporated
7
LMK04208
ZHCSFH1 –SEPTEMBER 2016
www.ti.com.cn
Electrical Characteristics (continued)
3.15 V ≤ VCC ≤ 3.45 V, –40 °C ≤ TA ≤ 85°C. Typical values represent most likely parametric norms at VCC = 3.3 V, TA = 25°C,
at the Recommended Operating Conditions at the time of product characterization and are not specified.(1)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX UNIT
PLL1 SPECIFICATIONS
fPD1
PLL1 phase detector frequency
40
MHz
µA
VCPout1 = VCC/2, PLL1_CP_GAIN = 0
VCPout1 = VCC/2, PLL1_CP_GAIN = 1
VCPout1 = VCC/2, PLL1_CP_GAIN = 2
VCPout1 = VCC/2, PLL1_CP_GAIN = 3
VCPout1=VCC/2, PLL1_CP_GAIN = 0
VCPout1=VCC/2, PLL1_CP_GAIN = 1
VCPout1=VCC/2, PLL1_CP_GAIN = 2
VCPout1=VCC/2, PLL1_CP_GAIN = 3
100
200
PLL1 charge
ICPout1SOURCE
Pump source current(6)
400
1600
–100
–200
–400
–1600
PLL1 charge
ICPout1SINK
µA
Pump sink current(6)
Charge pump
Sink/source mismatch
ICPout1%MIS
ICPout1VTUNE
ICPout1%TEMP
ICPout1 TRI
VCPout1 = VCC/2, T = 25 °C
3%
4%
4%
10%
Magnitude of charge pump current
variation vs. charge pump voltage
0.5 V < VCPout1 < VCC - 0.5 V
TA = 25 °C
Charge pump current vs.
temperature variation
Charge Pump TRI-STATE leakage
current
0.5 V < VCPout < VCC - 0.5 V
5
nA
PLL 1/f noise at 10 kHz offset.(7)
Normalized to 1 GHz Output Frequency
PLL1_CP_GAIN = 400 µA
PLL1_CP_GAIN = 1600 µA
PLL1_CP_GAIN = 400 µA
PLL1_CP_GAIN = 1600 µA
–117
–118
PN10kHz
PN1Hz
dBc/Hz
–221.5
–223
Normalized phase noise contribution(8)
dBc/Hz
PLL2 REFERENCE INPUT (OSCin) SPECIFICATIONS
fOSCin
PLL2 reference input(9)
500
2.4
MHz
V/ns
PLL2 reference clock minimum slew rate
on OSCin(4)
SLEWOSCin
20% to 80%
0.15
0.2
0.5
AC coupled; Single-ended (Unused
pin AC coupled to GND)
VOSCin
Input voltage for OSCin or OSCin*(4)
Vpp
VIDOSCin
VSSOSCin
0.2
0.4
1.55
3.1
|V|
Differential voltage swing (see Figure 8) AC coupled
Vpp
DC offset voltage between
VOSCin-offset
OSCin/OSCin*
Each pin AC coupled
20
mV
OSCinX* - OSCinX
EN_PLL2_REF_2X = 1;(10)
OSCin Duty Cycle 40% to 60%
fdoubler_max
Doubler input frequency(4)
155
MHz
(6) This parameter is programmable
(7) A specification in modeling PLL in-band phase noise is the 1/f flicker noise, LPLL_flicker(f), which is dominant close to the carrier. Flicker
noise has a 10 dB/decade slope. PN10kHz is normalized to a 10 kHz offset and a 1 GHz carrier frequency. PN10kHz = LPLL_flicker(10
kHz) - 20log(Fout / 1 GHz), where LPLL_flicker(f) is the single side band phase noise of only the flicker noise's contribution to total noise,
L(f). To measure LPLL_flicker(f) it is important to be on the 10 dB/decade slope close to the carrier. A high compare frequency and a clean
crystal are important to isolating this noise source from the total phase noise, L(f). LPLL_flicker(f) can be masked by the reference
oscillator performance if a low power or noisy source is used. The total PLL in-band phase noise performance is the sum of LPLL_flicker(f)
and LPLL_flat(f).
(8) A specification modeling PLL in-band phase noise. The normalized phase noise contribution of the PLL, LPLL_flat(f), is defined as:
PN1HZ=LPLL_flat(f) - 20log(N) - 10log(fPDX). LPLL_flat(f) is the single side band phase noise measured at an offset frequency, f, in a 1 Hz
bandwidth and fPDX is the phase detector frequency of the synthesizer. LPLL_flat(f) contributes to the total noise, L(f).
(9) FOSCin maximum frequency specified by characterization. Production tested at 200 MHz.
(10) The EN_PLL2_REF_2X bit (Register 13) enables/disables a frequency doubler mode for the PLL2 OSCin path.
8
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ZHCSFH1 –SEPTEMBER 2016
Electrical Characteristics (continued)
3.15 V ≤ VCC ≤ 3.45 V, –40 °C ≤ TA ≤ 85°C. Typical values represent most likely parametric norms at VCC = 3.3 V, TA = 25°C,
at the Recommended Operating Conditions at the time of product characterization and are not specified.(1)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX UNIT
CRYSTAL OSCILLATOR MODE SPECIFICATIONS
fXTAL
Crystal frequency range(4)
RESR < 40 Ω
6
20.5
MHz
µW
Vectron VXB1 crystal, 20.48 MHz,
RESR < 40 Ω
XTAL_LVL = 0
PXTAL
Crystal power dissipation(11)
100
6
Input capacitance of
LMK04208 OSCin port
CIN
-40 to +85 °C
pF
PLL2 PHASE DETECTOR and CHARGE PUMP SPECIFICATIONS
fPD2
Phase detector frequency
155
MHz
VCPout2=VCC/2, PLL2_CP_GAIN = 0
VCPout2=VCC/2, PLL2_CP_GAIN = 1
VCPout2=VCC/2, PLL2_CP_GAIN = 2
VCPout2=VCC/2, PLL2_CP_GAIN = 3
VCPout2=VCC/2, PLL2_CP_GAIN = 0
VCPout2=VCC/2, PLL2_CP_GAIN = 1
VCPout2=VCC/2, PLL2_CP_GAIN = 2
VCPout2=VCC/2, PLL2_CP_GAIN = 3
VCPout2=VCC/2, TA = 25 °C
100
400
ICPoutSOURCE
PLL2 charge pump source current(6)
µA
µA
1600
3200
–100
–400
–1600
–3200
3%
ICPoutSINK
PLL2 charge pump sink current(6)
Charge pump sink/source mismatch
ICPout2%MIS
ICPout2VTUNE
10%
10
Magnitude of charge pump current vs.
charge pump voltage variation
0.5 V < VCPout2 < VCC - 0.5 V
TA = 25 °C
4%
4%
Charge pump current vs.
Temperature variation
ICPout2%TEMP
ICPout2TRI
Charge pump leakage
0.5 V < VCPout2 < VCC - 0.5 V
PLL2_CP_GAIN = 400 µA
PLL2_CP_GAIN = 3200 µA
PLL2_CP_GAIN = 400 µA
PLL2_CP_GAIN = 3200 µA
nA
PLL 1/f Noise at 10 kHz offset(7)
Normalized to 1 GHz output frequency
–118
–121
PN10kHz
PN1Hz
dBc/Hz
–222.5
–227
Normalized Phase Noise Contribution(8)
dBc/Hz
MHz
INTERNAL VCO SPECIFICATIONS
fVCO
VCO tuning range
LMK04208
LMK04208
2750
3072
Fine tuning sensitivity
(The range displayed in the typical
column indicates the lower sensitivity is
typical at the lower end of the tuning
range, and the higher tuning sensitivity is
typical at the higher end of the tuning
range).
KVCO
20 to 36
MHz/V
°C
After programming R30 for lock, no
changes to output configuration are
permitted to ensure continuous lock
Allowable Temperature Drift for
Continuous Lock(12) (4)
|ΔTCL
|
125
(11) See Application Section discussion of Optional Crystal Oscillator Implementation (OSCin/OSCin*).
(12) Maximum Allowable Temperature Drift for Continuous Lock is how far the temperature can drift in either direction from the value it was
at the time that the R30 register was last programmed, and still have the part stay in lock. The action of programming the R30 register,
even to the same value, activates a frequency calibration routine. This implies the part works over the entire frequency range, but if the
temperature drifts more than the maximum allowable drift for continuous lock, then it is necessary to reload the R30 register to ensure it
stays in lock. Regardless of what temperature the part was initially programmed at, the temperature can never drift outside the
frequency range of -40 °C to 85 °C without violating specifications.
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Electrical Characteristics (continued)
3.15 V ≤ VCC ≤ 3.45 V, –40 °C ≤ TA ≤ 85°C. Typical values represent most likely parametric norms at VCC = 3.3 V, TA = 25°C,
at the Recommended Operating Conditions at the time of product characterization and are not specified.(1)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX UNIT
CLKout CLOSED LOOP JITTER SPECIFICATIONS USING a COMMERCIAL QUALITY VCXO(13)
Offset = 1 kHz
Offset = 10 kHz
–122.5
–132.9
–135.2
–143.9
–156.0
LMK04208
Offset = 100 kHz
fCLKout = 245.76 MHz
SSB Phase noise
Offset = 800 kHz
L(f)CLKout
dBc/Hz
Offset = 10 MHz; LVDS
Measured at clock outputs
Value is average for all output types(14)
Offset = 10 MHz; LVPECL 1600
mVpp
–157.5
Offset = 10 MHz; LVCMOS
BW = 12 kHz to 20 MHz
–157.1
111
JCLKout
LMK04208(14)
LVDS/LVPECL/ fCLKout = 245.76 MHz
LVCMOS Integrated RMS jitter
fs, RMS
BW = 100 Hz to 20 MHz
123
(15)
CLKout CLOSED LOOP JITTER SPECIFICATIONS USING THE INTEGRATED LOW NOISE CRYSTAL OSCILLATOR CIRCUIT
BW = 12 kHz to 20 MHz
XTAL_LVL = 3
192
450
LMK04208
fCLKout = 245.76 MHz
Integrated RMS jitter
fs rms
MHz
BW = 100 Hz to 20 MHz
XTAL_LVL = 3
DEFAULT POWER ON RESET CLOCK OUTPUT FREQUENCY
Default output clock frequency at device
fCLKout-startup
CLKout4, LVDS, LMK04208
90
110
130
power on(16)
(13) VCXO used is a 122.88-MHz Crystek CVHD-950-122.880.
(14) fVCO = 2949.12 MHz, PLL1 parameters: FPD1 = 1.024 MHz, ICP1 = 100 μA, loop bandwidth = 10 Hz. 122.88 MHz Crystek CVHD-
950–122.880. PLL2 parameters: PLL2_R = 1, FPD2 = 122.88 MHz, ICP2 = 3200 μA, C1 = 47 pF, C2 = 3.9 nF, R2 = 620 Ω, PLL2_C3_LF
= 0, PLL2_R3_LF = 0, PLL2_C4_LF = 0, PLL2_R4_LF = 0, CLKoutX_DIV = 12, and CLKoutX_ADLY_SEL = 0.
(15) Crystal used is a 20.48-MHz Vectron VXB1-1150-20M480 and Skyworks varactor diode, SMV-1249-074LF.
(16) CLKout3 and OSCout also oscillate at start-up at the frequency of the VCXO attached to OSCin port.
10
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ZHCSFH1 –SEPTEMBER 2016
Electrical Characteristics (continued)
3.15 V ≤ VCC ≤ 3.45 V, –40 °C ≤ TA ≤ 85°C. Typical values represent most likely parametric norms at VCC = 3.3 V, TA = 25°C,
at the Recommended Operating Conditions at the time of product characterization and are not specified.(1)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX UNIT
CLOCK SKEW and DELAY
LVDS-to-LVDS, T = 25 °C,
FCLK = 800 MHz, RL= 100 Ω
AC coupled
30
LVPECL-to-LVPECL,
T = 25 °C,
Maximum CLKoutX to CLKoutY(4)(17)
FCLK = 800 MHz, RL= 100 Ω
emitter resistors =
240 Ω to GND
|TSKEW
|
30
ps
AC coupled
Maximum skew between any two
LVCMOS outputs, same CLKout or
different CLKout(4)(17)
RL = 50 Ω, CL = 5 pF,
T = 25 °C, FCLK = 100 MHz.
100
Same device, T = 25 °C,
250 MHz
MixedTSKEW
LVDS or LVPECL to LVCMOS
750
ps
ps
MODE = 2
PLL1_R_DLY = 0; PLL1_N_DLY = 0
1850
MODE = 2
PLL1_R_DLY = 0; PLL1_N_DLY = 0;
VCO Frequency = 2949.12 MHz
Analog delay select = 0;
Feedback clock digital delay = 11;
Feedback clock half step = 1;
Output clock digital delay = 5;
Output clock half step = 0;
td0-DELAY
CLKin to CLKoutX delay(17)
0
LVDS CLOCK OUTPUTS (CLKoutX), CLKoutX_TYPE = 1
fCLKout
VOD
Maximum frequency(4)(18)
RL = 100 Ω
1536
250
MHz
400
800
450
|mV|
Differential output voltage (see Figure 9)
VSS
500
900 mVpp
Change in magnitude of VOD for
complementary output states
T = 25 °C, DC measurement
AC coupled to receiver input
R = 100-Ω differential termination
ΔVOD
VOS
–50
50
1.375
35
mV
V
Output offset voltage
1.125
1.25
200
Change in VOS for complementary output
states
ΔVOS
|mV|
Output rise time
Output fall time
20% to 80%, RL = 100 Ω
80% to 20%, RL = 100 Ω
TR / TF
ps
ISA
ISB
Output short circuit current
single-ended
Single-ended output shorted to GND
T = 25 °C
–24
–12
24
12
mA
mA
ISAB
Output short circuit current - differential
Complimentary outputs tied together
(17) Equal loading and identical clock output configuration on each clock output is required for specification to be valid. Specification is not
valid for CLKoutX or CLKoutY in analog delay mode.
(18) Refer to Typical Characteristics for output operation performance at higher frequencies than the minimum maximum output frequency.
Copyright © 2016, Texas Instruments Incorporated
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Electrical Characteristics (continued)
3.15 V ≤ VCC ≤ 3.45 V, –40 °C ≤ TA ≤ 85°C. Typical values represent most likely parametric norms at VCC = 3.3 V, TA = 25°C,
at the Recommended Operating Conditions at the time of product characterization and are not specified.(1)
PARAMETER
LVPECL CLOCK OUTPUTS (CLKoutX)
TEST CONDITIONS
MIN
TYP
MAX UNIT
fCLKout
Maximum frequency(4)(18)
1536
MHz
ps
20% to 80% output rise
RL = 100 Ω, emitter resistors = 240 Ω
to GND
CLKoutX_TYPE = 4 or 5
(1600 or 2000 mVpp)
TR / TF
150
80% to 20% output fall time
700-mVpp LVPECL CLOCK OUTPUTS (CLKoutX), CLKoutX_TYPE = 2
VCC
1.03
–
VOH
VOL
Output high voltage
Output low voltage
V
V
T = 25 °C, DC measurement
Termination = 50 Ω to
VCC - 1.4 V
VCC
1.41
–
VOD
VSS
305
610
380
760
440
|mV|
Output voltage (see Figure 9)
880 mVpp
1200-mVpp LVPECL CLOCK OUTPUTS (CLKoutX), CLKoutX_TYPE = 3
VCC
1.07
–
VOH
VOL
Output high voltage
Output low voltage
V
V
T = 25 °C, DC measurement
Termination = 50 Ω to
VCC - 1.7 V
VCC
1.69
–
VOD
VSS
545
625
705
|mV|
Output voltage (see Figure 9)
1090
1250
1410 mVpp
1600-mVpp LVPECL CLOCK OUTPUTS (CLKoutX), CLKoutX_TYPE = 4
VCC
1.10
–
VOH
VOL
Output high voltage
Output low voltage
V
V
T = 25 °C, DC Measurement
Termination = 50 Ω to
VCC - 2.0 V
VCC
1.97
–
VOD
VSS
660
870
965
|mV|
Output voltage (see Figure 9)
1320
1740
1930 mVpp
2000-mVpp LVPECL (2VPECL) CLOCK OUTPUTS (CLKoutX), CLKoutX_TYPE = 5
VCC
1.13
–
VOH
VOL
Output high voltage
Output low voltage
V
V
T = 25 °C, DC Measurement
Termination = 50 Ω to
VCC - 2.3 V
VCC
2.20
–
VOD
VSS
800
1070
2140
1200
|mV|
Output voltage Figure 9
1600
2400 mVpp
LVCMOS CLOCK OUTPUTS (CLKoutX)
fCLKout
Maximum frequency(4)(18)
5 pF Load
1 mA Load
250
MHz
V
VCC
–
VOH
Output high voltage
0.1
VOL
IOH
IOL
Output low voltage
1 mA Load
0.1
V
Output high current (source)
Output low current (sink)
VCC = 3.3 V, VO = 1.65 V
VCC = 3.3 V, VO = 1.65 V
28
28
mA
mA
VCC/2 to VCC/2, FCLK = 100 MHz
T = 25 °C
DUTYCLK
Output duty cycle(4)
Output rise time
Output fall time
45%
50%
400
400
55%
20% to 80%, RL = 50 Ω,
CL = 5 pF
TR
TF
ps
ps
80% to 20%, RL = 50 Ω,
CL = 5 pF
12
Copyright © 2016, Texas Instruments Incorporated
LMK04208
www.ti.com.cn
ZHCSFH1 –SEPTEMBER 2016
Electrical Characteristics (continued)
3.15 V ≤ VCC ≤ 3.45 V, –40 °C ≤ TA ≤ 85°C. Typical values represent most likely parametric norms at VCC = 3.3 V, TA = 25°C,
at the Recommended Operating Conditions at the time of product characterization and are not specified.(1)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX UNIT
DIGITAL OUTPUTS (Status_CLKinX, Status_LD, Status_Holdover, SYNC)
VCC
0.4
–
VOH
VOL
High-level output voltage
Low-level output voltage
IOH = -500 µA
IOL = 500 µA
V
0.4
V
DIGITAL INPUTS (Status_CLKinX, SYNC)
VIH
VIL
High-level input voltage
Low-level input voltage
1.6
VCC
0.4
V
V
Status_CLKinX_TYPE = 0
(High Impedance)
–5
–5
5
5
High-level input current
VIH = VCC
Status_CLKinX_TYPE = 1
(Pull-up)
IIH
µA
µA
Status_CLKinX_TYPE = 2
(Pull-down)
10
80
5
Status_CLKinX_TYPE = 0
(High Impedance)
–5
Low-level input current
VIL = 0 V
Status_CLKinX_TYPE = 1
(Pull-up)
IIL
–40
–5
-5
5
Status_CLKinX_TYPE = 2
(Pull-down)
DIGITAL INPUTS (CLKuWire, DATAuWire, LEuWire)
VIH
VIL
IIH
High-level input voltage
Low-level input voltage
High-level input current
Low-level input current
1.6
VCC
0.4
25
5
V
V
VIH = VCC
VIL = 0
5
µA
µA
IIL
–5
6.6 Timing Requirements
See Programming for additional information
MIN
25
25
8
NOM
MAX UNIT
TECS
TDCS
TCDH
TCWH
TCWL
TCES
TEWH
TCR
LE to clock set up time
Data to clock set up time
Clock to data hold time
Clock pulse width high
Clock pulse width low
Clock to LE set up time
LE pulse width
See Figure 1 through Figure 4
See Figure 1
ns
ns
ns
ns
ns
ns
ns
ns
See Figure 1
See Figure 1, Figure 2, and Figure 4
See Figure 1, Figure 2, and Figure 4
See Figure 1 through Figure 4
See Figure 1, Figure 2, and Figure 4
See Figure 4
25
25
25
25
25
Falling clock to readback time
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MSB
D26
LSB
A0
DATAuWire
CLKuWire
LEuWire
D25
D24
D23
D22
D0
A4
A1
t
CES
t
ECS
t
ECS
t
t
CWL
CWH
t
t
CDH
DCS
t
EWH
Figure 1. MICROWIRE Input Timing Diagram
MSB
D26
LSB
A0
DATAuWire
CLKuWire
LEuWire
t
ECS
t
CES
t
t
CWH
CWL
t
EWH
Figure 2. MICROWIRE Timing Diagram: Extra CLKuWire Pulses for R0 to R5
MSB
D26
LSB
A0
DATAuWire
CLKuWire
LEuWire
t
ECS
t
CES
t
CES
Figure 3. MICROWIRE Timing Diagram: Extra CLKuWire Pulses for R0 to R5 with LEuWire Asserted
MSB
D26
LSB
A0
DATAuWire
CLKuWire
t
CR
t
ECS
t
CWH
t
CR
t
CWL
LEuWire
t
CES
READBACK_LE = 0
t
EWH
t
ECS
LEuWire
READBACK_LE = 1
Readback Pin
RD26
RD25
RD24
RD23
RD0
Register Write
Register Read
Figure 4. MICROWIRE Readback Timing Diagram
14
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LMK04208
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ZHCSFH1 –SEPTEMBER 2016
6.7 Typical Characteristics
500
450
400
350
300
250
200
150
100
50
1200
1000
800
600
400
200
0
2000 mVpp
1600 mVpp
1200 mVpp
700 mVpp
0
0
500 1000 1500 2000 2500 3000
FREQUENCY (MHz)
0
500 1000 1500 2000 2500 3000
FREQUENCY (MHz)
Figure 5. LVDS VOD vs Frequency
Figure 6. LVPECL with 240-Ω Emitter Resistors
VOD vs Frequency
1200
1000
800
2000 mVpp
600
1600 mVpp
400
200
0
0
500 1000 1500 2000 2500 3000
FREQUENCY (MHz)
Figure 7. LVPECL with 120-Ω Emitter Resistors
VOD vs Frequency
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7 Parameter Measurement Information
7.1 Charge Pump Current Specification Definitions
I1 = Charge Pump Sink Current at VCPout = VCC - ΔV
I2 = Charge Pump Sink Current at VCPout = VCC/2
I3 = Charge Pump Sink Current at VCPout = ΔV
I4 = Charge Pump Source Current at VCPout = VCC - ΔV
I5 = Charge Pump Source Current at VCPout = VCC/2
I6 = Charge Pump Source Current at VCPout = ΔV
ΔV = Voltage offset from the positive and negative supply rails. Defined to be 0.5 V for this device.
7.1.1 Charge Pump Output Current Magnitude Variation Vs. Charge Pump Output Voltage
7.1.2 Charge Pump Sink Current Vs. Charge Pump Output Source Current Mismatch
7.1.3 Charge Pump Output Current Magnitude Variation vs. Ambient Temperature
16
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ZHCSFH1 –SEPTEMBER 2016
7.2 Differential Voltage Measurement Terminology
The differential voltage of a differential signal can be described by two different definitions causing confusion
when reading datasheets or communicating with other engineers. This section will address the measurement and
description of a differential signal so that the reader will be able to understand and discern between the two
different definitions when used.
The first definition used to describe a differential signal is the absolute value of the voltage potential between the
inverting and non-inverting signal. The symbol for this first measurement is typically VID or VOD depending on if
an input or output voltage is being described.
The second definition used to describe a differential signal is to measure the potential of the non-inverting signal
with respect to the inverting signal. The symbol for this second measurement is VSS and is a calculated
parameter. Nowhere in the IC does this signal exist with respect to ground, it only exists in reference to its
differential pair. VSS can be measured directly by oscilloscopes with floating references, otherwise this value can
be calculated as twice the value of VOD as described in the first description.
Figure 8 illustrates the two different definitions side-by-side for inputs and Figure 9 illustrates the two different
definitions side-by-side for outputs. The VID and VOD definitions show VA and VB DC levels that the non-inverting
and inverting signals toggle between with respect to ground. VSS input and output definitions show that if the
inverting signal is considered the voltage potential reference, the non-inverting signal voltage potential is now
increasing and decreasing above and below the non-inverting reference. Thus the peak-to-peak voltage of the
differential signal can be measured.
VID and VOD are often defined as volts (V) and VSS is often defined as volts peak-to-peak (VPP).
V
ID
Definition
V
Definition for Input
SS
Non-Inverting Clock
V
V
A
B
2·V
V
ID
ID
Inverting Clock
= | V - V
V
|
B
V
SS
= 2·V
ID
ID
A
GND
Figure 8. Two Different Definitions for Differential Input Signals
V
Definition
V
Definition for Output
SS
OD
Non-Inverting Clock
V
V
A
B
2·V
V
OD
OD
Inverting Clock
= | V - V
V
|
B
V
SS
= 2·V
OD
OD
A
GND
Figure 9. Two Different Definitions for Differential Output Signals
See AN-912, Common Data Transmission Parameters and their Definitions SNLA036, for more information.
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8 Detailed Description
8.1 Overview
In default mode of operation, dual PLL mode with internal VCO, the Phase Frequency Detector in PLL1
compares the active CLKinX reference divided by CLKinX_PreR_DIV and PLL1 R divider with the external
VCXO or crystal attached to the PLL2 OSCin port divided by PLL1 N divider. The external loop filter for PLL1
should be narrow to provide an ultra clean reference clock from the external VCXO or crystal to the
OSCin/OSCin* pins for PLL2.
The Phase Frequency Detector in PLL2 compares the external VCXO or crystal to the internal VCO after the
reference and feedback dividers. The VCXO or crystal on the OSCin input is divided by PLL2 R divider. The
feedback from the internal VCO is divided by the PLL2 Prescaler, the PLL2 N divider, and optionally the VCO
divider.
The bandwidth of the external loop filter for PLL2 should be designed to be wide enough to take advantage of
the low in-band phase noise of PLL2 and the low high offset phase noise of the internal VCO. The VCO output is
also placed on the distribution path for the Clock Distribution section. The clock distribution consists of 6 outputs.
Each clock output allows the user to select a divide value, a digital delay value, and an analog delay. The 6 clock
outputs drive programmable output buffers. Two clock outputs allow their input signal to be from the OSCin port
directly.
When a 0-delay mode is used, a clock output will be passed through the feedback mux to the PLL1 N Divider for
synchronization and 0-delay.
When an external VCO mode is used, the Fin port will be used to input an external VCO signal. PLL2 Phase
comparison will now be with this signal divided by the PLL2 N divider and N2 pre-scaler. The VCO divider may
not be used. One less clock input is available when using an external VCO mode.
When a single PLL mode is used, PLL1 is powered down. OSCin is used as a reference to PLL2.
8.1.1 System Architecture
The dual loop PLL architecture of the LMK04208 provides the lowest jitter performance over the widest range of
output frequencies and phase noise integration bandwidths. The first stage PLL (PLL1) is driven by an external
reference clock and uses an external VCXO or tunable crystal to provide a frequency accurate, low phase noise
reference clock for the second stage frequency multiplication PLL (PLL2). PLL1 typically uses a narrow loop
bandwidth (10 Hz to 200 Hz) to retain the frequency accuracy of the reference clock input signal while at the
same time suppressing the higher offset frequency phase noise that the reference clock may have accumulated
along its path or from other circuits. This cleaned reference clock provides the reference input to PLL2.
The low phase noise reference provided to PLL2 allows PLL2 to operate with a wide loop bandwidth (50 kHz to
200 kHz). The loop bandwidth for PLL2 is chosen to take advantage of the superior high offset frequency phase
noise profile of the internal VCO and the good low offset frequency phase noise of the reference VCXO or
tunable crystal.
Ultra low jitter is achieved by allowing the external VCXO or crystal’s phase noise to dominate the final output
phase noise at low offset frequencies and the internal (or external) VCO’s phase noise to dominate the final
output phase noise at high offset frequencies. This results in best overall phase noise and jitter performance.
The LMK04208 allows subsets of the device to be used to increase the flexibility of device. These different
modes are selected using MODE: Device Mode. For instance:
•
Dual Loop Mode - Typical use case of LMK04208. CLKinX used as reference input to PLL1, OSCin port is
connected to VCXO or tunable crystal.
•
•
Single Loop Mode - Powers down PLL1. OSCin port is used as reference input.
Clock Distribution Mode - Allows input of CLKin1 to be distributed to output with division, digital delay, and
analog delay.
See Device Functional Modes for more information on these modes.
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Overview (continued)
8.1.2 PLL1 Redundant Reference Inputs (CLKin0/CLKin0* and CLKin1/CLKin1*)
The LMK04208 has two reference clock inputs for PLL1: CLKin0 and CLKin1. Ref Mux selects CLKin0 or
CLKin1. Automatic or manual switching occurs between the inputs.
CLKin0 and CLKin1 each have input dividers. The input divider allows different clock input frequencies to be
normalized so that the frequency input to the PLL1 R divider remains constant during automatic switching. By
programming these dividers such that the frequency presented to the input of the PLL1 R divider is the same
prevents the user from needing to reprogram the PLL1 R divider when the input reference is changed to another
CLKin port with a different frequency.
CLKin1 is shared for use as an external 0-delay feedback (FBCLKin), or for use with an external VCO (Fin).
Fast manual switching between reference clocks is possible with external pins Status_CLKin0 and
Status_CLKin1.
8.1.3 PLL1 Tunable Crystal Support
The LMK04208 integrates a crystal oscillator on PLL1 for use with an external crystal and varactor diode to
perform jitter cleaning.
The LMK04208 must be programmed to enable Crystal mode.
8.1.4 VCXO/Crystal Buffered Output
The LMK04208 provides a dedicated output, OSCout, which is a buffered copy of the PLL2 reference input (see
Functional Block Diagram for a block diagram of this implementation). The PLL2 reference input is typically a low
noise VCXO or Crystal. When using a VCXO, this output can be used to clock external devices such as
microcontrollers, FPGAs, CPLDs, and so forth, before the LMK04208 is programmed. See Clock Output
Synchronization and MODE: Device Mode for further reference of these outputs
The OSCout buffer output type is programmable to LVDS, LVPECL, or LVCMOS.
The dedicated output buffer OSCout can output frequency lower than the VCXO or Crystal frequency by
programming the OSC Divider. The OSC Divider value range is 2 to 8.
Two clock outputs can also be programmed to be driven by OSCin. This allows a total of 2 additional differential
outputs to be buffered outputs of OSCin. When programmed in this way, a total of 3 differential or 6 single-ended
outputs can be driven by a buffered copy of OSCin.
VCXO/Crystal buffered outputs cannot be synchronized to the VCO clock distribution outputs. The assertion of
SYNC will still cause these outputs to become low temporarily. Since these outputs will turn off and on
asynchronously with respect to the VCO sourced clock outputs during a SYNC, it is possible for glitches to occur
on the buffered clock outputs when SYNC is asserted and unasserted. If the NO_SYNC_CLKoutX bits are set
these outputs will not be affected by the SYNC event except that the phase relationship will change with the
other synchronized clocks unless a buffered clock output is used as a qualification clock during SYNC.
8.1.5 Frequency Holdover
The LMK04208 supports holdover operation to keep the clock outputs on frequency with minimum drift when the
reference is lost until a valid reference clock signal is re-established.
8.1.6 Integrated Loop Filter Poles
The LMK04208 features programmable 3rd and 4th order loop filter poles for PLL2. These internal resistors and
capacitor values may be selected from a fixed range of values to achieve either a 3rd or 4th order loop filter
response. The integrated programmable resistors and capacitors compliment external components mounted near
the chip.
These integrated components can be effectively disabled by programming the integrated resistors and capacitors
to their minimum values.
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Overview (continued)
8.1.7 Internal VCO
The output of the internal VCO is routed to a mux which allows the user to select either the direct VCO output or
a divided version of the VCO for the Clock Distribution Path. This same selection is also fed back to the PLL2
phase detector through a prescaler and N-divider.
The mux selectable VCO divider has a divide range of 2 to 8 with 50% output duty cycle for both even and odd
divide values.
The primary use of the VCO divider is to achieve divides greater than the clock output divider supports alone.
8.1.8 External VCO Mode
The Fin/Fin* input allows an external VCO to be used with PLL2 of the LMK04208. An external VCO may be
needed to meet stringent output phase noise/jitter requirements in some applications, such as multi-carrier GSM.
An external VCO is permitted in single PLL, dual PLL, or 0-delay dual PLL mode. In 0-delay dual PLL mode, the
clock outputs driven from the external VCO can have deterministic phase with the clock input.
Using an external VCO reduces the number of available clock inputs by one. The VCO divider cannot be used
with an external VCO.
8.1.9 Clock Distribution
The LMK04208 features a total of 6 differential outputs driven from the internal or external VCO.
All VCO driven outputs have programmable output types. They can be programmed to LVPECL, LVDS, or
LVCMOS. When all distribution outputs are configured for LVCMOS or single ended LVPECL a total of 12
outputs are available.
If the buffered OSCin output OSCout is included in the total number of clock outputs the LMK04208 is able to
distribute, then up to 7 differential clocks or up to 14 single-ended clocks may be generated with the LMK04208.
The following sections discuss specific features of the clock distribution channels that allow the user to control
various aspects of the output clocks.
8.1.9.1 CLKout DIVIDER
Each clock has a single clock output divider. The divider supports a divide range of 1 to 1045 (even and odd)
with 50% output duty cycle. When divides of 26 or greater are used, the divider/delay block uses extended mode.
The VCO Divider may be used to reduce the divide needed by the clock group divider so that it may operate in
normal mode instead of extended mode. This can result in a small current saving if enabling the VCO Divider
allows 3 or more clock output divides to change from extended to normal mode.
8.1.9.2 CLKout Delay
See Clock Distribution section for details on both a fine (analog) and coarse (digital) delay for phase adjustment
of the clock outputs.
The fine (analog) delay allows a nominal 25-ps step size and range from 0 to 475 ps of total delay. Enabling the
analog delay adds a nominal 500 ps of delay in addition to the programmed value. When adjusting analog delay,
glitches may occur on the clock outputs being adjusted. Analog delay may not operate at frequencies above the
minimum-ensured maximum output frequency of 1536 MHz.
The coarse (digital) delay allows a group of outputs to be delayed by 4.5 to 12 clock distribution path cycles in
normal mode, or from 12.5 to 522 VCO cycles in extended mode. The delay step can be as small as half the
period of the clock distribution path by using the CLKoutX_HS bit provided the output divide value is greater than
1. For example, a 2-GHz VCO frequency without the use of the VCO divider results in 250 ps coarse tuning
steps. The coarse (digital) delay value takes effect on the clock outputs after a SYNC event.
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Overview (continued)
There are 3 different ways to use the digital (coarse) delay:
1. Fixed Digital Delay
2. Absolute Dynamic Digital Delay
3. Relative Dynamic Digital Delay
These are further discussed in Clock Distribution.
8.1.9.3 Programmable Output Type
For increased flexibility all LMK04208 clock outputs (CLKoutX) and OSCout can be programmed to an LVDS,
LVPECL, or LVCMOS output type.
Any LVPECL output type can be programmed to 700-, 1200-, 1600-, or 2000-mVpp amplitude levels. The 2000-
mVpp LVPECL output type is a Texas Instruments proprietary configuration that produces a 2000-mVpp
differential swing for compatibility with many data converters and is also known as 2VPECL.
8.1.9.4 Clock Output Synchronization
Using the SYNC input causes all active clock outputs to share a rising edge. See Clock Output Synchronization
(SYNC) for more information.
The SYNC event also causes the digital delay values to take effect.
8.1.10 0-Delay
The 0-delay mode synchronizes the input clock phase to the output clock phase. The 0-delay feedback may be
performed with an internal feedback loop from any of the clock groups or with an external feedback loop into the
FBCLKin port as selected by the FEEDBACK_MUX.
Without using 0-delay mode, there will be D possible fixed phase relationships from clock input to clock output
depending on the clock output divide value.
Using an external 0-delay feedback reduces the number of available clock inputs by one.
8.1.11 Default Startup Clocks
Before the LMK04208 is programmed, CLKout4 is enabled and operating at a nominal frequency and CLKout3
and OSCout are enabled and operating at the OSCin frequency. These clocks can be used to clock external
devices such as microcontrollers, FPGAs, CPLDs, and so forth, before the LMK04208 is programmed.
For CLKout3 and OSCout to work before the LMK04208 is programmed, the device must not be using Crystal
mode.
8.1.12 Status Pins
The LMK04208 provides status pins which can be monitored for feedback or in some cases used for input
depending upon device programming. For example:
•
•
•
•
The Status_Holdover pin may indicate if the device is in hold-over mode.
The Status_CLKin0 pin may indicate the LOS (loss-of-signal) for CLKin0.
The Status_CLKin0 pin may be an input for selecting the active clock input.
The Status_LD pin may indicate if the device is locked.
The status pins can be programmed to a variety of other outputs including analog lock detect, PLL divider
outputs, combined PLL lock detect signals, PLL1 Vtune railing, readback, and so forth. Refer to the Programming
of this datasheet for more information. Default pin programming is captured in Table 17.
8.1.13 Register Readback
Programmed registers may be read back using the MICROWIRE interface. For readback, one of the status pins
must be programmed for readback mode.
At no time may registers be programed to values other than the valid states defined in the datasheet.
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8.2 Functional Block Diagram
Status_LD
CLKin0 Divider
(1, 2, 4, or 8)
Ref
Mux
R1 Divider
(1 to 16,383)
Device
Control
CLKin0*
CLKin0
Status_Holdover
Status_CLKin0
Status_CLKin1
SYNC
R Delay
N Delay
Phase
Detector
PLL1
CLKin1*/Fin*
FBCLKin*
CLKin1/
CLKin1 Divider
(1, 2, 4, or 8)
N1 Divider
(1 to 16,383)
CLKuWire
Fin/FBCLKin
Fin/Fin*
FBMux
mWire
Port
Control
Registers
DATAuWire
LEuWire
CLKout0
CLKout1
CLKout2
CLKout3
CLKout4
CLKout5
Holdover
FB
Mux
Mode
Mux2
2X
Partially
Integrated
Loop Filter
2X
Mux
R2 Divider
(1 to 4,095)
Internal VCO
Phase
Detector
PLL2
OSCout
OSC Divider
(2 to 8)
OSCout_
MUX
OSCout*
N2 Divider
(1 to 262,143)
Mode
Mux3
FBMux
VCO
Mux
VCO Divider
(2 to 8)
N2 Prescaler
(2 to 8)
Clock Distribution Path Mode
Mux1
OSCin*
OSCin
Fin/Fin*
Osc
Mux1
CLKout0
CLKout0*
CLKout3
Divider
(1 to 1045)
Digital
Delay
Digital
Delay
Divider
(1 to 1045)
Mux
Mux
Mux
CLKout3*
Delay
Delay
Delay
Clock Buffer 1
Osc
Mux2
CLKout1
CLKout1*
CLKout4
CLKout4*
Divider
(1 to 1045)
Digital
Delay
Digital
Delay
Divider
(1 to 1045)
Mux
Mux
Delay
Delay
Clock Buffer 3
CLKout2
CLKout2*
CLKout5
CLKout5*
Divider
(1 to 1045)
Digital
Delay
Digital
Delay
Divider
(1 to 1045)
Mux
Delay
Clock Buffer 2
Clock Buffer 1
22
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8.3 Feature Description
8.3.1 Inputs / Outputs
8.3.1.1 PLL1 Reference Inputs (CLKin0 and CLKin1)
The reference clock inputs for PLL1 may be selected from either CLKin0 or CLKin1. The user has the capability
to manually select one of the inputs or to configure an automatic switching mode of operation. See Input Clock
Switching for more info.
CLKin0 and CLKin1 have dividers which allow the device to switch between reference inputs of different
frequencies automatically without needing to reprogram the PLL1 R divider. The CLKin pre-divider values are 1,
2, 4, and 8.
CLKin1 input can alternatively be used for external feedback in 0-delay mode (FBCLKin) or for an external VCO
input port (Fin).
8.3.1.2 PLL2 OSCin / OSCin* Port
The feedback from the external oscillator being locked with PLL1 drives the OSCin/OSCin* pins. Internally this
signal is routed to the PLL1 N Divider and to the reference input for PLL2.
This input may be driven with either a single-ended or differential signal and must be AC coupled. If operated in
single ended mode, the unused input must be connected to GND with a 0.1-µF capacitor.
8.3.1.3 Crystal Oscillator
The internal circuitry of the OSCin port also supports the optional implementation of a crystal based oscillator
circuit. A crystal, a varactor diode, and a small number of other external components may be used to implement
the oscillator. The internal oscillator circuit is enabled by setting the EN_PLL2_XTAL bit. See EN_PLL2_XTAL.
8.3.2 Input Clock Switching
Manual, pin select, and automatic are three different kinds clock input switching modes can be set with the
CLKin_SELECT_MODE register.
Below is information about how the active input clock is selected and what causes a switching event in the
various clock input selection modes.
8.3.2.1 Input Clock Switching - Manual Mode
When CLKin_SELECT_MODE is 0 or 1 then CLKin0 or CLKin1 respectively is always selected as the active
input clock. Manual mode will also override the EN_CLKinX bits such that the CLKinX buffer will operate even if
CLKinX is disabled with EN_CLKinX = 0.
•
Entering Holdover: If holdover mode is enabled, then holdover mode is entered if Digital lock detect of PLL1
goes low and DISABLE_DLD1_DET = 0.
•
Exiting Holdover: The active clock for automatic exit of holdover mode is the manually selected clock input.
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Feature Description (continued)
8.3.2.2 Input Clock Switching - Pin Select Mode
When CLKin_SELECT_MODE is 3, the pins Status_CLKin0 and Status_CLKin1 select which clock input is
active.
•
Clock Switch Event: Pins: Changing the state of Status_CLKin0 or Status_CLKin1 pins causes an input
clock switch event.
•
Clock Switch Event: PLL1 DLD: To prevent PLL1 DLD high to low transition from causing a input clock
switch event and causing the device to enter holdover mode, disable the PLL1 DLD detect by setting
DISABLE_DLD1_DET = 1. This is the preferred behavior for Pin Select Mode.
•
Configuring Pin Select Mode:
–
The Status_CLKin0_TYPE must be programmed to an input value for the Status_CLKin0 pin to function
as an input for pin select mode.
–
The Status_CLKin1_TYPE must be programmed to an input value for the Status_CLKin1 pin to function
as an input for pin select mode.
–
–
If the Status_CLKinX_TYPE is set as output, the input value is considered 0.
The polarity of Status_CLKin1 and Status_CLKin0 input pins cannot be inverted with the CLKin_SEL_INV
bit.
–
Table 1 defines which input clock is active depending on Status_CLKin0 and Status_CLKin1 state.
Table 1. Active Clock Input - Pin Select Mode
STATUS_CLKin1
STATUS_CLKin0
ACTIVE CLOCK
CLKin0
0
0
1
1
0
1
0
1
CLKin1
Reserved
Holdover
The pin select mode will override the EN_CLKinX bits such that the CLKinX buffer will operate even if CLKinX is
disabled with EN_CLKinX = 0. To switch as fast as possible, keep the clock input buffers enabled (EN_CLKinX =
1) that could be switched to.
8.3.2.2.1 Pin Select Mode and Host
When in the pin select mode, the host can monitor conditions of the clocking system which could cause the host
to switch the active clock input. The LMK04208 device can also provide indicators on the Status_LD and
Status_HOLDOVER like DAC Rail, PLL1 DLD, PLL1 and PLL2 DLD which the host can use in determining which
clock input to use as active clock input.
8.3.2.2.2 Switch Event without Holdover
When an input clock switch event is triggered and holdover mode is disabled, the active clock input immediately
switches to the selected clock. When PLL1 is designed with a narrow loop bandwidth, the switching transient is
minimized.
8.3.2.2.3 Switch Event with Holdover
When an input clock switch event is triggered and holdover mode is enabled, the device will enter holdover mode
and remain in holdover until a holdover exit condition is met as described in Holdover Mode. Then the device will
complete the reference switch to the pin selected clock input.
8.3.2.3 Input Clock Switching - Automatic Mode
When CLKin_SELECT_MODE is 4, the active clock is selected in priority order of enabled clock inputs starting
upon an input clock switch event. The priority order of the clocks is CLKin0 → CLKin1 → CLKin0, and so forth.
For a clock input to be eligible to be switched through, it must be enabled using EN_CLKinX.
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8.3.2.3.1 Starting Active Clock
Upon programming this mode, the currently active clock remains active if PLL1 lock detect is high. To ensure a
particular clock input is the active clock when starting this mode, program CLKin_SELECT_MODE to the manual
mode which selects the desired clock input (CLKin0 or 1). Wait for PLL1 to lock PLL1_DLD = 1, then select this
mode with CLKin_SELECT_MODE = 4.
8.3.2.3.2 Clock Switch Event: PLL1 DLD
A loss of lock as indicated by PLL1’s DLD signal (PLL1_DLD = 0) will cause an input clock switch event if
DISABLE_DLD1_DET = 0. PLL1 DLD must go high (PLL1_DLD = 1) in between input clock switching events.
8.3.2.3.3 Clock Switch Event: PLL1 Vtune Rail
If Vtune_RAIL_DET_EN is set and the PLL1 Vtune voltage crosses the DAC high or low threshold, holdover
mode will be entered. Since PLL1_DLD = 0 in holdover a clock input switching event will occur.
8.3.2.3.4 Clock Switch Event with Holdover
Clock switch event with holdover enabled is recommended in this input clock switching mode. When an input
clock switch event occurs, holdover mode is entered and the active clock is set to the clock input defined by the
Status_CLKinX pins. When the new active clock meets the holdover exit conditions, holdover is exited and the
active clock will continue to be used as a reference until another input clock switch event. PLL1 DLD must go
high in between input clock switching events.
8.3.2.4 Input Clock Switching - Automatic Mode with Pin Select
When CLKin_SELECT_MODE is 6, the active clock is selected using the Status_CLKinX pins upon an input
clock switch event according to Table 2.
8.3.2.4.1 Starting Active Clock
Upon programming this mode, the currently active clock remains active if PLL1 lock detect is high. To ensure a
particular clock input is the active clock when starting this mode, program CLKin_SELECT_MODE to the manual
mode which selects the desired clock input (CLKin0 or 1). Wait for PLL1 to lock PLL1_DLD = 1, then select this
mode with CLKin_SELECT_MODE = 6.
8.3.2.4.2 Clock Switch Event: PLL1 DLD
An input clock switch event is generated by a loss of lock as indicated by PLL1's DLD signal (PLL1 DLD = 0).
8.3.2.4.3 Clock Switch Event: PLL1 Vtune Rail
If Vtune_RAIL_DET_EN is set and the PLL1 Vtune voltage crosses the DAC threshold, holdover mode will be
entered. Since PLL1_DLD = 0 in holdover, a clock input switching event will occur.
8.3.2.4.4 Clock Switch Event with Holdover
Clock switch event with holdover enabled is recommended in this input clock switching mode. When an input
clock switch event occurs, holdover mode is entered and the active clock is set to the clock input defined by the
Status_CLKinX pins. When the new active clock meets the holdover exit conditions, holdover is exited and the
active clock will continue to be used as a reference until another input clock switch event. PLL1 DLD must go
high in between input clock switching events."
Table 2. Active Clock Input - Auto Pin Mode
STATUS_CLKin1(1)
STATUS_CLKin0
ACTIVE CLOCK
CLKin0
X
1
0
1
0
0
CLKin1
Reserved
(1) The polarity of Status_CLKin1 and Status_CLKin0 input pins can be inverted with the CLKin_SEL_INV bit.
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8.3.3 Holdover Mode
Holdover mode causes PLL2 to stay locked on frequency with minimal frequency drift when an input clock
reference to PLL1 becomes invalid. While in holdover mode, the PLL1 charge pump is TRI-STATED and a fixed
tuning voltage is set on CPout1 to operate PLL1 in open loop.
8.3.3.1 Enable Holdover
Program HOLDOVER_MODE to enable holdover mode. Holdover mode can be manually enabled by
programming the FORCE_HOLDOVER bit.
The holdover mode can be set to operate in 2 different sub-modes.
•
•
Fixed CPout1 (EN_TRACK = 0 or 1, EN_MAN_DAC = 1).
Tracked CPout1 (EN_TRACK = 1, EN_MAN_DAC = 0).
–
Not valid when EN_VTUNE_RAIL_DET = 1.
Updates to the DAC value for the Tracked CPout1 sub-mode occurs at the rate of the PLL1 phase detector
frequency divided by DAC_CLK_DIV. These updates occur any time EN_TRACK = 1.
The DAC update rate should be programmed for <= 100 kHz to ensure DAC holdover accuracy.
When tracking is enabled the current voltage of DAC can be readback, see DAC_CNT.
8.3.3.2 Entering Holdover
The holdover mode is entered as described in Input Clock Switching. Typically this is because:
•
•
FORCE_HOLDOVER bit is set.
PLL1 loses lock according to PLL1_DLD, and
–
–
HOLDOVER_MODE = 2
DISABLE_DLD1_DET = 0
•
CPout1 voltage crosses DAC high or low threshold, and
–
–
–
–
–
–
–
HOLDOVER_MODE = 2
EN_VTUNE_RAIL_DET = 1
EN_TRACK = 1
DAC_HIGH_TRIP = User Value
DAC_LOW_TRIP = User Value
EN_MAN_DAC = 1
MAN_DAC = User Value
8.3.3.3 During Holdover
PLL1 is run in open loop mode.
•
•
•
•
•
PLL1 charge pump is set to TRI-STATE.
PLL1 DLD will be unasserted.
The HOLDOVER status is asserted
During holdover If PLL2 was locked prior to entry of holdover mode, PLL2 DLD will continue to be asserted.
CPout1 voltage will be set to:
–
–
a voltage set in the MAN_DAC register (fixed CPout1).
a voltage determined to be the last valid CPout1 voltage (tracked CPout1).
•
PLL1 DLD will attempt to lock with the active clock input.
The HOLDOVER status signal can be monitored on the Status_HOLDOVER or Status_LD pin by programming
the HOLDOVER_MUX or LD_MUX register to Holdover Status.
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8.3.3.4 Exiting Holdover
Holdover mode can be exited in one of two ways.
•
•
Manually, by programming the device from the host.
Automatically, By a clock operating within a specified ppm of the current PLL1 frequency on the active clock
input. See Input Clock Switching for more detail on which clock input is active.
To exit holdover by programming, set HOLDOVER_MODE = Disabled. HOLDOVER_MODE can then be re-
enabled by programming HOLDOVER_MODE = Enabled. Care should be taken to ensure that the active clock
upon exiting holdover is as expected, otherwise the CLKin_SELECT_MODE register may need to be re-
programmed.
8.3.3.5 Holdover Frequency Accuracy and DAC Performance
When in holdover mode, PLL1 runs in open loop and the DAC sets the CPout1 voltage. If Fixed CPout1 mode is
used, then the output of the DAC is a voltage dependant upon the MAN_DAC register. If Tracked CPout1 mode
is used, then the output of the DAC is the voltage at the CPout1 pin before holdover mode was entered. When
using Tracked mode and EN_MAN_DAC = 1, during holdover the DAC value is loaded with the programmed
value in MAN_DAC, not the tracked value.
When in Tracked CPout1 mode, the DAC has a worst case tracking error of ±2 LSBs once PLL1 tuning voltage is
acquired. The step size is approximately 3.2 mV; therefore, the VCXO frequency error during holdover mode
caused by the DAC tracking accuracy is ±6.4 mV × Kv, where Kv is the tuning sensitivity of the VCXO in use.
Therefore, the accuracy of the system when in holdover mode in ppm is:
± 6.4 mV × Kv × 1e6
VCXO Frequency
Holdover accuracy (ppm) =
(1)
Example: Consider a system with a 19.2-MHz clock input, a 153.6-MHz VCXO with a Kv of 17 kHz/V. The
accuracy of the system in holdover in ppm is:
±6.4 mV ´ 17 kHz / V ´ 1e6
±0.71ppm =
153.6 MHz
(2)
It is important to account for this frequency error when determining the allowable frequency error window to
cause holdover mode to exit.
8.3.3.6 Holdover Mode - Automatic Exit of Holdover
The LMK04208 device can be programmed to automatically exit holdover mode when the accuracy of the
frequency on the active clock input achieves a specified accuracy. The programmable variables include
PLL1_WND_SIZE and DLD_HOLD_CNT.
See Digital Lock Detect Frequency Accuracy to calculate the register values to cause holdover to automatically
exit upon reference signal recovery to within a user specified ppm error of the holdover frequency.
It is possible for the time to exit holdover to vary because the condition for automatic holdover exit is for the
reference and feedback signals to have a time/phase error less than a programmable value. Because it is
possible for two clock signals to be very close in frequency but not close in phase, it may take a long time for the
phases of the clocks to align themselves within the allowable time/phase error before holdover exits.
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8.3.4 PLLs
8.3.4.1 PLL1
The maximum phase detector frequency (fPD1) of PLL1 is 40 MHz. Since a narrow loop bandwidth should be
used for PLL1, the need to operate at high phase detector rate to lower the in-band phase noise becomes
unnecessary. The maximum values for the PLL1 R and N dividers is 16,383. Charge pump current ranges from
100 to 1600 µA. PLL1 N divider may be driven by OSCin port through the OSCout_MUX output (default) or by
internal or external feedback as selected by Feedback Mux in 0-delay mode.
Low charge pump currents and phase detector frequencies aid design of low loop bandwidth loop filters with
reasonably sized components to allow the VCXO or PLL2 to dominate phase noise inside of PLL2 loop
bandwidth. High charge pump currents may be used by PLL1 when using VCXOs with leaky tuning voltage
inputs to improve system performance.
8.3.4.2 PLL2
PLL2's maximum phase detector frequency (fPD2) is 155 MHz. Operating at highest possible phase detector rate
will ensure low in-band phase noise for PLL2 which in turn produces lower total jitter. The in-band phase noise
from the reference input and PLL is proportional to N2. The maximum value for the PLL2 R divider is 4,095. The
maximum value for the PLL2 N divider is 262,143. The N2 Prescaler in the total N feedback path can be
programmed for values 2 to 8 (all divides even and odd). Charge pump current ranges from 100 to 3200 µA.
High charge pump currents help to widen the PLL2 loop bandwidth to optimize PLL2 performance.
8.3.4.2.1 PLL2 Frequency Doubler
The PLL2 reference input at the OSCin port may be routed through a frequency doubler before the PLL2 R
Divider. The frequency doubler feature allows the phase comparison frequency to be increased when a relatively
low frequency oscillator is driving the OSCin port. By doubling the PLL2 phase detector frequency, the in-band
PLL2 noise is reduced by about 3 dB.
When using the doubler, PLL2 R Divider may be used to reduce the phase detector frequency to the limit of the
PLL2 maximum phase detector frequency.
For applications in which the OSCin frequency and PLL2 phase detector frequency are equal, the best PLL2 in-
band noise can be achieved when the doubler is enabled (EN_PLL2_REF_2X = 1) and the PLL2 R divide value
is 2. Do not use doubler disabled (EN_PLL2_REF_2X = 0) and PLL2 R divide value of 1.
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8.3.4.3 Digital Lock Detect
Both PLL1 and PLL2 support digital lock detect. Digital lock detect compares the phase between the reference
path (R) and the feedback path (N) of the PLL. When the time error, which is phase error, between the two
signals is less than a specified window size (ε) a lock detect count increments. When the lock detect count
reaches a user specified value lock detect is asserted true. Once digital lock detect is true, a single phase
comparison outside the specified window will cause digital lock detect to be asserted false. This is illustrated in
Figure 10.
The incremental lock detect count feature functions as a digital filter to ensure that lock detect is not asserted for
only a brief time when the phases of R and N are within the specified tolerance for only a brief time during initial
phase lock.
The digital lock detect signal can be monitored on the Status_LD or Status_Holdover pin. The pin may be
programmed to output the status of lock detect for PLL1, PLL2, or both PLL1 and PLL2.
See Digital Lock Detect Frequency Accuracy for more detailed information on programming the registers to
achieve a specified frequency accuracy in ppm with lock detect.
The digital lock detect feature can also be used with holdover to automatically exit holdover mode. See Holdover
Mode for more info.
NO
NO
PLLX
Lock Detected = False
Lock Count = 0
YES
YES
Increment
PLLX Lock Count
PLLX
Lock Detected = True
PLLX Lock Count =
PLLX_DLD_CNT
START
Phase Error < g
Phase Error < g
YES
NO
Figure 10. Digital Lock Detect Flowchart
8.3.5 Status Pins
The Status_LD, Status_HOLDOVER, Status_CLKin0, Status_CLKin1, and SYNC pins can be programmed to
output a variety of signals for indicating various statuses like digital lock detect, holdover, several DAC indicators,
and several PLL divider outputs.
8.3.5.1 Logic Low
This is a very simple output. In combination with the output _MUX register, this output can be toggled between
high and low. Useful to confirm MICROWIRE programming or as a general purpose IO.
8.3.5.2 Digital Lock Detect
PLL1 DLD, PLL2 DLD, and PLL1 + PLL2 are selectable on certain output pins. See Digital Lock Detect for more
information.
8.3.5.3 Holdover Status
Indicates if the device is in Holdover mode. See HOLDOVER_MODE for more information.
8.3.5.4 DAC
Various flags for the DAC can be monitored including DAC Locked, DAC Rail, DAC Low, and DAC High.
When the PLL1 tuning voltage crosses the low threshold, DAC Low is asserted. When PLL1 tuning voltage
crosses the high threshold, DAC High is asserted. When either DAC Low or DAC High is asserted, DAC Rail will
also be asserted.
DAC Locked is asserted when EN_Track = 1 and DAC is closely tracking the PLL1 tuning voltage.
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8.3.5.5 PLL Divider Outputs
The PLL divider outputs are useful for debugging failure to lock issues. It allows the user to measure the
frequency the PLL inputs are receiving. The settings of PLL1_R, PLL1_N, PLL2_R, and PLL2_N output pulses at
the phase detector rate. The settings of PLL1_R / 2, PLL1_N / 2, PLL2_R / 2, and PLL2_N / 2 output a 50% duty
cycle waveform at half the phase detector rate.
8.3.5.6 CLKinX_LOS
The clock input loss of signal indicator is asserted when LOS is enabled (EN_LOS) and the clock no longer
detects an input as defined by the time-out threshold, LOS_TIMEOUT.
8.3.5.7 CLKinX Selected
If this clock is the currently selected/active clock, this pin will be asserted.
8.3.5.8 MICROWIRE Readback
The readback data can be output on any pin programmable to readback mode. For more information on
readback see Readback.
8.3.6 VCO
The integrated VCO uses a frequency calibration routine when register R30 is programmed to lock VCO to target
frequency. Register R30 contains the PLL2_N register.
During the frequency calibration the PLL2_N_CAL value is used instead of PLL2_N, this allows 0-delay modes to
have a separate PLL2 N value for VCO frequency calibration and regular operation. See Register 29, Register
30, and PLL Programming for more information.
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8.3.7 Clock Distribution
8.3.7.1 Fixed Digital Delay
This section discussing Fixed Digital delay and associated registers is fundamental to understanding digital delay
and dynamic digital delay.
Clock outputs may be delayed or advanced from one another by up to 517.5 clock distribution path periods. By
programming a digital delay value from 4.5 to 522 clock distribution path periods, a relative clock output delay
from 0 to 517.5 periods is achieved. The CLKoutX_DDLY (5 to 522) and CLKoutX_HS (–0.5 or 0) registers set
the digital delay as shown in Table 3.
Table 3. Possible Digital Delay Values
CLKoutX_DDLY
CLKoutX_HS
DIGITAL DELAY
5
5
1
0
1
0
1
0
...
0
1
0
1
0
4.5
5
6
5.5
6
6
7
6.5
7
7
...
...
520
521
521
522
522
520
520.5
521
521.5
522
NOTE
Digital delay values only take effect during a SYNC event and if the NO_SYNC_CLKoutX
bit is cleared for this clock group. See Clock Output Synchronization (SYNC) for more
information.
The resolution of digital delay is determined by the frequency of the clock distribution path. The clock distribution
path is the output of Mode Mux1 (Functional Block Diagram). The best resolution of digital delay is achieved by
bypassing the VCO divider.
VCO_DIV
Digital Delay Resolution
(with VCO Divider)
=
2 × VCO Frequency
(3)
Digital Delay Resolution
(VCO Divider bypassed or external VCO)
1
=
2 × VCO Frequency
(4)
The digital delay between clock outputs can be dynamically adjusted with no or minimum disruption of the output
clocks. See Dynamically Programming Digital Delay for more information.
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8.3.7.2 Fixed Digital Delay - Example
Given a VCO frequency of 2949.12 MHz and no VCO divider, by using digital delay the outputs can be adjusted
in 1 / (2 * 2949.12 MHz) = ~169.54 ps steps.
To achieve quadrature (90 degree shift) between the 122.88-MHz outputs on CLKout4 and CLKout6 from a VCO
frequency of 2949.12 MHz and bypassing the VCO divider, consider the following:
1. The frequency of 122.88 MHz has a period of ~8.14 ns.
2. To delay 90 degrees of a 122.88-MHz clock period requires a ~2.03 ns delay.
3. Given a digital delay step of ~169.54 ps, this requires a digital delay value of 12 steps (2.03 ns / 169.54 ps =
12).
4. Since the 12 steps are half period steps, CLKout3_DDLY is programmed 6 full periods beyond 5 for a total of
11.
This result in the following programming:
•
•
•
Clock output dividers to 24. CLKout2_DIV = 24 and CLKout3_DIV = 24.
Set first clock digital delay value. CLKout2_DDLY = 5, CLKout2_HS = 0.
Set second 90 degree shifted clock digital delay value. CLKout3_DDLY = 11, CLKout3_HS = 0.
Table 4 shows some of the possible phase delays in degrees achievable in the above example.
Table 4. Relative Phase Shift from CLKout4 and 5 to CLKout6 and 7(1)
CLKout3_DDLY
CLKout3_HS
RELATIVE DIGITAL DELAY
DEGREES of 122.88 MHz
5
5
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
...
-0.5
0.0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
6.5
7.0
7.5
8.0
8.5
...
-7.5°
0°
6
7.5°
6
15.0°
22.5°
30.0°
37.5°
45.0°
52.5°
60.0°
67.5°
75.0°
82.5°
90.0°
97.5°
105.0°
112.5°
120.0°
127.5°
...
7
7
8
8
9
9
10
10
11
11
12
12
13
13
14
...
(1) CLKout2_DDLY = 5 and CLKout2_HS = 0
Figure 12 illustrates clock outputs programmed with different digital delay values during a SYNC event.
Refer to Dynamically Programming Digital Delay for more information on dynamically adjusting digital delay.
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8.3.7.3 Clock Output Synchronization (SYNC)
The purpose of the SYNC function is to synchronize the clock outputs with a fixed and known phase relationship
between each clock output selected for SYNC. SYNC can also be used to hold the outputs in a low or 0 state.
The NO_SYNC_CLKoutX bits can be set to disable synchronization for a clock group.
To enable SYNC, EN_SYNC must be set. See EN_SYNC, Enable Synchronization.
The digital delay value set by CLKoutX_DDLY takes effect only upon a SYNC event. The digital delay due to
CLKoutX_HS takes effect immediately upon programming. See Dynamically Programming Digital Delay for more
information on dynamically changing digital delay.
During a SYNC event, clock outputs driven by the VCO are not synchronized to clock outputs driven by OSCin.
OSCout is always driven by OSCin. CLKout3 or CLKout4 may be driven by OSCin depending on the
CLKoutX_OSCin_Sel bit value. While SYNC is asserted, NO_SYNC_CLKoutX operates normally for CLKout3 or
CLKout4 under all circumstances. SYNC operates normally for CLKout3 or CLKout4 when driven by VCO.
8.3.7.3.1 Effect of SYNC
When SYNC is asserted, the outputs to be synchronized are held in a logic low state. When SYNC is
unasserted, the clock outputs to be synchronized are activated and will transition to a high state simultaneously
with one another except where different digital delay values have been programmed.
Refer to Dynamically Programming Digital Delay for SYNC functionality when SYNC_QUAL = 1.
Table 5. Steady State Clock Output Condition Given Specified Inputs
SYNC_TYPE
SYNC_POL
_INV
SYNC PIN
CLOCK OUTPUT STATE
0,1,2 (Input)
0,1,2 (Input)
0
0
1
1
0
1
0
Active
Low
1
0
0,1,2 (Input)
Low
0,1,2 (Input)
1
Active
Active
Low
3, 4, 5, 6 (Output)
3, 4, 5, 6 (Output)
0 or 1
0 or 1
8.3.7.3.2 Methods of Generating SYNC
There are five methods to generate a SYNC event:
•
Manual:
–
–
Asserting the SYNC pin according to the polarity set by SYNC_POL_INV.
Toggling the SYNC_POL_INV bit though MICROWIRE will cause a SYNC to be asserted.
•
Automatic:
–
–
–
If PLL1_SYNC_DLD or PLL2_SYNC_DLD is set, the SYNC pin will be asserted while DLD (digital lock
detect) is false for PLL1 or PLL2 respectively.
Programming Register R30, which contains PLL2_N will generate a SYNC event when using the internal
VCO.
Programming Register R0 through R5 when SYNC_EN_AUTO = 1.
NOTE
Due to the speed of the clock distribution path (as fast as ~325 ps period) and the slow
slew rate of the SYNC, the exact VCO cycle at which the SYNC is asserted or unasserted
by the SYNC is undefined. The timing diagrams show a sharp transition of the SYNC to
clarify functionality.
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8.3.7.3.3 Avoiding Clock Output Interruption Due to Sync
Any CLKout groups that have their NO_SYNC_CLKoutX bits set will be unaffected by the SYNC event. It is
possible to perform SYNC operation with the NO_SYNC_CLKoutX bits cleared, then set the
a
NO_SYNC_CLKoutX bits so that the selected clocks will not be affected by a future SYNC. Future SYNC events
will not effect these clocks but will still cause the newly synchronized clocks to be re-synchronized using the
currently programmed digital delay values. When this happens, the phase relationship between the first group of
synchronized clocks and the second group of synchronized clocks will be undefined unless the SYNC pulse is
qualified by an output clock. See Dynamically Programming Digital Delay.
8.3.7.3.4 SYNC Timing
When discussing the timing of the SYNC function, one cycle refers to one period of the clock distribution path.
CLKoutX_DDLY &
CLKoutX_HS
6 cycles
6 cycles
Distribution
Path
SYNC
(SYNC_POL
_INV=1)
CLKout0
CLKout1
CLKout2
D
A
B
C
CLKout0_DIV = 1 (valid only for external VCO mode)
CLKout1_DIV = 2
CLKout2_DIV = 4
The digital delay for all clock outputs is 5
The digital delay half step for all clock outputs is 0
SYNC_QUAL = 0 (No qualification)
Figure 11. Clock Output Synchronization Using the SYNC Pin (Active Low)
Refer to Figure 11 during this discussion on the timing of SYNC. SYNC must be asserted for greater than one
clock cycle of the clock distribution path to latch the SYNC event. After SYNC is asserted, the SYNC event is
latched on the rising edge of the distribution path clock, at time A. After this event has been latched, the outputs
will not reflect the low state for 6 cycles, at time B. Due to the asynchronous nature of SYNC with respect to the
output clocks, it is possible that a glitch pulse could be created when the clock output goes low from the SYNC
event. This is shown by CLKout4 in Figure 11 and CLKout2 in Figure 12. See Relative Dynamic Digital Delay for
more information on synchronizing relative to an output clock to eliminate or minimize this glitch pulse.
After SYNC becomes unasserted the event is latched on the following rising edge of the distribution path clock,
time C. The clock outputs will rise at time D, coincident with a rising distribution clock edge that occurs after 6
cycles plus as many more cycles as programmed by the digital delay for that clock output. Therefore, the
soonest a clock output will become high is 11 cycles after the SYNC unassertion event registration, time C, when
the smallest digital delay value of 5 is set. If CLKoutX_HS = 1 and CLKoutX_DDLY = 5, then the clock output will
rise 10.5 cycles after SYNC is unassertion event registration.
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CLKoutX_Y_DDLY & CLKoutX_HS
6 cycles
Distribution
Path
4.5
cycles
2.5
cycles
6 cycles
1 cycle
SYNC
(SYNC_POL
_INV=1)
CLKout0
CLKout1
CLKout2
C
D
E F
A
B
CLKout0_DIV = 2, CLKout0_DDLY = 5
CLKout1_DIV = 4, CLKout1_DDLY = 7
CLKout2_DIV = 4, CLKout2_DDLY = 8
CLKout0_HS = 1
CLKout1_HS = 0
CLKout2_HS = 0
SYNC_QUAL = 0 (No qualification)
Figure 12. Clock Output Synchronization Using the SYNC Pin (Active Low)
Figure 12 illustrates the timing with different digital delays programmed.
•
•
•
•
Time A) SYNC assertion event is latched.
Time B) SYNC unassertion latched.
Time C) All outputs toggle and remain low. A glitch pulse can occur at this time as shown by CLKout2.
Time D) After 6 + 4.5 = 10.5 cycles CLKout0 rises. This is the shortest time from SYNC unassertion
registration to clock rising edge possible.
•
•
Time E) After 6 + 7 = 13 cycles CLKout1 rises. CLKout1 and CLKout2 are programmed for quadrature
operation.
Time F) After 6 + 8 = 14 cycles CLKout2 rises.
8.3.7.4 Dynamically Programming Digital Delay
To use dynamic digital delay synchronization qualification set SYNC_QUAL = 1. This causes the SYNC pulse to
be qualified by a clock output so that the SYNC event occurs after a specified time from a clock output transition.
This allows the relative adjustment of clock output phase in real-time with no or minimum interruption of clock
outputs. Hence the term "dynamic digital delay."
Note that changing the phase of a clock output requires momentarily altering in the rate of change of the clock
output phase and therefore by definition results in a frequency distortion of the signal.
Without qualifying the SYNC with an output clock, the newly synchronized clocks would have a random and
unknown digital delay (or phase) with respect to clock outputs not currently being synchronized.
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8.3.7.4.1 Absolute vs. Relative Dynamic Digital Delay
The clock used for qualification of SYNC is selected with the feedback mux (FEEDBACK_MUX).
If the clock selected by the feedback mux has its NO_SYNC_CLKoutX = 1, then an absolute dynamic digital
delay adjustment will be performed during a SYNC event and the digital delay of the feedback clock will not be
adjusted.
If the clock selected by the feedback mux has its NO_SYNC_CLKoutX = 0, then a self-referenced or relative
dynamic digital delay adjustment will be performed during a SYNC event and the digital delay of the feedback
clock will be adjusted.
Clocks with NO_SYNC_CLKoutX = 1 always operate without interruption.
8.3.7.4.2 Dynamic Digital Delay and 0-Delay Mode
When using a 0-delay mode absolute dynamic digital delay is recommended. Using relative dynamic digital
delay with a 0-delay mode may result in a momentary clock loss on the adjusted clock also being used for 0-
delay feedback that may result in PLL1 DLD becoming low. This may result in HOLDOVER mode being activated
depending upon device configuration.
8.3.7.4.3 SYNC and Minimum Step Size
The minimum step size adjustment for digital delay is half a clock distribution path cycle. This is achieved by
using the CLKoutX_HS bit. The CLKoutX_HS bit change effect is immediate without the need for SYNC. To shift
digital delay using CLKoutX_DDLY a SYNC signal must be generated for the change to take effect.
8.3.7.4.4 Programming Overview
To dynamically adjust the digital delay with respect to an existing clock output the device should be programmed
as follows:
•
•
•
•
•
Set SYNC_QUAL = 1 for clock output qualification.
Set CLKout2_PD = 0. Required for proper operation of SYNC_QUAL = 1.
Set EN_FEEDBACK_MUX = 1 to enable the feedback buffer.
Set FEEDBACK_MUX to the clock output that the newly synchronized clocks will be qualified by.
Set NO_SYNC_CLKoutX = 1 for the output clocks that will continue to operate during the SYNC event. There
is no interruption of output on these clocks.
–
If FEEDBACK_MUX selects a clock output with NO_SYNC_CLKoutX = 1, then absolute dynamic digital
delay is performed.
–
If FEEDBACK_MUX selects a clock output with NO_SYNC_CLKoutX = 0, then self-referenced or relative
dynamic digital delay is performed.
•
•
The SYNC_EN_AUTO bit may be set to cause a SYNC event to begin when register R0 to R5 is
programmed. The auto SYNC feature is a convenience since does not require the application to manually
assert SYNC by toggling the SYNC_POL_INV bit or the SYNC pin when changing digital delay. However,
under the following condition a special programming sequence is required if SYNC_EN_AUTO = 1:
–
The CLKoutX_DDLY value being set in the programmed register is 13 or more.
Under the following condition a SYNC_EN_AUTO must = 0:
–
If the application requires a digital delay resolution of half a clock distribution path cycle in relative
dynamic digital delay mode because the HS bit must be fixed per Table 6 for a qualifying clock.
8.3.7.4.5 Internal Dynamic Digital Delay Timing
To dynamically adjust digital delay a SYNC must occur. Once the SYNC is qualified by an output clock, 3 cycles
later an internal one shot pulse will occur. The width of the one shot pulse is 3 cycles. This internal one shot
pulse will cause the outputs to turn off and then back on with a fixed delay with respect to the falling edge of the
qualification clock. This allows for dynamic adjustments of digital delay with respect to an output clock.
The qualified SYNC timing is shown in Figure 13 for absolute dynamic digital delay and Figure 14 for relative
dynamic digital delay.
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8.3.7.4.6 Other Timing Requirements
When adjusting digital delay dynamically, the falling edge of the qualifying clock selected by the
FEEDBACK_MUX must coincide with the falling edge of the clock distribution path. For this requirement to be
met, program the CLKoutX_HS value of the qualifying clock group according to Table 6.
Table 6. Half Step Programming Requirement of Qualifying Clock During Sync Event
DISTRIBUTION PATH FREQUENCY
CLKoutX_DIV VALUE
CLKoutX_HS
Even
Odd
Must = 1 during SYNC event.
Must = 0 during SYNC event.
Must = 0 during SYNC event.
Must = 1 during SYNC event.
≥ 1.8 GHz
Even
Odd
< 1.8 GHz
8.3.7.5 Absolute Dynamic Digital Delay
Absolute dynamic digital delay can be used to program a clock output to a specific phase offset from another
clock output.
Pros:
•
•
Simple direct phase adjustment with respect to another clock output.
CLKoutX_HS will remain constant for qualifying clock.
–
Can easily use auto sync feature (SYNC_EN_AUTO = 1) when digital delay adjustment requires half step
digital delay requirements.
•
Can be used with 0-delay mode.
Cons:
For some phase adjustments there may be a glitch pulse due to SYNC assertion.
For example, see CLKout4 in Figure 11 and CLKout2 in Figure 12.
•
–
8.3.7.5.1 Absolute Dynamic Digital Delay - Example
To illustrate the absolute dynamic digital delay adjust procedure, consider the following example.
System Requirements:
•
•
•
•
•
VCO Frequency = 2949.12 MHz
CLKout0 = 983.04 MHz (CLKout0_DIV = 3)
CLKout1 = 491.52 MHz (CLKout1_DIV = 6)
CLKout2 = 245.76 MHz (CLKout2_DIV = 12)
For all clock outputs during initial programming:
–
–
–
CLKoutX_DDLY = 5
CLKoutX_HS = 1
NO_SYNC_CLKoutX = 0
The application requires the 491.52 MHz clock to be stepped in 30 degree steps (~169.5 ps), which is the
minimum step resolution allowable by the clock distribution path requiring use of the half step bit (CLKoutX_HS).
That is 1 / 2949.52 MHz / 2 = ~169.5 ps. During the stepping of the 491.52-MHz clock, the 983.04-MHz and
245.76-MHz clock must not be interrupted.
1. The device is programmed from register R0 to R30 with values that result in the device being locked and
operating as desired ( see the system requirements above). The phase of all the output clocks are aligned
because all the digital delay and half step values were the same when the SYNC was generated by
programming register R30. The timing of this is as shown in Figure 11.
2. Now the registers will be programmed to prepare for changing digital delay (or phase) dynamically.
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Table 7. Register Setup for Absolute Dynamic Digital Delay Example
REGISTER
PURPOSE
Use a clock output for qualifying the SYNC pulse for dynamically
adjusting digital delay.
SYNC_QUAL = 1
EN_SYNC = 1 (default)
CLKout2_PD = 0
Required for SYNC functionality.
Required when SYNC_QUAL = 1.
CLKout2 may be powered down or in use.
Enable the feedback mux for SYNC operation for dynamically
adjusting digital delay.
EN_FEEDBACK_MUX = 1
FEEDBACK_MUX = 2 (CLKout2)
NO_SYNC_CLKout0 = 1
Use the fixed 245.76-MHz clock as the SYNC qualification clock.
This clock output (983.04 MHz) won't be affected by SYNC. It will
always operate without interruption.
This clock output (245.76 MHz) won't be affected by SYNC. It will
always operate without interruption.
NO_SYNC_CLKout2 = 1
This clock will also be the qualifying clock in this example.
Since CLKout2 is the qualifying clock and CLKoutX_DIV is even, the
half step bit must be set to 1. See Table 6.
CLKout2_HS = 1
SYNC_EN_AUTO = 1
Automatic generation of SYNC is allowed for this case.
After the registers in Table 7 have been programmed, the application may now dynamically adjust the digital
delay of CLKout1 (491.52 MHz).
3. Adjust digital delay of CLKout1.
Refer to Table 8 for the programming values to set a specified phase offset from the absolute reference clock.
Table 8 is dependant upon the qualifying clock divide value of 12, refer to Calculating Dynamic Digital Delay
Values for Any Divide for information on creating tables for any divide value.
Table 8. Programming for Absolute Digital Delay Adjustment
DEGREES OF ADJUSTMENT FROM INITIAL 491.52 MHz PHASE
PROGRAMMING
CLKout1_DDLY = 7; CLKout1_HS = 1
±0 or ±360 degrees
30 degrees
60 degrees
90 degrees
120 degrees
150 degrees
180 degrees
210 degrees
240 degrees
270 degrees
300 degrees
330 degrees
–330 degrees
–300 degrees
–270 degrees
–240 degrees
–210 degrees
–180 degrees
–150 degrees
–120 degrees
–90 degrees
–60 degrees
–30 degrees
CLKout1_DDLY = 7; CLKout1_HS = 0
CLKout1_DDLY = 8; CLKout1_HS = 1
CLKout1_DDLY = 8; CLKout1_HS = 0
CLKout1_DDLY = 9; CLKout1_HS = 1
CLKout1_DDLY = 9; CLKout1_HS = 0
CLKout1_DDLY = 10; CLKout1_HS = 1
CLKout1_DDLY = 10; CLKout1_HS = 0
CLKout1_DDLY = 5; CLKout1_HS = 1
CLKout1_DDLY = 5; CLKout1_HS = 0
CLKout1_DDLY = 6; CLKout1_HS = 1
CLKout1_DDLY = 6; CLKout1_HS = 0
After setting the new digital delay values, the act of programming R1 will start a SYNC automatically because
SYNC_EN_AUTO = 1.
If the user elects to reduce the number of SYNCs because they are not required when only CLKout1_HS is set,
then SYNC_EN_AUTO is = 0 and the SYNC may now be generated by toggling the SYNC pin or by toggling the
SYNC_POL_INV bit. Because of the internal one shot pulse, no strict timing of the SYNC pin or SYNC_POL_INV
bit is required.
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After the SYNC event, the clock output will adjust according to Table 8. See Figure 13 for a detailed view of the
timing diagram. The timing diagram critical points are:
•
•
•
•
•
•
•
•
Time A) SYNC assertion event is latched.
Time B) First qualifying falling clock output edge.
Time C) Second qualifying falling clock output edge.
Time D) Internal one shot pulse begins. 5 cycles later clock outputs will be forced low
Time E) Internal one shot pulse ends. 5 cycles + digital delay cycles later the synced clock outputs rise.
Time F) Clock outputs are forced low. (CLKout2 is already low).
Time G) Beginning of digital delay cycles.
Time H) For CLKout1_DDLY = 6; the clock output rises now.
CLKoutX_Y_DDLY and
CLKoutX_Y_HS
5 cycles
Distribution
Path
3 cycles 3 cycles
5.5 cycles
5.5 cycles
SYNC
Internal One
Shot Pulse
CLKout0 /6
HS = 1
CLKout2 /6
HS = 1
CLKout4 /12
HS = 1
1
2
AB
C
D
E
F
G
H
Figure 13. Absolute Dynamic Digital Delay Programming Example
(SYNC_QUAL = 1, Qualify with Clock Output)
8.3.7.6 Relative Dynamic Digital Delay
Relative dynamic digital delay can be used to program a clock output to a specific phase offset from another
clock output.
Pros:
•
•
Simple direct phase adjustment with respect to same clock output.
The clock output will always behave the same during digital delay adjustment transient. For some divide
values there will be no glitch pulse.
Cons:
•
•
For some clock divide values there may be a glitch pulse due to SYNC assertion.
Adjustments of digital delay requiring the half step bit (CLKoutX_HS) for finer digital delay adjust is
complicated.
•
Use with 0-delay mode may result in PLL1 DLD becoming low and HOLDOVER mode becoming activated.
–
DISABLE_DLD1_DET can be set to prevent HOLDOVER from becoming activated due to PLL1 DLD
becoming low.
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8.3.7.6.1 Relative Dynamic Digital Delay - Example
To illustrate the relative dynamic digital delay adjust procedure, consider the following example.
System Requirements:
•
•
•
•
•
VCO Frequency = 2949.12 MHz
CLKout0 = 983.04 MHz (CLKout0_DIV = 3)
CLKout1 = 491.52 MHz (CLKout1_DIV = 6)
CLKout2 = 491.52 MHz (CLKout2_DIV = 6)
For all clock outputs during initial programming:
–
–
–
CLKoutX_DDLY = 5
CLKoutX_HS = 0
NO_SYNC_CLKoutX = 0
The application requires the 491.52-MHz clock to be stepped in 30 degree steps (~169.5 ps), which is the
minimum step resolution allowable by the clock distribution path. That is 1 / 2949.52 MHz / 2 = ~169.5 ps. During
the stepping of the 491.52 MHz clocks the 983.04 MHz clock must not be interrupted.
1. The device is programmed from register R0 to R30 with values that result in the device being locked and
operating as desired, see the system requirements above. The phase of all the output clocks are aligned
because all the digital delay and half step values were the same when the SYNC was generated by
programming register R30. The timing of this is as shown in Figure 11.
2. Now the registers will be programmed to prepare for changing digital delay (or phase) dynamically.
Table 9. Register Setup for Relative Dynamic Digital Delay Adjustment
REGISTER
SYNC_QUAL = 1
PURPOSE
Use clock output for qualifying the SYNC pulse for dynamically adjusting digital delay.
Required for SYNC functionality.
EN_SYNC = 1 (default)
Required when SYNC_QUAL = 1.
CLKout4 and/or CLKout5 outputs may be powered down or in use.
CLKout2_PD = 0
EN_FEEDBACK_MUX = 1
Enable the feedback mux for SYNC operation for dynamically adjusting digital delay.
Use the clock itself as the SYNC qualification clock.
FEEDBACK_MUX = 1 (CLKout1)
This clock output (983.04 MHz) won't be affected by SYNC. It will always operate without
interruption.
NO_SYNC_CLKout0 = 1
NO_SYNC_CLKout2 = 1
CLKout3’s phase is not to change with respect to CLKout0.
Automatic generation of SYNC is not allowed because of the half step requirement in relative
dynamic digital delay mode.
SYNC_EN_AUTO = 0 (default)
SYNC must be generated manually by toggling the SYNC_POL_INV bit or the SYNC pin.
After the above registers have been programmed, the application may now dynamically adjust the digital
delay of the 491.52 MHz clocks.
3. Adjust digital delay of CLKout1 by one step which is 30 degrees or ~169.5 ps.
Refer to Table 10 for the programming sequence to step one half clock distribution period forward or backwards.
Refer to Calculating Dynamic Digital Delay Values for Any Divide for more information on how to calculate digital
delay and half step values for other cases.
To fulfill the qualifying clock output half step requirement in Table 6 when dynamically adjusting digital delay, the
CLKoutX_HS bit must be cleared for clocks with even divides. So before any dynamic digital delay adjustment,
CLKoutX_HS must be clear because the clock divide value is even. To achieve the final required digital delay
adjustment, the CLKoutX_HS bit may set after SYNC.
40
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Table 10. Programming Sequence for One Step Adjust
STEP DIRECTION and CURRENT HS STATE
PROGRAMMING SEQUENCE
1. CLKout1_HS = 1.
Adjust clock output one step forward.
CLKout1_HS is 0.
1. CLKout1_DDLY = 9.
2. Perform SYNC event.
3. CLKout1_HS = 0.
Adjust clock output one step forward.
CLKout1_HS is 1.
1. CLKout1_HS = 1.
2. CLKout1_DDLY = 5.
3. Perform SYNC event.
Adjust clock output one step backward.
CLKout1_HS is 0.
Adjust clock output one step backward.
CLKout1_HS is 1.
1. CLKout1_HS = 0.
After programing the updated CLKout1_DDLY and CLKout1_HS values, perform a SYNC event. The SYNC may
be generated by toggling the SYNC pin or by toggling the SYNC_POL_INV bit. Because of the internal one shot
pulse, no strict timing of the SYNC pin or SYNC_POL_INV bit is required. After the SYNC event, the clock output
will be at the specified phase. See Figure 14 for a detailed view of the timing diagram. The timing diagram critical
points are:
•
•
•
•
•
•
•
•
Time A) SYNC assertion event is latched.
Time B) First qualifying falling clock output edge.
Time C) Second qualifying falling clock output edge.
Time D) Internal one shot pulse begins. 5 cycles later clock outputs will be forced low.
Time E) Internal one shot pulse ends. 5 cycles + digital delay cycles later the synced clock outputs rise.
Time F) Clock outputs are forced low. (CLKouts are already low).
Time G) Beginning of digital delay cycles.
Time H) For CLKout1_DDLY = 9; the clock output rises now.
CLKoutX_DDLY and
5 cycles
CLKoutX_HS
Distribution
Path
3 cycles 3 cycles
5.5 cycles
8.5 cycles
SYNC
Internal One
Shot Pulse
CLKout0 /3
HS = 1
CLKout1 /6
HS = 1
1
2
CLKout2 /6
HS = 1
AB
C
D
E
F
G
H
(SYNC_QUAL = 1, Qualify with clock output)
Starting condition is after half step is removed (CLKout1_HS = 0).
Figure 14. Relative Dynamic Digital Delay Programming Example, 2nd Adjust
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8.3.8 0-Delay Mode
When 0-delay mode is enabled the clock output selected by the Feedback Mux is connected to the PLL1 N
counter to ensure a fixed phase relationship between the selected CLKin and the fed back CLKout. When all the
clock outputs are synced together, all the clock outputs will share the same fixed phase relationship between the
selected CLKin and the fed back CLKout. The feedback can be internal or external using FBCLKin port.
When 0-delay mode is enabled the lowest frequency clock output is fed back to the Feedback Mux to ensure a
repeatable fixed CLKin to CLKout phase relationship between all clock outputs.
If a clock output that is not the lowest frequency output is selected for feedback, then clocks with lower
frequencies will have an unknown phase relationship with respect the other clocks and clock input. There will be
a number of possible phase relationships equal to Feedback_Clock_Frequency / Lower_Clock_Frequency that
may occur.
The Feedback Mux selects the even clock output of any clock group for internal feedback or the FBCLKin port for
external 0-delay feedback. The even clock can remain powered down as long as the CLKoutX_PD bit is = 0 for
its clock group.
To use 0-delay mode, the bit EN_FEEDBACK_MUX must be set (=1) to power up the feedback mux.
When using an external VCO mode, internal 0-delay feedback must be used since the FBCLKin port is shared
with the Fin input.
See PLL Programming for more information on programming the various PLL and output dividers for 0-delay
mode.
Table 11 outlines several registers to program for 0-delay mode.
Table 11. Programming 0-Delay Mode
REGISTER
PURPOSE
Select one of the 0-delay modes:
Dual PLL, Internal VCO
Dual PLL, External VCO
Single PLL, Internal VCO
MODE = 2, 5, or 8
EN_FEEDBACK_MUX = 1
Enable feedback mux.
FEEDBACK_MUX = Application Specific
Select CLKout or FBCLKin for 0-delay feedback.
The divide value of the clock selected by FEEDBACK_MUX is important for
PLL1_N or PLL2_N value calculation for Dual PLL or Single PLL mode
respectively.
CLKoutX_DIV
PLL1_N or PLL2_N value used with CLKoutX_DIV in loop for Dual PLL or Single
PLL mode respectively.
PLL1_N or PLL2_N
42
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8.4 Device Functional Modes
8.4.1 Mode Selection
The LMK04208 is capable of operating in several different modes as programmed by MODE: Device Mode.
Table 12. Device Mode Selection
MODE
R11[31:27]
PLL1
PLL2
PLL2 VCO
0-DELAY
CLOCK DIST
0
2
X
X
X
X
X
X
X
X
X
X
X
Internal
Internal
External
External
Internal
Internal
External
X
X
X
X
X
X
X
X
X
X
X
3
5
6
8
11
16
In addition to selecting the device's mode of operation above, some modes require additional configuration. Also
there are other features including holdover and dynamic digital delay that can also be enabled.
Table 13. Registers to Further Configure Device Mode of Operation
DYNAMIC DIGITAL
REGISTER
HOLDOVER
0-DELAY
DELAY
HOLDOVER_MODE
EN_TRACK
2
—
—
User
User
User
User
User
User
User
0
—
—
DAC_CLK_DIV
EN_MAN_DAC
—
—
—
—
DISABLE_DLD1_DET
EN_VTUNE_RAIL_DET
DAC_HIGH_TRIP
DAC_LOW_TRIP
FORCE_HOLDOVER
SYNC_EN_AUTO
SYNC_QUAL
—
—
—
—
—
—
—
—
—
—
—
—
User
—
—
1
EN_SYNC
—
—
1
CLKout2_PD
—
—
0
EN_FEEDBACK_MUX
FEEDBACK_MUX
NO_SYNC_CLKoutX
—
1
Feedback Clock
—
1
—
Qualifying Clock
User
—
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8.4.2 Operating Modes
The LMK04208 is a flexible device that can be configured for many different use cases. The following simplified
block diagrams help show the user the different use cases of the device.
8.4.2.1 Dual PLL
Figure 15 illustrates the typical use case of the LMK04208 in dual loop mode. In dual loop mode the reference to
PLL1 is either CLKin0 or CLKin1. An external VCXO or tunable crystal will be used to provide feedback for the
first PLL and a reference to the second PLL. This first PLL cleans the jitter with the VCXO or low cost tunable
crystal by using a narrow loop bandwidth. The VCXO or tunable crystal output may be buffered through the
OSCout port and optionally on up to 2 of the CLKouts. The VCXO or tunable crystal is used as the reference to
PLL2 and may be doubled using the frequency doubler. The internal VCO drives up to six divide/delay blocks
which drive 12 clock outputs.
Holdover functionality is optionally available when the input reference clock is lost. Holdover works by fixing the
tuning voltage of PLL1 to the VCXO or tunable crystal.
It is also possible to use an external VCO in place of PLL2's internal VCO.
PLL1
PLL2
External
Loop Filter
OSCout
External VCXO
or Tunable
Crystal
OSCout*
CLKinX
CLKinX*
R
N
Phase
Detector
PLL1
CPout2
External
Loop Filter
2 inputs
R
N
Phase
Detector
PLL2
Divider
Digital Delay
Analog Delay
Partially
Integrated
Loop Filter
CLKoutX
CLKoutX*
6 outputs
Input
Buffer
Internal
VCO
6 blocks
LMK04208
Figure 15. Simplified Functional Block Diagram for Dual Loop Mode
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8.4.2.2 0-Delay Dual PLL
Figure 16 and Figure 17 illustrate the use case of 0-delay dual loop mode. This configuration is very similar to
Dual PLL except that the feedback to the first PLL is driven by a clock output. 0-Delay causes one clock output to
have deterministic phase with respect to the clock input. Since all the clock outputs can be synchronized
together, all the clock outputs can be in phase with the clock input signal.
When the internal VCO is used, the feedback to PLL1 can be connected internally as shown or externally using
FBCLKin (CLKin1) as an input port. When an external VCO is used, the feedback to PLL1 must be connected
internally since the external VCO drives the Fin (CLKin1) port and thus precludes the use of external feedback
via FBCLKin.
It is also possible to use an external VCO in place of PLL2's internal VCO.
PLL1
PLL2
External
Loop Filter
OSCout
OSCout*
External VCXO
CLKinX
CLKinX*
R
N
Phase
Detector
PLL1
CPout2
External
Loop Filter
2 inputs
R
N
Phase
Detector
PLL2
Divider
Digital Delay
Analog Delay
Partially
Integrated
Loop Filter
CLKoutX
CLKoutX*
Input
Buffer
6 outputs
Internal
VCO
6 blocks
Internal or external loopback, user programmable
LMK04208
Figure 16. Simplified Functional Block Diagram for 0-Delay Dual Loop Mode with Internal VCO
PLL1
PLL2
External
VCO
External
Loop Filter
OSCout
OSCout*
External VCXO
CLKin0
CLKin0*
Fin
Fin*
R
N
Phase
Detector
PLL1
CPout2
External
Loop Filter
1 input
R
N
Phase
Detector
PLL2
Divider
Digital Delay
Analog Delay
Partially
Integrated
Loop Filter
CLKoutX
CLKoutX*
Input
Buffer
6 outputs
6 blocks
Internal loopback (no external loopback with External VCO)
LMK04208
Figure 17. Simplified Functional Block Diagram for 0-Delay Dual Loop Mode with External VCO
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8.4.2.3 Single PLL
Figure 18 illustrates the use case of single PLL mode. In single PLL mode only PLL2 is used and PLL1 is
powered down. OSCin is used as the reference input. The internal VCO drives up to 6 divide/delay blocks which
drive 12 clock outputs. The reference at OSCin can be used to the OSCout port. OSCin can also optionally drive
up to 2 of the clock outputs.
It is also possible to use an external VCO in place of PLL2's internal VCO.
PLL2
External
Loop Filter
OSCout
OSCout*
CPout2
OSCin
OSCin*
R
N
Phase
Detector
PLL2
Divider
Digital Delay
Analog Delay
Partially
Integrated
Loop Filter
CLKoutX
CLKoutX*
6 outputs
Internal
VCO
6 blocks
LMK04208
Figure 18. Simplified Functional Block Diagram for Single Loop Mode
8.4.2.4 0-Delay Single PLL
Figure 19 illustrates the use case of 0-delay single PLL mode. This configuration is very similar to Single PLL
except that the feedback to PLL2 comes from a clock output. This causes the clock outputs to be in phase with
the reference input. Since all the clock outputs can be synchronized together, all the clock outputs can be in
phase with the clock input signal. The feedback to PLL2 can be performed internally as shown, or externally
using FBCLKin (CLKin1) as an input port.
It is also possible to use an external VCO in place of PLL2's internal VCO.
PLL2
External
Loop Filter
OSCout
OSCout*
CPout2
OSCin
OSCin*
R
N
Phase
Detector
PLL2
Divider
Digital Delay
Analog Delay
Partially
Integrated
Loop Filter
CLKoutX
CLKoutX*
6 outputs
Internal
VCO
6 blocks
Internal or external loopback, user programmable
LMK04208
Figure 19. Simplified Functional Block Diagram for 0-Delay Single Loop Mode
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8.4.2.5 Clock Distribution
Figure 20 illustrates the LMK04208 used for clock distribution. CLKin1 is used to drive up to 6 divide/delay blocks
which drive 12 outputs. OSCin can be used to drive the OSCout port. OSCin can also optionally drive up to 2 of
the clock outputs.
Divider
Digital Delay
Analog Delay
CLKoutX
CLKoutX*
6 outputs
CLKin1
CLKin1*
6 blocks
OSCout
OSCin
OSCout*
OSCin*
LMK04208
Figure 20. Simplified Functional Block Diagram for Mode Clock Distribution
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8.5 Programming
LMK04208 devices are programmed using 32-bit registers. Each register consists of a 5-bit address field and 27-
bit data field. The address field is formed by bits 0 through 4 (LSBs) and the data field is formed by bits 5 through
31 (MSBs). The contents of each register is clocked in MSB first (bit 31), and the LSB (bit 0) last. During
programming, the LEuWire signal should be held low. The serial data is clocked in on the rising edge of the
CLKuWire signal. After the LSB (bit 0) is clocked in the LEuWire signal should be toggled low-to-high-to-low to
latch the contents into the register selected in the address field. TI recommends programming registers in
numeric order, for example R0 to R16, and R24 to R31 to achieve proper device operation. Figure 1 illustrates
the serial data timing sequence.
To achieve proper frequency calibration, the OSCin port must be driven with a valid signal before programming
register R30. Changes to PLL2 R divider or the OSCin port frequency require register R30 to be reloaded in
order to activate the frequency calibration process.
A slew rate of at least 30 V/us is recommended for MICROWIRE signals.
After programming is complete the CLKuWire, DATAuWire, and LEuWire signals should be returned to a low
state. If the CLKuWire or DATAuWire lines are toggled while the VCO is in lock, as is sometimes the case when
these lines are shared with other parts, the phase noise may be degraded during programming of the other
devices.
At no time should the MICROWIRE registers be programmed to any value other than what is specified in the
datasheet.
8.5.1 Special Programming Case for R0 to R5 for CLKoutX_DIV and CLKoutX_DDLY
In some cases when programming register R0 to R5 to change the CLKoutX_DIV divide value or
CLKoutX_DDLY delay value, 3 additional CLKuWire cycles must occur after loading the register for the newly
programmed divide or delay value to take effect. These special cases include:
•
•
When CLKoutX_DIV is > 25.
When CLKoutX_DDLY is > 12. Note: loading the digital delay value only prepares for a future SYNC event.
Also, since SYNC_EN_AUTO bit = 1 automatically generates a SYNC on the falling edge of LE when R0 to R5 is
programmed, further programming considerations must be made when SYNC_EN_AUTO = 1.
These special programming cases requiring the additional three clock cycles may be properly programmed by
one of the following methods shown in Table 14.
Table 14. R0 to R5 Special Case
CLKoutX_DIV and
CLKoutX_DDLY
SYNC_EN_AUTO
PROGRAMMING METHOD
No Additional Clocks Required (Normal)
CLKoutX_DIV ≤ 25 and
CLKoutX_DDLY ≤ 12
0 or 1
CLKoutX_DIV > 25 or
CLKoutX_DDLY > 12
0
1
Three Extra CLKuWire Clocks (Or program another register)
Three Extra CLKuWire Clocks while LEuWire is High
CLKoutX_DIV > 25 or
CLKoutX_DDLY > 12
•
•
•
Method: No Additional Clocks Required (Normal) No special consideration to CLKuWire is required when
changing divide value to ≤ 25, digital delay value to ≤ 12, or when the digital delay and divide value do not
change. See MICROWIRE timing Figure 1.
Method: Three Extra CLKuWire Clocks Three extra clocks must be provided before CLKoutX_DIV > 25 or
CLKoutX_DDLY > 12 take effect. See MICROWIRE timing Figure 2. Also, by programming another register
the three clock requirement can be satisfied.
Method: Three Extra CLKuWire Clocks with LEuWire Asserted When SYNC_EN_AUTO = 1 the falling
edge of LEuWire will generate a SYNC event. CLKoutX_DIV and CLKoutX_DDLY values must be updated
before the SYNC event occurs. So 3 CLKuWire rising edges must occur before LEuWire goes low. See
MICROWIRE timing Figure 3.
•
Initial Programming Sequence During the recommended programming sequence the device is programmed
in order from R0 to R31, so it is expected at least one additional register will be programmed after
programming the last CLKoutX_DIV or CLKoutX_DDLY value in R0 to R5. This will result in the extra needed
CLKuWire rising edges, so this special note is of little concern. If programming R0 to R5 to change CLKout
48
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frequency or digital delay or dynamic digital delay at a later time in the application, take care to provide these
extra CLKuWire cycles to properly load the new divide and/or delay values.
8.5.1.1 Example
In this example, all registers have been programmed, the PLLs are locked. An LMK04208 has been generating a
clock output frequency of 61.44 MHz on CLKout4 using a VCO frequency of 2949.12 MHz and a divide value of
48. SYNC_EN_AUTO = 0. At a later time the application requires a 30.72-MHz output on CLKout4. By
reprogramming register R4 with CLKout4_DIV = 96 twice, the divide value of 96 is set for clock output 4 which
results in an output frequency of 30.72 MHz (2949.12 MHz / 96 = 30.72 MHz) on CLKout4.
In this example, the required 3 CLKuWire cycles were achieved by reprogramming the R4 register with the same
value twice.
8.5.2 Recommended Programming Sequence
Registers are programmed in numeric order with R0 being the first and R31 being the last register programmed.
The recommended programming sequence involves programming R0 with the reset bit (b17) set to 1 to ensure
the device is in a default state. If R0 is programmed again, the reset bit must be cleared to 0 during the
programming of R0.
8.5.2.1 Programming Sequence Overview
•
•
Program R0 with RESET bit = 1. This ensures that the device is configured with default settings. When
RESET = 1, all other R0 bits are ignored.
–
If R0 is programmed again during the initial configuration of the device, the RESET bit must be cleared.
R0 through R5: CLKouts.
–
Program as necessary to configure the clock outputs, CLKout0 to CLKout5 as desired. These registers
configure clock output controls such as powerdown, digital delay and divider value, analog delay select,
and clock source select.
•
R6 through R8: CLKouts.
–
Program as necessary to configure the clock outputs, CLKout0 to CLKout5 as desired. These registers
configure the output format for each clock output and the analog delay for the clock output groups.
•
•
R9: Required programming
Program this register as shown in the register map for proper operation.
R10: OSCout, VCO divider, and 0-delay.
–
–
–
–
Enable and configure clock outputs OSCout.
Set and select VCO divider (VCO bypass is recommended).
Set 0-delay feedback source if used.
•
•
R11: Part mode, SYNC, and XTAL.
–
Program to configure the mode of the part, to configure SYNC functionality and pin, and to enable crystal
mode.
R12: Pins, SYNC, and holdover mode.
–
–
Status_LD pin, more SYNC options to generate a SYNC upon PLL1 and/or PLL2 lock detect.
Enable clock features such as holdover.
•
•
•
•
R13: Pins, holdover mode, and CLKins.
–
–
Status_HOLDOVER, Status_CLKin0, and Status_CLKin1 pin controls.
Enable clock inputs for use in specific part modes.
R14: Pins, LOS, CLKins, and DAC.
–
–
Status_CLKin1 pin control.
Loss of signal detection, CLKin type, DAC rail detect enable and high and low trip points.
R15: DAC and holdover mode.
–
–
Program to enable and set the manual DAC value.
HOLDOVER mode options.
R16: Crystal amplitude.
Increasing XTAL_LVL can improve tunable crystal phase noise performance.
–
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•
R24: PLL1 and PLL2.
–
–
PLL1 N and R delay and PLL1 digital lock delay value.
PLL2 integrated loop filter.
•
•
•
R25: DAC and PLL1.
Program to configure DAC update clock divider and PLL1 digital lock detect count.
R26: PLL2.
–
–
Program to configure PLL2 options.
R27: CLKins and PLL1.
–
–
Clock input pre-dividers.
Program to configure PLL1 options.
•
•
R28: PLL1 and PLL2.
Program to configure PLL2 R and PLL1 N.
R29: OSCin and PLL2.
–
–
Program to configure oscillator input frequency, PLL2 fast phase detector frequency mode, and PLL2 N
calibration value.
•
•
R30: PLL2.
–
Program to configure PLL2 prescaler and PLL2 N value.
R31: uWire lock.
–
Program to set the uWire_LOCK bit.
50
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8.5.3 Readback
Readback from the MICROWIRE programming registers is available. The MICROWIRE readback function can
be enabled on the Status_LD, Status_HOLDOVER, Status_CLKin0, Status_CLKin1, or SYNC pin by
programming the corresponding MUX register to “uWire Readback” and the corresponding TYPE register to
"Output (push-pull)." Power on reset defaults the Status_HOLDOVER pin to “uWire Readback.”
For timing specifications, see Timing Requirements. Figure 4 shows timing for LEuWire for both READBACK_LE
= 1 and 0. The rising edges of CLKuWire during MICROWIRE readback continue to clock data on DATAuWire
into the device during readback. If after the readback, LEuWire transitions from low to high, this data will be
latched to the decoded register. The decoded register address consists of the last 5 bits clocked on DATAuWire
as shown in Figure 4.
NOTE
For debug of the MICROWIRE interface, TI recommends simply programming an output
pin mux to active low and then toggle the output type register between output and
inverting output while observing the output pin for a low to high transition. For example, to
verify MICROWIRE programming, set the LD_MUX = 0 (Low) and then toggle the
LD_TYPE register between 3 (Output, push-pull) and 4 (Output inverted, push-pull). The
result will be that the Status_LD pin will toggle from low to high.
To perform a readback operation first set the register to be read back by programming the READBACK_ADDR
register. Then after any MICROWIRE write operation, with the LEuWire pin held low continue to clock the
CLKuWire pin. On every rising edge of the CLKuWire pin a new data bit is clocked onto the any pins
programmed for uWire Readback. If the READBACK_LE bit is set, the LEuWire pin should be left high after
LEuWire rising edge while continuing to clock the CLKuWire pin.
It is allowable to perform a register read back in the same MICROWIRE operation which set the
READBACK_ADDR register value.
Figure 4 illustrates the serial data timing sequence for a readback operation for both cases of READBACK_LE =
0 (POR default) and READBACK_LE = 1.
Data is clocked out MSB first. After 27 clocks all the data values will have been read and the read operation is
complete. If READBACK_LE = 1, the LEuWire line may now be lowered. It is allowable for the CLKuWire pin to
be clocked additional cycles, but the data on the readback pin will be invalid. CLKuWire must be low before the
falling edge of LEuWire.
8.5.3.1 Readback - Example
To readback register R3 perform the following steps:
•
•
•
•
Write R31 with READBACK_ADDR = 3; READBACK_LE = 0. DATAuWire and CLKuWire are toggled as
shown in Figure 1 with new data being clocked in on rising edges of CLKuWire
Toggle LEuWire high and then low as shown in Figure 1 and Figure 4. LEuWire is returned low because
READBACK_LE = 0.
Toggle CLKuWire high and then low 27 times to read back all 27 bits of register R3. Data is read MSB first.
Data is valid on falling edge of CLKuWire.
Read operation is complete.
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ZHCSFH1 –SEPTEMBER 2016
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8.6 Register Maps
8.6.1 Register Map and Readback Register Map
Table 15 provides the register map for device programming. Normally any register can be read from the same data address it is written to. However,
READBACK_LE has a different readback address. Also, the DAC_CNT register is a read only register. Table 16 shows the address for
READBACK_LE and DAC_CNT. Bits marked as reserved are undefined upon readback.
Observe that only the DATA bits are readback during a readback which can result in an offset of 5 bits between the two register tables.
Table 15. Register Map
REG-
ISTER
31
30
29
28
27
26
25
24
23
22
21
20
19
18
17
16
15
14
13
12
11
10
9
8
7
6
5
4
3
2
1
0
Data [26:0]
Address [4:0]
CLKout0
_ADLY_SEL
R0
0
CLKout0_DDLY [27:18]
CLKout1_DDLY [27:18]
CLKout2_DDLY [27:18]
CLKout0_DIV [15:5]
CLKout1_DIV [15:5]
CLKout2_DIV [15:5]
0
0
0
0
0
0
0
CLKout1
_ADLY_SEL
R1
R2
0
0
0
0
0
0
0
1
1
0
CLKout2
_ADLY_SEL
0
0
CLKout3
_ADLY_SEL
R3
CLKout3_DDLY [27:18]
CLKout3_DIV [15:5]
0
0
0
1
1
CLKout4
_ADLY_SEL
R4
CLKout4_DDLY [27:18]
CLKout5_DDLY [27:18]
0
CLKout4_DIV [15:5]
0
0
1
0
0
CLKout5
_ADLY_SEL
R5
R6
0
0
0
0
CLKout5_DIV [15:5]
0
0
0
0
1
1
0
1
1
0
CLKout1_ADLY
[15:11]
CLKout0_ADLY
[9:5]
0
0
0
CLKout1_TYPE [27:24]
CLKout0_TYPE [23:20]
0
0
0
0
52
Copyright © 2016, Texas Instruments Incorporated
LMK04208
www.ti.com.cn
ZHCSFH1 –SEPTEMBER 2016
Register Maps (continued)
Table 15. Register Map (continued)
REG-
ISTER
31
30
29
28
27
26
25
24
23
22
21
20
19
18
Data [26:0]
0
17
16
15
14
13
12
11
10
9
8
7
6
5
4
3
2
1
0
Address [4:0]
CLKout3_ADLY
[15:11]
CLKout2_ADLY
[9:5]
R7
0
0
0
0
CLKout3_TYPE [27:24]
CLKout5_TYPE [27:24]
CLKout2_TYPE [23:20]
0
0
0
0
0
0
1
1
1
CLKout5_ADLY
[15:11]
CLKout4_ADLY
[9:5]
R8
R9
0
0
0
0
0
0
0
0
0
0
0
0
CLKout4_TYPE [19:16]
0
0
0
1
1
0
0
0
0
0
1
1
1
0
1
0
1
1
1
0
1
0
1
0
0
1
0
1
0
1
0
1
0
1
0
OSCout_DIV
[18:16]
VCO_DIV
[10:8]
FEEDBACK
_MUX [7:5]
R10
0
0
0
1
OSCout_TYPE [27:24]
0
0
1
0
0
1
0
1
0
SYNC_MUX
[19:18]
SYNC_TYPE
[14:12]
R11
R12
MODE [31:27]
0
0
0
0
0
0
0
0
0
0
0
1
1
0
1
1
0
1
0
HOLDOVER
_MODE
[7:6]
0
LD_MUX [31:27]
LD_TYPE [26:24]
0
1
1
0
0
0
0
0
0
1
(1)
Status_
CLKin1
_MUX
Status_
CLKin0
_TYPE
[18:16]
Status_
CLKin0
_MUX
CLKin
_Select
_MODE
[11:8]
HOLDOVER
_TYPE
[26:24]
HOLDOVER_MUX
[31:27]
R13
0
0
0
0
1
1
0
1
[22:20]
[14:12]
Status_
CLKin1
_TYPE
[26:24]
LOS_
TIMEOUT
[31:30]
DAC_HIGH_TRIP
[19:14]
DAC_LOW_TRIP
[11:6]
R14
0
0
0
0
0
0
0
1
1
1
0
(1) Although the value of 0 is written here, during readback the value of READBACK_LE will be read at this location. See Register Map and Readback Register Map.
Copyright © 2016, Texas Instruments Incorporated
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LMK04208
ZHCSFH1 –SEPTEMBER 2016
www.ti.com.cn
Register Maps (continued)
Table 15. Register Map (continued)
REG-
ISTER
31
30
29
28
27
26
25
24
23
22
21
20
19
18
17
16
15
14
13
12
11
10
9
8
7
6
5
4
3
2
1
0
Data [26:0]
Address [4:0]
MAN_DAC
[31:22]
HOLDOVER_DLD_CNT
[19:6]
R15
R16
0
0
0
1
1
1
0
1
1
0
XTAL_
LVL
0
0
0
0
0
1
0
0
1
1
0
0
1
0
1
0
0
0
0
0
0
0
1
0
0
0
0
0
0
0
PLL1_
PLL2_C4_LF
[31:28]
PLL2_C3_LF
[27:24]
PLL2_R4_LF
[22:20]
PLL2_R3_LF
[18:16]
PLL1_N_DLY
[14:12]
PLL1_R_DLY
[10:8]
WND_
SIZE
[7:6]
R24
R25
0
0
1
1
1
1
0
0
0
0
0
1
DAC_CLK_DIV [31:22]
0
0
PLL1_DLD_CNT [19:6]
PLL2_
WND_SIZE
[31:30]
PLL2_CP
_GAIN
[27:26]
PLL2_DLD_CNT
[19:6]
R26
1
1
1
0
1
0
1
1
0
1
0
PLL1_CP
_GAIN
[27:26]
CLKin1_
PreR_DIV
[23: 22]
CLKin0_
PreR_DIV
[21: 20]
PLL1_R
[19:6]
R27
R28
0
0
0
0
0
1
1
1
1
0
1
1
0
1
0
PLL2_R [31: 20]
PLL1_N [19:6]
PLL2_N_CAL [22:5]
PLL2_N [22:5]
0
OSCin_FREQ
[26:24]
R29
0
0
0
0
0
1
1
1
0
1
R30
R31
0
0
0
0
0
0
0
0
0
0
PLL2_P [26:24]
0
1
1
1
1
1
1
1
1
0
1
0
0
0
0
0
READBACK_ADDR [20:16]
0
0
0
0
0
0
0
0
0
0
54
Copyright © 2016, Texas Instruments Incorporated
LMK04208
www.ti.com.cn
REG-
ZHCSFH1 –SEPTEMBER 2016
Table 16. Readback Register Map
RD
26
RD
25
RD
24
RD
23
RD
22
RD
21
RD
20
RD
19
RD
18
RD
17
RD
16
RD
15
RD
14
RD
13
RD
12
RD
11
RD
10
RD
9
RD
8
RD
7
RD
6
RD
5
RD
4
RD
3
RD
2
RD
1
RD
0
ISTE
R
Data [26:0]
HOLDOVE
R_
MODE
[2:1]
RD
R12
LD_MUX [26:22]
LD_TYPE [21:19]
0
1
1
0
0
0
0
0
0
0
0
0
1
RD
R23
RESERVED
[26:24]
DAC_CNT [23:14]
RESERVED [13:0]
RD
R31
RESERVED [26:10]
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ZHCSFH1 –SEPTEMBER 2016
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8.6.2 Default Device Register Settings After Power On Reset
Table 17 illustrates the default register settings programmed in silicon for the LMK04208 after power on or
asserting the reset bit. Capital X and Y represent numeric values.
Table 17. Default Device Register Settings after Power On/Reset
DEFAULT
VALUE
(DECIMAL)
BIT
LOCATION
(MSB:LSB)
FIELD NAME
DEFAULT STATE
FIELD DESCRIPTION
REGISTER
CLKout0_PD
1
1
1
0
0
1
1
0
0
0
0
PD
PD
R0
R1
CLKout1_PD
CLKout2_PD
PD
R2
Powerdown control for analog and digital delay,
divider, and both output buffers
31
CLKout3_PD
Normal
Normal
PD
R3
CLKout4_PD
R4
CLKout5_PD
R5
CLKout3_OSCin_Sel
CLKout4_OSCin_Sel
CLKoutX_ADLY_SEL
CLKoutX_DDLY
RESET
OSCin
VCO
R3
30
30
Selects the clock source for a clock group from
internal VCO or external OSCin
R4
None
5
Add analog delay for clock output
Digital delay value
R0 to R5
R0 to R5
R0
28:29 [2]
27:18 [10]
17
Not in reset
Performs power on reset for device
Disabled
(device is active)
POWERDOWN
0
Device power down control
Half shift for digital delay
R1
17
16
CLKoutX_HS
CLKout0_DIV
CLKout1_DIV
CLKout2_DIV
CLKout3_DIV
CLKout4_DIV
CLKout5_DIV
CLKout1_TYPE
0
No shift
R0 to R5
R0
25
25
25
1
Divide-by-25
Divide-by-25
Divide-by-25
Divide-by-1
Divide-by-25
Divide-by-25
Powerdown
R1
R2
Divide for clock outputs
15:5 [11]
27:24 [4]
R3
25
25
0
R4
R5
R6
LVCMOS
(Norm/Norm)
CLKout3_TYPE
8
R7
CLKout5_TYPE
CLKout0_TYPE
0
0
Powerdown
Powerdown
R8
R6
R7
Individual clock output format. Select from
LVDS/LVPECL/LVCMOS.
23:20 [4]
19:16 [4]
CLKout2_TYPE
0
Powerdown
R8
CLKout4_TYPE
CLKoutX_ADLY
OSCout_TYPE
EN_OSCout
1
0
1
1
0
LVDS
No delay
LVDS
Analog delay setting for clock group
OSCout default clock output
R6 to R8
R10
15:11, 9:5 [5]
27:24 [4]
22
Enabled
Enable OSCout output buffer
Select OSCout divider or bypass
R10
OSCout_MUX
Bypass Divider
R10
20
Allows OSCin to be powered down. For use in clock
distribution mode.
PD_OSCin
0
OSCin powered
R10
19
OSCout_DIV
VCO_MUX
0
0
0
2
0
0
Divide-by-8
VCO
OSCout divider value
R10
R10
R10
R10
R10
R11
18:16 [3]
12
Select VCO or VCO Divider output
Feedback MUX is powered down.
VCO Divide value
EN_FEEDBACK_MUX
VCO_DIV
Disabled
11
Divide-by-2
CLKout0
10:8 [3]
7:5 [3]
31:27 [5]
FEEDBACK_MUX
MODE
Selects CLKout to feedback into the PLL1 N divider
Device mode
Internal VCO
56
Copyright © 2016, Texas Instruments Incorporated
LMK04208
www.ti.com.cn
ZHCSFH1 –SEPTEMBER 2016
Table 17. Default Device Register Settings after Power On/Reset (continued)
DEFAULT
VALUE
(DECIMAL)
BIT
LOCATION
(MSB:LSB)
FIELD NAME
DEFAULT STATE
FIELD DESCRIPTION
REGISTER
EN_SYNC
1
0
1
1
0
0
0
0
Enabled
Will sync
Enables synchronization circuitry.
R11
R11
R11
R11
R11
R11
R11
R11
26
25
NO_SYNC_CLKout5
NO_SYNC_CLKout4
NO_SYNC_CLKout3
NO_SYNC_CLKout2
NO_SYNC_CLKout1
NO_SYNC_CLKout0
SYNC_MUX
Will not sync
Will not sync
Will sync
24
23
Disable individual clock groups from becoming
synchronized.
22
Will sync
21
Will sync
20
Logic Low
Mux controlling SYNC pin when set to output
19:18 [2]
Allows SYNC operations to be qualified by a clock
output.
SYNC_QUAL
0
1
0
Not qualified
Logic Low
Manual
R11
R11
R11
17
16
15
SYNC_POL_INV
SYNC_EN_AUTO
Sets the polarity of the SYNC pin when input
SYNC is not started by programming a register R0 to
R5.
Input w/
Pull-up
SYNC_TYPE
1
SYNC IO pin type
R11
14:12 [3]
EN_PLL2_XTAL
LD_MUX
0
3
Disabled
Enable Crystal oscillator for OSCin
R11
R12
5
PLL1 and 2 DLD Lock detect mux selection when output
31:27 [5]
Output
LD_TYPE
3
LD IO pin type
(Push-Pull)
R12
26:24 [3]
SYNC_PLL2_DLD
SYNC_PLL1_DLD
EN_TRACK
0
0
1
2
7
Normal
Normal
Force synchronization mode until PLL2 locks
Force synchronization mode until PLL1 locks
R12
R12
R12
R12
R13
23
22
Enable Tracking DAC tracking of the PLL1 tuning voltage
Enable Holdover Causes holdover to activate when lock is lost
uWire Readback Holdover mux selection
8
HOLDOVER_MODE
HOLDOVER_MUX
7:6 [2]
31:27 [5]
Output
HOLDOVER_TYPE
3
HOLDOVER IO pin type
(Push-Pull)
R13
26:24 [3]
Status_CLKin1_MUX
Status_CLKin0_TYPE
0
2
Logic Low
Status_CLKin1 pin MUX selection
R13
R13
22:20 [3]
18:16 [3]
Input w/ Pull-down Status_CLKin0 IO pin type
Disables PLL1 DLD falling edge from causing
DISABLE_DLD1_DET
0
Not Disabled
R13
15
HOLDOVER mode to be entered
Status_CLKin0 pin MUX selection
Mode to use in determining reference CLKin for PLL1
Invert Status 0 and 1 pin polarity for input(1)
Set CLKin1 to be usable
Status_CLKin0_MUX
CLKin_SELECT_MODE
CLKin_Sel_INV
EN_CLKin1
0
3
0
1
1
0
1
2
0
0
Logic Low
Manual Select
Active High
Usable
R13
R13
R13
R13
R13
R14
R14
R14
R14
R14
14:12 [3]
11:9 [3]
8
6
EN_CLKin0
Usable
Set CLKin0 to be usable
5
31:30 [2]
28
LOS_TIMEOUT
EN_LOS
1200 ns, 420 kHz Time until no activity on CLKin asserts LOS
Enabled Loss of Signal Detect at CLKin
Input w/ Pull-down Status_CLKin1 pin IO pin type
Status_CLKin1_TYPE
CLKin1_BUF_TYPE
CLKin0_BUF_TYPE
26:24 [3]
21
Bipolar
Bipolar
CLKin1 Buffer Type
CLKin0 Buffer Type
20
(1) Inversion for Status 0 and 1 pins is only valid for CLKin_SELECT_MODE = 0x06
Copyright © 2016, Texas Instruments Incorporated
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ZHCSFH1 –SEPTEMBER 2016
www.ti.com.cn
Table 17. Default Device Register Settings after Power On/Reset (continued)
DEFAULT
VALUE
(DECIMAL)
BIT
LOCATION
(MSB:LSB)
FIELD NAME
DEFAULT STATE
FIELD DESCRIPTION
REGISTER
Voltage from Vcc at which holdover mode is entered if
EN_VTUNE_RAIL_DAC is enabled.
DAC_HIGH_TRIP
0
0
0
~50 mV from Vcc
~50 mV from GND
Disabled
R14
R14
R14
19:14 [6]
11:6 [6]
5
Voltage from GND at which holdover mode is entered
if EN_VTUNE_RAIL_DAC is enabled.
DAC_LOW_TRIP
Enable PLL1 unlock state when DAC trip points are
achieved
EN_VTUNE_RAIL_DET
Writing to this register will set the value for DAC when
in manual override.
MAN_DAC
512
3 V / 2
R15
31:22 [10]
Readback from this register is DAC value.
EN_MAN_DAC
0
Disabled
Set manual DAC override
R15
R15
20
Lock must be valid n many clocks of PLL1 PDF before
holdover mode is exited.
HOLDOVER_DLD_CNT
512
512 counts
19:6 [14]
Holdover not
forced
FORCE_HOLDOVER
0
Forces holdover mode.
R15
5
XTAL_LVL
0
0
0
0
0
1.65 Vpp
10 pF
Sets drive power level of Crystal
PLL2 integrated capacitor C4 value
PLL2 integrated capacitor C3 value
PLL2 integrated resistor R4 value
PLL2 integrated resistor R3 value
R16
R24
R24
R24
R24
31:30 [2]
31:28 [4]
27:24 [4]
22:20 [3]
18:16 [3]
PLL2_C4_LF
PLL2_C3_LF
PLL2_R4_LF
PLL2_R3_LF
10 pF
200 Ω
200 Ω
Delay in PLL1 feedback path to decrease lag from
input to output
PLL1_N_DLY
0
No delay
R24
14:12 [3]
Delay in PLL1 reference path to increase lag from
input to output
PLL1_R_DLY
0
3
4
No delay
40 ns
R24
R24
R25
10:8 [3]
7:6 [2]
PLL1_WND_SIZE
DAC_CLK_DIV
Window size used for digital lock detect for PLL1
DAC update clock divisor. Divides PLL1 phase
detector frequency.
Divide-by-4
31:22 [10]
Lock must be valid n many cycles before LD is
asserted
PLL1_DLD_CNT
PLL2_WND_SIZE
1024
0
1024 cycles
R25
R26
19:6 [14]
31:30 [2]
Reserved
Window size used for digital lock detect for PLL2
(2)
EN_PLL2_REF_2X
PLL2_CP_POL
0
0
3
Disabled, 1x
Negative
3.2 mA
Doubles reference frequency of PLL2.
Polarity of PLL2 Charge Pump
PLL2 Charge Pump Gain
R26
R26
R26
29
28
PLL2_CP_GAIN
27:26 [2]
Number of PDF cycles which phase error must be
within DLD window before LD state is asserted.
PLL2_DLD_CNT
8192
8192 Counts
R26
19:6 [14]
PLL2_CP_TRI
PLL1_CP_POL
PLL1_CP_GAIN
CLKin1_PreR_DIV
CLKin0_PreR_DIV
PLL1_R
0
1
Active
Positive
PLL2 Charge Pump Active
R26
R27
R27
R27
R27
R27
R27
R28
R28
R29
5
Polarity of PLL1 Charge Pump
PLL1 Charge Pump Gain
28
0
100 uA
27:26 [2]
23:22 [2]
21:20 [2]
19:6 [14]
5
0
Divide-by-1
Divide-by-1
Divide-by-96
Active
CLKin1 Pre-R divide value (1, 2, 4, or 8)
CLKin0 Pre-R divide value (1, 2, 4, or 8)
PLL1 R Divider (1 to 16383)
PLL1 Charge Pump Active
0
96
0
PLL1_CP_TRI
PLL2_R
4
Divide-by-4
Divide-by-192
448 to 511 MHz
PLL2 R Divider (1 to 4095)
31:20 [12]
19:6 [14]
26:24 [3]
PLL1_N
192
7
PLL1 N Divider (1 to 16383)
OSCin frequency range
OSCin_FREQ
PLL2 PDF > 100 When set, PLL2 PDF of greater than 100 MHz may be
PLL2_FAST_PDF
PLL2_N_CAL
1
R29
R29
23
MHz
used
Actual PLL2 N divider value used in calibration
routine.
48
Divide-by-48
22:5 [18]
PLL2_P
PLL2_N
2
Divide-by-2
PLL2 N Divider Prescaler (2 to 8)
PLL2 N Divider (1 to 262143)
R30
R30
26:24 [3]
22:5 [18]
48
Divide-by-48
(2) This register must be reprogrammed to a value of 2 (3.7 ns) during user programming.
58
Copyright © 2016, Texas Instruments Incorporated
LMK04208
www.ti.com.cn
ZHCSFH1 –SEPTEMBER 2016
Table 17. Default Device Register Settings after Power On/Reset (continued)
DEFAULT
VALUE
(DECIMAL)
BIT
LOCATION
(MSB:LSB)
FIELD NAME
DEFAULT STATE
FIELD DESCRIPTION
REGISTER
LEuWire Low for
Readback
READBACK_LE
0
State LEuWire pin must be in for readback
R31
21
READBACK_ADDR
uWire_LOCK
31
0
Register 31
Writable
Register to read back
R31
R31
20:16 [5]
5
The values of registers R0 to R30 are lockable
Copyright © 2016, Texas Instruments Incorporated
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LMK04208
ZHCSFH1 –SEPTEMBER 2016
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8.6.3 Register Descriptions
8.6.3.1 Registers R0 to R5
Registers R0 through R5 control the 12 clock outputs CLKout0 to CLKout5. Register R0 controls CLKout0 and
CLKout1, Register R1 controls CLKout2 and so on. All functions of the bits in these six registers are identical
except the different registers control different clock outputs.
The RESET bit is only in register R0.
The POWERDOWN bit is only in register R1.
The CLKoutX_OSCin_Sel bit is only in registers R3 and R4.
8.6.3.1.1 CLKoutX_PD, Powerdown CLKoutX Output Path
This bit powers down the clock as specified by CLKoutX. This includes the divider, digital delay, analog delay,
and output buffers.
Table 18. CLKoutX_PD
R0 to R5[31]
STATE
0
1
Power up clock group
Power down clock group
8.6.3.1.2 CLKoutX_OSCin_Sel, Clock Group Source
This bit sets the source for the clock output CLKoutX. The selected source will be either from a VCO via Mode
Mux1 or from the OSCin buffer.
This bit is valid only for registers R3 and R4, clock groups CLKout3 and CLKout4 respectively. All other clock
output groups are driven by a VCO via Mode Mux1.
Table 19. CLKoutX_OSCin_Sel
R3 to R4[30]
CLOCK GROUP SOURCE
0
1
VCO
OSCin
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8.6.3.1.3 CLKoutX_ADLY_SEL, Select Analog Delay
These bits individually select the analog delay block (CLKoutX_ADLY) for use with CLKoutX. Analog delay is
powered down when not selected. Analog delay may not operate at frequencies above the minimum-ensured
maximum output frequency of 1536 MHz.
Table 20. CLKoutX_ADLY_SEL
R0 to R5[28:29]
0 (0x00)
DEFINITION
Analog delay powered down
Reserved
1 (0x01)
2 (0x02)
Reserved
3 (0x03)
Analog delay selected
8.6.3.1.4 CLKoutX_DDLY, Clock Channel Digital Delay
CLKoutX_DDLY and CLKoutX_HS set the digital delay used for CLKoutX and CLKoutY. This value only takes
effect during a SYNC event and if the NO_SYNC_CLKoutX bit is cleared for this clock group. See Clock Output
Synchronization (SYNC).
Programming CLKoutX_DDLY can require special attention. See Special Programming Case for R0 to R5 for
CLKoutX_DIV and CLKoutX_DDLY for more details.
Using a CLKoutX_DDLY value of 13 or greater will cause the clock group to operate in extended mode
regardless of the clock group's divide value or the half step value.
One clock cycle is equal to the period of the clock distribution path. The period of the clock distribution path is
equal to VCO Divider value divided by the frequency of the VCO. If the VCO divider is disabled or an external
VCO is used, the VCO divide value is treated as 1.
tclock distribution path = VCO divide value / fVCO
Table 21. CLKoutX_DDLY, 10 Bits
R0 to R5[27:18]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
4 (0x04)
5 (0x05)
6 (0x06)
7 (0x07)
...
DELAY
5 clock cycles
5 clock cycles
5 clock cycles
5 clock cycles
5 clock cycles
5 clock cycles
6 clock cycles
7 clock cycles
...
POWER MODE
Normal Mode
12 (0x0C)
13 (0x0D)
...
12 clock cycles
13 clock cycles
...
520 (0x208)
521 (0x209)
522 (0x20A)
520 clock cycles
521 clock cycles
522 clock cycles
Extended Mode
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8.6.3.1.5 Reset
The RESET bit is located in register R0 only. Setting this bit will cause the silicon default values to be loaded.
When programming register R0 with the RESET bit set, all other programmed values are ignored. After resetting
the device, the register R0 must be programmed again (with RESET = 0) to set non-default values in register R0.
The reset occurs on the falling edge of the LEuWire pin which loaded R0 with RESET = 1.
The RESET bit is automatically cleared upon writing any other register. For instance, when R0 is written to again
with default values.
Table 22. RESET
R0[17]
STATE
0
1
Normal operation
Reset (automatically cleared)
8.6.3.1.6 POWERDOWN
The POWERDOWN bit is located in register R1 only. Setting the bit causes the device to enter powerdown
mode. Normal operation is resumed by clearing this bit via MICROWIRE.
Table 23. POWERDOWN
R1[17]
STATE
0
1
Normal operation
Powerdown
8.6.3.1.7 CLKoutX_HS, Digital Delay Half Shift
This bit subtracts a half clock cycle of the clock distribution path period to the digital delay of CLKoutX and
CLKoutY. CLKoutX_HS is used together with CLKoutX_DDLY to set the digital delay value.
When changing CLKoutX_HS, the digital delay immediately takes effect without a SYNC event.
Table 24. CLKoutX_HS
R0 to R5[16]
STATE
0
Normal
Subtract half of a clock distribution path period from the total digital
delay
1
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8.6.3.1.8 CLKoutX_DIV, Clock Output Divide
CLKoutX_DIV sets the divide value for the clock group. The divide may be even or odd. Both even and odd
divides output a 50% duty cycle clock.
Using a divide value of 26 or greater will cause the clock group to operate in extended mode regardless of the
clock group's digital delay value.
Programming CLKoutX_DIV can require special attention. See section Special Programming Case for R0 to R5
for CLKoutX_DIV and CLKoutX_DDLY for more details.
Table 25. CLKoutX_DIV, 11 Bits
R0 to R5[15:5]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
4 (0x04)
5 (0x05)
6 (0x06)
...
DIVIDE VALUE
POWER MODE
Reserved
(1)
1
(2)
2
3
(2)
4
Normal Mode
(2)
5
6
...
24 (0x18)
25 (0x19)
26 (0x1A)
27 (0x1B)
...
24
25
26
27
...
Extended Mode
1044 (0x414)
1045 (0x415)
1044
1045
(1) CLKoutX_HS must = 0 for divide by 1.
(2) After programming PLL2_N value, a SYNC must occur on channels using this divide value. Programming PLL2_N does generate a
SYNC event automatically which satisfies this requirement, but NO_SYNC_CLKoutX must be set to 0 for these clock groups.
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8.6.3.2 Registers R6 to R8
Registers R6 to R8 set the clock output types and analog delays.
8.6.3.2.1 CLKoutX_TYPE
The clock output types of the LMK04208 are individually programmable. The CLKoutX_TYPE registers set the
output type of an individual clock output to LVDS, LVPECL, LVCMOS, or powers down the output buffer. Note
that LVPECL supports four different amplitude levels and LVCMOS supports single LVCMOS outputs, inverted,
and normal polarity of each output pin for maximum flexibility.
Table 26 shows at what register and address the specified clock output CLKoutX_TYPE register is located.
The CLKoutX_TYPE table shows the programming definition for these registers.
Table 26. CLKoutX_TYPE Programming Addresses
CLKoutX
CLKout0
CLKout1
CLKout2
CLKout3
CLKout4
CLKout5
PROGRAMMING ADDRESS
R6[23:20]
R6[27:24]
R7[23:20]
R7[27:24]
R8[19:16]
R8[27:24]
Table 27. CLKoutX_TYPE, 4 Bits
R6-R8[27:24, 23:20]
0 (0x00)
DEFINITION
Power down
1 (0x01)
LVDS
2 (0x02)
LVPECL (700 mVpp)
LVPECL (1200 mVpp)
LVPECL (1600 mVpp)
LVPECL (2000 mVpp)
LVCMOS (Norm/Inv)
LVCMOS (Inv/Norm)
LVCMOS (Norm/Norm)
LVCMOS (Inv/Inv)(1)
LVCMOS (Low/Norm)(1)
LVCMOS (Low/Inv)(1)
LVCMOS (Norm/Low)(1)
LVCMOS (Inv/Low)(1)
LVCMOS (Low/Low)(1)
3 (0x03)
4 (0x04)
5 (0x05)
6 (0x06)
7 (0x07)
8 (0x08)
9 (0x09)
10 (0x0A)
11 (0x0A)
12 (0x0C)
13 (0x0D)
14 (0x0E)
(1) To reduce supply switching and crosstalk noise, TI recommends using a complementary LVCMOS output type such as 6 or 7. See
Section Vcc2, Vcc3, Vcc10, Vcc11, Vcc12, Vcc13 (CLKout Vccs) for more information
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8.6.3.2.2 CLKoutX_ADLY
These registers control the analog delay of the clock group CLKoutX. Adding analog delay to the output will
increase the noise floor of the output. For this analog delay to be active for a clock output, it must be selected
with CLKoutX_ADLY_SEL. If neither clock output in a clock group selects the analog delay, then the analog
delay block is powered down. Analog delay may not operate at frequencies above the minimum-ensured
maximum output frequency of 1536 MHz.
In addition to the programmed delay, a fixed 500 ps of delay will be added by engaging the delay block.
The programming addresses table shows at what register and address the specified clock output
CLKoutX_ADLY register is located.
The CLKoutX_ADLY table shows the programming definition for these registers.
Table 28. CLKoutX_ADLY Programming Addresses
CLKoutX_ADLY
CLKout0_ADLY
CLKout1_ADLY
CLKout2_ADLY
CLKout3_ADLY
CLKout4_ADLY
CLKout5_ADLY
PROGRAMMING ADDRESS
R6[9:5]
R6[15:11]
R7[9:5]
R7[15:11]
R8[9:5]
R8[15:11]
Table 29. CLKoutX_ADLY, 5 Bits
R6-R8[15:11, 9:5]
0 (0x00)
DEFINITION
500 ps + No delay
500 ps + 25 ps
500 ps + 50 ps
500 ps + 75 ps
500 ps + 100 ps
500 ps + 125 ps
500 ps + 150 ps
500 ps + 175 ps
500 ps + 200 ps
500 ps + 225 ps
500 ps + 250 ps
500 ps + 275 ps
500 ps + 300 ps
500 ps + 325 ps
500 ps + 350 ps
500 ps + 375 ps
500 ps + 400 ps
500 ps + 425 ps
500 ps + 450 ps
500 ps + 475 ps
500 ps + 500 ps
500 ps + 525 ps
500 ps + 550 ps
500 ps + 575 ps
1 (0x01)
2 (0x02)
3 (0x03)
4 (0x04)
5 (0x05)
6 (0x06)
7 (0x07)
8 (0x08)
9 (0x09)
10 (0x0A)
11 (0x0B)
12 (0x0C)
13 (0x0D)
14 (0x0E)
15 (0x0F)
16 (0x10)
17 (0x11)
18 (0x12)
19 (0x13)
20 (0x14)
21 (0x15)
22 (0x16)
23 (0x17)
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8.6.3.3 Register R10
8.6.3.3.1 OSCout_TYPE
The OSCout clock output has a programmable output type. The OSCout_TYPE register sets the output type to
LVDS, LVPECL, LVCMOS, or powers down the output buffer. Note that LVPECL supports four different
amplitude levels and LVCMOS supports dual and single LVCMOS outputs with inverted, and normal polarity of
each output pin for maximum flexibility.
To turn on the output, the OSCout_TYPE must be set to a non-power down setting and enabled with
EN_OSCout, OSCout Output Enable.
Table 30. OSCout_TYPE, 4 Bits
R10[27:24]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
4 (0x04)
5 (0x05)
6 (0x06)
7 (0x07)
8 (0x08)
9 (0x09)
10 (0x0A)
11 (0x0B)
12 (0x0C)
13 (0x0D)
14 (0x0E)
DEFINITION
Powerdown
LVDS
LVPECL (700 mVpp)
LVPECL (1200 mVpp)
LVPECL (1600 mVpp)
LVPECL (2000 mVpp)
LVCMOS (Norm/Inv)
LVCMOS (Inv/Norm)
LVCMOS (Norm/Norm)(1)
LVCMOS (Inv/Inv)(1)
LVCMOS (Low/Norm)(1)
LVCMOS (Low/Inv)(1)
LVCMOS (Norm/Low)(1)
LVCMOS (Inv/Low)(1)
LVCMOS (Low/Low)(1)
(1) To reduce supply switching and crosstalk noise, TI recommends using a complementary LVCMOS output type such as 6 or 7. See
Vcc2, Vcc3, Vcc10, Vcc11, Vcc12, Vcc13 (CLKout Vccs) for more information"
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8.6.3.3.2 EN_OSCout, OSCout Output Enable
EN_OSCout is used to enable an oscillator buffered output.
Table 31. EN_OSCout
R10[22]
OUTPUT STATE
0
1
OSCout Disabled
OSCout Enabled
Note: In addition to enabling the output with EN_OSCout. The OSCout_TYPE must be programmed to a non-
power down value for the output buffer to power up.
8.6.3.3.3 OSCout_MUX, Clock Output Mux
Sets OSCout buffer to output a divided or bypassed OSCin signal. The divisor is set by OSCout_DIV, Oscillator
Output Divide.
Table 32. OSCout_MUX
R10[20]
MUX OUTPUT
Bypass divider
Divided
0
1
8.6.3.3.4 PD_OSCin, OSCin Powerdown Control
Except in clock distribution mode, the OSCin buffer must always be powered up.
In clock distribution mode, the OSCin buffer must be powered down if not used.
Table 33. PD_OSCin
R10[19]
OSCin BUFFER
0
1
Normal Operation
Powerdown
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8.6.3.3.5 OSCout_DIV, Oscillator Output Divide
The OSCout divider can be programmed from 2 to 8. Divide by 1 is achieved by bypassing the divider with
OSCout_MUX, Clock Output Mux.
Note that OSCout_DIV will be in the PLL1 N feedback path if OSCout_MUX selects divided as an output. When
OSCout_DIV is in the PLL1 N feedback path, the OSCout_DIV divide value must be accounted for when
programming PLL1 N.
See PLL Programming for more information on programming PLL1 to lock.
Table 34. OSCout_DIV, 3 Bits
R10[18:16]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
4 (0x04)
5 (0x05)
6 (0x06)
7 (0x07)
DIVIDE
8
2
2
3
4
5
6
7
8.6.3.3.6 VCO_MUX
When the internal VCO is used, the VCO divider can be selected to divide the VCO output frequency to reduce
the frequency on the clock distribution path. TI recommends using the VCO directly unless:
•
•
Very low output frequencies are required.
If using the VCO divider results in three or more clock output divider/delays changing from extended to
normal power mode, a small power savings may be achieved by using the VCO divider.
A consequence of using the VCO divider is a small degradation in phase noise.
Table 35. VCO_MUX
R10[12]
DEFINITION
0
VCO selected
1
VCO divider selected
8.6.3.3.7 EN_FEEDBACK_MUX
When using 0-delay or dynamic digital delay (SYNC_QUAL = 1), EN_FEEDBACK_MUX must be set to 1 to
power up the feedback mux.
Table 36. EN_FEEDBACK_MUX
R10[11]
DEFINITION
0
1
Feedback mux powered down
Feedback mux enabled
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8.6.3.3.8 VCO_DIV, VCO Divider
Divide value of the VCO Divider.
See PLL Programming for more information on programming PLL2 to lock.
Table 37. VCO_DIV, 3 Bits
R10[10:8]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
4 (0x04)
5 (0x05)
6 (0x06)
7 (0x07)
DIVIDE
8
2
2
3
4
5
6
7
8.6.3.3.9 FEEDBACK_MUX
When in 0-delay mode, the feedback mux selects the clock output to be fed back into the PLL1 N Divider.
Table 38. FEEDBACK_MUX, 3 Bits
R10[7:5]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
4 (0x04)
5 (0x05)
6 (0x06)
MUX OUTPUT
CLKout0
CLKout1
CLKout2
CLKout3
CLKout4
CLKout5
FBCLKin/FBCLKin*
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8.6.3.4 Register R11
8.6.3.4.1 MODE: Device Mode
MODE determines how the LMK04208 operates from a high level. Different blocks of the device can be powered
up and down for specific application requirements from a dual loop architecture to clock distribution.
The LMK04208 can operate in:
•
•
•
Dual PLL mode with the internal VCO or an external VCO.
Single PLL mode uses PLL2 and powers down PLL1. OSCin is used for PLL reference input.
Clock Distribution mode allows use of CLKin1 to distribute to clock outputs CLKout0 through CLKout5, and
OSCin to distribute to OSCout, and optionally CLKout3 and CLKout4.
For the PLL modes, deterministic phase delay with respect to the input can be achieved with the 0-delay mode.
For the PLL modes it is also possible to use an external VCO.
Table 39. MODE, 5 Bits
R11[31:27]
0 (0x00)
VALUE
Dual PLL, Internal VCO
Reserved
1 (0x01)
Dual PLL, Internal VCO,
0-Delay
2 (0x02)
3 (0x03)
4 (0x04)
5 (0x05)
6 (0x06)
7 (0x07)
Dual PLL, External VCO (Fin)
Reserved
Dual PLL, External VCO (Fin), 0-Delay
PLL2, Internal VCO
Reserved
PLL2, Internal VCO,
0–Delay
8 (0x08)
9 (0x09)
10 (0x0A)
11 (0x0B)
12 (0x0C)
13 (0x0D)
14 (0x0E)
15 (0x0F)
16 (0x10)
Reserved
Reserved
PLL2, External VCO (Fin)
Reserved
Reserved
Reserved
Reserved
Clock Distribution
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8.6.3.4.2 EN_SYNC, Enable Synchronization
The EN_SYNC bit (default on) must be enabled for synchronization to work. Synchronization is required for
dynamic digital delay.
The synchronization enable may be turned off once the clocks are operating to save current. If EN_SYNC is set
after it has been cleared (a transition from 0 to 1), a SYNC is generated that can disrupt the active clock outputs.
Setting the NO_SYNC_CLKoutX bits will prevent this SYNC pulse from affecting the output clocks. Setting the
EN_SYNC bit is not a valid method for synchronizing the clock outputs. See the Clock Output Synchronization
section for more information on synchronization.
Table 40. EN_SYNC
R11[26]
DEFINITION
0
1
Synchronization disabled
Synchronization enabled
8.6.3.4.3 NO_SYNC_CLKoutX
The NO_SYNC_CLKoutX bits prevent individual clock groups from becoming synchronized during a SYNC
event. A reason to prevent individual clock groups from becoming synchronized is that during synchronization,
the clock output is in a fixed low state or can have a glitch pulse.
By disabling SYNC on a clock group, it will continue to operate normally during a SYNC event.
Digital delay requires a SYNC operation to take effect. If NO_SYNC_CLKoutX is set before a SYNC event, the
digital delay value will be unused.
Setting the NO_SYNC_CLKoutX bit has no effect on clocks already synchronized together.
Table 41. NO_SYNC_CLKoutX Programming Addresses
NO_SYNC_CLKoutX
CLKout0
PROGRAMMING ADDRESS
R11:20
R11:21
R11:22
R11:23
R11:24
R11:25
CLKout1
CLKout2
CLKout3
CLKout4
CLKout5
Table 42. NO_SYNC_CLKoutX
R11[25, 24, 23, 22, 21, 20]
DEFINITION
0
1
CLKoutX will synchronize
CLKoutX will not synchronize
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8.6.3.4.4 SYNC_MUX
Mux controlling SYNC pin when type is an output.
All the outputs logic is active high when SYNC_TYPE = 3 (Output). All the outputs logic is active low when
SYNC_TYPE = 4 (Output Inverted). For example, when SYNC_MUX = 0 (Logic Low) and SYNC_TYPE = 3
(Output) then SYNC outputs a logic low. When SYNC_MUX = 0 (Logic Low) and SYNC_TYPE = 4 (Output
Inverted) then SYNC outputs a logic high.
Table 43. SYNC_MUX, 2 Bits
R11[19:18]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
SYNC PIN OUTPUT
Logic Low
Reserved
Reserved
uWire Readback
8.6.3.4.5 SYNC_QUAL
When SYNC_QUAL is set, clock outputs will be synchronized to an existing clock output selected by
FEEDBACK_MUX. By using the NO_SYNC_CLKoutX bits, selected clock outputs will not be interrupted during
the SYNC event.
Qualifying the SYNC by an output clock means that the pulse which turns the clock outputs off and on will have a
fixed time relationship to the qualifying output clock.
SYNC_QUAL = 1 requires CLKout2_PD = 0 for proper operation. CLKout2_TYPE may be set to Powerdown
mode.
See Clock Output Synchronization (SYNC) for more information.
Table 44. SYNC_QUAL
R11[17]
MODE
0
No qualification
Qualification by clock output from feedback mux
(Must set CLKout2_PD = 0)
1
8.6.3.4.6 SYNC_POL_INV
Sets the polarity of the SYNC pin when input. When SYNC is asserted the clock outputs will transition to a low
state.
See Clock Output Synchronization (SYNC) for more information on SYNC. A SYNC event can be generated by
toggling this bit through the MICROWIRE interface.
Table 45. SYNC_POL_INV
R11[16]
POLARITY
0
1
SYNC is active high
SYNC is active low
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8.6.3.4.7 SYNC_EN_AUTO
When set, causes a SYNC event to occur when programming register R0 to R5 to adjust digital delay values.
The SYNC event will coincide with the LEuWire pin falling edge.
Refer to Special Programming Case for R0 to R5 for CLKoutX_DIV and CLKoutX_DDLY for more information on
possible special programming considerations when SYNC_EN_AUTO = 1.
Table 46. SYNC_EN_AUTO
R11[15]
MODE
0
1
Manual SYNC
SYNC Internally Generated
8.6.3.4.8 SYNC_TYPE
Sets the IO type of the SYNC pin.
Table 47. SYNC_TYPE, 3 Bits
R11[14:12]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
4 (0x04)
5 (0x05)
6 (0x06)
POLARITY
Input
Input w/ pull-up resistor
Input w/ pull-down resistor
Output (push-pull)
Output inverted (push-pull)
Output (open source)
Output (open drain)
When in output mode, the SYNC input is forced to 0 regardless of the SYNC_MUX setting. A synchronization
can then be activated by uWire by programming the SYNC_POL_INV register to active low to assert SYNC.
SYNC can then be released by programming SYNC_POL_INV to active high. Using this uWire programming
method to create a SYNC event saves the need for an IO pin from another device.
8.6.3.4.9 EN_PLL2_XTAL
If an external crystal is being used to implement a discrete VCXO, the internal feedback amplifier must be
enabled with this bit in order to complete the oscillator circuit.
Table 48. EN_PLL2_XTAL
R11[5]
OSCILLATOR AMPLIFIER STATE
0
1
Disabled
Enabled
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8.6.3.5 Register R12
8.6.3.5.1 LD_MUX
LD_MUX sets the output value of the LD pin.
All the outputs logic is active high when LD_TYPE = 3 (Output). All the outputs logic is active low when
LD_TYPE = 4 (Output Inverted). For example, when LD_MUX = 0 (Logic Low) and LD_TYPE = 3 (Output) then
Status_LD outputs a logic low. When LD_MUX = 0 (Logic Low) and LD_TYPE = 4 (Output Inverted) then
Status_LD outputs a logic high.
Table 49. LD_MUX, 5 Bits
R12[31:27]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
4 (0x04)
5 (0x05)
6 (0x06)
7 (0x07)
8 (0x08)
9 (0x09)
10 (0x0A)
11 (0x0B)
12 (0x0C)
13 (0x0D)
14 (0x0E)
15 (0x0F)
16 (0x10)
17 (0x11)
18 (0x12)
MODE
Logic Low
PLL1 DLD
PLL2 DLD
PLL1 and PLL2 DLD
Holdover Status
DAC Locked
Reserved
uWire Readback
DAC Rail
DAC Low
DAC High
PLL1_N
PLL1_N/2
PLL2 N
PLL2 N/2
PLL1_R
PLL1_R/2
(1)
PLL2 R
(1)
PLL2 R/2
(1) Only valid when HOLDOVER_MUX is not set to 2 (PLL2_DLD) or 3 (PLL1 and PLL2 DLD).
8.6.3.5.2 LD_TYPE
Sets the IO type of the LD pin.
Table 50. LD_TYPE, 3 Bits
R12[26:24]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
POLARITY
Reserved
Reserved
Reserved
Output (push-pull)
4 (0x04)
5 (0x05)
6 (0x06)
Output inverted (push-pull)
Output (open source)
Output (open drain)
74
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8.6.3.5.3 SYNC_PLLX_DLD
By setting SYNC_PLLX_DLD a SYNC mode will be engaged (asserted SYNC) until PLL1 and/or PLL2 locks.
SYNC_QUAL must be 0 to use this functionality.
Table 51. SYNC_PLL2_DLD
R12[23]
SYNC MODE FORCED
0
1
No
Yes
Table 52. SYNC_PLL1_DLD
R12[22]
SYNC MODE FORCED
0
1
No
Yes
8.6.3.5.4 EN_TRACK
Enable the DAC to track the PLL1 tuning voltage. For optional use in in holdover mode.
Tracking can be used to monitor PLL1 voltage by readback of DAC_CNT register in any mode.
Table 53. EN_TRACK
R12[8]
DAC TRACKING
Disabled
0
1
Enabled
8.6.3.5.5 HOLDOVER_MODE
Enable the holdover mode.
Table 54. HOLDOVER_MODE, 2 Bits
R12[7:6]
HOLDOVER MODE
Reserved
0
1
2
3
Disabled
Enabled
Reserved
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8.6.3.6 Register R13
8.6.3.6.1 HOLDOVER_MUX
HOLDOVER_MUX sets the output value of the Status_Holdover pin.
The outputs are active high when HOLDOVER_TYPE = 3 (Output). The outputs are active low when
HOLDOVER_TYPE = 4 (Output Inverted).
Table 55. HOLDOVER_MUX, 5 Bits
R13[31:27]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
4 (0x04)
5 (0x05)
6 (0x06)
7 (0x07)
8 (0x08)
9 (0x09)
10 (0x0A)
11 (0x0B)
12 (0x0C)
13 (0x0D)
14 (0x0E)
15 (0x0F)
16 (0x10)
17 (0x11)
18 (0x12)
DEFINITION
Logic Low
PLL1 DLD
PLL2 DLD
PLL1 and PLL2 DLD
Holdover Status
DAC Locked
Reserved
uWire Readback
DAC Rail
DAC Low
DAC High
PLL1 N
PLL1 N/2
PLL2 N
PLL2 N/2
PLL1 R
PLL1 R/2
(1)
PLL2 R
(1)
PLL2 R/2
(1) Only valid when LD_MUX is not set to 2 (PLL2_DLD) or 3 (PLL1 and PLL2 DLD).
8.6.3.6.2 HOLDOVER_TYPE
Sets the IO mode of the Status_Holdover pin.
Table 56. HOLDOVER_TYPE, 3 Bits
R13[26:24]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
POLARITY
Reserved
Reserved
Reserved
Output (push-pull)
4 (0x04)
5 (0x05)
6 (0x06)
Output inverted (push-pull)
Output (open source)
Output (open drain)
76
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8.6.3.6.3 Status_CLKin1_MUX
Status_CLKin1_MUX sets the output value of the Status_CLKin1 pin. If Status_CLKin1_TYPE is set to an input
type, this register has no effect. This MUX register only sets the output signal.
The outputs are active high when Status_CLKin1_TYPE = 3 (Output). The outputs are active low when
Status_CLKin1_TYPE = 4 (Output Inverted).
Table 57. Status_CLKin1_MUX, 3 Bits
R13[22:20]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
4 (0x04)
5 (0x05)
6 (0x06)
DEFINITION
Logic Low
CLKin1 LOS
CLKin1 Selected
DAC Locked
DAC Low
DAC High
uWire Readback
8.6.3.6.4 Status_CLKin0_TYPE
Status_CLKin0_TYPE sets the IO type of the Status_CLKin0 pin.
Table 58. Status_CLKin0_TYPE, 3 Bits
R13[18:16]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
4 (0x04)
5 (0x05)
6 (0x06)
DEFINITION
Input
Input w/ pull-up resistor
Input w/ pull-down resistor
Output (push-pull)
Output inverted (push-pull)
Output (open source)
Output (open drain)
8.6.3.6.5 DISABLE_DLD1_DET
DISABLE_DLD1_DET disables the HOLDOVER mode from being activated when PLL1 lock detect signal
transitions from high to low.
When using Pin Select Mode as the input clock switch mode, this bit should normally be set.
Table 59. DISABLE_DLD1_DET
R13[15]
HOLDOVER DLD1 DETECT
PLL1 DLD causes clock switch event
PLL1 DLD does not cause clock switch event
0
1
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8.6.3.6.6 Status_CLKin0_MUX
CLKin0_MUX sets the output value of the Status_CLKin0 pin. If Status_CLKin0_TYPE is set to an input type, this
register has no effect. This MUX register only sets the output signal.
The outputs logic is active high when Status_CLKin0_TYPE = 3 (Output). The outputs logic is active low when
Status_CLKin0_TYPE = 4 (Output Inverted).
Table 60. Status_CLKin0_MUX, 3 Bits
R13[14:12]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
4 (0x04)
5 (0x05)
6 (0x06)
DEFINITION
Logic Low
CLKin0 LOS
CLKin0 Selected
DAC Locked
DAC Low
DAC High
uWire Readback
8.6.3.6.7 CLKin_SELECT_MODE
CLKin_SELECT_MODE sets the mode used in determining reference CLKin for PLL1.
Table 61. CLKin_SELECT_MODE, 3 Bits
R13[11:9]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
4 (0x04)
5 (0x05)
6 (0x06)
7 (0x07)
MODE
CLKin0 Manual
CLKin1 Manual
Reserved
Pin Select Mode
Auto Mode
Reserved
Auto mode and next clock pin select
Reserved
8.6.3.6.8 CLKin_Sel_INV
CLKin_Sel_INV sets the input polarity of Status_CLKin0 and Status_CLKin1 pins.
Inversion for Status 0 and 1 pins is only valid for CLKin_SELECT_MODE = 0x06.
Table 62. CLKin_Sel_INV
R13[8]
INPUT
0
1
Active High
Active Low
78
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8.6.3.6.9 EN_CLKinX
Each clock input can individually be enabled to be used during auto-switching CLKin_SELECT_MODE. Clock
input switching priority is always CLKin0 → CLKin1.
Table 63. EN_CLKin1
R13[6]
ENABLED
No
0
1
Yes
Table 64. EN_CLKin0
R13[5]
ENABLED
No
0
1
Yes
8.6.3.7 Register 14
8.6.3.7.1 LOS_TIMEOUT
This bit controls the amount of time in which no activity on a CLKin causes LOS (Loss-of-Signal) to be asserted.
Table 65. LOS_TIMEOUT, 2 Bits
R14[31:30]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
TIMEOUT
1200 ns, 420 kHz
206 ns, 2.5 MHz
52.9 ns, 10 MHz
23.7 ns, 22 MHz
8.6.3.7.2 EN_LOS
Enables the LOS (Loss-of-Signal) timeout control.
Table 66. EN_LOS
R14[28]
LOS
0
1
Disabled
Enabled
8.6.3.7.3 Status_CLKin1_TYPE
Sets the IO type of the Status_CLKin1 pin.
Table 67. Status_CLKin1_TYPE, 3 Bits
R14[26:24]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
4 (0x04)
5 (0x05)
6 (0x06)
POLARITY
Input
Input w/ pull-up resistor
Input w/ pull-down resistor
Output (push-pull)
Output inverted (push-pull)
Output (open source)
Output (open drain)
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8.6.3.7.4 CLKinX_BUF_TYPE, PLL1 CLKinX/CLKinX* Buffer Type
There are two input buffer types for the PLL1 reference clock inputs: either bipolar or CMOS. Bipolar is
recommended for differential inputs such as LVDS and LVPECL. CMOS is recommended for DC coupled single
ended inputs.
When using bipolar, CLKinX and CLKinX* input pins must be AC coupled when using a differential or single
ended input.
When using CMOS, CLKinX and CLKinX* input pins may be AC or DC coupled with a differential input.
When using CMOS in single ended mode, the unused clock input pin (CLKinX or CLKinX*) must be AC
grounded. The used clock input pin (CLKinX* or CLKinX) may be AC or DC coupled to the signal source.
The programming addresses table shows at what register and address the specified CLKinX_BUF_TYPE bit is
located.
The CLKinX_BUF_TYPE table shows the programming definition for these registers.
Table 68. CLKinX_BUF_TYPE Programming Addresses
CLKinX_BUF_TYPE
CLKin1_BUF_TYPE
CLKin0_BUF_TYPE
PROGRAMMING ADDRESS
R14[21]
R14[20]
Table 69. CLKinX_BUF_TYPE
R14[21, 20]
CLKinX BUFFER TYPE
0
1
Bipolar
CMOS
8.6.3.7.5 DAC_HIGH_TRIP
Voltage from Vcc at which holdover mode is entered if EN_VTUNE_RAIL_DAC is enabled. This will also set flags
which can be monitored out Status_LD/Status_Holdover pins.
Step size is ~51 mV.
Table 70. DAC_HIGH_TRIP, 6 Bits
R14[19:14]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
4 (0x04)
...
TRIP VOLTAGE FROM VCC (V)
1 × Vcc / 64
2 × Vcc / 64
3 × Vcc / 64
4 × Vcc / 64
5 × Vcc / 64
...
61 (0x3D)
62 (0x3E)
63 (0x3F)
62 × Vcc / 64
63 × Vcc / 64
64 × Vcc / 64
80
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8.6.3.7.6 DAC_LOW_TRIP
Voltage from GND at which holdover mode is entered if EN_VTUNE_RAIL_DAC is enabled. This will also set
flags which can be monitored out Status_LD/Status_Holdover pins.
Step size is ~51 mV
Table 71. DAC_LOW_TRIP, 6 Bits
R14[11:6]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
4 (0x04)
...
TRIP VOLTAGE from GND (V)
1 × Vcc / 64
2 × Vcc / 64
3 × Vcc / 64
4 × Vcc / 64
5 × Vcc / 64
...
61 (0x3D)
62 (0x3E)
63 (0x3F)
62 × Vcc / 64
63 × Vcc / 64
64 × Vcc / 64
8.6.3.7.7 EN_VTUNE_RAIL_DET
Enables the DAC Vtune rail detection. When the DAC achieves a specified Vtune, if this bit is enabled, the
current clock input is considered invalid and an input clock switch event is generated.
Table 72. EN_VTUNE_RAIL_DET
R14[5]
STATE
Disabled
Enabled
0
1
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8.6.3.8 Register 15
8.6.3.8.1 MAN_DAC
Sets the DAC value when in manual DAC mode in ~3.2 mV steps.
Table 73. MAN_DAC, 10 Bits
R15[31:22]
0 (0x00)
1 (0x01)
2 (0x02)
...
DAC VOLTAGE
0 × Vcc / 1023
1 × Vcc / 1023
2 × Vcc / 1023
...
1023 (0x3FF)
1023 × Vcc / 1023
8.6.3.8.2 EN_MAN_DAC
This bit enables the manual DAC mode.
Table 74. EN_MAN_DAC
R15[20]
DAC MODE
Automatic
Manual
0
1
8.6.3.8.3 HOLDOVER_DLD_CNT
Lock must be valid for this many clocks of PLL1 PDF before holdover mode is exited.
Table 75. HOLDOVER_DLD_CNT, 14 Bits
R15[19:6]
0 (0x00)
EXIT COUNTS
Reserved
1 (0x01)
1
2
2 (0x02)
...
...
16,383 (0x3FFF)
16,383
8.6.3.8.4 FORCE_HOLDOVER
This bit forces the holdover mode.
When holdover is forced, if in fixed CPout1 mode (EN_TRACK = 0 or 1, EN_MAN_DAC = 1) , then the DAC will
set the programmed MAN_DAC value. If in tracked CPout1 mode (EN_TRACK = 1, EN_MAN_DAC = 0,
EN_VTUNE_RAIL_DET = 0), then the DAC will set the current tracked DAC value.
Setting FORCE_HOLDOVER does not constitute a clock input switch event unless DISABLE_DLD1_DET = 0,
since when in holdover mode, PLL1_DLD = 0 will trigger the clock input switch event.
Table 76. FORCE_HOLDOVER
R15[5]
HOLDOVER
Disabled
0
1
Enabled
82
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8.6.3.9 Register 16
8.6.3.9.1 XTAL_LVL
Sets the peak amplitude on the tunable crystal.
Increasing this value can improve the crystal oscillator phase noise performance at the cost of increased current
and higher crystal power dissipation levels.
Table 77. XTAL_LVL, 2 Bits
R15[31:22]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
PEAK AMPLITUDE(1)
1.65 Vpp
1.75 Vpp
1.90 Vpp
2.05 Vpp
(1) At crystal frequency of 20.48 MHz
8.6.3.10 Register 23
This register must not be programmed, it is a readback only register.
8.6.3.10.1 DAC_CNT
The DAC_CNT register is 10 bits in size and located at readback bit position R23[23:14]. When using tracking
mode for holdover, the DAC value can be readback at this address.
8.6.3.11 Register 24
8.6.3.11.1 PLL2_C4_LF, PLL2 Integrated Loop Filter Component
Internal loop filter components are available for PLL2, enabling either 3rd or 4th order loop filters without
requiring external components.
Internal loop filter capacitor C4 can be set according to Table 78.
Table 78. PLL2_C4_LF, 4 Bits
R24[31:28]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
4 (0x04)
5 (0x05)
6 (0x06)
7 (0x07)
8 (0x08)
9 (0x09)
10 (0x0A)
11 (0x0B)
12 (0x0C)
13 (0x0D)
14 (0x0E)
15 (0x0F)
LOOP FILTER CAPACITANCE (pF)
10 pF
15 pF
29 pF
34 pF
47 pF
52 pF
66 pF
71 pF
103 pF
108 pF
122 pF
126 pF
141 pF
146 pF
Reserved
Reserved
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8.6.3.11.2 PLL2_C3_LF, PLL2 Integrated Loop Filter Component
Internal loop filter components are available for PLL2, enabling either 3rd or 4th order loop filters without
requiring external components.
Internal loop filter capacitor C3 can be set according to Table 79.
Table 79. PLL2_C3_LF, 4 Bits
R24[27:24]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
4 (0x04)
5 (0x05)
6 (0x06)
7 (0x07)
8 (0x08)
9 (0x09)
10 (0x0A)
11 (0x0B)
12 (0x0C)
13 (0x0D)
14 (0x0E)
15 (0x0F)
LOOP FILTER CAPACITANCE (pF)
10 pF
11 pF
15 pF
16 pF
19 pF
20 pF
24 pF
25 pF
29 pF
30 pF
33 pF
34 pF
38 pF
39 pF
Reserved
Reserved
8.6.3.11.3 PLL2_R4_LF, PLL2 Integrated Loop Filter Component
Internal loop filter components are available for PLL2, enabling either 3rd or 4th order loop filters without
requiring external components.
Internal loop filter resistor R4 can be set according to Table 80.
Table 80. PLL2_R4_LF, 3 Bits
R24[22:20]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
4 (0x04)
5 (0x05)
6 (0x06)
7 (0x07)
RESISTANCE
200 Ω
1 kΩ
2 kΩ
4 kΩ
16 kΩ
Reserved
Reserved
Reserved
84
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8.6.3.11.4 PLL2_R3_LF, PLL2 Integrated Loop Filter Component
Internal loop filter components are available for PLL2, enabling either 3rd or 4th order loop filters without
requiring external components.
Internal loop filter resistor R3 can be set according to Table 81.
Table 81. PLL2_R3_LF, 3 Bits
R24[18:16]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
4 (0x04)
5 (0x05)
6 (0x06)
7 (0x07)
RESISTANCE
200 Ω
1 kΩ
2 kΩ
4 kΩ
16 kΩ
Reserved
Reserved
Reserved
8.6.3.11.5 PLL1_N_DLY
Increasing delay of PLL1_N_DLY will cause the outputs to lead from CLKinX. For use in 0-delay mode.
Table 82. PLL1_N_DLY, 3 Bits
R24[14:12]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
4 (0x04)
5 (0x05)
6 (0x06)
7 (0x07)
DEFINITION
0 ps
205 ps
410 ps
615 ps
820 ps
1025 ps
1230 ps
1435 ps
8.6.3.11.6 PLL1_R_DLY
Increasing delay of PLL1_R_DLY will cause the outputs to lag from CLKinX. For use in 0-delay mode.
Table 83. PLL1_R_DLY, 3 Bits
R24[10:8]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
4 (0x04)
5 (0x05)
6 (0x06)
7 (0x07)
DEFINITION
0 ps
205 ps
410 ps
615 ps
820 ps
1025 ps
1230 ps
1435 ps
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8.6.3.11.7 PLL1_WND_SIZE
PLL1_WND_SIZE sets the window size used for digital lock detect for PLL1. If the phase error between the
reference and feedback of PLL1 is less than specified time, then the PLL1 lock counter increments.
Refer to Digital Lock Detect Frequency Accuracy for more information.
Table 84. PLL1_WND_SIZE, 2 Bits
R24[7:6]
DEFINITION
5.5 ns
0
1
2
3
10 ns
18.6 ns
40 ns
8.6.3.12 Register 25
8.6.3.12.1 DAC_CLK_DIV
The DAC update clock frequency is the PLL1 phase detector frequency divided by the divisor listed in Table 85.
Table 85. DAC_CLK_DIV, 10 Bits
R25[31:22]
0 (0x00)
DIVIDE
Reserved
1 (0x01)
1
2
2 (0x02)
3 (0x03)
3
...
...
1,022 (0x3FE)
1,023 (0x3FF)
1022
1023
8.6.3.12.2 PLL1_DLD_CNT
The reference and feedback of PLL1 must be within the window of phase error as specified by PLL1_WND_SIZE
for this many phase detector cycles before PLL1 digital lock detect is asserted.
Refer to Digital Lock Detect Frequency Accuracy for more information.
Table 86. PLL1_DLD_CNT, 14 Bits
R25[19:6]
0 (0x0000)
1 (0x0001)
2 (0x0002)
3 (0x0003)
...
VALUE
Reserved
1
2
3
...
16,382 (0x3FFE)
16,383 (0x3FFF)
16,382
16,383
86
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8.6.3.13 Register 26
8.6.3.13.1 PLL2_WND_SIZE
PLL2_WND_SIZE sets the window size used for digital lock detect for PLL2. If the phase error between the
reference and feedback of PLL2 is less than specified time, then the PLL2 lock counter increments. This value
must be programmed to 2 (3.7 ns).
Refer to Digital Lock Detect Frequency Accuracy for more information.
Table 87. PLL2_WND_SIZE, 2 Bits
R26[31:30]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
DEFINITION
Reserved
Reserved
3.7 ns
Reserved
8.6.3.13.2 EN_PLL2_REF_2X, PLL2 Reference Frequency Doubler
Enabling the PLL2 reference frequency doubler allows for higher phase detector frequencies on PLL2 than would
normally be allowed with the given VCXO or Crystal frequency.
Higher phase detector frequencies reduces the PLL N values which makes the design of wider loop bandwidth
filters possible.
Table 88. EN_PLL2_REF_2X
R26[29]
DESCRIPTION
0
1
Reference frequency normal(1)
Reference frequency doubled (2x)(2)
(1) When the doubler is not enabled, PLL2_R should not be programmed to 1.
(2) See PLL2 Frequency Doubler
8.6.3.13.3 PLL2_CP_POL, PLL2 Charge Pump Polarity
PLL2_CP_POL sets the charge pump polarity for PLL2. The internal VCO requires the negative charge pump
polarity to be selected. Many VCOs use positive slope.
A positive slope VCO increases output frequency with increasing voltage. A negative slope VCO decreases
output frequency with increasing voltage.
Table 89. PLL2_CP_POL
R26[28]
DESCRIPTION
0
1
Negative Slope VCO/VCXO
Positive Slope VCO/VCXO
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8.6.3.13.4 PLL2_CP_GAIN, PLL2 Charge Pump Current
This bit programs the PLL2 charge pump output current level. Table 90 also illustrates the impact of the PLL2
TRI-STATE bit in conjunction with PLL2_CP_GAIN.
Table 90. PLL2_CP_GAIN, 2 Bits
PLL2_CP_TRI
R26[27:26]
CHARGE PUMP CURRENT (µA)
R26[5]
X
1
0
0
0
0
Hi-Z
100
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
400
1600
3200
8.6.3.13.5 PLL2_DLD_CNT
The reference and feedback of PLL2 must be within the window of phase error as specified by PLL2_WND_SIZE
for PLL2_DLD_CNT cycles before PLL2 digital lock detect is asserted.
Refer to Digital Lock Detect Frequency Accuracy for more information
Table 91. PLL2_DLD_CNT, 14 Bits
R26[19:6]
0 (0x00)
VALUE
Reserved
1 (0x01)
1
2 (0x02)
2
3
3 (0x003)
...
...
16,382 (0x3FFE)
16,383 (0x3FFF)
16,382
16,383
8.6.3.13.6 PLL2_CP_TRI, PLL2 Charge Pump TRI-STATE
This bit allows for the PLL2 charge pump output pin, CPout2, to be placed into TRI-STATE.
Table 92. PLL2_CP_TRI
R26[5]
DESCRIPTION
0
1
PLL2 CPout2 is active
PLL2 CPout2 is at TRI-STATE
88
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8.6.3.14 Register 27
8.6.3.14.1 PLL1_CP_POL, PLL1 Charge Pump Polarity
PLL1_CP_POL sets the charge pump polarity for PLL1. Many VCXOs use positive slope.
A positive slope VCXO increases output frequency with increasing voltage. A negative slope VCXO decreases
output frequency with increasing voltage.
Table 93. PLL1_CP_POL
R27[28]
DESCRIPTION
0
1
Negative Slope VCO/VCXO
Positive Slope VCO/VCXO
8.6.3.14.2 PLL1_CP_GAIN, PLL1 Charge Pump Current
This bit programs the PLL1 charge pump output current level. Table 94 also illustrates the impact of the PLL1
TRI-STATE bit in conjunction with PLL1_CP_GAIN.
Table 94. PLL1_CP_GAIN, 2 Bits
PLL1_CP_TRI
R26[27:26]
CHARGE PUMP CURRENT (µA)
R27[5]
X
1
0
0
0
0
Hi-Z
100
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
200
400
1600
8.6.3.14.3 CLKinX_PreR_DIV
The pre-R dividers before the PLL1 R divider can be programmed such that when the active clock input is
switched, the frequency at the input of the PLL1 R divider will be the same. This allows PLL1 to stay in lock
without needing to re-program the PLL1 R register when different clock input frequencies are used. This is
especially useful in the auto CLKin switching modes.
Table 95. CLKinX_PreR_DIV Programming Addresses
CLKinX_PreR_DIV
CLKin1_PreR_DIV
CLKin0_PreR_DIV
PROGRAMMING ADDRESS
R27[23:22]
R27[21:20]
Table 96. CLKinX_PreR_DIV, 2 Bits
R27[23:22, 21:20]
0 (0x00)
DIVIDE
1
2
4
8
1 (0x01)
2 (0x02)
3 (0x03)
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8.6.3.14.4 PLL1_R, PLL1 R Divider
The reference path into the PLL1 phase detector includes the PLL1 R divider. Refer to PLL Programming for
more information on how to program the PLL dividers to lock the PLL.
The valid values for PLL1_R are shown in Table 97.
Table 97. PLL1_R, 14 Bits
R27[19:6]
0 (0x00)
DIVIDE
Reserved
1 (0x01)
1
2 (0x02)
2
3
3 (0x03)
...
...
16,382 (0x3FFE)
16,383 (0x3FFF)
16,382
16,383
8.6.3.14.5 PLL1_CP_TRI, PLL1 Charge Pump TRI-STATE
This bit allows for the PLL1 charge pump output pin, CPout1, to be placed into TRI-STATE.
Table 98. PLL1_CP_TRI
R27[5]
DESCRIPTION
0
1
PLL1 CPout1 is active
PLL1 CPout1 is at TRI-STATE
90
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8.6.3.15 Register 28
8.6.3.15.1 PLL2_R, PLL2 R Divider
The reference path into the PLL2 phase detector includes the PLL2 R divider.
Refer to PLL Programming for more information on how to program the PLL dividers to lock the PLL.
The valid values for PLL2_R are shown in Table 99.
Table 99. PLL2_R, 12 Bits
R28[31:20]
0 (0x00)
DIVIDE
Not Valid
1 (0x01)
1(1)
2
.
2 (0x02)
3 (0x03)
3
...
...
4,094 (0xFFE)
4,095 (0xFFF)
4,094
4,095
(1) When using PLL2_R divide value of 1, the PLL2 reference doubler should be used (EN_PLL2_REF_2X = 1). See PLL2 Frequency
Doubler.
8.6.3.15.2 PLL1_N, PLL1 N Divider
The feedback path into the PLL1 phase detector includes the PLL1 N divider.
Refer to PLL Programming for more information on how to program the PLL dividers to lock the PLL.
The valid values for PLL1_N are shown in Table 100.
Table 100. PLL1_N, 14 Bits
R28[19:6]
0 (0x00)
1 (0x01)
2 (0x02)
...
DIVIDE
Not Valid
1
2
...
4,095 (0xFFF)
4,095
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8.6.3.16 Register 29
8.6.3.16.1 OSCin_FREQ, PLL2 Oscillator Input Frequency Register
The frequency of the PLL2 reference input to the PLL2 Phase Detector (OSCin/OSCin* port) must be
programmed in order to support proper operation of the frequency calibration routine which locks the internal
VCO to the target frequency.
Table 101. OSCin_FREQ, 3 Bits
R29[26:24]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
4 (0x04)
OSCin FREQUENCY
0 to 63 MHz
>63 MHz to 127 MHz
>127 MHz to 255 MHz
Reserved
>255 MHz to 400 MHz
8.6.3.16.2 PLL2_FAST_PDF, High PLL2 Phase Detector Frequency
When PLL2 phase detector frequency is greater than 100 MHz, set the PLL2_FAST_PDF to ensure proper
operation of device.
Table 102. PLL2_FAST_PDF
R29[23]
PLL2 PDF
Less than or
equal to 100 MHz
0
1
Greater than 100 MHz
8.6.3.16.3 PLL2_N_CAL, PLL2 N Calibration Divider
During the frequency calibration routine, the PLL uses the divide value of the PLL2_N_CAL register instead of
the divide value of the PLL2_N register to lock the VCO to the target frequency.
This is only used for internal PLL2 VCO modes.
NOTE: Unless in 0-delay mode or external VCO mode, PLL2_N_CAL should be set equal to PLL2_N.
Refer to PLL Programming for more information on how to program the PLL dividers to lock the PLL.
Table 103. PLL2_N_CAL, 18 Bits
R29[22:5]
0 (0x00)
DIVIDE
Not Valid
Not Valid
2
1 (0x01)
2 (0x02)
...
...
262,143 (0x3FFFF)
262,143
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8.6.3.17 Register 30
If an internal VCO mode is used, programming Register 30 triggers the frequency calibration routine. This
calibration routine will also generate a SYNC event. See Clock Output Synchronization (SYNC) for more details
on a SYNC.
8.6.3.17.1 PLL2_P, PLL2 N Prescaler Divider
The PLL2 N Prescaler divides the output of the VCO as selected by Mode_MUX1VCO_MUX and is connected to
the PLL2 N divider.
Refer to PLL Programming for more information on how to program the PLL dividers to lock the PLL.
Table 104. PLL2_P, 3 Bits
R30[26:24]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
4 (0x04)
5 (0x05)
6 (0x06)
7 (0x07)
DIVIDE VALUE
8
2
2
3
4
5
6
7
8.6.3.17.2 PLL2_N, PLL2 N Divider
The feeback path into the PLL2 phase detector includes the PLL2 N divider.
Each time register 30 is updated via the MICROWIRE interface and the internal VCO is used, a frequency
calibration routine runs to lock the VCO to the target frequency. During this calibration PLL2_N is substituted with
PLL2_N_CAL.
Refer to PLL Programming for more information on how to program the PLL dividers to lock the PLL.
The valid values for PLL2_N are shown in Table 105.
Table 105. PLL2_N, 18 Bits
R30[22:5]
0 (0x00)
1 (0x01)
2 (0x02)
DIVIDE
Not Valid
1(1)
2
...
262,143 (0x3FFFF)
262,143
(1) Valid in PLL2 external VCO mode. When using internal PLL2 VCO, be aware that PLL2_N_CAL cannot be 1. Some PLL2 internal VCO
0-delay cases may allow PLL2_N = 1 as PLL2_N_CAL will be greater than 1. If PLL2_N = 1 requires PLL2_N_CAL = 1, then this setting
cannot be used.
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8.6.3.18 Register 31
8.6.3.18.1 READBACK_LE
Sets the required state of the LEuWire pin when performing register readback.
Refer to Readback.
Table 106. READBACK_LE
R31[21]
DEFINITION
0
1
LE must be low for readback
LE must be high for readback
8.6.3.18.2 READBACK_ADDR
Sets the address of the register to read back when performing readback.
When reading register 12, the READBACK_ADDR will be read back at R12[20:16].
When reading back from R31 bits 6 to 31 should be ignored. Only uWire_LOCK is valid.
Refer to Register Readback for more information on readback.
Table 107. READBACK_ADDR, 5 Bits
R31[20:16]
0 (0x00)
1 (0x01)
2 (0x02)
3 (0x03)
4 (0x04)
5 (0x05)
6 (0x06)
7 (0x07)
8 (0x08)
9 (0x09)
10 (0x0A)
11 (0x0B)
12 (0x0C)
13 (0x0D)
14 (0x0E)
15 (0x0F)
16 (0x10)
17 (0x11)
...
REGISTER
R0
R1
R2
R3
R4
R5
R6
R7
R8
Reserved
R10
R11
R12
R13
R14
R15
R16
Reserved
...
22 (0x16)
23 (0x17)
24 (0x18)
25 (0x19)
26 (0x1A)
27 (0x1B)
28 (0x1C)
29 (0x1D)
30 (0x1E)
31 (0x1F)
Reserved
Reserved
R24
R25
R26
R27
R28
R29
R30
R31
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8.6.3.18.3 uWire_LOCK
Setting uWire_LOCK will prevent any changes to uWire registers R0 to R30. Only by clearing the uWire_LOCK
bit in R31 can the uWire registers be unlocked and written to once more.
It is not necessary to lock the registers to perform a readback operation.
Table 108. uWire_LOCK
R31[5]
STATE
0
1
Registers unlocked
Registers locked, Write-protect
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9 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
9.1 Application Information
To assist customers in frequency planning and design of loop filters, Texas Instruments provides Clock Architect.
9.1.1 Loop Filter
Each PLL of the LMK04208 requires a dedicated loop filter.
LMK04208 PLL1
LMK04208 PLL2
PLL2 Internal Loop Filter
Internal VCO
PLL2
Phase
R3
R4
PLL1
Phase
Detector
Detector
C3
C4
CPout1
CPout2
External VCXO
to h{/in
PLL1 External Loop
Filter
PLL2 External Loop
Filter
LF1_C2
LF1_R2
LF2_C2
LF1_C1
LF2_C1
LF2_R2
Figure 21. PLL1 and PLL2 Loop Filters
9.1.1.1 PLL1
The loop filter for PLL1 must be connected to the CPout1 pin. Figure 21 shows a simple 2-pole loop filter. The
output of the filter drives an external VCXO module or discrete implementation of a VCXO using a crystal
resonator and external varactor diode. Higher order loop filters may be implemented using additional external R
and C components. TI recommends that the loop filter for PLL1 result in a total closed loop bandwidth in the
range of 10 Hz to 200 Hz. The design of the loop filter is application specific and highly dependent on
parameters such as the phase noise of the reference clock, VCXO phase noise, and phase detector frequency
for PLL1. TI's Clock Conditioner Owner’s Manual covers this topic in detail and Texas Instruments Clock
Architect can be used to simulate loop filter designs for both PLLs. These resources may be found at:
Clock and Timing landing page.
9.1.1.2 PLL2
As shown in Figure 21, the charge pump for PLL2 is directly connected to the optional internal loop filter
components, which are normally used only if either a third or fourth pole is needed. The first and second poles
are implemented with external components. The loop must be designed to be stable over the entire application-
specific tuning range of the VCO. The designer should note the range of KVCO listed in the table of Electrical
Characteristics and how this value can change over the expected range of VCO tuning frequencies. Because
loop bandwidth is directly proportional to KVCO, the designer should model and simulate the loop at the expected
extremes of the desired tuning range, using the appropriate values for KVCO
.
When designing with the integrated loop filter of the LMK04208, considerations for minimum resistor thermal
noise often lead one to the decision to design for the minimum value for integrated resistors, R3 and R4.
96
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Application Information (continued)
Both the integrated loop filter resistors (R3 and R4) and capacitors (C3 and C4) also restrict the maximum loop
bandwidth. However, these integrated components do have the advantage that they are closer to the VCO and
can therefore filter out some noise and spurs better than external components. For this reason, a common
strategy is to minimize the internal loop filter resistors and then design for the largest internal capacitor values
that permit a wide enough loop bandwidth. In situations where spur requirements are very stringent and there is
margin on phase noise, a feasible strategy would be to design a loop filter with integrated resistor values larger
than their minimum value.
9.1.2 Driving CLKin and OSCin Inputs
9.1.2.1 Driving CLKin Pins with a Differential Source
Both CLKin ports can be driven by differential signals. TI recommends that the input mode be set to bipolar
(CLKinX_BUF_TYPE = 0) when using differential reference clocks. The LMK04208 internally biases the input
pins so the differential interface should be AC coupled. The recommended circuits for driving the CLKin pins with
either LVDS or LVPECL are shown in Figure 22 and Figure 23.
CLKinX
0.1 mF
100 W Trace
LMK04208
Input
LVDS
(Differential)
0.1 mF
CLKinX*
Figure 22. CLKinX/X* Termination for an LVDS Reference Clock Source
CLKinX
0.1 mF
0.1 mF
0.1 mF
0.1 mF
100 W Trace
(Differential)
LVPECL
Ref Clk
LMK04208
Input
CLKinX*
Figure 23. CLKinX/X* Termination for an LVPECL Reference Clock Source
Finally, a reference clock source that produces a differential sine wave output can drive the CLKin pins using the
following circuit. Note: the signal level must conform to the requirements for the CLKin pins listed in Electrical
Characteristics.
CLKinX
0.1 mF
100 W Trace
LMK04208
Input
(Differential)
0.1 mF
Differential
Sinewave Clock
Source
CLKinX*
Figure 24. CLKinX/X* Termination for a Differential Sinewave Reference Clock Source
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Application Information (continued)
9.1.2.2 Driving CLKin Pins with a Single-Ended Source
The CLKin pins of the LMK04208 can be driven using a single-ended reference clock source, for example, either
a sine wave source or an LVCMOS/LVTTL source. Either AC coupling or DC coupling may be used. In the case
of the sine wave source that is expecting a 50-Ω load, TI recommends that AC coupling be used as shown in
Figure 25 with a 50-Ω termination.
NOTE
The signal level must conform to the requirements for the CLKin pins listed in Electrical
Characteristics. CLKinX_BUF_TYPE in Register 11 is recommended to be set to bipolar
mode (CLKinX_BUF_TYPE = 0).
0.1 mF
50 W Trace
CLKinX
50 W
LMK04208
Clock Source
CLKinX*
0.1 mF
Figure 25. CLKinX/X* Single-Ended Termination
If the CLKin pins are being driven with a single-ended LVCMOS/LVTTL source, either DC coupling or AC
coupling may be used. If DC coupling is used, the CLKinX_BUF_TYPE should be set to MOS buffer mode
(CLKinX_BUF_TYPE = 1) and the voltage swing of the source must meet the specifications for DC coupled,
MOS-mode clock inputs given in the table of Electrical Characteristics. If AC coupling is used, the
CLKinX_BUF_TYPE should be set to the bipolar buffer mode (CLKinX_BUF_TYPE = 0). The voltage swing at
the input pins must meet the specifications for AC coupled, bipolar mode clock inputs given in the table of
Electrical Characteristics. In this case, some attenuation of the clock input level may be required. A simple
resistive divider circuit before the AC coupling capacitor is sufficient.
50 W Trace
CLKinX
LMK04208
LVCMOS/LVTTL
Clock Source
CLKinX*
0.1 mF
Figure 26. DC Coupled LVCMOS/LVTTL Reference Clock
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Application Information (continued)
9.1.3 Termination and Use of Clock Output (Drivers)
When terminating clock drivers keep in mind these guidelines for optimum phase noise and jitter performance:
•
•
Transmission line theory should be followed for good impedance matching to prevent reflections.
Clock drivers should be presented with the proper loads. For example:
–
–
LVDS drivers are current drivers and require a closed current loop.
LVPECL drivers are open emitters and require a DC path to ground.
•
Receivers should be presented with a signal biased to their specified DC bias level (common mode voltage)
for proper operation. Some receivers have self-biasing inputs that automatically bias to the proper voltage
level. In this case, the signal should normally be AC coupled.
It is possible to drive a non-LVPECL or non-LVDS receiver with an LVDS or LVPECL driver as long as the above
guidelines are followed. Check the datasheet of the receiver or input being driven to determine the best
termination and coupling method to be sure that the receiver is biased at its optimum DC voltage (common mode
voltage). For example, when driving the OSCin/OSCin* input of the LMK04208 , OSCin/OSCin* should be AC
coupled because OSCin/OSCin* biases the signal to the proper DC level (See Figure 40) This is only slightly
different from the AC coupled cases described in Driving CLKin Pins with a Single-Ended Source because the
DC blocking capacitors are placed between the termination and the OSCin/OSCin* pins, but the concept remains
the same. The receiver (OSCin/OSCin*) sets the input to the optimum DC bias voltage (common mode voltage),
not the driver.
9.1.3.1 Termination for DC Coupled Differential Operation
For DC coupled operation of an LVDS driver, terminate with 100 Ω as close as possible to the LVDS receiver as
shown in Figure 27.
CLKoutX
100W Trace
(Differential)
LVDS
Receiver
LVDS
Driver
CLKoutX*
Figure 27. Differential LVDS Operation, DC Coupling, No Biasing of the Receiver
For DC coupled operation of an LVPECL driver, terminate with 50 Ω to VCC - 2 V as shown in Figure 28.
Alternatively, terminate with a Thevenin equivalent circuit (120-Ω resistor connected to VCC and an 82-Ω resistor
connected to ground with the driver connected to the junction of the 120-Ω and 82-Ω resistors) as shown in
Figure 29 for VCC = 3.3 V.
Vcc - 2 V
CLKoutX
100W Trace
(Differential)
LVPECL
Driver
LVPECL
Receiver
CLKoutX*
Vcc - 2 V
Figure 28. Differential LVPECL Operation, DC Coupling
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Application Information (continued)
Vcc
CLKoutX
100W Trace
(Differential)
LVPECL
Driver
LVPECL
Receiver
CLKoutX*
Vcc
Figure 29. Differential LVPECL Operation, DC Coupling, Thevenin Equivalent
9.1.3.2 Termination for AC Coupled Differential Operation
AC coupling allows for shifting the DC bias level (common mode voltage) when driving different receiver
standards. Since AC coupling prevents the driver from providing a DC bias voltage at the receiver it is important
to ensure the receiver is biased to its ideal DC level.
When driving non-biased LVDS receivers with an LVDS driver, the signal may be AC coupled by adding DC
blocking capacitors, however the proper DC bias point needs to be established at the receiver. One way to do
this is with the termination circuitry in Figure 30.
0.1 mF
100W Trace
CLKoutX
(Differential)
LVDS
Receiver
LVDS
Driver
Vbias
CLKoutX*
0.1 mF
Figure 30. Differential LVDS Operation, AC Coupling,
External Biasing at the Receiver
Some LVDS receivers may have internal biasing on the inputs. In this case, the circuit shown in Figure 30 is
modified by replacing the 50-Ω terminations to Vbias with a single 100-Ω resistor across the input pins of the
receiver, as shown in Figure 31. When using AC coupling with LVDS outputs, there may be a startup delay
observed in the clock output due to capacitor charging. The previous figures employ a 0.10-µF capacitor. This
value may need to be adjusted to meet the startup requirements for a particular application.
0.1 mF
100W Trace
(Differential)
LVDS
Receiver
LVDS
Driver
0.1 mF
Figure 31. LVDS Termination for a Self-Biased Receiver
100
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Application Information (continued)
LVPECL drivers require a DC path to ground. When AC coupling an LVPECL signal use 120-Ω emitter resistors
close to the LVPECL driver to provide a DC path to ground as shown in Figure 32. For proper receiver operation,
the signal should be biased to the DC bias level (common mode voltage) specified by the receiver. The typical
DC bias voltage for LVPECL receivers is 2 V. A Thevenin equivalent circuit (82-Ω resistor connected to VCC and
a 120-Ω resistor connected to ground with the driver connected to the junction of the 82-Ω and 120-Ω resistors)
is a valid termination as shown in Figure 32 for VCC = 3.3 V. Note this Thevenin circuit is different from the DC
coupled example in Figure 29.
Vcc
CLKoutX
0.1 mF
100W Trace
(Differential)
LVPECL
Receiver
LVPECL
Driver
0.1 mF
CLKoutX*
Vcc
Figure 32. Differential LVPECL Operation, AC Coupling, Thevenin Equivalent,
External Biasing at the Receiver
9.1.3.3 Termination for Single-Ended Operation
A balun can be used with either LVDS or LVPECL drivers to convert the balanced, differential signal into an
unbalanced, single-ended signal.
It is possible to use an LVPECL driver as one or two separate 800 mVpp signals. When using only one LVPECL
driver of a CLKoutX/CLKoutX* pair, be sure to properly terminate the unused driver. When DC coupling one of
the LMK04208 clock LVPECL drivers, the termination should be 50 Ω to VCC - 2 V as shown in Figure 33. The
Thevenin equivalent circuit is also a valid termination as shown in Figure 34 for Vcc = 3.3 V.
Vcc - 2V
CLKoutX
50W Trace
LVPECL
Driver
Vcc - 2V
Load
CLKoutX*
50W
Figure 33. Single-Ended LVPECL Operation, DC Coupling
Vcc
CLKoutX
Vcc
50W Trace
LVPECL
Driver
CLKoutX*
Load
Figure 34. Single-Ended LVPECL Operation, DC Coupling,
Thevenin Equivalent
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Application Information (continued)
When AC coupling an LVPECL driver use a 120-Ω emitter resistor to provide a DC path to ground and ensure a
50-Ω termination with the proper DC bias level for the receiver. The typical DC bias voltage for LVPECL
receivers is 2 V (See Driving CLKin Pins with a Single-Ended Source). If the companion driver is not used it
should be terminated with either a proper AC or DC termination. This latter example of AC coupling a single-
ended LVPECL signal can be used to measure single-ended LVPECL performance using a spectrum analyzer or
phase noise analyzer. When using most RF test equipment no DC bias point (0 VDC) is required for safe and
proper operation. The internal 50-Ω termination of the test equipment correctly terminates the LVPECL driver
being measured as shown in Figure 35.
CLKoutX
50W Trace
0.1 mF
LVPECL
Driver
0.1 mF
CLKoutX*
Load
Figure 35. Single-Ended LVPECL Operation, AC Coupling
102
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Application Information (continued)
9.1.4 Frequency Planning with the LMK04208
Calculating the value of the output dividers for use with the LMK04208 is simple due to the architecture of the
LMK04208. That is, the VCO divider may be bypassed and the clock output dividers allow for even and odd
output divide values from 2 to 1045. For most applications, TI recommends bypassing the VCO divider.
The procedure for determining the needed LMK04208 device and clock output divider values for a set of clock
output frequencies is straightforward.
1. Calculate the least common multiple (LCM) of the clock output frequencies.
2. Determine which VCO ranges will support the target clock output frequencies given the LCM.
3. Determine the clock output divide values based on VCO frequency.
4. Determine the PLL2_P, PLL2_N, and PLL2_R divider values given the OSCin VCXO or crystal frequency
and VCO frequency.
For example, given the following target output frequencies: 200 MHz, 120 MHz, and 25 MHz with a VCXO
frequency of 40 MHz:
•
First determine the LCM of the three frequencies. LCM(200 MHz, 120 MHz, 25 MHz) = 600 MHz. The LCM
frequency is the lowest frequency for which all of the target output frequencies are integer divisors of the
LCM. Note: if there is one frequency which causes the LCM to be very large, greater than 3 GHz for
example, determine if there is a single frequency requirement which causes this. It may be possible to select
the VCXO/crystal frequency to satisfy this frequency requirement through OSCout or CLKout3/4 driven by
OSCin. In this way it is possible to get non-integer related frequencies at the outputs.
•
•
Second, since the LCM is not in a VCO frequency range supported by the LMK04208, multiply the LCM
frequency by an integer which causes it to fall into a valid VCO frequency range of an LMK04208 device. In
this case 600 MHz * 5 = 3000 MHz which is valid for the LMK04208.
Third, continuing the example by using a VCO frequency of 3000 MHz and the LMK04208, the CLKout
dividers can be calculated by simply dividing the VCO frequency by the output frequency. To output 200 MHz,
120 MHz, and 25 MHz the output dividers will be 12, 20, and 96 respectively.
–
–
–
3000 MHz / 200 MHz = 15
3000 MHz / 120 MHz = 25
3000 MHz / 25 MHz = 120
•
Fourth, PLL2 must be locked to its input reference. Refer to PLL Programming for more information on this
topic. By programming the clock output dividers and the PLL2 dividers the VCO can lock to the frequency of
3000 MHz and the clock outputs dividers will each divide the VCO frequency down to the target output
frequencies of 200 MHz, 120 MHz, and 25 MHz.
Refer to AN-1865, Frequency Synthesis and Planning for PLL Architectures for more information on this topic
and LCM calculations.
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Application Information (continued)
9.1.5 PLL Programming
To lock a PLL the divided reference and divided feedback from VCO or VCXO must result in the same phase
detector frequency. The tables below illustrate how the divides are structured for the reference path (R) and
feedback path (N) depending on the MODE of the device.
Table 109. PLL1 Phase Detector Frequency — Reference Path (R)
MODE
PLL1 PDF (R) =
All
CLKinX Frequency / (CLKinX_PreR_DIV * PLL1_R)
Table 110. PLL1 Phase Detector Frequency — Feedback Path (N)
MODE
VCO_MUX
—
OSCout
Bypass
Divided
—
PLL1 PDF (N) =
VCXO Frequency / PLL1_N
Dual PLL, Internal VCO
—
VCXO Frequency / (OSCin_DIV * PLL1_N)
VCO Frequency / (CLKoutX_DIV * PLL1_N)
(1)
Bypass
Divided
—
Dual PLL, Internal VCO, 0-Delay
Dual PLL, External VCO, 0-Delay
(1)
—
VCO Frequency / (VCO_DIV * CLKoutX_DIV * PLL1_N)
(1)
—
VCO Frequency / (CLKoutX_DIV * PLL1_N)
(1) The actual CLKoutX_DIV used is selected by the feedback mux. See EN_FEEDBACK_MUX.
Table 111. PLL2 Phase Detector Frequency — Reference Path (R)
EN_PLL2_REF_2X
Disabled
PLL2 PDF (R) =
OSCin Frequency / PLL2_R(1)(2)
Enabled
OSCin Frequency * 2 / PLL2_R(1)(2)
(1) For applications in which the OSCin frequency and PLL2 phase detector frequency are equal, the best PLL2 in-band noise can be
achieved when the doubler is enabled (EN_PLL2_REF_2X = 1) and the PLL2 R divide value is 2. Do not use doubler disabled
(EN_PLL2_REF_2X = 0) and PLL2 R divide value of 1.
(2) See PLL2 Frequency Doubler
Table 112. PLL2 Phase Detector Frequency — Feedback Path (N)
MODE
VCO_MUX
PLL2 PDF (N) =
Dual PLL, Internal VCO
Dual PLL, Internal VCO, 0-Delay
Single PLL, Internal VCO
Dual PLL, Internal VCO
VCO
VCO Frequency / (PLL2_P * PLL2_N)
Dual PLL, Internal VCO, 0-Delay
Single PLL, Internal VCO
Dual PLL, External VCO
Dual PLL External VCO, 0-Delay
Single PLL, External VCO
VCO Divider VCO Frequency / (VCO_DIV * PLL2_P * PLL2_N)
—
VCO Frequency / (PLL2_P * PLL2_N)
VCO
VCO Frequency / (CLKoutX_DIV * PLL2_N)
Single PLL, Internal VCO, 0-Delay
VCO Divider VCO Frequency / (VCO_DIV * CLKoutX_DIV * PLL2_N)
Table 113. PLL2 Phase Detector Frequency — Feedback Path (N) During VCO Frequency Calibration
MODE
VCO_MUX
VCO
PLL2 PDF (N_CAL) =
VCO Frequency / (PLL2_P * PLL2_N_CAL)
VCO Frequency / (VCO_DIV * PLL2_P * PLL2_N_CAL)
All Internal VCO
Modes
VCO Divider
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9.1.5.1 Example PLL2 N Divider Programming
To program PLL2 to lock an LMK04208 using Dual PLL mode to a VCO frequency of 3000 MHz using a 40 MHz
VCXO reference, first determine the total PLL2 N divide value. This is VCO Frequency / PLL2 phase detector
frequency. This example assumes the PLL2 reference frequency doubler is enabled and a PLL2 R divide value
of 2 (see Note 1 in Table 111) which results in PLL2 phase detector frequency the same as PLL2 reference
frequency (40 MHz). 3000 MHz / 40 MHz = 75, so the total PLL2 N divide value is 75.
The dividers in the PLL2 N feedback path for Dual PLL mode include PLL2_P and PLL2_N. PLL2_P can be
programmed from 2 to 8, including both even and odd values. PLL2_N can be programmed from 1 to 263,143,
including both even and odd values. Since the total PLL2 N divide value of 75 contains the factors 3, 5, and 5, it
would be allowable to program PLL2_P to 3 or 5. It is simplest to use the smallest divide, so PLL2_P = 3, and
PLL2_N = 25 which results in a Total PLL2 N = 75.
For this example and in most cases, PLL2_N_CAL will have the same value as PLL2_N. However when using
Single PLL mode with 0-delay, the values will differ. When using an external VCO, PLL2_N_CAL value is
unused.
9.1.6 Digital Lock Detect Frequency Accuracy
The digital lock detect circuit is used to determine PLL1 locked, PLL2 locked, and holdover exit events. A window
size and lock count register are programmed to set a ppm frequency accuracy of reference to feedback signals
of the PLL for each event to occur. When a PLL digital lock event occurs the PLL's digital lock detect is asserted
true. When the holdover exit event occurs, the device will exit holdover mode.
Table 114. Digital Lock Detect Frequency Accuracy Table
EVENT
PLL
PLL1
PLL2
PLL1
WINDOW SIZE
PLL1_WND_SIZE
PLL2_WND_SIZE
PLL1_WND_SIZE
LOCK COUNT
PLL1_DLD_CNT
PLL1 Locked
PLL2 Locked
Holdover exit
PLL2_DLD_CNT
HOLDOVER_DLD_CNT
For a digital lock detect event to occur there must be a “lock count” number of phase detector cycles of PLLX
during which the time/phase error of the PLLX_R reference and PLLX_N feedback signal edges are within the
user programmable "window size." Since there must be at least "lock count" phase detector events before a lock
event occurs, a minimum digital lock event time can be calculated as "lock count" / fPDX where X = 1 for PLL1 or
2 for PLL2.
By using Equation 5, values for a "lock count" and "window size" can be chosen to set the frequency accuracy
required by the system in ppm before the digital lock detect event occurs:
2e6 × PLLX_WND_SIZE × fPDX
ppm =
PLLX_DLD_CNT
(5)
The effect of the "lock count" value is that it shortens the effective lock window size by dividing the "window size"
by "lock count".
If at any time the PLLX_R reference and PLLX_N feedback signals are outside the time window set by "window
size", then the “lock count” value is reset to 0.
9.1.6.1 Minimum Lock Time Calculation Example
The minimum time for PLL2 digital lock to be asserted can be calculated by Equation 6:
PLL2_DLD_CNT / Phase detector frequency
(6)
Given a PLL2 phase detector frequency of 40 MHz and PLL2_DLD_CNT value of 10,000, the minimum digital
lock detect time of PLL2 will be 10,000 / 40 MHz = 250 μs.
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9.1.7 Calculating Dynamic Digital Delay Values for Any Divide
This section explains how to calculate the dynamic digital delay for any divide value.
Dynamic digital delay allows the time offset between two or more clock outputs to be adjusted with no or minimal
interruption of clock outputs. Since the clock outputs are operating at a known frequency, the time offset can also
be expressed as a phase shift. When dynamically adjusting the digital delay of clock outputs with different
frequencies the phase shift should be expressed in terms of the higher frequency clock. The step size of the
smallest time adjustment possible is equal to half the period of the Clock Distribution Path, which is the VCO
frequency (Equation 3) or the VCO frequency divided by the VCO divider (Equation 4) if not bypassed. The
smallest degree phase adjustment with respect to a clock frequency will be 360 * the smallest time adjustment *
the clock frequency. The total number of phase offsets that the LMK04208 is able to achieve using dynamic
digital delay is equal 1 / (higher clock frequency * the smallest phase adjustment).
Equation 7 calculates the digital delay value that must be programmed for a synchronizing clock to achieve a 0
time/phase offset from the qualifying clock. Once this digital delay value is known, it is possible to calculate the
digital delay value for any phase offset. The qualifying clock for dynamic digital delay is selected by the
FEEDBACK_MUX. When dynamic digital delay is engaged with same clock output used for the qualifying clock
and the new synchronized clock, it is termed relative dynamic digital delay since causing another SYNC event
with the same digital delay value will offset the clock by the same phase once again. The important part of
relative dynamic digital delay is that the CLKoutX_HS must be programmed correctly when the SYNC event
occurs (Table 6). This can result in needing to program the device twice. Once to set the new CLKoutX_DDLY
with CLKoutX_HS as required for the SYNC event, and again to set the CLKoutX_HS to its desired value.
Digital delay values are programmed using the CLKoutX_DDLY and CLKoutX_HS registers as shown in
Equation 8. For example, to achieve a digital delay of 13.5, program CLKoutX_DDLY = 14 and CLKoutX_HS = 1.
≈
∆
’
≈
∆
’
◊
»
…
ÿ
Ÿ
16
0 digital delay =
ì CLKoutX_Y_DIV
+ 0.5
÷- 11.5
÷
CLKoutX_Y_DIV
…
Ÿ
«
«
◊
(7)
Equation 7 uses the ceiling operator. To find the ceiling of a fractional number round up. An integer remains the
same value.
Digital delay = CLKoutX_DDLY - (0.5 * CLKoutX_HS)
(8)
Note: since the digital delay value for 0 time/phase offset is a function of the qualifying clock's divide value, the
resulting digital delay value can be used for any clock output operating at any frequency to achieve a 0
time/phase offset from the qualifying clock. Therefore the calculated time shift table will also be the same as in
Table 115.
9.1.7.1 Example
Consider a system with:
•
•
•
•
A VCO frequency of 3000 MHz.
The VCO divider is bypassed, therefore the clock distribution path frequency is 3000 MHz.
CLKout0_DIV = 10 resulting in a 300 MHz frequency on CLKout0.
CLKout1_DIV = 20 resulting in a 150 MHz frequency on CLKout2.
For this system the minimum time adjustment is ~0.16667 ns, which is 0.5 / (3000 MHz). Since the higher
frequency is 300 MHz, phase adjustments will be calculated with respect to the 300 MHz frequency. The 0.25 ns
minimum time adjustment results in a minimum phase adjustment of 18 degrees, which is 360 degrees / 200
MHz * 0.25 ns.
To calculate the digital delay value to achieve a 0 time/phase shift of CLKout2 when CLKout0 is the qualifying
clock. Solve Equation 7 using the divide value of 10. To solve the equation 16/10 = 1.6, the ceiling of 1.6 is 2.
Then to finish solving the equation solve (2 + 0.5) * 10 - 11.5 = 13.5. A digital delay value of 13.5 is programmed
by setting CLKout1_DDLY = 14 and CLKout1_HS = 1.
To calculate the digital delay value to achieve a 0 time/phase shift of CLKout0 when CLKout2 is the qualifying
clock, solve Equation 7 using the divide value of CLKout2, which is 20. This results in a digital delay of 18.5
which is programmed as CLKout0_DDLY = 19 and CLKout0_HS = 1.
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Once the 0 time/phase shift digital delay programming value is known a table can be constructed with the digital
delay value to be programmed for any time/phase offset by decrementing or incrementing the digital delay value
by 0.5 for the minimum time/phase adjustment.
A complete filled out table for use of CLKout0 as the qualifying clock is shown in Table 115. It was created by
entering a digital delay of 13.5 for 0 degree phase shift, then decrementing the digital delay down to the minimum
value of 4.5. Since this did not result in all the possible phase shifts, the digital delay was then incremented from
13.5 to 14.0 to complete all possible phase shifts.
Table 115. Example Digital Delay Calculation
CALCULATED TIME SHIFT
(ns)
RELATIVE TIME SHIFT
to 300 MHz (ns)
PHASE SHIFT
of 300 MHz (Degrees)
DIGITAL DELAY
4.5
5
-3.0
-2.833...
-2.666...
-2.5
0.333...
0.5
36
54
5.5
6
0.666...
0.833...
1.0
72
90
6.5
7
-2.333
-2.166
-2.0
108
126
144
162
180
198
216
234
252
270
288
306
324
342
0
1.166...
1.333...
1.5
7.5
8
-1.833
-1.666...
-1.5
8.5
9
1.666...
1.833...
2.0
9.5
10
-1.333...
-1.166...
-1.0
2.166...
2.333...
2.5
10.5
11
-0.833...
-0.666...
-0.5
11.5
12
2.666...
2.833...
3.0
12.5
13
-0.333...
-0.166...
0
3.166...
0
13.5
14
0.166...
0.333...
0.166...
0.333...
18
14.5
36
Observe that the digital delay value of 4.5 and 14.5 will achieve the same relative time shift/phase delay.
However programming a digital delay of 14.5 will result in a clock off time for the synchronizing clock to achieve
the same phase time shift/phase delay.
Digital delay value is programmed as CLKoutX_DDLY - (0.5 * CLKoutX_HS). So to achieve a digital delay of
13.5, program CLKoutX_DDLY = 14 and CLKoutX_HS = 1. To achieve a digital delay of 14, program
CLKoutX_DDLY = 14 and CLKoutX_HS = 0.
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9.1.8 Optional Crystal Oscillator Implementation (OSCin/OSCin*)
The LMK04208 features supporting circuitry for a discretely implemented oscillator driving the OSCin port pins.
Figure 36 illustrates a reference design circuit for a crystal oscillator:
OSCin*
C
C1
= 2.2 nF
C
opt
R1 = 4.7k
SMV1249-074LF
R3 = 10k
LMK04208
XTAL
1 nF
R2 = 4.7k
C
C2
= 2.2 nF
OSCin
C
opt
PLL1 Loop Filter
Figure 36. Reference Design Circuit for Crystal Oscillator Option
This circuit topology represents a parallel resonant mode oscillator design. When selecting a crystal for parallel
resonance, the total load capacitance, CL, must be specified. The load capacitance is the sum of the tuning
capacitance (CTUNE), the capacitance seen looking into the OSCin port (CIN), and stray capacitance due to PCB
parasitics (CSTRAY), and is given by Equation 9.
CSTRAY
CL = CTUNE + CIN
+
2
(9)
CTUNE is provided by the varactor diode shown in Figure 36, Skyworks model SMV1249-074LF. A dual diode
package with common cathode provides the variable capacitance for tuning. The single diode capacitance
ranges from approximately 31 pF at 0.3 V to 3.4 pF at 3 V. The capacitance range of the dual package (anode to
anode) is approximately 15.5 pF at 3 V to 1.7 pF at 0.3 V. The desired value of VTUNE applied to the diode should
be VCC/2, or 1.65 V for VCC = 3.3 V. The typical performance curve from the data sheet for the SMV1249-074LF
indicates that the capacitance at this voltage is approximately 6 pF (12 pF / 2).
The nominal input capacitance (CIN) of the LMK04208 OSCin pins is 6 pF. The stray capacitance (CSTRAY) of the
PCB should be minimized by arranging the oscillator circuit layout to achieve trace lengths as short as possible
and as narrow as possible trace width (50 Ω characteristic impedance is not required).
As an example, assume that CSTRAY is 4 pF. The total load capacitance is nominally:
4
2
= 14 pF
CL = 6 + 6 +
(10)
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Consequently the load capacitance specification for the crystal in this case should be nominally 14 pF.
The 2.2-nF capacitors shown in the circuit are coupling capacitors that block the DC tuning voltage applied by
the 4.7-kΩ and 10-kΩ resistors. The value of these coupling capacitors should be large, relative to the value of
CTUNE (CC1 = CC2 >> CTUNE), so that CTUNE becomes the dominant capacitance.
For a specific value of CL, the corresponding resonant frequency (FL) of the parallel resonant mode circuit is:
1
+ 1
C1
2(C0 + CL1
C0
CL
C1
+ 1
= FS À
≈
’
÷
◊
FL = FS
À
)
+
2 ∆C
1
«
where
•
•
•
•
FS = Series resonant frequency
C1 = Motional capacitance of the crystal
CL = Load capacitance
C0 = Shunt capacitance of the crystal, specified on the crystal datasheet
(11)
(12)
The normalized tuning range of the circuit is closely approximated by:
1
1
-
FCL1 - FCL2
C1
2
1
2
1
1
DF
F
C0 CL1
+
C0 CL2
+
À
-
≈
∆
«
’ ≈
’
÷
◊
À
=
=
=
(C0 + CL1
)
(C0 + CL2)
FFCL1
÷ ∆C
C1
C1
C1
1
◊ «
CL1, CL2 = The endpoints of the circuit’s load capacitance range, assuming a variable capacitance element is one
component of the load. FCL1, FCL2 = parallel resonant frequencies at the extremes of the circuit’s load
capacitance range.
A common range for the pullability ratio, C0/C1, is 250 to 280. The ratio of the load capacitance to the shunt
capacitance is ~(n * 1000), n < 10. Hence, picking a crystal with a smaller pullability ratio supports a wider tuning
range because this allows the scale factors related to the load capacitance to dominate.
9.1.8.1 Examples of Phase Noise and Jitter Performance
Examples of the phase noise and jitter performance of the LMK04208 with a crystal oscillator are shown in
Table 116. This table illustrates the clock output phase noise when a 20.48-MHz crystal is paired with PLL1.
Performance of other LMK04208 devices will be similar.
Table 116. Example RMS Jitter and Clock Output Phase Noise for LMK04208
(1)
with a 20.48 MHz Crystal Driving OSCin (T = 25 °C, VCC = 3.3 V)
PLL2 PDF = 20.48 MHz
(EN_PLL2_REF2X = 0,
XTAL_LVL = 3)
PLL2 PDF = 40.96 MHz
(EN_PLL2_REF2X = 1, XTAL_LVL = 3)
INTEGRATION
BANDWIDTH
CLOCK OUTPUT TYPE
fCLK = 245.76 MHz
fCLK = 122.88 MHz
fCLK = 245.76 MHz
RMS JITTER (fs, RMS)
LVCMOS
LVDS
374
419
460
226
231
226
412
421
448
195
205
191
382
372
440
190
194
188
100 Hz – 20 MHz
LVPECL 1.6 Vpp
LVCMOS
10 kHz – 20 MHz
LVDS
LVPECL 1.6 Vpp
(1) Performance data and crystal specifications contained in this section are based on Vectron model VXB1-1150-20M480, 20.48 MHz.
PLL1 has a narrow loop bandwidth, PLL2 loop parameters are: C1 = 150 pF, C2 = 120 nF, R2 = 470 Ω, Charge Pump current = 3.2 mA,
Phase detector frequency = 20.48 MHz or 40.96 MHz, VCO frequency = 2949.12 MHz. Loop filter was optimized for 40.96 MHz phase
detector performance.
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Table 116. Example RMS Jitter and Clock Output Phase Noise for LMK04208
with a 20.48 MHz Crystal Driving OSCin (T = 25 °C, VCC = 3.3 V) (1) (continued)
PLL2 PDF = 20.48 MHz
(EN_PLL2_REF2X = 0,
XTAL_LVL = 3)
PLL2 PDF = 40.96 MHz
INTEGRATION
BANDWIDTH
(EN_PLL2_REF2X = 1, XTAL_LVL = 3)
fCLK = 122.88 MHz fCLK = 245.76 MHz
CLOCK OUTPUT TYPE
fCLK = 245.76 MHz
PHASE NOISE (dBc/Hz)
PLL2 PDF = 20.48 MHz
(EN_PLL2_REF2X = 0,
XTAL_LVL = 3)
PLL2 PDF = 40.96 MHz
(EN_PLL2_REF2X = 1, XTAL_LVL = 3)
Offset
Clock Output Type
fCLK = 245.76 MHz
-87
fCLK = 122.88 MHz
-93
fCLK = 245.76 MHz
-87
LVCMOS
LVDS
100 Hz
1 kHz
-86
-91
-86
LVPECL 1.6 Vpp
LVCMOS
-86
-92
-85
-115
-121
-115
LVDS
-115
-123
-116
LVPECL 1.6 Vpp
LVCMOS
-114
-122
-116
-117
-128
-122
10 kHz
100 kHz
1 MHz
LVDS
-117
-128
-122
LVPECL 1.6 Vpp
LVCMOS
-117
-128
-122
-130
-135
-129
LVDS
-130
-135
-129
LVPECL 1.6 Vpp
LVCMOS
-129
-135
-129
-150
-154
-148
LVDS
-149
-153
-148
LVPECL 1.6 Vpp
LVCMOS
-150
-154
-148
-159
-162
-159
40 MHz
LVDS
-157
-159
-157
LVPECL 1.6 Vpp
-159
-161
-159
110
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Example crystal specifications are presented in Table 117.
Table 117. Example Crystal Specifications
PARAMETER
VALUE
Nominal Frequency (MHz)
20.48
Frequency Stability, T = 25 °C
Operating temperature range
Frequency Stability, -40 °C to +85 °C
Load Capacitance
± 10 ppm
-40 °C to +85 °C
± 15 ppm
14 pF
Shunt Capacitance (C0)
Motional Capacitance (C1)
Equivalent Series Resistance
Drive level
5 pF Maximum
20 fF ± 30%
25 Ω Maximum
2 mWatts Maximum
225 typical, 250 Maximum
C0/C1 ratio
See Figure 37 for a representative tuning curve.
180
140
100
60
20
-20
-60
-100
-140
-180
0.0 0.4 0.8 1.2 1.6 2.0 2.4 2.8 3.2
V
(V)
TUNE
Figure 37. Example Tuning Curve, 20.48-MHz Crystal
The tuning curve achieved in the user's application may differ from the curve shown above due to differences in
PCB layout and component selection.
This data is measured on the bench with the crystal integrated with the LMK04208. Using a voltmeter to monitor
the VTUNE node for the crystal, the PLL1 reference clock input frequency is swept in frequency and the resulting
tuning voltage generated by PLL1 is measured at each frequency. At each value of the reference clock
frequency, the lock state of PLL1 should be monitored to ensure that the tuning voltage applied to the crystal is
valid.
The curve shows over the tuning voltage range of 0.3 VDC to 3.0 VDC, the frequency range is -140 to +91 ppm;
or equivalently, a tuning range of -2850 Hz to +1850 Hz. The measured tuning voltage at the nominal crystal
frequency (20.48 MHz) is 1.7 V. Using the diode data sheet tuning characteristics, this voltage results in a tuning
capacitance of approximately 6.5 pF.
The tuning curve data can be used to calculate the gain of the oscillator (KVCO). The data used in the calculations
is taken from the most linear portion of the curve, a region centered on the crossover point at the nominal
frequency (20.48 MHz). For a well designed circuit, this is the most likely operating range. In this case, the tuning
range used for the calculations is ± 1000 Hz (± 0.001 MHz), or ± 81.4 ppm. The simplest method is to calculate
the ratio:
DF2 - DF1
≈
= ∆V
«
’
DF
DV
MHz
V
÷,
◊
KVCO
=
TUNE2 - VTUNE1
(13)
111
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ΔF2 and ΔF1 are in units of MHz. Using data from the curve this becomes:
0.001 - (-0.001)
2.03 - 0.814
MHz
V
= 0.00164
(14)
A second method uses the tuning data in units of ppm:
FNOM À (Dppm2 - Dppm1)
KVCO
=
DV À 106
(15)
(16)
FNOM is the nominal frequency of the crystal and is in units of MHz. Using the data, this becomes:
12.288 À (81.4 - (-81.4))
MHz
= 0.00164,
(2.03 - 0.814) À 106
V
In order to ensure startup of the oscillator circuit, the equivalent series resistance (ESR) of the selected crystal
should conform to the specifications listed in the table of Electrical Characteristics.
It is also important to select a crystal with adequate power dissipation capability, or drive level. If the drive level
supplied by the oscillator exceeds the maximum specified by the crystal manufacturer, the crystal will undergo
excessive aging and possibly become damaged. Drive level is directly proportional to resonant frequency,
capacitive load seen by the crystal, voltage and equivalent series resistance (ESR).
For more complete coverage of crystal oscillator design, see:
Clocks and Timers or AN-1939 Crystal Based Oscillator Design with the LMK04000 Family.
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9.2 Typical Applications
Normal use case of the LMK04208 device is as a dual loop jitter cleaner. This section will discuss a design
example to illustrate the various functional aspects of the LMK04208 device.
PLL1
PLL2
External
Loop Filter
OSCout
External VCXO
or Tunable
Crystal
OSCout*
CLKinX
CLKinX*
R
N
Phase
Detector
PLL1
CPout2
External
Loop Filter
2 inputs
R
N
Phase
Detector
PLL2
Divider
Digital Delay
Analog Delay
Partially
Integrated
Loop Filter
CLKoutX
CLKoutX*
6 outputs
Input
Buffer
Internal
VCO
6 blocks
LMK04208
Figure 38. Simplified Functional Block Diagram for Dual Loop Mode
9.2.1 Design Requirements
Given a remote radio head (RRU) type application which needs to clock some ADCs, DACs, FPGA, SERDES,
and an LO. The input clock will be a recovered clock which needs jitter cleaning. The FPGA clock should have a
clock output on power up. A summary of clock input and output requirements are as follows:
Clock Input:
•
30.72-MHz recovered clock.
Clock Outputs:
•
•
•
•
•
1x 245.76-MHz clock for ADC, LVPECL
2x 983.04-MHz clock for DAC, LVPECL
1x 122.88-MHz clock for FPGA, LVDS. POR clock
1x 122.88-MHz clock for SERDES, LVPECL
2x 122.88-MHz clock for LO, LVCMOS
It is also desirable to have the holdover feature engage if the recovered clock reference is ever lost. The
following information reviews the steps to produce this design.
9.2.2 Detailed Design Procedure
Design of all aspects of the LMK04208 are quite involved and software has been written to assist in part
selection, part programming, loop filter design, and simulation. This design procedure will give a quick outline of
the process.
Note that this information is current as of the date of the release of this datasheet. Design tools receive
continuous improvements to add features and improve model accuracy. Refer to software instructions or training
for latest features.
1. Device Selection
–
the key to device selection is required VCO frequency given required output frequencies. The device
must be able to produce the VCO frequency that can be divided down to required output frequencies.
–
The software design tools will take into account VCO frequency range for specific devices based on the
application's required output frequencies. Using an external VCO provides increased flexibility regarding
valid designs.
–
To understand the process better, refer to Frequency Planning with the LMK04208 for more detail on
calculating valid VCO frequency when using integer dividers using the least common multiple (LCM) of
the output frequencies.
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Typical Applications (continued)
2. Device Configuration
–
There are many possible permutations of dividers and other registers to get same input and output
frequencies from a device. However there are some optimizations and trade-offs to be considered.
–
If more than one divider is in series, for instance VCO divider to CLKout divider, or VCO divider to PLL
prescaler to PLL N. It is possible although not assured that some crosstalk/mixing could be created
when using some divides.
–
–
The design software normally attempts to maximize phase detector frequency, use smallest dividers, and
maximizes PLL charge pump current.
When an external VCXO or crystal is used for jitter cleaning, the design software will choose the
maximum frequency value, depending on design software options, this max frequency may be limited to
standard value VCXOs/Crystals. Note, depending on application, different frequency VCXOs may be
chosen to generate some of the required output frequencies.
–
–
Refer to PLL Programming for divider equations need to ensure PLL is locked. The design software is
able to configure the device for most cases, but at this time for advanced features like 0-delay, the
user must take care to ensure proper PLL programming.
These guidelines may be followed when configuring PLL related dividers or other related registers:
–
For lowest possible in-band PLL flat noise, maximize phase detector frequency to minimize N divide
value.
–
For lowest possible in-band PLL flat noise, maximize charge pump current. The highest value charge
pump currents often have similar performance due to diminishing returns.
–
–
To reduce loop filter component sizes, increase N value and/or reduce charge pump current.
Large capacitors help reduce phase detector spurs at phase detector frequency caused by external
VCOs/VCXOs with low input impedance.
–
As rule of thumb, keeping the phase detector frequency approximately between 10 * PLL loop
bandwidth and 100 * PLL loop bandwidth. A phase detector frequency less than 5 * PLL bandwidth
may be unstable and a phase detector frequency > 100 * loop bandwidth may experience increased
lock time due to cycle slipping. However for clock generation/jitter cleaning applications, lock time is
typically not critical and large phase detector frequencies typically result in reduced PLL noise, so
cycle-slipping during lock is acceptable.
3. PLL Loop Filter Design
–
–
TI recommends using Clock Architect to design your loop filter.
Best loop filter design and simulation can be achieved when:
–
–
Custom reference and VCXO phase noise profiles are loaded into the software.
VCO gain of the external VCXO or possible external VCO device are entered.
–
The design tool will return solutions with high reference/phase detector frequencies and high charge
pump currents by default. It is possible to reduce the phase detector frequency charge pump current in
Clock Architect. Due to the narrow loop bandwidth used on PLL1, it is common to lower the phase
detector frequency and/or charge pump current on PLL1 to reduce component size.
–
–
While designing loop filter, adjusting the charge pump current or N value can help with loop filter
component selection. Lower charge pump currents and larger N values result in smaller component
values but may increase impacts of leakage and reduce PLL phase noise performance.
More detailed understanding of loop filter design can found in Dean Banerjee's PLL Performance,
Simulation, and Design (www.ti.com/tool/pll_book).
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Typical Applications (continued)
4. Clock Output Assignment
–
At this time the design software does not take into account frequency assignment to specific outputs
except to ensure that the output frequencies can be achieved. It is best to consider proximity of each
clock output to each other and other PLL circuitry when choosing final clock output locations. Here are
some guidelines to help achieve best performance when assigning outputs to specific CLKout/OSCout
pins.
–
–
Group common frequencies together.
PLL charge pump circuitry can cause crosstalk at charge pump frequency. Place outputs sharing
charge pump frequency or lower priority outputs not sensitive to charge pump frequency spurs
together.
–
Muxes can create a path for noise coupling. Consider all frequencies which may have some bleed
through from non-selected mux inputs.
–
For example, LMK04208 CLKout3 and CLKout4 share a mux with OSCin.
–
–
Some clock targets require low close-in phase noise. If possible, use a VCXO based PLL1 output from
CLKout3 or CLKout4 for such a clock target. An example is a clock to a PLL reference.
Some clock targets require excellent noise floor performance. Outputs driven by the internal VCO have
the best noise floor performance. An example is an ADC or DAC.
5. Other device specific configuration. For LMK04208, consider the following:
–
PLL lock time based on programming:
–
In addition to the time it takes the device to lock to frequency, there is a digital filter to avoid false lock
time detects which can also be used to ensure a specific PPM frequency accuracy. This also impacts
the time it takes for the digital lock detect (DLD) pin to be asserted. Refer to Digital Lock Detect
Frequency Accuracy for more information.
–
Holdover configuration:
–
Specific PPM frequency accuracy required to exit holdover can be programmed. Refer to Digital Lock
Detect Frequency Accuracy for more information.
–
–
Digital delay: phase alignment of the output clocks.
Analog delay: another method to shift phases of clocks with finer resolution with the penalty of increase
noise floor.
–
Dynamic digital delay: ability to shift phase alignment of clocks with minimum disruption during operation.
6. Device Programming
The software tool TICS Pro for EVM programming can be used to setup the device in the desired
configuration, then export a hex register map suitable for use in application.
–
Some additional information on each part of the design procedure for the RRU example is below.
9.2.2.1 Device Selection
Use the WEBENCH Clock Architect Tool. Enter the required frequencies and formats into the tool. To use this
device, find a solution using the LMK04208.
9.2.2.1.1 Clock Architect
When viewing resulting solutions, it is possible to narrow the parts used in the solution by setting a filter.
Filtering of a specific device can be done by selecting the device from the filter combo box. Also, regular
expressions can be typed into filter combo box. LMK04208 will filter for only the LMK04208 device.
To simulate single loop solutions with dual loop device, set PLL1 loop filter to a very narrow or "0 Hz LBW."
9.2.2.1.2 Calculation Using LCM
In this example, the LCM(245.76 MHz, 983.04 MHz, 122.88 MHz) = 983.04 MHz. A valid VCO frequency for
LMK04208 is 2949.12 MHz = 3 * 983.04 MHz. Therefore the LMK04208 may be used to produce these output
frequencies.
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Typical Applications (continued)
9.2.2.2 Device Configuration
The tools automatically configure the simulation to meet the input and output frequency requirements given and
make assumptions about other parameters to give some default simulations. The assumptions made are to
maximize input frequencies, phase detector frequencies, and charge pump currents while minimizing VCO
frequency and divider values.
9.2.2.2.1 PLL LO Reference
PLL1 outputs have the best phase noise performance for LO references. As such OSCout, or CLKout3/CLKout4
(with CLKout#_OSCin_Sel field selecting OSCin clock source) can be used to provide the 122.88 MHz LO
reference clock. To achieve this with a 245.76 MHz VCXO the OSCout_DIV can be set to 2 to provide 122.88
MHz at OSCout. CLKout3/4_DIV can be set to 2 for 122.88 MHz output if LO references are clocked from
CLKout3/4.
In the next section it is determined that for the POR clock, a 122.88 MHz VCXO will be used. This means no
division will be needed to provide 122.88 MHz.
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Typical Applications (continued)
9.2.2.2.2 POR Clock
If OSCout is to be used for LVDS POR 122.88 MHz clock, the POR value of the OSCout_DIV is 1, so a 122.88
MHz VCXO frequency must be chosen. This may be desired anyway since the phase detector frequency is
limited to 122.88 MHz and lower frequency VCXOs tend to cost less. With this change the OSCin frequency and
phase detector frequency are the same, so the doubler must be enabled and the PLL2 R divider programmed =
2 to follow the rule stated in PLL2 Frequency Doubler.
Note: it is possible to set the PLL2 R = 0.5 to simulate the doubler in-case lower frequency VCXOs would like to
be simulated. For example a 61.44 MHz VCXO could be used while retaining a 122.88 MHz phase detector
frequency. However, it would reduce the LO reference frequency and POR clock frequency to 61.44 MHz.
At this time, the VCXO frequency and phase detector frequency is chosen, so loop filter design may begin.
9.2.2.3 PLL Loop Filter Design
The PLL structure for the LMK04208 is illustrated in Loop Filter.
At this time the user may choose to make adjustments to the simulation tools for more accurate simulations to
their application. For example:
•
Clock Architect allows loading a custom phase noise plot for any block. Typically, a custom phase noise plot
is entered for CLKin to match the reference phase noise to the device; a phase noise plot for the VCXO can
additionally be provided to match the performance of VCXO used. For improved accuracy in simulation and
optimum loop filter design, be sure to load these custom noise profiles for use in application. After loading a
phase noise plot, user should recalculate the recommended loop filter design.
•
The Clock Architect will return solutions with high reference/phase detector frequencies by default. The user
may decrease the phase detector frequency if desired. Be sure to decrease by integer relationships with the
reference as an integer divider will be used. Due to the narrow loop bandwidth used on PLL1, it is common to
reduce the phase detector frequency on PLL1 by increasing PLL1 R.
For this example, for PLL1 to perform jitter cleaning and to minimize jitter from PLL2 used for frequency
multiplication:
•
PLL1: A narrow loop bandwidth PLL1 filter was design by updating the loop bandwidth to 50 Hz and phase
margin to 50 degrees.
•
PLL2:
–
VCXO noise profile is measured, then loaded into VCXO phase noise profile in Clock Architect. Be sure
that the VCO frequency of PLL1 is as desired. If changing the VC(X)O frequency of PLL1, be sure to the
PLL2 Phase Detector frequency aligns with an integer divider.
–
The recommended loop filter is redesigned. Updates to the PLL1 loop filter and VCXO phase noise may
change the loop filter recommendation, so PLL2 loop filter may need to be recalculated.
The next two sections will discuss PLL1 and PLL2 loop filter design specific to this example using default phase
noise profiles.
NOTE
Clock Architect provides some recommend loop filters upon first load of the simulation.
Anytime PLL related inputs change like an input phase noise, charge pump current,
divider values, and so forth. it is best to re-design the PLL1 loop filter to the recommended
design or your desired parameters. After PLL1, then update the PLL2 loop filter in the
same way to keep the loop filters designed and optimized for the application. Since PLL1
loop filter design may impact PLL2 loop filter design, be sure to update the designs in
order.
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Typical Applications (continued)
9.2.2.3.1 PLL1 Loop Filter Design
For this example, in the Clock Architect tool update the loop bandwidth for 0.05 kHz and the phase margin for 50
degrees and press "Choose RC Components for me." With the 30.72 MHz phase detector frequency and 1.6 mA
charge pump; the designed loop filter's largest capacitor, C2, is 27 µF. Supposing a goal of < 10 µF; setting PLL1
R = 4 and pressing the calculate again shows that C2 is 6.8 µF. Suppose that a reduction to < 1 µF is desired,
continuing to increase the PLL1 R to 8 resulting in a phase detector frequency of 3.84 MHz and reducing the
charge pump current from 1.6 mA to 0.4 mA and calculating again shows that C2 is 820 nF. As N was increased
and charge pump decreased, this final design has R2 = 12 kΩ. The first design with low N value and high charge
pump current result in R2 = 390 Ω. The impact of the thermal resistance is calculated in the tool. Viewing the
simulation of the loop filter with the 12-kΩ resistor shows that the thermal noise in the loop is not impacting
performance.
It may be desired to design a 3rd order loop filter for additional attenuation input noise and spurs
With the PLL1 loop filter design complete, PLL2's loop filter is ready to be designed.
9.2.2.3.2 PLL2 Loop Filter Design
In Clock Architect, select LOOPFILTER2 tab under Loop Filters tab. Click "Choose RC Components for me." For
PLL2's loop filter maximum phase detector frequency and maximum charge pump current are typically used.
Typically the jitter integration bandwidth includes the loop filter bandwidth for PLL2. The recommended loop filter
by the tools are designed to minimize jitter. The integrated loop filter components are minimized with this
recommendation as to allow maximum flexibility in achieve wide loop bandwidths for low PLL noise. With the
recommended loop filter calculated, this loop filter is ready to be simulated.
If using integrated components is desired, make adjustments to the integrated components. The effective loop
bandwidth and phase margin with these updates is calculated every time "Update Actual Loop Parameters" is
clicked. The integrated loop filter components are good to use when attempting to eliminate some spurs since
they provide filtering after the bond wires. The recommended procedure is to increase C3/C4 capacitance, then
R3/R4 resistance. Large R3/R4 resistance can result in degraded VCO phase noise performance.
9.2.2.4 Clock Output Assignment
At this time Clock Architect only assign outputs to specific clock outputs numerically; not necessarily by optimum
configuration. The user may wish to make some educated re-assignment of outputs.
During device configuration, some output assignment was discussed since it impacted the part's configuration
relating to loop filter design, such as:
•
In this example, OSCout can be used to provide the power on reset (POR) start-up clock to the FPGA at
122.88 MHz since the VCXO frequency is the same as the output frequency.
•
Since PLL1 outputs have best in-band noise, CLKout3 is used with CLKout3_OSCin_Sel = 0x01 (OSCin) to
provide a PLL1 based output. LVCMOS (Norm/Inv) is used instead of LVCMOS (Norm/Norm) to reduce
crosstalk. If OSCout was not needed for FPGA start-up clock, OSCout could have been used to provide the
LO reference clocks with lower noise floor, but close-in noise is typically of more concern since noise above
the loop bandwidth of the LO will be dominated by the VCO of the LO. See Figure 39.
Since CLKout3 and CLKout4 have a mux allowing them to be driven by the VCXO and due there is a chance for
some 122.88 MHz crosstalk from the VCXO. The 122.88 MHz SERDES clock will be placed on CLKout4 since it
will not be sensitive to crosstalk as it is operating at the same frequency.
Three converter clocks still need to be assigned. The 245.76 MHz ADC clock and two 983.04 MHz DAC. There
are four remaining clock outputs. To maximize distance of the ADC clock from other clocks which could create
sub-harmonic spurs, CLKout0 is chosen for ADC at 245.76 MHz clock. CLKout1 and CLKout2 are chosen for the
DAC 983.04 MHz clocks. Because the ADC clock is often the most sensitive to sub-harmonic spurs, the goal
was to place the ADC clock as far as possible from other clocks which could result in sub-harmonic spurs.
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Typical Applications (continued)
9.2.2.5 Other Device Specific Configuration
9.2.2.5.1 Digital Lock Detect
Digital lock time for PLL1 will ultimately depend upon the programming of the PLL1_DLD_CNT register as
discussed in Digital Lock Detect Frequency Accuracy. Since the PLL1 phase detector frequency in this example
is 3.84 MHz, the lock time will = 1 / (PLL1_DLD_CNT * 3.84 MHz)
Digital lock time for PLL1 if PLL1_DLD_CNT = 10000 is just over 2.6 ms. When using holdover, it is very
important to program the PLL1_DLD_CNT to a value large enough to prevent false digital lock detect signals.
If PLL1_DLD_CNT is too small, when the device exits holdover and is re-locking, the DLD will go high while the
phase of the reference and feedback are within the specified window size because the programmed
PLL1_DLD_CNT will be satisfied. However, if the loop has not yet settled to without the window size, when the
phases of the reference and feedback once again exceed the window size, the DLD will return low. Provided that
DISABLE_DLD1_DET = 0, the device once again enter holdover. Assuming that the reference clock is valid
because holdover was just exited, the exit criteria will again be met, holdover will exit, and PLL1 will start locking.
Unfortunately, the same sequence of events will repeat resulting in oscillation out-of and back-into holdover.
Setting the PLL1_DLD_CNT to an appropriately large value prevents chattering of the PLL1 DLD signal and
stable holdover operation can be achieved.
Refer to Digital Lock Detect Frequency Accuracy for more detail on calculating exit times and how the
PLL1_DLD_CNT and PLL1_WND_SIZE work together.
9.2.2.5.2 Holdover
For this example, when the recovered clock is lost, the goal is to set the VCXO to Vcc/2 until the recovered clock
returns. Holdover Mode contains detailed information on how to program holdover.
To achieve the above goal, fixed holdover will be used. Program:
•
•
•
•
•
HOLDOVER_MODE = 2 (Holdover enabled)
EN_TRACK = 0 (Tracking disabled)
EN_MAN_DAC = 1 (Use manual DAC for holdover voltage value)
MAN_DAC = 512 (Approximately Vcc/2)
DISABLE_DLD1_DET = 0 (Use PLL1 DLD = Low to start holdover)
9.2.2.6 Device Programming
The TICS Pro software is used to program the LMK04208 evaluation board using the LMK04208 profile. It also
allows the exporting of a register map which can be used to program the device to the user’s desired
configuration.
Once a configuration has been achieved using the TICS Pro to meet the requested input/output frequencies with
the desired performance, the TICS Pro software is manually updated with this information to meet the required
application. At this time no automatic import exists.
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Typical Applications (continued)
9.2.3 Application Curve
-130
-135
-140
-145
-150
-155
-160
-165
VCO CLKoutX
VCXO CLKout3,4
VCXO OSCout
VCXO Direct
-170
1k
10k
100k
1M
10M
Frequency Offset (Hz)
D001
Figure 39. LVPECL Phase Noise, 122.88 MHz
Illustration of Different Performance Depending on Signal Path
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9.3 System Examples
9.3.1 System Level Diagram
Figure 40 and Figure 41 show an LMK04208 device with external circuitry for clocking and for power supply to
serve as a guideline for good practices when designing with the LMK04208. Refer to Pin Connection
Recommendations for more details on the pin connections and bypassing recommendations. Also refer to the
evaluation board in LMK0480x Evaluation Board Instructions. PCB design will also play a role in device
performance. As discussed in PLL LO Reference , the LO clocks at 122.88 MHz may be moved to CLKout5 if the
VCXO frequency will support 122.88 MHz output.
Status_CLKin0
240 Ö
Status_CLKin1
0.1 mF
CLKout0
CLKout0*
Status_LD
Status_HOLDOVER
SYNC
LVPECL clock
to ADC at
245.76 MHz
To Host
processor
0.1 mF
240 Ö
240 Ö
LEuWire
CLKuWire
DATAuWire
0.1 mF
CLKout1, 2
CLKout1*,2*
2x LVPECL
clocks to DAC at
983.04 MHz
Recovered
Reference
Clock
0.1 mF
CLKin0
0.1 mF
CLKin0*
240 Ö
50 Ö
0.1 mF
LMK04208
2x LVCMOS
clocks to LO at
122.88 MHz
CLKout3
CLKout3*
0.1 mF
CLKin1
100 Ö
CLKin1*
1x LVDS clock to
SERDES at
122.88 MHz
CLKout4
CLKout4*
TCXO
0.1 mF
0.1 mF
CLKout 3 and 4 active at startup
OSCin*
OSCin
CLKout5
CLKout5*
Unused
0.1 mF
R
term
VCXO
LDObyp1
LDObyp2
OSCout
1x LVDS clock
to FPGA at
OSCout*
122.88 MHz
OSCout on at start-up at OSCin
frequency as LVDS output
10 mF
0.1 mF
LF1_R3
LF1_C3
LF1_C2
LF1_R2
LF1_C1
LF2_C2
LF2_R2
LF2_C1
PLL1 Loop Filter
PLL2 External
Loop Filter
Figure 40. Example Application – System Schematic Except for Power
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System Examples (continued)
Figure 40 shows the primary reference clock input is at CLKin0/0*. A secondary reference clock is driving
CLKin1/1*. Both clocks are depicted as AC coupled drivers. The VCXO attached to the OSCin/OSCin* port is
configured as an AC coupled single-ended driver. Any of the input ports (CLKin0/0*, CLKin1/1*, or
OSCin/OSCin*) may be configured as either differential or single-ended. These options are discussed later in the
data sheet.
See Loop Filter for more information on PLL1 and PLL2 loop filters.
In the system shown in Figure 40, LVPECL clocks are AC coupled via 0.1 µF capacitors and LVDS clocks are
DC coupled. Some clock outputs are depicted as LVPECL with 240-Ω emitter resistors and some clock outputs
as LVDS. The appropriate output termination on each output should be implemented according to the output
format to be programmed by the user. Later sections of this data sheet illustrate alternative methods for AC
coupling, DC coupling, and terminating the clock outputs. PCB design will influence crosstalk performance.
Tightly coupled clock traces will have less crosstalk than loosely coupled clock traces. Also proximity to other
clocks traces will influence crosstalk.
PLL Supply Plane
Vcc1
VCO LDO
Digital
FB
Vcc4
Vcc5
Vcc7
Vcc9
CLKin
10 µF, 1 µF, 0.1 µF
1 µF, 0.1 µF, 10 nF
OSCin/OSCout/
PLL2 Circuitry
PLL2 N Divider
LDO
LP3878-ADJ
Vcc6
PLL1 CP
FB
0.1 µF
0.1 µF
0.1 µF
PLL2 CP
FB
0.1 µF
FB = Ferrite bead
Vcc8
LMK04208
Clock Supply Plane
10 µF, 1 µF, 0.1 µF
Vcc13
CLKout0
FB
FB
Example
Frequency 1
Vcc2
Vcc3
CLKout1
CLKout2
FB
FB
Example
Frequency 2
Do not directly copy schematic for
CLKout Vcc13/2/3/10/11/12. This
is for example frequency plan only.
Vcc10
Vcc11
Vcc12
CLKout3
CLKout4
CLKout5
Recommendation is to group
supplies by same frequency and
share a ferrite bead among outputs
of the same frequency.
Example
Frequency 3
Figure 41. Example Application – Power System Schematic
Figure 41 shows an example decoupling and bypassing scheme for the LMK04208, which could apply to
configuration shown in Figure 40. The ferrite beads and capacitors drawn in dotted lines are optional (see Pin
Connection Recommendations). Two power planes are used in these example designs, one for the clock outputs
and one for PLL circuits. It is possible to reduce the number of decoupling components by tying together clock
output Vcc pins for CLKouts that share the same frequency or otherwise can tolerate potential crosstalk between
outputs with different frequencies. In the two examples, Vcc2 and Vcc3 can be tied together since CLKout1 and
CLKout2 will operate at the same frequencies. Vcc10, Vcc11, and Vcc12 can be tied together since potential
crosstalk between the FPGA/SerDes clocks and low-frequency synchronization clocks will not impact the
performance of these digital interfaces, which typically have less stringent jitter requirements. PCB design will
influence impedance to the supply. Vias and traces will increase the impedance to the power supply. Ensure
good direct return current paths.
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9.4 Do's and Don'ts
9.4.1 LVCMOS Complementary vs. Non-Complementary Operation
•
•
•
TI recommends using a complementary LVCMOS output format such as LVCMOS (Norm/Inv) to reduce
switching noise and crosstalk when using LVCMOS.
If only a single LVCMOS output is required, the complementary LVCMOS output format can still be used by
leaving the unused LVCMOS output floating.
A non-complimentary format such as LVCMOS (Norm/Norm) is not recommended as increased switching
noise is present.
9.4.2 LVPECL Outputs
When using an LVPECL output it is not recommended to place a capacitor to ground on the output as might be
done when using a capacitor input LC lowpass filter. The capacitor will appear as a short to the LVPECL output
drivers which are able to supply large amounts of switching current. The effect of the LVPECL sourcing large
switching currents can result in the following:
1. Large switching currents through the Vcc pin of the LVPECL power supply resulting in more Vcc noise and
possible Vcc spikes.
2. Large switching currents injected into the ground plane through the capacitor which could couple onto other
Vcc pins with bypass capacitors to ground resulting in more Vcc noise and possible Vcc spikes.
9.4.3 Sharing MICROWIRE (SPI) Lines
When CLKuWire and DATAuWire toggle and an internal VCO mode is used, there may some spurious content
on the phase noise plot related to the frequency of the CLKuWire and DATAuWire pins.
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10 Power Supply Recommendations
10.1 Pin Connection Recommendations
10.1.1 Vcc Pins and Decoupling
All Vcc pins must always be connected.
Integrated capacitance on the LMK04208 makes external high frequency decoupling capacitors (≤ 1 nF)
unnecessary. The internal capacitance is more effective at filtering high frequency noise than off device bypass
capacitance because there is no bond wire inductance between the LMK04208 circuit and the bypass capacitor.
10.1.1.1 Vcc2, Vcc3, Vcc10, Vcc11, Vcc12, Vcc13 (CLKout Vccs)
Each of these pins has an internal 200 pF of capacitance.
Ferrite beads may be used to reduce crosstalk between different clock output frequencies on the same
LMK04208 device. Ferrite beads placed between the power supply and a clock Vcc pin will reduce noise
between the Vcc pin and the power supply. When several output clocks share the same frequency a single ferrite
bead can be used between the power supply and each same frequency CLKout Vcc pin.
When using ferrite beads on CLKout Vcc pins, consider the following guidelines to ensure the power supply will
source the needed switching current:
•
•
In most cases a ferrite bead may be placed and the internal capacitance is sufficient.
If a ferrite bead is used with a low frequency output (typically ≤ 30 MHz) and a high current switching clock
output format such as non-complementary LVCMOS or high swing LVPECL is used, then:
–
The ferrite bead can be removed to the lower impedance to the main power supply and bypass capacitors,
or
–
Localized capacitance can be placed between the ferrite bead and Vcc pin to support the switching
current.
–
Note: the decoupling capacitors used between the ferrite bead and a CLKout Vcc pin can permit high
frequency switching noise to couple through the capacitors into the ground plane and onto other
CLKout Vcc pins with decoupling capacitors. This can degrade crosstalk performance.
–
TI recommends using a complementary LVCMOS output format such as LVCMOS (Norm/Inv) to reduce
switching noise and crosstalk when using LVCMOS. If only a single LVCMOS output is required, the
complementary LVCMOS output format can still be used by leaving the unused LVCMOS output floating.
10.1.1.2 Vcc1 (VCO), Vcc4 (Digital), and Vcc9 (PLL2)
Each of these pins has internal bypass capacitance.
Ferrite beads should not be used between these pins and the power supply/large bypass capacitors because
these Vcc pins don’t produce much noise and a ferrite bead can cause phase noise disturbances and
resonances.
The typical application diagram in Figure 41 shows all these Vccs connected to together to Vcc without a ferrite
bead.
10.1.1.3 Vcc6 (PLL1 Charge Pump) and Vcc8 (PLL2 Charge Pump)
Each of these pins has an internal bypass capacitor.
Use of a ferrite bead between the power supply/large bypass capacitors and PLL1 is optional. PLL1 charge
pump can be connected directly to Vcc along with Vcc1, Vcc4, and Vcc9. Depending on the application, a 0.1 µF
capacitor may be placed close to PLL1 charge pump Vcc pin.
A ferrite bead should be placed between the power supply/large bypass capacitors and Vcc8. Most applications
have high PLL2 phase detector frequencies and (> 50 MHz) such that the internal bypassing is sufficient and a
ferrite bead can be used to isolate this switching noise from other circuits. For lower phase detector frequencies
a ferrite bead is optional and depending on application a 0.1 µF capacitor may be added on Vcc8.
124
Copyright © 2016, Texas Instruments Incorporated
LMK04208
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ZHCSFH1 –SEPTEMBER 2016
Pin Connection Recommendations (continued)
10.1.1.4 Vcc5 (CLKin), Vcc7 (OSCin and OSCout)
Each of these pins has an internal 100 pF of capacitance. No ferrite bead should be placed between the power
supply/large bypass capacitors and Vcc5 or Vcc7.
These pins are unique since they supply an output clock and other circuitry.
Vcc5 supplies CLKin.
Vcc7 supplies OSCin, OSCout, and PLL2 circuitry.
Impacts of excessive noise on PLL2 circuitry may impact PLL2 DLD operation.
TI recommends using a complementary LVCMOS output format such as LVCMOS (Norm/Inv) to reduce
switching noise and crosstalk when using LVCMOS. If only a single LVCMOS output is required, the
complementary LVCMOS output format can still be used by leaving the unused LVCMOS output floating.
10.1.2 LVPECL Outputs
When using an LVPECL output it is not recommended to place a capacitor to ground on the output as might be
done when using a capacitor input LC lowpass filter. The capacitor will appear as a short to the LVPECL output
drivers which are able to supply large amounts of switching current. The effect of the LVPECL sourcing large
switching currents can result in:
1. Large switching currents through the Vcc pin of the LVPECL power supply resulting in more Vcc noise and
possible Vcc spikes.
2. Large switching currents injected into the ground plane through the capacitor which could couple onto other
Vcc pins with bypass capacitors to ground resulting in more Vcc noise and possible Vcc spikes.
10.1.3 Unused Clock Outputs
Leave unused clock outputs floating and powered down.
10.1.4 Unused Clock Inputs
Unused clock inputs can be left floating.
10.1.5 LDO Bypass
The LDObyp1 and LDObyp2 pins should be connected to GND through external capacitors, as shown in
Figure 41.
Copyright © 2016, Texas Instruments Incorporated
125
LMK04208
ZHCSFH1 –SEPTEMBER 2016
www.ti.com.cn
10.2 Current Consumption and Power Dissipation Calculations
From Table 118 the current consumption can be calculated for any configuration.
For example, the current for the entire device with 1 LVDS (CLKout0) and 1 LVPECL 1.6 Vpp w/ 240-Ω emitter
resistors (CLKout1) output active with a clock output divide = 1, and no other features enabled can be calculated
by adding up the following blocks: core current, clock buffer, one LVDS output buffer current, and one LVPECL
output buffer current. There will also be one LVPECL output drawing emitter current, which means some of the
power from the current draw of the device is dissipated in the external emitter resistors which doesn't add to the
power dissipation budget for the device but is important for LDO ICC calculations.
For total current consumption of the device, add up the significant functional blocks. In this example, 228.1 mA
equals the sum of the following:
•
•
•
•
•
•
140 mA (core current)
17.3 mA (base clock distribution)
25.5 mA (CLKout0 divider)
25.5 mA (CLKout1 divider)
14.3 mA (LVDS buffer)
31 mA (LVPECL 1.6 Vpp buffer w/ 240-Ω emitter resistors)
Once total current consumption has been calculated, power dissipated by the device can be calculated. The
power dissipation of the device is equation to the total current entering the device multiplied by the voltage at the
device minus the power dissipated in any emitter resistors connected to any of the LVPECL outputs. If no emitter
resistors are connected to the LVPECL outputs, this power will be 0 watts. Continuing the above example which
has 253.6 mA total Icc and one output with 240-Ω emitter resistors. Total IC power = 801.88 mW = 3.3 V * 253.6
mA - 35 mW.
Table 118. Typical Current Consumption for Selected Functional Blocks
(TA = 25 °C, VCC = 3.3 V)
POWER
POWER
TYPICAL
ICC
DISSIPATED
EXTERNALL
Y(1)(2)(3)
DISSIPATED
in DEVICE
(mW)
BLOCK
CONDITION
(mA)
(mW)
CORE and FUNCTIONAL BLOCKS
MODE = 0: Dual Loop, Internal
VCO
PLL1 and PLL2 locked
140
155
127
142
116
103
462
512
419
469
383
340
-
-
-
-
-
-
MODE = 2: Dual Loop, Internal
VCO, 0-Delay
PLL1 and PLL2 locked; Includes
EN_FEEDBACK_MUX = 1
MODE = 3: Dual Loop, External
VCO
PLL1 and PLL2 locked
MODE = 5: Dual PLL, 0-DELAY,
External VCO
PLL1 and PLL2 locked; Includes
EN_FEEDBACK_MUX = 1
Core
MODE = 6: Single Loop (PLL2),
Internal VCO
PLL2 locked
PLL2 locked
MODE = 11: Single Loop (PLL2),
External VCO
PD_OSCin = 0
PD_OSCin = 1
42
34.5
2
139
114
6.6
-
-
-
MODE = 16: Clock Distribution
EN_TRACK
Tracking is enabled (EN_TRACK = 1)
At least 1 CLKoutX_PD = 0
Base Clock
Distribution
17.3
2.8
57.1
9.2
-
-
CLKout Group
Each CLKout group (CLKout0 and 5, CLKout1 and 2, CLKout 3 and 4)
(1) Power is dissipated externally in LVPECL emitter resistors. The externally dissipated power is calculated as twice the DC voltage level
of one LVPECL clock output pin squared over the emitter resistance. That is to say power dissipated in emitter resistors = 2 * Vem2
Rem.
/
(2) Assuming R θJA = 15 °C/W, the total power dissipated on chip must be less than (125 °C – 85 °C) / 16 °C/W = 2.5 W to ensure a
junction temperature is less than 125 °C.
(3) Worst case power dissipation can be estimated by multiplying typical power dissipation with a factor of 1.15.
126
Copyright © 2016, Texas Instruments Incorporated
LMK04208
www.ti.com.cn
ZHCSFH1 –SEPTEMBER 2016
Current Consumption and Power Dissipation Calculations (continued)
Table 118. Typical Current Consumption for Selected Functional Blocks
(TA = 25 °C, VCC = 3.3 V) (continued)
POWER
DISSIPATED
EXTERNALL
Y(1)(2)(3)
POWER
TYPICAL
ICC
(mA)
DISSIPATED
in DEVICE
(mW)
BLOCK
CONDITION
(mW)
When a clock output is enabled, this contributes the divider/delay block
Divider / digital delay in extended mode
VCO Divider current
25.5
29.6
7.7
84.1
97.7
25.4
7.2
-
-
-
-
Clock Divider/
Digital Delay
VCO Divider
HOLDOVER mode
When in holdover mode
2.2
Feedback mux must be enabled for 0-delay modes and digital delay
mode (SYNC_QUAL = 1)
Feedback Mux
SYNC Asserted
EN_SYNC = 1
SYNC_QUAL = 1
4.9
1.7
6
16.1
5.6
-
-
-
While SYNC is asserted, this extra current is drawn
Required for SYNC functionality. May be turned off once SYNC is
complete to save power.
19.8
Delay enabled, delay > 7 (CLKout_MUX = 2, 3)
XTAL_LVL = 0
8.7
1.8
2.7
3.6
4.5
2.8
3.4
3.8
4.2
4.7
5.2
28.7
5.9
-
-
-
-
-
-
-
-
-
-
-
XTAL_LVL = 1
Enabling the Crystal Oscillator
XTAL_LVL = 2
9
Crystal Mode
12
XTAL_LVL = 3
EN_PLL2_REF_2X = 1
15
OSCin Doubler
9.2
CLKoutX_ANLG_DLY = 0 to 3
11.2
12.5
13.9
15.5
17.2
CLKoutX_ANLG_DLY = 4 to 7
Analog Delay
Analog Delay Value
CLKoutX_ANLG_DLY = 8 to 11
CLKoutX_ANLG_DLY = 12 to 15
CLKoutX_ANLG_DLY = 16 to 23
CLOCK OUTPUT BUFFERS
LVDS
100-Ω differential termination
LVPECL 2.0 Vpp, AC coupled using 240-Ω emitter resistors
14.3
32
47.2
70.6
67.3
91.8
59
-
35
35
60
40
40
-
LVPECL 1.6 Vpp, AC coupled using 240-Ω emitter resistors
LVPECL 1.6 Vpp, AC coupled using 120-Ω emitter resistors
LVPECL 1.2 Vpp, AC coupled using 240-Ω emitter resistors
LVPECL 0.7 Vpp, AC coupled using 240-Ω emitter resistors
31
LVPECL
46
30
29
55.7
79.2
87.5
120.5
49.5
52.8
71
LVCMOS Pair (CLKoutX_TYPE
= 6 to 9)
CL = 5 pF
3 MHz
24
30 MHz
150 MHz
26.5
36.5
15
-
-
LVCMOS
LVCMOS Single (CLKoutX_TYPE 3 MHz
-
= 10 to 13)
CL = 5 pF
30 MHz
16
-
150 MHz
21.5
-
Copyright © 2016, Texas Instruments Incorporated
127
LMK04208
ZHCSFH1 –SEPTEMBER 2016
www.ti.com.cn
11 Layout
11.1 Layout Guidelines
Power consumption of the LMK04208 device can be high enough to require attention to thermal management.
For reliability and performance reasons the die temperature should be limited to a maximum of 125 °C. That is,
as an estimate, TA (ambient temperature) plus device power consumption times RθJA should not exceed 125 °C.
The package of the device has an exposed pad that provides the primary heat removal path as well as excellent
electrical grounding to a printed circuit board. To maximize the removal of heat from the package a thermal land
pattern including multiple vias to a ground plane must be incorporated on the PCB within the footprint of the
package. The exposed pad must be soldered down to ensure adequate heat conduction out of the package.
A recommended land and via pattern is shown at the end of the datasheet, 机械、封装和可订购信息. More
information on soldering WQFN packages can be obtained from www.ti.com/packaging/.
To minimize junction temperature, TI recommends that a simple heat sink be built into the PCB (if the ground
plane layer is not exposed). This is done by including a copper area on the opposite side of the PCB from the
device. This copper area may be plated or solder coated to prevent corrosion, but should not have conformal
coating (if possible), which could provide thermal insulation.
128
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LMK04208
www.ti.com.cn
ZHCSFH1 –SEPTEMBER 2016
11.2 Layout Example
CLKin and OSCin paths œ if differential input (preferred) route trace
Charge pump output œ shorter traces are better
tightly coupled like clock outputs. If single ended, have at least 3 trace
width (of CLKin/OSCin trace) separation from other RF traces.
Example shown is hybrid for both differential and single ended œ not
tightly couple to compromise for both configurations. RF Terminations
should be placed as close to IC as possible. When using CLKin1 for
high frequency input for external VCO or distribution, a 3 dB pi pad is
suggested for termination.
Place all resistors and caps closer to IC except
a single capacitor next to VCXO. In a 2nd order
filter place C1 close to VCXO Vtune pin. In a 3r
and 4th order filter place C3 or C4 respectively
close to VCXO.
Clock outputs œ differential signals, should be
routed tightly coupled to minimize PCB crosstal
Trace impedance and terminations should be
designed according to output type being used (
LVDS, LVPECL...)
For CLKout Vcc‘s place ferrite beads on top layer close to pins to choke
high frequency noise from via.
Figure 42. LMK04208 Layout Example
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129
LMK04208
ZHCSFH1 –SEPTEMBER 2016
www.ti.com.cn
12 器件和文档支持
12.1 器件支持
12.1.1 开发支持
如需支持，请访问以下网站：
•
•
时钟架构（仿真）：http://www.ti.com/lsds/ti/analog/webench/clock-architect.page
TICS Pro（EVM 编程和寄存器生成工具）：
12.2 文档支持
12.2.1 相关文档
更多信息，请参见以下文档：
•
•
•
《AN-912 常用数据传输参数及其定义》
《AN-1939 采用 LMK04000 系列器件的晶体振荡器设计》
《AN-1865 PLL 频率合成和架构规划》
12.3 接收文档更新通知
如需接收文档更新通知，请访问 www.ti.com.cn 网站上的器件产品文件夹。点击右上角的提醒我 (Alert me) 注册
后，即可每周定期收到已更改的产品信息。有关更改的详细信息，请查阅已修订文档中包含的修订历史记录。
12.4 社区资源
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
12.5 商标
PLLatinum, E2E are trademarks of Texas Instruments.
All other trademarks are the property of their respective owners.
12.6 静电放电警告
这些装置包含有限的内置 ESD 保护。 存储或装卸时，应将导线一起截短或将装置放置于导电泡棉中，以防止 MOS 门极遭受静电损
伤。
12.7 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
13 机械、封装和可订购信息
以下页中包括机械、封装和可订购信息。这些信息是针对指定器件可提供的最新数据。这些数据会在无通知且不对
本文档进行修订的情况下发生改变。欲获得该数据表的浏览器版本，请查阅左侧的导航栏。
130
版权 © 2016, Texas Instruments Incorporated
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status Package Type Package Pins Package
Eco Plan
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
Samples
Drawing
Qty
(1)
(2)
(3)
(4/5)
(6)
LMK04208NKDR
LMK04208NKDT
ACTIVE
ACTIVE
WQFN
WQFN
NKD
NKD
64
64
2000 RoHS & Green
250 RoHS & Green
SN
Level-3-260C-168 HR
Level-3-260C-168 HR
-40 to 85
-40 to 85
K04208NKD
K04208NKD
SN
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based
flame retardants must also meet the <=1000ppm threshold requirement.
(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6)
Lead finish/Ball material - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead finish/Ball material values may wrap to two
lines if the finish value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 1
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
Addendum-Page 2
GENERIC PACKAGE VIEW
NKD 64
9 x 9, 0.5 mm pitch
WQFN - 0.8 mm max height
PLASTIC QUAD FLATPACK - NO LEAD
This image is a representation of the package family, actual package may vary.
Refer to the product data sheet for package details.
4229637/A
www.ti.com
PACKAGE OUTLINE
NKD0064A
WQFN - 0.8 mm max height
S
C
A
L
E
1
.
6
0
0
WQFN
9.1
8.9
A
B
PIN 1 INDEX AREA
0.5
0.3
9.1
8.9
0.3
0.2
DETAIL
OPTIONAL TERMINAL
TYPICAL
0.8 MAX
C
SEATING PLANE
(0.1)
TYP
7.2 0.1
SEE TERMINAL
DETAIL
17
32
60X 0.5
33
16
4X
7.5
1
48
0.3
64X
PIN 1 ID
64
49
0.2
(OPTIONAL)
0.1
C A
C
B
0.5
0.3
64X
0.05
4214996/A 08/2013
NOTES:
1. All linear dimensions are in millimeters. Dimensions in parenthesis are for reference only. Dimensioning and tolerancing
per ASME Y14.5M.
2. This drawing is subject to change without notice.
3. The package thermal pad must be soldered to the printed circuit board for thermal and mechanical performance.
www.ti.com
EXAMPLE BOARD LAYOUT
NKD0064A
WQFN - 0.8 mm max height
WQFN
(
7.2)
SYMM
64X (0.6)
64X (0.25)
SEE DETAILS
49
64
1
48
60X (0.5)
SYMM
(8.8)
(1.36)
TYP
8X (1.31)
33
(
0.2) VIA
TYP
16
17
32
(1.36) TYP
8X (1.31)
(8.8)
LAND PATTERN EXAMPLE
SCALE:8X
0.07 MAX
ALL AROUND
0.07 MIN
ALL AROUND
METAL
SOLDER MASK
OPENING
SOLDER MASK
OPENING
METAL
NON SOLDER MASK
DEFINED
SOLDER MASK
DEFINED
(PREFERRED)
SOLDER MASK DETAILS
4214996/A 08/2013
NOTES: (continued)
4. This package is designed to be soldered to a thermal pad on the board. For more information, refer to QFN/SON PCB application note
in literature No. SLUA271 (www.ti.com/lit/slua271).
www.ti.com
EXAMPLE STENCIL DESIGN
NKD0064A
WQFN - 0.8 mm max height
WQFN
SYMM
(1.36) TYP
49
64X (0.6)
64X (0.25)
64
1
48
(1.36)
TYP
60X (0.5)
SYMM
(8.8)
METAL
TYP
16
33
17
32
25X (1.16)
(8.8)
SOLDERPASTE EXAMPLE
BASED ON 0.125mm THICK STENCIL
EXPOSED PAD
65% PRINTED SOLDER COVERAGE BY AREA
SCALE:10X
4214996/A 08/2013
NOTES: (continued)
5. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate
design recommendations.
www.ti.com
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