LMR64010XMFE/NOPB [TI]

LMR64010 SIMPLE SWITCHERÂ® 40Vout, 1A Step-Up Voltage Regulator in SOT-23;


型号：	LMR64010XMFE/NOPB
厂家：	TEXAS INSTRUMENTS
描述：	LMR64010 SIMPLE SWITCHERÂ® 40Vout, 1A Step-Up Voltage Regulator in SOT-23 开关光电二极管
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LMR64010

LMR64010 SIMPLE SWITCHER ® 40Vout, 1A Step-Up Voltage Regulator in SOT-23

Literature Number: SNVS736A

November 16, 2011

LMR64010

SIMPLE SWITCHER^®40Vout, 1A Step-Up Voltage

Regulator in SOT-23

Features

Performance Benefits

Input voltage range of 2.7V to 14V

Extremely easy to use

■

Output voltage up to 40V

Tiny overall solution reduces system cost

Switch current up to 1A

Applications

1.6 MHz switching frequency

Low shutdown Iq, <1 µA

Boost Conversions from 3.3V, 5V, and 12V Rails

■

Cycle-by-cycle current limiting

Space Constrained Applications

Internally compensated

Embedded Systems

SOT23-5 packaging (2.92 x 2.84 x 1.08mm)

LCD Displays

Fully enabled for WEBENCH® Power Designer

LED Applications

■

System Performance

30167557

30167558

30167501

301675

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Connection Diagram

Top View

30167502

5-Lead SOT-23 Package

See NS Package Number MF05A

Ordering Information

Order Number

LMR64010XMFE

LMR64010XMF

LMR64010XMFX

Package Type

Package Drawing

Supplied As

Package ID

250 Units, Tape and Reel

1000 Units, Tape and Reel

3000 Units, Tape and Reel

SF9B

SOT23-5

MF05A

Pin Descriptions

Name

Function

Drain of the internal FET switch.

Analog and power ground.

GND

Feedback point that connects to external resistive divider.

Shutdown control input. Connect to V_INif this feature is not used.

Analog and power input.

SHDN

V_IN

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SW Pin Voltage

Input Supply Voltage

SHDN Pin Voltage

−0.4V to +40V

−0.4V to +14.5V

−0.4V to VIN + 0.3V

Absolute Maximum Ratings (Note 1)

If Military/Aerospace specified devices are required,

please contact the Texas Instruments Sales Office/

Distributors for availability and specifications.

_J-A(SOT23-5)

265°C/W

ESD Rating (Note 3)

Human Body Model

Machine Model

For soldering specifications: see product folder at

www.national.com and www.national.com/ms/MS/MS-

SOLDERING.pdf

Storage Temperature Range

Operating Junction

Temperature Range

Lead Temp. (Soldering, 5 sec.)

Power Dissipation (Note 2)

FB Pin Voltage

−65°C to +150°C

2 kV

200V

−40°C to +125°C

300°C

Internally Limited

−0.4V to +6V

Electrical Characteristics

Limits in standard typeface are for T_J= 25°C, and limits in boldface type apply over the full operating temperature range

(−40°C ≤ T_J≤ +125°C). Unless otherwise specified: V_IN= 5V, V_SHDN= 5V, I_L= 0A.

Min

(Note 4)

Typical

(Note 5)

Max

(Note 4)

Symbol

V_IN

Parameter

Input Voltage

Conditions

Units

2.7

I_SW

Switch Current Limit

Switch ON Resistance

Shutdown Threshold

(Note 6)

1.0

1.5

R_DS(ON)

SHDN_TH

I_SW= 100 mA

Device ON

Device OFF

V_SHDN= 0

500

650

0.50

mΩ

1.5

µA

I_SHDN

Shutdown Pin Bias Current

V_SHDN= 5V

V_IN= 3V

V_FB

Feedback Pin Reference

Voltage

1.205

1.230

1.255

I_FB

I_Q

Feedback Pin Bias Current

Quiescent Current

V_FB= 1.23V

2.1

V_SHDN= 5V, Switching

V_SHDN= 5V, Not Switching

V_SHDN= 0

3.0

500

400

µA

0.024

FB Voltage Line Regulation

2.7V ≤ V_IN≤ 14V

0.02

%/V

F_SW

D_MAX

I_L

Switching Frequency

Maximum Duty Cycle

Switch Leakage

1.15

1.6

1.85

MHz

Not Switching V_SW= 5V

µA

Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the component may occur. Electrical specifications do not apply when operating the

device outside of the limits set forth under the operating ratings which specify the intended range of operating conditions.

Note 2: The maximum power dissipation which can be safely dissipated for any application is a function of the maximum junction temperature, T_J(MAX) = 125°

C, the junction-to-ambient thermal resistance for the SOT-23 package, θ_J-A= 265°C/W, and the ambient temperature, T_A. The maximum allowable power

dissipation at any ambient temperature for designs using this device can be calculated using the formula:

If power dissipation exceeds the maximum specified above, the internal thermal protection circuitry will protect the device by reducing the output voltage as

required to maintain a safe junction temperature.

Note 3: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin. The machine model is a 200 pF capacitor discharged

directly into each pin.

Note 4: Limits are guaranteed by testing, statistical correlation, or design.

Note 5: Typical values are derived from the mean value of a large quantity of samples tested during characterization and represent the most likely expected value

of the parameter at room temperature.

Note 6: Switch current limit is dependent on duty cycle (see Typical Performance Characteristics). Limits shown are for duty cycles ≤ 50%.

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Typical Performance Characteristics Unless otherwise specified: V_IN= 5V, SHDN pin is tied to V_IN.

Iq V_IN(Active) vs Temperature

Oscillator Frequency vs Temperature

30167508

30167510

Max. Duty Cycle vs Temperature

Feedback Voltage vs Temperature

30167506

30167555

R_DS(ON) vs Temperature

Current Limit vs Temperature

30167509

30167507

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R_DS(ON)vs V_IN

Efficiency vs Load Current (V_OUT= 12V)

30167514

30167523

Efficiency vs Load Current (V_OUT= 15V)

Efficiency vs Load Current (V_OUT= 20V)

30167545

30167546

Efficiency vs Load Current (V_OUT= 25V)

Efficiency vs Load Current (V_OUT= 30V)

30167547

30167548

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Efficiency vs Load Current (V_OUT= 35V)

Efficiency vs Load Current (V_OUT= 40V)

30167550

30167549

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Block Diagram

30167503

currents flowing through Q1 and Q2 will be equal, and the

General Description

feedback loop will adjust the regulated output to maintain this.

Because of this, the regulated output is always maintained at

a voltage level equal to the voltage at the FB node "multiplied

up" by the ratio of the output resistive divider.

The LMR64010 switching regulators is a current-mode boost

converter operating at a fixed frequency of 1.6 MHz.

The use of SOT-23 package, made possible by the minimal

power loss of the internal 1A switch, and use of small induc-

tors and capacitors result in the industry's highest power

density. The 40V internal switch makes these solutions per-

fect for boosting to voltages of 16V or greater.

The current limit comparator feeds directly into the flip-flop,

that drives the switch FET. If the FET current reaches the limit

threshold, the FET is turned off and the cycle terminated until

the next clock pulse. The current limit input terminates the

pulse regardless of the status of the output of the PWM com-

parator.

These parts have a logic-level shutdown pin that can be used

to reduce quiescent current and extend battery life.

Protection is provided through cycle-by-cycle current limiting

and thermal shutdown. Internal compensation simplifies de-

sign and reduces component count.

Application Hints

SELECTING THE EXTERNAL CAPACITORS

The best capacitors for use with the LMR64010 are multi-lay-

er ceramic capacitors. They have the lowest ESR (equivalent

series resistance) and highest resonance frequency which

makes them optimum for use with high frequency switching

converters.

Theory of Operation

The LMR64010 is a switching converter IC that operates at a

fixed frequency (1.6 MHz) using current-mode control for fast

transient response over a wide input voltage range and in-

corporates pulse-by-pulse current limiting protection. Be-

cause this is current mode control, a 50 mΩ sense resistor in

series with the switch FET is used to provide a voltage (which

is proportional to the FET current) to both the input of the

pulse width modulation (PWM) comparator and the current

limit amplifier.

When selecting a ceramic capacitor, only X5R and X7R di-

electric types should be used. Other types such as Z5U and

Y5F have such severe loss of capacitance due to effects of

temperature variation and applied voltage, they may provide

as little as 20% of rated capacitance in many typical applica-

tions. Always consult capacitor manufacturer’s data curves

before selecting a capacitor.

At the beginning of each cycle, the S-R latch turns on the FET.

As the current through the FET increases, a voltage (propor-

tional to this current) is summed with the ramp coming from

the ramp generator and then fed into the input of the PWM

comparator. When this voltage exceeds the voltage on the

other input (coming from the Gm amplifier), the latch resets

and turns the FET off. Since the signal coming from the Gm

amplifier is derived from the feedback (which samples the

voltage at the output), the action of the PWM comparator

constantly sets the correct peak current through the FET to

keep the output volatge in regulation.

SELECTING THE OUTPUT CAPACITOR

A single ceramic capacitor of value 4.7 µF to 10 µF will provide

sufficient output capacitance for most applications. For output

voltages below 10V, a 10 µF capacitance is required. If larger

amounts of capacitance are desired for improved line support

and transient response, tantalum capacitors can be used in

parallel with the ceramics. Aluminum electrolytics with ultra

low ESR such as Sanyo Oscon can be used, but are usually

prohibitively expensive. Typical AI electrolytic capacitors are

not suitable for switching frequencies above 500 kHz due to

significant ringing and temperature rise due to self-heating

Q1 and Q2 along with R3 - R6 form a bandgap voltage refer-

ence used by the IC to hold the output in regulation. The

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from ripple current. An output capacitor with excessive ESR

can also reduce phase margin and cause instability.

SELECTING THE INPUT CAPACITOR

An input capacitor is required to serve as an energy reservoir

for the current which must flow into the coil each time the

switch turns ON. This capacitor must have extremely low

ESR, so ceramic is the best choice. We recommend a nomi-

nal value of 2.2 µF, but larger values can be used. Since this

capacitor reduces the amount of voltage ripple seen at the

input pin, it also reduces the amount of EMI passed back

along that line to other circuitry.

FEED-FORWARD COMPENSATION

Although internally compensated, the feed-forward capacitor

Cf is required for stability (see Basic Application Circuit).

Adding this capacitor puts a zero in the loop response of the

converter. Without it, the regulator loop can oscillate. The

recommended frequency for the zero fz should be approxi-

mately 8 kHz. Cf can be calculated using the formula:

30167522

Cf = 1 / (2 X π X R1 X fz)

Recommended PCB Component Layout

SELECTING DIODES

Some additional guidelines to be observed:

The external diode used in the typical application should be

a Schottky diode. If the switch voltage is less than 15V, a 20V

diode such as the MBR0520 is recommended. If the switch

voltage is between 15V and 25V, a 30V diode such as the

MBR0530 is recommended. If the switch voltage exceeds

25V, a 40V diode such as the MBR0540 should be used.

1. Keep the path between L1, D1, and C2 extremely short.

Parasitic trace inductance in series with D1 and C2 will

increase noise and ringing.

2. The feedback components R1, R2 and CF must be kept

close to the FB pin of U1 to prevent noise injection on the

FB pin trace.

The MBR05XX series of diodes are designed to handle a

maximum average current of 0.5A. For applications exceed-

ing 0.5A average but less than 1A, a Toshiba CRS08 can be

used.

3. If internal ground planes are available (recommended)

use vias to connect directly to ground at pin 2 of U1, as

well as the negative sides of capacitors C1 and C2.

LAYOUT HINTS

SETTING THE OUTPUT VOLTAGE

High frequency switching regulators require very careful lay-

out of components in order to get stable operation and low

noise. All components must be as close as possible to the

LMR64010 device. It is recommended that a 4-layer PCB be

used so that internal ground planes are available.

The output voltage is set using the external resistors R1 and

R2 (see Basic Application Circuit). A value of approximately

13.3 kΩ is recommended for R2 to establish a divider current

of approximately 92 µA. R1 is calculated using the formula:

R1 = R2 X (V_OUT/1.23 − 1)

As an example, a recommended layout of components is

shown:

30167505

Basic Application Circuit

DUTY CYCLE

This applies for continuous mode operation.

The maximum duty cycle of the switching regulator deter-

mines the maximum boost ratio of output-to-input voltage that

the converter can attain in continuous mode of operation. The

duty cycle for a given boost application is defined as:

The equation shown for calculating duty cycle incorporates

terms for the FET switch voltage and diode forward voltage.

The actual duty cycle measured in operation will also be af-

fected slightly by other power losses in the circuit such as wire

losses in the inductor, switching losses, and capacitor ripple

current losses from self-heating. Therefore, the actual (effec-

tive) duty cycle measured may be slightly higher than calcu-

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lated to compensate for these power losses. A good

approximation for effctive duty cycle is :

inductor current will begin touching the zero axis which means

it will be in discontinuous mode. A similar analysis can be

performed on any boost converter, to make sure the ripple

current is reasonable and continuous operation will be main-

tained at the typical load current values.

DC (eff) = (1 - Efficiency x (V_IN/V_OUT))

Where the efficiency can be approximated from the curves

provided.

INDUCTANCE VALUE

The first question we are usually asked is: “How small can I

make the inductor?” (because they are the largest sized com-

ponent and usually the most costly). The answer is not simple

and involves tradeoffs in performance. Larger inductors mean

less inductor ripple current, which typically means less output

voltage ripple (for a given size of output capacitor). Larger

inductors also mean more load power can be delivered be-

cause the energy stored during each switching cycle is:

E =L/2 X (lp)²

30167524

Typical Application, 5V–12V Boost

Where “lp” is the peak inductor current. An important point to

observe is that the LMR64010 will limit its switch current

based on peak current. This means that since lp(max) is fixed,

increasing L will increase the maximum amount of power

available to the load. Conversely, using too little inductance

may limit the amount of load current which can be drawn from

the output.

MAXIMUM SWITCH CURRENT

The maximum FET swtch current available before the current

limiter cuts in is dependent on duty cycle of the application.

This is illustrated in the graphs below which show both the

typical and guaranteed values of switch current as a function

of effective (actual) duty cycle:

Best performance is usually obtained when the converter is

operated in “continuous” mode at the load current range of

interest, typically giving better load regulation and less output

ripple. Continuous operation is defined as not allowing the in-

ductor current to drop to zero during the cycle. It should be

noted that all boost converters shift over to discontinuous op-

eration as the output load is reduced far enough, but a larger

inductor stays “continuous” over a wider load current range.

To better understand these tradeoffs, a typical application cir-

cuit (5V to 12V boost with a 10 µH inductor) will be analyzed.

We will assume:

V_IN= 5V, V_OUT= 12V, V_DIODE= 0.5V, V_SW= 0.5V

Since the frequency is 1.6 MHz (nominal), the period is ap-

proximately 0.625 µs. The duty cycle will be 62.5%, which

means the ON time of the switch is 0.390 µs. It should be

noted that when the switch is ON, the voltage across the in-

ductor is approximately 4.5V.

Using the equation:

V = L (di/dt)

30167525

Switch Current Limit vs Duty Cycle

We can then calculate the di/dt rate of the inductor which is

found to be 0.45 A/µs during the ON time. Using these facts,

we can then show what the inductor current will look like dur-

ing operation:

CALCULATING LOAD CURRENT

As shown in the figure which depicts inductor current, the load

current is related to the average inductor current by the rela-

tion:

I_LOAD= I_IND(AVG) x (1 - DC)

Where "DC" is the duty cycle of the application. The switch

current can be found by:

I_SW= I_IND(AVG) + ½ (I_RIPPLE

)

Inductor ripple current is dependent on inductance, duty cy-

cle, input voltage and frequency:

I_RIPPLE= DC x (V_IN-V_SW) / (f x L)

combining all terms, we can develop an expression which al-

lows the maximum available load current to be calculated:

30167512

10 µH Inductor Current,5V–12V Boost

During the 0.390 µs ON time, the inductor current ramps up

0.176A and ramps down an equal amount during the OFF

time. This is defined as the inductor “ripple current”. It can also

be seen that if the load current drops to about 33 mA, the

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The equation shown to calculate maximum load current takes

into account the losses in the inductor or turn-OFF switching

losses of the FET and diode. For actual load current in typical

applications, we took bench data for various input and output

voltages and displayed the maximum load current available

for a typical device in graph form:

possible inductance value for cost and size savings. The con-

verter will operate in discontinuous mode in such a case.

The minimum inductance should be selected such that the

inductor (switch) current peak on each cycle does not reach

the 1A current limit maximum. To understand how to do this,

an example will be presented.

In the example, minimum switching frequency of 1.15 MHz

will be used. This means the maximum cycle period is the

reciprocal of the minimum frequency:

T_ON(max)= 1/1.15M = 0.870 µs

We will assume the input voltage is 5V, V_OUT= 12V, V_SW

0.2V, V_DIODE= 0.3V. The duty cycle is:

Duty Cycle = 60.3%

Therefore, the maximum switch ON time is 0.524 µs. An in-

ductor should be selected with enough inductance to prevent

the switch current from reaching 1A in the 0.524 µs ON time

interval (see below):

30167534

Max. Load Current vs V_IN

Discontinuous Design, 5V–12V Boost ^30167513

DESIGN PARAMETERS V_SWAND I_SW

The value of the FET "ON" voltage (referred to as V_SWin the

equations) is dependent on load current. A good approxima-

tion can be obtained by multiplying the "ON Resistance" of

the FET times the average inductor current.

The voltage across the inductor during ON time is 4.8V. Min-

imum inductance value is found by:

V = L X dl/dt, L = V X (dt/dl) = 4.8 (0.524µ/1) = 2.5 µH

FET on resistance increases at V_INvalues below 5V, since

the internal N-FET has less gate voltage in this input voltage

range (see Typical performance Characteristics curves).

Above V_IN= 5V, the FET gate voltage is internally clamped to

5V.

In this case, a 2.7 µH inductor could be used assuming it pro-

vided at least that much inductance up to the 1A current value.

This same analysis can be used to find the minimum induc-

tance for any boost application.

When selecting an inductor, make certain that the continuous

current rating is high enough to avoid saturation at peak cur-

rents. A suitable core type must be used to minimize core

(switching) losses, and wire power losses must be considered

when selecting the current rating.

The maximum peak switch current the device can deliver is

dependent on duty cycle. The minimum value is guaranteed

to be > 1A at duty cycle below 50%. For higher duty cycles,

see Typical performance Characteristics curves.

THERMAL CONSIDERATIONS

SHUTDOWN PIN OPERATION

At higher duty cycles, the increased ON time of the FET

means the maximum output current will be determined by

power dissipation within the LMR64010 FET switch. The

switch power dissipation from ON-state conduction is calcu-

lated by:

The device is turned off by pulling the shutdown pin low. If this

function is not going to be used, the pin should be tied directly

to V_IN. If the SHDN function will be needed, a pull-up resistor

must be used to V_IN(approximately 50k-100kΩ recommend-

ed). The SHDN pin must not be left unterminated.

P_(SW)= DC x I_IND(AVE)²x R_DSON

There will be some switching losses as well, so some derating

needs to be applied when calculating IC power dissipation.

MINIMUM INDUCTANCE

In some applications where the maximum load current is rel-

atively small, it may be advantageous to use the smallest

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Physical Dimensions inches (millimeters) unless otherwise noted

5-Lead SOT-23 Package

Order Number LMR64010XMF, LMR64010XMFX

NS Package Number MF05A

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