OPA820IDBVRG4 [TI]
单位增益稳定、低噪声、电压反馈运算放大器 | DBV | 5 | -40 to 85;
| 型号： | OPA820IDBVRG4 |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | 单位增益稳定、低噪声、电压反馈运算放大器 | DBV | 5 | -40 to 85 放大器 光电二极管 运算放大器 |
| 文件： | 总50页 (文件大小：1399K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
Sample &
Buy
Support &
Community
Product
Folder
Tools &
Software
Technical
Documents
OPA820
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
OPA820 Unity-Gain Stable, Low-Noise, Voltage-Feedback Operational Amplifier
1 Features
3 Description
The OPA820 device provides a wideband, unity-gain
stable, voltage-feedback amplifier with a very-low
input-noise voltage and high-output current using a
low 5.6-mA supply current. At unity-gain, the OPA820
device gives more than 800-MHz bandwidth with less
than 1-dB peaking. The OPA820 device complements
this high-speed operation with excellent DC precision
in a low-power device. A worst-case input-offset
voltage of ±750 µV and an offset current of ±400 nA
provide excellent absolute DC precision for pulse
amplifier applications.
1
•
High Bandwidth (240 MHz, G = 2)
High-Output Current (±110 mA)
Low-Input Noise (2.5 nV/√Hz)
Low-Supply Current (5.6 mA)
Flexible Supply Voltage:
•
•
•
•
–
–
Dual ±2.5 V to ±6 V
Single 5 V to 12 V
•
Excellent DC Accuracy:
–
–
Maximum 25°C Input Offset Voltage = ±750 µV
Maximum 25°C Input Offset Current = ±400 nA
Minimal input and output voltage-swing headroom
allow the OPA820 device to operate on a single 5-V
supply with more than 2-VPP output swing. While not
a rail-to-rail (RR) output, this swing supports most
emerging analog-to-digital converter (ADC) input
ranges with lower power and noise than typical RR
output op amps.
2 Applications
•
•
•
•
•
•
•
•
•
Low-Cost Video Line Drivers
ADC Preamplifiers
Active Filters
Exceptionally low dG/dP (0.01% or 0.03°) supports
low-cost composite-video line-driver applications.
Existing designs can use the industry-standard
pinout, 8-pin SOIC package while emerging high-
density portable applications can use the 5-pin SOT-
23. Offering the lowest thermal impedance of the
Low-Noise Integrators
Portable Test Equipment
Optical Channel Amplifiers
Low-Power, Baseband Amplifiers
CCD Imaging Channel Amplifiers
OPA650 and OPA620 Upgrade
industry in
a
SOT package, along with full
specification over both the commercial and industrial
temperature ranges, provides solid performance over
a wide temperature range.
Device Information(1)
PART NUMBER
PACKAGE
SOIC (8)
SOT-23 (5)
BODY SIZE (NOM)
4.90 mm × 3.91 mm
2.90 mm × 1.60 mm
OPA820
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
AC-Coupled, 14-Bit ADS850 Interface
5 V
5 V
REFT
(3 V)
R
2 kꢀ
2 kꢀ
S
0.1 mF
V
IN
24.9 ꢀ
+
OPA820
IN
IN
50 ꢀ
100 pF
ADS850
14-Bit
10 MSPS
œ5 V
2 kꢀ
402 ꢀ
402 ꢀ
0.1 µF
(2 V) (1 V)
REFB VREF
2 kꢀ
SEL
Copyright © 2016, Texas Instruments Incorporated
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
OPA820
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
www.ti.com
Table of Contents
9.3 Device Functional Modes........................................ 24
10 Application and Implementation........................ 28
10.1 Application Information.......................................... 28
10.2 Typical Applications .............................................. 28
11 Power Supply Recommendations ..................... 34
12 Layout................................................................... 35
12.1 Layout Guidelines ................................................. 35
12.2 Layout Example .................................................... 36
13 Device and Documentation Support ................. 37
13.1 Device Support...................................................... 37
13.2 Documentation Support ........................................ 37
13.3 Receiving Notification of Documentation Updates 37
13.4 Community Resources.......................................... 38
13.5 Trademarks........................................................... 38
13.6 Electrostatic Discharge Caution............................ 38
13.7 Glossary................................................................ 38
1
2
3
4
5
6
7
Features.................................................................. 1
Applications ........................................................... 1
Description ............................................................. 1
Revision History..................................................... 2
Device Comparison Table..................................... 3
Pin Configuration and Functions......................... 3
Specifications......................................................... 3
7.1 Absolute Maximum Ratings ...................................... 3
7.2 ESD Ratings.............................................................. 4
7.3 Recommended Operating Conditions....................... 4
7.4 Thermal Information.................................................. 4
7.5 Electrical Characteristics: VS = ±5 V......................... 4
7.6 Electrical Characteristics: VS = 5 V........................... 7
7.7 Typical Characteristics............................................ 10
Parameter Measurement Information ................ 18
Detailed Description ............................................ 20
9.1 Overview ................................................................. 20
9.2 Feature Description................................................. 20
8
9
14 Mechanical, Packaging, and Orderable
Information ........................................................... 38
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision C (August 2008) to Revision D
Page
•
Added Pin Configuration and Functions section, ESD Ratings table, Feature Description section, Device Functional
Modes, Application and Implementation section, Power Supply Recommendations section, Layout section, Device
and Documentation Support section, and Mechanical, Packaging, and Orderable Information section ............................... 1
Deleted Ordering Information table; see Package Option Addendum at the end of the data sheet...................................... 1
Deleted Lead temperature (soldering, 300°C maximum) from Absolute Maximum Ratings table......................................... 3
Added Thermal Information table ........................................................................................................................................... 4
Changed Open-loop voltage gain test condition in Electrical Characteristics: VS = ±5 V table From: VO To: VCM ................ 5
Changed Open-loop voltage gain test condition in Electrical Characteristics: VS = 5 V table From: VO To: VCM .................. 8
•
•
•
•
•
•
Changed R2 From: 505 Ω To: 517 Ω, C1 From: 150 pF To: 100 pF, and C2 From: 100 pF To: 160 pF in 5-MHz
Butterworth Low-Pass Active Filter image............................................................................................................................ 28
Changes from Revision B (March 2006) to Revision C
Page
•
Changed Storage Temperature minimum value from –40°C to –65°C.................................................................................. 3
Changes from Revision A (July 2004) to Revision B
Page
•
Changed the board part number in the Design-In Tools section.......................................................................................... 37
2
Submit Documentation Feedback
Copyright © 2004–2016, Texas Instruments Incorporated
Product Folder Links: OPA820
OPA820
www.ti.com
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
5 Device Comparison Table
Table 1. Related Products
SINGLE CHANNEL
DUAL CHANNEL
OPA2354
OPA2690
OPA2652
OPA2822
—
TRIPLE CHANNEL
QUAD CHANNEL
OPA4354
FEATURES
CMOS RR output
High-slew rate
8-Pin SOT23
Low noise
OPA354
OPA690
—
—
OPA3690
—
—
—
—
—
—
—
—
OPA4820
Quad OPA820
6 Pin Configuration and Functions
D Package
8-Pin SOIC
Top View
DBV Package
5-Pin SOT-23
Top View
NC
Inverting Input
1
2
3
4
8
7
6
5
NC
+V
Output
1
2
3
5
4
+V
S
S
Noninverting Input
Output
Noninverting Input
Inverting Input
œV
S
Noninverting Input
+
NC
Noninverting Input
œV
S
Pin Functions
PIN
I/O
DESCRIPTION
NAME
SOIC
SOT-23
Disable
Inverting Input
NC
8
—
4
I
Disable the op amp (Low = Disable; High = Enable)
Inverting input
2
I
1, 5, 8
—
3
—
I
No connection
Noninverting Input
3
6
7
4
Noninverting input
Output
+VS
1
O
—
—
Output of amplifier
5
Positive power supply
Negative power supply
–VS
2
7 Specifications
7.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted)(1)
MIN
MAX
±6.5
UNIT
Power supply
VDC
Internal power dissipation
Differential input voltage
Input common-mode voltage
Junction temperature, TJ
Storage temperature, Tstg
See Thermal Information
±1.2
±VS
150
V
V
°C
°C
–65
125
(1) Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
Copyright © 2004–2016, Texas Instruments Incorporated
Submit Documentation Feedback
3
Product Folder Links: OPA820
OPA820
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
www.ti.com
7.2 ESD Ratings
VALUE
±3000
±1000
±300
UNIT
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001(1)
Charged-device model (CDM), per JEDEC specification JESD22-C101(2)
V(ESD)
Electrostatic discharge
V
Machine model (MM)
(1) JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
(2) JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
7.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
5
NOM
10
MAX
12
UNIT
V
VS
TA
Total supply voltage
Operating ambient temperature
–45
25
85
°C
7.4 Thermal Information
OPA820
THERMAL METRIC(1)
DBV (SOT-23)
5 PINS
150
D (SOIC)
UNIT
8 PINS
125
RθJA
RθJC(top)
RθJB
ψJT
Junction-to-ambient thermal resistance
Junction-to-case (top) thermal resistance
Junction-to-board thermal resistance
°C/W
°C/W
°C/W
°C/W
°C/W
141.1
42.9
72.6
68.2
28.1
67.8
Junction-to-top characterization parameter
Junction-to-board characterization parameter
23.5
ψJB
42
(1) For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
7.5 Electrical Characteristics: VS = ±5 V
RF = 402 Ω, RL = 100 Ω, G = 2, and TA = 25°C (unless otherwise noted)(1)(2)(3)
PARAMETER
AC PERFORMANCE
TEST CONDITIONS
MIN
TYP MAX
UNIT
MHz
MHz
G = 1, VO = 0.1 VPP, RF = 0 Ω, Test level = C
800
240
TA = 25°C
170
160
155
23
G = 2, VO = 0.1 VPP
,
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 25°C
Test level = B
Small-signal bandwidth
30
G = 10, VO = 0.1 VPP
,
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 25°C
21
Test level = B
20
220
204
200
280
Gain-bandwidth product
G ≥ 20, Test level = B
TA = 0°C to 70°C
TA = –40°C to 85°C
Bandwidth for 0.1-dB gain flatness G = 2, VO = 0.1 VPP, Test level = C
38
0.5
85
MHz
dB
Peaking at a gain of 1
Large-signal bandwidth
VO = 0.1 VPP, RF = 0 Ω, Test level = C
G = 2, VO = 2 VPP, Test level = C
MHz
TA = 25°C
192
186
180
240
G = 2, 2-V step,
Test level = B
Slew rate
TA = 0°C to 70°C
V/µs
ns
TA = –40°C to 85°C
Rise time and fall time
G = 2, VO = 0.2-V step, Test level = C
1.5
(1) Test levels: (A) 100% tested at 25°C. Overtemperature limits by characterization and simulation. (B) Limits set by characterization and
simulation. (C) Typical value only for information.
(2) TJ = TA for 25°C specifications.
(3) TJ = TA at low temperature limits; TJ = TA + 9°C at high temperature limit for over temperature.
4
Submit Documentation Feedback
Copyright © 2004–2016, Texas Instruments Incorporated
Product Folder Links: OPA820
OPA820
www.ti.com
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
Electrical Characteristics: VS = ±5 V (continued)
RF = 402 Ω, RL = 100 Ω, G = 2, and TA = 25°C (unless otherwise noted)(1)(2)(3)
PARAMETER
TEST CONDITIONS
MIN
TYP MAX
22
UNIT
To 0.02%
G = 2, VO = 2-V step,
Settling time
ns
Test level = C
To 0.1%
18
TA = 25°C
–85 –81
–80
G = 2, f = 1 MHz,
VO = 2 VPP, RL = 200 Ω,
Test level = B
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 25°C
–79
Harmonic distortion, 2nd-harmonic
dBc
dBc
–90 –85
–83
G = 2, f = 1 MHz,
VO = 2 VPP, RL ≥ 500 Ω,
Test level = B
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 25°C
–81
–95 –90
–89
G = 2, f = 1 MHz,
VO = 2 VPP, RL = 200 Ω,
Test level = B
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 25°C
–88
Harmonic distortion, 3rd-harmonic
–110 –105
–102
G = 2, f = 1 MHz,
VO = 2 VPP, RL ≥ 500 Ω,
Test level = B
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 25°C
–100
2.5
1.7
2.7
2.8
2.9
2.6
2.8
3
Input voltage noise
Input current noise
f > 100 kHz, Test level = B
f > 100 kHz, Test level = B
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 25°C
nV/√Hz
pA/√Hz
°
TA = 0°C to 70°C
TA = –40°C to 85°C
Differential gain
G = 2, PAL, VO = 1.4 VPP, RL = 150 Ω, Test level = C
G = 2, PAL, VO = 1.4 VPP, RL = 150 Ω, Test level = C
0.01%
0.03
Differential phase
DC PERFORMANCE(4)
TA = 25°C
62
61
60
66
AOL
Open-loop voltage gain
Input offset voltage
VCM = 0 V, Test level = A
TA = 0°C to 70°C
dB
TA = –40°C to 85°C
±0.7
5
TA = 25°C
±0.2
VCM = 0 V, Test level = A
mV
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 25°C
±1
±1.2
4
Average input offset voltage drift
Input bias current
VCM = 0 V, Test level = B
VCM = 0 V, Test level = A
VCM = 0 V, Test level = B
VCM = 0 V, Test level = A
VCM = 0 V, Test level = B
µV/°C
µA
4
–9 –17
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 25°C
–19
–23
30
Average input bias current drift
Input offset current
nA/°C
nA
50
±100 ±400
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 0°C to 70°C
TA = –40°C to 85°C
±600
±700
5
Inverting input bias-current drift
nA/°C
5
(4) Current is considered positive out-of-node. VCM is the input common-mode voltage.
Copyright © 2004–2016, Texas Instruments Incorporated
Submit Documentation Feedback
5
Product Folder Links: OPA820
OPA820
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
www.ti.com
Electrical Characteristics: VS = ±5 V (continued)
RF = 402 Ω, RL = 100 Ω, G = 2, and TA = 25°C (unless otherwise noted)(1)(2)(3)
PARAMETER
TEST CONDITIONS
MIN
TYP MAX
UNIT
V
INPUT
TA = 25°C
±3.8
±3.7
±3.6
76
±4
CMIR
Common-mode input range(5)
Common-mode rejection ratio
Test level = A
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 25°C
85
VCM = 0 V, Input-referred,
Test level = A
CMRR
TA = 0°C to 70°C
TA = –40°C to 85°C
75
dB
73
Input impedance, differential mode VCM = 0 V, TA = 25°C, Test level = C
18 || 0.8
kΩ || pF
MΩ || pF
Input impedance, common mode
Output voltage swing
Output current
VCM = 0 V, TA = 25°C, Test level = C
6 || 1
OUTPUT
TA = 25°C
±3.5
±3.7
±3.4
2
No load, Test level = A
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 25°C
±3.4
±3.5
V
±3.6
±3.4
5
RL = 100 Ω, Test level = A
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 25°C
±3.4
±90
±80
±75
±110
VO = 0 V, Test level = A
TA = 0°C to 70°C
TA = –40°C to 85°C
mA
Short-circuit output current
Output shorted to ground, Test level = C
±125
0.04
mA
Closed-loop output impedance
G = 2, f ≤ 100 kHz, Test level = C
Ω
POWER SUPPLY
TA = 25°C
5.45
5
5.6 5.75
Quiescent current
VS = ±5 V, Test level = A
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 25°C
6.2
6.4
mA
dB
4.8
64
63
62
72
PSRR
Power-supply rejection ratio
Input referred, Test level = A
TA = 0°C to 70°C
TA = –40°C to 85°C
(5) Tested at less than 3 dB below the minimum specified CMRR at ± CMIR limits.
6
Submit Documentation Feedback
Copyright © 2004–2016, Texas Instruments Incorporated
Product Folder Links: OPA820
OPA820
www.ti.com
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
7.6 Electrical Characteristics: VS = 5 V
RF = 402 Ω, RL = 100 Ω, G = 2, and TA = 25°C (unless otherwise noted)(1)(2)(3)
PARAMETER
AC PERFORMANCE
TEST CONDITIONS
MIN
TYP
MAX
UNIT
MHz
MHz
G = 1, VO = 0.1 VPP, RF = 0 Ω, Test level = C
550
230
TA = 25°C
168
155
151
21
G = 2, VO = 0.1 VPP
,
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 25°C
Test level = B
Small-signal bandwidth
28
G = 10, VO = 0.1 VPP
,
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 25°C
20
Test level = B
19
200
190
185
260
Gain-bandwidth product
G ≥ 20, Test level = B
TA = 0°C to 70°C
TA = –40°C to 85°C
Peaking at a gain of 1
Large-signal bandwidth
VO = 0.1 VPP, RF = 0 Ω, Test level = C
0.5
70
dB
G = 2, VO = 2 VPP, Test level = C
MHz
TA = 25°C
145
140
135
200
G = 2, 2-V step,
Test level = B
Slew rate
TA = 0°C to 70°C
TA = –40°C to 85°C
G = 2, VO = 2-V step, Test level = C
V/µs
Rise time and fall time
Settling time
1.7
24
ns
ns
To 0.02%
G = 2, VO = 2-V step,
Test level = C
To 0.1%
21
TA = 25°C
–80
–76
–75
–74
–79
–77
–75
–92
–91
–90
–95
–93
–92
2.8
G = 2, f = 1 MHz,
VO = 2 VPP, RL = 200 Ω,
Test level = B
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 25°C
Harmonic distortion, 2nd-harmonic
dBc
dBc
–83
–100
–98
2.5
G = 2, f = 1 MHz,
VO = 2 VPP, RL ≥ 500 Ω,
Test level = B
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 25°C
G = 2, f = 1 MHz,
VO = 2 VPP, RL = 200 Ω,
Test level = B
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 25°C
Harmonic distortion, 3rd-harmonic
G = 2, f = 1 MHz,
VO = 2 VPP, RL ≥ 500 Ω,
Test level = B
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 25°C
Input voltage noise
Input current noise
f > 100 kHz, Test level = B TA = 0°C to 70°C
TA = –40°C to 85°C
2.9 nV/√Hz
3
TA = 25°C
1.6
2.5
f > 100 kHz, Test level = B TA = 0°C to 70°C
TA = –40°C to 85°C
2.7 pA/√Hz
2.9
(1) Test levels: (A) 100% tested at 25°C. Overtemperature limits by characterization and simulation. (B) Limits set by characterization and
simulation. (C) Typical value only for information.
(2) TJ = TA for 25°C specifications.
(3) TJ = TA at low temperature limits; TJ = TA + 9°C at high temperature limit for over temperature.
Copyright © 2004–2016, Texas Instruments Incorporated
Submit Documentation Feedback
7
Product Folder Links: OPA820
OPA820
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
www.ti.com
Electrical Characteristics: VS = 5 V (continued)
RF = 402 Ω, RL = 100 Ω, G = 2, and TA = 25°C (unless otherwise noted)(1)(2)(3)
PARAMETER
DC PERFORMANCE(4)
TEST CONDITIONS
MIN
TYP
MAX
UNIT
TA = 25°C
60
59
58
65
AOL
Open-loop voltage gain
VO = 2.5 V, Test level = A
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 25°C
dB
±0.3
±1.1
±1.4
±1.6
4
Input offset voltage
VCM = 2.5 V, Test level = A TA = 0°C to 70°C
TA = –40°C to 85°C
mV
µV/°C
µA
TA = 0°C to 70°C
VCM = 2.5 V, Test level = B
Average input offset voltage drift
Input bias current
TA = –40°C to 85°C
4
TA = 25°C
VCM = 2.5 V, Test level = A TA = 0°C to 70°C
TA = –40°C to 85°C
–8
–16
–18
–22
30
TA = 0°C to 70°C
VCM = 2.5 V, Test level = B
Average input bias current drift
Input offset current
nA/°C
nA
TA = –40°C to 85°C
50
TA = 25°C
VCM = 2.5 V, Test level = A TA = 0°C to 70°C
TA = –40°C to 85°C
±100
±400
±600
±700
5
TA = 0°C to 70°C
VCM = 2.5 V, Test level = B
Inverting input bias-current drift
nA/°C
TA = –40°C to 85°C
5
INPUT
TA = 25°C
0.9
4.5
1.1
1.2
1.3
Lease positive input voltage
Most positive input voltage
Common-mode rejection ratio
Test level = A
Test level = A
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 25°C
V
V
4.2
4.1
4
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 25°C
74
73
72
83
VCM = 2.5 V, Input-referred,
Test level = A
CMRR
TA = 0°C to 70°C
TA = –40°C to 85°C
dB
Input impedance, differential mode VCM = 2.5 V, TA = 25°C, Test level = C
Input impedance, common mode VCM = 2.5 V, TA = 25°C, Test level = C
(4) Current is considered positive out-of-node. VCM is the input common-mode voltage.
15 || 1
kΩ || pF
MΩ || pF
5 ||
1.3
8
Submit Documentation Feedback
Copyright © 2004–2016, Texas Instruments Incorporated
Product Folder Links: OPA820
OPA820
www.ti.com
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
Electrical Characteristics: VS = 5 V (continued)
RF = 402 Ω, RL = 100 Ω, G = 2, and TA = 25°C (unless otherwise noted)(1)(2)(3)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
OUTPUT
TA = 25°C
3.8
3.75
3.7
3.9
No load, Test level = A
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 25°C
Most positive output voltage
V
3.7
3.8
1.2
RL = 100 Ω to 2.5 V,
Test level = A
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 25°C
3.65
3.6
1.3
1.35
1.4
No load, Test level = A
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 25°C
Least positive output voltage
V
1.2
1.3
RL = 100 Ω to 2.5 V,
Test level = A
TA = 0°C to 70°C
TA = –40°C to 85°C
TA = 25°C
1.35
1.4
±80
±70
±65
±105
Output current
VO = 2.5 V, Test level = A
TA = 0°C to 70°C
TA = –40°C to 85°C
mA
Short-circuit output current
Output shorted to ground, Test level = C
±115
0.04
mA
Closed-loop output impedance
G = 2, f ≤ 100 kHz, Test level = C
Ω
POWER SUPPLY
TA = 25°C
4.4
4.25
4.1
5
5.4
5.5
5.6
Quiescent current
VS = ±5 V, Test level = A
TA = 0°C to 70°C
TA = –40°C to 85°C
mA
dB
PSRR
Power-supply rejection ratio
Input referred, TA = 25°C, Test level = A
68
Copyright © 2004–2016, Texas Instruments Incorporated
Submit Documentation Feedback
9
Product Folder Links: OPA820
OPA820
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
www.ti.com
7.7 Typical Characteristics
7.7.1 ±5-V Supply Voltage
VS = ±5 V, RF = 402 Ω, RL = 100 Ω, G = 2, and TA = 25°C (unless otherwise noted)
3
0
3
0
−
3
6
9
−
3
6
9
−
−
−
−
−
12
−
15
−
18
−
12
−
15
−
18
G = 1
G = 2
G = 5
G = 10
G = –1
G = –2
G = –5
G = –10
1M
10M
100M
1G
1
10
100
500
Frequency (Hz)
Frequency (MHz)
See Figure 56
VO = 0.1 VPP
RF = 0 Ω at G = 1
See Figure 55
VO = 0.1 VPP
Figure 1. Noninverting Small-Signal Frequency Response
Figure 2. Inverting Small-Signal Frequency Response
9
6
3
0
3
0
−
3
6
9
−
−
−
−
−
3
6
9
−
−
−
12
15
18
VO = 0.5 VPP
VO = 1 VPP
VO = 2 VPP
VO = 4 VPP
VO = 0.5 VPP
VO = 1 VPP
VO = 2 VPP
VO = 4 VPP
−
12
1
10
100
500
1
10
100
500
Frequency (MHz)
See Figure 55
Frequency (MHz)
See Figure 56
G = 2
G = –1
Figure 3. Noninverting Large-Signal Frequency Response
Figure 4. Inverting Large-Signal Frequency Response
0.4
2.0
0.4
0.3
2.0
Large Signal 1 V
Small Signal 100 mV
0.3
0.2
1.5
1.5
1.0
0.2
1.0
0.1
0.5
0.1
0.5
0
0
0
0
−0.1
−0.2
−0.3
−0.4
−0.5
−1.0
−1.5
−2.0
−0.1
−0.2
−0.3
−0.4
−0.5
−1.0
−1.5
−2.0
Large Signal 1 V
Small Signal 100 mV
Time (10 ns/div)
Time (10 ns/div)
G = 2
See Figure 55
G = –1
See Figure 56
Figure 5. Noninverting Pulse Response
Figure 6. Inverting Pulse Response
10
Submit Documentation Feedback
Copyright © 2004–2016, Texas Instruments Incorporated
Product Folder Links: OPA820
OPA820
www.ti.com
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
±5-V Supply Voltage (continued)
VS = ±5 V, RF = 402 Ω, RL = 100 Ω, G = 2, and TA = 25°C (unless otherwise noted)
−
−
−
−
−
75
80
85
90
95
−
−
−
−
−
−
70
75
80
85
90
95
2nd−Harmonic
3rd−Harmonic
2nd−Harmonic
3rd−Harmonic
−
−
100
105
−
100
100
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
1k
Supply Voltage ( VS)
Resistance (Ω)
VO = 2 VPP
VO = 2 VPP
RL = 200 Ω
G = 2 V/V
f = 1 MHz
G = 2 V/V
See Figure 55
See Figure 55
Figure 8. 1-MHz Harmonic Distortion vs Supply Voltage
Figure 7. Harmonic Distortion vs Load Resistance
−
−
−
−
−
−
−
−
−
−
−
−
−
75
80
85
90
95
60
2nd−Harmonic
3rd−Harmonic
2nd−Harmonic
3rd−Harmonic
65
70
75
80
85
90
95
−
100
105
110
−
−
−
100
105
−
0.1
1
10
0.1
1
10
Output Voltage (VPP
)
Frequency (MHz)
G = 2 V/V
VO = 2 VPP
f = 1 MHz
RL = 200 Ω
G = 2 V/V
See Figure 55
See Figure 55
Figure 9. Harmonic Distortion vs Frequency
Figure 10. Harmonic Distortion vs Output Voltage
−
−
70
70
75
80
85
90
95
2nd−Harmonic
3rd−Harmonic
2nd−Harmonic
3rd−Harmonic
−
−
−
−
−
−
−
−
−
−
75
80
85
90
95
−
−
−
100
105
110
−
100
1
10
1
10
Gain (V/V)
Gain (|V/V|)
RL = 200 Ω
f = 1 MHz
RL = 200 Ω
VO = 2 VPP
f = 1 MHz
VO = 2 VPP
See Figure 55
See Figure 56
Figure 11. Harmonic Distortion vs Noninverting Gain
Figure 12. Harmonic Distortion vs Inverting Gain
Copyright © 2004–2016, Texas Instruments Incorporated
Submit Documentation Feedback
11
Product Folder Links: OPA820
OPA820
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
www.ti.com
±5-V Supply Voltage (continued)
VS = ±5 V, RF = 402 Ω, RL = 100 Ω, G = 2, and TA = 25°C (unless otherwise noted)
50
45
40
35
30
25
20
15
100
10
1
Voltage Noise (2.5 nV/√HZ)
Current Noise (1.7 pA/√HZ)
0
5
10
15
20
25
30
10
100
1k
10k
100k
1M
10M
Frequency (MHz)
See Figure 48
Frequency (Hz)
Figure 14. Two-Tone, 3rd-Order Intermodulation Intercept
Figure 13. Input Voltage and Current Noise
100
8
7
6
5
4
3
2
1
10
0
CL = 10 pF
CL = 22 pF
CL = 47 pF
CL = 100 pF
−
1
2
3
−
−
1
1
10
100
1000
1
10
100
400
Capacitive Load (pF)
Frequency (MHz)
See Figure 49
0-dB peaking targeted
Figure 15. Recommended RS vs Capacitive Load
Figure 16. Frequency Response vs Capacitive Load
90
80
0
20 log (AOL
∠ AOL
)
80
70
60
50
40
30
20
10
0
70
60
50
40
30
20
10
0
−20
−40
−60
−80
−100
−120
−140
−160
−180
CMRR
+PSRR
–PSRR
−10
1k
10k
100k
1M
10M
100M
100
1k
10k
100k
1M
10M 100M
1G
Frequency (Hz)
Frequency (Hz)
Figure 17. CMRR and PSRR vs Frequency
Figure 18. Open-Loop Gain and Phase
12
Submit Documentation Feedback
Copyright © 2004–2016, Texas Instruments Incorporated
Product Folder Links: OPA820
OPA820
www.ti.com
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
±5-V Supply Voltage (continued)
VS = ±5 V, RF = 402 Ω, RL = 100 Ω, G = 2, and TA = 25°C (unless otherwise noted)
5
4
3
2
1
0
10
1-W Internal
Power Limit
Output Current
Limit
1
−
1
2
3
4
5
0.1
0.01
−
−
−
−
Output Current
Limit
1-W Internal
RL = 25 Ω
RL = 50 Ω
RL = 100 Ω
50
Power Limit
1k
10k
100k
1M
10M
100M
−
−
−
50
150
100
0
100 150
Frequency (Hz)
Output Current (mA)
Figure 20. Closed-Loop Output Impedance vs Frequency
Figure 19. Output Voltage and Current Limitations
5
4
3
2
1
0
5
4
3
8
6
4
2
0
4
Input
3
2
1
0
−
−
−
−
Output
2
Input
1
0
−
−
−
−
−
−
1
2
3
4
5
1
2
3
4
5
−
2
4
6
8
1
2
3
4
Output
−
−
−
−
−
−
−
Time (40 ns/div)
Time (40 ns/div)
G = 2 V/V
See Figure 55
G = 2 V/V
See Figure 56
Figure 21. Noninverting Overdrive Recovery
Figure 22. Inverting Overdrive Recovery
0.20
0.18
0.16
0.14
0.12
0.10
0.08
0.06
0.04
0.02
0
0.40
0.36
0.32
0.28
0.24
0.20
0.16
0.12
0.08
0.04
0
1.0
0.5
0
20
10
0
10× Input Offset Current (IOS
)
)
dG Negative Video
dG Positive Video
dP Negative Video
dP Positive Video
Input Offset Voltage (VOS
Input Bias current (IB)
−
0.5
−
10
−1.0
−50
−20
−25
0
25
50
75
100
125
1
2
3
4
Ambient Temperature (°C)
Video Loads
G = 2 V/V
Figure 24. Typical DC Drift Over Temperature
Figure 23. Composite Video dG/dP
Copyright © 2004–2016, Texas Instruments Incorporated
Submit Documentation Feedback
13
Product Folder Links: OPA820
OPA820
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
www.ti.com
±5-V Supply Voltage (continued)
VS = ±5 V, RF = 402 Ω, RL = 100 Ω, G = 2, and TA = 25°C (unless otherwise noted)
125
100
75
12
6
5
4
3
2
1
0
Supply Current
Source Output Current
Sink Output Current
–VIN
+VIN
–VOUT
+VOUT
9
6
50
3
25
0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
−
50
−
25
0
25
50
75
100
125
Supply Voltage ( VS)
Ambient Temperature (°C)
Figure 25. Supply and Output Current vs Temperature
Figure 26. Common-Mode Input Range and Output Swing
vs Supply Voltage
10M
1M
2500
2000
1500
1000
500
100k
10k
1k
Common-Mode Input Impedance
Differential Input Impedance
0
100
1k
10k
100k
1M
10M
100M
Frequency (Hz)
µ
Input Offset Voltage ( V)
Mean = –30 µV
Total count = 6115
Standard deviation = 80 µV
Figure 27. Common-Mode and Differential Input
Impedance
Figure 28. Typical Input Offset Voltage Distribution
2000
1800
1600
1400
1200
1000
800
600
400
200
0
Input Offset Current (nA)
Mean = 26 nA
Standard deviation = 57 nA
Total count = 6115
Figure 29. Typical Input Offset Current Distribution
14
Submit Documentation Feedback
Copyright © 2004–2016, Texas Instruments Incorporated
Product Folder Links: OPA820
OPA820
www.ti.com
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
7.7.2 5-V Supply Voltage
VS = 5 V, RF = 402 Ω, RL = 100 Ω, G = 2, and TA = 25°C (unless otherwise noted)
3
0
3
0
−
3
6
9
−
3
6
9
−
−
−
−
−
12
−
15
−
18
−
12
−
15
−
18
G = 1
G = 2
G = 5
G = 10
G = –1
G = –2
G = –5
G = –10
1M
10M
100M
1G
1
10
100
500
Frequency (Hz)
Frequency (MHz)
See Figure 58
VO = 0.1 VPP
See Figure 57
VO = 0.1 VPP
Figure 30. Noninverting Small-Signal Frequency Response
Figure 31. Inverting Small-Signal Frequency Response
9
6
3
0
3
0
−
−
−
3
6
9
−
−
−
3
6
9
−
12
15
18
VO = 0.5 VPP
VO = 1 VPP
VO = 2 VPP
VO = 4 VPP
VO = 0.5 VPP
VO = 1 VPP
VO = 2 VPP
VO = 4 VPP
−
−
−
12
1
10
100
600
1
10
100
500
Frequency (MHz)
Frequency (MHz)
See Figure 58
G = 2 V/V
See Figure 57
G = –1
Figure 32. Noninverting Large-Signal Frequency Response
Figure 33. Inverting Large-Signal Frequency Response
2.9
2.8
2.7
2.6
2.5
2.4
2.3
2.2
2.1
4.5
4.0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
2.9
4.5
4.0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
Large Signal 1 V
Small Signal 100 mV
2.8
2.7
2.6
2.5
2.4
2.3
2.2
2.1
Large Signal 1 V
Small Signal 100 mV
Time (10 ns/div)
Time (10 ns/div)
G = 2
See Figure 57
G = –1
See Figure 58
Figure 34. Noninverting Pulse Response
Figure 35. Inverting Pulse Response
Copyright © 2004–2016, Texas Instruments Incorporated
Submit Documentation Feedback
15
Product Folder Links: OPA820
OPA820
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
www.ti.com
5-V Supply Voltage (continued)
VS = 5 V, RF = 402 Ω, RL = 100 Ω, G = 2, and TA = 25°C (unless otherwise noted)
−
−
−
−
−
−60
75
80
85
90
95
2nd−Harmonic
3rd−Harmonic
2nd−Harmonic
3rd−Harmonic
−
−
−
70
80
90
−
100
110
−
100
−
105
−
100
1k
0.1
1
10
Resistance (Ω)
f = 1 MHz
Frequency (MHz)
RL = 200 Ω
See Figure 57
G = 2 V/V
VO = 2 VPP
G = 2 V/V
VO = 2 VPP
See Figure 57
Figure 36. Harmonic Distortion vs Load Resistance
Figure 37. Harmonic Distortion vs Frequency
−
−
−
−
60
70
80
90
−
70
2nd−Harmonic
3rd−Harmonic
2nd−Harmonic
3rd−Harmonic
−
−
80
90
−
100
110
−
100
110
−
−
1
10
0.1
1
10
Gain (V/V)
Output Voltage Swing (VPP
)
f = 1 MHz
RL = 200 Ω
VO = 2 VPP
f = 1 MHz
RL = 200 Ω
VO = 2 VPP
G = 2 V/V
Figure 39. Harmonic Distortion vs Noninverting Gain
Figure 38. Harmonic Distortion vs Output Voltage
−
70
40
2nd−Harmonic
3rd−Harmonic
−
−
−
−
−
75
80
85
90
95
35
30
25
20
15
−
100
1
10
0
5
10
15
20
25
30
Gain (|V/V|)
Frequency (MHz)
See Figure 50
f = 1 MHz
RL = 200 Ω
VO = 2 VPP
Figure 40. Harmonic Distortion vs Inverting Gain
Figure 41. Two-Tone, 3rd-Order Intermodulation Intercept
16
Submit Documentation Feedback
Copyright © 2004–2016, Texas Instruments Incorporated
Product Folder Links: OPA820
OPA820
www.ti.com
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
5-V Supply Voltage (continued)
VS = 5 V, RF = 402 Ω, RL = 100 Ω, G = 2, and TA = 25°C (unless otherwise noted)
100
10
1
8
7
6
5
4
3
2
1
0
CL = 10 pF
CL = 22 pF
CL = 47 pF
CL = 100 pF
−
1
2
3
−
−
1
10
100
1000
1
10
100
300
Capacitive Load (pF)
Frequency (MHz)
See Figure 51
0-dB peaking targeted
Figure 42. Recommended RS vs Capacitive Load
Figure 43. Frequency Response vs Capacitive Load
1.5
15
10
5
125
12
10× Input Offset Current (IOS
)
)
Supply Current
Source Output Current
Sink Output Current
9
Input Offset Voltage (VOS
Input Bias current (IB)
1.0
0.5
100
75
0
0
6
3
0
−0.5
−1.0
−1.5
−5
−10
−15
50
25
−
50
−
25
0
25
50
75
100
125
−50
−25
0
25
50
75
100
125
Ambient Temperature ( C)
Ambient Temperature (°C)
Figure 45. Supply and Output Current vs Temperature
Figure 44. Typical DC Drift Over Temperature
3500
3000
2500
2000
1500
1000
500
2000
1800
1600
1400
1200
1000
800
600
400
200
0
0
Input Offset Voltage (mV)
Input Offset Current (nA)
Mean = 490 µV
Total count = 6115
Mean = 43 nA
Total count = 6115
Standard deviation = 90 µV
Standard deviation = 50 nA
Figure 46. Typical Input Offset Voltage Distribution
Figure 47. Typical Input Offset Current Distribution
Copyright © 2004–2016, Texas Instruments Incorporated
Submit Documentation Feedback
17
Product Folder Links: OPA820
OPA820
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
www.ti.com
8 Parameter Measurement Information
P
I
+
OPA820
P
O
50 ꢀ
200 ꢀ
402 ꢀ
402 ꢀ
Copyright © 2016, Texas Instruments Incorporated
Figure 48. Circuit for ±5-V Two-Tone, 3rd-Order Intermodulation Intercept (Figure 14)
V
I
R
S
+
OPA820
V
O
50 ꢀ
(1)
C
L
1 kꢀ
402 ꢀ
402 ꢀ
Copyright © 2016, Texas Instruments Incorporated
(1) 1 kΩ is optional.
Figure 49. Circuit for ±5-V Frequency Response vs Capacitive Load (Figure 55)
5 V
806 ꢀ
0.01 µF
P
I
+
OPA820
P
O
57.6 ꢀ
806 ꢀ
200 kꢀ
402 ꢀ
402 ꢀ
0.01 µF
Copyright © 2016, Texas Instruments Incorporated
Figure 50. Circuit for 5-V Two-Tone, 3rd-Order Intermodulation Intercept (Figure 41)
18
Submit Documentation Feedback
Copyright © 2004–2016, Texas Instruments Incorporated
Product Folder Links: OPA820
OPA820
www.ti.com
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
Parameter Measurement Information (continued)
5 V
806 ꢀ
0.01 µF
V
I
R
S
+
OPA820
V
O
57.6 ꢀ
806 ꢀ
(1)
C
L
1 kꢀ
402 ꢀ
402 ꢀ
0.01 µF
Copyright © 2016, Texas Instruments Incorporated
(1) This resistor is optional.
Figure 51. Circuit for 5-V Frequency Response vs Capacitive Load (Figure 43)
Copyright © 2004–2016, Texas Instruments Incorporated
Submit Documentation Feedback
19
Product Folder Links: OPA820
OPA820
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
www.ti.com
9 Detailed Description
9.1 Overview
The OPA820 provides an exceptional combination of DC precision, wide bandwidth, and low noise while
consuming 5.6 mA of quiescent current. With excellent performance extending from DC to high frequencies, the
OPA820 can be used in a variety of applications ranging from driving the inputs of high-precision SAR ADCs to
video distributions systems.
9.2 Feature Description
9.2.1 Input and ESD Protection
The OPA820 device is built using a very high-speed complementary bipolar process. The internal junction
breakdown voltages are relatively low for these very small geometry devices. These breakdowns are reflected in
Absolute Maximum Ratings. All device pins are protected with internal ESD-protection diodes to the power
supplies, as shown in Figure 52.
V
CC
External
Pin
œV
CC
Copyright © 2016, Texas Instruments Incorporated
Figure 52. Internal ESD Protection
These diodes provide moderate protection to input overdrive voltages above the supplies as well. The protection
diodes can typically support 30-mA continuous current. Where higher currents are possible (for example, in
systems with ±15-V supply parts driving into the OPA820 device), add current-limiting series resistors into the
two inputs. Keep these resistor values as low as possible because high values degrade both noise performance
and frequency response. Figure 53 shows an example protection circuit for I/O voltages that may exceed the
supplies.
5 V
50-ꢀ Source
Power-supply
decoupling not shown.
174 ꢀ
V
1
+
50 ꢀ
D1
D2
50 ꢀ
OPA820
V
O
œ
50 ꢀ
R
F
301 ꢀ
R
301 ꢀ
G
œ5 V
Copyright © 2016, Texas Instruments Incorporated
D1 = D2; IN5911 (or equivalent)
Figure 53. Gain of 2 With Input Protection
20
Submit Documentation Feedback
Copyright © 2004–2016, Texas Instruments Incorporated
Product Folder Links: OPA820
OPA820
www.ti.com
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
Feature Description (continued)
9.2.2 Bandwidth versus Gain
Voltage-feedback op amps exhibit decreasing closed-loop bandwidth as the signal gain is increased. In theory,
this relationship is described by the GBP listed in Specifications. Ideally, dividing GBP by the noninverting signal
gain (also called the noise gain, or NG) predicts the closed-loop bandwidth. In practice, this prediction only holds
true when the phase margin approaches 90°, as it does in high-gain configurations. At low signal gains, most
amplifiers exhibit a more complex response with lower phase margin. The OPA820 device is optimized to give a
maximally-flat, 2nd-order Butterworth response in a gain of 2. In this configuration, the OPA820 device has
approximately 64° of phase margin and shows a typical –3-dB bandwidth of 240 MHz. When the phase margin is
64°, the closed-loop bandwidth is approximately √2 greater than the value predicted by dividing GBP by the
noise gain.
Increasing the gain causes the phase margin to approach 90° and the bandwidth to more closely approach the
predicted value of GBP / NG. At a gain of 10, the 30-MHz bandwidth shown in Electrical Characteristics: VS = ±5
V matches the prediction of the simple formula using the typical GBP of 280 MHz.
9.2.3 Output Drive Capability
The OPA820 device has been optimized to drive the demanding load of a doubly-terminated transmission line.
When a 50-Ω line is driven, a series 50-Ω source resistor leading into the cable and a terminating 50-Ω load
resistor at the end of the cable are used. Under these conditions, the cable impedance seems resistive over a
wide frequency range, and the total effective load on the OPA820 device is 100 Ω in parallel with the resistance
of the feedback network. Specifications lists a ±3.6-V swing into this load—which is then reduced to a ±1.8-V
swing at the termination resistor. The ±75-mA output drive over temperature provides adequate current-drive
margin for this load. Higher voltage swings (and lower distortion) are achievable when driving higher impedance
loads.
A single video load typically appears as a 150-Ω load (using standard 75-Ω cables) to the driving amplifier. The
OPA820 device provides adequate voltage and current drive to support up to three parallel video loads (50-Ω
total load) for an NTSC signal. With only one load, the OPA820 device achieves an exceptionally low 0.01% or
0.03° dG/dP error.
9.2.4 Driving Capacitive Loads
One of the most demanding, and yet very common, load conditions for an op amp is capacitive loading. A high-
speed, high open-loop gain amplifier like the OPA820 device can be very susceptible to decreased stability and
closed-loop response peaking when a capacitive load is placed directly on the output pin. In simple terms, the
capacitive load reacts with the open-loop output resistance of the amplifier to introduce an additional pole into the
loop and thereby decrease the phase margin. This issue has become a popular topic of application notes and
articles, and several external solutions to this problem have been suggested. When the primary considerations
are frequency response flatness, pulse response fidelity, distortion, or a combination, the simplest and most
effective solution is to isolate the capacitive load from the feedback loop by inserting a series isolation resistor
between the amplifier output and the capacitive load. This solution does not eliminate the pole from the loop
response, but rather shifts it and adds a zero at a higher frequency. The additional zero acts to cancel the phase
lag from the capacitive load pole, thus increasing the phase margin and improving stability.
Figure 15 (±5 V) and Figure 42 (5 V) show the recommended RS versus capacitive load and the resulting
frequency response at the load. The criterion for setting the recommended resistor is the maximum-bandwidth,
flat-frequency response at the load. Because a passive low-pass filter is now between the output pin and the
load capacitance, the response at the output pin is typically somewhat peaked, and becomes flat after the roll-off
action of the RC network. This response is not a concern in most applications, but can cause clipping if the
desired signal swing at the load is very close to the swing limit of the amplifier. Such clipping most likely to
occurs in pulse response applications where the frequency peaking is manifested as an overshoot in the step
response.
Parasitic capacitive loads greater than 2 pF can begin to degrade the performance of the OPA820 device. Long
printed-circuit board traces, unmatched cables, and connections to multiple devices can easily cause this value
to be exceeded. Always consider this effect carefully, and add the recommended series resistor as close as
possible to the OPA820 output pin (see Layout Guidelines).
Copyright © 2004–2016, Texas Instruments Incorporated
Submit Documentation Feedback
21
Product Folder Links: OPA820
OPA820
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
www.ti.com
Feature Description (continued)
9.2.5 Distortion Performance
The OPA820 device is capable of delivering an exceptionally-low distortion signal at high frequencies and low
gains. The distortion plots in Typical Characteristics show the typical distortion under a wide variety of conditions.
Most of these plots are limited to 100-dB dynamic range. The OPA820 distortion does not rise above –90 dBc
until either the signal level exceeds 0.9 V, the fundamental frequency exceeds 500 kHz, or both occur. Distortion
in the audio band is less than or equal to –100 dBc.
Generally, until the fundamental signal reaches very high frequencies or powers, the 2nd-harmonic dominates
the distortion with a negligible 3rd-harmonic component. Focusing then on the 2nd-harmonic, increasing the load
impedance improves distortion directly. Remember that the total load includes the feedback network—in the
noninverting configuration this is the sum of RF + RG, whereas in the inverting configuration this is just RF (see
Figure 55). Increasing the output voltage swing directly increases harmonic distortion. Increasing the signal gain
also increases the 2nd-harmonic distortion. Again, a 6-dB increase in gain increases the 2nd and 3rd-harmonic
by 6 dB even with a constant output power and frequency. Finally, the distortion increases as the fundamental
frequency increases because of the roll-off in the loop gain with frequency. Conversely, the distortion improves
going to lower frequencies down to the dominant open-loop pole at approximately 100 kHz. Starting from the
–85-dBc 2nd-harmonic for 2 VPP into 200 Ω, G = 2 distortion at 1 MHz (from Typical Characteristics), the 2nd-
harmonic distortion does not show any improvement below 100 kHz and then becomes Equation 1.
–100 dB – 20log (1 MHz / 100 kHz) = –105 dBc
(1)
9.2.6 Noise Performance
The OPA820 device complements low harmonic distortion with low input-noise terms. Both the input-referred
voltage noise and the two input-referred current noise terms combine to give a low output noise under a wide
variety of operating conditions. Figure 54 shows the op amp noise analysis model with all the noise terms
included. In this model, all the noise terms are taken to be noise voltage or current density terms in either nV/√Hz
or pA/√Hz.
E
NI
OPA820
E
O
+
R
S
I
BN
E
RS
R
F
4kTRS
4kTRF
R
G
I
BI
4kT
RG
4kT = 1.6E œ 20J
at 290°K
Copyright © 2016, Texas Instruments Incorporated
Figure 54. Op Amp Noise Analysis Model
The total output spot noise voltage is computed as the square root of the squared contributing terms to the
output noise voltage. This computation is adding all the contributing noise powers at the output by superposition,
then taking the square root to get back to a spot noise voltage. Equation 2 shows the general form for this output
noise voltage using the terms presented in Figure 54.
ENI + I R + 4kTR NG + I R 2 + 4kTRFNG
» ÿ
BN S S BI F
2
2
EO
=
(
)
(
)
⁄
(2)
22
Submit Documentation Feedback
Copyright © 2004–2016, Texas Instruments Incorporated
Product Folder Links: OPA820
OPA820
www.ti.com
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
Feature Description (continued)
Dividing this expression by the noise gain (NG = 1 + RF / RG) gives the equivalent input referred spot noise
voltage at the noninverting input, as shown in Equation 3.
2
I R
4kTRF
NG
≈
’
2
EN = EN2I + IBNRS + 4kTRS +
+
BI
F
(
)
∆
«
÷
◊
NG
(3)
Evaluating these two equations for the OPA820 circuit shown in Figure 55 gives a total output spot noise voltage
of 6.44 nV/√Hz and an equivalent input spot noise voltage of 3.22 nV/√Hz.
9.2.7 DC Offset Control
The OPA820 device can provide excellent DC-signal accuracy because of high open-loop gain, high common-
mode rejection, high power-supply rejection, low input-offset voltage, and low bias-current offset errors. To take
full advantage of this low input-offset voltage, careful attention to input bias-current cancellation is also required.
The high-speed input stage for the OPA820 device has a moderately high input bias current (9 µA typical into the
pins) but with a very close match between the two input currents—typically 100-nA input offset current. The total
output-offset voltage can be considerably reduced by matching the source impedances looking out of the two
inputs. For example, one way to add bias current cancellation to the circuit of Figure 55 is to insert a 175-Ω
series resistor into the noninverting input from the 50-Ω terminating resistor. When the 50-Ω source resistor is
DC-coupled, the source impedance for the noninverting input bias current increases to 200 Ω. Because this value
is now equal to the impedance looking out of the inverting input (RF || RG), the circuit cancels the gains for the
bias currents to the output leaving only the offset current times the feedback resistor as a residual DC error term
at the output. Using a 402-Ω feedback resistor, this output error is now less than ±0.4 µA × 402 Ω = ±160 µV at
25°C.
9.2.8 Thermal Analysis
The OPA820 device does not require heat sinking or airflow in most applications. The maximum desired junction
temperature sets the maximum allowed internal power dissipation as described in this section. Make sure that
the maximum junction temperature does not exceed 150°C.
Use Equation 4 to calculate the operating junction temperature (TJ).
TA + PD × RθJA
(4)
The total internal power dissipation (PD) is the sum of quiescent power (PDQ) and additional power dissipated in
the output stage (PDL) to deliver load power. Quiescent power is the specified no-load supply current times the
total supply voltage across the part. PDL depends on the required output signal and load but, for a grounded
resistive load, is at a maximum when the output is fixed at a voltage equal to ½ of either supply voltage (for equal
bipolar supplies). Under this worst-case condition, use Equation 5 to calculate PDL
.
PDL = VS2 / (4 × RL)
where
•
RL includes feedback network loading.
(5)
NOTE
The power in the output stage and not in the load that determines internal power
dissipation.
As a worst-case example, compute the maximum TJ using an OPA820IDBV (SOT23-5 package) in the circuit of
Figure 55 operating at the maximum specified ambient temperature of 85°C.
PD = 10 V (6.4 mA) + 52 / (4 × (100 Ω || 800 Ω)) = 134 mW
(6)
(7)
Maximum TJ = 85°C + (134 mW × 150°C/W) = 105°C
Copyright © 2004–2016, Texas Instruments Incorporated
Submit Documentation Feedback
23
Product Folder Links: OPA820
OPA820
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
www.ti.com
9.3 Device Functional Modes
9.3.1 Wideband Noninverting Operation
The combination of speed and dynamic range offered by the OPA820 device is easily achieved in a wide variety
of application circuits, providing that simple principles of good design practice are observed. For example, good
power-supply decoupling, as shown in Figure 55, is essential to achieve the lowest-possible harmonic distortion
and smooth frequency response.
Proper PCB layout and careful component selection maximize the performance of the OPA820 device in all
applications, as discussed in the following sections of this data sheet.
Figure 55 shows the gain of 2 configuration used as the basis for most of the typical characteristics. Most of the
curves in Typical Characteristics were characterized using signal sources with a 50-Ω driving impedance and
with measurement equipment presenting 50-Ω load impedance. In Figure 55, the 50-Ω shunt resistor at the VI
terminal matches the source impedance of the test generator while, the 50-Ω series resistor at the VO terminal
provides a matching resistor for the measurement equipment load. Generally, data sheet specifications refer to
the voltage swings at the output pin (VO in Figure 55). The 100-Ω load, combined with the 804-Ω total feedback
network load, presents the OPA820 device with an effective load of approximately 90 Ω in Figure 55.
5 V
+V
S
+
0.1 µF
2.2 µF
50-ꢀ Source
50-ꢀ Load
50 ꢀ
V
+
V
O
IN
OPA820
50 ꢀ
R
F
402 ꢀ
R
G
402 ꢀ
+
0.1 µF
2.2 µF
œV
S
œ5 V
Copyright © 2016, Texas Instruments Incorporated
Figure 55. Gain of 2, High-Frequency Application and Characterization Circuit
9.3.2 Wideband Inverting Operation
Operating the OPA820 device as an inverting amplifier has several benefits and is particularly useful when a
matched 50-Ω source and input impedance is required. Figure 56 shows the inverting gain of –1 circuit used as
the basis of the inverting mode curves in Typical Characteristics.
24
Submit Documentation Feedback
Copyright © 2004–2016, Texas Instruments Incorporated
Product Folder Links: OPA820
OPA820
www.ti.com
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
Device Functional Modes (continued)
5 V
+
0.1 µF
2.2 µF
50-ꢀ Load
50 ꢀ
+
V
O
OPA820
R
205 ꢀ
T
0.01 µF
R
402 ꢀ
R
F
402 ꢀ
G
50-ꢀ Source
V
I
R
M
57.6 ꢀ
+
0.1 µF
2.2 µF
œ5 V
Copyright © 2016, Texas Instruments Incorporated
Figure 56. Inverting G = –1 Specifications and Test Circuit
In the inverting case, only the feedback resistor appears as part of the total output load in parallel with the actual
load. For the 100-Ω load used in the curves in Typical Characteristics, this results in a total load of 80 Ω in this
inverting configuration. The gain resistor is set to get the desired gain (in this case 402 Ω for a gain of –1) while
an additional input-matching resistor (RM) can be used to set the total input impedance equal to the source if
desired. In this case, RM is 57.6 Ω in parallel with the 402-Ω gain setting resistor results in a matched input
impedance of 50 Ω. This matching is only required when the input must be matched to a source impedance, as
in the characterization testing done using the circuit of Figure 56.
The OPA820 device offers extremely good DC accuracy as well as low noise and distortion. To take full
advantage of that DC precision, the total DC impedance at each of the input nodes must be matched to get bias
current cancellation. For the circuit of Figure 56, this matching requires the 205-Ω resistor to ground on the
noninverting input. The calculation for this resistor includes a DC-coupled 50-Ω source impedance along with RG
and RM. Although this resistor provides cancellation for the bias current, it must be well decoupled (0.01 µF in
Figure 56) to filter the noise contribution of the resistor and the input current noise.
As the required RG resistor approaches 50 Ω at higher gains, the bandwidth for the circuit in Figure 56 exceeds
the bandwidth at that same gain magnitude for the noninverting circuit of Figure 55 which occurs because of the
lower noise gain for the circuit of Figure 56 when the 50-Ω source impedance is included in the analysis. For
instance, at a signal gain of –10 (RG = 50 Ω, RM = open, RF = 499 Ω) the noise gain for the circuit of Figure 56 is
shown in Equation 8.
1 + 499 Ω / (50 Ω + 50 Ω) = 6
(8)
Equation 8 is a result of adding the 50-Ω source in the noise gain equation which results in a considerably higher
bandwidth than the noninverting gain of 10. Using the 240-MHz gain bandwidth product for the OPA820 device,
an inverting gain of –10 from a 50-Ω source to a 50-Ω RG gives 55-MHz bandwidth, whereas the noninverting
gain of 10 gives 30 MHz.
Copyright © 2004–2016, Texas Instruments Incorporated
Submit Documentation Feedback
25
Product Folder Links: OPA820
OPA820
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
www.ti.com
Device Functional Modes (continued)
9.3.3 Wideband Single-Supply Operation
Figure 57 shows the AC-coupled, single 5-V supply, gain of 2-V/V circuit configuration used as a basis only for
the 5-V specifications in Specifications. The most important requirement for single-supply operation is to maintain
input and output-signal swings within the useable voltage ranges at both the input and the output. The circuit of
Figure 57 establishes an input midpoint bias using a simple resistive divider from the 5-V supply (two 806-Ω
resistors) to the noninverting input. The input signal is then AC-coupled into this midpoint voltage bias. The input
voltage can swing to within 0.9 V of the negative supply and 0.5 V of the positive supply, giving a 3.6-VPP input-
signal range. The input impedance-matching resistor (57.6 Ω) used in Figure 57 is adjusted to give a 50-Ω input
match when the parallel combination of the biasing divider network is included. The gain resistor (RG) is AC-
coupled, giving the circuit a DC gain of 1 which puts the input DC bias voltage (2.5 V) on the output as well. On a
single 5-V supply, the output voltage can swing to within 1.3 V of either supply pin while delivering more than 80-
mA output current giving 2.4-V output swing into 100 Ω (5.6 dBm maximum at the matched load).
Figure 58 shows the AC-coupled, single 5-V supply, gain of –1-V/V circuit configuration used as a basis only for
the 5-V specifications in Specifications. In this case, the midpoint DC bias on the noninverting input is also
decoupled with an additional 0.01-µF decoupling capacitor which reduces the source impedance at higher
frequencies for the noninverting-input bias-current noise. This 2.5-V bias on the noninverting input pin appears
on the inverting input pin and, because RG is DC blocked by the input capacitor, also appears at the output pin.
The single-supply test circuits of Figure 57 and Figure 58 show 5-V operation. These same circuits can be used
with a single-supply of 5 V to 12 V. Operating on a single 12-V supply, with the absolute-maximum supply-
voltage specification of 13 V, gives adequate design margin for the typical ±5% supply tolerance.
5 V
+V
S
+
0.1 µF
6.8 µF
50-ꢀ Source
806 ꢀ
DIS
100 ꢀ
0.01 µF
V
I
+
V
O
OPA820
V /2
S
57.6 ꢀ
806 ꢀ
R
F
402 ꢀ
R
G
402 ꢀ
0.01 µF
Copyright © 2016, Texas Instruments Incorporated
Figure 57. AC-Coupled, G = 2 V/V, Single-Supply Specifications and Test Circuit
26
Submit Documentation Feedback
Copyright © 2004–2016, Texas Instruments Incorporated
Product Folder Links: OPA820
OPA820
www.ti.com
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
Device Functional Modes (continued)
5 V
+V
S
+
0.1 µF
6.8 µF
806 ꢀ
DIS
100 ꢀ
+
V
O
OPA820
V /2
S
806 ꢀ
0.01 µF
0.01 µF
R
G
R
F
402 ꢀ
402 ꢀ
V
I
Copyright © 2016, Texas Instruments Incorporated
Figure 58. AC-Coupled, G = –1 V/V, Single-Supply Specifications and Test Circuit
Copyright © 2004–2016, Texas Instruments Incorporated
Submit Documentation Feedback
27
Product Folder Links: OPA820
OPA820
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
www.ti.com
10 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
10.1 Application Information
10.1.1 Optimizing Resistor Values
Because the OPA820 device is a unity-gain stable, voltage-feedback op amp, a wide range of resistor values can
be used for the feedback and gain-setting resistors. The primary limits on these values are set by dynamic range
(noise and distortion) and parasitic capacitance considerations. Usually, the feedback resistor value is from
200 Ω to 1 kΩ. At less than 200 Ω, the feedback network presents additional output loading which can degrade
the harmonic distortion performance of the OPA820 device. At greater than 1 kΩ, the typical parasitic
capacitance (approximately 0.2 pF) across the feedback resistor can cause unintentional band limiting in the
amplifier response. A direct short is suggested as a feedback for AV = 1 V/V.
A good design practice is to target the parallel combination of RF and RG (see Figure 55) to be less than
approximately 200 Ω. The combined impedance RF || RG interacts with the inverting input capacitance, placing an
additional pole in the feedback network, and thus a zero in the forward response. Assuming a total parasitic of
2 pF on the inverting node, holding RF || RG < 200 Ω keeps this pole above 400 MHz. This constraint implies that
the feedback resistor RF can increase to several kΩ at high gains which is acceptable as long as the pole formed
by RF and any parasitic capacitance appearing in parallel is kept out of the frequency range of interest.
In the inverting configuration, an additional design consideration must be considered. RG becomes the input
resistor and therefore the load impedance to the driving source. If impedance matching is desired, RG can be set
equal to the required termination value. However, at low inverting gains, the resulting feedback resistor value can
present a significant load to the amplifier output. For example, an inverting gain of 2 with a 50-Ω input matching
resistor (RG) requires a 100-Ω feedback resistor, which contributes to output loading in parallel with the external
load. In such a case, increasing both the RF and RG values is preferable, and then achieve the input matching
impedance with a third resistor to ground (see Figure 56). The total input impedance becomes the parallel
combination of RG and the additional shunt resistor.
10.2 Typical Applications
10.2.1 Active Filter Design
Most active filter topologies have exceptional performance using the broad bandwidth and unity-gain stability of
the OPA820 device. Topologies employing capacitive feedback require a unity-gain stable, voltage-feedback op
amp. Sallen-Key filters simply use the op amp as a noninverting gain stage inside an RC network. Either current
feedback or voltage-feedback op amps can be used in Sallen-Key implementations.
28
Submit Documentation Feedback
Copyright © 2004–2016, Texas Instruments Incorporated
Product Folder Links: OPA820
OPA820
www.ti.com
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
Typical Applications (continued)
C
2
R
3
160 pF
500 ꢀ
+
OPA820
R
4
500 ꢀ
5 V
R
124 ꢀ
R
2
517 ꢀ
1
R
158 ꢀ
5
C
V
+
2
1
V
OPA820
1000 pF
O
R
158 ꢀ
2
C
1
R
1
100 pF
V
OPA820
OUT
1.58 kꢀ
R
402 ꢀ
F
V
IN
+
C
1
1000 pF
R
402 ꢀ
G
œ5 V
Copyright © 2016, Texas Instruments Incorporated
Copyright © 2016, Texas Instruments Incorporated
Figure 59. 5-MHz Butterworth Low-Pass Active
Filter
Figure 60. High-Q 1-MHz Bandpass Filter
10.2.1.1 Design Requirements
The design requirements for the active filters are given in Table 2:
Table 2. Design Requirements
FILTER CUTOFF FREQUENCY
f-3dB (MHz)
FILTER TYPE
Q
DC GAIN (dB)
Second order Butterworth, low-pass filter
High-Q, bandpass filter
0.707
10
5
1
6
—
10.2.1.2 Detailed Design Procedure
10.2.1.2.1 High-Q Bandpass Filter Design Procedure
The transfer function of a high-Q bandpass filter shown in Figure 64 is given by Equation 9.
R3 + R4
R1R4C1
S
VOUT
=
R3
1
V
S2 + S
+
IN
R1C1 R2R4R5C1C2
(9)
R3
2
wO
=
R2R4R5C1C2
(10)
wO
Q
1
=
R1C1
(11)
(12)
wO
fO
=
; 1MHz
2p
Use Equation 11 and Equation 12, along with the filter specifications in table to find the relationship between ω0,
Q, R1, and C1. Set C1 = 1000 pF, which results in R1= 1.5915 kΩ. The closest E96 standard value resistor value
is 1.58 kΩ.
Notice that the DC load driven by the OPA820 driving the output VOUT = R3 + R4. Select the total load to be 1 kΩ
and R3 = R4, which results in a value of 500 Ω.
To simplify the filter design, set C1 = C2 = 1000 pF.
Copyright © 2004–2016, Texas Instruments Incorporated
Submit Documentation Feedback
29
Product Folder Links: OPA820
OPA820
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
www.ti.com
Plugging the values of R3, R4, C1, and C2 into Equation 10 and assuming R2 = R5 results in a value of 159.15 Ω.
The closest standard E96 value is 158 Ω.
See Figure 61 for the frequency response of the filter shown in Figure 60.
10.2.1.2.2 Low-Pass Butterworth Filter Design Procedure
The transfer function of a low-pass Butterworth filter shown in Figure 59 is given by Equation 13.
VOUT
K
=
s2 R R C C + s R C + R C + R C 1-K +1
V
(
)
(
)
)
IN
(
1
2
1
2
1
1
2
1
1 2
where
æ
ö
÷
ø
RF
K = 1+
ç
RG
è
•
is the low-frequency DC gain
(13)
The values for RF and RG are the standard recommended values in the data sheet.
The cutoff frequency is in Equation 14.
1
w0 =
R C R C
2
(
)
1
1
2
(14)
(15)
The Q of the filter is given by Equation 15.
R1R2C1C2
Q =
R1C1 + R2C1 + R1C2 1-K
(
)
From Table 1, Q = 0.707 and ω0 = 2π × 5 MHz. To aid in solving this circuit, assume C1 = 100 pF and
R1 = 124 Ω. Plugging these values into Equation 14 and Equation 15 and finding the closest standard value
components results in R2 = 517 Ω and C2 = 160 pF. See Figure 62 for the frequency response of the filter shown
in Figure 59.
10.2.1.3 Application Curves
9
6
6
0
−
6
12
18
24
30
36
42
48
54
60
66
72
−
−
−
−
−
−
−
−
−
−
−
3
0
-3
-6
-9
100k
1M
10M
Frequency (Hz)
100M
10k
100k
1M
10M
Frequency (Hz)
D001
Figure 61. High-Q 1-MHz Bandpass Filter Frequency
Response
Figure 62. Frequency Response
10.2.2 Buffering High-Performance ADCs
To achieve full performance from a high-dynamic range ADC, take considerable care in the design of the input-
amplifier interface circuit. The example circuit in Figure 63 shows a typical AC-coupled interface to a very-high
dynamic-range converter. This AC-coupled example allows the OPA820 device to operate using a signal range
that swings symmetrically around ground (0 V). The 2-VPP swing is then level-shifted through the blocking
capacitor to a midscale reference level, which is created by a well-decoupled resistive divider off the internal
reference voltages of the converter. To have a negligible effect (1 dB) on the rated spurious-free dynamic range
30
Submit Documentation Feedback
Copyright © 2004–2016, Texas Instruments Incorporated
Product Folder Links: OPA820
OPA820
www.ti.com
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
(SFDR) of the converter, the SFDR of the amplifier must be at least 18 dB greater than the converter. The
OPA820 device has a minimal effect on the rated distortion of the ADS850 device, given the 79-dB SFDR at 2
VPP, 1 MHz of the ADS850 device. The greater than 90-dB (<1 MHz) SFDR for the OPA820 device in this
configuration implies a less than 3-dB degradation (for the system) from the specification of the converter. For
additional SFDR improvement with the OPA820 device, use a differential configuration.
Successful application of the OPA820 device for ADC driving requires careful selection of the series resistor at
the amplifier output and the additional shunt capacitor at the ADC input. To some extent, the selection of this RC
network is determined empirically for each converter. Many high-performance CMOS ADCs, such as the
ADS850, perform better with the shunt capacitor at the input pin. This capacitor provides low source impedance
for the transient currents produced by the sampling process. Improved SFDR is often obtained by adding this
external capacitor, whose value is often recommended in the data sheet of the converter. The external capacitor,
in combination with the built-in capacitance of the ADC input, presents a significant capacitive load to the
OPA820 device. Without a series isolation resistor, an undesirable peaking or loss of stability in the amplifier can
occur.
Because the DC bias current of the CMOS ADC input is negligible, the resistor has no effect on overall gain or
offset accuracy. See Figure 15 (±5 V) and Figure 42 (5 V) to obtain a good starting value for the series resistor
which ensures a flat-frequency response to the ADC input. Increasing the external capacitor value allows the
series resistor to be reduced. Intentionally band limiting using this RC network can also be used to limit noise at
the converter input.
5 V
5 V
REFT
(3 V)
R
2 kꢀ
2 kꢀ
S
0.1 mF
V
IN
24.9 ꢀ
+
OPA820
IN
IN
50 ꢀ
100 pF
ADS850
14-Bit
10 MSPS
œ5 V
2 kꢀ
402 ꢀ
402 ꢀ
0.1 µF
(2 V) (1 V)
REFB VREF
2 kꢀ
SEL
Copyright © 2016, Texas Instruments Incorporated
Figure 63. High Dynamic-Range Converter
10.2.3 Video Line Driving
Most video distribution systems are designed with 75-Ω series resistors to drive a matched 75-Ω cable. To
deliver a net gain of 1 to the 75-Ω matched load, the amplifier is typically set up for a voltage gain of 2,
compensating for the 6-dB attenuation of the voltage divider formed by the series and shunt 75-Ω resistors at
either end of the cable.
The circuit of Figure 55 applies to this requirement if all references to 50-Ω resistors are replaced by 75-Ω
values. Often, the amplifier gain is further increased to 2.2, which recovers the additional DC loss of a typical
long cable run. This change would require the gain resistor (RG) in Figure 55 to be reduced from 402 Ω to 335 Ω.
In either case, both the gain flatness and the differential gain and phase performance of the OPA820 device
provides exceptional results in video-distribution applications. Differential gain and phase measure the change in
overall small-signal gain and phase for the color sub-carrier frequency (3.58 MHz in NTSC systems) versus
changes in the large-signal output level (which represents luminance information in a composite video signal).
The OPA820 device, with the typical 150-Ω load of a single-matched video cable, shows less than 0.01%
differential gain and 0.01° phase errors over the standard luminance range for a positive video (negative sync)
signal. Similar performance can be observed for multiple video signals, as shown in Figure 64.
Copyright © 2004–2016, Texas Instruments Incorporated
Submit Documentation Feedback
31
Product Folder Links: OPA820
OPA820
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
www.ti.com
335 ꢀ
402 ꢀ
75-ꢀ Transmission Line
75 ꢀ
OPA820
+
V
OUT
Video
Input
75 ꢀ
75 ꢀ
75 ꢀ
V
V
OUT
75 ꢀ
75 ꢀ
OUT
75 ꢀ
Copyright © 2016, Texas Instruments Incorporated
High output current drive capability allows three back-terminated 75-Ω transmission lines to be simultaneously driven.
Figure 64. Video Distribution Amplifier
10.2.4 Single Differential Op Amp
The voltage-feedback architecture of the OPA820 device, with the high common-mode rejection ratio (CMRR),
provides exceptional performance in differential amplifier configurations. Figure 65 shows a typical configuration.
The starting point for this design is the selection of the RF value from 200 Ω to 2 kΩ. Lower values reduce the
required RG, increasing the load on the V2 source and on the OPA820 output. Higher values increase output
noise as well as the effects of parasitic board and device capacitances. Following the selection of RF, RG must
be set to achieve the desired inverting gain for V2. Remember that the bandwidth is set approximately by the
gain bandwidth product (GBP) divided by the noise gain (1 + RF / RG). For accurate differential operation (that is,
good CMRR), the ratio R2 / R1 must be set equal to RF / RG.
Usually, setting the absolute values of R2 and R1 equal to RF and RG (respectively) is best. This setting equalizes
the divider resistances and cancels the effect of input bias currents. However, scaling the values of R2 and R1 to
adjust the loading on the driving source, V1, can be useful. In most cases, the achievable low-frequency CMRR
is limited by the accuracy of the resistor values. The 85-dB CMRR of the OPA820 device does not determine the
overall circuit CMRR unless the resistor ratios are matched to better than 0.003%. If trimming the CMRR is
required, R2 is the suggested adjustment point.
5 V
Power-supply decoupling not shown.
R
1
V
V
50 ꢀ
1
2
+
RF
OPA820
VO
=
(V1 - V2)
RG
R
2
R2 RF
when
=
R1 RG
R
R
F
G
œ5 V
Copyright © 2016, Texas Instruments Incorporated
Figure 65. High-Speed, Single Differential Amplifier
32
Submit Documentation Feedback
Copyright © 2004–2016, Texas Instruments Incorporated
Product Folder Links: OPA820
OPA820
www.ti.com
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
10.2.5 Triple Differencing Op Amp (Instrumentation Topology)
The primary drawback of the single differential amplifier is the relatively low input impedances of the topology.
Where high impedance is required at the differential input, a standard instrumentation amplifier (INA) topology
can be built using the OPA820 device as the differencing stage. Figure 66 shows an example of this, in which
the two input amplifiers are packaged together as a dual voltage-feedback op amp, the OPA2822 device.
This approach saves board space, cost, and power compared to using two additional OPA820 devices, and still
achieves very good noise and distortion performance as a result of the moderate loading on the input amplifiers.
In this circuit, the common-mode gain to the output is always 1, because of the four matched 500-Ω resistors,
whereas the differential gain is set by Equation 16 which is equal to 2 using the values in Figure 66.
1 + 2RF1 / RG
(16)
The differential to single-ended conversion is still performed by the OPA820 output stage. The high-impedance
inputs allow the V1 and V2 sources to be terminated or impedance-matched as required. If the V1 and V2 inputs
are already truly differential, such as the output from a signal transformer, then a single-matching termination
resistor can be used between them. Remember, however, that a defined DC signal path must always exist for
the V1 and V2 inputs; for the transformer case, a center-tapped secondary connected to ground would provide an
optimum DC operating point.
5 V
V
+
1
OPA2822
5 V
R
F1
500 ꢀ
500 ꢀ
+
V
OPA820
O
500 ꢀ
R
G
500 ꢀ
R
F1
500 ꢀ
œ5 V
500 ꢀ
500 ꢀ
OPA2822
V
+
2
œ5 V
Copyright © 2016, Texas Instruments Incorporated
Figure 66. Wideband 3-Differencing Amplifier
10.2.6 DAC Transimpedance Amplifier
High-frequency digital-to-analog converters (DACs) require a low-distortion output amplifier to retain the SFDR
performance into practical loads. Figure 67 shows a single-ended output-drive implementation. In this circuit, only
one side of the complementary output drive signal is used. Figure 67 shows the signal output current connected
into the virtual ground-summing junction of the OPA820 device, which is set up as a transimpedance stage or I-V
converter. The unused current output of the DAC is connected to ground. If the DAC requires the outputs to be
terminated to a compliance voltage other than ground for operation, then the appropriate voltage level can be
applied to the noninverting input of the OPA820 device.
The DC gain for this circuit is equal to RF. At high frequencies, the DAC output capacitance (CD) produces a zero
in the noise gain for the OPA820 device that can cause peaking in the closed-loop frequency response. CF is
added across RF to compensate for this noise-gain peaking. To achieve a flat transimpedance-frequency
response, this pole in the feedback network must be set to the value shown in Equation 17 which gives a corner
frequency f–3 dB of approximately as shown in Equation 18.
Copyright © 2004–2016, Texas Instruments Incorporated
Submit Documentation Feedback
33
Product Folder Links: OPA820
OPA820
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
www.ti.com
1
GBP
=
2pRFCF
4pRFCD
(17)
(18)
GBP
2pRFCD
f-3 dB
=
+
OPA820
V = I R
O D F
High-Speed
DAC
500 ꢀ
C
F
I
C
D
D
GBP : Gain Bandwidth
Product (Hz) for the OPA820.
I
D
Copyright © 2016, Texas Instruments Incorporated
Figure 67. Wideband, Low-Distortion DAC Transimpedance Amplifier
11 Power Supply Recommendations
High-speed amplifiers require low-inductance power supply traces and low-ESR bypass capacitors. When
possible both power and ground planes must be used in the printed-circuit board design and the power plane
must be adjacent to the ground plane in the board stack-up. The power supply voltage must be centered on the
desired amplifier output voltage; so for ground referenced output signals, split supplies are required. The power
supply voltage must be from 5 V to 12 V.
34
Submit Documentation Feedback
Copyright © 2004–2016, Texas Instruments Incorporated
Product Folder Links: OPA820
OPA820
www.ti.com
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
12 Layout
12.1 Layout Guidelines
Achieving optimum performance with a high-frequency amplifier such as the OPA820 device requires careful
attention to board layout parasitics and external component types. This section lists recommendations to
optimize performance.
12.1.1 Minimizing Parasitic Capacitance
Minimize parasitic capacitance to any AC ground for all of the signal I/O pins. Parasitic capacitance on the output
and inverting input pins can cause instability. On the noninverting input, parasitic capacitance can react with the
source impedance to cause unintentional band limiting. To reduce unwanted capacitance, a window around the
signal I/O pins must be opened in all of the ground and power planes around those pins. Otherwise, ground and
power planes must be unbroken elsewhere on the board.
12.1.2 Minimizing Distance from Power Supply to Decoupling Capacitors
Minimize the distance, less than 0.25 inches, from the power-supply pins to high-frequency 0.1-µF decoupling
capacitors. At the device pins, the ground and power-plane layout must not be in close proximity to the signal I/O
pins. Avoid narrow power and ground traces to minimize inductance between the pins and the decoupling
capacitors. The power-supply connections must always be decoupled with these capacitors. Larger (2.2-µF to
6.8-µF) decoupling capacitors, effective at lower frequency, must also be used on the main supply pins. Place
these capacitors somewhat farther from the device. These capacitors can be shared among several devices in
the same area of the PCB.
12.1.3 Selecting and Placing External Components
Careful selection and placement of external components preserves the high-frequency performance of the
OPA820 device. Resistors must be a very-low reactance type. Surface-mount resistors work best and allow a
tighter overall layout. Metal-film and carbon composition, axially leaded resistors can also provide good high-
frequency performance. Again, keep the leads and PCB trace length as short as possible. Never use wire-wound
type resistors in a high-frequency application. Because the output pin and inverting input pin are the most
sensitive to parasitic capacitance, always position the feedback and series output resistor, if any, as close as
possible to the output pin. Other network components, such as noninverting input termination resistors, must also
be placed close to the package. Where double-side component mounting is allowed, place the feedback resistor
directly under the package on the other side of the board between the output and inverting input pins. Even with
a low parasitic capacitance shunting the external resistors, excessively high resistor values can create significant
time constants that can degrade performance. Good axial metal-film or surface-mount resistors have
approximately 0.2 pF in shunt with the resistor. For resistor values greater than 1.5 kΩ, this parasitic capacitance
can add a pole, a zero, or both below 500 MHz that can effect circuit operation. Keep resistor values as low as
possible consistent with load-driving considerations. A good starting point for design is to set RG || RF = 200 Ω.
Using this setting automatically keeps the resistor noise terms low, and minimizes the effect of the parasitic
capacitance.
12.1.4 Connecting Other Wideband Devices
Connections to other wideband devices on the board can be made with short, direct traces or through onboard
transmission lines. For short connections, consider the trace and the input to the next device as a lumped
capacitive load. Relatively wide traces (50 mils to 100 mils) must be used, preferably with ground and power
planes opened up around them. Estimate the total capacitive load and set RS from the plot of Figure 15 (±5 V)
and Figure 42 (5 V). Low parasitic capacitive loads (<5 pF) may not require an RS because the OPA820 device
is nominally compensated to operate with a 2-pF parasitic load. Higher parasitic capacitive loads without an RS
are allowed as the signal gain increases (increasing the unloaded phase margin). If a long trace is required, and
the 6-dB signal loss intrinsic to a doubly-terminated transmission line is acceptable, implement a matched
impedance transmission line using microstrip or stripline techniques (consult an ECL design handbook for
microstrip and stripline layout techniques). A 50-Ω environment is normally not necessary onboard, and in fact, a
higher impedance environment improves distortion as shown in Figure 7 and Figure 36. With a characteristic
board-trace impedance defined based on board material and trace dimensions, a matching series resistor into
the trace from the output of the OPA820 device is used as well as a terminating shunt resistor at the input of the
destination device. Remember also that the terminating impedance is the parallel combination of the shunt
Copyright © 2004–2016, Texas Instruments Incorporated
Submit Documentation Feedback
35
Product Folder Links: OPA820
OPA820
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
www.ti.com
Layout Guidelines (continued)
resistor and input impedance of the destination device; this total effective impedance must be set to match the
trace impedance. If the 6-dB attenuation of a doubly-terminated transmission line is unacceptable, a long trace
can be series-terminated at the source end only. Treat the trace as a capacitive load in this case and set the
series resistor value as shown in the plot of Figure 15 (±5 V) and Figure 42 (5 V) which does not preserve signal
integrity as well as a doubly-terminated line. If the input impedance of the destination device is low, some signal
attenuation occurs because of the voltage divider formed by the series output into the terminating impedance.
12.1.5 Socketing
TI does not recommend socketing a high-speed part like the OPA820 device. The additional lead length and pin-
to-pin capacitance introduced by the socket can create an extremely troublesome parasitic network, which can
make achieving a smooth, stable frequency response almost impossible. The best results are obtained by
soldering the OPA820 device onto the board.
12.2 Layout Example
Ground and power plane exist on
inner layers
Ground and power plane removed
Place output resistors close
to output pins to minimize
parasitic capacitance
from inner layers
1
2
3
6
5
4
Place bypass capacitors
close to power pins
Place bypass capacitors
close to power pins
Place input resistor close to pin 4 to
minimize stray capacitance
Place feedback resistor on the bottom
of PCB between pins 4 and 6
Remove GND and Power plane
under pins 1 and 4 to minimize
stray PCB capacitance
Figure 68. OPA820 Layout Example
36
Submit Documentation Feedback
Copyright © 2004–2016, Texas Instruments Incorporated
Product Folder Links: OPA820
OPA820
www.ti.com
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
13 Device and Documentation Support
13.1 Device Support
13.1.1 Design-In Tools
13.1.1.1 Demonstration Fixtures
Two printed-circuit boards (PCBs) are available to assist in the initial evaluation of circuit performance using the
OPA820 device in the two package options. Both of these boards are offered free of charge as unpopulated
PCBs, delivered with a user’s guide. Table 3 lists a summary information for these fixtures.
Table 3. Demonstration Fixtures
DEVICE NUMBER
OPA820
OPA820
PACKAGE
ORDERING NUMBER
DEM-OPA-SO-1A
USER'S GUIDE
SOIC (8)
SOT-23 (5)
DEM-OPA-SO-1A Demonstration Fixture
DEM-OPA-SOT-1A Demonstration Fixture
DEM-OPA-SOT-1A
The demonstration fixtures can be requested through the OPA820 product folder.
13.1.1.2 Macromodels and Applications Support
Computer simulation of circuit performance using SPICE is often a quick way to analyze the performance of the
OPA820 device and the device circuit designs. This is particularly true for video and RF amplifier circuits where
parasitic capacitance and inductance can play a major role on circuit performance. A SPICE model for the
OPA820 device is available through www.ti.com. The applications department is also available for design
assistance. These models predict typical small-signal AC, transient steps, DC performance, and noise under a
wide variety of operating conditions. The models include the noise terms found in the electrical specifications of
the data sheet. These models do not attempt to distinguish between the package types in their small-signal AC
performance.
13.1.2 Development Support
For the OPA820 PSpice Model, see SBOC048.
For the OPA820 TINA-TI Reference Design, see SBOC094.
For the OPA820 TINA-TI Spice Model, see SBOM176.
13.2 Documentation Support
13.2.1 Related Documentation
For related documentation, see the following:
•
•
•
•
•
•
•
•
ADS850 14-Bit, 10MSPS Self-Calibrating Analog-to-Digital Converter (SBAS154)
DEM-OPA-SO-1A Demonstration Fixture (SBOU009)
DEM-OPA-SOT-1A Demonstration Fixture (SBOU010)
Measuring Board Parasitics in High-Speed Analog Design (SBOA094)
Noise Analysis for High-Speed Op Amps (SBOA066)
OPA2822 Dual, Wideband, Low-Noise Operational Amplifier (SBOS188)
RLC Filter Design for ADC Interface Applications (SBAA108)
Wideband Complementary Current Output DAC to Single-Ended Interface: Improved Matching for the Gain
and Compliance Voltage Swing (SBAA135)
13.3 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper
right corner, click on Alert me to register and receive a weekly digest of any product information that has
changed. For change details, review the revision history included in any revised document.
Copyright © 2004–2016, Texas Instruments Incorporated
Submit Documentation Feedback
37
Product Folder Links: OPA820
OPA820
SBOS303D –JUNE 2004–REVISED DECEMBER 2016
www.ti.com
13.4 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
13.5 Trademarks
E2E is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
13.6 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
13.7 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
14 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
38
Submit Documentation Feedback
Copyright © 2004–2016, Texas Instruments Incorporated
Product Folder Links: OPA820
PACKAGE OPTION ADDENDUM
www.ti.com
29-Jun-2023
PACKAGING INFORMATION
Orderable Device
Status Package Type Package Pins Package
Eco Plan
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
Samples
Drawing
Qty
(1)
(2)
(3)
(4/5)
(6)
OPA820ID
ACTIVE
SOIC
D
8
5
75
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
Level-2-260C-1 YEAR
-40 to 85
-40 to 85
OPA
820
Samples
Samples
OPA820IDBVR
ACTIVE
SOT-23
DBV
3000 RoHS & Green
3000 RoHS & Green
NIPDAU
NSO
OPA820IDBVRG4
OPA820IDBVT
LIFEBUY
ACTIVE
SOT-23
SOT-23
DBV
DBV
5
5
NIPDAU
NIPDAU
Level-2-260C-1 YEAR
Level-2-260C-1 YEAR
-40 to 85
-40 to 85
NSO
NSO
250
RoHS & Green
Samples
Samples
OPA820IDBVTG4
OPA820IDR
LIFEBUY
ACTIVE
SOT-23
SOIC
DBV
D
5
8
250
RoHS & Green
NIPDAU
NIPDAU
Level-2-260C-1 YEAR
Level-2-260C-1 YEAR
-40 to 85
-40 to 85
NSO
2500 RoHS & Green
OPA
820
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based
flame retardants must also meet the <=1000ppm threshold requirement.
(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6)
Lead finish/Ball material - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead finish/Ball material values may wrap to two
lines if the finish value exceeds the maximum column width.
Addendum-Page 1
PACKAGE OPTION ADDENDUM
www.ti.com
29-Jun-2023
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
3-Jun-2022
TAPE AND REEL INFORMATION
REEL DIMENSIONS
TAPE DIMENSIONS
K0
P1
W
B0
Reel
Diameter
Cavity
A0
A0 Dimension designed to accommodate the component width
B0 Dimension designed to accommodate the component length
K0 Dimension designed to accommodate the component thickness
Overall width of the carrier tape
W
P1 Pitch between successive cavity centers
Reel Width (W1)
QUADRANT ASSIGNMENTS FOR PIN 1 ORIENTATION IN TAPE
Sprocket Holes
Q1 Q2
Q3 Q4
Q1 Q2
Q3 Q4
User Direction of Feed
Pocket Quadrants
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
B0
K0
P1
W
Pin1
Diameter Width (mm) (mm) (mm) (mm) (mm) Quadrant
(mm) W1 (mm)
OPA820IDBVR
OPA820IDBVT
OPA820IDR
SOT-23
SOT-23
SOIC
DBV
DBV
D
5
5
8
3000
250
180.0
180.0
330.0
8.4
8.4
3.15
3.15
6.4
3.1
3.1
5.2
1.55
1.55
2.1
4.0
4.0
8.0
8.0
8.0
Q3
Q3
Q1
2500
12.4
12.0
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
3-Jun-2022
TAPE AND REEL BOX DIMENSIONS
Width (mm)
H
W
L
*All dimensions are nominal
Device
Package Type Package Drawing Pins
SPQ
Length (mm) Width (mm) Height (mm)
OPA820IDBVR
OPA820IDBVT
OPA820IDR
SOT-23
SOT-23
SOIC
DBV
DBV
D
5
5
8
3000
250
210.0
210.0
356.0
185.0
185.0
356.0
35.0
35.0
35.0
2500
Pack Materials-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
3-Jun-2022
TUBE
T - Tube
height
L - Tube length
W - Tube
width
B - Alignment groove width
*All dimensions are nominal
Device
Package Name Package Type
SOIC
Pins
SPQ
L (mm)
W (mm)
T (µm)
B (mm)
OPA820ID
D
8
75
506.6
8
3940
4.32
Pack Materials-Page 3
PACKAGE OUTLINE
DBV0005A
SOT-23 - 1.45 mm max height
S
C
A
L
E
4
.
0
0
0
SMALL OUTLINE TRANSISTOR
C
3.0
2.6
0.1 C
1.75
1.45
1.45
0.90
B
A
PIN 1
INDEX AREA
1
2
5
(0.1)
2X 0.95
1.9
3.05
2.75
1.9
(0.15)
4
3
0.5
5X
0.3
0.15
0.00
(1.1)
TYP
0.2
C A B
NOTE 5
0.25
GAGE PLANE
0.22
0.08
TYP
8
0
TYP
0.6
0.3
TYP
SEATING PLANE
4214839/G 03/2023
NOTES:
1. All linear dimensions are in millimeters. Any dimensions in parenthesis are for reference only. Dimensioning and tolerancing
per ASME Y14.5M.
2. This drawing is subject to change without notice.
3. Refernce JEDEC MO-178.
4. Body dimensions do not include mold flash, protrusions, or gate burrs. Mold flash, protrusions, or gate burrs shall not
exceed 0.25 mm per side.
5. Support pin may differ or may not be present.
www.ti.com
EXAMPLE BOARD LAYOUT
DBV0005A
SOT-23 - 1.45 mm max height
SMALL OUTLINE TRANSISTOR
PKG
5X (1.1)
1
5
5X (0.6)
SYMM
(1.9)
2
3
2X (0.95)
4
(R0.05) TYP
(2.6)
LAND PATTERN EXAMPLE
EXPOSED METAL SHOWN
SCALE:15X
SOLDER MASK
OPENING
SOLDER MASK
OPENING
METAL UNDER
SOLDER MASK
METAL
EXPOSED METAL
EXPOSED METAL
0.07 MIN
ARROUND
0.07 MAX
ARROUND
NON SOLDER MASK
DEFINED
SOLDER MASK
DEFINED
(PREFERRED)
SOLDER MASK DETAILS
4214839/G 03/2023
NOTES: (continued)
6. Publication IPC-7351 may have alternate designs.
7. Solder mask tolerances between and around signal pads can vary based on board fabrication site.
www.ti.com
EXAMPLE STENCIL DESIGN
DBV0005A
SOT-23 - 1.45 mm max height
SMALL OUTLINE TRANSISTOR
PKG
5X (1.1)
1
5
5X (0.6)
SYMM
(1.9)
2
3
2X(0.95)
4
(R0.05) TYP
(2.6)
SOLDER PASTE EXAMPLE
BASED ON 0.125 mm THICK STENCIL
SCALE:15X
4214839/G 03/2023
NOTES: (continued)
8. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate
design recommendations.
9. Board assembly site may have different recommendations for stencil design.
www.ti.com
PACKAGE OUTLINE
D0008A
SOIC - 1.75 mm max height
SCALE 2.800
SMALL OUTLINE INTEGRATED CIRCUIT
C
SEATING PLANE
.228-.244 TYP
[5.80-6.19]
.004 [0.1] C
A
PIN 1 ID AREA
6X .050
[1.27]
8
1
2X
.189-.197
[4.81-5.00]
NOTE 3
.150
[3.81]
4X (0 -15 )
4
5
8X .012-.020
[0.31-0.51]
B
.150-.157
[3.81-3.98]
NOTE 4
.069 MAX
[1.75]
.010 [0.25]
C A B
.005-.010 TYP
[0.13-0.25]
4X (0 -15 )
SEE DETAIL A
.010
[0.25]
.004-.010
[0.11-0.25]
0 - 8
.016-.050
[0.41-1.27]
DETAIL A
TYPICAL
(.041)
[1.04]
4214825/C 02/2019
NOTES:
1. Linear dimensions are in inches [millimeters]. Dimensions in parenthesis are for reference only. Controlling dimensions are in inches.
Dimensioning and tolerancing per ASME Y14.5M.
2. This drawing is subject to change without notice.
3. This dimension does not include mold flash, protrusions, or gate burrs. Mold flash, protrusions, or gate burrs shall not
exceed .006 [0.15] per side.
4. This dimension does not include interlead flash.
5. Reference JEDEC registration MS-012, variation AA.
www.ti.com
EXAMPLE BOARD LAYOUT
D0008A
SOIC - 1.75 mm max height
SMALL OUTLINE INTEGRATED CIRCUIT
8X (.061 )
[1.55]
SYMM
SEE
DETAILS
1
8
8X (.024)
[0.6]
SYMM
(R.002 ) TYP
[0.05]
5
4
6X (.050 )
[1.27]
(.213)
[5.4]
LAND PATTERN EXAMPLE
EXPOSED METAL SHOWN
SCALE:8X
SOLDER MASK
OPENING
SOLDER MASK
OPENING
METAL UNDER
SOLDER MASK
METAL
EXPOSED
METAL
EXPOSED
METAL
.0028 MAX
[0.07]
.0028 MIN
[0.07]
ALL AROUND
ALL AROUND
SOLDER MASK
DEFINED
NON SOLDER MASK
DEFINED
SOLDER MASK DETAILS
4214825/C 02/2019
NOTES: (continued)
6. Publication IPC-7351 may have alternate designs.
7. Solder mask tolerances between and around signal pads can vary based on board fabrication site.
www.ti.com
EXAMPLE STENCIL DESIGN
D0008A
SOIC - 1.75 mm max height
SMALL OUTLINE INTEGRATED CIRCUIT
8X (.061 )
[1.55]
SYMM
1
8
8X (.024)
[0.6]
SYMM
(R.002 ) TYP
[0.05]
5
4
6X (.050 )
[1.27]
(.213)
[5.4]
SOLDER PASTE EXAMPLE
BASED ON .005 INCH [0.125 MM] THICK STENCIL
SCALE:8X
4214825/C 02/2019
NOTES: (continued)
8. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate
design recommendations.
9. Board assembly site may have different recommendations for stencil design.
www.ti.com
IMPORTANT NOTICE AND DISCLAIMER
TI PROVIDES TECHNICAL AND RELIABILITY DATA (INCLUDING DATA SHEETS), DESIGN RESOURCES (INCLUDING REFERENCE
DESIGNS), APPLICATION OR OTHER DESIGN ADVICE, WEB TOOLS, SAFETY INFORMATION, AND OTHER RESOURCES “AS IS”
AND WITH ALL FAULTS, AND DISCLAIMS ALL WARRANTIES, EXPRESS AND IMPLIED, INCLUDING WITHOUT LIMITATION ANY
IMPLIED WARRANTIES OF MERCHANTABILITY, FITNESS FOR A PARTICULAR PURPOSE OR NON-INFRINGEMENT OF THIRD
PARTY INTELLECTUAL PROPERTY RIGHTS.
These resources are intended for skilled developers designing with TI products. You are solely responsible for (1) selecting the appropriate
TI products for your application, (2) designing, validating and testing your application, and (3) ensuring your application meets applicable
standards, and any other safety, security, regulatory or other requirements.
These resources are subject to change without notice. TI grants you permission to use these resources only for development of an
application that uses the TI products described in the resource. Other reproduction and display of these resources is prohibited. No license
is granted to any other TI intellectual property right or to any third party intellectual property right. TI disclaims responsibility for, and you
will fully indemnify TI and its representatives against, any claims, damages, costs, losses, and liabilities arising out of your use of these
resources.
TI’s products are provided subject to TI’s Terms of Sale or other applicable terms available either on ti.com or provided in conjunction with
such TI products. TI’s provision of these resources does not expand or otherwise alter TI’s applicable warranties or warranty disclaimers for
TI products.
TI objects to and rejects any additional or different terms you may have proposed. IMPORTANT NOTICE
Mailing Address: Texas Instruments, Post Office Box 655303, Dallas, Texas 75265
Copyright © 2023, Texas Instruments Incorporated
相关型号：
SI9130DB
5- and 3.3-V Step-Down Synchronous ConvertersWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9135LG-T1
SMBus Multi-Output Power-Supply ControllerWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9135LG-T1-E3
SMBus Multi-Output Power-Supply ControllerWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9135_11
SMBus Multi-Output Power-Supply ControllerWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9136_11
Multi-Output Power-Supply ControllerWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9130CG-T1-E3
Pin-Programmable Dual Controller - Portable PCsWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9130LG-T1-E3
Pin-Programmable Dual Controller - Portable PCsWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9130_11
Pin-Programmable Dual Controller - Portable PCsWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9137
Multi-Output, Sequence Selectable Power-Supply Controller for Mobile ApplicationsWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9137DB
Multi-Output, Sequence Selectable Power-Supply Controller for Mobile ApplicationsWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9137LG
Multi-Output, Sequence Selectable Power-Supply Controller for Mobile ApplicationsWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
SI9122E
500-kHz Half-Bridge DC/DC Controller with Integrated Secondary Synchronous Rectification DriversWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
-
VISHAY
©2020 ICPDF网 联系我们和版权申明