THS4130IDGK [TI]
HIGH SPEED LOW NOISE, FULLY DIFFERENTIAL I/O AMPLIFIERS; 高速低噪声，全差分I / O放大器
| 型号： | THS4130IDGK |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | HIGH SPEED LOW NOISE, FULLY DIFFERENTIAL I/O AMPLIFIERS |
| 文件： | 总27页 (文件大小：566K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
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SLOS318E − MAY 2000 − REVISED JANUARY 2004
features
key applications
D
High Performance
− 150 MHz −3 dB Bandwidth (V
− 51 V/µs Slew Rate
− −100 dB Third Harmonic Distortion at
250 kHz
D
Single-Ended To Differential Conversion
Differential ADC Driver
= 15 V)
CC
D
D
D
D
Differential Antialiasing
Differential Transmitter And Receiver
Output Level Shifter
D
D
Low Noise
− 1.3 nV/√Hz Input-Referred Noise
Differential-Input/Differential-Output
− Balanced Outputs Reject Common-Mode
Noise
THS4130
THS4131
D, DGN, OR DGK PACKAGE
(TOP VIEW)
D, DGN, OR DGK PACKAGE
(TOP VIEW)
− Reduced Second Harmonic Distortion
Due to Differential Output
V
V
V
V
IN+
1
2
3
4
8
7
6
5
IN−
IN+
IN−
1
2
3
4
8
7
6
5
V
PD
V
V
NC
V
OCM
OCM
D
D
Wide Power Supply Range
V
V
CC+
CC−
CC+
CC−
− V
= 5 V Single Supply to 15 V Dual
CC
V
V
V
V
OUT+
OUT− OUT+
OUT−
Supply
I
= 860 µA in Shutdown Mode
CC(SD)
(THS4130)
HIGH-SPEED DIFFERENTIAL I/O FAMILY
NUMBER OF
description
DEVICE
SHUTDOWN
CHANNELS
The THS413x is one in a family of fully-differential
input/differential output devices fabricated using
Texas Instruments’ state-of-the-art BiComI
complementary bipolar process.
THS4130
THS4131
1
1
X
−
The THS413x is made of a true fully-differential
signal path from input to output. This design leads
to an excellent common-mode noise rejection and
improved total harmonic distortion.
TOTAL HARMONIC DISTORTION
vs
FREQUENCY
−20
−30
−40
−50
−60
−70
−80
−90
−100
RELATED DEVICES
V
= 2 V
PP
OUT
DEVICE
THS412x
THS414x
THS415x
DESCRIPTION
100 MHz, 43 V/µs, 3.7 nV/√Hz
160 MHz, 450 V/µs, 6.5 nV/√Hz
180 MHz, 850 V/µs, 9 nV/√Hz
typical A/D application circuit
V
DD
V
CC
= 5 V to 5 V
5 V
AV
DD
DV
V
V
IN
DD
+
−
A
A
V
OCM
IN
V
CC
=
15 V
IN
DIGITAL
OUTPUT
+
AV
SS
−
ref
100k
1M
f − Frequency − Hz
10M
−5 V
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
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Copyright 2001 − 2004, Texas Instruments Incorporated
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1
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
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SLOS318E − MAY 2000 − REVISED JANUARY 2004
AVAILABLE OPTIONS
PACKAGED DEVICES
EVALUATION
MODULES
MSOP PowerPAD
(DGN) SYMBOL
MSOP
T
A
SMALL OUTLINE
(D)
(DGK)
SYMBOL
ATP
THS4130CD
THS4131CD
THS4130ID
THS4131ID
THS4130CDGN
THS4131CDGN
THS4130IDGN
THS4131IDGN
AOB
AOD
AOC
AOE
THS4130CDGK
THS4131CDGK
THS4130IDGK
THS4131IDGK
THS4130EVM
0°C to 70°C
ATQ
THS4131EVM
ASO
−
−
−40°C to 85°C
ASP
†
absolute maximum ratings over operating free-air temperature range (unless otherwise noted)
Supply voltage, V
to V
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33 V
CC−
CC+
Input voltage, V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
V
CC
I
Output current, I (see Note 1) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 150 mA
O
Differential input voltage, V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 V
ID
Continuous total power dissipation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . See Dissipation Rating Table
Maximum junction temperature, T (see Note 2) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 150°C
J
Maximum junction temperature, continuous operation, long term reliability, T (see Note 3) . . . . . . . . 125°C
J
Operating free-air temperature, T :C suffix . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0°C to 70°C
A
I suffix . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −40°C to 85°C
Storage temperature, T
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −65°C to 150°C
Lead temperature 1,6 mm (1/16 Inch) from case for 10 seconds . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 300°C
stg
ESD ratings:
HBM . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2500 V
CDM . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1500 V
MM . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 200 V
†
Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
NOTE 1: The THS413x may incorporate a PowerPad on the underside of the chip. This acts as a heatsink and must be connected to a thermally
dissipative plane for proper power dissipation. Failure to do so may result in exceeding the maximum junction temperature which could
permanently damage the device. See TI technical brief SLMA002 and SLMA004 for more information about utilizing the PowerPad
thermally enhanced package.
NOTE 2: The absolute maximum temperature under any condition is limited by the constraints of the silicon process.
NOTE 3: The maximum junction temperature for continuous operation is limited by package constraints. Operation above this temperature may
result in reduced reliability and/or lifetime of the device.
DISSIPATION RATING TABLE
§
‡
POWER RATING
= 25°C T = 85°C
A
θ
θ
JC
(°C/W)
JA
PACKAGE
(°C/W)
T
A
D
97.5
58.4
260
38.3
4.7
1.02 W
1.71 W
410 mW
685 mW
154 mW
DGN
DGK
54.2
385 mW
‡
§
This data was taken using the JEDEC standard High−K test PCB.
Power rating is determined with a junction temperature of 125°C. This is the point where distortion starts to
substantially increase. Thermal management of the final PCB should strive to keep the junction temperature at or
below 125°C for best performance and long term reliability.
recommended operating conditions
MIN
2.5
TYP
MAX
15
UNIT
Dual supply
Single supply
C suffix
Supply voltage, V
CC+
to V
CC−
V
5
0
30
70
85
Operating free-air temperature, T
°C
A
I suffix
−40
PowerPAD is a trademark of Texas Instruments.
2
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
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SLOS318E − MAY 2000 − REVISED JANUARY 2004
†
electrical characteristics, V
= 5 V, R = 800 Ω, T = 25°C (unless otherwise noted)
CC
L
A
dynamic performance
PARAMETER
TEST CONDITIONS
MIN
TYP
125
MAX
UNIT
V
CC
V
CC
V
CC
V
CC
V
CC
V
CC
= 5
=
Gain = 1, R = 390 Ω
f
Small signal bandwidth (−3 dB),
Single ended input, differential output, V = 63 mV
I
5
Gain = 1, R = 390 Ω
135
150
80
f
PP
=
15
Gain = 1, R = 390 Ω
f
BW
SR
MHz
= 5
=
Gain = 2, R = 750 Ω
f
Small signal bandwidth (−3 dB),
Single ended input, differential output, V = 63 mV
5
Gain = 2, R = 750 Ω
85
f
I
PP
=
15
Gain = 2, R = 750 Ω
90
f
Slew rate (see Note 2)
Settling time to 0.1%
Settling time to 0.01%
Gain = 1
52
V/µs
ns
78
t
s
Step voltage = 2 V, Gain = 1
213
ns
†
The full range temperature is 0°C to 70°C for the C suffix, and −40°C to 85°C for the I suffix.
NOTE 4: Slew rate is measured from an output level range of 25% to 75%.
distortion performance
PARAMETER
TEST CONDITIONS
f = 250 kHz
MIN
TYP
MAX
UNIT
−95
−81
−96
−80
−97
−80
−91
−75
−91
−75
97
V
V
V
V
V
= 5
=
CC
CC
CC
CC
CC
f = 1 MHz
Total harmonic distortion,
Differential input, differential output,
f = 250 kHz
5
f = 1 MHz
Gain = 1, R = 390 Ω, R = 800 Ω, V = 2 V
f
L
O
PP
f = 250 kHz
THD
=
15
dBc
f = 1 MHz
f = 250 kHz
=
5
f = 1 MHz
V
O
= 4 V
PP
f = 250 kHz
=
15
f = 1 MHz
V
CC
V
CC
V
CC
V
CC
V
CC
=
=
=
=
=
2.5
5
98
V
O
= 2 V
pp
pp
Spurious free dynamic range (SFDR),
Differential input, differential output,
15
5
99
dB
Gain = 1, R = 390 Ω,
R = 800 Ω, f = 250 kHz
L
f
93
V
V
= 4 V
O
15
95
= 4 V,
G = 1,
I(PP)
Third intermodulation distortion
Third order intercept
−53
41.5
dBc
dB
F1 = 3 MHz,
F2 = 3.5 MHz
V
= 4 V,
I(PP)
F1 = 3 MHz,
G = 1,
F2 = 3.5 MHz
†
The full range temperature is 0°C to 70°C for the C suffix, and −40°C to 85°C for the I suffix.
3
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
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SLOS318E − MAY 2000 − REVISED JANUARY 2004
†
electrical characteristics, V
noise performance
= 5 V, R = 800 Ω, T = 25°C (unless otherwise noted) (continued)
L A
CC
PARAMETER
TEST CONDITIONS
f = 10 kHz
f = 10 kHz
MIN
TYP
1.3
MAX
UNIT
V
Input voltage noise
Input current noise
nV/√Hz
pA/√Hz
n
I
n
1
†
The full range temperature is 0°C to 70°C for the C suffix, and −40°C to 85°C for the I suffix.
dc performance
PARAMETER
TEST CONDITIONS
= 25°C
MIN
TYP
MAX
UNIT
T
71
69
78
A
Open loop gain
dB
T
A
= full range
= 25°C
T
A
0.2
2
Input offset voltage
T
= full range
= 25°C
A
3
mV
A
V
(OS)
Common mode input offset voltage, referred to V
Input offset voltage drift
Input bias current
T
0.2
4.5
2
3.5
OCM
T
= full range
= full range
A
µV/°C
µA
A
I
I
T
6
IB
Input offset current
100
2
500
nA
T
A
= full range
OS
Offset drift
nA/°C
†
The full range temperature is 0°C to 70°C for the C suffix, and −40°C to 85°C for the I suffix.
input characteristics
PARAMETER
TEST CONDITIONS
= full range
MIN
80
−3.77 −4 to
TYP
MAX
UNIT
CMRR
Common-mode rejection ratio
T
A
95
dB
V
ICR
Common-mode input voltage range
V
to 4.3
4.5
34
4
R
C
Input resistance
Measured into each input terminal
Open loop
MΩ
pF
Ω
I
I
Input capacitance, closed loop
Output resistance
r
41
o
†
The full range temperature is 0°C to 70°C for the C suffix, and −40°C to 85°C for the I suffix.
output characteristics
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
T
= 25°C
1.2 to 3.8 0.9 to 4.1
1.3 to 3.7
A
V
V
= 5 V
CC
T
A
= full range
= 25°C
T
A
3.7
3.6
10.5
10.2
25
4
12.4
45
Output voltage swing
=
=
5 V
V
CC
T
A
= full range
= 25°C
T
A
V
V
15 V
CC
T
= full range
= 25°C
A
A
T
= 5 V,
= 7 Ω
CC
R
T
= full range
T = 25°C
A
20
L
A
30
55
V
R
=
5 V,
CC
I
O
Output current
mA
= 7 Ω
T
= full range
T = 25°C
A
28
L
A
60
85
V
R
=
15 V,
CC
= 7 Ω
T
A
= full range
50
L
†
The full range temperature is 0°C to 70°C for the C suffix, and −40°C to 85°C for the I suffix.
4
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
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SLOS318E − MAY 2000 − REVISED JANUARY 2004
†
electrical characteristics, V
CC
= 5 V, R = 800 Ω, T = 25°C (unless otherwise noted) (continued)
L A
power supply
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
33
UNIT
Single supply
Split supply
4
V
CC
Supply voltage range
V
2
16.5
15
T
= 25°C
12.3
A
V
V
=
=
5 V
CC
T
A
= full range
= 25°C
16
I
Quiescent current
mA
CC
15 V
T
A
14
CC
T
= 25°C
0.86
1.4
1.5
A
I
Quiescent current (shutdown) (THS4130 only)
Power supply rejection ratio (dc)
V
PD
= −5 V
mA
dB
CC(SD)
T
A
= full range
= 25°C
T
A
73
70
98
PSRR
T
A
= full range
†
The full range temperature is 0°C to 70°C for the C suffix, and −40°C to 85°C for the I suffix.
TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
Small signal frequency response
1, 2
3
Small signal frequency response (various supplies)
Small signal frequency response (various C )
4
F
Small signal frequency response (various C )
L
5
Large signal transient response (differential in/single out)
Large signal frequency response
6
7
CMMR
Common mode rejection ratio
vs Frequency
8
vs Free-air temperature
9
I
I
Supply current
CC
vs Free-air temperature (shutdown state)
vs Free-air temperature
10
Input bias current
11
IB
Settling time
12
PSRR
THD
Power supply rejection ratio
Large signal transient response
Total harmonic distortion
vs Frequency (differential out)
13
14
vs Frequency
15
vs Frequency
16, 17
18, 19
20, 21
22, 23
24
Second harmonic distortion
Third harmonic distortion
vs Output voltage
vs Frequency
vs Output voltage
vs Frequency
V
n
Voltage noise
I
n
Current noise
vs Frequency
25
V
Input offset voltage
Output voltage
Output impedance
vs Common-mode output voltage
vs Differential load resistance
vs Frequency
26
(OS)
O
V
27
z
28
o
5
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
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SLOS318E − MAY 2000 − REVISED JANUARY 2004
TYPICAL CHARACTERISTICS
SMALL SIGNAL FREQUENCY RESPONSE
SMALL SIGNAL FREQUENCY RESPONSE
25
20
3
2
R
V
= 800 Ω,
Gain = 1,
L
R = 620 Ω
f
Gain = 10_R = 4 kΩ
f
=
5 V,
R
V
= 800 Ω,
CC
CC
L
V = 63 mV
I
=
5 V,
PP
1
0
V = 63 mV
PP
I
Gain = 5_R = 2 kΩ
f
15
10
−1
R = 390 Ω
f
−2
−3
−4
Gain = 2_R = 750 Ω
f
5
Gain = 1_R = 390 Ω
f
0
−5
−6
−5
−7
−8
−10
100 k
1 M
10 M
100 M
1 G
100 k
1 M
10 M
100 M
1 G
f − Frequency − Hz
f − Frequency − Hz
Figure 1
Figure 2
SMALL SIGNAL FREQUENCY RESPONSE
(VARIOUS SUPPLIES)
SMALL SIGNAL FREQUENCY RESPONSE
(VARIOUS C )
F
2
1
3
V
CC
= 15
2
1
C
= 0 pF
F
0
0
−1
−2
−3
−1
V
CC
= 5
−2
−3
−4
C
= 1 pF
F
−4
−5
−5
−6
−7
−8
−6
−7
−8
Gain = 1,
= 800 Ω,
R
L
R = 390 Ω,
f
V = 63 mV
I
PP
−9
−10
100 k
100 k
1 M
10 M
f − Frequency − Hz
100 M
1 G
1 M
10 M
100 M
1 G
f − Frequency − Hz
Figure 3
Figure 4
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SLOS318E − MAY 2000 − REVISED JANUARY 2004
TYPICAL CHARACTERISTICS
SMALL SIGNAL FREQUENCY RESPONSE
LARGE SIGNAL TRANSIENT RESPONSE
(DIFFERENTIAL IN/SINGLE OUT)
(VARIOUS C )
L
5
4
3
1
0.5
0
Gain = 1,
C
= 10 pF
L
V
O+
R
= 800 Ω,
L
V
CC
= 5 V,
V = 63 mV
,
I
PP
2
1
R = 390 Ω
f
V
O−
−0
−0.5
0.5
0
C
= 0 pF
L
−1
−2
−3
−4
−5
−6
V (Diff)
I
−0.5
−1
−7
−8
100 k
1 M
10 M
100 M
1 G
0.2
0.3
0.4
0.5
0.6
0
0.1
f − Frequency − Hz
t − Time − µs
Figure 5
Figure 6
COMMON MODE REJECTION RATIO
vs
FREQUENCY
LARGE SIGNAL FREQUENCY RESPONSE
−50
5
R = 1 kΩ,
f
−55
−60
−65
−70
−75
−80
−85
−90
−95
−100
V
CC
= 5 V
V
CC
= 15 V
0
−5
−10
−15
−20
−25
V
=
5 V
CC
Gain = 1
R = 390 Ω,
f
L
F
V
= 5 V
CC
R
C
= 800 Ω,
= 0 pF,
V = 0.2 V
I RMS
100 k
1 M
10 M
100 M
1 G
100 k
1 M
10 M
100 M
f − Frequency − Hz
f − Frequency − Hz
Figure 7
Figure 8
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SLOS318E − MAY 2000 − REVISED JANUARY 2004
TYPICAL CHARACTERISTICS
SUPPLY CURRENT
vs
FREE-AIR TEMPERATURE
(SHUTDOWN STATE)
SUPPLY CURRENT
vs
FREE-AIR TEMPERATURE
15
14.5
14
940
920
900
880
860
840
820
800
V
= 15 V
CC
13.5
13
12.5
12
V
= 5 V
CC
11.5
11
10.5
10
−40
−20
0
20
40
60
80
100
−50
−25
0
25
50
75
100
T
A
− Free-Air Temperature − °C
T
A
− Free-Air Temperature (Shutdown State) − °C
Figure 9
Figure 10
INPUT BIAS CURRENT
vs
FREE-AIR TEMPERATURE
SETTLING TIME
2.4
2.35
2.3
2.04
2.02
2
I
IB+
R
C
= 510 Ω
= 1 pF,
= 5 V
F
F
2.25
2.2
1.98
1.96
1.94
V
V
CC
O
= 4 V
= 800 Ω
PP
R
L
2.15
2.1
I
IB−
1.92
1.9
2.05
−50
−25
0
25
50
75
100
0
25
50
75
100
125
150
T
A
− Free-Air Temperature − °C
t − Time − ns
Figure 11
Figure 12
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SLOS318E − MAY 2000 − REVISED JANUARY 2004
TYPICAL CHARACTERISTICS
POWER SUPPLY REJECTION RATIO
vs
FREQUENCY (DIFFERENTIAL OUT)
LARGE SIGNAL TRANSIENT RESPONSE
2.5
2
−40
−50
−60
−70
−80
−90
−100
V
+
Gain = 1,
O
R = 330 Ω,
f
L
R
= 400 Ω
1.5
1
G = 1,
R = 390 Ω,
f
L
F
L
R
C
C
V
V
T
= 800 Ω,
= 0 pF,
= 10 pF,
= 2 V,
15 V
= 25°C
5
V
= 5 V
CC
0
I_Peak
CC
A
−5
−1
−1.5
−2
−2.5
=
V
= −5 V
CC
V
−
O
10 k
100 k
1 M
10 M
100 M
0
40
80
120
160
200
f − Frequency (Differential Out) − Hz
t − Time − nS
Figure 13
Figure 14
TOTAL HARMONIC DISTORTION
vs
FREQUENCY
−20
−30
−40
−50
−60
−70
−80
−90
−100
V
OUT
= 2 V
PP
V
CC
= 5 V to 5 V
V
CC
=
15 V
100k
1M
10M
f − Frequency − Hz
Figure 15
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SLOS318E − MAY 2000 − REVISED JANUARY 2004
TYPICAL CHARACTERISTICS
SECOND HARMONIC DISTORTION
SECOND HARMONIC DISTORTION
vs
vs
FREQUENCY
FREQUENCY
−30
−40
−30
−40
V
R
= 2 V
PP,
Single Ended Input
Differential Output
Single Ended Input
Differential Output
O
L
f
V
R
= 4 V
PP,
O
L
f
= 800 Ω,
= 800 Ω,
R = 390 Ω,
G = 1
R = 390 Ω,
G = 1
−50
−50
V
CC
= 5 V
−60
−60
V
CC
= 5 V
−70
−70
−80
−80
V
CC
= 15 V
−90
−90
−100
−110
−100
−110
V
= 15V, 5V
CC
100 k
1 M
10 M
100 k
1 M
10 M
f − Frequency − Hz
f − Frequency − Hz
Figure 16
Figure 17
SECOND HARMONIC DISTORTION
SECOND HARMONIC DISTORTION
vs
vs
OUTPUT VOLTAGE
OUTPUT VOLTAGE
−88
−90
−92
−94
f = 250 KHz
V
CC
= 15 V
V
=
5 V
CC
CC
R
= 800 Ω,
L
f
R = 390 Ω,
G = 1
−92
−96
V
CC
= 5 V
−94
V
CC
= 5 V
−98
−96
V
=
15 V
−98
−100
−102
−104
−106
V
CC
= 5 V
−100
−102
−104
−106
f = 500 KHz
= 800 Ω,
R
L
Single Ended Input
Differential Output
Single Ended Input
Differential Output
R = 390 Ω,
f
G = 1
0
1
2
3
4
5
6
7
0
1
2
3
4
5
6
7
V
O
− Output Voltage − V
V
O
− Output Voltage − V
Figure 18
Figure 19
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SLOS318E − MAY 2000 − REVISED JANUARY 2004
TYPICAL CHARACTERISTICS
THIRD HARMONIC DISTORTION
THIRD HARMONIC DISTORTION
vs
vs
FREQUENCY
FREQUENCY
−30
−40
−30
−40
V
R
= 4 V
PP
V
R
= 2 V ,
PP
O
L
f
O
L
f
= 800 Ω,
= 800 Ω,
R = 390 Ω,
G = 1
R = 390 Ω,
Gain = 1
V
CC
= 5 V
−50
−50
−60
Single Ended Input
Differential Output
−60
V
CC
= 15 V
−70
−70
V
CC
= 15 V
−80
−80
V
CC
= 5 V
−90
−90
−100
−110
V
= 5 V
CC
−100
−110
Single Ended Input
Differential Output
100 k
1 M
10 M
100 k
1 M
10 M
f − Frequency − Hz
f − Frequency − Hz
Figure 20
Figure 21
THIRD HARMONIC DISTORTION
THIRD HARMONIC DISTORTION
vs
vs
OUTPUT VOLTAGE
OUTPUT VOLTAGE
−88
−90
−88
−90
f = 250 KHz
= 800 Ω,
V
CC
= 15 V
R
L
R = 390 Ω,
f
V
CC
=
5 V
−92
G = 1
−92
−94
−94
V
CC
= 5 V
−96
−96
V
CC
= 5 V
V
CC
= 5 V
−98
−98
V
CC
= 15 V
f = 500 KHz
= 800 Ω,
−100
−102
−104
−106
−100
−102
−104
−106
R
L
R = 390 Ω,
f
G = 1
Single Ended Input
Differential Output
Single Ended Input
Differential Output
0
1
2
3
4
5
6
7
0
1
2
3
4
5
6
7
V
O
− Output Voltage − V
V
O
− Output Voltage − V
Figure 22
Figure 23
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SLOS318E − MAY 2000 − REVISED JANUARY 2004
TYPICAL CHARACTERISTICS
VOLTAGE NOISE
vs
FREQUENCY
10
1
10
100
1 k
10 k
100 k
f − Frequency − Hz
Figure 24
CURRENT NOISE
vs
INPUT OFFSET VOLTAGE
vs
FREQUENCY
COMMON-MODE OUTPUT VOLTAGE
7E−12
6E−12
1000
800
600
400
200
0
R = 1 k,
f
R
G = 1
= 800 Ω,
L
V
CC
= 2.5 V
5E−12
4E−12
V
CC
= 5 V
3E−12
2E−12
V
CC
= 15 V
−200
1E−12
0
−400
−600
1
10
100
1 k
10 k
100 k
−12
−9
−6
−3
0
3
6
9
12
f − Frequency − Hz
V
OCM
− Common-Mode Output Voltage − V
Figure 25
Figure 26
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SLOS318E − MAY 2000 − REVISED JANUARY 2004
TYPICAL CHARACTERISTICS
OUTPUT VOLTAGE
vs
OUTPUT IMPEDANCE
vs
DIFFERENTIAL LOAD RESISTANCE
FREQUENCY
15
10
5
100
10
R = 1 k
G = 2
f
V
CC
= 5 V
V
=
=
15 V
5 V
CC
CC
V
OUT+
V
V
OUT+
0
V
OUT−
V
CC
=
5 V
−5
1
V
OUT−
−10
−15
V
CC
=
15 V
0.1
100
1000
10 k
100 k
100 k
1 M
10 M
100 M
1 G
R
− Differential Load Resistance − Ω
f − Frequency − Hz
Figure 28
L
Figure 27
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SLOS318E − MAY 2000 − REVISED JANUARY 2004
APPLICATION INFORMATION
resistor matching
Resistor matching is important in fully differential amplifiers. The balance of the output on the reference voltage
depends on matched ratios of the resistor. CMRR, PSRR, and cancellation of the second harmonic distortion
will diminish if resistor mismatch occurs. Therefore, it is recommended to use 1% tolerance resistors or better
to keep the performance optimized.
V
sets the dc level of the output signals. If no voltage is applied to the V
pin, it will be set to the midrail
OCM
OCM
voltage internally defined as:
ǒV Ǔ ǒ
Ǔ
CC–
) V
CC)
2
In the differential mode, the V
mode is the same as the input in the gain of 1. V
on the two outputs cancel each other. Therefore, the output in the differential
OCM
has a high bandwidth capability up to the typical operation
OCM
range of the amplifier. For the prevention of noise going through the device, use a 0.1 µF capacitor on the V
pin as a bypass capacitor. The following graph shows the simplified diagram of the THS413x.
OCM
V
CC+
Output Buffer
V
IN−
x1
V
V
OUT+
C
R
R
V
IN+
Vcm Error
Amplifier
+
_
C
x1
OUT−
Output Buffer
V
CC+
30 kΩ
V
CC−
30 kΩ
CC−
V
V
OCM
Figure 29. THS413x Simplified Diagram
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SLOS318E − MAY 2000 − REVISED JANUARY 2004
APPLICATION INFORMATION
data converters
Data converters are one of the most popular applications for the fully differential amplifiers. The following
schematic shows a typical configuration of a fully differential amplifier attached to a differential ADC.
V
DD
V
CC
5 V
AV
DD
IN1
DV
DD
V
IN
+
−
A
A
V
OCM
+
IN2
AV
0.1 µF
−
V
ref
SS
−5 V
V
CC
−
Figure 30. Fully Differential Amplifier Attached to a Differential ADC
Fully differential amplifiers can operate with a single supply. V defaults to the midrail voltage, V /2. The
OCM
CC
differential output may be fed into a data converter. This method eliminates the use of a transformer in the circuit.
If the ADC has a reference voltage output (V ), then it is recommended to connect it directly to the V of
ref
OCM
the amplifier using a bypass capacitor for stability. For proper operation, the input common-mode voltage to the
input terminal of the amplifier should not exceed the common-mode input voltage range.
V
DD
V
CC
5 V
AV
DD
IN1
DV
DD
V
IN
+
−
A
A
V
OCM
+
IN2
AV
−
0.1 µF
V
ref
SS
Figure 31. Fully Differential Amplifier Using a Single Supply
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SLOS318E − MAY 2000 − REVISED JANUARY 2004
APPLICATION INFORMATION
data converters (continued)
Some single supply applications may require the input voltage to exceed the common-mode input voltage
range. In such cases, the following circuit configuration is suggested to bring the common-mode input voltage
within the specifications of the amplifier.
V
DD
V
CC
R
f
V
CC
R
PU
5 V
R
g
V
V
OUT
AV
DD
DV
V
IN
DD
+
−
V
A
A
P
IN1
V
OCM
THS1206
+
IN2
AV
−
0.1 µF
V
SS
ref
R
OUT
g
R
PU
V
CC
R
f
Figure 32. Circuit With Improved Common-Mode Input Voltage
The following equation is used to calculate R
:
PU
V
– V
P
ǒVIN PǓ
ǒCC
PǓ
R
+
PU
1
RG
1
RF
– V
) V
– V
OUT
driving a capacitive load
Driving capacitive loads with high performance amplifiers is not a problem as long as certain precautions are
taken. The first is to realize that the THS413x has been internally compensated to maximize its bandwidth and
slew rate performance. When the amplifier is compensated in this manner, capacitive loading directly on the
output will decrease the device’s phase margin leading to high frequency ringing or oscillations. Therefore, for
capacitive loads of greater than 10 pF, it is recommended that a resistor be placed in series with the output of
the amplifier, as shown in Figure 33. A minimum value of 20 Ω should work well for most applications. For
example, in 50-Ω transmission systems, setting the series resistor value to 50 Ω both isolates any capacitance
loading and provides the proper line impedance matching at the source end.
390 Ω
20 Ω
Output
390 Ω
THS413x
20 Ω
390 Ω
Output
390 Ω
Figure 33. Driving a Capacitive Load
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SLOS318E − MAY 2000 − REVISED JANUARY 2004
APPLICATION INFORMATION
Active antialias filtering
For signal conditioning in ADC applications, it is important to limit the input frequency to the ADC. Low-pass
filters can prevent the aliasing of the high frequency noise with the frequency of operation. The following figure
presents a method by which the noise may be filtered in the THS413x.
C1
R2
V
CC
R4
+
C3
C3
R1
R3
R3
−
+
V
−
V
V
+
IN
IN
R
THS413x
(t)
C2
THS1050
−
Vs
−
+
IN
V
+
V
OCM
IN
V
OCM
R1
V
IC
R4
V
CC
−
+
C1
R2
Figure 34. Antialias Filtering
The transfer function for this filter circuit is:
ȡ
ȣ
Rt
ȡ
ȣ
2R4 ) Rt
K
R2
R1
ȧ
ȧx
Where K +
H (f) + ȧ
ȧ
ȧ
j2πfR4RtC3ȧ
d
2
jf
1 )
Ȣ
f
1
2R4 ) Rt Ȥ
–ǒFSF x fcǓ
)
) 1
Ȣ
Ȥ
Q FSF x fc
Ǹ
2 x R2R3C1C2
R3C1 ) R2C1 ) KR3C1
1
FSF x fc +
and Q +
Ǹ
2π 2 x R2R3C1C2
K sets the pass band gain, fc is the cutoff frequency for the filter, FSF is a frequency scaling factor, and Q is the
quality factor.
2
ǸRe ) Im
2
|
|
2
2
ǸRe ) Im
|
|
FSF +
and Q +
2Re
where Re is the real part, and Im is the imaginary part of the complex pole pair. Setting R2 = R, R3 = mR,
C1 = C, and C2 = nC results in:
Ǹ
2 x mn
1
FSF x fc +
and Q +
Ǹ
(
)
1 ) m 1 ) K
2πRC 2 x mn
Start by determining the ratios, m and n, required for the gain and Q of the filter type being designed, then select
C and calculate R for the desired fc.
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SLOS318E − MAY 2000 − REVISED JANUARY 2004
PRINCIPLES OF OPERATION
theory of operation
The THS413x is a fully differential amplifier. Differential amplifiers are typically differential in/single out, whereas
fully differential amplifiers are differential in/differential out.
Differential Amplifier
THS413x
Fully differential Amplifier
R
f
V
CC+
R
R
(g)
(g)
_
_
+
V
V
V
IN−
O+
+
_
V
IN+
O−
+
R
f
V
OCM
V
CC−
Figure 35. Differential Amplifier Versus a Fully Differential Amplifier
To understand the THS413x fully differential amplifiers, the definition for the pinouts of the amplifier are
provided.
ǒV Ǔ ǒV Ǔ
)
I)
I–
ǒVI)Ǔ – ǒVI–Ǔ
Input voltage definition
V
+
V
+
ID
IC
V
2
ǒVO)
Ǔ
ǒVO–Ǔ
)
ǒVO)Ǔ – ǒVO–Ǔ
Output voltage definition
Transfer function
V
+
+
OD
OD
OC
2
V
+ V
x A
ID
+ V
ǒ Ǔ
f
Output common mode voltage V
OC
OCM
Differential Structure Rejects
Coupled Noise at The Input
Differential Structure Rejects
Coupled Noise at The Output
V
CC+
_
V
V
V
IN−
O+
+
_
V
IN+
O−
+
Differential Structure Rejects
Coupled Noise at The Power Supply
V
OCM
CC−
V
Figure 36. Definition of the Fully Differential Amplifier
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SLOS318E − MAY 2000 − REVISED JANUARY 2004
PRINCIPLES OF OPERATION
theory of operation (continued)
The following schematics depict the differences between the operation of the THS413x, fully differential
amplifier, in two different modes. Fully differential amplifiers can work with differential input or can be
implemented as single in/differential out.
R
f
V
CC+
R
R
(g)
(g)
V
IN−
IN+
−
+
V
V
O+
+
−
Vs
O−
V
V
OCM
V
CC−
R
f
Note: For proper operation, maintain symmetry by setting
R 1 = R 2 = R and R 1 = R 2 = R ⇒ A = R /R
f
f
f
(g) (g) (g) (g)
f
Figure 37. Amplifying Differential Signals
R
f
V
CC+
RECOMMENDED RESISTOR VALUES
R
R
(g)
(g)
V
V
IN−
GAIN
R
Ω
R Ω
f
(g)
−
+
V
V
O+
+
−
1
2
5
10
390
374
402
402
390
750
2010
4020
O−
IN+
V
OCM
Vs
V
CC−
R
f
Figure 38. Single In With Differential Out
If each output is measured independently, each output is one-half of the input signal when gain is 1. The
following equations express the transfer function for each output:
1
2
V
+
V
O
I
The second output is equal and opposite in sign:
1
V
+ –
V
O
I
2
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SLOS318E − MAY 2000 − REVISED JANUARY 2004
PRINCIPLES OF OPERATION
theory of operation (continued)
Fully differential amplifiers may be viewed as two inverting amplifiers. In this case, the equation of an inverting
amplifier holds true for gain calculations. One advantage of fully differential amplifiers is that they offer twice as
much dynamic range compared to single-ended amplifiers. For example, a 1-V ADC can only support an input
PP
signal of 1 V . If the output of the amplifier is 2 V , then it will not be practical to feed a 2-V signal into the
PP
PP
PP
targeted ADC. Using a fully differential amplifier enables the user to break down the output into two 1-V signals
PP
with opposite signs and feed them into the differential input nodes of the ADC. In practice, the designer has been
able to feed a 2-V peak-to-peak signal into a 1-V differential ADC with the help of a fully differential amplifier.
The final result indicates twice as much dynamic range. Figure 39 illustrates the increase in dynamic range. The
gain factor should be considered in this scenario. The THS413x fully differential amplifier offers an improved
CMRR and PSRR due to its symmetrical input and output. Furthermore, second harmonic distortion is
improved. Second harmonics tend to cancel because of the symmetrical output.
a
V
OD
= 1−0 = 1
V
CC+
+1
_
V
V
V
IN−
O+
+
_
0
V
+1
IN+
O−
+
0
V
OCM
CC−
V
OD
= 0−1 = −1
V
b
Figure 39. Fully Differential Amplifier With Two 1-V Signals
PP
Similar to the standard inverting amplifier configuration, input impedance of a fully differential amplifier is
selected by the input resistor, R . If input impedance is a constraint in design, the designer may choose to
(g)
implement the differential amplifier as an instrumentation amplifier. This configuration improves the input
impedance of the fully differential amplifier. The following schematic depicts the general format of
instrumentation amplifiers.
The general transfer function for this circuit is:
V
R
OD
– V
f
2R2
R1
ǒ1 )
Ǔ
+
V
R
IN1
IN2
(g)
THS4012
R
R
(g)
f
+
V
IN1
_
R2
_
+
R1
R2
THS413x
_
+
V
IN2
R
R
THS4012
(g)
f
Figure 40. Instrumentation Amplifier
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SLOS318E − MAY 2000 − REVISED JANUARY 2004
PRINCIPLES OF OPERATION
circuit layout considerations
To achieve the levels of high frequency performance of the THS413x, follow proper printed-circuit board high
frequency design techniques. A general set of guidelines is given below. In addition, a THS413x evaluation
board is available to use as a guide for layout or for evaluating the device performance.
D
Ground planes—It is highly recommended that a ground plane be used on the board to provide all
components with a low inductive ground connection. However, in the areas of the amplifier inputs and
output, the ground plane can be removed to minimize the stray capacitance.
D
Proper power supply decoupling—Use a 6.8-µF tantalum capacitor in parallel with a 0.1-µF ceramic
capacitor on each supply terminal. It may be possible to share the tantalum among several amplifiers
depending on the application, but a 0.1-µF ceramic capacitor should always be used on the supply terminal
of every amplifier. In addition, the 0.1-µF capacitor should be placed as close as possible to the supply
terminal. As this distance increases, the inductance in the connecting trace makes the capacitor less
effective. The designer should strive for distances of less than 0.1 inches between the device power
terminals and the ceramic capacitors.
D
D
Sockets—Sockets are not recommended for high-speed operational amplifiers. The additional lead
inductance in the socket pins will often lead to stability problems. Surface-mount packages soldered directly
to the printed-circuit board is the best implementation.
Short trace runs/compact part placements—Optimum high frequency performance is achieved when stray
series inductance has been minimized. To realize this, the circuit layout should be made as compact as
possible, thereby minimizing the length of all trace runs. Particular attention should be paid to the inverting
input of the amplifier. Its length should be kept as short as possible. This will help to minimize stray
capacitance at the input of the amplifier.
D
Surface-mount passive components—Using surface-mount passive components is recommended for high
frequency amplifier circuits for several reasons. First, because of the extremely low lead inductance of
surface-mount components, the problem with stray series inductance is greatly reduced. Second, the small
size of surface-mount components naturally leads to a more compact layout thereby minimizing both stray
inductance and capacitance. If leaded components are used, it is recommended that the lead lengths be
kept as short as possible.
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SLOS318E − MAY 2000 − REVISED JANUARY 2004
PRINCIPLES OF OPERATION
power-down mode
The power-down mode is used when power saving is required. The power-down terminal (PD) found on the
THS413x is an active low terminal. If it is left as a no-connect terminal, the device will always stay on due to an
internal 50 kΩ resistor to V . The threshold voltage for this terminal is approximately 1.4 V above V
. This
CC
CC−
means that if the PD terminal is 1.4 V above V
, the device is active. If the PD terminal is less than 1.4 V above
CC−
V
, the device is off. For example, if V
= −5 V, then the device is on when PD reaches −3.6 V, (−5 V +
CC−
CC−
1.4 V = −3.6 V). By the same calculation, the device is off below −3.6 V. It is recommended to pull the terminal
to V in order to turn the device off. The following graph shows the simplified version of the power-down
CC−
circuit. While in the power-down state, the amplifier goes into a high impedance state. The amplifier output
impedance is typically greater than 1 MΩ in the power-down state.
V
CC
50 kΩ
To Internal Bias
Circuitry Control
PD
V
CC−
Figure 41. Simplified Power-Down Circuit
Due to the similarity of the standard inverting amplifier configuration, the output impedance appears to be very
low while in the power-down state. This is because the feedback resistor (R ) and the gain resistor (R ) are
f
(g)
still connected to the circuit. Therefore, a current path is allowed between the input of the amplifier and the output
of the amplifier. An example of the closed loop output impedance is shown in Figure 42.
OUTPUT IMPEDANCE (IN POWER DOWN)
vs
FREQUENCY
2200
V
= 5 V
CC
G = 1
R = 1 kΩ
f
PD = V
CC−
1200
200
100 k
1 M
10 M
100 M
1 G
f − Frequency − Hz
Figure 42
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SLOS318E − MAY 2000 − REVISED JANUARY 2004
PRINCIPLES OF OPERATION
general PowerPAD design considerations
The THS413x is available packaged in a thermally-enhanced DGN package, which is a member of the
PowerPAD family of packages. This package is constructed using a downset leadframe upon which the die is
mounted [see Figure 43(a) and Figure 43(b)]. This arrangement results in the lead frame being exposed as a
thermal pad on the underside of the package [see Figure 43(c)]. Because this thermal pad has direct thermal
contact with the die, excellent thermal performance can be achieved by providing a good thermal path away
from the thermal pad.
The PowerPAD package allows for both assembly and thermal management in one manufacturing operation.
During the surface-mount solder operation (when the leads are being soldered), the thermal pad can also be
soldered to a copper area underneath the package. Through the use of thermal paths within this copper area,
heat can be conducted away from the package into either a ground plane or other heat dissipating device.
The PowerPAD package represents a breakthrough in combining the small area and ease of assembly of the
surface mount with the, heretofore, awkward mechanical methods of heatsinking.
More complete details of the PowerPAD installation process and thermal management techniques can be found
in the Texas Instruments Technical Brief, PowerPAD Thermally Enhanced Package (SLMA002). This document
can be found at the TI web site (www.ti.com) by searching on the key word PowerPAD. The document can also
be ordered through your local TI sales office. Refer to literature number SLMA002 when ordering.
DIE
Side View (a)
Thermal
Pad
DIE
End View (b)
Bottom View (c)
NOTE A: The thermal pad is electrically isolated from all terminals in the package.
Figure 43. Views of Thermally Enhanced DGN Package
23
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