TPS40020PWPRG4 [TI]
Enhanced Low Input (2.25V-5.5V) up to 1MHz Frequency, Sync. Buck Controller, source only 16-HTSSOP -40 to 85;型号: | TPS40020PWPRG4 |
厂家: | TEXAS INSTRUMENTS |
描述: | Enhanced Low Input (2.25V-5.5V) up to 1MHz Frequency, Sync. Buck Controller, source only 16-HTSSOP -40 to 85 开关 光电二极管 |
文件: | 总38页 (文件大小:930K) |
中文: | 中文翻译 | 下载: | 下载PDF数据表文档文件 |
Not Recommended for New Designs
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SLUS535D − MARCH 2003 − REVISED JULY 2007
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FEATURES
DESCRIPTION
D
D
D
D
D
D
Operating Input Voltage 2.25 V to 5.5 V
The TPS4002x family of dc-to-dc controllers are designed
for non-isolated synchronous buck regulators, providing
enhanced operation and design flexability through user
programmability.
Output Voltage as Low as 0.7 V
1% Internal 0.7 V Reference
Predictive Gate Drive N-Channel MOSFET
Drivers for Higher Efficiency
The TPS4002x utilizes a proprietary Predictive Gate
Drive technology to minimize the diode conduction
losses associated with the high-side and synchronous
rectifier N-channel MOSFET transistions. The integrated
charge pump with boost circuit provides a regulated 5-V
gate drive for both the high side and synchronous rectifier
N-channel MOSFETs. The use of the Predictive Gate
Drive technology and charge pump/boost circuits
combine to provide a highly efficient, smaller and less
expensive converter.
Externally Adjustable Soft-Start and Short
Circuit Current Limit
Programmable Fixed-Frequency
100 KHz-to-1 MHz Voltage-Mode Control
D
Source-Only Current or Source/Sink Current
D
Quick Response Output Transient
Comparators with Power Good Indication
Provide Output Status
Design flexibility is provided through user programmability
of such functions as: operating frequency, short circuit
current detection thresholds, soft-start ramp time, and
external synchronization frequency. The operating
frequency is programmable using a single resistor over a
frequency range of 100 kHz to 1 MHz. Higher operating
frequencies yield smaller component values for a given
converter power level as well as faster loop closure.
D
16-Pin PowerPAD Package
APPLICATIONS
D
D
D
D
D
Networking Equipment
Telecom Equipment
Base Stations
Servers
DSP Power
VDD
VOUT
TPS40020
VDD 2.25 V − 5.5 V
ILIM/
SYNC
1
2
3
4
5
6
7
8
16
15
BOOT1
HDRV
VOUT
VDD
OSNS
FB
SW 14
BOOT2 13
PVDD 12
LDRV 11
PGND 10
COMP
SS/SD
RT
SGND
9
PWRGD
UDG−02094
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments
semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAD and Predictive Gate Drive are trademarks of Texas Instruments.
ꢁꢘ ꢏ ꢌꢓ ꢋ ꢀꢒ ꢏ ꢈ ꢌ ꢊꢀꢊ ꢛꢜ ꢝꢞ ꢟ ꢠꢡ ꢢꢛꢞꢜ ꢛꢣ ꢤꢥ ꢟ ꢟ ꢦꢜꢢ ꢡꢣ ꢞꢝ ꢧꢥꢨ ꢩꢛꢤ ꢡꢢꢛ ꢞꢜ ꢪꢡ ꢢꢦꢫ ꢁꢟ ꢞꢪꢥ ꢤꢢꢣ
ꢤ ꢞꢜ ꢝꢞꢟ ꢠ ꢢꢞ ꢣ ꢧꢦ ꢤ ꢛ ꢝꢛ ꢤ ꢡ ꢢꢛ ꢞꢜꢣ ꢧꢦ ꢟ ꢢꢬꢦ ꢢꢦ ꢟ ꢠꢣ ꢞꢝ ꢀꢦꢭ ꢡ ꢣ ꢒꢜꢣ ꢢꢟ ꢥꢠ ꢦꢜꢢ ꢣ ꢣꢢ ꢡꢜꢪ ꢡꢟ ꢪ ꢮ ꢡꢟ ꢟ ꢡ ꢜꢢꢯꢫ
ꢁꢟ ꢞ ꢪꢥꢤ ꢢ ꢛꢞ ꢜ ꢧꢟ ꢞ ꢤ ꢦ ꢣ ꢣ ꢛꢜ ꢰ ꢪꢞ ꢦ ꢣ ꢜꢞꢢ ꢜꢦ ꢤꢦ ꢣꢣ ꢡꢟ ꢛꢩ ꢯ ꢛꢜꢤ ꢩꢥꢪ ꢦ ꢢꢦ ꢣꢢꢛ ꢜꢰ ꢞꢝ ꢡꢩ ꢩ ꢧꢡ ꢟ ꢡꢠ ꢦꢢꢦ ꢟ ꢣꢫ
Copyright 2004, Texas Instruments Incorporated
Not Recommended for New Designs
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SLUS535D − MARCH 2003 − REVISED JULY 2007
DESCRIPTION (CONTINUED)
The short circuit current detection is programmable through a single resistor, allowing the short circuit current limit
detection threshold to be easily tailored to accommodate different size (R ) MOSFETs. The short circuit current
DS(on)
function provides pulse-by-pulse current limiting during soft-start and short term transient conditions as well as a fault
counter to handle longer duration short circuit current conditions. If a fault is detected the controller shuts down for
a period of time determined by six (6) consecutive soft-start cycles. The controller automatically retries the output
th
every seventh (7 ) soft-start cycle.
In addition to determining the off time during a fault condition, the soft-start ramp provides a closed loop controlled
ramp of the converter output during startup. Programmability allows the ramp rate to be adjusted for a wide variety
of output L-C component values.
The output voltage transient comparators provide a quick response , first strike, approach to output voltage transients. The
output voltage is sensed through a resistor divider at the OSNS pin. If an overvoltage condition is detected the HDRV gate
drive is shut-off and the LDRV gate drive is turned on until the output is returned to regulation. Similarly, if an output
undervoltage condition is sensed the HDRV gate drive goes to 95% duty cycle to pump the output back up quickly. In either
case, the PowerGood open drain output pulls low to indicate an output voltage out of regulation condition. The PowerGood
output can be daisy-chained to the SS/SD pin or enable pin of other controllers or converters for output voltage sequencing.
The transient comparators can be disabled by simply tying the OSNS pin to VDD.
The TPS4002x can be externally synchronized through the ILIM/SYNC pin up to 1.5× the free-running frequency. This
allows multiple contollers to be synchronized to eliminate EMI concerns due to input beat frequencies between controllers.
INTERNAL BLOCK DIAGRAM
VDD
2
3
VDD
0.719 V
OSNS
VDD
13 BOOT2
12 PVDD
16 BOOT1
15 HDRV
14 SW
SS
CHARGE
PUMP
ACTIVE
PWRGD
9
0.659 V
FB
4
5
0.69 V
+
+
DRV
PREDICTIVE
GATE
PWM
COMP
DRIVE(tm)
PWM
UVLO
OSC
CLK
LOGIC
PVDD
RT
7
UVLO
IRT
DRV
11 LDRV
10 PGND
FAULT
CLK
VDD
IRT
SS
ACTIVE
CURRENT LIMIT
COMPARATOR
SOFT
START
FAULT
COUNTER
I
−
SS
OC
SS/SD
SGND
6
8
1
ILIM/SYNC
DCHG
+
UVLO
SD
SYNC
0.28 V
1 V
VDD
UVLO
UVLO
+
DISABLE
V
DD
1.4 V
UDG−02092
2
Not Recommended for New Designs
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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during
storage or handling to prevent electrostatic damage to the MOS gates.
ORDERING INFORMATION
(1)
T
LOAD CURRENT
PACKAGE
PART NUMBER
TPS40020PWP
TPS40021PWP
A
(2)
(2)
SOURCE
Plastic HTSSOP (PWP)
−40°C to 85°C
SOURCE/SINK
Plastic HTSSOP (PWP)
(1)
(2)
See page 7 for explanation.
The PWP package is also available taped and reeled. Add an R suffix to the device type (i.e., TPS40020PWPR). See the application section of
the data sheet for PowerPAD drawing and layout information.
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature range unless otherwise noted
(4)
TPS4002X
UNIT
SS/SD, VDD, PVDD, OSNS
BOOT2, BOOT1
SW
−0.3 to 6
V
SW
+ 6
−3.0 to 10.5
−5
Input voltage range, V
IN
V
SWT (SW transient < 50 ns)
FB, ILIM
−0.3 to 6.0
−0.3 to 6
10
Output voltage range, V
OUT
COMP, PWRGD, RT
PWRGD
Sink current, I
mA
S
Operating virtual junction temperature range, T
−40 to 125
−55 to 150
260
J
Storage temperature, T
stg
°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds
(4)
Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only,
and functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is
not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
RECOMMENDED OPERATING CONDITIONS
MIN NOM MAX UNIT
Input voltage, V
IN
2.25
−40
5.5
85
V
Operating junction temperature, T
°C
J
(5)(6)
PWP PACKAGE
(TOP VIEW)
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
ILIM/SYNC
VDD
BOOT1
HDRV
SW
BOOT2
PVDD
LDRV
OSNS
FB
COMP
SS/SD
RT
THERMAL
PAD
PGND
PWRGD
SGND
(5) For more information on the PWP package, refer to TI Technical Brief, Literature No. SLMA002.
(6) PowerPADt heat slug must be connected to SGND (Pin 8), or electrically isolated from all other pins.
3
Not Recommended for New Designs
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SLUS535D − MARCH 2003 − REVISED JULY 2007
ELECTRICAL CHARACTERISTICS
T = −40°C to 85°C, T = T
V
= 5.0 V (unless otherwise noted)
J
J
A, DD
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
INPUT SUPPLY
V
V
Input voltage range, VDD
PVDD pin voltage
Switching current
Quiescent current
Shutdown current
Minimum on-voltage
Hysteresis
2.25
5.50
5.2
DD
V
V
= 3.3 V
4.9
3.5
PVDD
DD
500 kHz, No load on HDRV, LDRV
FB = 0.8 V
5.0
2.0
3.0
I
mA
DD
SS/SD = 0 V, Outputs OFF
0.38
2.05
130
1.00
2.15
200
1.95
80
V
V
UVLO
mV
OSCILLATOR
2.25 V ≤ V
2.25 V ≤ V
≤ 5.00 V,
≤ 5.00 V,
R
= 69.8 kΩ
= 34.8 kΩ
425
800
500
950
575
1100
1.07
0.41
DD
T
f
Accuracy
kHz
V
OSC
R
T
DD
V
V
Ramp voltage
V
−V
0.80
0.24
0.93
0.31
RAMP
PEAK VAL
Ramp valley voltage
VAL
PWM
V
V
= V , R = 34.8 kΩ,
DD
OSNS
DD
T
85%
90%
94%
95%
= 3.3 V, FB = 0 V
d
d
Maximum duty cycle
MAX
V
V
= V , R = 70 kΩ,
DD
OSNS
DD
T
= 5.0 V, FB = 0 V
Minimum duty cycle
0%
MIN
MIN
(2)
t
Minimum HDRV on-time
250
ns
ERROR AMPLIFIER
V
Feedback input voltage
Input bias current
−40°C ≤ T ≤ 85°C, 2.25V ≤ V
DD
≤ 5.00V
0.685 0.690
30
0.697
130
V
FB
A
I
nA
BIAS
V
High-level output voltage
I
I
= 0.5 mA, V
= GND
2.0
2.5
0.08
7
OH
OL
OH
FB
V
V
Low-level output voltage
= 0.5 mA, V
= GND
= V
DD
0.15
OL
FB
I
I
High-level output source current
Low-level output sink current
V
V
3
3
OH
FB
mA
= V
DD
8
OL
FB
(1)
Gain bandwidth
G
5
10
85
MHz
dB
BW
(1)
Open loop gain
A
OL
55
CURRENT LIMIT
I
Current limit sink current
2.25 V ≤ V
DD
≤ 5.00 V,
R
T
= 69.8 kΩ
165
−20
190
0
215
20
µA
SINK
V
Current limit offset voltage
mV
OS
ON
ON
SS
t
t
t
Minimum HDRV on−time in overcurrent
Switch leading-edge blanking pulse time
Soft-start cycles
V
DD
= 3.3 V
200
140
6
300
ns
(1)
cycles
V
V
Current limit input voltage range
2
VDD
5.4
ILIM
SOFT START
I
Soft-start source current
Outputs = OFF
2.0
3.3
µA
SS
(1)
(2)
Ensured by design. Not production tested.
Operation below the minimum on-time could result in overlap of the HDRV and LDRV outputs.
4
Not Recommended for New Designs
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ELECTRICAL CHARACTERISTICS (continued)
T = −40°C to 85°C, T = T
V
= 5.0 V (unless otherwise noted)
J
J
A, DD
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
SHUTDOWN
V
V
Shutdown threshold voltage
0.22
0.25
0.26
0.28
0.29
0.32
SD
V
Device enable threshold voltage
EN
OUTPUT DRIVER
V
− V
(SW)
= 3.3 V,
= 3.3 V,
(BOOT1)
R
R
High-side driver pull-up resistance
High-side driver pull-down resistance
1.0
0.8
2.5
1.5
5.0
3.0
HDHI
I
100mA
SOURCE =
V
− V
(SW)
100mA
(BOOT1)
HDLO
Ω
I
SINK =
R
R
Low-side driver pull-up resistance
Low-side driver pull-down resistance
Low-side driver rise time
P
P
= 3.3 V, I
100 mA
SOURCE =
1.0
2.5
0.80
15
5.0
1.50
35
LDHI
VDD
= 3.3 V, I 100 mA
SINK =
0.45
LDLO
VDD
t
t
t
t
LRISE
LFALL
HRISE
HFALL
Low-side driver fall time
10
25
C
= 1 nF
ns
LOAD
High-side driver rise time
15
35
High-side driver fall time
10
25
THERMAL SHUTDOWN
Shutdown temperature
(1)
(1)
165
15
T
SD
°C
Hysteresis
CHARGE PUMP
R
VB2
R
B2P
R
PB1
R
R
R
VDD to BOOT2
BOOT2 to PVDD
PVDD to BOOT1
V
DD
V
DD
V
DD
= 5.0 V,
I
I
I
10 mA
2.8
2.8
2.9
6.6
5.6
5.9
10.4
8.4
DS(on)
DS(on)
DS(on)
SOURCE =
SOURCE =
SOURCE =
= 5.0 V,
= 5.0 V,
10 mA
10 mA
Ω
8.9
POWER GOOD
V
V
= 0.8 V,
I
0.5 mA,
OSNS
= 3.3 V
PWRGD =
V
Pull-down voltage
50
6
90
10
140
14
mV
PGD
DD
Output sense high to power good low delay
time
0.7 V≤ V
≤ 0.8 V, I
≤ 0.7 V, I
0.5 mA,
PWRGD =
OSNS
t
ONHPL
V
DD
= 3.3 V
Output sense low to power good low delay
time
0.6 V≤ V
0.5 mA,
PWRGD =
OSNS
= 3.3 V
t
6
10
14
ONLPL
V
DD
µs
V
V
= 0.7 V, I
= 3.3 V, 0.0 V ≤ V
0.5 mA,
≤ 0.4 V
OSNS
DD
PWRGD =
t
Shutdown high to power good high delay time
Shutdown low to power good low delay time
2
4
6
SDHPH
SS/SD
V
V
= 0.7 V, I
= 3.3 V, 0.0 V ≤ V
SS/SD
0.5 mA,
≤ 0.4 V
OSNS
DD
PWRGD =
t
0.5
140
140
1.5
500
500
3.0
SDLPL
Output sense high to nominal to power good
high delay time
0.7 V≤ V
≤ 0.8 V, I
0.5 mA,
PWRGD =
OSNS
= 3.3 V
t
1000
1000
ONHPH
V
DD
ns
Output sense low to nominal to power good
high delay time
0.6 V≤ V
≤ 0.7 V, I
0.5 mA,
PWRGD =
OSNS
= 3.3 V
t
ONLPH
V
DD
TRANSIENT COMPARATORS
Overvoltage output threshold voltage
23
8
29
15
35
22
V
OV
Hysteresis
Referenced to V
Referenced to V
mV
V
FB
Undervoltage output threshold voltage
Hysteresis
−37
8
−31
15
−25
22
V
V
UV
OSNS minimum disable voltage
0.5
DIS
DD
(1)
Ensured by design. Not production tested.
5
Not Recommended for New Designs
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ELECTRICAL CHARACTERISTICS (continued)
T = −40°C to 85°C, T = T = 5.0 V (unless otherwise noted)
V
J
J
A, DD
PARAMETER
SYNCHRONIZATION
TEST CONDITIONS
MIN
TYP
MAX
UNIT
V
Synchronization enable low threshold voltage
0.7
50
ENSY
BLNK
MIN
V
Synchronization current limit enable threshold
voltage
V
Referenced to VDD
−0.7
t
Minimum synchronization input pulse width
35
ns
PREDICTIVE DELAY
V
Sense voltage to modulate delay
Maximum delay modulation
Counter delay/bit time
−200
65
mV
ns
SWP
LDRV OFF-to-HDRV ON
LDRV OFF-to-HDRV ON
HDRV OFF-to-LDRV ON
HDRV OFF-to-LDRV ON
40
2.5
55
90
6.2
105
6.5
t
LDHD
4.5
80
Maximum delay modulation
Counter delay/bit time
t
HDLD
2.2
5.0
RECTIFIER ZERO CURRENT COMPARATOR
Sense voltage to turn off
rectifier MOSFET
V
TPS40020 LDRV output = OFF
−5
−2.5
150
2
mV
ns
SW
(1)
t
Zero current blanking time
ZBLNK
(1)
Ensured by design. Not production tested.
TERMINAL FUNCTIONS
TERMINAL
I/O
DESCRIPTION
NAME
NO.
This pin provides a bootstrapped supply for the high side FET driver, enabling the gate of the high side FET to be
driven above the input supply rail. Connect a capacitor from this pin to the SW pin.
BOOT1
16
I
I
This pin provides a secondary bootstrapping necessary for generation of PVDD. Connect a capacitor from this
pin to SW.
BOOT2
13
COMP
FB
5
4
O
I
Output of the error amplifier. Refer to Electrical Characteristics table for loading constraints.
Inverting input of the error amplifier. In normal operation, V
FB
is equal to the internal reference level of 690 mV.
The gate drive output for the high side N-channel MOSFET switch is bootstrapped to near PVDD for good
enhancementof the high-side switch. The HDRV switches from BOOT1 to SW.
HDRV
15
O
The current limit pin is used to set the current limit threshold. A current sink from this pin to GND sets the threshold
voltage for output short circuit current across a resistor connected to VDD. Synchronization is accomplished by
pulling IMAX to less than 1 V for a period greater than the minimum pulse width and then releasing. An open
collector or drain device should be used. These pulses must be of higher frequency than the free running frequency
of the local oscillator.
ILIM/SYNC
1
I
Gate drive output for the low-side synchronous rectifier N-channel MOSFET. LDRV switches from PVDD to
PGND.
LDRV
OSNS
11
3
O
O
The output sense pin is connected to a resistor divider from VOUT to GND (identical to the main feedback loop)
and is used to sense power good condition and provides reference for the transient comparators.
PGND
10
9
O
−
Power (high-current) ground used by LDRV.
PWRGD
Power good. This is an open-drain output which connects to the supply via an external resistor.
This pin is the regulated output of the charge-pump and provides the supply voltage for the LDRV driver stage.
PVDD also drives the bootstrap circuit which generates the voltage on BOOT1.
PVDD
12
O
RT
7
8
I
External pin for programming the oscillator frequency. Connnected a resistor between this pin and GND.
Signal ground
SGND
−
The soft-start/shutdown pin provides user programmable soft-start timing and shutdown capability for the
controller.
SS/SD
6
I
This pin, used for overcurrent, zero-current, and in the anti-cross conduction sensing is connected to the switched
node on the converter. Output short circuit is detected by sensing the voltage at this pin with respect to VDD while
the high-side switch is on. Zero current is detected by sensing the pin voltage with respect to ground when the
low-side rectifier MOSFET is on.
SW
14
2
I
I
VDD
Power input for the device. Maximum voltage is 5.5 V. De-coupling of this pin is required.
6
Not Recommended for New Designs
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APPLICATION INFORMATION
The TPS4002x series of devices are low-input voltage, synchronous, voltage mode-buck controllers. A typical
application circuit is shown in Figure 1. These controllers are designed to allow construction of
high-performance dc-to-dc converters with input voltages from 2.25 V to 5.5 V, and output voltages as low as
690 mV. Using a top side N-channel MOSFET for the primary buck switch results in lower switch resistance
for a given gate charge.
The device controls the delays from main switch off to rectifier turn on and from rectifier turn off to main switch
turn on in a way that minimizes diode losses (both conduction and recovery) in the synchronous rectifier. The
reduction in these losses is significant and can mean that for a given converter power level, smaller FETs can
be used, or that heat sinking can be reduced or even eliminated.
The TPS40021 is the controller of choice for most general purpose synchronous buck designs, operating in two
quadrant mode (i.e. source or sink current) full time. This choice provides the best performance for output
voltage load transient response over the widest load current range.
The TPS40020 operates in single quadrant mode (source current only) full time, allowing the paralleling of
converters. Single quadrant operation ensures one converter does pull current from a paralleled converter. A
converter using one of these controllers emulates a non-synchronous buck converter at light loads. When
current in the output inductor attempts to reverse, an internal zero-current detection circuit turns OFF the
synchronous rectifier and causes the current flow in the inductor to become discontinuous. At average load
currents greater than the peak amplitude of the inductor ripple current, the converter returns to operation as
a synchronous buck converter to maximize efficiency.
The controller provides for a coarse short circuit current-limit function that provides pulse-by-pulse current
limiting, as well as integrates short circuit current pulses to determine the existence of a persistant fault state
at the converter output. If a fault is detected, the converter shuts down for a period of time (determined by six
soft-start cycles) and then restarts. The current-limit threshold is adjustable with a single resistor connected
from VDD to the ILIM/SYNC pin. This overcurrent function is designed to protect against catastrophic faults
only, and cannot be guaranteed to protect against all overcurrent conditions.
The controller implements a closed-loop soft start function. Startup ramp time is set by a single external
capacitor connected to the SS/SD pin. The SS/SD pin also doubles as a shutdown function.
VOLTAGE REFERENCE
The bandgap cell is designed with a trimmed, curvature corrected (< 1%) 0.69-V output, allowing output
voltages as low as 690 mV to be obtained.
Oscillator
The ramp waveform is a saw-tooth form at the PWM frequency with a peak voltage of 1.25 V, and a valley of
0.3 V. The PWM duty cycle is limited to a maximum of 97%, allowing the bootstrap and charge pump capacitors
to charge during every cycle.
7
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APPLICATION INFORMATION
Bootstrap/Charge Pump
The TPS4002X series includes a charge pump to boost the drive voltage to the power MOSFET’s to higher
levels when the input supply is low. A capacitor connected from PVDD to PGND is the storage cap for the pump.
A capacitor connected from SW to BOOT2 gets charged every switching cycle while LDRV is high and its charge
is dumped on the PVDD capacitor when HDRV goes high. An internal switch disables the charge pump when
the voltage on PVDD reaches approximately 4.8 V and enables pumping when PVDD falls to approximately
4.6 V. The high-side driver uses the capacitor from SW to BOOT1 as its power supply. When SW is low, this
capacitor charges from the PVDD capacitor. When the SW pin goes high, this capacitor provides above-rail
drive for the high-side N-channel FET.
PVDD, BOOT1 and BOOT2 are pre-charged to the VDD voltage during a shutdown condition. For low-input
voltage converters, utilizing higher gate threshold voltage MOSFETs, it may be necessary to add an Schottky
diode from VDD (anode) to BOOT1 to guarantee sufficient voltage for initial start up. Once switching starts the
charge pump reverses bias on the Schottky diode.
When operating the TPS40020 under no load or extremely light-load conditions the controller will be operating
in discontinuous Mode (DCM); reverse current is prevented from flowing in the synchronous rectifier. In DCM
the on times for both the HDRV and LDRV pulses can become too narrow to provide adequate charging of
PVDD and BOOT1 outputs, causing their voltages to collapse. Insufficient PVDD and BOOT1 voltages prevent
the external MOSFETS from becomming fully enhanced, causing loss of converter output regulation. Schottky
diodes from VIN (anode) to PVDD, and VIN (anode) to BOOT1, as well as a pre-load can be added to maintain
PVDD and BOOT1 at voltage levels sufficient enough to fully enhance the external MOSFETs. The amount of
pre-load typically ranges from 50 mA to 100 mA depending on operating conditions and external MOSFET
selection.
Drivers
The HDRV and LDRV MOSFET drivers are capable of driving gate-to-source voltages up to 5.0 V. Using
appropriate MOSFETs, a 25-A converter can be achieved. The LDRV driver switches between VDD and
ground, while the HDRV driver is referenced to SW and switches between BOOT1 and SW. The maximum
voltage between BOOT1 and SW is 5.0 V when PVDD is in regulation.
8
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Figure 1. Typical Application
9
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APPLICATION INFORMATION
Synchronous Rectification and Predictive Gate Delay
In a normal buck converter, when the high−side switch turns off, current is flowing in the inductor. Since this
current cannot be stopped immediately a rectifier or catch device is used to give this current a path to flow and
maintain voltage levels at a safe level. This device can be a simple diode or it can be an actively−controlled
transistor if a control signal is available to drive it. The TPS4002X provides a signal to drive an N−channel
MOSFET as a synchronous rectifier. This control signal is carefully coordinated with the drive signal for the
main switch so that there is absolute minimum dead−time between the turn off of one FET and the turn on of
the other. This TI−patented function, predictive gate delay, uses information from the current switching cycle
to adjust the delays for the next cycle virtually eliminating diode conduction while preventing cross−conduction
or shoot through. Figure 2 shows the switch−node voltage waveform for a synchronously rectified buck
converter during the synchronous rectification period. Illustrated are the relative effects of a fixed delay drive
scheme (constant, pre−set delays for the turn−off to turn−on intervals), an adaptive delay drive scheme
(variable delays based on voltages sensed on the current switching cycle) and TI’s predictive delay drive
scheme. Since the diode voltage drop is greater than the conduction drop of the FET, the longer time spent
in diode conduction, the more power dissipated in the rectifier and the lower the efficiency. Also, not shown
in the figure, is the fact that the predictive delay circuit can actually prevent the body diode from becoming
forward biased at all, avoiding reverse recovery and its associated losses. This results in a significant power
savings when the main FET turns on.
The predictive gate drive architecture on the TPS40020/21 requires a minimum pulse width of greater than
150 ns for proper operation. At pulse widths below 150 ns, the low−side FET turn−on could overlap the high−side
FET turn−off leading to cross conduction in the power stage.
GND
Channel Conduction
Body Diode Conduction
Fixed Delay
Adaptive Delay
Predictive Delay
UDG−01144
Figure 2. Switch Node Waveforms for Synchronous Buck Converter
10
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Output Short Circuit Protection
Output short circuit protection in the TPS4002x is sensed by looking at the voltage across the main FET while
it is on. If the voltage exceeds a pre-set threshold, the current pulse is terminated, and a counter inside the
device is incremented. If this counter fills up, a fault condition is declared and the chip disables switching for
a period of time and then attempts to restart the converter with a full soft-start cycle. The more detailed
explanation follows.
In each switching cycle, a comparator looks at the voltage across the top side FET while it is on. If the voltage
across that FET exceeds a programmable threshold voltage, then the current switching pulse is terminated and
a 3-bit counter (eight counts) is incremented by one count. If during the switching cycle the top side FET voltage
does not exceed a preset threshold, then this counter is decremented by one count. (The counter does not wrap
around from seven to zero or from zero to seven). If the counter reaches a full count of seven, the device
declares that a fault condition exists at the output of the converter. In this state, switching stops and the soft-start
capacitor is discharged. The counter is decremented by one by the soft start cap discharge. When the soft-start
capacitor is fully discharged, the discharge circuit is turned off and the cap is allowed to charge up at the nominal
charging rate, When the soft-start capacitor reaches approximately 1.3 V, it is discharged again and the
overcurrent counter is decremented by one count. The capacitor is charged and discharged, and the counter
decremented until the count reaches zero (a total of six times). When this happens, the outputs are again
enabled as the soft-start capacitor generates a reference ramp for the converter to follow while attempting to
restart. During this soft-start interval (whether or not the controller is attempting to do a fault recovery or starting
for the first time), pulse-by-pulse current limiting is in effect, but overcurrent pulses are not counted to declare
a fault until the soft-start cycle has been completed. It is possible to have a supply try to bring up a short circuit
for the duration of the soft-start period plus seven switching cycles. Power stage designs should take this into
account if it makes a difference thermally. Figure 3 shows the details of the overcurrent operation.
(+)
V
TS
(−)
Overcurrent
Threshold
Voltgage
Internal PWM
V
TS
0 V
External
Main Drive
Normal Cycle
Overcurrent
Cycle
UDG−03029
Figure 3. Switch Node Waveforms for Synchronous Buck Converter
11
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APPLICATION INFORMATION
Figure 4 shows the behavior of key signals during initial startup, during a fault and a successfully fault recovery.
At time t0, power is applied to the converter. The voltage on the soft-start capacitor (V ) begins to ramp up
CSS
At t1, the soft-start period is over and the converter is regulating its output at the desired voltage level. From
t0 to t1, pulse-by-pulse current limiting was in effect, and from t1 onward, overcurrent pulses are counted for
purposes of determining if a fault exists. At t2, a heavy overload is applied to the converter. This overload is
in excess of the overcurrent threshold, the converter starts limiting current and the output voltage falls to some
level depending on the overload applied. During the period from t2 to t3, the counter is counting overcurrent
pulses and at time t3 reaches a full count of 7. The soft-start capacitor is then discharged, the outputs are
disabled, the counter decremented, and a fault condition is declared.
V
DD
ꢀ 1.3 V
ꢀ 0.6 V
0.6 V
V
CSS
FAULT
I
LOAD
V
OUT
t
t4
t0
t1
t2 t3
t5
t6
t7
t8
t9
t10
Counter
0
6
5
4
3
2
1
0
1
2
3
4
5
6
7
cycles
UDG−03187
Figure 4. Overcurrent/Fault Waveforms
When the soft-start capacitor is fully discharged, it begins charging again at the same rate that it does on startup,
with a nominal 3-µA current source. As the capacitor voltage reaches full charge, it is discharged again and the
counter is decremented by one count. These transitions occur at t3 through t9. At t9, the counter has been
decremented to zero. Now the fault logic is cleared, the outputs are enabled and the converter attempts to
restart with a full soft-start cycle. The converter comes into regulation at t10.
The internal SS signal is a diode drop below V
. When V
reaches one diode drop above ground, (≅0.6 V)
CSS
CSS
the output (V
) begins it’s soft-start ramp.
OUT
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Setting the Short Circuit Current Limit Threshold
Connecting a resistor from VDD to ILIM sets the current limit. A current sink in the chip causes a voltage drop
across the resistor connected to ILIM. This voltage drop is the short circuit current threshold for the part. The
current that the ILIM pin sinks is dependent on the value of the resistor connected to RT and is given by:
0.69 V
I
+ 19.0
ILIM
R
T
(1)
The tolerance of the current sink is too loose to do an accurate current limit. The main purpose is for hard fault
protection of the power switches. Given the tolerance of the ILIM sink current, and the R range for a
DS(on)
MOSFET, it is generally possible to apply a load that thermally damages the converter. This device is intended
for embedded converters where load characteristics are defined and can be controlled. A small capacitor can
be added between ILIM and VDD for filtering. However, capacitors should not be used if the synchronization
function is to be used.
Soft-Start and Shutdown
The soft−start and shutdown functions are common to the SS/SD pin. The voltage at this pin over−rides the
reference voltage on the error amplifier during startup. This controls the output voltage slew rate and the surge
current required to charge the output capacitor at startup, allowing for a smooth startup with no overshoot of
the output voltage. Initial HDRV pulse widths during Soft−Start are typically very narrow, likely less than 150ns.
As a result, HDRV and LDRV can be on simultaneously, resulting in cross−conduction the MOSFETs of the
power stage. To minimize cross−conduction during soft−start, the soft−start time when the pulse widths are
less than 150ns should be kept to a minimum. A shutdown feature can be implemented by pulling SS/SD to
GND via a transistor as shown in Figure 5.
3.3 µA
6
SS/SD
C
SS
SHUTDOWN
TPS4002x
Figure 5. Shutdown Implementation
I
SS
C
+
t
(F)
SS
SS
V
FB
(2)
where
D t is the start up time in seconds
SS
Switching Frequency
The switching frequency is programmed by a resistor from RT to SGND. Nominal switching frequency can be
calculated by:
3
37.736 10
(
)
R (kW) +
* 5.09 kW
T
(
)
kHz
f
OSC
(3)
13
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Synchronization
The TPS4002x can be synchronized to an external reference frequency higher than the free running oscillator
frequency. The recommended method is to use a diode and a push pull drive signal as shown in Figure 6.
PREFERRED
VDD
ALTERNATE
VDD
Minumize
Output/Stray
Capacitance
on ILIM Node
TPS4002XPWP
TPS4002XPWP
1
ILIM/SYNC
VDD
1
2
ILIM/SYNC
VDD
2
50 ns to 100 ns
UDG−03032
50 ns to 100 ns
Figure 6. Synchronization Methods
This design allows synchronization up to the maximum operating frequency of 1 MHz. For best results the
nominal operating frequency of a converter that is to be synchronized should be kept as close as practicable
to the synchronization frequency to avoid excessive noise induced pulse width jitter. A good target is to shoot
for the free run frequency to be 80% of the synchronized frequency. This ensures that the synchronization
source is the frequency determining element in the system and not to adversely affect noise immunity.
Other methods of implementing the synchronization function include using an open collector or open drain
output device directly, or discreet devices to pull the ILIM/SYNC pin down. These do work but performance can
suffer at high frequency because the ILIM/SYNC pin must rise to (V
− 1.0 V) before the next switching cycle
DD
begins. Any time that this requires is directly subtracted from the maximum pulse width available and should
be considered when choosing devices to drive ILIM/SYNC. Consequently, the lowest output capacitance
devices work best.
During a synchronization cycle, the current sink on the ILIM/SYNC pin becomes disabled when ILIM/SYNC is
pulled below 1.0 V. The ILIM/SYNC current sink remains disabled until ILIM/SYNC reaches (V
−1.0 V) This
DD
removes the load on the ILIM/SYNC pin to allow the voltage to slew rapidly depending on the ILIM resistor and
any stray capacitance on the pin. To maximize this slew rate, minimize stray capacitance on this pin.
The duration of the synchronization pulse pulling ILIM/SYNC low shoud be between 50 ns and 100 ns. Longer
durations may limit the maximum obtainable duty cycle.
14
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APPLICATION INFORMATION
Transient Comparators and Power Good
The TPS4002x makes use of a separate pin, OSNS, to monitor output voltage for these two functions. In normal
operation, OSNS is connected to the output via a resistor divider. It is important to make this divider the same
ratio as the divider for the feedback network so that in normal operation the voltage at OSNS is the same as
the voltage at FB, 0.69 V nominal.
The PWRGD pin is an open drain output that is pulled low when the voltage at OSNS falls outside 0.69 V 4.6%
(approximately). A delay has been purposely built into the PWRGD pin pulling low in response to an out of band
voltage on OSNS, to minimize the need for filtering the signal in the event of a noise glitch causing a brief out
of band OSNS voltage. The PWRGD signal returns to high when the OSNS signal returns to approximately 1%
of nominal (0.69 V 1%).
The transient comparators override the conventional voltage control loop when the output voltage exceeds a
4.6% window. If the output transition is high (i.e. load steps down from 90% load to 10 % load) then the HDRV
gate drive is terminated, 0% duty cycle, the LDRV gate drive is turned on to sink output current until V
returns
OUT
to within 1% of nominal. Conversely when V
drops outside the window (i.e. step load increases from 10%
OUT
load to 90% load) HDRV increases to maximum duty cycle until V
Figure 7.)
returns to within 1% of nominal. (See
OUT
During start-up, the transient comparators control the state of PWRGD as previously described. However, the
operation of the gate drive outputs is not affected. (See Figure 8)
The transient comparators provide an improvement in load transient recovery time if used properly. In some
situations, recovery time may be one half of the time required without transient comparators. Keep in mind that
the transient comparator concept is a double-edged sword. While they provide improved transient recovery
time, they can also lead to instability if incorrectly applied. For proper functionality, design a feedback loop for
the converter that places the closed loop unity gain frequency at least five times higher than the 0 dB frequency
of the output L-C filter. If not, the feedback loop cannot respond to the ring of the L-C on a transient event. The
ring is likely to be large enough to disturb the transient comparators and the result is a power oscillator. Another
helpful action is to ground the feedback loop divider and the OSNS divider at the SGND pin. Make sure both
dividers measure the same physical location on the output bus. These help avoid problems with resistive drops
at higher loads causing problems.
Connecting OSNS to VDD disables the transient comparators. This also disables the PWRGD function.
Alternatively, OSNS and FB can be tied together. This connection allows a proper PWRGD at startup, though
transient performance diminishes.
< 10 µs
4.6%
1%
FB
−1%
− 4.6%
− 4.6%
10 µs
PWRGD
µ
500 n s
500 n s
10
s
SW
98 % Duty Cycle
0 % Duty Cycle
98 % Duty Cycle
0 % Duty Cycle
UDG−03181
Figure 7. Duty Cycle Waveforms
15
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V
DD
1.3 V
0.6 V
0.3 V
SS/SD
V
OUT
− 1%
500 ns
V
OUT
1.5 µs
Transient Comparators Enabled
4 µs
PWRGD
Transient Comparators Disabled
UDG−03181
Figure 8. Transient Comparator Waveforms
Layout Considerations
Successful operation of the TPS4002x family of controllers is dependent upon the proper converter layout and
grounding techniques. High current returns for the SR MOSFET’s source, input capacitance, output
capacitance, PVDD capacitance, and input bypass capacitors (if applicable), should be kept on a single ground
plane or wide trace connected to the PGND (pin10) through a short wide trace. Control components connected
to signal ground, as well as the PowerPad thermal pad, should be connected to a single ground plane connected
to SGND (Pin 8) through a short trace. SGND and PGND should be connected at a single point using a narrow
trace.
Proper operation of Predictive Gate Drive technology and I
functions are dependent upon detecting
ZERO
low-voltage thresholds on the SW node. To ensure that the signal at the SW pin accurately represents the
voltage at the main switching node, the connection from SW (pin 14) to the main switching node of the converter
should be kept as short and wide as possible and should ideally be kept on the top level with the power
components. If the SW trace must traverse multiple board layers between the TPS4002x and the main
switching node, multiple vias should be used to minimize the trace impedance.
Gate drive outputs, LDRV and HDRV (pins 11 and 15, respectively) should be kept as short as possible to
minimize inductances in the traces. If the gate drive outputs need to traverse multiple board layers multiple vias
should be used.
Charge pump components, BOOT1, BOOT2, PVDD, and any input bypass capacitors (if required), should be
kept as close as possible to their respective pins. Ceramic bypass capacitors should be used if the input
capacitors are located more than a couple of inches away from the TPS4002X. If a bypass capacitor is not
needed the trace from the input capacitors to VDD (pin2) should be kept as short and wide as possible to
minimize trace impedance. If multiple board layers are traversed multiple vias should be used.
16
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Manufacturer’s instructions should be followed for proper layout of the external MOSFETs. Thermal
impedances given in the manufacturer’s datasheets are for a given mounting technique with a specified surface
area under the drain of the MOSFET. PowerPad package information can be found in the APPLICATION
INFORMATION section of this datasheet.
Refer to TPS40021 EVM−001 High Efficiency Synchronous Buck Converter with PWM Controller Evaluation
Module (HPA009) User’s Guide, (Literature No. sluu144A) for a typical board layout.
The PowerPAD package provides low thermal impedance for heat removal from the device. The PowerPAD
derives its name and low thermal impedance from the large bonding pad on the bottom of the device. The circuit
board must have an area of solder-tinned-copper underneath the package. The dimensions of this area
depends on the size of the PowerPAD package. For a 16-pin TSSOP (PWP) package the area is
5 mm x 3.4 mm [3].
Thermal vias connect this area to internal or external copper planes and should have a drill diameter sufficiently
small so that the via hole is effectively plugged when the barrel of the via is plated with copper. This plug is
needed to prevent wicking the solder away from the interface between the package body and the solder-tinned
area under the device during solder reflow. Drill diameters of 0.33 mm (13 mils) works well when 1-oz copper
is plated at the surface of the board while simultaneously plating the barrel of the via. If the thermal vias are
not plugged when the copper plating is performed, then a solder mask material should be used to cap the vias
with a diameter equal to the via diameter of 0.1 mm minimum. This capping prevents the solder from being
wicked through the thermal vias and potentially creating a solder void under the package. Refer to PowerPAD
[3]
Thermally Enhanced Package for more information on the PowerPAD package.
17
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TYPICAL CHARACTERISTICS
OSCILLATOR FREQUENCY
MAXIMUM DUTY CYCLE
vs
vs
JUNCTION TEMPERATURE
JUNCTION TEMPERATURE
99
1000
950
900
850
800
750
700
650
600
550
500
450
98
97
R
T
= 35 kΩ
96
95
V
= 3.3 V, R = 69.8 kΩ
T
DD
R
T
= 69.8 kΩ
94
93
V
DD
= 5 V, R = 35 kΩ
T
−50
−25
0
25
50
75
100
125
−50
−25
0
25
50
75
100
125
T
J
− Junction Temperature − °C
T
J
− Junction Temperature − °C
Figure 10
Figure 9
SHUTDOWN SUPPLY CURRENT
vs
REFERENCE VOLTAGE
vs
JUNCTION TEMPERATURE
JUNCTION TEMPERATURE
0.700
0.675
697
695
693
0.650
0.625
0.600
0.575
691
689
687
685
683
0.550
0.525
0.500
−50
−25
0
25
50
75
100
125
−50
−25
0
25
50
75
100
125
T
J
− Junction Temperature − °C
T
J
− Junction Temperature − °C
Figure 11
Figure 12
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TYPICAL CHARACTERISTICS
SOFT-START CURRENT
vs
JUNCTION TEMPERATURE
TIMING RESISTANCE
vs
SWITCHING FREQUENCY
3.50
3.45
3.40
1000
900
800
700
600
V
IN
= 3.9 V
3.35
3.30
3.25
500
400
3.20
3.15
3.10
300
200
3.05
3.00
100
20
40
60
80
100
120
140
160
−50
−25
0
25
50
75
100
125
R
T
− Timing Resistance − kΩ
T
J
− Junction Temperature − °C
Figure 13
Figure 14
SHUTDOWN THRESHOLD VOLTAGE
ILIM OFFSET VOLTAGE
vs
JUNCTION TEMPERATURE
vs
JUNCTION TEMPERATURE
0.30
20
15
10
5
0.29
0.28
V
DD
= 2.0 V
Enable
0.27
0.26
0.25
0
Disable
0.24
0.23
V
= 3.2 V
DD
−5
−10
−15
0.22
0.21
0.20
V
= 4.9 V
DD
−20
−50
−50
−25
0
25
50
75
100
125
−25
0
25
50
75
100
125
T
J
− Junction Temperature − °C
T
J
− Junction Temperature − °C
Figure 15
Figure 16
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TYPICAL CHARACTERISTICS
ILIM SINK CURRENT
vs
JUNCTION TEMPERATURE
200
V
V
= 3.3 V
= 1.5 V
= 300 kHz
DD
OUT
V
R
= 3 V
DD
= 69.8 kΩ
V
OUT
198
196
T
f
SW
194
192
190
LDRV
SW
188
186
184
182
180
t − Time − 1 µs/div
−50
−25
0
25
50
75
100
125
T
J
− Junction Temperature − °C
Figure 18. TPS40020 Discontinuous Mode (DCM)
Figure 17
V
V
DD
= 3.3 V
DD
= 1.5 V
f
SW
= 300 kHz
SS/SD
V
OUT
V
OUT
LDRV
SW
SW
t − Time − 1 µs/div
t − Time − 20 ms/div
Figure 20. Output Current Fault Operation
Figure 19. TPS40020 I
Detection − DCM
ZERO
20
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TYPICAL CHARACTERISTICS
V
V
= 3.3 V
V
V
= 3.3 V
DD
DD
SS/SD
= 1.5 V
= 300 kHz
= 1.5 V
= 300 kHz
OUT
OUT
f
I
f
I
SW
SW
PVDD
= 5 A
= 5 A
LOAD
LOAD
SW
V
OUT
PWRGD
V
OUT
t − Time − 25 µs/div
t − Time − 1 ms/div
Figure 21. Start−Up Operation Without
Transient Comparators
Figure 22. PVDD Hysteresis
V
= 3.3 V
= 1.5 V
= 300 kHz
= 5 A
DD
V
V
= 3.3 V
DD
SD
V
OUT
= 1.5 V
= 300 kHz
OUT
f
SW
f
I
SW
I
LOAD
= 5 A
LOAD
SS/SD
COMP
SW
VOUT
PWRGD
t − Time − 1 ms/div
t − Time − 200 µs/div
Figure 23. Start−Up Operation With
Transient Comparators
Figure 24. COMP Shutdown Operation
21
Not Recommended for New Designs
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SLUS535D − MARCH 2003 − REVISED JULY 2007
TYPICAL CHARACTERISTICS
V
V
= 3.3 V
= 1.5 V
= 300 kHz
DD
OUT
V
V
f
= 3.3 V
DD
ILIM
SW
= 1.5 V
OUT
SYNC
f
SW
= 330 kHz
SS/SD
PWRGD
LDRV
t − Time − 500 ns/div
t − Time − 1 µs/div
Figure 25. PWRGD Shutdown Operation
Figure 26. External Synchronization
22
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REFERENCE DESIGN
This design used the TPS40020 PWM controller to facilitate a step-down application from 3.3-V to 1.5 V. (see
Figure 29) Design specifications include:
D Input voltage: 2.5 V ≤ V ≤ 5.0 V
IN
D Nominal output voltage: 3.3 V
D Output voltage V
D Output current I
: 1.5 V
OUT
: 20 A
OUT
D Switching frequency: 300 kHz
DESIGN PROCEDURE
Setting the Frequency
Choosing the optimum switching frequency is complicated. The higher the frequency, the smaller the
inductance and capacitance needed, so the smaller the size, but then the the switching losses are higher, the
efficiency is poorer. For this evaluation module, 300 kHz is chosen for reasonable efficiency and size.
A resistor R4, which is connected from pin 7 to ground, programs the oscillator frequency. The approximate
operating frequency is calculated in equation (3)
3
37.736 10
(
)
R (kW) +
* 5.09 kW
T
(
)
kHz
f
OSC
(4)
Using equation (2), R is calculated to be 120 kΩ and a 118-kΩ resistor is chosen for 300 kHz operation.
T
Inductance Value
The inductance value can be calculated by equation (2).
V
V
OUT
OUT
L
+
1 *
ǒ Ǔ
(min)
f I
V
RIPPLE
IN(max)
(5)
where I
losses.
is the ripple current flowing through the inductor, which affects the output voltage ripple and core
RIPPLE
Based on 24% ripple current and 300 kHz, the inductance value is calculated to 0.71 µH and a 0.75-µH inductor
(part number is CDEP149−0R7) is chosen. The DCR of this inductor is 1.1 mΩ and the loss is 440 mW, which
is approximately 1.5% of output power.
I
RIPPLE
C
+
OUT(min)
8 f V
RIPPLE
(6)
(7)
V
RIPPLE
RIPPLE
ESR
+
OUT
I
With 1.2% output voltage ripple, the needed capacitance is at least 109 µF and its ESR should be less than
3.75 mΩ. Three 2-V, 470-µF, POSCAP capacitors from Sanyo are used. The ESR is 10 mΩ each.
The required input capacitance is calculated in equation (5). The calculated value is approximately 390 µF for
a 100-mV input ripple. Three 6.0-V, 330-µF POSCAP capacitors with 10 mΩ ESR are used to handle 10 A of
RMS input current. Additionally, two ceramic capacitors are used to reduce the switching ripple current.
1
V
C
+ I
D
IN(min)
OUT(max)
(max)
f
OSC
IN(ripple)
(8)
23
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REFERENCE DESIGN
UDG−05002
Figure 27. Reference Design Schematic
24
Not Recommended for New Designs
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SLUS535D − MARCH 2003 − REVISED JULY 2007
REFERENCE DESIGN
Input and Output Capacitors
The output capacitance and its ESR needed are calculated in equations (5) and (6).
Compensation Design
Voltage-mode control is used in this evaluation module, using R2, R7, R8, C14, C15, and C16 to form a Type-III
compensator network. The L-C frequency of the power stage is approximately 4.9-kHz and the ESR-zero is
around 34 kHz. The overall crossover frequency, f
, is chosen at 43-kHz for reasonable transient response
0db
and stability. Two zeros f and f from the compensator are set at 2.4 kHz and 4 kHz. The two poles, f and
Z1
Z2
P1
f
are set at 34 kHz and 115 kHz. The frequency of poles and zeros are defined by the following equations:
P2
1
f
f
f
f
+
+
+
+
Z1
Z2
P1
P2
2p R7 C14
(9)
1
(assuming R2 ơ R8)
2p R2 C16
(10)
(11)
(12)
1
2p R8 C16
1
(assuming C14 ơ C15)
2p R7 C15
The transfer function for the compensator is calculated in equation (10).
(
)
[
(
)]
1 ) s ǒC14 R7 1 ) s C16 R2 ) R3
Ǔ) s R7 C15 1 ) s R8 C16
A(s) +
C15
C14
ƪ 1 )
ƫ
(
)
s R2 C14
(13)
Figure 28 shows the close loop gain and phase. The overall crossover frequency is approximately 30 kHz. The
phase margin is 57°.
OVERALL GAIN AND PHASE
vs
OSCILLATOR FREQUENCY
50
200
Gain
40
30
150
100
20
10
0
50
0
Phase
−10
−20
−30
−40
−50
−100
−50
100
−150
100 k
1 k
10 k
f
− Oscillator Frequency − kHz
OSC
Figure 28.
25
Not Recommended for New Designs
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SLUS535D − MARCH 2003 − REVISED JULY 2007
REFERENCE DESIGN
MOSFETs and Diodes
For a 1.5-V output voltage, the lower the R
of the MOSFET, the higher the efficiency. Due to the high
DS(on)
current and high conduction loss, the MOSFET should have very low conduction resistance (R
) and
DS(on)
thermal resistance. Si7858DP is chosen for its low R
(between 3 mΩ and 4 mΩ) and Power-Pak package.
DS(on)
Current Limiting
Resistor R3 sets the short current limit threshold. The R
of the upper MOSFET is used as a current sensor.
DS(on)
The current limit, I
is initialized at 30% above the maximum output current, I
, which is 28 A. Then
OUT(CL)
OUT(max)
R3 can be calculated in equation (11) and yields a value of 1.4 kΩ. An R3 of 1.43 kΩ is selected.
V
0.69 V
118 kW
FB
+ ǒ19
Ǔ+ 111.1 mA
(
)
+ ǒ Ǔ
I
19
LIM
R4
(14)
(15)
K R
I
DS(on)
I
OUT(CL)
1.5 0.004 28 A
(
)
R3 +
+
+ 1.4 kW
I
LIM
LIM
where
D R
is the on-resistor of Q1 (4 mΩ)
DS(on)
D Temperature coefficient, K=1.5
D V = 0.69 V
FB
D R4=118 kΩ
Voltage Sense Regulator
R1 and R2 operate as the output voltage divider. The error amplifier reference voltage (V ) is 0.69 V. The
FB
relationship between the output voltage and divider is described in equation (8). Using a 10-kΩ resistor for R2
and 1.5-V output regulation, R1 is calculated as 8.52 kΩ, 8.66 kΩ is selected for R1.
V
V
OUT
0.69 V
R1
1.5 V
R1 ) 10 kW
FB
+
³
+
³ R1 + 8.52 kW
R1
R1 ) R2
(16)
Transient Comparator
The output voltage transient comparators provide a quick response, first strike, approach to output voltage
transients. The output voltage is sensed through a resistor divider at the OSNS pin, using R5 and R6 shown
in Figure 27. If an overvoltage condition is detected, the HDRV gate drive is shut off and the LDRV gate drive
is turned on until the output is returned to regulation. Similarly, if an output undervoltage condition is sensed,
the HDRV gate drive goes to 95% duty cycle to pump the output back up quickly. The voltage divider should
be exactly the same as resistors R1 and R2 discussed previously. Resistor R5=8.66 kΩ and R6=10 kΩ in this
evaluation module.
26
Not Recommended for New Designs
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SLUS535D − MARCH 2003 − REVISED JULY 2007
REFERENCE DESIGN
TEST RESULTS
Efficiency Curves
The tested efficiency at different loads and input voltages are shown in Figure 29. The maximum efficiency is
as high as 93% at 1.5-V output. The efficiency is around 88% when the load current (I
) is 20 A.
LOAD
EFFICIENCY
vs
OUTPUT LOAD CURRENT
0.95
V
IN
= 2.5 V
0.90
0.85
V
IN
= 4.0 V
0.80
0.75
V
IN
= 5.0 V
V
IN
= 3.3 V
0.70
0
5
10
− Load Current − A
15
20
I
LOAD
Figure 29.
Typical Operating Waveforms
Typical operating waveforms are shown in Figure 30 and 31.
V
I
= 3.3 V
V
I
= 3.3 V
IN
IN
= 20 A
= 20 A
LOAD
LOAD
V
(10 mV/div)
OUTac
V
SW
(2 V/div)
t − Time − 1 µs/div
t − Time − 1 µs/div
Figure 31
Figure 30
27
Not Recommended for New Designs
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SLUS535D − MARCH 2003 − REVISED JULY 2007
REFERENCE DESIGN
Transient Response and Output Ripple Voltage
The output ripple is about 15 mV
voltage is about 35 mV.
at 20-A output. When the load changes from 4 A to 20 A, the overshooting
P−P
Figures 32 and 33 show the transient waveform with and without the transient comparator. Using the transient
comparator yields a settling time of 10-µs faster than without.
The output ripple is about 15 mV
at 20-A output which is shown in Figure 32. When the load changes from
P−P
0 A to 13 A, the overshoot voltage is approximately 80 mV, and the undershoot is is approximately 60 mV as
shown in Figure 34. When the transient comparator is triggered, the powergood (PWRGD) signal goes low.
t − Time − 200 µs/div
V
IN
= 3.3 V
WIthout Transient
Comparator
WIth Transient
Comparator
WIth Transient
Comparator
WIthout Transient
Comparator
I
(10 A/div)
OUT
I
(10 A/div)
OUT
t − Time − 10 µs/div
t − Time − 10 µs/div
Figure 32. Transient Response Undershoot
Figure 33. Transient Response Overshoot
28
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REFERENCE DESIGN
V
PWRGD
(2.5 V/div)
V
(100 mV/div)
OUTac
I
(10 A/div)
OUT
t − Time − 200 µs/div
Figure 34. Transient Response
29
PACKAGE OPTION ADDENDUM
www.ti.com
12-Jun-2013
PACKAGING INFORMATION
Orderable Device
TPS40020PWP
Status Package Type Package Pins Package
Eco Plan Lead/Ball Finish
MSL Peak Temp
Op Temp (°C)
-40 to 85
-40 to 85
-40 to 85
-40 to 85
-40 to 85
-40 to 85
-40 to 85
-40 to 85
Device Marking
Samples
Drawing
Qty
(1)
(2)
(3)
(4/5)
NRND
HTSSOP
HTSSOP
HTSSOP
HTSSOP
HTSSOP
HTSSOP
HTSSOP
HTSSOP
PWP
16
16
16
16
16
16
16
16
90
Green (RoHS
& no Sb/Br)
CU NIPDAU
CU NIPDAU
CU NIPDAU
CU NIPDAU
CU NIPDAU
CU NIPDAU
CU NIPDAU
CU NIPDAU
Level-2-260C-1 YEAR
Level-2-260C-1 YEAR
Level-2-260C-1 YEAR
Level-2-260C-1 YEAR
Level-2-260C-1 YEAR
Level-2-260C-1 YEAR
Level-2-260C-1 YEAR
Level-2-260C-1 YEAR
40020
40020
40020
40020
40021
40021
40021
40021
TPS40020PWPG4
TPS40020PWPR
TPS40020PWPRG4
TPS40021PWP
NRND
NRND
NRND
NRND
NRND
NRND
NRND
PWP
PWP
PWP
PWP
PWP
PWP
PWP
90
2000
2000
90
Green (RoHS
& no Sb/Br)
Green (RoHS
& no Sb/Br)
Green (RoHS
& no Sb/Br)
Green (RoHS
& no Sb/Br)
TPS40021PWPG4
TPS40021PWPR
TPS40021PWPRG4
90
Green (RoHS
& no Sb/Br)
2000
2000
Green (RoHS
& no Sb/Br)
Green (RoHS
& no Sb/Br)
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
Addendum-Page 1
PACKAGE OPTION ADDENDUM
www.ti.com
12-Jun-2013
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
OTHER QUALIFIED VERSIONS OF TPS40021 :
Enhanced Product: TPS40021-EP
•
NOTE: Qualified Version Definitions:
Enhanced Product - Supports Defense, Aerospace and Medical Applications
•
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
14-Jul-2012
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
B0
K0
P1
W
Pin1
Diameter Width (mm) (mm) (mm) (mm) (mm) Quadrant
(mm) W1 (mm)
TPS40020PWPR
TPS40021PWPR
HTSSOP PWP
HTSSOP PWP
16
16
2000
2000
330.0
330.0
12.4
12.4
6.9
6.9
5.6
5.6
1.6
1.6
8.0
8.0
12.0
12.0
Q1
Q1
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
14-Jul-2012
*All dimensions are nominal
Device
Package Type Package Drawing Pins
SPQ
Length (mm) Width (mm) Height (mm)
TPS40020PWPR
TPS40021PWPR
HTSSOP
HTSSOP
PWP
PWP
16
16
2000
2000
367.0
367.0
367.0
367.0
35.0
35.0
Pack Materials-Page 2
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LOW PIN COUNT, LOW VIN (2.5 V TO 5.5 V) SYNCHRONOUS BUCK DC-TO-DC CONTROLLER WITH ENABLE
TI
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