TPS51116-EP [TI]
COMPLETE DDR, DDR2, DDR3, AND LPDDR3 MEMORY POWER SOLUTION SYNCHRONOUS BUCK CONTROLLER, 1-A LDO, BUFFERED REFERENCE;
| 型号： | TPS51116-EP |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | COMPLETE DDR, DDR2, DDR3, AND LPDDR3 MEMORY POWER SOLUTION SYNCHRONOUS BUCK CONTROLLER, 1-A LDO, BUFFERED REFERENCE 双倍数据速率 光电二极管 |
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TPS51116
www.ti.com
SLUS609A–MAY 2004–REVISED JUNE 2004
COMPLETE DDR AND DDR2 MEMORY POWER SOLUTION
SYNCHRONOUS BUCK CONTROLLER, 3-A LDO, BUFFERED REFERENCE
FEATURES
DESCRIPTION
•
Synchronous Buck Controller (VDDQ)
The TPS51116 provides a complete power supply for
both DDR/SSTL-2 and DDR2/SSTL-18 memory sys-
tems. It integrates a synchronous buck controller with
a 3-A sink/source tracking linear regulator and
buffered low noise reference. The TPS51116 offers
the lowest total solution cost in systems where space
is at a premium. The TPS51116 synchronous control-
ler runs fixed 400kHz pseudo-constant frequency
PWM with an adaptive on-time control that can be
configured in D-CAP™ Mode for ease of use and
fastest transient response or in current mode to
support ceramic output capacitors. The 3-A
sink/source LDO maintains fast transient response
only requiring 20-µF (2 × 10 µF) of ceramic output
capacitance. In addition, the LDO supply input is
available externally to significantly reduce the total
power losses. The TPS51116 supports all of the
sleep state controls placing VTT at high-Z in S3
(suspend to RAM) and discharging VDDQ, VTT and
VTTREF (soft-off) in S4/S5 (suspend to disk). The
TPS51116 has all of the protection features including
thermal shutdown and is in a 20-pin HTSSOP
PowerPAD™ package.
– Wide-Input Voltage Range: 3.0-V to 28-V
– D–CAP™ Mode with 100-ns Load Step Re-
sponse
– Current Mode Option Supports Ceramic
Output Capacitors
– Supports Soft-Off in S4/S5 States
– Current Sensing from RDS(on) or Resistor
– 2.5-V (DDR), 1.8-V (DDR2) or Adjustable
Output (1.5-V to 3.0-V)
– Equipped with Powergood, Overvoltage Pro-
tection and Undervoltage Protection
•
3-A LDO (VTT), Buffered Reference (VREF)
– Capable to Sink and Source 3 A
– LDO Input Available to Optimize Power
Losses
– Requires only 20-µF Ceramic Output Ca-
pacitor
– Buffered Low Noise 10-mA Output
– Accuracy ±20 mV for both VREF and VTT
– Supports High-Z in S3 and Soft-Off in S4/S5
– Thermal Shutdown
APPLICATIONS
•
DDR/DDR2 Memory Power Supplies
•
SSTL-2 SSTL-18 and HSTL Termination
TYPICAL APPLICATION
(DDR2)
C1
Ceramic
V
IN
TPS51116PWP
C5
Ceramic
2 × 10 µF
1
2
3
VLDOIN VBST 20
L1
1 µH
VTT
0.9 V
2 A
0.1 µF
VTT
DRVH 19
VDDQ
1.8 V
10 A
VTTGND
LL 18
GND
C8
SP−CAP
2 × 150 µF
4
5
6
7
8
9
VTTSNS DRVL 17
C3
Ceramic
GND
PGND 16
CS 15
2 × 10 µF
R1
C4
MODE
VTTREF
Ceramic
0.033 µF
V5IN 14
VREF
5V_IN
0.9 V
R2
10 mA
COMP PGOOD 13
VDDQSNS S5 12
C2
PGOOD
S5
Ceramic
4.7 µF
GND
10 VDDQSET S3 11
S3
UDG−04058
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2004, Texas Instruments Incorporated
TPS51116
www.ti.com
SLUS609A–MAY 2004–REVISED JUNE 2004
These devices have limited built-in ESD protection. The leads should be shorted together or the device
placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates.
ORDERING INFORMATION
TA
PLASTIC HTTSOP PowerPAD (PWP)(1)
-40°C to 85°C
TPS51116PWP
(1) The PWP package is also available taped and reeled. Add an R
suffix to the device type (TPS51116PWPR)
ABSOLUTE MAXIMUM RATINGS(1)
over operating free-air temperature range unless otherwise noted
TPS51116
-0.3 to 36
-0.3 to 6
UNITS
VBST
VBST wrt LL
VIN
Input voltage range
V
CS, MODE, S3, S5, VTTSNS, VDDQSNS, V5IN, VLDOIN, VDDQSET
-0.3 to 6
PGND, VTTGND
DRVH
-0.3 to 0.3
-1.0 to 36
-1.0 to 30
-0.3 to 6
VOUT Output voltage range LL
COMP, DRVL, PGOOD, VTT, VTTREF
Operating ambient temperature range
Storage temperature
V
TA
-40 to 85
-55 to 150
°C
Tstg
(1) Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under "recommended operating
conditions" is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. All voltage
values are with respect to the network ground terminal unless otherwise noted.
DISSIPATION RATINGS
TA < 25°C POWER
DERATING FACTOR
ABOVE TA = 25°C
PACKAGE
TA = 85°C POWER RATING
RATING
20-pin PWP
2.53 W
25.3 mW/°C
1.01 W
RECOMMENDED OPERATING CONDITIONS
MIN
4.75
-0.1
-0.6
-0.1
-0.1
-0.1
MAX
5.25
34
UNIT
Supply voltage, V5IN
V
VBST, DRVH
LL
28
VLDOIN, VTT, VTTSNS, VDDQSNS
VTTREF
3.6
1.8
0.1
Voltage range
V
PGND, VTTGND
S3, S5, MODE, VDDQSET, CS, COMP, PGOOD,
DRVL
-0.1
-40
5.25
85
Operating free-air temperature, TA
°C
2
TPS51116
www.ti.com
SLUS609A–MAY 2004–REVISED JUNE 2004
ELECTRICAL CHARACTERISTICS
over operating free-air temperature range, VV5IN = 5 V, VLDOIN is connected to VDDQ output (unless otherwise noted)
PARAMETER
SUPPLY CURRENT
TEST CONDITIONS
MIN
TYP
MAX
UNIT
TA = 25°C, No load, VS3 = VS5 = 5 V,
COMP connected to capacitor
IV5IN1
IV5IN2
IV5IN3
Supply current 1, V5IN
Supply current 2, V5IN
Supply current 3, V5IN
0.8
300
240
2
600
500
mA
TA = 25°C, No load, VS3 = 0 V, VS5 = 5 V,
COMP connected to capacitor
TA = 25°C, No load, VS3 = 0 V, VS5 = 5 V,
VCOMP = 5 V
µA
IV5INSDN
IVLDOIN1
IVLDOIN2
IVLDOINSDN
Shutdown current, V5IN
Supply current 1, VLDOIN
Supply current 2, VLDOIN
Standby current, VLDOIN
TA = 25°C, No load, VS3 = VS5 = 0 V
TA = 25°C, No load, VS3 = VS5 = 5 V
TA = 25°C, No load, VS3 = 5 V, VS5 = 0 V,
TA = 25°C, No load, VS3 = VS5 = 0 V
0.1
1
1.0
10
0.1
0.1
10
1.0
VTTREF OUTPUT
VVTTREF
Output voltage, VTTREF
VVDDQSNS/2
V
-10 mA < IVTTREF < 10 mA, VVDDQSNS = 2.5 V,
Tolerance to VVDDQSNS/2
-20
-18
20
18
VVTTREFTOL
Output voltage tolerance
mV
-10 mA < IVTTREF < 10 mA, VVDDQSNS = 1.8 V,
Tolerance to VVDDQSNS/2
VVTTREFSRC
VVTTREFSNK
VTT OUTPUT
Source current
Sink current
VVDDQSNS = 2.5 V, VVTTREF = 0 V
VVDDQSNS = 2.5 V, VVTTREF = 2.5 V
-20
20
-40
40
-80
80
mA
V
VS3 = VS5 = 5 V, VVLDOIN = VVDDQSNS = 2.5 V
VS3 = VS5 = 5 V, VVLDOIN = VVDDQSNS = 1.8 V
VS3 = VS5 = 5 V, IVTT = 0 A
1.25
0.9
VVTTSNS
Output voltage, VTT
-20
-30
-40
-20
-30
-40
20
30
40
20
30
40
VTT output voltage tolerance
to VTTREF
VVTTTOL25
VS3 = VS5 = 5 V, |IVTT| = 0 A < 1.5 A
VS3 = VS5 = 5 V, |IVTT| = 0 A < 3 A
VS3 = VS5 = 5 V, IVTT = 0 A
mV
VTT output voltage tolerance
to VTTREF
VVTTTOL18
VS3 = VS5 = 5 V, |IVTT| = 0 A < 1 A
VS3 = VS5 = 5 V, |IVTT| = 0 A < 2 A
VVLDOIN = VVDDQSNS = 2.5 V, VVTT = VVTTSNS
1.19 V, PGOOD = HI
=
3.0
1.5
3.0
3.8
2.2
3.6
2.2
6.0
3.0
6.0
IVTTTOCLSRC
Source current limit, VTT
Sink current limit, VTT
VVLDOIN = VVDDQSNS = 2.5 V, VVTT = 0 V
A
VVLDOIN = VVDDQSNS = 2.5 V, VVTT = VVTTSNS
1.31 V, PGOOD = HI
=
IVTTTOCLSNK
VVLDOIN = VVDDQSNS = 2.5 V, VVTT = VVDDQ
VS3 = 0 V, VS5 = 5 V, VVTT = VVDDQSNS /2
VS3 = 5 V, VVTTSNS = VVDDQSNS /2
1.5
-10
-1
3.0
10
1
IVTTLK
Leakage current, VTT
IVTTBIAS
IVTTSNSLK
Input bias current, VTTSNS
Leakage current, VTTSNS
-0.1
17
µA
VS3 = 0 V, VS5 = 5 V, VVTT = VVDDQSNS /2
-1
1
TA = 25°C, VS3 = VS5 = VVDDQSNS = 0 V,
VVTT = 0.5 V
IVTTDisch
Discharge current, VTT
10
mA
VDDQ OUTPUT
TA = 25°C, VVDDQSET = 0 V, No load
0°C ≤ TA≤ 85°C, VVDDQSET = 0 V, No load(1)
2.465
2.457
2.440
1.776
1.769
1.764
1.5
2.500
2.500
2.500
1.800
1.800
1.800
2.535
2.543
2.550
1.824
1.831
1.836
3.0
(1)
-40°C ≤ TA≤ 85°C, VVDDQSET = 0 V, No load
TA = 25°C, VVDDQSET = 5 V, No load(1)
0°C ≤ TA≤ 85°C, VVDDQSET = 5V, No load(1)
-40°C ≤ TA≤ 85°C, VVDDQSET = 5V, No load(1)
-40°C ≤ TA≤ 85°C, Adjustable mode, No load(1)
VVDDQ
Output voltage, VDDQ
V
(1) Ensured by design. Not production tested.
3
TPS51116
www.ti.com
SLUS609A–MAY 2004–REVISED JUNE 2004
ELECTRICAL CHARACTERISTICS (continued)
over operating free-air temperature range, VV5IN = 5 V, VLDOIN is connected to VDDQ output (unless otherwise noted)
PARAMETER
TEST CONDITIONS
TA = 25°C, Adjustable mode
MIN
742.5
740.2
738.0
TYP
750.0
750.0
750.0
215
MAX
757.5
759.8
762.0
UNIT
VVDDQSET
VDDQSET regulation voltage 0°C ≤ TA≤ 85°C, Adjustable mode
-40°C ≤ TA≤ 85°C, Adjustable mode
VVDDQSET = 0 V
mV
RVDDQSNS
Input impedance, VDDQSNS VVDDQSET = 5 V
Adjustable mode
180
kΩ
460
VVDDQSET = 0.78 V, COMP = Open
Input current, VDDQSET
-0.04
-0.06
IVDDQSET
µA
VVDDQSET = 0.78 V, COMP = 5 V
VS3 = VS5 = 0 V, VVDDQSNS = 0.5 V,
IVDDQDisch
Discharge current, VDDQ
VMODE = 0 V
10
40
mA
VS3 = VS5 = 0 V, VVDDQSNS = 0.5 V,
Discharge current, VLDOIN
IVDDOINDisch
700
VMODE = 0.5 V
TRANSCONDUCTANCE AMPLIFIER
gm
Gain
TA = 25°C
240
300
13
360
µS
µA
COMP maximum sink cur-
rent
VS3 = 0 V, VS5 = 5 V, VVDDQSET = 0 V,
VVDDQSNS = 2.7 V, VCOMP = 1.28 V
ICOMPSNK
COMP maximum source cur- VS3 = 0 V, VS5 = 5 V, VVDDQSET = 0 V,
rent
ICOMPSRC
VCOMPHI
VCOMPLO
-13
1.34
1.21
VVDDQSNS = 2.3 V, VCOMP = 1.28 V
VS3 = 0 V, VS5 = 5 V, VVDDQSET = 0 V,
VVDDQSNS = 2.3 V, VCS = 0 V
COMP high clamp voltage
1.31
1.18
1.37
1.24
V
VS3 = 0 V, VS5 = 5 V, VVDDQSET = 0 V,
VVDDQSNS = 2.7 V, VCS = 0 V
COMP low clamp voltage
DUTY CONTROL
TON
Operating on-time
VIN = 12 V, VVDDQSET = 0 V
VIN = 12 V, VVDDQSNS = 0 V
520
125
100
350
TON0
Startup on-time
Minimum on-time
Minimum off-time
ns
(2)
TON(min)
TOFF(min)
TA = 25°C
TA = 25°C(2)
OUTPUT DRIVERS
Source, IDRVH = -100 mA
Sink, IDRVH = 100 mA
Source, IDRVL = -100 mA
Sink, IDRVL = 100 mA
3
0.9
3
6
3
6
3
RDRVH
RDRVL
TD
DRVH resistance
Ω
DRVL resistance
Dead time
0.9
10
20
(2)
LL-low to DRVL-on
ns
DRVL-off to DRVH-on(2)
INTERNAL BST DIODE
VFBST
Forward voltage
VV5IN-VBST , IF = 10 mA, TA = 25°C
0.7
0.8
0.1
0.9
1.0
V
VVBST = 34 V, VLL = 28 V, VVDDQ = 2.6 V,
TA = 25°C
IVBSTLK
VBST leakage current
µA
(2) Ensured by design. Not production tested.
4
TPS51116
www.ti.com
SLUS609A–MAY 2004–REVISED JUNE 2004
ELECTRICAL CHARACTERISTICS (continued)
over operating free-air temperature range, VV5IN = 5 V, VLDOIN is connected to VDDQ output (unless otherwise noted)
PARAMETER
PROTECTIONS
TEST CONDITIONS
MIN
TYP
MAX
UNIT
VPGND-CS , PGOOD = HI, VCS < 0.5 V
VPGND-CS , PGOOD = LO, VCS < 0.5 V
TA = 25°C, VCS > 4.5 V, PGOOD = HI
TA = 25°C, VCS > 4.5 V, PGOOD = LO
50
20
9
60
30
10
5
70
40
11
6
VOCL
Current limit threshold
mV
ITRIP
Current sense sink current
µA
4
TRIP current temperature co- RDS(on) sense scheme, On the basis
TCITRIP
VOCL(off)
VR(trip)
4500
0
ppm/°C
(3)
efficient
of TA = 25°C
Overcurrent protection
COMP offset
(VV5IN-CS - VPGND-LL), VV5IN-CS = 60 mV,
VCS > 4.5 V
-5
5
mV
Current limit threshold setting
range
(3)
VV5IN-CS
30
150
POWERGOOD COMPARATOR
PG in from lower
93%
95%
105%
5%
97%
VTVDDQPG
VDDQ powergood threshold PG in from higher
103%
107%
PG hysteresis
IPG(max)
TPG(del)
PGOOD sink current
PGOOD delay time
VVTT = 0 V, VPGOOD = 0.5 V
Delay for PG in
2.5
80
7.5
mA
µs
130
200
UNDERVOLTAGE LOCKOUT/LOGIC THRESHOLD
Wake up
3.7
0.2
4.7
4.0
0.3
4.3
0.4
V5IN UVLO threshold volt-
age
VUVV5IN
Hysteresis
No discharge
Non-tracking discharge
2.5 V output
1.8 V output
S3, S5
VTHMODE
MODE threshold
0.1
0.25
4.5
0.08
3.5
0.15
4.0
V
VTHVDDQSET
VDDQSET threshold voltage
VIH
High-level input voltage
Low-level input voltage
Hysteresis voltage
2.2
VIL
S3, S5
0.3
VIHYST
VINLEAK
VINVDDQSET
S3, S5
0.2
Logic input leakage current
Input leakage/ bias current
S3, S5, MODE
VDDQSET
-1
-1
1
1
µA
µs
UNDERVOLTAGE AND OVERVOLTAGE PROTECTION
OVP detect
Hysteresis
110%
115%
5%
120%
VDDQ OVP trip threshold
voltage
VOVP
VDDQ OVP propagation de-
lay
TOVPDEL
VUVP
TUVPDEL
TUVPEN
1.5
(3)
UVP detect
Hysteresis
70%
10%
Output UVP trip threshold
Output UVP propagation de-
lay(3)
Output UVP enable delay(3)
32
cycle
1007
THERMAL SHUTDOWN
Shutdown temperature
Hysteresis
160
10
(3)
TSDN Thermal SDN threshold
°C
(3) Ensured by design. Not production tested.
5
TPS51116
www.ti.com
SLUS609A–MAY 2004–REVISED JUNE 2004
DEVICE INFORMATION
PWP PACKAGE
(TOP VIEW)
1
2
3
4
5
6
7
8
9
10
20
19
18
17
16
15
14
13
12
11
VLDOIN
VTT
VTTGND
VTTSNS
GND
MODE
VTTREF
COMP
VBST
DRVH
LL
DRVL
PGND
CS
V5IN
PGOOD
S5
VDDQSNS
VDDQSET
S3
TERMINAL FUNCTIONS
TERMINAL
I/O
DESCRIPTION
NAME
NO.
Output of the transconductance amplifier for phase compensation. Connect to V5IN to disable
gm amplifier and use D-CAP™ mode.
COMP
8
I/O
I/O
Current sense comparator input (-) for resistor current sense scheme. Or over current trip voltage
setting input for RDS(on) current sense scheme if connected to V5IN through the voltage setting
resistor.
CS
15
DRVH
DRVL
GND
19
17
5
O
O
-
Switching (top) MOSFET gate drive output.
Rectifying (bottom) MOSFET gate drive output.
Signal ground. Connect to minus terminal of the VTT LDO output capacitor.
Switching (top) MOSFET gate driver return. Current sense comparator input (-) for RDS(on) current
sense.
LL
18
I/O
MODE
PGND
6
I
Discharge mode setting pin. See VDDQ and VTT Discharge Control section.
16
-
Ground for rectifying (bottom) MOSFET gate driver. Also current sense comparator input (+).
Powergood signal open drain output, In HIGH state when VDDQ output voltage is within the
target range.
PGOOD
13
O
S3
11
12
14
20
10
I
S3 signal input.
S5
I
S5 signal input.
V5IN
VBST
VDDQSET
I
I/O
I
5-V power supply input for internal circuits.
Switching (top) MOSFET driver bootstrap voltage input.
VDDQ output voltage setting pin. See VDDQ Output Voltage Selection section.
VDDQ reference input for VTT and VTTREF. Power supply for the VTTREF. Discharge current
sinking terminal for VDDQ Non-tracking discharge. Output voltage feedback input for VDDQ
output if VDDQSET pin is connected to V5IN or GND.
VDDQSNS
9
I/O
VLDOIN
VTT
1
2
3
7
4
I
O
-
Power supply for the VTT LDO.
Power output for the VTT LDO.
VTTGND
VTTREF
VTTSNS
Power ground output for the VTT LDO.
O
I
VTTREF buffered reference output.
Voltage sense input for the VTT LDO. Connect to plus terminal of the VTT LDO output capacitor.
6
TPS51116
www.ti.com
SLUS609A–MAY 2004–REVISED JUNE 2004
FUNCTIONAL BLOCK DIAGRAM
7
TPS51116
www.ti.com
SLUS609A–MAY 2004–REVISED JUNE 2004
DETAILED DESCRIPTION
The TPS51116 is an integrated power management solution which combines a synchronous buck controller, a
10-mA buffered reference and a high-current sink/source low-dropout linear regulator (LDO) in a small 20-pin
HTSSOP package. Each of these rails generates VDDQ, VTTREF and VTT that required with DDR/DDR2
memory systems. The switch mode power supply (SMPS) portion employs external N-channel MOSFETs to
support high current for DDR/DDR2 memory’s VDD/VDDQ. The output voltage is preset and selectable from 2.5
V or 1.8 V. User defined output voltage is also possible and can be adjustable from 1.5 V to 3 V. Input voltage
range of the SMPS is 3 V to 28 V. The SMPS runs an adaptive on-time PWM operation at high-load condition
and automatically reduces frequency to keep excellent efficiency down to several mA. Current sensing scheme
uses either RDS(on) of the external rectifying MOSFET for a low-cost, loss-less solution, or an optional sense
resistor placed in series to the rectifying MOSFET for more accurate current limit. The output of the switcher is
sensed by VDDQSNS pin to generate one-half VDDQ for the 10-mA buffered reference (VTTREF) and the VTT
active termination supply. The VTT LDO can source and sink up to 3-A peak current with only 20-µF (two 10 µF
in parallel) ceramic output capacitors. VTTREF tracks VDDQ/2 within ±20 mV. VTT output tracks VTTREF within
±20 mV at no load condition while ±40 mV at full load. The LDO input can be separated from VDDQ and
optionally connected to a lower voltage by using VLDOIN pin. This helps reducing power dissipation in sourcing
phase. TheTPS51116 is fully compatible to JEDEC DDR/DDR2 specifications at S3/S5 sleep state (see Table 2).
The part has two options of output discharge function when both VTT and VDDQ are disabled. The tracking
discharge mode discharges VDDQ and VTT outputs through the internal LDO transistors and then VTT output
tracks half of VDDQ voltage during discharge. The non-tracking discharge mode discharges outputs using
internal discharge MOSFETs which are connected to VDDQSNS and VTT. The current capability of these
discharge FETs are limited and discharge occurs more slowly than the tracking discharge. These discharge
functions can be disabled by selecting non-discharge mode.
VDDQ SMPS, Dual PWM Operation Modes
The main control loop of the SMPS is designed as an adaptive on-time pulse width modulation (PWM) controller.
It supports two control schemes which are a current mode and a proprietary D-CAP™ mode. D-CAP™ mode
uses internal compensation circuit and is suitable for low external component count configuration with an
appropriate amount of ESR at the output capacitor(s). Current mode control has more flexibility, using external
compensation network, and can be used to achieve stable operation with very low ESR capacitor(s) such as
ceramic or specialty polymer capacitors.
These control modes are selected by the COMP terminal connection. If the COMP pin is connected to V5IN,
TPS51116 works in the D-CAP™ mode, otherwise it works in the current mode. VDDQ output voltage is
monitored at a feedback point voltage. If VDDQSET is connected to V5IN or GND, this feedback point is the
output of the internal resistor divider inside VDDQSNS pin. If an external resistor divider is connected to
VDDQSET pin, VDDQSET pin itself becomes the feedback point (see VDDQ Output Voltage Selection section).
At the beginning of each cycle, the synchronous top MOSFET is turned on, or becomes ON state. This MOSFET
is turned off, or becomes OFF state, after internal one shot timer expires. This one shot is determined by VIN and
VOUT to keep frequency fairly constant over input voltage range, hence it is called adaptive on-time control (see
PWM Frequency and Adaptive On-Time Control section). The MOSFET is turned on again when feedback
information indicates insufficient output voltage and inductor current information indicates below the over current
limit. Repeating operation in this manner, the controller regulates the output voltage. The synchronous bottom or
the rectifying MOSFET is turned on each OFF state to keep the conduction loss minimum. The rectifying
MOSFET is turned off when inductor current information detects zero level. This enables seamless transition to
the reduced frequency operation at light load condition so that high efficiency is kept over broad range of load
current.
In the current mode control scheme, the transconductance amplifier generates a target current level
corresponding to the voltage difference between the feedback point and the internal 750 mV reference. During
the OFF state, the PWM comparator monitors the inductor current signal as well as this target current level, and
when the inductor current signal comes lower than the target current level, the comparator provides SET signal
to initiate the next ON state. The voltage feedback gain is adjustable outside the controller device to support
various types of output MOSFETs and capacitors. In the D-CAP™ mode, the transconductance amplifier is
disabled and the PWM comparator compares the feedback point voltage and the internal 750 mV reference
during the OFF state. When the feedback point comes lower than the reference voltage, the comparator provides
SET signal to initiate the next ON state.
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TPS51116
www.ti.com
SLUS609A–MAY 2004–REVISED JUNE 2004
DETAILED DESCRIPTION (continued)
VDDQ SMPS, Light Load Condition
TPS51116 automatically reduces switching frequency at light load condition to maintain high efficiency. This
reduction of frequency is achieved smoothly and without increase of VOUTripple or load regulation. Detail
operation is described as follows. As the output current decreases from heavy load condition, the inductor current
is also reduced and eventually comes to the point that its valley touches zero current, which is the boundary
between continuous conduction and discontinuous conduction modes. The rectifying MOSFET is turned off when
this zero inductor current is detected. As the load current further decreased, the converter runs in discontinuous
conduction mode and it takes longer and longer to discharge the output capacitor to the level that requires next
ON cycle. The ON-time is kept the same as that in the heavy load condition. In reverse, when the output current
increase from light load to heavy load, switching frequency increases to the constant 400 kHz as the inductor
current reaches to the continuous conduction. The transition load point to the light load operation IOUT(LL) (i.e. the
threshold between continuous and discontinuous conduction mode) can be calculated in Equation 1:
(V
V
)
V
IN
OUT
OUT
1
I
+
OUT(LL)
2 L f
V
IN
(1)
where
•
f is the PWM switching frequency (400 kHz)
Switching frequency versus output current in the light load condition is a function of L, f, VIN and VOUT, but it
decreases almost proportional to the output current from the IOUT(LL) given above. For example, it is 40 kHz at
IOUT(LL)/10 and 4 kHz at IOUT(LL)/100.
Low-Side Driver
The low-side driver is designed to drive high-current, low-RDS(on), N-channel MOSFET(s). The drive capability is
represented by its internal resistance, which are 3 Ω for V5IN to DRVL and 0.9 Ω for DRVL to PGND. A
dead-time to prevent shoot through is internally generated between top MOSFET off to bottom MOSFET on, and
bottom MOSFET off to top MOSFET on. 5-V bias voltage is delivered from V5IN supply. The instantaneous drive
current is supplied by an input capacitor connected between V5IN and GND. The average drive current is equal
to the gate charge at VGS = 5 V times switching frequency. This gate drive current as well as the high-side gate
drive current times 5 V makes the driving power which needs to be dissipated from TPS51116 package.
High-Side Driver
The high-side driver is designed to drive high-current, low-RDS(on) N-channel MOSFET(s). When configured as a
floating driver, 5-V bias voltage is delivered from V5IN supply. The average drive current is also calculated by the
gate charge at VGS = 5V times switching frequency. The instantaneous drive current is supplied by the flying
capacitor between VBST and LL pins. The drive capability is represented by its internal resistance, which are 3 Ω
for VBST to DRVH and 0.9 Ω for DRVH to LL.
Current Sensing Scheme
In order to provide both good accuracy and cost effective solution, TPS51116 supports both of external resistor
sensing and MOSFET RDS(on) sensing. For resistor sensing scheme, an appropriate current sensing resistor
should be connected between the source terminal of the bottom MOSFET and PGND. CS pin is connected to the
MOSFET source terminal node. The inductor current is monitored by the voltage between PGND pin and CS pin.
For RDS(on) sensing scheme, CS pin should be connected to V5IN through the trip voltage setting resistor, RTRIP
.
In this scheme, CS terminal sinks 10-µA ITRIP current and the trip level is set to the voltage across the RTRIP. The
inductor current is monitored by the voltage between PGND pin and LL pin so that LL pin should be connected to
the drain terminal of the bottom MOSFET. ITRIP has 4500ppm/°C temperature slope to compensate the
temperature dependency of the RDS(on). In either scheme, PGND is used as the positive current sensing node so
that PGND should be connected to the proper current sensing device, i.e. the sense resistor or the source
terminal of the bottom MOSFET.
9
TPS51116
www.ti.com
SLUS609A–MAY 2004–REVISED JUNE 2004
DETAILED DESCRIPTION (continued)
PWM Frequency and Adaptive On-Time Control
TPS51116 employs adaptive on-time control scheme and does not have a dedicated oscillator on board.
However, the device runs with fixed 400-kHz pseudo-constant frequency by feed-forwarding the input and output
voltage into the on-time one-shot timer. The on-time is controlled inverse proportional to the input voltage and
proportional to the output voltage so that the duty ratio is kept as VOUT/VIN technically with the same cycle time.
Although the TPS51116 does not have a pin connected to VIN, the input voltage is monitored at LL pin during
the ON state. This helps pin count reduction to make the part compact without sacrificing its performance. In
order to secure minimum ON-time during startup, feed-forward from the output voltage is enabled after the output
becomes 750 mV or larger.
VDDQ Output Voltage Selection
TPS51116 can be used for both of DDR (VVDDQ = 2.5 V) and DDR2 (VVDDQ = 1.8 V) power supply and adjustable
output voltage (1.5 V < VVDDQ < 3 V) by connecting VDDQSET pin as shown in Table 1.
Table 1. VDDQSET and Output Voltages
VDDQSET
GND
VDDQ (V)
2.5
VTTREF and VTT
VVDDQSNS/2
NOTE
DDR
V5IN
1.8
VVDDQSNS/2
DDR2
FB Resistors
Adjustable
VVDDQSNS/2
1.5 V < VVDDQ < 3 V
VTT Linear Regulator and VTTREF
TPS51116 integrates high performance low-dropout linear regulator that is capable of sourcing and sinking
current up to 3 A. This VTT linear regulator employs ultimate fast response feedback loop so that small ceramic
capacitors are enough to keep tracking the VTTREF within ±40 mV at all conditions including fast load transient.
To achieve tight regulation with minimum effect of wiring resistance, a remote sensing terminal, VTTSNS, should
be connected to the positive node of VTT output capacitor(s) as a separate trace from VTT pin. For stable
operation, total capacitance of the VTT output terminal can be equal to or greater than 20 µF. It is recommended
to attach two 10-µF ceramic capacitors in parallel to minimize the effect of ESR and ESL. If ESR of the output
capacitor is greater than 2 mΩ, insert an RC filter between the output and the VTTSNS input to achieve loop
stability. The RC filter time constant should be almost the same or slightly lower than the time constant made by
the output capacitor and its ESR. VTTREF block consists of on-chip 1/2 divider, LPF and buffer. This regulator
also has sink and source capability up to 10 mA. Bypass VTTREF to GND by a 0.033-µF ceramic capacitor for
stable operation.
Outputs Management by S3, S5 Control
In the DDR/DDR2 memory applications, it is important to keep VDDQ always higher than VTT/VTTREF including
both start-up and shutdown. TPS51116 provides this management by simply connecting both S3 and S5
terminals to the sleep-mode signals such as SLP_S3 and SLP_S5 in the notebook PC system. All of VDDQ,
VTTREF and VTT are turned on at S0 state (S3 = S5 = high). In S3 state (S3 = low, S5 = high), VDDQ and
VTTREF voltages are kept on while VTT is turned off and left at high impedance (high-Z) state. The VTT output
is floated and does not sink or source current in this state. In S4/S5 states (S3 = S5 = low), all of the three
outputs are disabled. Outputs are discharged to ground according to the discharge mode selected by MODE pin
(see VDDQ and VTT Discharge Control section). Each state code represents as follow; S0 = full ON, S3 =
suspend to RAM (STR), S4 = suspend to disk (STD), S5 = soft OFF. (See Table 2)
Table 2. S3 and S5 Control
STATE
S0
S3
HI
S5
HI
VDDQ
VTTREF
VTT
On
On
On
On
On
S3
LO
LO
HI
Off (Hi-Z)
Off (Discharge)
S4/S5
LO
Off (Discharge)
Off (Discharge)
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TPS51116
www.ti.com
SLUS609A–MAY 2004–REVISED JUNE 2004
DETAILED DESCRIPTION (continued)
Soft-Start and Powergood
The soft start function of the SMPS is achieved by ramping up reference voltage and two-stage current clamp. At
the starting point, the reference voltage is set to 650 mV (87% of its target value) and the overcurrent threshold
is set half of the nominal value. When UVP comparator detects VDDQ become greater than 80% of the target,
the reference voltage is raised toward 750 mV using internal 4-bit DAC. This takes approximately 85 µs. The
overcurrent threshold is released to nominal value at the end of this period. The powergood signal waits another
45 µs after the reference voltage reaches 750 mV and the VDDQ voltage becomes good (above 95% of the
target voltage), then turns off powergood open-drain MOSFET.
The soft-start function of the VTT LDO is achieved by current clamp. The current limit threshold is also changed
in two stages using an internal powergood signal dedicated for LDO. During VTT is below the powergood
threshold, the current limit level is cut into 60% (2.2 A).This allows the output capacitors to be charged with low
and constant current that gives linear ramp up of the output. When the output comes up to the good state, the
overcurrent limit level is released to normal value (3.8 A). TPS51116 has an independent counter for each
output, but the PGOOD signal indicates only the status of VDDQ and does not indicate VTT powergood
externally. See Figure 1.
100%
87%
80%
V
VDDQ
V
OCL
V
PGOOD
V
S5
85 µs
45 µs
UDG−04066
Figure 1. VDDQ Soft-Start and Powergood Timing
Soft-start duration, TVDDQSS, TVTTSS are functions of output capacitances.
2 C
V
0.8
VDDQ
VDDQ
T
+
) 85 ms
VDDQSS
I
VDDQOCP
(2)
(3)
where IVDDQOCP is the current limit value for VDDQ switcher calculated by Equation 5.
C
V
VTT
VTT
T
+
VTTSS
I
VTTOCL
where, IVTTOCL = 2.2 A (typ). In each of the two previous calculations, no load current during start-up are
assumed. Note that both switchers and the LDO do not start up with full load condition.
VDDQ and VTT Discharge Control
TPS51116 discharges VDDQ, VTTREF and VTT outputs during S3 and S5 are both low. There are two different
discharge modes. The discharge mode can be set by connecting MODE pin as shown in Table 3.
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TPS51116
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SLUS609A–MAY 2004–REVISED JUNE 2004
DETAILED DESCRIPTION (continued)
Table 3. Discharge Selection
MODE
V5IN
DISCHARGE MODE
No discharge
VDDQ
S4/GND
Tracking discharge
Non-tracking discharge
When in tracking-discharge mode, TPS51116 discharges outputs through the internal VTT regulator transistors
and VTT output tracks half of VDDQ voltage during this discharge. Note that VDDQ discharge current flows via
VLDOIN to LDOGND thus VLDOIN must be connected to VDDQ output in this mode. The internal LDO can
handle up to 3 A and discharge quickly. After VDDQ is discharged down to 0.2 V, the internal LDO is turned off
and the operation mode is changed to the non-tracking-discharge mode.
When in non-tracking-discharge mode, TPS51116 discharges outputs using internal MOSFETs which are
connected to VDDQSNS and VTT. The current capability of these MOSFETs are limited to discharge slowly.
Note that VDDQ discharge current flows from VDDQSNS to PGND in this mode. In case of non-tracking mode,
TPS51116 does not discharge output charge at all.
Current Protection for VDDQ
The SMPS has cycle-by-cycle over current limiting control. The inductor current is monitored during the OFF
state and the controller keeps the OFF state during the inductor current is larger than the over current trip level.
The trip level and current sense scheme are determined by CS pin connection (see Current Sensing Scheme
section). For resistor sensing scheme, the trip level, VTRIP, is fixed value of 60 mV.
For RDS(on) sensing scheme, CS terminal sinks 10 µA and the trip level is set to the voltage across this RTRIP
resistor.
V
(mV)
R
(kW) 10 (mA)
TRIP
TRIP
(4)
As the comparison is done during the OFF state, VTRIP sets valley level of the inductor current. Thus, the load
current at over current threshold, IOCP, can be calculated as shown in Equation 5.
ǒV
Ǔ
V
* V
V
I
V
TRIP
IN
OUT
OUT
TRIP
RIPPLE
2
1
I
+
)
+
)
OCP
R
R
2 L f
V
DS(on)
DS(on)
IN
(5)
In an overcurrent condition, the current to the load exceeds the current to the output capacitor thus the output
voltage tends to fall down. If the output voltage becomes less than Powergood level, the VTRIP is cut into half and
the output voltage tends to be even lower. Eventually, it crosses the undervoltage protection threshold and
shutdown.
Current Protection for VTT
The LDO has an internally fixed constant over current limiting of 3.8 A while operating at normal condition. This
trip point is reduced to 2.2 A before the output voltage comes within ±5% of the target voltage or goes outside of
±10% of the target voltage.
Overvoltage and Undervoltage Protection for VDDQ
TPS51116 monitors a resistor divided feedback voltage to detect overvoltage and undervoltage. If VDDQSET is
connected to V5IN or GND, the feedback voltage is made by an internal resistor divider inside VDDQSNS pin. If
an external resistor divider is connected to VDDQSET pin, the feedback voltage is VDDQSET voltage itself.
When the feedback voltage becomes higher than 115% of the target voltage, the OVP comparator output goes
high and the circuit latches as the top MOSFET driver OFF and the bottom MOSFET driver ON.
Also, TPS51116 monitors VDDQSNS voltage directly and if it becomes greater than 4 V TPS51116 turns off the
top MOSFET driver. When the feedback voltage becomes lower than 70% of the target voltage, the UVP
comparator output goes high and an internal UVP delay counter begins counting. After 32 cycles, TPS51116
latches OFF both top and bottom MOSFETs. This function is enabled after 1007 cycles of SMPS operation to
ensure startup.
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TPS51116
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SLUS609A–MAY 2004–REVISED JUNE 2004
DETAILED DESCRIPTION (continued)
V5IN Undervoltage Lockout (UVLO) Protection
TPS51116 has V5IN undervoltage lock out protection (UVLO). When the V5IN voltage is lower than UVLO
threshold voltage, SMPS, VTTLDO and VTTREF are shut off. This is a non-latch protection.
V5IN Input Capacitor
Add a ceramic capacitor with a value between 1.0 µF and 4.7 µF placed close to the V5IN pin to stabilize 5 V
from any parasitic impedance from the supply.
Thermal Shutdown
TPS51116 monitors the temperature of itself. If the temperature exceeds the threshold value, 160°C (typ),
SMPS, VTTLDO and VTTREF are shut off. This is a non-latch protection and the operation is resumed when the
device is cooled down by about 10°C.
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TPS51116
www.ti.com
SLUS609A–MAY 2004–REVISED JUNE 2004
APPLICATION INFORMATION
Loop Compensation and External Parts Selection
Current Mode Operation
A buck converter using TPS51116 current mode operation can be partitioned into three portions, a voltage
divider, an error amplifier and a switching modulator. By linearizing the switching modulator, we can derive the
transfer function of the whole system. Since current mode scheme directly controls the inductor current, the
modulator can be linearized as shown in Figure 2.
Figure 2. Linearizing the Modulator
Here, the inductor is located inside the local feedback loop and its inductance does not appear in the small signal
model. As a result, a modulated current source including the power inductor can be modeled as a current source
with its transconductance of 1/RS and the output capacitor represent the modulator portion. This simplified model
is applicable in the frequency space up to approximately a half of the switching frequency. One note is, although
the inductance has no influence to small signal model, it has influence to the large signal model as it limits slew
rate of the current source. This means the buck converter’s load transient response, one of the large signal
behaviors, can be improved by using smaller inductance without affecting the loop stability.
Total open loop transfer function of the whole system is given by Equation 6.
H(s)
H (s) H (s) H (s)
1 2 3
(6)
Assuming RL>>ESR, RO>>RC and CC>>CC2, each transfer function of the three blocks is shown starting with
Equation 7.
R2
R2 ) R1
H (s) +
1
(
)
(7)
O ǒ1 ) s C RCǓ
R
C
H (s) + * gm
2
ǒ1 ) s C R Ǔ ǒ1 ) s C
R
C2
CǓ
C
O
(8)
(9)
(1 ) s C ESR)
O
RL
H (s) +
3
R
ǒ1 ) s C RLǓ
S
O
There are three poles and two zeros in H(s). Each pole and zero is given by the following five equations.
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TPS51116
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SLUS609A–MAY 2004–REVISED JUNE 2004
APPLICATION INFORMATION (continued)
1
w
+ ǒC ROǓ
P1
C
(10)
(11)
(12)
(13)
(14)
1
w
+ ǒC RLǓ
P2
O
1
R
w
+ ǒC
CǓ
P3
C2
1
+ ǒC RCǓ
w
Z1
C
1
+ ǒC ESRǓ
w
Z2
O
Usually, each frequency of those poles and zeros is lower than the 0 dB frequency, f0. However, the f0 should be
kept under 1/3 of the switching frequency to avoid effect of switching circuit delay. The f0 is given by Equation 15.
R
R
gm
gm
C
S
0.75
C
S
1
2p
R1
R1 ) R2
1
2p
f +
+
0
C
R
V
C
R
O
OUT
O
(15)
Based on small signal analysis above, the external components can be selected by following manner.
1. Choose the inductor. The inductance value should be determined to give the ripple current of
approximately 1/4 to 1/3 of maximum output current.
ǒV
Ǔ
ǒV
Ǔ
* V
V
* V
V
IN(max)
OUT
OUT
IN(max)
OUT OUT
3
1
L +
+
I
f
V
I
f
V
IND(ripple)
IN(max)
OUT(max)
IN(max)
(16)
The inductor also needs to have low DCR to achieve good efficiency, as well as enough room above peak
inductor current before saturation. The peak inductor current can be estimated as shown in Equation 17.
ǒV
Ǔ
* V
V
OUT
V
IN(max)
OUT
TRIP
1
I
+
)
IND(peak)
R
L f
V
DS(on)
IN(max)
(17)
2. Choose rectifying (bottom) MOSFET. When RDS(on) sensing scheme is selected, the rectifying MOSFET’s
on-resistance is used as this RS so that lower RDS(on) does not always promise better performance. In order
to clearly detect inductor current, minimum RS recommended is to give 15 mV or larger ripple voltage with
the inductor ripple current. This promises smooth transition from CCM to DCM or vice versa. Upper side of
the RDS(on) is of course restricted by the efficiency requirement, and usually this resistance affects efficiency
more at high-load conditions. When using external resistor current sensing, there is no restriction for low
RDS(on). However, the current sensing resistance RS itself affects the efficiency
3. Choose output capacitor(s). In cases of organic semiconductor capacitors (OS-CON) or specialty polymer
capacitors (SP-CAP), ESR to achieve required ripple value at stable state or transient load conditions
determines the amount of capacitor(s) need, and capacitance is then enough to satisfy stable operation. The
peak-to-peak ripple value can be estimated by ESR times the inductor ripple current for stable state, or ESR
times the load current step for a fast transient load response. In case of ceramic capacitor(s), usually ESR is
small enough to meet ripple requirement. On the other hand, transient undershoot and overshoot driven by
output capacitance becomes the key factor to determine the capacitor(s).
4. Determine f0 and calculate RC using Equation 18. Note that higher RC shows faster transient response in
cost of unstableness. If the transient response is not enough even with high RC value, try increasing the out
put capacitance. Recommended f0 is fOSC/4. Then RC can be derived by Equation 19.
15
TPS51116
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SLUS609A–MAY 2004–REVISED JUNE 2004
APPLICATION INFORMATION (continued)
V
C
gm
OUT
O
R
v 2p f
R
0
C
S
0.75
(18)
(19)
R
2.8
V
C
[mF] R [mW]
C
OUT
O
S
5. Calculate CC2. Purpose of this capacitance is to cancel zero caused by ESR of the output capacitor. In case
of ceramic capacitor(s) is used, no need for CC2
.
1
1
w
+ w
p3
+ ǒC
Ǔ
+ ǒC
CǓ
R
z2
ESR
O
C2
(20)
ǒC ESRǓ
O
C
+
C2
R
C
(21)
6. Calculate CC. The purpose of CC is to cut DC component to obtain high DC feedback gain. However, as it
causes phase delay, another zero to cancel this effect at f0 frequency is need. This zero, ωz1, is determined
by Cc and Rc. Recommended ωz1 is 10 times lower to the f0 frequency.
f
0
1
f
+
+
z1
10
2p C R
C
C
(22)
7. When using adjustable mode, determine the value of R1 and R2. Recommended R2 value is from 100
kΩ to 300 kΩ. Determine R1 using Equation 23.
V
0.75
OUT
R1 +
R2
0.75
(23)
D-CAP™ Mode Operation
A buck converter system using D-CAP™ Mode can be simplified as below.
Figure 3. Linearizing the Modulator
The VDDQSNS voltage is compare with internal reference voltage after divider resistors. The PWM comparator
determines the timing to turn on top MOSFET. The gain and speed of the comparator is high enough to keep the
voltage at the beginning of each on cycle (or the end of off cycle) substantially constant. The DC output voltage
may have line regulation due to ripple amplitude that slightly increases as the input voltage increase.
For the loop stability, the 0-dB frequency, f0, defined below need to be lower than 1/3 of the switching frequency.
16
TPS51116
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SLUS609A–MAY 2004–REVISED JUNE 2004
APPLICATION INFORMATION (continued)
f
SW
3
1
f +
v
0
2p ESR C
O
(24)
As f0 is determined solely by the output capacitor’s characteristics, loop stability of D-CAP™ mode is determined
by the capacitor’s chemistry. For example, specialty polymer capacitors (SP-CAP) have CO in the order of
several 100 µF and ESR in range of 10 mΩ. These makes f0 in the order of 100 kHz or less and the loop is then
stable. However, ceramic capacitors have f0 at more than 700 kHz, which is not suitable for this operational
mode.
Although D-CAP™ mode provides many advantages such as ease-of-use, minimum external components
configuration and extremely short response time, due to not employing an error amplifier in the loop, sufficient
amount of feedback signal needs to be provided by external circuit to reduce jitter level.
The required signal level is approximately 15 mV at comparing point. This gives VRIPPLE = (VOUT/0.75) x 15 (mV)
at the output node. The output capacitor’s ESR should meet this requirement.
The external components selection is much simple in D-CAP™ mode.
1. Choose inductor. This section is the same as the current mode. Please refer to the instructions in the
Current Mode Operation section.
2. Choose output capacitor(s).Organic semiconductor capacitor(s) or specialty polymer capacitor(s) are
recommended. Determine ESR to meet required ripple voltage above. A quick approximation is shown in
Equation 25.
V
0.015
OUT
VOUT
ESR +
[
60 [mW]
I
0.75
I
RIPPLE
OUT(max)
(25)
Thermal Design
Primary power dissipation of TPS51116 is generated from VTT regulator. VTT current flow in both source and
sink directions generate power dissipation from the part. In the source phase, potential difference between
VLDOIN and VTT times VTT current becomes the power dissipation, WDSRC
.
+ ǒV
Ǔ
I
W
* V
DSRC
VLDOIN
VTT
VTT
(26)
In this case, if VLDOIN is connected to an alternative power supply lower than VDDQ voltage, power loss can be
decreased.
For the sink phase, VTT voltage is applied across the internal LDO regulator, and the power dissipation, WDSNK
,
is calculated by Equation 27:
W
V
I
VTT
DSNK
VTT
(27)
Since this device does not sink AND source the current at the same time and IVTT varies rapidly with time, actual
power dissipation need to be considered for thermal design is an average of above value. Another power
consumption is the current used for internal control circuitry from V5IN supply and VLDOIN supply. V5IN
supports both the internal circuit and external MOSFETs drive current. The former current is in the VLDOIN
supply can be estimated as 1.5 mA or less at normal operational conditions.
These powers need to be effectively dissipated from the package. Maximum power dissipation allowed to the
package is calculated by Equation 28,
T
T
A(max)
J(max)
W
+
PKG
q
JA
(28)
where
•
•
•
TJ(max) is 125°C
TA(max) is the maximum ambient temperature in the system
θJA is the thermal resistance from the silicon junction to the ambient
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TPS51116
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SLUS609A–MAY 2004–REVISED JUNE 2004
APPLICATION INFORMATION (continued)
This thermal resistance strongly depends on the board layout. TPS51116 is assembled in a thermally enhanced
PowerPAD™ package that has exposed die pad underneath the body. For improved thermal performance, this
die pad needs to be attached to ground trace via thermal land on the PCB. This ground trace acts as a heat
sink/spread. The typical thermal resistance, 39.6°C/W, is achieved based on a 6.5 mm × 3.4 mm thermal land
with eight vias without air flow. It can be improved by using larger thermal land and/or increasing vias number.
Further information about PowerPAD™ and its recommended board layout is described in (SLMA002). This
document is available at <www.ti.com>.
Layout Considerations
Certain points must be considered before designing a layout using the TPS51116.
•
•
•
•
PCB trace defined as LL node, which connects to source of switching MOSFET, drain of rectifying MOSFET
and high-voltage side of the inductor, should be as short and wide as possible.
Consider adding a small snubber circuit, consists of 3 Ω and 1 nF, between LL and PGND in case a
high-frequency surge is observed on the LL voltage waveform.
All sensitive analog traces such as VDDQSNS, VTTSNS and CS should placed away from high-voltage
switching nodes such as LL, DRVL or DRVH nodes to avoid coupling.
VLDOIN should be connected to VDDQ output with short and wide trace. If different power source is used for
VLDOIN, an input bypass capacitor should be placed to the pin as close as possible with short and wide
connection.
•
•
The output capacitor for VTT should be placed close to the pin with short and wide connection in order to
avoid additional ESR and/or ESL of the trace.
VTTSNS should be connected to the positive node of VTT output capacitor(s) as a separate trace from the
high current power line and is strongly recommended to avoid additional ESR and/or ESL. If it is needed to
sense the voltage of the point of the load, it is recommended to attach the output capacitor(s) at that point.
Also, it is recommended to minimize any additional ESR and/or ESL of ground trace between GND pin and
the output capacitor(s).
•
•
Consider adding LPF at VTTSNS in case ESR of the VTT output capacitor(s) is larger than 2 mΩ.
VDDQSNS can be connected separately from VLDOIN. Remember that this sensing potential is the
reference voltage of VTTREF. Avoid any noise generative lines.
•
•
Negative node of VTT output capacitor(s) and VTTREF capacitor should be tied together by avoiding
common impedance to the high current path of the VTT source/sink current.
GND (Signal GND) pin node represents the reference potential for VTTREF and VTT outputs. Connect GND
to negative nodes of VTT capacitor(s), VTTREF capacitor and VDDQ capacitor(s) with care to avoid
additional ESR and/or ESL. GND and PGND (power ground) should be connected together at a single point.
•
In order to effectively remove heat from the package, prepare thermal land and solder to the package’s
thermal pad. Wide trace of the component-side copper, connected to this thermal land, helps heat spreading.
Numerous vias with a 0.33-mm diameter connected from the thermal land to the internal/solder-side ground
plane(s) should be used to help dissipation. Do NOT connect PGND to this thermal land underneath the
package.
18
TPS51116
www.ti.com
SLUS609A–MAY 2004–REVISED JUNE 2004
APPLICATION INFORMATION (continued)
Figure 4. D-CAP™ Mode
Table 4. D-CAP™ Mode Schematic Components
SYMBOL
R1
SPECIFICATION
5.1 kΩ
MANUFACTURER
PART NUMBER
-
R2
100 kΩ
-
R3
(100 ×VVDDQ - 75) kΩ
75 kΩ
-
R4
-
M1
30 V, 13 mΩ
30 V, 5 mΩ
International Rectifier
International Rectifier
IRF7821
IRF7832
M2
19
TPS51116
www.ti.com
SLUS609A–MAY 2004–REVISED JUNE 2004
Figure 5. Current Mode
Table 5. Current Mode
SYMBOL
R0
SPECIFICATION
MANUFACTURER
Vishay
PART NUMBER
6 mΩ, 1%
100 kΩ
WSL-2521 0.006
R2
-
-
M0
30 V, 13 mΩ
30 V, 5 mΩ
International Rectifier
International Rectifier
IRF7821
IRF7832
M1
20
TPS51116
www.ti.com
SLUS609A–MAY 2004–REVISED JUNE 2004
TYPICAL CHARACTERISTICS
V5IN SUPPLY CURRENT
V5IN SHUTDOWN CURRENT
vs
JUNCTION TEMPERATURE
vs
JUNCTION TEMPERATURE
1.0
0.9
0.8
0.7
2.0
1.8
1.6
1.4
0.6
0.5
0.4
1.2
1.0
0.8
0.3
0.2
0.6
0.4
0.1
0
0.2
0
−50
−50
0
50
100
150
0
50
100
150
T − Junction Temperature − °C
J
T − Junction Temperature − °C
J
Figure 6.
Figure 7.
V5IN SUPPLY CURRENT
vs
VLDOIN SUPPLY CURRENT
vs
LOAD CURRENT
TEMPERATURE
10
9
1.0
DDR2
= 0.9 V
V
VTT
0.9
0.8
0.7
8
7
6
0.6
0.5
0.4
5
4
3
2
0.3
0.2
1
0.1
0
0
−50
−2
−1
0
1
2
0
50
100
150
I
− VTT Current − A
T − Junction Temperature − °C
J
VTT
Figure 8.
Figure 9.
21
TPS51116
www.ti.com
SLUS609A–MAY 2004–REVISED JUNE 2004
TYPICAL CHARACTERISTICS (continued)
CS CURRENT
vs
JUNCTION TEMPERATURE
VDDQ DISCHARGE CURRENT
vs
JUNCTION TEMPERATURE
16
14
80
70
60
50
PGOOD = HI
12
10
8
40
30
6
PGOOD = LO
4
20
10
2
0
−50
0
50
100
150
−50
0
50
100
150
T − Junction Temperature − °C
J
T − Junction Temperature − °C
J
Figure 10.
Figure 11.
VTT DISCHARGE CURRENT
vs
JUNCTION TEMPERATURE
OVERVOLTAGE AND UNDERVOLTAGE THRESHOLD
vs
JUNCTION TEMPERATURE
30
140
120
100
80
25
20
15
10
V
OVP
V
UVP
60
−50
−50
0
50
100
150
0
50
100
150
T − Junction Temperature − °C
J
T − Junction Temperature − °C
J
Figure 12.
Figure 13.
22
TPS51116
www.ti.com
SLUS609A–MAY 2004–REVISED JUNE 2004
TYPICAL CHARACTERISTICS (continued)
SWITCHING FREQUENCY
vs
SWITCHING FREQUENCY
vs
INPUT VOLTAGE
OUTPUT CURRENT
430
450
400
D-CAP Mode
DDR2
I
= 7 A
VDDQ
420
410
350
300
250
DDR
400
390
380
200
150
100
50
DDR2
DDR
D−CAP Mode
V
IN
= 12 V
370
0
4
8
12
16
20
24
28
0
2
4
6
8
10
I
− VDDQ Output Current − A
V
IN
− Input Voltage − V
VDDQ
Figure 14.
Figure 15.
VDDQ OUTPUT VOLTAGE
vs
OUTPUT CURRENT (DDR)
VDDQ OUTPUT VOLTAGE
vs
INPUT VOLTAGE (DDR2)
1.820
1.815
1.810
1.805
1.800
1.795
1.790
1.785
1.780
1.820
1.815
1.810
1.805
1.800
1.795
1.790
1.785
1.780
D−CAP Mode
I
= 0 A
VDDQ
I
= 10 A
VDDQ
D−CAP Mode
V
IN
= 12 V
0
2
4
6
8
10
4
8
12
16
20
24
30
I
− VDDQ Output Current − A
VDDQ
V
IN
− Input Voltage − V
Figure 16.
Figure 17.
23
TPS51116
www.ti.com
SLUS609A–MAY 2004–REVISED JUNE 2004
TYPICAL CHARACTERISTICS (continued)
VTT OUTPUT VOLTAGE
vs
OUTPUT CURRENT (DDR)
VTT OUTPUT VOLTAGE
vs
OUTPUT CURRENT (DDR2)
0.94
0.93
0.92
0.91
0.90
0.89
0.88
0.87
0.86
1.30
1.29
1.28
1.27
1.26
V
= 1.8 V
VLDOIN
V
= 2.5 V
VLDOIN
1.25
1.24
1.23
1.22
V
= 1.5 V
VLDOIN
V
= 1.2 V
VLDOIN
V
= 1.8 V
0
VLDOIN
1.21
1.20
−3
−2
−1
0
1
2
3
−5 −4 −3 −2 −1
1
2
3
4
5
I
− VTT Output Current − A
I
− VTT Output Current − A
VTT
VTT
Figure 18.
Figure 19.
VTTREF OUTPUT VOLTAGE
vs
OUTPUT CURRENT (DDR)
VTTREF OUTPUT VOLTAGE
vs
OUTPUT CURRENT (DDR2)
1.252
1.251
1.250
1.249
1.248
0.904
0.903
0.902
0.901
0.900
DDR2
DDR
1.247
1.246
0.899
0.898
1.245
1.244
0.897
0.896
−10
−5
0
5
10
−10
−5
0
5
10
I
− VTTREF Current − A
I
− VTTREF Current − A
VTTREF
VTTREF
Figure 20.
Figure 21.
24
TPS51116
www.ti.com
SLUS609A–MAY 2004–REVISED JUNE 2004
TYPICAL CHARACTERISTICS (continued)
VDDQ EFFICIENCY (DDR)
vs
VDDQ EFFICIENCY (DDR2)
vs
VDDQ CURRENT
VDDQ CURRENT
100
100
V
IN
= 8 V
V
IN
= 8 V
V
VDDQ
= 2.5 V
V
VDDQ
= 1.8 V
90
80
90
80
V
IN
= 12 V
V
IN
= 20 V
V
IN
= 12 V
V
IN
= 20 V
70
60
70
60
50
50
0.001
0.01
0.1
1
10
0.001
0.01
I
0.1
1
10
I
− VDDQ Current − A
− VDDQ Current − A
VDDQ
VDDQ
Figure 22.
Figure 23.
V
VDDQ
(50 mV/div)
I
(2 A/div)
VDDQ
V
(10 mV/div)
(10 mV/div)
VTTREF
V
VTT
t − Time − 2 µs/div
t − Time − 20 µs/div
Figure 25. VDDQ Load Transient Response
Figure 24. Ripple Waveforms - Heavy Load Condition
25
TPS51116
www.ti.com
SLUS609A–MAY 2004–REVISED JUNE 2004
TYPICAL CHARACTERISTICS (continued)
V
VDDQ
(50 mV/div)
V
VTT
(20 mV/div)
S5
VDDQ
V
VTTREF
(20 mV/div)
VTTREF
PGOOD
I
VTT
(2 A/div)
I
= I
VTTREF
= 0 A
VDDQ
t − Time − 100 µs/div
Figure 27. VDDQ, VTT, and VTTREF Start-Up Waveforms
t − Time − 20 µs/div
Figure 26. VTT Load Transient Response
VDDQ
VDDQ
VTTREF
VTTREF
VTT
S5
VTT
S5
I
= I
VTT
= I
VTTREF
= 0 A
VDDQ
I
= I
VTT
= I
VTTREF
= 0 A
VDDQ
t − Time − 1 ms/div
Figure 29. Soft-Stop Waveforms Non-Tracking Discharge
t − Time − 200 µs/div
Figure 28. Soft-Start Waveforms Tracking Discharge
26
TPS51116
www.ti.com
SLUS609A–MAY 2004–REVISED JUNE 2004
TYPICAL CHARACTERISTICS (continued)
VDDQ BODE PLOT (CURRENT MODE)
VTT BODE PLOT, SOURCE (DDR2)
GAIN AND PHASE
vs
GAIN AND PHASE
vs
FREQUENCY
FREQUENCY
80
60
180
135
90
80
180
135
90
60
Phase
Phase
40
40
45
20
45
20
0
0
0
0
Gain
−45
−90
−135
−180
−20
−40
−60
−80
−20
−45
−90
−135
−180
Gain
−40
−60
I
= −1 A
VTT
I
= 7 A
VDDQ
−80
10 k
10 k
1 M
100 k
10 M
1 M
100 k
10 M
f − Frequency − Hz
f − Frequency − Hz
Figure 30.
Figure 31.
VTT BODE PLOT, SINK (DDR2)
GAIN AND PHASE
vs
FREQUENCY
80
60
180
135
90
40
Phase
Gain
20
45
0
0
−20
−40
−60
−80
−45
−90
−135
−180
I
= 1 A
VTT
10 k
1 M
100 k
10 M
f − Frequency − Hz
Figure 32.
27
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