TPS61041DRVR [TI]
LOW-POWER DC/DC BOOST CONVERTER IN SOT-23 AND SON PACKAGES; 低功耗DC / DC升压转换器采用SOT -23和儿子套餐
| 型号： | TPS61041DRVR |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | LOW-POWER DC/DC BOOST CONVERTER IN SOT-23 AND SON PACKAGES |
| 文件： | 总22页 (文件大小：528K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
TPS61040
TPS61041
www.ti.com
SLVS413C–OCTOBER 2002–REVISED AUGUST 2007
LOW-POWER DC/DC BOOST CONVERTER IN SOT-23 AND SON PACKAGES
1
FEATURES
DESCRIPTION
•
•
•
1.8-V to 6-V Input Voltage Range
The TPS61040/41 is
a
high-frequency boost
converter dedicated for small to medium LCD bias
supply and white LED backlight supplies. The device
is ideal to generate output voltages up to 28 V from a
dual cell NiMH/NiCd or a single cell Li-Ion battery.
The part can also be used to generate standard
3.3-V/5-V to 12-V power conversions.
Adjustable Output Voltage Range up to 28 V
400-mA (TPS61040) and 250-mA (TPS61041)
Internal Switch Current
•
•
•
•
•
Up to 1-MHz Switching Frequency
28-μA Typical No-Load Quiescent Current
1-μA Typical Shutdown Current
Internal Soft Start
The TPS61040/41 operates with
a
switching
frequency up to 1 MHz. This allows the use of small
external components using ceramic as well as
tantalum output capacitors. Together with the thin
SON package, the TPS61040/41 gives a very small
overall solution size. The TPS61040 has an internal
400 mA switch current limit, while the TPS61041 has
a 250-mA switch current limit, offering lower output
voltage ripple and allows the use of a smaller form
factor inductor for lower power applications. The low
quiescent current (typically 28 μA) together with an
optimized control scheme, allows device operation at
very high efficiencies over the entire load current
range.
Available in SOT23-5 and 2 × 2 × 0.8-mm SON
Packages
APPLICATIONS
•
•
•
•
•
•
•
LCD Bias Supply
White-LED Supply for LCD Backlights
Digital Still Camera
PDAs, Organizers, and Handheld PCs
Cellular Phones
Internet Audio Player
Standard 3.3-V/5-V to 12-V Conversion
DBV PACKAGE
(Top View)
DRV PACKAGE
(Top View)
1
2
3
5
4
1
2
3
6
5
4
GND
VIN
SW
NC
FB
VIN
EN
SW
GND
EN
FB
TYPICAL APPLICATION
EFFICIENCY
vs
OUTPUT CURRENT
90
88
86
84
82
80
78
76
74
72
70
L1
10 µH
V
= 18 V
D1
O
V = 5 V
I
V
V
OUT
V
IN
to 28 V
IN
1.8 V to 6.0 V
V = 3.6 V
I
C
FF
R1
1
3
2
5
4
V
SW
IN
C
1 µF
O
V = 2.4 V
I
FB
C
4.7 µF
IN
EN
GND
R2
0.10
1
10
100
I
- Output Current - mA
O
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2002–2007, Texas Instruments Incorporated
TPS61040
TPS61041
www.ti.com
SLVS413C–OCTOBER 2002–REVISED AUGUST 2007
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
(1)
ORDERING INFORMATION
SWITCH CURRENT
LIMIT, mA
PACKAGE
MARKING
TA
PART NUMBER
PACKAGE
TPS61040DBV
TPS61041DBV
TPS61040DRV
TPS61041DRV
400
250
400
250
SOT23-5
SOT23-5
PHOI
PHPI
CCL
–40°C to
85°C
SON-6 2×2
SON-6 2×2
CAW
(1) The devices are available in tape and reel and in tubes. Add R suffix to the part number (e.g.,
TPS61040DRVR) to order quantities of 3000 parts in tape and reel or add suffix T (e.g.,
TPS61040DRVT) to order a tube with 250 pieces..
FUNCTIONAL BLOCK DIAGRAM
SW
Under Voltage
Lockout
Bias Supply
400 ns Min
VIN
FB
Off Time
Error Comparator
-
S
Power MOSFET
N-Channel
+
RS Latch
Logic
Gate
Driver
V
REF
= 1.233 V
R
Current Limit
R
SENSE
+
_
6 µs Max
On Time
EN
Soft
Start
GND
2
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Product Folder Link(s): TPS61040 TPS61041
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SLVS413C–OCTOBER 2002–REVISED AUGUST 2007
Table 1. Terminal Functions
TERMINAL
NAME DBV NO. DRV NO.
I/O
DESCRIPTION
This is the enable pin of the device. Pulling this pin to ground forces the device into shutdown
mode reducing the supply current to less than 1 μA. This pin should not be left floating and needs
to be terminated.
EN
4
3
I
I
This is the feedback pin of the device. Connect this pin to the external voltage divider to program
the desired output voltage.
FB
3
4
GND
NC
2
–
1
5
–
–
Ground
No connection
Connect the inductor and the Schottky diode to this pin. This is the switch pin and is connected to
the drain of the internal power MOSFET.
SW
VIN
1
5
6
2
I
I
Supply voltage pin
DETAILED DESCRIPTION
OPERATION
The TPS61040/41 operates with an input voltage range of 1.8 V to 6 V and can generate output voltages up to
28 V. The device operates in a pulse-frequency-modulation (PFM) scheme with constant peak current control.
This control scheme maintains high efficiency over the entire load current range, and with a switching frequency
up to 1 MHz, the device enables the use of very small external components.
The converter monitors the output voltage, and as soon as the feedback voltage falls below the reference voltage
of typically 1.233 V, the internal switch turns on and the current ramps up. The switch turns off as soon as the
inductor current reaches the internally set peak current of typically 400 mA (TPS61040) or 250 mA (TPS61041).
See the Peak Current Control section for more information. The second criteria that turns off the switch is the
maximum on-time of 6 μs (typical). This is just to limit the maximum on-time of the converter to cover for extreme
conditions. As the switch is turned off the external Schottky diode is forward biased delivering the current to the
output. The switch remains off for a minimum of 400 ns (typical), or until the feedback voltage drops below the
reference voltage again. Using this PFM peak current control scheme the converter operates in discontinuous
conduction mode (DCM) where the switching frequency depends on the output current, which results in very high
efficiency over the entire load current range. This regulation scheme is inherently stable, allowing a wider
selection range for the inductor and output capacitor.
PEAK CURRENT CONTROL
The internal switch turns on until the inductor current reaches the typical dc current limit (ILIM) of 400 mA
(TPS61040) or 250 mA (TPS61042). Due to the internal propagation delay of typical 100 ns, theactualcurrent
exceeds the dc current limit threshold by a small amount. The typical peak current limit can be calculated:
V
IN
I
+ I
)
100 ns
peak(typ)
LIM
L
V
IN
I
I
+ 400 mA )
+ 250 mA )
100 ns for the TPS61040
peak(typ)
peak(typ)
L
V
IN
100 ns for the TPS61041
L
(1)
The higher the input voltage and the lower the inductor value, the greater the peak.
By selecting the TPS61040 or TPS61041, it is possible to tailor the design to the specific application current limit
requirements. A lower current limit supports applications requiring lower output power and allows the use of an
inductor with a lower current rating and a smaller form factor. A lower current limit usually has a lower output
voltage ripple as well.
Copyright © 2002–2007, Texas Instruments Incorporated
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SLVS413C–OCTOBER 2002–REVISED AUGUST 2007
SOFT START
All inductive step-up converters exhibit high inrush current during start-up if no special precaution is made. This
can cause voltage drops at the input rail during start up and may result in an unwanted or early system shut
down.
I
LIM
4
The TPS61040/41 limits this inrush current by increasing the current limit in two steps starting from
for 256
I
LIM
2
cycles to
for the next 256 cycles, and then full current limit (see Figure 14).
ENABLE
Pulling the enable (EN) to ground shuts down the device reducing the shutdown current to 1 μA (typical).
Because there is a conductive path from the input to the output through the inductor and Schottky diode, the
output voltage is equal to the input voltage during shutdown. The enable pin needs to be terminated and should
not be left floating. Using a small external transistor disconnects the input from the output during shutdown as
shown in Figure 18.
UNDERVOLTAGE LOCKOUT
An undervoltage lockout prevents misoperation of the device at input voltages below typical 1.5 V. When the
input voltage is below the undervoltage threshold, the main switch is turned off.
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature (unless otherwise noted)
(1)
UNIT
(2)
Supply voltages on pin VIN
–0.3 V to 7 V
–0.3 V to VIN + 0.3 V
30 V
(2)
Voltages on pins EN, FB
Switch voltage on pin SW
(2)
Continuous power dissipation
Operating junction temperature
Storage temperature
See Dissipation Rating Table
–40°C to 150°C
–65°C to 150°C
260°C
TJ
Tstg
Lead temperature (soldering 10 seconds)
(1) Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under recommended operating
conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
(2) All voltage values are with respect to network ground terminal.
DISSIPATION RATING TABLE
DERATING
T
A ≤ 25°C
FACTOR
ABOVE
TA = 70°C
POWER RATING POWER RATING
TA = 85°C
PACKAGE
RθJA
POWER RATING
TA = 25°C
DBV
DRV
250°C/W
76°C/W
357 mW
3.5 mW/°C
13 mW/°C
192 mW
688 mW
140 mW
500 mW
1300 mW
4
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SLVS413C–OCTOBER 2002–REVISED AUGUST 2007
RECOMMENDED OPERATING CONDITIONS
MIN
TYP
MAX UNIT
VIN
VOUT
L
Input voltage range
Output voltage range
Inductor(1)
1.8
6
V
V
28
2.2
10
μH
MHz
μF
μF
°C
f
Switching frequency(1)
1
(1)
CIN
COUT
TA
Input capacitor
4.7
(1)
Output capacitor
1
–40
–40
Operating ambient temperature
Operating junction temperature
85
TJ
125
°C
(1) See application section for further information.
ELECTRICAL CHARACTERISTICS
VIN = 2.4 V, EN = VIN, TA = –40°C to 85°C, typical values are at TA = 25°C (unless otherwise noted)
PARAMETER
SUPPLY CURRENT
TEST CONDITIONS
MIN
TYP
MAX UNIT
VIN
Input voltage range
1.8
6
50
1
V
μA
μA
V
IQ
Operating quiescent current
Shutdown current
IOUT = 0 mA, not switching, VFB = 1.3 V
EN = GND
28
0.1
1.5
ISD
VUVLO
ENABLE
VIH
Under-voltage lockout threshold
1.7
EN high level input voltage
EN low level input voltage
EN input leakage current
1.3
V
V
VIL
0.4
1
II
EN = GND or VIN
0.1
μA
POWER SWITCH AND CURRENT LIMIT
Vsw
toff
Maximum switch voltage
Minimum off time
30
550
7.5
V
250
4
400
6
ns
ton
Maximum on time
μs
RDS(on)
RDS(on)
MOSFET on-resistance
MOSFET on-resistance
MOSFET leakage current
MOSFET current limit
MOSFET current limit
VIN = 2.4 V; ISW = 200 mA; TPS61040
VIN = 2.4 V; ISW = 200 mA; TPS61041
VSW = 28 V
600
750
1
1000
1250
10
mΩ
mΩ
μA
mA
mA
ILIM
TPS61040
350
215
400
250
450
285
ILIM
TPS61041
OUTPUT
VOUT
Vref
Adjustable output voltage range
Internal voltage reference
Feedback input bias current
Feedback trip point voltage
VIN
28
1
V
V
1.233
IFB
VFB = 1.3 V
μA
V
VFB
1.8 V ≤ VIN ≤ 6 V
1.208 1.233 1.258
1.8 V ≤ VIN ≤ 6 V; VOUT = 18 V; Iload = 10 mA;
CFF = not connected
(1)
Line regulation
0.05
0.15
%/V
Load regulation(1)
VIN = 2.4 V; VOUT = 18 V; 0 mA ≤ IOUT ≤ 30 mA
%/mA
(1) The line and load regulation depend on the external component selection. See the application section for further information.
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SLVS413C–OCTOBER 2002–REVISED AUGUST 2007
TYPICAL CHARACTERISTICS
Table 2. Table of Graphs
FIGURE
vs Load current
1, 2, 3
4
η
Efficiency
vs Input voltage
IQ
Quiescent current
Feedback voltage
Switch current limit
vs Input voltage and temperature
vs Temperature
5
VFB
ISW
6
vs Temperature
7
vs Supply voltage, TPS61041
vs Supply voltage, TPS61040
vs Temperature
8
ICL
Switch current limit
RDS(on)
9
10
11
12
13
14
RDS(on)
vs Supply voltage
Line transient response
Load transient response
Start-up behavior
EFFICIENCY
EFFICIENCY
vs
LOAD CURRENT
vs
OUTPUT CURRENT
90
88
86
84
82
80
78
76
74
72
70
90
88
86
84
82
80
78
76
74
72
70
V
O
= 18 V
L = 10 µH
= 18 V
V
O
V = 5 V
I
TPS61040
V = 3.6 V
I
TPS61041
V = 2.4 V
I
0.10
1
10
100
0.10
1
10
100
I
O
- Output Current - mA
I - Load Current - mA
L
Figure 1.
Figure 2.
6
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SLVS413C–OCTOBER 2002–REVISED AUGUST 2007
EFFICIENCY
vs
LOAD CURRENT
EFFICIENCY
vs
INPUT VOLTAGE
90
V
90
88
86
84
82
80
78
76
74
= 18 V
L = 10 µH
= 18 V
O
88
V
O
I
O
= 10 mA
86
84
82
L = 10 µH
I
O
= 5 mA
L = 3.3 µH
80
78
76
74
72
70
72
70
1
2
3
4
5
6
0.10
1
10
100
I - Load Current - mA
L
V - Input Voltage - V
I
Figure 3.
Figure 4.
TPS61040
QUIESCENT CURRENT
vs
FEEDBACK VOLTAGE
vs
FREE-AIR TEMPERATIRE
INPUT VOLTAGE
40
1.24
T
= 85°C
= 27°C
= -40°C
A
35
30
25
20
15
10
1.238
T
A
1.236
1.234
T
A
V
CC
= 2.4 V
1.232
1.23
5
0
1.8
2.4
3
3.6
4.2
4.8
5.4
6
-40 -20
0
20
40
60
80 100 120
T
A
- Temperature - °C
V - Input Voltage - V
I
Figure 5.
Figure 6.
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SLVS413C–OCTOBER 2002–REVISED AUGUST 2007
TPS61040/41
SWITCH CURRENT LIMIT
vs
TPS61041
CURRENT LIMIT
vs
FREE-AIR TEMPERATURE
SUPPLY VOLTAGE
260
258
256
254
252
430
TPS61040
410
390
370
350
330
310
290
T
A
= 27°C
250
248
246
244
270
TPS61041
250
230
242
240
1.8
2.4
3
3.6
4.2
4.8
5.4
6
-40 -30 -20 -10
0
10 20 30 40 50 60 70 80 90
- Temperature - °C
T
A
V
CC
- Supply Voltage - V
Figure 7.
Figure 8.
TPS61040
TPS61040/41
CURRENT LIMIT
vs
STATIC DRAIN-SOURCE ON-STATE RESISTANCE
vs
SUPPLY VOLTAGE
FREE-AIR TEMPERATURE
420
1200
415
410
405
400
395
390
1000
800
600
400
TPS61041
T
A
= 27°C
TPS61040
200
0
385
380
−40−30 −20 −10 0 10 20 30 40 50 60 70 80 90
1.8
2.4
3
3.6
4.2
4.8
5.4
6
T
A
− Temperature − °C
V
CC
- Supply Voltage - V
Figure 9.
Figure 10.
8
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SLVS413C–OCTOBER 2002–REVISED AUGUST 2007
TPS61040/41
STATIC DRAIN-SOURCE ON-STATE RESISTANCE
vs
SUPPLY VOLTAGE
1000
V
O
= 18 V
900
800
700
600
500
400
300
200
V
I
2.4 V to 3.4 V
TPS61041
TPS61040
V
O
100 mV/div
100
0
1.8
2.4
3
3.6
4.2
4.8
5.4
6
200 µS/div
Figure 12. Line Transient Response
V
− Supply Voltage − V
CC
Figure 11.
V
O
= 18 V
V
O
= 18 V
V
O
V
O
100 mA/div
5 V/div
EN
1 V/div
V
O
1 mA to 10 mA
I
I
50 mA/div
200 µS/div
Figure 13. Load Transient Tresponse
Figure 14. Start-Up Behavior
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SLVS413C–OCTOBER 2002–REVISED AUGUST 2007
APPLICATION INFORMATION
INDUCTOR SELECTION, MAXIMUM LOAD CURRENT
Because the PFM peak current control scheme is inherently stable, the inductor value does not affect the stability
of the regulator. The selection of the inductor together with the nominal load current, input and output voltage of
the application determines the switching frequency of the converter. Depending on the application, inductor
values between 2.2 μH and 47 μH are recommended. The maximum inductor value is determined by the
maximum on time of the switch, typically 6 μs. The peak current limit of 400 mA/250 mA (typically) should be
reached within this 6-μs period for proper operation.
The inductor value determines the maximum switching frequency of the converter. Therefore, select the inductor
value that ensures the maximum switching frequency at the converter maximum load current is not exceeded.
The maximum switching frequency is calculated by the following formula:
V
(V
* V
IN(min)
OUT
L V
IN)
fS
+
max
I
P
OUT
(2)
Where:
IP = Peak current as described in the Peak Current Control section
L = Selected inductor value
VIN(min) = The highest switching frequency occurs at the minimum input voltage
If the selected inductor value does not exceed the maximum switching frequency of the converter, the next step
is to calculate the switching frequency at the nominal load current using the following formula:
2 I
(V
* V ) Vd)
load
OUT
IN
fSǒIloadǓ+
2
I
L
P
(3)
Where:
IP = Peak current as described in the Peak Current Control section
L = Selected inductor value
Iload = Nominal load current
Vd = Rectifier diode forward voltage (typically 0.3V)
A smaller inductor value gives a higher converter switching frequency, but lowers the efficiency.
The inductor value has less effect on the maximum available load current and is only of secondary order. The
best way to calculate the maximum available load current under certain operating conditions is to estimate the
expected converter efficiency at the maximum load current. This number can be taken out of the efficiency
graphs shown in Figure 1 through Figure 4. The maximum load current can then be estimated as follows:
2
I
L fS
max
* V
P
I
+ h
load max
2 (V
OUT
IN)
(4)
Where:
IP = Peak current as described in the Peak Current Control section
L = Selected inductor value
fSmax = Maximum switching frequency as calculated previously
η = Expected converter efficiency. Typically 70% to 85%
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SLVS413C–OCTOBER 2002–REVISED AUGUST 2007
The maximum load current of the conveter is the current at the operation point where the coverter starts to enter
the continuous conduction mode. Usually the converter should always operate in discontinuous conduction
mode.
Last, the selected inductor should have a saturation current that meets the maximum peak current of the
converter (as calculated in the Peak Current Control section). Use the maximum value for ILIMfor this calculation.
Another important inductor parameter is the dc resistance. The lower the dc resistance, the higher the efficiency
of the converter. See Table 3 and the typical applications for the inductor selection.
Table 3. Recommended Inductor for Typical LCD Bias Supply (see Figure 15)
DEVICE
INDUCTOR VALUE
10 μH
COMPONENT SUPPLIER
Sumida CR32-100
COMMENTS
High efficiency
10 μH
Sumida CDRH3D16-100
Murata LQH4C100K04
Sumida CDRH3D16-4R7
Murata LQH3C4R7M24
High efficiency
TPS61040
10 μH
High efficiency
4.7 μH
Small solution size
Small solution size
4.7 μH
High efficiency
Small solution size
TPS61041
10 μH
Murata LQH3C100K24
SETTING THE OUTPUT VOLTAGE
The output voltage is calculated as:
R1
R2
+ 1.233 V ǒ1 ) Ǔ
V
OUT
(5)
For battery-powered applications, a high-impedance voltage divider should be used with a typical value for R2 of
≤200 kΩ and a maximum value for R1 of 2.2 MΩ. Smaller values might be used to reduce the noise sensitivity of
the feedback pin.
A feedforward capacitor across the upper feedback resistor R1 is required to provide sufficient overdrive for the
error comparator. Without a feedforward capacitor, or one whose value is too small, the TPS61040/41 shows
double pulses or a pulse burst instead of single pulses at the switch node (SW), causing higher output voltage
ripple. If this higher output voltage ripple is acceptable, the feedforward capacitor can be left out.
The lower the switching frequency of the converter, the larger the feedforward capacitor value required. A good
starting point is to use a 10-pF feedforward capacitor. As a first estimation, the required value for the feedforward
capacitor at the operation point can also be calculated using the following formula:
1
C
+
FF
fS
20
2 p
R1
(6)
Where:
R1 = Upper resistor of voltage divider
fS = Switching frequency of the converter at the nominal load current (See the INDUCTOR SELECTION,
MAXIMUM LOAD CURRENT section for calculating the switching frequency)
CFF = Choose a value that comes closest to the result of the calculation
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The larger the feedforward capacitor the worse the line regulation of the device. Therefore, when concern for line
regulation is paramount, the selected feedforward capacitor should be as small as possible. See the LINE AND
LOAD REGULATION section for more information about line and load regulation.
LINE AND LOAD REGULATION
The line regulation of the TPS61040/41 depends on the voltage ripple on the feedback pin. Usually a 50 mV
peak-to-peak voltage ripple on the feedback pin FB gives good results.
Some applications require a very tight line regulation and can only allow a small change in output voltage over a
certain input voltage range. If no feedforward capacitor CFF is used across the upper resistor of the voltage
feedback divider, the device has the best line regulation. Without the feedforward capacitor the output voltage
ripple is higher because the TPS61040/41 shows output voltage bursts instead of single pulses on the switch pin
(SW), increasing the output voltage ripple. Increasing the output capacitor value reduces the output voltage
ripple.
If a larger output capacitor value is not an option, a feedforward capacitor CFF can be used as described in the
previous section. The use of a feedforward capacitor increases the amount of voltage ripple present on the
feedback pin (FB). The greater the voltage ripple on the feedback pin (≥50 mV), the worse the line regulation.
There are two ways to improve the line regulation further:
1. Use a smaller inductor value to increase the switching frequency which will lower the output voltage ripple,
as well as the voltage ripple on the feedback pin.
2. Add a small capacitor from the feedback pin (FB) to ground to reduce the voltage ripple on the feedback pin
down to 50 mV again. As a starting point, the same capacitor value as selected for the feedforward capacitor
CFF can be used.
OUTPUT CAPACITOR SELECTION
For best output voltage filtering, a low ESR output capacitor is recommended. Ceramic capacitors have a low
ESR value but tantalum capacitors can be used as well, depending on the application.
Assuming the converter does not show double pulses or pulse bursts on the switch node (SW), the output
voltage ripple can be calculated as:
I
L
I
out
1
P
DV
+
–
) I ESR
ǒ
Ǔ
out
P
C
fS(Iout) Vout ) Vd–Vin
out
(7)
where:
IP = Peak current as described in the Peak Current Control section
L = Selected inductor value
Iout = Nominal load current
fS (Iout) = Switching frequency at the nominal load current as calculated previously
Vd = Rectifier diode forward voltage (typically 0.3 V)
Cout = Selected output capacitor
ESR = Output capacitor ESR value
See Table 4 and the typical applications section for choosing the output capacitor.
Table 4. Recommended Input and Output Capacitors
DEVICE
CAPACITOR
4.7 μF/X5R/0805
10 μF/X5R/0805
1 μF/X7R/1206
1 μF/X5R/1206
4.7 μF/X5R/1210
VOLTAGE RATING
COMPONENT SUPPLIER
Tayo Yuden JMK212BY475MG
Tayo Yuden JMK212BJ106MG
Tayo Yuden TMK316BJ105KL
Tayo Yuden GMK316BJ105KL
Tayo Yuden TMK325BJ475MG
COMMENTS
CIN/COUT
CIN/COUT
COUT
6.3 V
6.3 V
25 V
35 V
25 V
TPS61040/41
COUT
COUT
12
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Product Folder Link(s): TPS61040 TPS61041
TPS61040
TPS61041
www.ti.com
SLVS413C–OCTOBER 2002–REVISED AUGUST 2007
INPUT CAPACITOR SELECTION
For good input voltage filtering, low ESR ceramic capacitors are recommended. A 4.7 μF ceramic input capacitor
is sufficient for most of the applications. For better input voltage filtering this value can be increased. See Table 4
and typical applications for input capacitor recommendations.
DIODE SELECTION
To achieve high efficiency a Schottky diode should be used. The current rating of the diode should meet the
peak current rating of the converter as it is calculated in the Peak Current Control section. Use the maximum
value for ILIM for this calculation. See Table 5 and the typical applications for the selection of the Schottky diode.
Table 5. Recommended Schottky Diode for Typical LCD Bias Supply (see Figure 15)
DEVICE
REVERSE VOLTAGE
COMPONENT SUPPLIER
ON Semiconductor MBR0530
ON Semiconductor MBR0520
ON Semiconductor MBRM120L
Toshiba CRS02
COMMENTS
30 V
20 V
20 V
30 V
TPS61040/41
High efficiency
LAYOUT CONSIDERATIONS
Typical for all switching power supplies, the layout is an important step in the design; especially at high peak
currents and switching frequencies. If the layout is not carefully done, the regulator might show noise problems
and duty cycle jitter.
The input capacitor should be placed as close as possible to the input pin for good input voltage filtering. The
inductor and diode should be placed as close as possible to the switch pin to minimize the noise coupling into
other circuits. Because the feedback pin and network is a high-impedance circuit, the feedback network should
be routed away from the inductor. The feedback pin and feedback network should be shielded with a ground
plane or trace to minimize noise coupling into this circuit.
Wide traces should be used for connections in bold as shown in Figure 15. A star ground connection or ground
plane minimizes ground shifts and noise.
D1
L1
V
O
C
FF
R1
V
V
IN
SW
FB
IN
C
O
C
IN
R2
EN
GND
Figure 15. Layout Diagram
Copyright © 2002–2007, Texas Instruments Incorporated
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TPS61040
TPS61041
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SLVS413C–OCTOBER 2002–REVISED AUGUST 2007
L1
10 µH
D1
V
18 V
OUT
V
IN
1.8 V to 6 V
TPS61040
C
22 pF
FF
R1
2.2 MW
V
IN
SW
FB
C2
1 µF
C1
4.7 µF
L1:
D1:
C1:
C2:
Sumida CR32-100
Motorola MBR0530
Tayo Yuden JMK212BY475MG
Tayo Yuden TMK316BJ105KL
EN
GND
R2
160 kW
Figure 16. LCD Bias Supply
L1
10 µH
D1
V
O
18 V
TPS61040
C
FF
R1
2.2 MW
22 pF
V
V
IN
SW
FB
IN
C2
1 µF
1.8 V to 6 V
C1
4.7 µF
DAC or Analog Voltage
0 V = 25 V
1.233 V = 18 V
EN
GND
R2
160 kW
L1:
Sumida CR32-100
D1:
C1:
C2:
Motorola MBR0530
Tayo Yuden JMK212BY475MG
Tayo Yuden GMK316BJ105KL
Figure 17. LCD Bias Supply With Adjustable Output Voltage
R3
200 kW
BC857C
L1
10 µH
D1
V
IN
V
OUT
1.8 V to 6 V
18 V / 10 mA
TPS61040
R1
C
FF
2.2 MW
22 pF
V
IN
SW
FB
C2
1 µF
C3
0.1 µF
(Optional)
C1
4.7 µF
R2
160 kW
EN
GND
L1:
D1:
C1:
C2:
Sumida CR32-100
Motorola MBR0530
Tayo Yuden JMK212BY475MG
Tayo Yuden TMK316BJ105KL
Figure 18. LCD Bias Supply With Load Disconnect
14
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Product Folder Link(s): TPS61040 TPS61041
TPS61040
TPS61041
www.ti.com
SLVS413C–OCTOBER 2002–REVISED AUGUST 2007
D3
V2 = -10 V/15 mA
D2
C4
4.7 µF
C3
1 µF
L1
D1
6.8 µH
V1 = 10 V/15 mA
TPS61040
C
FF
22 pF
R1
1.5 MW
V
SW
FB
IN
V
IN
= 2.7 V to 5 V
C2
1 µF
C1
4.7 µF
L1:
Murata LQH4C6R8M04
D1, D2, D3: Motorola MBR0530
EN
GND
R2
C1:
Tayo Yuden JMK212BY475MG
210 kW
C2, C3, C4: Tayo Yuden EMK316BJ105KF
Figure 19. Positive and Negative Output LCD Bias Supply
L1
6.8 µH
D1
V
O =
12 V/35 mA
TPS61040
C
4.7 pF
FF
R1
1.8 MW
V
IN
3.3 V
V
IN
SW
FB
C2
4.7 µF
C1
10 µF
L1:
Murata LQH4C6R8M04
Motorola MBR0530
Tayo Yuden JMK212BJ106MG
Tayo Yuden EMK316BJ475ML
EN
GND
R2
205 kW
D1:
C1:
C2:
Figure 20. Standard 3.3-V to 12-V Supply
D1
3.3 µH
TPS61040
5 V/45 mA
C
FF
3.3 pF
R1
620 kW
V
1.8 V to 4 V
SW
FB
IN
C2
4.7 µF
C1
4.7 µF
R2
200 kW
EN
GND
L1:
D1:
Murata LQH4C3R3M04
Motorola MBR0530
C1, C2: Tayo Yuden JMK212BY475MG
Figure 21. Dual Battery Cell to 5-V/50-mA Conversion
Efficiency Approx. Equals 84% at VIN = 2.4 V to Vo = 5 V/45 mA
Copyright © 2002–2007, Texas Instruments Incorporated
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Product Folder Link(s): TPS61040 TPS61041
TPS61040
TPS61041
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SLVS413C–OCTOBER 2002–REVISED AUGUST 2007
L1
D1
10 µH
D2
24 V
(Optional)
V
CC
= 2.7 V to 6 V
V
IN
SW
FB
C1
4.7 µF
L1:
D1:
C1:
C2:
Murata LQH4C100K04
Motorola MBR0530
Tayo Yuden JMK212BY475MG
Tayo Yuden TMK316BJ105KL
C2
1 µF
R
82 Ω
EN
PWM
100 Hz to 500 Hz
GND
S
Figure 22. White LED Supply With Adjustable Brightness Control
Using a PWM Signal on the Enable Pin, Efficiency Approx. Equals 86% at VIN = 3 V, ILED = 15 mA
D1
L1
MBRM120L
10 µH
†
C2
D2
24 V
(Optional)
V
CC
= 2.7 V to 6 V
100 nF
V
SW
FB
IN
C1
4.7 µF
R1
120 kW
EN
GND
R
S
110 W
L1:
D1:
C1:
C2:
Murata LQH4C3R3M04
Motorola MBR0530
Tayo Yuden JMK212BY475MG
Standard Ceramic Capacitor
Analog Brightness Control
3.3 V Led Off
R2 160 kW
0 V Iled = 20 mA
A. A smaller output capacitor value for C2 causes a larger LED ripple.
Figure 23. White LED Supply With Adjustable Brightness Control
Using an Analog Signal on the Feedback Pin
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Product Folder Link(s): TPS61040 TPS61041
PACKAGE OPTION ADDENDUM
www.ti.com
15-Oct-2007
PACKAGING INFORMATION
Orderable Device
TPS61040DBVR
Status (1)
ACTIVE
ACTIVE
Package Package
Pins Package Eco Plan (2) Lead/Ball Finish MSL Peak Temp (3)
Qty
Type
Drawing
SOT-23
DBV
5
3000 Green (RoHS & CU NIPDAU Level-1-260C-UNLIM
no Sb/Br)
TPS61040DBVRG4
SOT-23
DBV
5
3000 Green (RoHS & CU NIPDAU Level-1-260C-UNLIM
no Sb/Br)
TPS61040DRVR
TPS61041DBVR
PREVIEW
ACTIVE
SON
DRV
DBV
6
5
3000
TBD
Call TI
Call TI
SOT-23
3000 Green (RoHS & CU NIPDAU Level-1-260C-UNLIM
no Sb/Br)
TPS61041DBVRG4
ACTIVE
SOT-23
DBV
5
3000 Green (RoHS & CU NIPDAU Level-1-260C-UNLIM
no Sb/Br)
TPS61041DRVR
TPS61041DRVT
PREVIEW
PREVIEW
SON
SON
DRV
DRV
5
5
3000
250
TBD
TBD
Call TI
Call TI
Call TI
Call TI
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check
http://www.ti.com/productcontent for the latest availability information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements
for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered
at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and
package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS
compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame
retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder
temperature.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is
provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the
accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take
reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on
incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited
information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI
to Customer on an annual basis.
Addendum-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
5-Oct-2007
TAPE AND REEL BOX INFORMATION
Device
Package Pins
Site
Reel
Reel
A0 (mm)
B0 (mm)
K0 (mm)
P1
W
Pin1
Diameter Width
(mm) (mm) Quadrant
(mm)
179
(mm)
TPS61040DBVR
TPS61041DBVR
DBV
DBV
5
5
SITE 48
SITE 48
8
8
3.2
3.2
3.2
3.2
1.4
1.4
4
4
8
8
Q3
Q3
179
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
5-Oct-2007
Device
Package
Pins
Site
Length (mm) Width (mm) Height (mm)
TPS61040DBVR
TPS61041DBVR
DBV
DBV
5
5
SITE 48
SITE 48
195.0
195.0
200.0
200.0
45.0
45.0
Pack Materials-Page 2
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