TPS62231DRYT [TI]
3 MHz Ultra Small Step Down Converter in 1x1.5 SON Package; 3 MHz的超小型降压转换器， 1x1.5 SON封装
| 型号： | TPS62231DRYT |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | 3 MHz Ultra Small Step Down Converter in 1x1.5 SON Package |
| 文件： | 总28页 (文件大小：1307K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
TPS62230
TPS62231
TPS62232
www.ti.com ..................................................................................................................................................................................................... SLVS941–APRIL 2009
3 MHz Ultra Small Step Down Converter in 1x1.5 SON Package
1
FEATURES
DESCRIPTION
•
•
•
•
•
•
3 MHz switch frequency
Up to 94% efficiency
The TPS6223X device family is a high frequency
synchronous step down DC-DC converter optimized
for battery powered portable applications. It supports
up to 500mA output current and allows the use of tiny
and low cost chip inductors and capacitors.
Output Peak Current up to 500mA
Excellent AC and Transient Load Regulation
High PSRR (up to 90dB)
Small External Output Filter Components
1.0µH/ 4.7µF
With a wide input voltage range of 2.05V to 6V the
device supports applications powered by Li-Ion
batteries with extended voltage range. The minimum
input voltage of 2.05V allows as well the operation
from Li-primary or two alkaline batteries. Different
fixed output voltage versions are available from 1.2V
to 2.5V.
•
•
VIN range from 2.05V to 6V
Optimized Power Save Mode For Low Output
Ripple Voltage
•
•
•
•
•
•
•
Forced PWM Mode Operation
Typ. 22 µA Quiescent Current
The TPS6223X series features switch frequency up
to 3.8MHz. At medium to heavy loads, the converter
operates in PWM mode and automatically enters
Power Save Mode operation at light load currents to
maintain high efficiency over the entire load current
range.
100% Duty Cycle for Lowest Dropout
Small 1 × 1.5 × 0.6mm3 SON Package
12 mm2 Minimum Solution Size
Supports 0.6 mm Maximum Solution Height
Soft Start with typ. 100µs Start Up Time
Because of its excellent PSRR and AC load
regulation performance, the device is also suitable to
replace linear regulators to obtain better power
conversion efficiency.
APPLICATIONS
•
•
•
•
•
•
LDO Replacement
Portable Audio, Portable Media
Cell Phones
Low Power Wireless
Low Power DSP Core Supply
Digital Cameras
The Power Save Mode in TPS6223X reduces the
quiescent current consumption down to 22µA during
light load operation. It is optimized to achieve very
low output voltage ripple even with small external
component and features excellent ac load regulation.
For very noise sensitive applications, the device can
be forced to PWM Mode operation over the entire
load range by pulling the MODE pin high. In the
shutdown mode, the current consumption is reduced
to less than 1µA. The TPS6223X is available in a 1 ×
1.5mm2 6 pin SON package.
V
IN
L
TPS62231
1/2.2 mH
V
2.05 V - 6 V
OUT
VIN
SW
1.8 V
C
FB
EN
IN
C
OUT
MODE
GND
2.2 mF
4.7 mF
Total area
12mm²
L1
V
IN
C2
GND
V OUT
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
UNLESS OTHERWISE NOTED this document contains
PRODUCTION DATA information current as of publication date.
Products conform to specifications per the terms of Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2009, Texas Instruments Incorporated
TPS62230
TPS62231
TPS62232
SLVS941–APRIL 2009 ..................................................................................................................................................................................................... www.ti.com
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
ORDERING INFORMATION
PACKAGE
DESIGNATOR
PACKAGE
MARKING
TA
PART NUMBER(1)
TPS62230
OUTPUT VOLTAGE(2)
ORDERING
2.5 V
1.8 V
1.2 V
1.0 V
1.3 V
1.5 V
2.0 V
2.1 V
2.25 V
2.3 V
2.7 V
2.9 V
3.0 V
DRY
DRY
DRY
DRY
DRY
DRY
DRY
DRY
DRY
DRY
DRY
DRY
DRY
TPS62230DRY
TPS62231DRY
TPS62232DRY
GV
GW
GX
TPS62231
TPS62232
TPS6223-1.0(3)
TPS6223-1.3(3)
TPS6223-1.5(3)
TPS6223-2.0(3)
TPS6223-2.1(3)
TPS6223-2.25(3)
TPS6223-2.3(3)
TPS6223-2.7(3)
TPS6223-2.9(3)
TPS6223-3.0(3)
–40°C to 85°C
(1) The DRY package is available in tape on reel. Add R suffix to order quantities of 3000 parts per reel, T suffix for 250 parts per reel.
(2) Contact TI for other fixed output voltage options
(3) Device status is product preview, contact TI for more details
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature range (unless otherwise noted)
(1)
VALUE
–0.3 to 7
UNIT
Voltage at VIN and SW Pin(2)
Voltage at EN, MODE Pin(2)
V
V
V
A
VI
–0.3 to VIN +0.3, ≤7
–0.3 to 3.6
internally limited
2
(2)
Voltage at FB Pin
Peak output current
ESD rating(3)
HBM Human body model
CDM Charge device model
Machine model
kV
V
1
200
Power dissipation
Internally limited
–40 to 125
–65 to 150
TJ
Maximum operating junction temperature
Storage temperature range
°C
°C
Tstg
(1) Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only and functional operation of the device at these or any other conditions beyond those indicated under recommended operating
conditions is not implied. Exposure to absolute–maximum–rated conditions for extended periods may affect device reliability.
(2) All voltage values are with respect to network ground terminal.
(3) The human body model is a 100-pF capacitor discharged through a 1.5-kΩ resistor into each pin. The machine model is a 200-pF
capacitor discharged directly into each pin.
DISSIPATION RATINGS(1)
POWER RATING
FOR TA ≤ 25°C
DERATING FACTOR
ABOVE TA = 25°C
PACKAGE
RθJA
1 × 1.5 SON
234°C/W(2)
420 mW
4.2 mW/°C
(1) Maximum power dissipation is a function of TJ(max), θJA and TA. The maximum allowable power dissipation at any allowable ambient
temperature is PD = [TJ(max) – TA] /θJA
(2) This thermal data is measured with high-K board (4 layers board according to JESD51-7 JEDEC standard).
.
2
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TPS62230
TPS62231
TPS62232
www.ti.com ..................................................................................................................................................................................................... SLVS941–APRIL 2009
RECOMMENDED OPERATING CONDITIONS
operating ambient temperature TA = –40 to 85°C (unless otherwise noted)(1)
MIN
NOM
MAX UNIT
(2)
Supply voltage VIN
2.05
6
V
Effective inductance
Effective capacitance
2.2
4.7
3.0
2.5
µH
µF
2.0
(4)
V
OUT ≤ VIN -1 V(3)
500 mA maximum IOUT
3.6
2.7
Recommended minimum
supply voltage
(5)
350mA maximum IOUT
60 mA maximum output current(5)
V
VOUT ≤ 1.8V
2.05
125
Operating virtual junction temperature range, TJ
–40
°C
(1) In applications where high power dissipation and/or poor package thermal resistance is present, the maximum ambient temperature may
have to be derated. Maximum ambient temperature (TA(max)) is dependent on the maximum operating junction temperature (TJ(max)), the
maximum power dissipation of the device in the application (PD(max)), and the junction-to-ambient thermal resistance of the part/package
in the application (θJA), as given by the following equation: TA(max) = TJ(max) – (θJA × PD(max)).
(2) The minimum required supply voltage for startup is 2.05 V. The part is functional down to the falling UVL (Under Voltage Lockout)
threshold.
(3) For a voltage difference between minimum VIN and VOUT of ≥ 1 V
(4) Typical value applies for TA = 25°C, maximum value applies for TA = 70°C with TJ ≤ 125°C, PCB layout needs to support proper thermal
performance.
(5) Typical value applies for TA = 25°C, maximum value applies for TA = 85°C with TJ ≤ 125°C, PCB layout needs to support proper thermal
performance.
Copyright © 2009, Texas Instruments Incorporated
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TPS62230
TPS62231
TPS62232
SLVS941–APRIL 2009 ..................................................................................................................................................................................................... www.ti.com
ELECTRICAL CHARACTERISTICS
VIN = 3.6V, VOUT = 1.8V, EN = VIN, MODE = GND, TA = –40°C to 85°C(1) typical values are at TA = 25°C (unless otherwise
noted), CIN = 2.2µF, L = 2.2µH, COUT = 4.7µF, see parameter measurement information
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX UNIT
SUPPLY
(2)
VIN
Input voltage range
2.05
6
V
IOUT = 0mA. PFM mode enabled (Mode = 0)
device not switching
22
25
3
40
µA
IOUT = 0mA. PFM mode enabled (Mode = 0)
device switching, VIN = 3.6V, VOUT = 1.2V
µA
IQ
Operating quiescent current
IOUT = 0 mA. Switching with no load
(MODE/DATA = VIN), PWM operation,
VOUT = 1.8V, L = 2.2µH
mA
ISD
Shutdown current
EN = GND(3)
0.1
1.8
1.9
1
1.9
µA
V
Falling
VUVLO
Undervoltage lockout threshold
Rising
2.05
V
ENABLE, MODE THRESHOLD
VIH TH Threshold for detecting high EN, MODE 2.05 V ≤ VIN ≤ 6V , rising edge
VIL TH HYS Threshold for detecting low EN, MODE 2.05 V ≤ VIN ≤ 6V , falling edge
0.8
0.6
1
V
V
0.4
IIN
Input bias Current, EN, MODE
EN, MODE = GND or VIN = 3.6V
0.01
0.5
µA
POWER SWITCH
High side MOSFET on-resistance
600
350
850
850
480
RDS(ON)
VIN = 3.6V, TJmax = 85°C; RDS(ON) max value
VIN = 3.6V, open loop
mΩ
Low Side MOSFET on-resistance
Forward current limit MOSFET
high-side
690
550
1050 mA
ILIMF
TSD
Forward current limit MOSFET low side
Thermal shutdown
840
150
20
1220 mA
Increasing junction temperature
Decreasing junction temperature
°C
°C
Thermal shutdown hysteresis
CONTROLLER
TONmin
TOFFmin
OUTPUT
VREF
Minimum ON time
VIN 3.6V, VOUT = 1.8V, Mode = high, IOUT = 0 mA
135
40
ns
ns
Minimum OFF time
Internal Reference Voltage
Output voltage accuracy(4)
0.70
0%
V
VOUT
VIN = 3.6V, Mode = GND, device operating in PFM
Mode, IOUT = 0mA
VIN = 3.6V, MODE = VIN
IOUT = 0 mA
,
TA = 25°C
–2.0%
2.0%
2.5%
%/mA
%/V
TA = –40°C to 85°C –2.5%
DC output voltage load regulation
DC output voltage line regulation
Start-up Time
PWM operation, Mode = VIN = 3.6V, VOUT = 1.8 V
0.001
0
IOUT = 0 mA, Mode = VIN, 2.05V ≤ VIN ≤ 6V
tStart
Time from active EN to VOUT = 1.8V, VIN = 3.6V,
100
µs
10Ω load
ILK_SW
Leakage current into SW pin
VIN = VOUT = VSW = 3.6 V, EN = GND(5)
0.1
0.5
µA
(1) In applications where high power dissipation and/or poor package thermal resistance is present, the maximum ambient temperature may
have to be derated. Maximum ambient temperature (TA(max)) is dependent on the maximum operating junction temperature (TJ(max)), the
maximum power dissipation of the device in the application (PD(max)), and the junction-to-ambient thermal resistance of the
part/package in the application (θJA), as given by the following equation: TA(max) = TJ(max) – (θJA × PD(max)).
(2) The minimum required supply voltage for startup is 2.05V. The part is functional down to the falling UVL (Under Voltage Lockout)
threshold
(3) Shutdown current into VIN pin, includes internal leakage
(4) VIN = VO + 1.0 V
(5) The internal resistor divider network is disconnected from FB pin.
4
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Product Folder Link(s): TPS62230 TPS62231 TPS62232
TPS62230
TPS62231
TPS62232
www.ti.com ..................................................................................................................................................................................................... SLVS941–APRIL 2009
DRY PACKAGE
(TOP VIEW)
6
5
4
1
2
3
MODE
SW
FB
EN
VIN
GND
PIN FUNCTIONS
PIN
I/O
DESCRIPTION
NAME
VIN
NO
3
PWR VIN power supply pin.
PWR GND supply pin
GND
EN
4
5
IN
This is the enable pin of the device. Pulling this pin to low forces the device into shutdown mode. Pulling
this pin to high enables the device. This pin must be terminated.
SW
2
OUT
This is the switch pin and is connected to the internal MOSFET switches. Connect the inductor to this
terminal
FB
6
1
IN
IN
Feedback Pin for the internal regulation loop. Connect this pin directly to the output capacitor.
MODE
MODE pin = high forces the device to operate in PWM mode. Mode = low enables the Power Save Mode
with automatic transition from PFM (Pulse frequency mode) to PWM (pulse width modulation) mode.
FUNCTIONAL BLOCK DIAGRAM
VIN
Undervoltage
Lockout
VREF
0.70 V
Current
Bandgap
Limit Comparator
Limit
High Side
MODE
Softstart
MODE
PMOS
VIN
Gate Driver
Anti
Shoot-Through
Min. On Time
Control
Logic
SW
FB
EN
Min. OFF Time
VREF
NMOS
FB
Limit
Low Side
Integrated
Feed Back
Network
Error
Comparator
Thermal
Shutdown
Zero/Negative
Current Limit Comparator
EN
GND
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TPS62230
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SLVS941–APRIL 2009 ..................................................................................................................................................................................................... www.ti.com
PARAMETER MEASUREMENT INFORMATION
TPS6223X
L = 1/2.2 mH
V
= 2.05 V to 6 V
IN
V
OUT
VIN
SW
FB
C
C
EN
IN
OUT
4.7 mF
2.2 mF
MODE
GND
C
: Murata GRM155R60J225ME15D 2.2 mF 0402 size
IN
C
C
: Murata GRM188R60J475ME 4.7 mF 0603 size, VOUT >= 1.8 V
: Taiyo Yuden AMK105BJ475MV 4.7 mF 0402 size, VOUT = 1.2 V
OUT
OUT
l: Murata LQM2HPN1R0MJ0 1 mH, LQM2HPN2R2MJ0 2.2 mH,
size 2.5x2.0x1.2mm3
6
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TPS62230
TPS62231
TPS62232
www.ti.com ..................................................................................................................................................................................................... SLVS941–APRIL 2009
TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
η
Efficiency
vs Load current
1, 2, 3, 4, 5, 6, 7
VO
Output voltage
vs Output current
8, 9, 10, 11, 12, 13
14, 15, 16, 17, 18, 19,
20
Switching frequency
vs Output current
IQ
Quiescent current
Shutdown current
vs Ambient temperature
vs Ambient temperature
21
22
ISD
PMOS Static drain-source on-state
resistance
vs Supply voltage and ambient temperature
23
24
rDS(ON)
PSRR
NMOS Static drain-source on-state
resistance
vs Supply voltage and ambient temperature
vs Frequency
Power supply rejection ratio
Typical operation
25
26, 27, 28
29
PFM
Line transient response
PWM
30
Mode transition PFM / forced PWM
AC - load regulation performance
Load transient response
Start-up
31
32, 33, 34
35, 36, 37, 38
39, 40
–
–
100
100
V
= 3.6 V
= 2.9 V
IN
V
= 2.9 V
IN
90
80
70
60
90
80
70
60
50
40
30
20
10
0
V
= 3.6 V
IN
V
= 4.2 V
IN
V
IN
V
= 4.2 V
IN
V
= 5 V
IN
V
= 5 V
IN
50
40
30
20
10
0
MODE = GND,
= 2.5V,
MODE = V
,
V
IN
OUT
L = 2.2 mH (LQM2HPN2R2MJ0)
V
= 2.5 V,
OUT
L = 2.2 mH (LQM2HPN2R2MJ0)
= 4.7 mF
C
= 4.7 mF
OUT
C
OUT
0.1
1
I
10
- Output Current - mA
100
1000
1
10
I - Output Current - mA
O
100
1000
O
Figure 1. Efficiency PFM/PWM Mode 2.5V Output Voltage
Figure 2. Efficiency Forced PWM Mode 2.5V Output
Voltage
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100
90
80
70
60
50
100
90
80
70
60
50
40
30
20
10
0
V
= 2.3 V
IN
V
= 2.3 V
IN
V
= 2.7 V
V
= 2.7 V
IN
IN
V
= 3.3 V
IN
V
= 3.3 V
V
= 3.6 V
IN
IN
V
= 3.6 V
IN
V
= 4.2 V
IN
V
= 4.2 V
V
= 5 V
IN
IN
40
30
20
V
= 5 V
IN
MODE = GND,
= 1.8 V,
MODE = V
,
IN
= 1.8 V,
V
OUT
V
OUT
L = 2.2 mH (MIPSA25202R2),
= 4.7 mF
L = 2.2 mH (MIPSA25202R2),
= 4.7 mF
C
10
0
OUT
C
OUT
0.1
1
I
10
- Output Current - mA
100
1000
1
10
100
1000
I - Output Current - mA
O
O
Figure 3. Efficiency PFM/PWM MODE 1.8V Output Voltage
Figure 4. Efficiency Forced PWM Mode 1.8V Output
voltage
100
90
80
70
60
100
90
80
70
60
50
V
= 2.3 V
IN
V
= 2.3 V
IN
V
= 2.7 V
IN
V
= 2.7 V
IN
V
= 3.6 V
IN
V
= 3.6 V
IN
V
IN
= 4.2 V
V
= 4.2 V
50
IN
V
= 5 V
IN
V
= 5 V
40
30
20
10
0
40
30
20
IN
MODE = GND,
= 1.2 V,
MODE = V
,
IN
= 1.2 V,
V
OUT
L = 2.2 mH MIPSZ2012 2R2 (2012 size),
V
OUT
L = 2.2 mH MIPSZ2012 2R2 (2012 size),
= 4.7 mF
C
= 4.7 mF
10
0
OUT
C
OUT
1
10 100
- Output Current - mA
1000
0.1
1
10
- Output Current - mA
100
1000
I
I
O
O
Figure 5. Efficiency PFM/PWM Mode 1.2V Output voltage
Figure 6. Efficiency Forced PWM Mode 1.2V Output
Voltage
8
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2.575
2.55
90
85
80
75
70
65
60
55
50
MODE = V
,
IN
= 2.5 V,
V
OUT
L = 1 mH,
C
T
= 4.7 mF,
OUT
MIPSA25202R2
L = 2.2 mH
(2.5x2x1.2mm3)
= 25°C
A
V
= 3.3 V
MIPSD1R0
L = 1 mH 0805
(2x1.25x1mm3)
IN
2.525
2.5
V
= 3.6 V
LQM2HPN1R0MJ0
L = 1 mH
(2.5x2x1.2mm3)
IN
MIPSZ2012D2R2
L = 2.2 mH 0805
(2x1.25x1mm3)
V
= 4.2 V
IN
V
= 5 V
LQM21PN2R2
L = 2.2 mH 0805
(2x1.25x0.55mm3)
2.475
IN
MODE = GND,
C
C
V
= 2.2 mF (0402),
IN
= 4.7 mF (0402),
OUT
2.45
= 1.8 V,
100
OUT
V
= 3.6 V
IN
2.425
0.1
1
I
10
- Output Current - mA
100
1000
0.1
1
10
1000
I
- Output Current - mA
O
O
Figure 7. Comparison Efficiency vs Inductor Value and
Size
Figure 8. 2.5V Output Voltage Accuracy forced PWM Mode
2.575
1.854
MODE = GND,
= 2.5 V,
MODE = GND,
= 1.8 V,
V
OUT
L = 1 mH,
V
OUT
L = 2.2 mH,
2.55
2.525
2.5
1.836
1.818
1.8
C
T
= 4.7 mF,
OUT
= 25°C
C
T
= 4.7 mF,
OUT
= 25°C
A
A
V
= 4.2 V
= 3.3 V
IN
V
= 5 V
IN
V
= 3.6 V
= 4.2 V
V
= 3.3 V
IN
IN
V
IN
V
= 3.6 V
IN
V
IN
2.475
1.782
1.764
1.746
V
= 5 V
IN
2.45
2.425
0.1
1
10
- Output Current - mA
100
1000
0.01
0.1
1
10
- Output Current - mA
100
1000
I
I
O
O
Figure 9. 2.5V Output Voltage Accuracy PFM/PWM Mode
Figure 10. 1.8V Output Voltage Accuracy PFM/PWM Mode
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1.854
1.836
1.818
1.8
1.236
1.224
1.212
1.2
MODE = V
,
MODE = V
,
IN
= 1.2 V,
IN
= 1.8 V,
V
V
OUT
L = 2.2 mH,
OUT
L = 1 mH,
C
T
= 4.7 mF,
C
T
= 4.7 mF,
OUT
= 25°C
OUT
= 25°C
V
= 3.3 V
A
A
V
= 3.3 V
IN
IN
V
= 3.6 V
IN
V
= 3.6 V
IN
V
= 5 V
IN
V
= 4.2 V
V
IN
V
= 4.2 V
1.188
1.782
IN
= 5 V
IN
1.176
1.164
1.764
1.746
0.1
1
I
10
- Output Current - mA
100
1000
0.1
1
I
10
- Output Current - mA
100
1000
O
O
Figure 11. 1.8V Output Voltage Accuracy Forced PWM
MODE
Figure 12. 1.2V Output Voltage Accuracy Forced PWM
MODE
1.236
4000
MODE = GND,
= 1.2 V,
V
OUT
L = 2.2 mH,
3500
3000
2500
2000
1500
1000
V
= 5 V
IN
1.224
1.212
1.2
C
T
= 4.7 mF,
OUT
= 25°C
V
= 4.2 V
IN
A
V
= 3.6 V
V
= 3.3 V
IN
IN
V
= 3.3 V
IN
V
= 3.6 V
IN
V
= 4.2 V
IN
1.188
V
= 5 V
IN
MODE = GND,
= 1.8 V,
V
V
= 2.7 V
OUT
L = 2.2 mH,
IN
1.176
1.164
V
= 2.3 V
500
0
C
T
= 4.7 mF,
IN
OUT
= 25°C
A
0
100
200 300
- Output Current - mA
400
500
0.01
0.1
1
10
- Output Current - mA
100
1000
I
I
O
O
Figure 13. 1.2V Output Voltage Accuracy PFM/PWM MODE
Figure 14. Switching Frequency vs Output Current, 1.8V
Output Voltage MODE = GND
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4000
4000
V
= 5 V
V
IN
V
= 5 V
V
IN
= 4.2 V
V
= 4.2 V
IN
3500
3000
2500
2000
1500
3500
3000
2500
2000
1500
1000
500
IN
V
= 3.6 V
V
= 3.6 V
V
IN
IN
= 3.3 V
IN
= 3.3 V
IN
MODE = GND,
= 1.8 V,
MODE = V
,
IN
= 1.8 V,
V
1000
500
0
V
IN
= 2.7 V
OUT
L = 1 mH,
V
OUT
L = 2.2 mH,
V
= 2.7 V
IN
V
= 2.3 V
V
= 2.3 V
100
IN
IN
C
T
= 4.7 mF,
OUT
= 25°C
C
T
= 4.7 mF,
OUT
= 25°C
A
A
0
0
100
I
200 300
- Output Current - mA
400
500
0
200
- Output Current - mA
O
300
400
500
I
O
Figure 15. Switching Frequency vs Output Current, 1.8V
Output Voltage MODE = GND
Figure 16. Switching Frequency vs Output Current, 1.8V
Output Voltage MODE = VIN
4000
4000
V
= 5 V
V
MODE = V
,
IN
MODE = GND,
= 2.5 V,
IN
= 2.5 V,
V
= 4.2 V
V
OUT
L = 2.2 mH,
IN
OUT
L = 2.2 mH,
3500
3000
2500
2000
1500
1000
3500
3000
2500
2000
1500
V
= 5 V
IN
V
= 3.6 V
IN
V
= 4.2 V
C
T
= 4.7 mF,
C
T
= 4.7 mF,
IN
OUT
= 25°C
OUT
= 25°C
V
= 3.3 V
IN
A
A
V
= 3.6 V
IN
V
= 3.3 V
IN
1000
500
0
V
= 3 V
100
V
= 3 V
500
0
IN
IN
0
200 300
- Output Current - mA
400
500
0
100
200 300
- Output Current - mA
400
500
I
I
O
O
Figure 17. Switching Frequency vs Output Current, 2.5V
Output Voltage MODE = GND
Figure 18. Switching Frequency vs Output Current, 2.5V
Output Voltage MODE = VIN
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3000
2500
2000
1500
1000
3500
3000
2500
2000
1500
1000
500
V
= 5 V
IN
V
= 5 V
IN
V
= 4.2 V
IN
V
= 4.2 V
V
IN
V
= 3.3 V
IN
= 3.6 V
IN
V
= 3.6 V
V
IN
V
= 3.3 V
IN
= 2.7 V
IN
V
= 2.3 V
IN
V
= 2.7 V
V
= 2 V
IN
IN
MODE = GND,
= 1.2 V,
V
= 2.3 V
IN
MODE = V
,
IN
= 1.2 V,
V
OUT
V
= 2 V
IN
V
OUT
L = 2.2 mH,
500
0
L = 2.2 mH,
C
T
= 4.7 mF,
OUT
C
T
= 4.7 mF,
OUT
= 25°C
A
= 25°C
A
0
0
100
200 300
- Output Current - mA
400
500
0
100
200
300
400
500
I
I - Output Current - mA
O
O
Figure 19. Switching Frequency vs Output Current, 1.2V
Output Voltage MODE = GND
Figure 20. Switching Frequency vs Output Current, 1.2V
Output Voltage MODE = VIN
35
30
25
20
15
10
0.2
0.18
0.16
0.14
0.12
0.1
T
= 85°C
T
= 85°C
A
A
T
= 60°C
T
= 25°C
A
A
0.08
0.06
0.04
T = -40°C
A
T
A
= -40°C
T
= 25°C
T
= 60°C
A
A
0.02
0
2
2.5
3
3.5
4
4.5
- Input Voltage - V
5
5.5
6
2
2.5
3
3.5
4
4.5
- Input Voltage - V
5
5.5
6
V
V
IN
IN
Figure 21. Quiescent Current IQ vs Ambient Temperature
TA
Figure 22. Shutdown Current ISD vs Ambient Temperature
TA
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2
1.8
1.6
1.4
1.2
1
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0
PMOS
NMOS
T
= 85°C
T
T
= 85°C
A
A
= 60°C
T
T
= 60°C
A
A
= 25°C
T
= 25°C
A
A
T
A
= -40°C
T
A
= -40°C
0.8
0.6
0.4
0.2
0
2
2.5
3
3.5
4
4.5
- Input Voltage - V
5
5.5
6
2
2.5
3
3.5 4.5
- Input Voltage - V
4
5
5.5
6
V
V
IN
IN
Figure 23. PMOS RDSON vs Supply Voltage VIN and
Ambient Temperature TA
Figure 24. NMOS RDSON vs Supply Voltage VIN and
Ambient Temperature TA
100
90
80
70
60
SW
2 V/div
I
= 50 mA,
OUT
MODE = 0,
forced PWM
V
= 3.6V
IN
C
= 4.7 mF
V
= 2.5V
OUT
L = 1 mH
OUT
20 mV/Div
I
= 50 mA,
OUT
MODE = 1,
PFM/PWM
50
40
30
20
10
0
MODE = GND
I
= 150 mA,
OUT
PWM Mode
I
= 10 mA
OUT
V
V
= 3.6 V,
I
IN
L
= 1.8 V,
= 2.2 mF,
= 4.7 mF,
200 mA/Div
OUT
C
C
IN
OUT
L = 2.2 mH
t - Time - 1 ms/div
10
100
1k 10k
f - Frequency - kHz
100k
1M
Figure 25. TPS62231 1.8V PSRR
Figure 26. PFM Mode Operation IOUT = 10mA
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V
= 3.6 V
V
= 3.6 V
MODE = V
IN
IN
IN
C
= 4.7 mF
V
= 2.5V
I
= 10 mA
C
= 4.7 mF
OUT
OUT
V
= 2.5 V
OUT
OUT
OUT
20 mV/div
L = 1 mH
L = 2.2 mH
20 mV/div
MODE = GND
SW
SW
I
= 10 mA
OUT
2 V/div
2 V/div
I
L
I
200 mA/div
L
200 mA/div
t - Time - 500 ns/div
t - Time - 1 ms/div
Figure 27. PFM Mode Operation IOUT = 10mA
Figure 28. Forced PWM Mode Operation IOUT = 10mA
V
= 3.6 V to 4.2 V
IN
V
= 3.6 V to 4.2 V
200 mV/div
IN
200 mV/div
C
= 4.7 mF
V
= 1.8 V
C
= 4.7 mF
OUT
L = 2.2 mH
MODE = V
V
= 1.8 V
OUT
20 mV/div
OUT
L = 2.2 mH
MODE = GND
= 50 mA
OUT
20 mV/div
IN
= 50 mA
I
I
OUT
OUT
t - Time - 10 ms/div
t - Time - 100 ms/div
Figure 29. Line Transient Response PFM Mode
Figure 30. Line Transient Response PWM Mode
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V
= 3.6 V
IN
V
= 2.5 V
MODE: 0 V to 3.6 V
2 V/div
Forced PWM
Mode Operation
OUT
50 mV/div
C
= 4.7 mF
OUT
L = 2.2 mH
MODE = GND
PFM Mode Operation
V
SW
2 V/div
I
= 5 mA to 200 mA
OUT
sinusoidal
100 mA/div
V
= 3.6 V,
IN
C
= 4.7 mF
OUT
L = 1 mH
= 10 mA
I
COIL
200 mA/div
I
OUT
I
L
200 mA/div
V
= 1.8 V
OUT
20 mV/div
t - Time - 1 ms/div
Figure 31. Mode Transition PFM / Forced PWM Mode
t - Time - 5 ms/div
Figure 32. AC – Load Regulation Performance 2.5V VOUT
PFM Mode
V
= 3.6 V
V
= 2.5 V
IN
OUT
C
= 4.7 mF
50 mV/div
OUT
V
= 1.8 V
OUT
50 mV/div
L = 2.2 mH
MODE = GND
V
= 3.6 V
IN
I
= 5 mA to 150 mA, 50 kHz
C
= 4.7 mF
OUT
IOUT = 5mA to 200mA
sinusoidal
OUT
L = 2.2 mH
MODE = V
sinusoidal 100 mA/div
100mA/Div
IN
I
L
I
L
200 mA/div
200 mA/div
t - Time - 4 ms/div
t - Time - 5 ms/div
Figure 33. AC – Load Regulation Performance 2.5V VOUT
PWM Mode
Figure 34. AC – Load Regulation Performance 1.8V VOUT
PFM Mode
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V
= 2.5 V
V
= 2.5 V
OUT
50 mV/div
OUT
50 mV/div
V
= 3.6 V
IN
V
= 3.6 V
IN
C
= 4.7 mF
OUT
L = 1 mH
MODE = GND
C
= 4.7 mF
OUT
I
= 5 mA to 200 mA
100 mA/div
I
= 5 mA to 200 mA
100 mA/div
OUT
OUT
L = 1 mH
MODE = V
IN
I
L
I
L
200 mA/div
200 mA/div
t - Time - 5 ms/div
t - Time - 5 ms/div
Figure 35. Load Transient Response 5mA to 200mA PFM
to PWM Mode, VOUT 2.5V
Figure 36. Load Transient Response 5mA to 200mA,
Forced PWM Mode, VOUT 2.5V
V
= 3.6 V
V
= 3.6 V
IN
IN
C
= 4.7 mF
C
= 4.7 mF
OUT
L = 2.2 mH
MODE = GND
OUT
L = 2.2 mH
MODE = V
V
= 1.8 V
V
= 1.8 V
OUT
50 mV/div
OUT
50 mV/div
IN
I
= 5 mA to 150 mA
I
= 5 mA to 150 mA
OUT
OUT
100 mA/div
100 mA/div
I
L
I
L
200 mA/div
200 mA/div
t - Time - 10 ms/div
t - Time - 10 ms/div
Figure 37. Load Transient Response 5mA to 150mA, PFM
to PWM Mode, VOUT 1.8V
Figure 38. Load Transient Response 5mA to 150mA,
Forced PWM Mode, VOUT 1.8V
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EN 2 V/div
EN
2 V/div
SW
2 V/div
V
= 1.8 V
V
Pre Bias = 1V
OUT
1 V/div
OUT
V
= 3.6 V
IN
V
= 0 V to 2.5 V
C
= 4.7 mF
OUT
OUT
1 V/div
L = 1 mH
SW 5 V/div
MODE = GND
Load = 20 R
V
= 3.6 V
IN
C
= 4.7 mF
OUT
L = 2.2 mH
MODE = GND
= 0 mA
I
IN
IL
50 mA/div
200 mA/div
I
OUT
Time Base - 20 ms/div
Figure 40. Startup in 1V Pre-biased Output
t - Time - 20 ms/div
Figure 39. Start Up into 20Ω Load, VOUT 2.5V
DETAILED DESCRIPTION
The TPS6223X synchronous step down converter family includes a unique hysteric PWM controller scheme
which enables switch frequencies over 3MHz, excellent transient and ac load regulation as well as operation with
tiny and cost competitive external components.
The controller topology supports forced PWM Mode as well as Power Save Mode operation. Power Save Mode
operation reduces the quiescent current consumption down to 22µA and ensures high conversion efficiency at
light loads by skipping switch pulses.
In forced PWM Mode, the device operates on a quasi fixed frequency, avoids pulse skipping and allows therefore
easy filtering of the switch noise by external filter components.
The TPS6223X devices offer fixed output voltage options featuring smallest solution size by using only three
external components.
The internal switch current limit of typical 850mA supports output currents of up to 500mA, depending on the
operating condition.
A significant advantage of TPS6223X compared to other hysteretic PWM controller topologies is its excellent DC
and AC load regulation capability in combination with low output voltage ripple over the entire load range which
makes this part well suited for audio and RF applications.
OPERATION
Once the output voltage falls below the threshold of the error comparator a switch pulse is initiated and the high
side switch is turned on. It remains turned on until a minimum on time of TONmin expires and the output voltage
trips the threshold of the error comparator or the inductor current reaches the high side switch current limit. Once
the high side switch turns off, the low side switch rectifier is turned on and the inductor current ramps down until
the high side switch turns on again or the inductor current reaches zero.
In forced PWM Mode operation negative inductor current is allowed to enable continuous conduction mode even
at no load condition.
POWER SAVE MODE
Connecting the MODE pin to GND enables the automatic PWM and power-save mode operation. The converter
operates in quasi fixed frequency PWM mode at moderate to heavy loads and in the PFM (Pulse Frequency
Modulation) mode during light loads, which maintains high efficiency over a wide load current range.
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In PFM Mode the device starts to skip switch pulses and generates only single pulses with an on time of TONmin
The PFM Mode frequency depends on the load current and the external inductor and output capacitor values.
The PFM Mode of TPS6223X is optimized for low output voltage ripple if small and tiny external components are
used. Even at low output currents, the PFM frequency is above the audible noise spectrum and makes this
operation mode suitable for audio applications.
.
The on time TONmin can be estimated to:
V
OUT
T
=
´ 260 ns
ONmin
V
IN
(1)
(2)
Therefore the peak inductor current in PFM mode is approximately:
(V - V
)
OUT
IN
I
=
´ T
ONmin
LPFMpeak
L
With
TON: High side switch on time [ns]
VIN: Input voltage [V]
VOUT: Output voltage [V]
L : Inductance [µH]
ILPFMpeak : PFM inductor peak current [mA]
FORCED PWM MODE
Pulling the MODE pin high forces the converter to operate in a continuous conduction PWM mode even at light
load currents. The advantage is that the converter operates with a quasi fixed frequency that allows simple
filtering of the switching frequency for noise-sensitive applications. In this mode, the efficiency is lower compared
to the power-save mode during light loads.
For additional flexibility, it is possible to switch from power-save mode to forced PWM mode during operation.
This allows efficient power management by adjusting the operation of the converter to the specific system
requirements.
100% DUTY CYCLE LOW DROPOUT OPERATION
The device starts to enter 100% duty cycle mode once the input voltage comes close to the nominal output
voltage. In order to maintain the output voltage, the High Side switch is turned on 100% for one or more cycles.
With further decreasing VIN the High Side MOSFET switch is turned on completely. In this case the converter
offers a low input-to-output voltage difference. This is particularly useful in battery-powered applications to
achieve longest operation time by taking full advantage of the whole battery voltage range.
The minimum input voltage to maintain regulation depends on the load current and output voltage, and can be
calculated as:
Vinmin = Voutmax + Ioutmax
´
RDSonmax + RL
(
)
(3)
With:
Ioutmax = maximum output current plus inductor ripple current
RDSonmax = maximum P-channel switch RDSon.
RL = DC resistance of the inductor
Voutmax = nominal output voltage plus maximum output voltage tolerance
UNDER-VOLTAGE LOCKOUT
The under voltage lockout circuit prevents the device from misoperation at low input voltages. It prevents the
converter from turning on the switch or rectifier MOSFET under undefined conditions. The TPS6223X devices
have a UVLO threshold set to 1.8V (typical). Fully functional operation is permitted for input voltage down to the
falling UVLO threshold level. The converter starts operation again once the input voltage trips the rising UVLO
threshold level.
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SOFT START
The TPS6223X has an internal soft-start circuit that controls the ramp up of the output voltage and limits the
inrush current during start-up. This limits input voltage drops when a battery or a high-impedance power source
is connected to the input of the converter.
The soft-start system generates a monotonic ramp up of the output voltage and reaches the nominal output
voltage typically 100µs after EN pin was pulled high.
Should the output voltage not have reached its target value by this time, such as in the case of heavy load, the
converter then operates in a current limit mode set by its switch current limits.
TPS6223X is able to start into a pre-biased output capacitor. The converter starts with the applied bias voltage
and ramps the output voltage to its nominal value.
ENABLE / SHUTDOWN
The device starts operation when EN is set high and starts up with the soft start as previously described. For
proper operation, the EN pin must be terminated and must not be left floating.
Pulling the EN pin low forces the device into shutdown, with a shutdown quiescent current of typically 0.1µA. In
this mode, the P and N-channel MOSFETs are turned off, the internal resistor feedback divider is disconnected,
and the entire internal-control circuitry is switched off.
The EN input can be used to control power sequencing in a system with various DC/DC converters. The EN pin
can be connected to the output of another converter, to drive the EN pin high and getting a sequencing of supply
rails.
SHORT-CIRCUIT PROTECTION
The TPS6223X integrates a High Side and Low Side MOSFET current limit to protect the device against heavy
load or short circuit. The current in the switches is monitored by current limit comparators. When the current in
the P-channel MOSFET reaches its current limit, the P-channel MOSFET is turned off and the N-channel
MOSFET is turned on to ramp down the current in the inductor. The High Side MOSFET switch can only turn on
again, once the current in the Low Side MOSFET switch has decreased below the threshold of its current limit
comparator.
THERMAL SHUTDOWN
As soon as the junction temperature, TJ, exceeds 150°C (typical) the device goes into thermal shutdown. In this
mode, the High Side and Low Side MOSFETs are turned-off. The device continues its operation when the
junction temperature falls below the thermal shutdown hysteresis.
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APPLICATION INFORMATION
L
V
IN
TPS62230
1/2.2 mH
V
2.7 V - 6 V
OUT
VIN
SW
FB
2.5 V
EN
C
C
OUT
4.7 mF
IN
2.2
MODE
GND
mF
Figure 41. TPS62230 2.5V Output
L
V
IN
TPS62231
1/2.2 mH
2.05 V - 6 V
V
OUT
SW
FB
VIN
EN
1.8 V
C
C
IN
OUT
4.7 mF
GND
2.2 mF
MODE
Figure 42. TPS62231 1.8V Output
L
V
IN
TPS62232
1/2.2 mH
2.05 V - 6 V
V
OUT
SW
FB
VIN
EN
1.2 V
C
C
IN
OUT
4.7 mF
GND
2.2 mF
MODE
Figure 43. TPS62232 1.2V Output
OUTPUT FILTER DESIGN (INDUCTOR AND OUTPUT CAPACITOR)
The TPS6223X is optimized to operate with effective inductance values in the range of 0.7µH to 4.3µH and with
effective output capacitance in the range of 2.0µF to 15µF. The internal compensation is optimized to operate
with an output filter of L = 1.0µH/2.2µH and COUT = 4.7µF. Larger or smaller inductor/capacitor values can be
used to optimize the performance of the device for specific operation conditions. For more details, see the
CHECKING LOOP STABILITY section.
INDUCTOR SELECTION
The inductor value affects its peak-to-peak ripple current, the PWM-to-PFM transition point, the output voltage
ripple and the efficiency. The selected inductor has to be rated for its dc resistance and saturation current. The
inductor ripple current (ΔIL) decreases with higher inductance and increases with higher VIN or VOUT. Equation 4
calculates the maximum inductor current under static load conditions. The saturation current of the inductor
should be rated higher than the maximum inductor current as calculated with Equation 5. This is recommended
because during heavy load transient the inductor current will rise above the calculated value.
Vout
1-
Vin
DIL = Vout ´
L ´ ¦
(4)
DI
L
I
= I
+
Lmax
outmax
2
(5)
With:
f = Switching Frequency
L = Inductor Value
ΔIL= Peak to Peak inductor ripple current
ILmax = Maximum Inductor current
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In high-frequency converter applications, the efficiency is essentially affected by the inductor AC resistance (i.e.,
quality factor) and to a smaller extent by the inductor DCR value. To achieve high efficiency operation, care
should be taken in selecting inductors featuring a quality factor above 25 at the switching frequency. Increasing
the inductor value produces lower RMS currents, but degrades transient response. For a given physical inductor
size, increased inductance usually results in an inductor with lower saturation current.
The total losses of the coil consist of both the losses in the DC resistance, R(DC), and the following
frequency-dependent components:
•
•
•
•
The losses in the core material (magnetic hysteresis loss, especially at high switching frequencies)
Additional losses in the conductor from the skin effect (current displacement at high frequencies)
Magnetic field losses of the neighboring windings (proximity effect)
Radiation losses
The following inductor series from different suppliers have been used with the TPS6223X converters.
Table 1. List of inductors
INDUCTANCE
DIMENSIONS
[mm3]
INDUCTOR TYPE
SUPPLIER
[µH]
1.0/2.2
2.2
2.5 × 2.0 × 1.2
2.0 × 1.2 × 0.55
2.0 × 1.2 × 1.0
2.0 × 2.5 × 1.2
2.0 × 1.2 × 1.0
LQM2HPN1R0MJ0
LQM21PN2R2
MIPSZ2012
Murata
Murata
1.0/2.2
1.0/2.2
1.0/2.2
FDK
MIPSA2520
FDK
KSLI2012 series
Hitachi Metal
OUTPUT CAPACITOR SELECTION
The unique hysteric PWM control scheme of the TPS62230 allows the use of tiny ceramic capacitors. Ceramic
capacitors with low ESR values have the lowest output voltage ripple and are recommended. The output
capacitor requires either an X7R or X5R dielectric. Y5V and Z5U dielectric capacitors, aside from their wide
variation in capacitance over temperature, become resistive at high frequencies.
At light load currents the converter operate in Power Save Mode and the output voltage ripple is dependent on
the output capacitor value and the PFM peak inductor current. Higher output capacitor values minimize the
voltage ripple in PFM Mode and tighten DC output accuracy in PFM Mode.
INPUT CAPACITOR SELECTION
Because of the nature of the buck converter having a pulsating input current, a low ESR input capacitor is
required for best input voltage filtering and minimizing the interference with other circuits caused by high input
voltage spikes. For most applications a 2.2µF to 4.7µF ceramic capacitor is recommended. The input capacitor
can be increased without any limit for better input voltage filtering. Because ceramic capacitor loses up to 80% of
its initial capacitance at 5V, it is recommended to use 4.7µF input capacitors for input voltages > 4.5V.
Take care when using only small ceramic input capacitors. When a ceramic capacitor is used at the input and the
power is being supplied through long wires, such as from a wall adapter, a load step at the output or VIN step on
the input can induce ringing at the VIN pin. This ringing can couple to the output and be mistaken as loop
instability or could even damage the part by exceeding the maximum ratings.
Table 2 shows a list of tested input/output capacitors.
Copyright © 2009, Texas Instruments Incorporated
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Product Folder Link(s): TPS62230 TPS62231 TPS62232
TPS62230
TPS62231
TPS62232
SLVS941–APRIL 2009 ..................................................................................................................................................................................................... www.ti.com
Table 2. List of Capacitor
CAPACITANCE
SIZE
CAPACITOR TYPE
SUPPLIER
[µF]
2.2
4.7
4.7
4.7
4.7
0402
0402
0402
0402
0603
GRM155R60J225
AMK105BJ475MV
GRM155R60J475
CL05A475MQ5NRNC
GRM188R60J475
Murata
Taiyo Yuden
Murata
Samsung
Murata
CHECKING LOOP STABILITY
The first step of circuit and stability evaluation is to look from a steady-state perspective at the following signals:
•
•
•
Switching node, SW
Inductor current, IL
Output ripple voltage, VOUT(AC)
These are the basic signals that need to be measured when evaluating a switching converter. When the
switching waveform shows large duty cycle jitter or the output voltage or inductor current shows oscillations, the
regulation loop may be unstable. This is often a result of board layout and/or L-C combination.
As a next step in the evaluation of the regulation loop, the load transient response is tested. The time between
the application of the load transient and the turn on of the P-channel MOSFET, the output capacitor must supply
all of the current required by the load. VOUT immediately shifts by an amount equal to ΔI(LOAD) x ESR, where ESR
is the effective series resistance of COUT. ΔI(LOAD) begins to charge or discharge CO generating a feedback error
signal used by the regulator to return VOUT to its steady-state value. The results are most easily interpreted when
the device operates in PWM mode.
During this recovery time, VOUT can be monitored for settling time, overshoot or ringing that helps judge the
converter’s stability. Without any ringing, the loop has usually more than 45° of phase margin.
Because the damping factor of the circuitry is directly related to several resistive parameters (e.g., MOSFET
rDS(on)) that are temperature dependant, the loop stability analysis has to be done over the input voltage range,
load current range, and temperature range.
LAYOUT CONSIDERATIONS
As for all switching power supplies, the layout is an important step in the design. Proper function of the device
demands careful attention to PCB layout. Care must be taken in board layout to get the specified performance. If
the layout is not carefully done, the regulator could show poor line and/or load regulation, stability issues as well
as EMI problems. It is critical to provide a low inductance, impedance ground path. Therefore, use wide and
short traces for the main current paths. The input capacitor should be placed as close as possible to the IC pins
as well as the inductor and output capacitor.
Use a common Power GND node and a different node for the Signal GND to minimize the effects of ground
noise. Keep the common path to the GND PIN, which returns the small signal components and the high current
of the output capacitors as short as possible to avoid ground noise. The FB line should be connected to the
output capacitor and routed away from noisy components and traces (e.g. SW line).
22
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Copyright © 2009, Texas Instruments Incorporated
Product Folder Link(s): TPS62230 TPS62231 TPS62232
TPS62230
TPS62231
TPS62232
www.ti.com ..................................................................................................................................................................................................... SLVS941–APRIL 2009
Total area
L1
is less than
12mm²
V IN
C
1
C2
GND
VOUT
Figure 44. Recommended PCB Layout for TPS6223X
Copyright © 2009, Texas Instruments Incorporated
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Product Folder Link(s): TPS62230 TPS62231 TPS62232
PACKAGE OPTION ADDENDUM
www.ti.com
6-May-2009
PACKAGING INFORMATION
Orderable Device
TPS62230DRYR
TPS62230DRYT
TPS62231DRYR
TPS62231DRYT
TPS62232DRYR
TPS62232DRYT
Status (1)
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
Package Package
Pins Package Eco Plan (2) Lead/Ball Finish MSL Peak Temp (3)
Qty
Type
Drawing
SON
DRY
6
6
6
6
6
6
5000 Green (RoHS & CU NIPDAU Level-1-260C-UNLIM
no Sb/Br)
SON
SON
SON
SON
SON
DRY
DRY
DRY
DRY
DRY
250 Green (RoHS & CU NIPDAU Level-1-260C-UNLIM
no Sb/Br)
5000 Green (RoHS & CU NIPDAU Level-1-260C-UNLIM
no Sb/Br)
250 Green (RoHS & CU NIPDAU Level-1-260C-UNLIM
no Sb/Br)
5000 Green (RoHS & CU NIPDAU Level-1-260C-UNLIM
no Sb/Br)
250 Green (RoHS & CU NIPDAU Level-1-260C-UNLIM
no Sb/Br)
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check
http://www.ti.com/productcontent for the latest availability information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements
for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered
at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and
package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS
compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame
retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder
temperature.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is
provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the
accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take
reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on
incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited
information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI
to Customer on an annual basis.
Addendum-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
6-May-2009
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0 (mm)
B0 (mm)
K0 (mm)
P1
W
Pin1
Diameter Width
(mm) W1 (mm)
(mm) (mm) Quadrant
TPS62230DRYR
TPS62230DRYT
TPS62231DRYR
TPS62231DRYT
TPS62232DRYR
TPS62232DRYT
SON
SON
SON
SON
SON
SON
DRY
DRY
DRY
DRY
DRY
DRY
6
6
6
6
6
6
5000
250
179.0
179.0
179.0
179.0
179.0
179.0
8.4
8.4
8.4
8.4
8.4
8.4
1.2
1.2
1.2
1.2
1.2
1.2
1.65
1.65
1.65
1.65
1.65
1.65
0.7
0.7
0.7
0.7
0.7
0.7
4.0
4.0
4.0
4.0
4.0
4.0
8.0
8.0
8.0
8.0
8.0
8.0
Q1
Q1
Q1
Q1
Q1
Q1
5000
250
5000
250
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
6-May-2009
*All dimensions are nominal
Device
Package Type Package Drawing Pins
SPQ
Length (mm) Width (mm) Height (mm)
TPS62230DRYR
TPS62230DRYT
TPS62231DRYR
TPS62231DRYT
TPS62232DRYR
TPS62232DRYT
SON
SON
SON
SON
SON
SON
DRY
DRY
DRY
DRY
DRY
DRY
6
6
6
6
6
6
5000
250
220.0
220.0
220.0
220.0
220.0
220.0
205.0
205.0
205.0
205.0
205.0
205.0
50.0
50.0
50.0
50.0
50.0
50.0
5000
250
5000
250
Pack Materials-Page 2
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