V62/09616-01XE [TI]
增强型产品 4:1 高速多路复用器 | D | 14 | -55 to 125;
| 型号： | V62/09616-01XE |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | 增强型产品 4:1 高速多路复用器 | D | 14 | -55 to 125 开关 信号电路 复用器 复用器或开关 |
| 文件： | 总24页 (文件大小：853K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
OPA4872-EP
www.ti.com........................................................................................................................................................................................... SBOS444–DECEMBER 2008
4:1 HIGH-SPEED MULTIPLEXER
1
FEATURES
APPLICATIONS
•
•
•
•
Video Router
LCD and Plasma Display
High Speed PGA
2
•
500-MHz Small-Signal Bandwidth
•
•
•
•
•
•
•
•
•
500-MHz, 2-VPP Bandwidth
0.1-dB Gain Flatness to 120 MHz
10-ns Channel-Switching Time
Low Switching Glitch: 40 mVPP
2300-V/µs Slew Rate
Drop-In Upgrade to AD8174
SUPPORTS DEFENSE, AEROSPACE,
AND MEDICAL APPLICATIONS
•
•
•
•
0.035%/0.005° Differential Gain, Phase
Quiescent Current = 10.6 mA
1.1-mA Quiescent Current in Shutdown Mode
Controlled Baseline
One Assembly/Test Site
One Fabrication Site
Available in Military (–55°C/125°C),
Temperature Range(1)
88-dB Off Isolation in Disable or Shutdown
(10 MHz)
•
•
•
Extended Product Life Cycle
Extended Product-Change Notification
Product Traceability
(1) Additional temperature ranges available - contact factory
DESCRIPTION
The OPA4872 offers a very wideband 4:1 multiplexer in an SO-14 package. Using only 10.6 mA, the OPA4872
provides a user-settable output amplifier gain with greater than 500-MHz large-signal bandwidth (2 VPP). The
switching glitch is improved over earlier solutions using a new (patented) input stage switching approach. This
technique uses current steering as the input switch while maintaining an overall closed-loop design. The
OPA4872 exhibits an off isolation of 88dB in either Disable or Shutdown mode. With greater than 500-MHz
small-signal bandwidth at a gain of 2, the OPA4872 gives a typical 0.1-dB gain flatness to greater than 120 MHz.
System power may be optimized using the chip-enable feature for the OPA4872. Taking the chip enable (EN)
line high powers down the OPA4872 to less than 3.4 mA total supply current. Further power reduction to 1.1mA
quiescent current can be achieved by bringing the shutdown (SD) line high. Muxing multiple OPA4872s outputs
together, then using the chip enable to select which channels are active, increases the number of possible
inputs.
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
2
All trademarks are the property of their respective owners.
UNLESS OTHERWISE NOTED this document contains
PRODUCTION DATA information current as of publication date.
Products conform to specifications per the terms of Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2008, Texas Instruments Incorporated
OPA4872-EP
SBOS444–DECEMBER 2008........................................................................................................................................................................................... www.ti.com
+5V
50W
OPA695
G = 1V/V
-5V
+5V
523W
50W
IN0
+5V
OPA4872
SD
EN
50W
OPA695
G = 2V/V
-5V
IN1
IN2
511W
50W
511W
523W
+5V
To 50W Load
50W
523W
OPA695
G = 4V/V
Logic
-5V
453W
IN3
A0 A1
-5V
149W
+5V
50W
OPA695
G = 8V/V
-5V
402W
57.6W
2-Bit, High-Speed PGA, Greater Than 300MHz Channel Bandwidth
2
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OPA4872-EP
www.ti.com........................................................................................................................................................................................... SBOS444–DECEMBER 2008
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
ORDERING INFORMATION(1)
SPECIFIED
PACKAGE
TEMPERATURE
RANGE
PACKAGE
MARKING
ORDERING
NUMBER
TRANSPORT
MEDIA, QUANTITY
PACKAGE-LEAD
DESIGNATOR(2)
SO-14
D
–55°C to 125°C
OPA4872M
OPA4872MDREP
Tape and Reel, 2500
(1) For the most current package and ordering information see the Package Option Addendum at the end of this document, or see the TI
web site at www.ti.com.
(2) Package drawings, standard packing quantities, thermal data, symbolization, and PCB design guidelines are available at
www.ti.com/sc/package.
ABSOLUTE MAXIMUM RATINGS(1)
Over operating free-air temperature range, unless otherwise noted.
OPA4872
UNIT
Power supply
±6.5
V
Internal power dissipation
See Thermal Characteristics
Input voltage range
±VS
–65 to +125
+260
V
°C
°C
°C
°C
V
Storage temperature range
Lead temperature (soldering, 10s)
Junction temperature (TJ)
+150
Junction temperature: continuous operation, long-term reliability
Human body model (HBM)
+140
1500
ESD rating
Charged device model (CDM)
Machine model (MM)
1000
V
200
V
(1) Stresses above these ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may
degrade device reliability. These are stress ratings only, and functional operation of the device at these or any other conditions beyond
those specified is not implied.
PIN CONFIGURATION
SO-14
Top View
OPA4872
V+
IN0
14
13
1
2
3
4
5
6
7
OUT
GND
IN1
12 FB
11 SD
10 EN
GND
IN2
9
8
A1
A0
V-
IN3
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ELECTRICAL CHARACTERISTICS: VS = ±5 V
At TA = +25°C, G = +2 V/V, RF = 523 Ω, and RL = 150 Ω, unless otherwise noted.
MIN/MAX OVER
TYP
TEMPERATURE
–55°C to
MIN/
MAX
TEST
PARAMETER
CONDITIONS
+25°C
+25°C(2)
+125°C(3)
UNITS
LEVEL(1)
AC PERFORMANCE
Small-signal bandwidth
Bandwidth for 0.1 dB flatness
Large-signal bandwidth
Slew rate
VO = 500 mVPP, RL = 150 Ω
VO = 500 mVPP, RL = 150 Ω
VO = 2 VPP, RL = 150 Ω
4 V step
500
120
500
2300
1.25
15
MHz
MHz
MHz
V/µs
ns
min
typ
B
C
B
B
B
C
B
B
min
min
max
typ
Rise time and fall time
Settling time
4 V step
to 0.05%
to 0.1%
2 V step
ns
2 V step
14
ns
max
max
Channel switching time
Harmonic distortion
10
ns
G = +2 V/V, f = 10 MHz, VO = 2 VPP
RL = 150 Ω
2nd-harmonic
-60
-78
dBc
dBc
max
max
max
max
max
typ
B
B
B
B
B
C
C
C
C
3rd-harmonic
RL = 150 Ω
Input voltage noise
f > 100 kHz
4.5
nV/√Hz
pA/√Hz
pA/√Hz
%
Noninverting input current noise
Inverting input current noise
Differential gain
f > 100 kHz
4.0
f > 100 kHz
19
G = +2 V/V, PAL, VO = 1.4 VP
G = +2 V/V, PAL, VO = 1.4 VP
Three channels driven at 5 MHz, 1 VPP
Three channels driven at 30 MHz, 1 VPP
0.035
0.005
-80
Differential phase
°
typ
All hostile crosstalk, input-referred
dB
typ
-66
dB
typ
DC PERFORMANCE
Open-loop transimpedance (ZOL
)
VO = 0 V, RL = 100 Ω
VCM = 0 V
103
±1
92
±5
60
±10.5
±30
kΩ
mV
min
max
max
max
max
max
max
max
A
A
B
A
A
B
A
B
Input offset voltage
Average Input offset voltage drift
Input offset voltage matching
VCM = 0 V
µV/°C
mV
VCM = 0 V
±1
±4
±5
±10.5
±20
Noninverting input bias current
VCM = 0 V
±14
µA
Average noninverting input bias current
Inverting bias current
Average inverting input bias current
INPUT
VCM = 0 V
±48
nA/°C
µA
VCM = 0 V
±4
±18
±35
VCM = 0 V
±125
nA/°C
Common-mode input range (CMIR)
Common-mode rejection ratio (CMRR)
Input resistance
Each noninverting input
±2.7
56
±2.55
50
±2.4
43
V
min
min
A
A
VCM = 0 V, input-referred, noninverting input
dB
Noninverting
Channel enabled
open loop
2.5
70
MΩ
typ
typ
C
C
Inverting
Ω
Input capacitance
Noninverting
Channel selected
Channel deselected
Chip disabled
0.9
0.9
0.9
pF
pF
pF
typ
typ
typ
C
C
C
OUTPUT
Output voltage swing
R
L ≥ 1 kΩ
±4
±3.9
±3.55
±48
±3.55
±3.35
±38
V
V
min
min
min
typ
A
A
A
C
C
RL = 150 Ω
VO = 0 V
±3.7
±75
Output current
mA
mA
Ω
Short-circuit output current
Closed-Loop output impedance
Output shorted to ground
G = +2 V/V, f ≤ 100 kHz
±100
0.03
typ
(1) Test levels: (A) 100% tested at +25°C. Over temperature limits set by characterization and simulation. (B) Limits set by characterization
and simulation. (C) Typical value only for information.
(2) Junction temperature = ambient for +25°C tested specifications.
(3) Junction temperature = ambient at low temperature limit; junction temperature = ambient +9°C at high temperature limit for over
temperature specifications.
4
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www.ti.com........................................................................................................................................................................................... SBOS444–DECEMBER 2008
ELECTRICAL CHARACTERISTICS: VS = ±5 V (continued)
At TA = +25°C, G = +2 V/V, RF = 523 Ω, and RL = 150 Ω, unless otherwise noted.
MIN/MAX OVER
TYP
TEMPERATURE
–55°C to
MIN/
MAX
TEST
PARAMETER
CONDITIONS
+25°C
+25°C(2)
3.6
+125°C(3)
4.3
UNITS
LEVEL(1)
ENABLE (EN)
Power-down supply current
Disable time
VEN = 0 V
VIN = ±0.25 VDC
3.4
25
6
mA
ns
max
typ
typ
typ
typ
typ
A
C
C
C
C
C
Enable time
VIN = ±0.25 VDC
ns
Off isolation
G = +2 V/V, f = 10 MHz
88
14
2.5
dB
MΩ
pF
Output resistance in disable
Output capacitance in disable
DIGITAL INPUTS
Maximum logic 0
Minimum logic 1
A0, A1, EN, SD
0.8
2.0
40
0.8
2.0
55
V
max
min
max
typ
B
B
A
C
C
C
A0, A1, EN, SD
V
Logic input current
Output switching glitch
A0 , A1, EN, SD, input = 0V each line
Channel selection, at matched load
Channel disable, at matched load
Shutdown, at matched load
32
µA
mV
mV
mV
±20
±40
±40
typ
typ
SHUTDOWN
Shutdown supply current
Shutdown time
VSD = 0 V
VIN = ±0.25 VDC
1.1
75
15
88
14
2.5
1.3
2.0
mA
ns
max
typ
typ
typ
typ
typ
A
C
C
C
C
C
Enable time
VIN = ±0.25 VDC
ns
Off isolation
G = +2 V/V, f = 10 MHz
dB
MΩ
pF
Output resistance in shutdown
Output capacitance in shutdown
POWER SUPPLY
Specified operating voltage
Minimum operating voltage
Maximum operating voltage
Maximum quiescent current
Minimum quiescent current
±5
V
V
typ
min
max
max
min
min
min
C
B
A
A
A
A
A
±3.5
±6.0
11
±3.5
±6.0
12.5
8.25
–42
V
VS = ±5 V
VS = ±5 V
10.6
10.6
–56
–57
mA
mA
dB
dB
10
(+PSRR)
(–PSRR)
Input-referred
Input-referred
–50
–51
Power-supply rejection ratio
–43
THERMAL CHARACTERISTICS
Specified operating range, D package
Thermal resistance, θJA
D
–55 to +125
80
°C
typ
typ
C
C
Junction-to-ambient
SO-14
°C/W
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TYPICAL CHARACTERISTICS
At TA = +25°C, G = +2 V/V, RF = 523 Ω, and RL = 150 Ω, unless otherwise noted.
SMALL-SIGNAL FREQUENCY RESPONSE
SMALL-SIGNAL FREQUENCY RESPONSE
7
6
5
4
3
2
1
0
0.3
6
3
G = +1V/V
VO = 500mVPP
Bandwidth
0.2
0.1
0
Normalized
0
Flatness
-3
-6
-9
-12
G = +2V/V
-0.1
-0.2
-0.3
-0.4
VO = 500mVPP
RL = 150W
G = +4V/V
G = +2V/V
1M
10M
100M
Frequency (Hz)
1G
1M
10M
100M
Frequency (Hz)
Figure 2.
1G 2G
Figure 1.
LARGE-SIGNAL FREQUENCY RESPONSE
NONINVERTING PULSE RESPONSE
7
6
5
4
3
2
1
0
0.5
2.5
RL = 150W
G = +2V/V
0.4
0.3
2.0
VO = 2VPP
Large-Signal 4VPP
Right Scale
1.5
0.2
1.0
VO = 1VPP
Small-Signal 0.4VPP
Left Scale
0.1
0.5
VO = 0.5VPP
0
0
-0.1
-0.2
-0.3
-0.4
-0.5
-0.5
-1.0
-1.5
-2.0
-2.5
VO = 4VPP
-1
-2
-3
0
200M
400M
600M
800M
1G
Time (10ns/div)
Frequency (Hz)
Figure 3.
Figure 4.
RECOMMENDED RS vs CAPACITIVE LOAD
FREQUENCY RESPONSE vs CAPACITIVE LOAD
100
90
80
70
60
50
40
30
20
10
0
8
7
CL = 10pF
6
5
4
VI
3
75W
2
CL = 22pF
RS
+
75W
75W
75W
1
VO
1kW(1)
-
CL
523W
0
CL = 47pF
523W
-1
-2
-3
NOTE: (1) Optional.
CL = 100pF
1
10
Capacitive Load (pF)
Figure 5.
100
1000
1
10
100
300
Frequency (MHz)
Figure 6.
6
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TYPICAL CHARACTERISTICS (continued)
At TA = +25°C, G = +2 V/V, RF = 523 Ω, and RL = 150 Ω, unless otherwise noted.
HARMONIC DISTORTION vs LOAD RESISTANCE
HARMONIC DISTORTION vs SUPPLY VOLTAGE
-55
-60
-65
-70
-75
-80
-85
-90
-95
-55
-60
-65
-70
-75
-80
-85
-90
2nd-Harmonic
VO = 2VPP
f = 5MHz
2nd-Harmonic
3rd-Harmonic
3rd-Harmonic
VO = 2VPP
RL = 150W
dBc = dB Below Carrier
dBc = dB Below Carrier
f = 5MHz
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
100
1k
Supply Voltage (±VS)
Load Resistance (W)
Figure 7.
Figure 8.
HARMONIC DISTORTION vs FREQUENCY
HARMONIC DISTORTION vs OUTPUT VOLTAGE
-40
-45
-50
-55
-60
-65
-70
-75
-80
-85
-90
-95
-50
-55
-60
-65
-70
-75
-80
-85
-90
-95
-100
VO = 2VPP
RL = 150W
RL = 150W
f = 5MHz
2nd-Harmonic
2nd-Harmonic
3rd-Harmonic
3rd-Harmonic
dBc = dB Below Carrier
dBc = dB Below Carrier
1
10
100
0.5
1.5
2.5
3.5
4.5
5.5
6.5
7.5
Frequency (MHz)
Output Voltage Swing (VPP
)
Figure 9.
Figure 10.
DISABLE AND SHUTDOWN FEEDTHROUGH vs
FREQUENCY
OUTPUT VOLTAGE AND CURRENT LIMITATIONS
6
5
-20
-30
1W Internal
Power Limit
Input-referred
4
-40
100W Load
3
-50
2
Shutdown Feedthrough
1
-60
50W Load
25W Load
0
-70
-1
-2
-3
-4
-5
-80
Disable Feedthrough
1W Internal
Power Limit
-90
-100
-110
-200
-100
0
100
200
300
1M
10M
100M
Frequency (Hz)
1G
Output Current (mA)
Figure 11.
Figure 12.
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TYPICAL CHARACTERISTICS (continued)
At TA = +25°C, G = +2 V/V, RF = 523 Ω, and RL = 150 Ω, unless otherwise noted.
CHANNEL-TO-CHANNEL SWITCHING
CHANNEL-TO-CHANNEL SWITCHING GLITCH
75
50
0.75
0.50
0.25
0
At Matched Load
25
Output Voltage
0
-25
-50
-75
-0.25
-0.50
-0.75
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0
VIN_Ch0 = 200MHz, 0.7VPP
VIN_Ch1 = 0VDC
A0
A0
Time (10ns/div)
Time (10ns/div)
Figure 13.
Figure 14.
DISABLE/ENABLE TIME
DISABLE/ENABLE SWITCHING GLITCH
75
50
0.75
0.50
0.25
0
At Matched Load
25
Output
0
-25
-50
-75
-0.25
-0.50
-0.75
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0
VIN_Ch0 = 200MHz, 0.10VPP
EN
EN
Time (10ns/div)
Time (10ns/div)
Figure 15.
Figure 16.
SHUTDOWN/START-UP TIME
SHUTDOWN GLITCH
75
50
0.75
0.50
0.25
0
At Matched Load
25
Output
0
-25
-50
-75
-0.25
-0.50
-0.75
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0
VIN_Ch0 = 200MHz, 0.7VPP
SD
SD
Time (20ns/div)
Time (20ns/div)
Figure 17.
Figure 18.
8
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TYPICAL CHARACTERISTICS (continued)
At TA = +25°C, G = +2 V/V, RF = 523 Ω, and RL = 150 Ω, unless otherwise noted.
OPEN-LOOP TRANSIMPEDANCE GAIN AND PHASE vs
ALL HOSTILE CROSSTALK vs FREQUENCY
FREQUENCY
0
-10
-20
-30
-40
-50
-60
-70
-80
-90
-100
1M
100k
10k
1k
0
Input-referred
< ZOL
-45
-90
-135
-180
-225
½ZOL
½
100
10
10k
100k
1M
10M
100M
1G
1M
10M
100M
Frequency (Hz)
1G
Frequency (Hz)
Figure 19.
Figure 20.
CLOSED-LOOP OUTPUT IMPEDANCE vs FREQUENCY
1M
INPUT IMPEDANCE vs FREQUENCY
100M
10M
1M
Disabled
100k
Disabled (or Shutdown)
10k
1k
Enabled
100k
10k
1k
100
10
1
Enabled
10M
0.1
100
100k
1M
100M
1G
10k
100k
1M
10M
100M
1G
Frequency (Hz)
Frequency (Hz)
Figure 21.
Figure 22.
PSRR vs FREQUENCY
OUTPUT AND SUPPLY CURRENT vs TEMPERATURE
15.00
78.00
76.75
75.50
74.25
73.00
71.75
70.50
69.25
68.00
60
50
40
30
20
10
0
13.75
Input-referred
12.50
-PSRR
Supply Current (IQ)
11.25
10.00
8.75
+PSRR
+IOUT
7.50
6.25
-IOUT
5.00
-50
-25
0
25
50
75
100
125
1k
10k
100k
1M
10M
100M
1G
Ambient Temperature (°C)
Frequency (Hz)
Figure 23.
Figure 24.
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TYPICAL CHARACTERISTICS (continued)
At TA = +25°C, G = +2 V/V, RF = 523 Ω, and RL = 150 Ω, unless otherwise noted.
TYPICAL DC DRIFT OVER TEMPERATURE
INPUT VOLTAGE AND CURRENT NOISE
2.0
1.5
9
8
7
6
5
4
3
2
1
0
-1
300
100
1.0
VOS
Inverting Input Current Noise (19pA/ÖHz)
0.5
0
Ibn
-0.5
-1.0
-1.5
-2.0
-2.5
-3.0
10
Input Voltage Noise (4.5nV/ÖHz)
Ibi
Noninverting Input Current Noise (4pA/ÖHz)
1
-50
-25
0
25
50
75
100
125
10
100
1k
10k
100k
1M
10M
Ambient Temperature (°C)
Frequency (Hz)
Figure 26.
Figure 25.
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APPLICATION INFORMATION
input of a current feedback amplifier. Depending on
the logic applied to channel control pins A0 and A1,
one switch is on at all times. Figure 27 represents the
OPA4872 in this configuration. The truth table for
channel selection is shown in Table 1.
WIDEBAND MULTIPLEXER OPERATION
The OPA4872 gives a new level of performance in
wideband multiplexers. Figure 27 shows the
dc-coupled, gain of +2 V/V, dual power-supply circuit
used as the basis of the ±5-V Electrical
Characteristics and Typical Characteristic curves. For
test purposes, the input impedance is set to 75 Ω with
a resistor to ground and the output impedance is set
to 75 Ω with a series output resistor. Voltage swings
reported in the specifications are taken directly at the
input and output pins while load powers (in dBm) are
defined at a matched 75-Ω load. For the circuit of
Figure 27, the total effective load will be 150 Ω ||
1046 Ω = 131 Ω. Logic pins A0 and A1 control which
of the four inputs is selected while EN and SD allow
for power reduction. One optional component is
included in Figure 27. In addition to the usual
power-supply decoupling capacitors to ground, a
0.01-µF capacitor is included between the two
power-supply pins. In practical printed circuit board
(PCB) layouts, this optional added capacitor typically
improves the 2nd-harmonic distortion performance by
3 dB to 6 dB for bipolar supply operation.
Table 1. TRUTH TABLE
A0
0
A1
0
EN
0
SD
0
VOUT
IN0
1
0
0
0
IN1
IN2
0
1
0
0
1
1
0
0
IN3
X
X
X
X
1
0
High-Z, IQ = 3.4 mA
High-Z, IQ = 1.1 mA
X
1
The OPA4872 is in disable mode, with a quiescent
current of 3.4mA typical, when the EN pin is set to
0V. After being placed in disable mode, the OPA4872
is fully enabled in 6ns. For further power savings, the
SD pin can be used. Setting the SD pin to 5V places
the device in shutdown mode with a standing
quiescent current of 1.1 mA. Note that in this
shutdown mode, the OPA4872 requires 15ns to be
fully powered again. The truth table for disable and
shutdown modes can be found in Table 1.
Even though the internal architecture of the OPA4872
includes current steering, it is advantageous to look
at it as four switches looking into the noninverting
+5V
0.1mF
6.8mF
+
OPA4872
SD
EN
IN0
VIN0
75W
IN1
VIN1
75W
75W
VOUT
To 75W Load
IN2
523W
VIN2
75W
523W
IN3
VIN3
A0
A1
75W
Optional
0.1mF
6.8mF
0.01mF
+
-5V
Figure 27. DC-Coupled, G = +2V/V Bipolar Specification and Test Circuit (Channel 0 Selected)
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2-BIT HIGH-SPEED PGA
When channel 0 is selected, the overall gain to the
matched load of the OPA4872 is 0dB. When channel
1 is selected, this circuit delivers 6 dB of gain to the
matched load. When channel 2 is selected, this circuit
delivers 12 dB of gain to the matched load. When
channel 3 is selected, this circuit delivers 18 dB of
gain to the matched load.
The OPA4872 can be used as a 2-bit, high-speed
programmable gain amplifier (PGA) when used in
conjunction with another amplifier. Figure 28 shows
one OPA695 used in series with each OPA4876 input
and configured with gains of +1 V/V, +2 V/V, +4 V/V,
and +8 V/V, respectively.
+5V
50W
OPA695
G = 1V/V
-5V
+5V
523W
50W
IN0
+5V
OPA4872
SD
EN
50W
OPA695
G = 2V/V
-5V
IN1
IN2
511W
50W
511W
523W
+5V
To 50W Load
50W
523W
OPA695
G = 4V/V
Logic
-5V
453W
IN3
A0 A1
-5V
149W
+5V
50W
OPA695
G = 8V/V
-5V
402W
57.6W
Figure 28. 2-Bit, High-Speed PGA, Greater Than 300MHz Channel Bandwidth
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2-BIT, HIGH-SPEED ATTENUATOR
8-TO-1 VIDEO MULTIPLEXER
In contrast to the PGA,
a
two-bit high-speed
Two OPA4872s can be used together to form an
8-input video multiplexer. The multiplexer is shown in
Figure 31.
attenuator can be implemented by using an R-2R
ladder together with the OPA4872. Figure 29 shows
such an implementation.
Channel 0 sees the full input signal amplitude, where
as channel 1 sees 1/2 VIN, channel 2 see 1/4 VIN and
channel 3 sees 1/8 VIN.
OPA4872
EN
SD
VIN
OPA4872
SD
2R
IN0
EN
RO = 69W
R
2R
523W
IN1
To 75W Load
50W
R
523W
VOUT
2R
IN2
523W
To 50W Load
Logic
R
523W
R
IN3
Logic
OPA4872
EN
A0 A1
SD
Figure 29. 2-Bit, High-Speed Attenuator,
500MHz Channel Bandwidth
4-INPUT RGB ROUTER
RO = 69W
523W
Three OPA4872s can be used together to form a
four-input RGB router. The router for the red
component is shown in Figure 30. Identical stages
would be used for the green and blue channels.
523W
Logic
+5V
OPA4872
EN
Figure 31. 8-to-1 Video Multiplexer
SD
R1
When connecting OPA4872 outputs together,
maintain a gain of +1V/V at the load. The OPA4872
configuration shown is a gain of +6 dB; thus, the
matching resistance must be selected to achieve
–6 dB.
R2
75W
Red Out
523W
The set of equations to solve is shown in Equation 1
and Equation 2. Here, the impedance of interest is
75 Ω.
R3
To 75W Load
523W
R4
RO = ZO || (RO + RF + RG)
RF
Logic
1 +
= 2
A0 A1
RG
(1)
(2)
-5V
RF + RG = 1046W
Figure 30. 4-Input RGB Router
(Red Channel Shown)
RF = RG
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Solving for RO, with n devices connected together,
results in Equation 3:
OPERATING SUGGESTIONS
SETTING RESISTOR VALUES TO OPTIMIZE
BANDWIDTH
75 ´ (n - 1) + 804
241200
[75 ´ (n - 1) + 804]2
RO
=
´
1 +
- 1
2
The output stage of the OPA4872 is
a
current-feedback op amp, meaning it can hold an
almost constant bandwidth over signal gain settings
with the proper adjustment of the external resistor
values. This performance is shown in the Typical
Characteristic curves; the small-signal bandwidth
decreases only slightly with increasing gain. These
curves also show that the feedback resistor has been
changed for each gain setting. The resistor values on
the feedback path can be treated as frequency
response compensation elements while the ratio sets
the signal gain of the feedback resistor divided by the
gain resistor. Figure 32 shows the small-signal
frequency response analysis circuit for a current
feedback amplifier.
(3)
Results for n varying from 2 to 6 are given in Table 2.
Table 2. Series Resistance vs
Number of Parallel Outputs
NUMBER OF OPA4872s
RO (Ω)
69
2
3
4
5
6
63.94
59.49
55.59
52.15
The two major limitations of this circuit are the device
requirements for each OPA4872 and the acceptable
return loss resulting from the mismatch between the
load and the matching resistor.
VI
a
VO
DESIGN-IN TOOLS
RI
Z(S) iERR
DEMONSTRATION FIXTURE
iERR
RF
A printed circuit board (PCB) is available to assist in
the initial evaluation of circuit performance using the
OPA4872. The fixture is offered free of charge as an
unpopulated PCB, delivered with a user's guide. The
summary information for this fixture is shown in
Table 3.
RG
Figure 32. Recommended Feedback Resistor vs
Noise Gain
Table 3. OPA4872 Demonstration Fixture
LITERATURE
PACKAGE
ORDERING NUMBER
NUMBER
The key elements of this current-feedback op amp
model are:
SO-14
DEM-OPA-SO-1E
SBOU045
α → Buffer gain from the noninverting input to the
inverting input
The demonstration fixture can be requested at the
Texas Instruments web site at (www.ti.com) through
the OPA4872 product folder.
RI → Buffer output impedance
iERR → Feedback error current signal
MACROMODELS AND APPLICATIONS SUPPORT
Z(s) Frequency-dependent
transimpedance gain from iERR to VO
→
open-loop
Computer simulation of circuit performance using
SPICE is often useful when analyzing the
performance of analog circuits and systems. This
practice is particularly true for video and RF amplifier
circuits, where parasitic capacitance and inductance
can have a major effect on circuit performance. A
SPICE model for the OPA4872 is available through
the Texas Instruments web site at www.ti.com. This
model does a good job of predicting small-signal ac
and transient performance under a wide variety of
operating conditions. It does not do as well in
predicting the harmonic distortion or dG/dP
characteristics.
The buffer gain is typically very close to 1.00 and is
normally neglected from signal gain considerations. It
will, however, set the CMRR for a single op amp
differential amplifier configuration. For a buffer gain
α < 1.0, the CMRR = –20 × log (1 – α) dB.
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RI, the buffer output impedance, is a critical portion of
the bandwidth control equation. RI for the OPA4872 is
typically about 30 Ω. A current-feedback op amp
senses an error current in the inverting node (as
opposed to a differential input error voltage for a
voltage-feedback op amp) and passes this on to the
output through an internal frequency dependent
transimpedance gain. The Typical Characteristics
show this open-loop transimpedance response. This
open-loop response is analogous to the open-loop
voltage gain curve for a voltage-feedback op amp.
Developing the transfer function for the circuit of
Figure 32 gives Equation 4:
The OPA4872 is internally compensated to give a
maximally flat frequency response for RF = 523 Ω at
NG = 2 on ±5-V supplies. Evaluating the denominator
of Equation 5 (which is the feedback transimpedance)
gives an optimal target of 663 Ω. As the signal gain
changes, the contribution of the NG × RI term in the
feedback transimpedance will change, but the total
can be held constant by adjusting RF. Equation 6
gives an approximate equation for optimum RF over
signal gain:
RF = 663W - NG x RI
(6)
As the desired signal gain increases, this equation
will eventually predict a negative RF. A somewhat
subjective limit to this adjustment also can be set by
holding RG to a minimum value of 20 Ω. Lower values
load both the buffer stage at the input and the output
stage, if RF gets too low, actually decreasing the
bandwidth. Figure 33 shows the recommended RF
versus NG for ±5-V operation. The values for RF
versus gain shown here are approximately equal to
the values used to generate the Typical
Characteristics. They differ in that the optimized
values used in the Typical Characteristics are also
correcting for board parasitics not considered in the
simplified analysis leading to Equation 5. The values
shown in Figure 33 give a good starting point for
design where bandwidth optimization is desired.
RF
a 1+
VO
VI
RG
RF
aNG
RF + RI NG
Z(S)
=
=
RF + RI
1+
1+
RG
1+
Z(S)
where:
RF
= 1+
NG
RG
(4)
This formula is written in a loop-gain analysis format,
where the errors arising from a noninfinite open-loop
gain are shown in the denominator. If Z(S) were
infinite over all frequencies, the denominator of
Equation 4 would reduce to 1 and the ideal desired
signal gain shown in the numerator would be
achieved. The fraction in the denominator of
Equation 4 determines the frequency response.
Equation 5 shows this as the loop-gain equation:
600
550
500
450
400
350
300
250
200
150
100
Z(S)
= Loop Gain
RF + RI NG
(5)
If 20 × log(RF + NG × RI) were drawn on top of the
open-loop transimpedance plot, the difference
between the two calculations would be the loop gain
at a given frequency. Eventually, Z(S) rolls off to equal
the denominator of Equation 5, at which point the
loop gain reduces to 1 (and the curves intersect).
This point of equality is where the amplifier
closed-loop frequency response given by Equation 4
starts to roll off, and is exactly analogous to the
frequency at which the noise gain equals the
open-loop voltage gain for a voltage-feedback op
amp. The difference here is that the total impedance
in the denominator of Equation 5 may be controlled
somewhat separately from the desired signal gain (or
NG).
0
5
10
15
20
Noise Gain
Figure 33. Feedback Resistor vs Noise Gain
The total impedance going into the inverting input
may be used to adjust the closed-loop signal
bandwidth. Inserting a series resistor between the
inverting input and the summing junction increases
the feedback impedance (denominator of Equation 4),
decreasing the bandwidth.
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DRIVING CAPACITIVE LOADS
VOSO_envelope = VOS ´ G ± Ibi x RF ± (RS ´ Ib) ´ G
PSRR+
± ½5 - (VS+)½ ´ 10-
One of the most demanding, yet very common load
conditions, is capacitive loading. Often, the capacitive
load is the input of an analog-to-digital converter
(ADC)—including additional external capacitance that
may be recommended to improve ADC linearity. A
high-speed device such as the OPA4872 can be very
susceptible to decreased stability and closed-loop
response peaking when a capacitive load is placed
directly on the output pin. When the device open-loop
output resistance is considered, this capacitive load
introduces an additional pole in the signal path that
can decrease the phase margin. Several external
solutions to this problem have been suggested. When
the primary considerations are frequency response
flatness, pulse response fidelity, and/or distortion, the
simplest and most effective solution is to isolate the
capacitive load from the feedback loop by inserting a
series isolation resistor between the amplifier output
and the capacitive load. This isolation resistor does
not eliminate the pole from the loop response, but
rather shifts it and adds a zero at a higher frequency.
The additional zero acts to cancel the phase lag from
the capacitive load pole, thus increasing the phase
margin and improving stability.
20
PSRR-
±½-5 - (VS-)½ ´ 10-
20
(7)
Where:
RS: Input resistance seen by R0, R1, G0, G1, B0,
or B1.
Ib: Noninverting input bias current
Ibi: Inverting input bias current
G: Gain
VS+: Positive supply voltage
VS–: Negative supply voltage
PSRR+: Positive supply PSRR
PSRR–: Negative supply PSRR
VOS: Input Offset Voltage
Evaluating the front-page schematic, using
a
worst-case, +25°C offset voltage, bias current and
PSRR specifications and operating at ±6 V, gives a
worst-case output equal to Equation 8:
±10mV + 75W ´ ±14mA ´ 2
50
±½5 - 6½ ´ 10-
20
+523W ´ ±18mA
±½-5 - (-6)½ ´ 10-
51
20
The Typical Characteristics show the recommended
RS versus capacitive load and the resulting frequency
response at the load; see Figure 5. Parasitic
capacitive loads greater than 2 pF can begin to
degrade the performance of the OPA4872. Long PCB
traces, unmatched cables, and connections to
multiple devices can easily cause this value to be
exceeded. Always consider this effect carefully, and
add the recommended series resistor as close as
possible to the OPA4872 output pin (see the Board
Layout Guidelines section).
= 29.2mV
(8)
DISTORTION PERFORMANCE
The OPA4872 provides good distortion performance
into a 150-Ω load on ±5-V supplies. Relative to
alternative solutions, it provides exceptional
performance into lighter loads. Generally, until the
fundamental signal reaches very high frequency or
power levels, the 2nd harmonic dominates the
distortion with a negligible 3rd harmonic component.
Focusing then on the 2nd harmonic, increasing the
load impedance directly improves distortion. Also,
providing an additional supply decoupling capacitor
(0.01 µF) between the supply pins (for bipolar
operation) improves the 2nd-order distortion slightly
(3 dB to 6 dB).
DC ACCURACY
The OPA4872 offers excellent dc signal accuracy.
Parameters that influence the output dc offset voltage
are:
•
•
•
•
•
Output offset voltage
Input bias current
Gain error
Power-supply rejection ratio
Temperature
In most op amps, increasing the output voltage swing
increases harmonic distortion directly. The Typical
Characteristics show the 2nd harmonic increasing at
a little less than the expected 2X rate while the 3rd
harmonic increases at a little less than the expected
3X rate. Where the test power doubles, the 2nd
harmonic increases only by less than the expected 6
dB, whereas the 3rd harmonic increases by less than
the expected 12 dB.
Leaving both temperature and gain error parameters
aside, the output offset voltage envelope can be
described as shown in Equation 7:
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NOISE PERFORMANCE
EO
=
ENI2 + (IBNRS)2 + 4kTRS NG2 + (IBIRF)2 + 4kTRFNG
The OPA4872 offers an excellent balance between
voltage and current noise terms to achieve low output
noise. The inverting current noise (19 pA/√Hz) is
significantly lower than earlier solutions, while the
input voltage noise (4.5 nV/√Hz) is lower than most
unity-gain stable, wideband, voltage-feedback op
amps. As long as the ac source impedance looking
out of the noninverting node is less than 100 Ω, this
current noise will not contribute significantly to the
total output noise. The op amp input voltage noise
and the two input current noise terms combine to give
low output noise under a wide variety of operating
conditions. Figure 34 shows the OPA4872 noise
analysis model with all the noise terms included. In
this model, all noise terms are taken to be noise
voltage or current density terms in either nV/√Hz or
pA/√Hz.
(9)
Dividing this expression by the noise gain (NG = (1 +
RF/RG)) gives the equivalent input-referred spot noise
voltage at the noninverting input, as shown in
Equation 10.
2
EO = ENI2 + (IBNRS)2 + 4kTRS +
IBIRF 4kTRF
+
NG
NG
(10)
Evaluating these two equations for the OPA4872
circuit and component values (see Figure 27) gives a
total output spot noise voltage of 14.2 nV/√Hz and a
total equivalent input spot noise voltage of 7.1
nV/√Hz. This total input-referred spot noise voltage is
higher than the 4.5-nV/√Hz specification for the
OPA4872 voltage noise alone. This voltage reflects
the noise added to the output by the inverting current
noise times the feedback resistor. If the feedback
resistor is reduced in high-gain configurations, the
total input-referred voltage noise given by
Equation 10 approaches only the 4.5 nV/√Hz of the
op amp itself. For example, going to a gain of +10
using RF = 178 Ω gives a total input-referred noise of
4.7 nV/√Hz.
ENI
EO
OPA4872
RS
IBN
ERS
RF
Ö4kTRS
Ö4kTRF
IBI
RG
4kT
RG
20
4kT = 1.6 x 10
at 290K
J
Figure 34. Op Amp Noise Analysis Model
The total output spot noise voltage can be computed
as the square root of the sum of all squared output
noise voltage contributors. Equation 9 shows the
general form for the output noise voltage using the
terms shown in Figure 35.
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THERMAL ANALYSIS
load power. Quiescent power is simply the specified
no-load supply current times the total supply voltage
across the part. PDL depends on the required output
signal and load; for a grounded resistive load, PDL is
at a maximum when the output is fixed at a voltage
equal to 1/2 of either supply voltage (for equal bipolar
Heatsinking or forced airflow may be required under
extreme operating conditions. Maximum desired
junction temperature sets the maximum allowed
internal power dissipation as discussed in this
document. In no case should the maximum junction
temperature be allowed to exceed +150°C.
2
supplies). Under this condition PDL = VS /(4 × RL),
where RL includes feedback network loading.
Operating junction temperature (TJ) is given by TA
+
Note that it is the power in the output stage and not in
the load that determines internal power dissipation.
PD × θJA. The total internal power dissipation (PD) is
the sum of quiescent power (PDQ) and additional
power dissipated in the output stage (PDL) to deliver
-20
en
4kT = 1.6 x 10
J
OPA4872
at 290K
RS
ini
VRS = Ö4kTRS
eo
RF
VRF = Ö4kTRF
4kT
iin
iRG
RG
Ö
RG
Figure 35. OPA4872 Noise Analysis Model
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As a worst-case example, compute the maximum TJ
using an OPA4872ID in the circuit of Figure 27
operating at the maximum specified ambient
temperature of +85°C with its output driving a
grounded 100-Ω load to +2.5 V:
Again, keep their leads and PCB trace length as short
as possible. Never use wirewound type resistors in a
high-frequency
application.
Other
network
components, such as noninverting input termination
resistors, should also be placed close to the package.
PD = 10V ´ 11.7mA + (52/[4 ´ (150W || 1046W)]) = 165mW
d) Connections to other wideband devices on the
board may be made with short direct traces or
through onboard transmission lines. For short
connections, consider the trace and the input to the
next device as a lumped capacitive load. Relatively
wide traces (50mils to 100mils) should be used,
preferably with ground and power planes opened up
around them.
Maximum TJ = +85°C + (165mW ´ 80°C/W) = 98°C
This worst-case condition does not exceed the
maximum junction temperature. Normally, this
extreme case is not encountered.
BOARD LAYOUT GUIDELINES
Estimate the total capacitive load and set RS from the
plot of Figure 5. Low parasitic capacitive loads
(greater than 5 pF) may not need an RS because the
OPA4872 is nominally compensated to operate with a
2-pF parasitic load. If a long trace is required, and the
6dB signal loss intrinsic to a doubly-terminated
transmission line is acceptable, implement a matched
impedance transmission line using microstrip or
stripline techniques (consult an ECL design handbook
for microstrip and stripline layout techniques). A 50-Ω
environment is normally not necessary on the board,
Achieving
optimum
performance
with
a
high-frequency amplifier such as the OPA4872
requires careful attention to board layout parasitics
and external component types. Recommendations to
optimize performance include:
a) Minimize parasitic capacitance to any ac
ground for all of the signal I/O pins. Parasitic
capacitance on the output pin can cause instability;
on the noninverting input, it can react with the source
impedance to cause unintentional bandlimiting. To
reduce unwanted capacitance, a window around the
signal I/O pins should be opened in all of the ground
and power planes around those pins. Otherwise,
ground and power planes should be unbroken
elsewhere on the board.
and in fact,
a higher impedance environment
improves distortion as shown in the Distortion versus
Load plot; see Figure 7. With a characteristic board
trace impedance defined based on board material
and trace dimensions, a matching series resistor into
the trace from the output of the OPA4872 is used as
well as a terminating shunt resistor at the input of the
destination device. Remember also that the
terminating impedance is the parallel combination of
the shunt resistor and the input impedance of the
destination device; this total effective impedance
should be set to match the trace impedance. The
high output voltage and current capability of the
OPA4872 allow multiple destination devices to be
handled as separate transmission lines, each with its
own series and shunt terminations. If the 6-dB
attenuation of a doubly-terminated transmission line
b) Minimize the distance (< 0.25") from the
power-supply pins to high frequency 0.1-µF
decoupling capacitors. At the device pins, the
ground and power plane layout should not be in close
proximity to the signal I/O pins. Avoid narrow power
and ground traces to minimize inductance between
the pins and the decoupling capacitors. The
power-supply connections (on pins 9, 11, 13, and 15)
should always be decoupled with these capacitors.
An optional supply decoupling capacitor across the
two power supplies (for bipolar operation) improves
2nd harmonic distortion performance. Larger (2.2 µF
to 6.8 µF) decoupling capacitors, effective at lower
frequency, should also be used on the main supply
pins. These capacitors may be placed somewhat
farther from the device and may be shared among
several devices in the same area of the PCB.
is
unacceptable,
a
long
trace
can
be
series-terminated at the source end only. Treat the
trace as a capacitive load in this case and set the
series resistor value as shown in Figure 5. This
configuration does not preserve signal integrity as
well as
a doubly-terminated line. If the input
impedance of the destination device is low, there will
be some signal attenuation because of the voltage
divider formed by the series output into the
terminating impedance.
c) Careful selection and placement of external
components
preserves
the
high-frequency
performance of the OPA4872. Resistors should be
a very low reactance type. Surface-mount resistors
work best and allow a tighter overall layout. Metal-film
and carbon composition, axially-leaded resistors can
also provide good high-frequency performance.
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e) Socketing a high-speed part like the OPA4872
+VCC
is not recommended. The additional lead length and
pin-to-pin capacitance introduced by the socket can
create an extremely troublesome parasitic network
that can make it almost impossible to achieve a
smooth, stable frequency response. Best results are
obtained by soldering the OPA4872 onto the board.
External
Pin
Internal
Circuitry
-VCC
INPUT AND ESD PROTECTION
Figure 36. Internal ESD Protection
The OPA4872 is built using a very high-speed
complementary bipolar process. The internal junction
breakdown voltages are relatively low for these very
small geometry devices. These breakdowns are
reflected in the Absolute Maximum Ratings table. All
device pins have limited ESD protection using internal
diodes to the power supplies as shown in Figure 36.
These diodes provide moderate protection to input
overdrive voltages above the supplies as well. The
protection diodes can typically support 30-mA
continuous current. Where higher currents are
possible (for example, in systems with ±15-V supply
parts driving into the OPA4872), current-limiting
series resistors should be added into the two inputs.
Keep these resistor values as low as possible
because high values degrade both noise performance
and frequency response.
20
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Copyright © 2008, Texas Instruments Incorporated
Product Folder Link(s): OPA4872-EP
PACKAGE MATERIALS INFORMATION
www.ti.com
16-Dec-2008
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0 (mm)
B0 (mm)
K0 (mm)
P1
W
Pin1
Diameter Width
(mm) W1 (mm)
(mm) (mm) Quadrant
OPA4872MDREP
SOIC
D
14
2500
330.0
16.4
6.5
9.0
2.1
8.0
16.0
Q1
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
16-Dec-2008
*All dimensions are nominal
Device
Package Type Package Drawing Pins
SOIC 14
SPQ
Length (mm) Width (mm) Height (mm)
346.0 346.0 33.0
OPA4872MDREP
D
2500
Pack Materials-Page 2
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