TP2350 [TRIPATH]
Interface Circuit;
| 型号： | TP2350 |
| 厂家： | 
TRIPATH TECHNOLOGY INC. |
| 描述： | Interface Circuit |
| 文件： | 总36页 (文件大小：700K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
Tripath Technology, Inc. - Technical Information
TK2350
STEREO 300W (4Ω) CLASS-T DIGITAL AUDIO AMPLIFIER DRIVER
USING DIGITAL POWER PROCESSINGTM TECHNOLOGY
T e c h n i c a l I n f o r m a t i o n
R e vi s i o n 1 . 4 – Ap r i l 2 0 0 3
GENERAL DESCRIPTION
The TK2350 (TC2001/TP2350B chipset) is a two-channel, 300W (4Ω) per channel Amplifier Driver that uses
Tripath’s proprietary Digital Power Processing (DPPTM) technology. Class-T amplifiers offer both the audio
fidelity of Class-AB and the power efficiency of Class-D amplifiers.
Applications
Features
ꢀ
ꢀ
ꢀ
ꢀ
Audio/Video Amplifiers & Receivers
Pro-audio Amplifiers
ꢀ
ꢀ
ꢀ
ꢀ
Class-T architecture
Pin compatible with Tripath TK2150
Proprietary Digital Power Processing technology
“Audiophile” Sound Quality
Automobile Power Amplifiers
Subwoofer Amplifiers
ꢀ
ꢀ
0.02% THD+N @ 50W, 8Ω
0.03% IHF-IM @ 30W, 8Ω
Benefits
ꢀ
ꢀ
High Efficiency
ꢀ
ꢀ
ꢀ
Reduced system cost with smaller/less
ꢀ
ꢀ
95% @ 150W @ 8Ω
90% @ 275W @ 4Ω
expensive power supply and heat sink
Signal fidelity equal to high quality Class-AB
amplifiers
Supports wide range of output power levels
ꢀ
Up to 300W/channel (4Ω), single-ended
outputs
High dynamic range compatible with digital
media such as CD and DVD
ꢀ
Up to 1000W (4Ω), bridged outputs
ꢀ
ꢀ
ꢀ
Output over-current protection
Over- and under-voltage protection
Over-temperature protection
Typical Performance for TK2350
THD+N versus Output Power versus Supply Voltage
RL = 4Ω
10
5
f = 1kHz
BBM = 80nS
BW = 22Hz - 22kHz
2
1
39V
45V
54V
0.5
0.2
0.1
0.05
0.02
0.01
1
2
5
10
20
50
100
200
500
Output Power (W)
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Tripath Technology, Inc. - Technical Information
Absolute Maximum Ratings TC2001 (Note 1)
SYMBOL
V5
Vlogic
TA
TSTORE
TJMAX
ESDHB
PARAMETER
Value
6
V5+0.3V
-40° to +85°
-55° to 150°
150°
UNITS
V
V
°C
°C
5V Power Supply
Input Logic Level
Operating Free-air Temperature Range
Storage Temperature Range
Maximum Junction Temperature
ESD Susceptibility – Human Body Model (Note 2)
All pins
°C
2000
V
Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur.
See the table below for Operating Conditions.
Note 2: Human body model, 100pF discharged through a 1.5KΩ resistor.
Absolute Maximum Ratings TP2350B (Note 3)
SYMBOL
VPP, VNN Supply Voltage
VN10
TSTORE
TA
PARAMETER
Value
+/- 70
VNN+13
-55º to 150º
-40º to 85º
150º
UNITS
V
V
C
C
C
Voltage for FET drive
Storage Temperature Range
Operating Free-air Temperature Range
Junction Temperature
TJ
ESDHB
ESD Susceptibility – Human Body Model (Note 4)
All pins
2000
200
V
V
ESDMM
ESD Susceptibility – Machine Model (Note 5)
All pins
Note 3: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur.
See the table below for Operating Conditions.
Note 4: Human body model, 100pF discharged through a 1.5KΩ resistor.
Note 5: Machine model, 220pF – 240pF discharged through all pins.
Operating Conditions TC2001 (Note 6)
SYMBOL
V5
VHI
VLO
TA
PARAMETER
MIN.
4.5
V5-1.0
TYP.
5
MAX. UNITS
Supply Voltage
Logic Input High
Logic Input Low
5.5
V
V
V
C
1
Operating Temperature Range
-40°
25°
85°
Note 6: Recommended Operating Conditions indicate conditions for which the device is functional.
See Electrical Characteristics for guaranteed specific performance limits.
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Tripath Technology, Inc. - Technical Information
Operating Conditions TP2350B (Note 7)
SYMBOL
VPP, VNN Supply Voltage
VN10
PARAMETER
MIN.
+/- 15 +/-45 +/- 65
10 12
TYP.
MAX. UNITS
V
V
Voltage for FET drive (Volts above VNN)
9
Note 7: Recommended Operating Conditions indicate conditions for which the device is functional.
See Electrical Characteristics for guaranteed specific performance limits.
Operating Characteristics TC2001 (Note 8)
SYMBOL
I5
PARAMETER
MIN.
TYP.
50
MAX. UNITS
Supply Current
mA
fsw
VIN
VOUTHI
VOUTLO
RIN
Switching Frequency
Input Sensitivity
High Output Voltage
Low Output Voltage
Input Impedance
Input DC Bias
650
kHz
V
V
0
1.5
V5-0.5
100
mV
2
2.4
k
Ω
V
Note 8: Recommended Operating Conditions indicate conditions for which the device is functional.
See Electrical Characteristics for guaranteed specific performance limits.
Thermal Characteristics TC2001
SYMBOL
PARAMETER
Value
UNITS
Junction-to-ambient Thermal Resistance (still air)
C/W
θ
JA
80°
Thermal Characteristics TP2350B
SYMBOL
PARAMETER
Value
UNITS
Junction-to-case Thermal Resistance (Note 9)
C/W
θ
JC
3°
Note 9: Recommended Operating Conditions indicate conditions for which the device is functional.
See Electrical Characteristics for guaranteed specific performance limits.
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Tripath Technology, Inc. - Technical Information
Electrical Characteristics TC2001 (Note 10)
TA = 25 °C. See Application/Test Circuit on page 7. Unless otherwise noted, the supply voltage is
VPP=|VNN|=45V.
SYMBOL
Iq
PARAMETER
CONDITIONS
MIN.
TYP.
45
MAX. UNITS
Quiescent Current
V5 = 5V
V5 = 5V
60
mA
(Mute = 0V)
IMUTE
Mute Supply Current
(Mute = 5V)
20
25
mA
VIH
VIL
VOH
VOL
VTOC
High-level input voltage (MUTE)
3.5
4.0
V
V
V
V
V
Low-level input voltage (MUTE)
High-level output voltage (HMUTE) IOH = 3mA
Low-level output voltage (HMUTE) IOL = 3mA
1.0
0.5
1.09
Over Current Sense Voltage
TBD
0.85
0.97
Threshold
IVPPSENSE VPPSENSE Threshold Currents
Over-voltage turn on (muted)
Over-voltage turn off (mute off)
Under-voltage turn off (mute off)
Under-voltage turn on (muted)
Over-voltage turn on (muted)
Over-voltage turn off (mute off)
Under-voltage turn off (mute off)
Under-voltage turn on (muted)
Over-voltage turn on (muted)
Over-voltage turn off (mute off)
Under-voltage turn off (mute off)
Under-voltage turn on (muted)
162
154
79
178
87
µA
µA
µA
µA
V
138
62
72
VVPPSENSE Threshold Voltages with
RVPPSENSE = 422KΩ
68.4
65.0
33.3
30.4
174
169
86
75.8
37.1
191
95
57.6
25.9
152
65
V
V
(Note 11, Note 12)
V
IVNNSENSE VNNSENSE Threshold Currents
µA
µA
µA
µA
V
77
VVNNSENSE Threshold Voltages with
RVNNSENSE = 392KΩ
Over-voltage turn on (muted)
Over-voltage turn off (mute off)
Under-voltage turn off (mute off)
Under-voltage turn on (muted)
-68.2
-66.2
-33.7
-30.2
-75.6
-37.6
-59.0
-25.2
V
V
(Note 11, Note 12)
V
Note 10: Minimum and maximum limits are guaranteed but may not be 100% tested.
Note 11: These supply voltages are calculated using the IVPPSENSE and IVNNSENSE values shown in the Electrical
Characteristics table. The typical voltage values shown are calculated using a RVPPSENSE and RVNNSENSE value
of 422kohm without any tolerance variation. The minimum and maximum voltage limits shown include either a +1% or
–1% (+1% for Over-voltage turn on and Under-voltage turn off, -1% for Over-voltage turn off and Under-voltage turn on)
variation of RVPPSENSE or RVNNSENSE off the nominal 422kohm and 392kohm values. These voltage
specifications are examples to show both typical and worst case voltage ranges for a given RVPPSENSE and
RVNNSENSE resistor values of 422kohm and 392kohm. Please refer to the Application Information section for a more
detailed description of how to calculate the over and under voltage trip voltages for a given resistor value.
Note 12: The fact that the over-voltage turn on specifications exceed the absolute maximum of +/-70V for the TK2350 does not
imply that the part will work at these elevated supply voltages. It also does not imply that the TK2350 is tested or
guaranteed at these supply voltages. The supply voltages are simply a calculation based on the process spread of the
IVPPSENSE and IVNNSENSE currents (see note 7). The supply voltage must be maintained below the absolute
maximum of +/-70V or permanent damage to the TK2350 may occur.
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Electrical Characteristics TK2350 (Note 14)
TA = 25 °C. See Application/Test Circuit on page 7. Unless otherwise noted, the supply voltage is
VPP=|VNN|=45V.
SYMBOL
Iq
PARAMETER
CONDITIONS
VPP = +45V
MIN.
TYP.
MAX. UNITS
Quiescent Current
90
mA
mA
mA
(No load, BBM0=1,BBM1=0,
Mute = 0V)
VNN = -45V (using external VN10)
VNN = -45V (using SMPSO pin to
drive IRF9510 for generating VN10)
VN10 = 10V
90
TBD
200
1
TBD
mA
mA
mA
mA
IMUTE
Mute Supply Current
(No load, Mute = 5V)
VPP = +45V
VNN = -45V
1
VN10 = 10V
1
Note 14: Minimum and maximum limits are guaranteed but may not be 100% tested.
Performance Characteristics TK2350 – Single Ended
TA = 25 °C. Unless otherwise noted, the supply voltage is VPP=|VNN|=45V, the input frequency is 1kHz and the
measurement bandwidth is 20kHz. See Application/Test Circuit.
SYMBOL
POUT
PARAMETER
Output Power
(continuous Power/Channel)
CONDITIONS
THD+N = 0.1%, RL = 8Ω
MIN.
TYP.
MAX. UNITS
100
190
120
220
0.02
0.03
102
W
W
W
W
RL = 4Ω
THD+N = 1%,
RL = 8Ω
RL = 4Ω
THD + N Total Harmonic Distortion Plus
Noise
IHF-IM
SNR
CS
%
POUT = 50W/Channel, RL = 8Ω
IHF Intermodulation Distortion
%
19kHz, 20kHz, 1:1 (IHF), RL = 8Ω
POUT = 30W/Channel
Signal-to-Noise Ratio
dB
A Weighted, RL = 4Ω,
POUT = 275W/Channel
Channel Separation
Power Efficiency
Amplifier Gain
97
95
10.7
dB
%
V/V
0dBr = 30W, RL = 8Ω, f = 1kHz
POUT = 150W/Channel, RL = 8Ω
η
AV
P
OUT = 10W/Channel, RL = 4Ω
See Application / Test Circuit
AVERROR
eNOUT
Channel to Channel Gain Error
Output Noise Voltage
0.5
1.0
dB
P
OUT = 10W/Channel, RL = 4Ω
See Application / Test Circuit
A Weighted, no signal, input shorted,
DC offset nulled to zero
260
µV
VOFFSET
Output Offset Voltage
No Load, Mute = Logic Low
0.1% RFBA, RFBB, RFBC resistors
-1.0
V
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Tripath Technology, Inc. - Technical Information
TK2350 Block Diagram
LC
Output Left
Input Left
TC2001
Audio
Filter
TP2350B
MOSFET
Driver
Output
MOSFETs
Signal
LC
Filter
Processor
Input Right
Output Right
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Tripath Technology, Inc. - Technical Information
TC2001 Pinout
28-pin SOIC
(Top View)
28
27
26
1
2
3
4
5
6
7
8
BIASCAP
FBKGND2
DCMP
FBKOUT2
VPWR
FBKGND1
FBKOUT1
HMUTE
Y1
INV2
OAOUT2
BBM0
BBM1
MUTE
INV1
OAOUT1
V5
AGND
VPPSENSE
OVRLDB
VNNSENSE
OCD1
REF
25
24
23
22
21
20
19
18
9
10
11
12
13
14
Y1B
Y2B
Y2
NC
17
16
15
OCD2
TP2350B Pinout
64-pin LQFP
(Top View)
51 50 49 48 47
40 39 38
36 35 34 33
37
46 45 44 43 42 41
52
53
54
55
56
57
58
59
60
61
62
63
64
32
31
30
29
28
27
26
25
24
23
22
21
20
NC
OCS1LN
NC
OCS2LN
OCS2LP
NC
NC
VBOOT2
OCS1LP
NC
NC
VBOOT1
NC
SW -FB
SMPSO
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
1
2
3
4
5
6
7
8
9
15 16 17 18 19
10 11 12 13 14
Please note that the heatslug on the bottom of the package is connected to VNN.
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Tripath Technology, Inc. - Technical Information
TC2001 Audio Signal Processor Pin Descriptions
Pin
Function
Description
1
BIASCAP
Bandgap reference times two (typically 2.5VDC). Used to set the
common mode voltage for the input op amps. This pin is not capable of
driving external circuitry.
2, 6
3
FBKGND2,
FBKGND1
DCMP
Ground Kelvin feedback (Channels 1 & 2)
Internal mode selection. This pin must be grounded for proper device
operation.
4, 7
FBKOUT2,
FBKOUT1
VPWR
Switching feedback (Channels 1 & 2)
5
8
Test pin. Must be left floating.
HMUTE
Logic output. A logic high indicates both amplifiers are muted, due to the
mute pin state, or a “fault”.
9, 12
10, 11
13
Y1, Y2
Y1B, Y2B
NC
Non-inverted switching modulator outputs.
Inverted switching modulator outputs.
No connect
14, 16
15
OCD2, OCD1
REF
Over Current Detect pins.
Internal bandgap reference voltage; approximately 1.2 VDC.
Negative supply voltage sense input. This pin is used for both over and
under voltage sensing for the VNN supply.
A logic low output indicates the input signal has overloaded the amplifier.
Positive supply voltage sense input. This pin is used for both over and
under voltage sensing for the VPP supply.
Analog Ground.
17
VNNSENSE
18
19
OVRLDB
VPPSENSE
20
AGND
V5
21
5 Volt power supply input.
22, 27
23, 28
OAOUT1, OAOUT2 Input stage output pins.
INV1, INV2
Single-ended inputs. Inputs are a “virtual” ground of an inverting opamp
with approximately 2.4VDC bias.
24
MUTE
When set to logic high, both amplifiers are muted and in idle mode.
When low (grounded), both amplifiers are fully operational. If left floating,
the device stays in the mute mode. Ground if not used.
Break-before-make timing control to prevent shoot-through in the output
MOSFETs.
25, 26
BBM1, BBM0
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Tripath Technology, Inc. - Technical Information
TP2350B Pin Description
Pin
5
Function
Description
AGND
V5
Analog ground.
5V power supply input.
6
7
9
10
13,17
14,16
27,57
OCD1
Over-current threshold output (Channel 1)
This a test pin for the TP2350B. This pin should be left floating.
Over-current threshold output (Channel 2)
Non-inverted switching modulator inputs
Inverted switching modulator inputs
Bootstrapped voltage to supply drive to gate of high-side FET
TSS
OCD2
Y2, Y1
Y2B, Y1B
VBOOT2, VBOOT1
(Channel 2 & 1)
30,31
33,34
36,48
37,47
39,45
40,44
41,43
OCS2LP, OCS2LN
OCS2HP, OCS2HN
HO2, HO1
Over Current Sense inputs, Channel 2 low-side
Over Current Sense inputs, Channel 2 high-side
High side gate drive output (Channel 2 & 1)
HO2COM, HO1COM
LO2COM, LO1COM
LO2, LO1
Kelvin connection to source of high-side transistor (Channel 2 & 1)
Kelvin connection to source of low-side transistor (Channel 2 & 1)
Low side gate drive output (Channel 2 & 1)
VN10
“Floating” supply input for the FET drive circuitry. This voltage must be stable
and referenced to VNN.
42
50,51
53,54
59
VNN
OCS1HN, OCS1HP
OCS1LN ,OCS1LP
SW-FB
Negative supply voltage.
Over Current Sense inputs, Channel 1 high-side
Over Current Sense inputs, Channel 1 low-side
Feedback for regulating switching power supply output for VN10
Switching power supply output for VN10
Not connected (bonded) internally. Leave these pins floating.
60
SMPSO
NC
1,2,3,4,8,
11,12,15,
18,19,20,
21,22,23,
24,25,26,
28,29,32,
35,38,46,
49,52,55,
56,58,61,
62,63,64
Please note that the heatslug on the bottom of the package is connected to VNN.
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Application/Test Circuit
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External Components Description (Refer to the Application/Test Circuit)
Components Description
RI
RF
CI
Inverting input resistance to provide AC gain in conjunction with RF. This input is
biased at the BIASCAP voltage (approximately 2.5VDC).
Feedback resistor to set AC gain in conjunction with RI. Please refer to the Amplifier
Gain paragraph, in the Application Information section.
AC input coupling capacitor which, in conjunction with RI, forms a highpass filter at
.
fC = 1 (2πRICI)
RFBA
RFBB
Feedback divider resistor connected to V5. This resistor is normally set at 1kΩ.
Feedback divider resistor connected to AGND. This value of this resistor depends
on the supply voltage setting and helps set the TK2350 gain in conjunction with RI,
RF, RFBA, and RFBC. Please see the Modulator Feedback Design paragraphs in the
Application Information Section.
RFBC
Feedback resistor connected from either the OUT1(OUT2) to FBKOUT1(FBKOUT2)
or speaker ground to FBKGND1(FBKGND2). The value of this resistor depends on
the supply voltage setting and helps set the TK2350 gain in conjunction with RI, RF,
RFBA,, and RFBB. It should be noted that the resistor from OUT1(OUT2) to
2
FBKOUT1(FBKOUT2) must have a power rating of greater than
.
P
= VPP (2RFBC)
DISS
Please see the Modulator Feedback Design paragraphs in the Application
Information Section.
CFB
Feedback delay capacitor that both lowers the idle switching frequency and filters
very high frequency noise from the feedback signal, which improves amplifier
performance. The value of CFB should be offset between channel 1 and channel 2
so that the idle switching difference is greater than 40kHz. Please refer to the
Application / Test Circuit.
ROFA
ROFB
Potentiometer used to manually trim the DC offset on the output of the TK2350.
Resistor that limits the manual DC offset trim range and allows for more precise
adjustment.
RREF
CA
Bias resistor. Locate close to pin 15 of the TC2001 and ground at pin 20 of the
TC2001.
BIASCAP decoupling capacitor. Should be located close to pin 1 of the TC2001 and
grounded at pin 20 of the TC2001.
DB
Bootstrap diode. This diode charges up the bootstrap capacitors when the output is
low (at VNN) to drive the high side gate circuitry. A fast or ultra fast recovery diode
is recommended for the bootstrap circuitry. In addition, the bootstrap diode must be
able to sustain the entire VPP-VNN voltage. Thus, for most applications, a 150V (or
greater) diode should be used.
CB
High frequency bootstrap capacitor, which filters the high side gate drive supply.
This capacitor must be located as close to VBOOT1 (pin 57 of the TP2350B) or
VBOOT2 (pin 27 of the TP2350B) for reliable operation. The “negative” side of CB
should be connected directly to the HO1COM (pin 47 of the TP2350B) or HO2COM
(pin 37 of the TP2350B). Please refer to the Application / Test Circuit.
Bulk bootstrap capacitor that supplements CB during “clipping” events, which result
in a reduction in the average switching frequency.
CBAUX
RB
Bootstrap resistor that limits CBAUX charging current during TK2350 power up
(bootstrap supply charging).
CS
Supply decoupling for the power supply pins. For optimum performance, these
components should be located close to the TC2001 and TP2350B and returned to
their respective ground as shown in the Application/Test Circuit.
RVNN1
Main overvoltage and undervoltage sense resistor for the negative supply (VNN).
Please refer to the Electrical Characteristics Section for the trip points as well as the
hysteresis band. Also, please refer to the Over / Under-voltage Protection section in
the Application Information for a detailed discussion of the internal circuit operation
and external component selection.
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RVNN2
RVPP1
RVPP2
Secondary overvoltage and undervoltage sense resistor for the negative supply
(VNN). This resistor accounts for the internal VNNSENSE bias of 1.25V. Nominal
resistor value should be three times that of RVNN1. Please refer to the Over / Under-
voltage Protection section in the Application Information for a detailed discussion of
the internal circuit operation and external component selection.
Main overvoltage and undervoltage sense resistor for the positive supply (VPP).
Please refer to the Electrical Characteristics Section for the trip points as well as the
hysteresis band. Also, please refer to the Over / Under-voltage Protection section in
the Application Information for a detailed discussion of the internal circuit operation
and external component selection.
Secondary overvoltage and undervoltage sense resistor for the positive supply
(VPP). This resistor accounts for the internal VPPSENSE bias of 2.5V. Nominal
resistor value should be equal to that of RVPP1. Please refer to the Over / Under-
voltage Protection section in the Application Information for a detailed discussion of
the internal circuit operation and external component selection.
RS
Over-current sense resistor. Please refer to the section, Setting the Over-current
Threshold, in the Application Information for a discussion of how to choose the value
of RS to obtain a specific current limit trip point.
ROCR
Over-current “trim” resistor, which, in conjunction with RS, sets the current trip point.
Please refer to the section, Setting the Over-current Threshold, in the Application
Information for a discussion of how to calculate the value of ROCR
.
COCR
Over-current filter capacitor, which filters the overcurrent signal at the OCR pins to
account for the half-wave rectified current sense circuit internal to the TC2001. A
typical value for this component is 220pF. In addition, this component should be
located near pin 14 or pin 16 of the TC2001 as possible.
CHBR
Supply decoupling for the high current Half-bridge supply pins. These components
must be located as close to the output MOSFETs as possible to minimize output
ringing which causes power supply overshoot. By reducing overshoot, these
capacitors maximize both the TP2350B and output MOSFET reliability. These
capacitors should have good high frequency performance including low ESR and
low ESL. In addition, the capacitor rating must be twice the maximum VPP voltage.
Panasonic EB capacitors are ideal for the bulk storage (nominally 33uF) due to their
high ripple current and high frequency design.
RG
DG
Gate resistor, which is used to control the MOSFET rise/ fall times. This resistor
serves to dampen the parasitics at the MOSFET gates, which, in turn, minimizes
ringing and output overshoots. The typical power rating is 1 watt.
Gate diode, placed in parallel to the gate resistor. This diode will help discharge the
parasitic capacitance at the MOSFET gates, thus decreasing the MOSFET fall time.
This helps reduce shoot through current between the top side and bottom side
output MOSFETs.
CZ
RZ
Zobel capacitor, which in conjunction with RZ, terminates the output filter at high
frequencies. Use a high quality film capacitor capable of sustaining the ripple current
caused by the switching outputs.
Zobel resistor, which in conjunction with CZ, terminates the output filter at high
frequencies. The combination of RZ and CZ minimizes peaking of the output filter
under both no load conditions or with real world loads, including loudspeakers which
usually exhibit a rising impedance with increasing frequency. Depending on the
program material, the power rating of RZ may need to be adjusted. The typical
power rating is 2 watts.
LO
Output inductor, which in conjunction with CO, demodulates (filters) the switching
waveform into an audio signal. Forms a second order filter with a cutoff frequency
of
and a quality factor of
.
LO CO
fC = 1 (2π LO C O
)
Q = RL CO
CO
Output capacitor, which, in conjunction with LO, demodulates (filters) the switching
waveform into an audio signal. Forms a second order low-pass filter with a cutoff
frequency of
and a quality factor of
)
. Use
LO CO
fC = 1 (2π LO C O
Q = RL CO
a high quality film capacitor capable of sustaining the ripple current caused by the
switching outputs.
DD
Drain diode. This diode must be connected from the drain of the high side output
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MOSFET to the drain of the low side output MOSFET. This diode absorbs any high
frequency overshoots caused by the output inductor LO during high output current
conditions. In order for this diode to be effective it must be connected directly to the
drains of both the top and bottom side output MOSFET. An ultra fast recovery diode
that can sustain the entire VPP-VNN voltage should be used here. In most
applications a 150V or greater diode must be used.
DS
Source diode. This diode must be connected from the source of the high side
output MOSFET to the source of the low side output MOSFET. This diode absorbs
any high frequency undershoots caused by the output inductor LO during high output
current conditions. In order for this diode to be effective it must be connected
directly to the sources of both the top and bottom sides output MOSFETs. An ultra
fast recovery diode that can sustain the entire VPP-VNN voltage should be used
here. In most applications a 150V or greater diode must be used.
RB
CB
Output MOSFET snubber resistor. This resistor forms a low pass filter with CO with
a frequency of
. This RC filter removes any high frequency overshoots
fC = 1 (2πRBCB )
that can be present on the switching output waveform. This RC filter must be
connected right across the drain and source of the low side output MOSFET.
Output MOSFET snubber capacitor. This resistor forms a low pass filter with RB
with a frequency of
. This RC filter removes any high frequency
fC = 1 (2πRBCB )
overshoots that can be present on the switching output waveform. This RC filter
must be connected right across the drain and source of the low side output
MOSFET.
RS
Source resistor. This resistor is in series between HOCOM and the source of the
top side output MOSFET. This resistor serves to limit the voltage swing at the
HOCOM pin (pins 37 and 47 of the TP2350B) to protect the TP2350B during any
output overshoots/undershoots. Since this resistor alters the rise and fall times of
the gate on the high side output MOSFET an additional resistor of the same value is
placed in series with LOCOM and the source of the bottom side output MOSFET to
match the rise and fall times of the top side to the bottom side.
RPG
Gate resistor for the output MOSFET for the switchmode power supply. Controls
the rise time, fall time, and reduces ringing for the gate of the output MOSFET for
the switchmode power supply.
QB
Output MOSFET for the switchmode power supply to generate the VN10. This
output MOSFET must be a P channel device.
DSW
LSW
Flywheel diode for the internal VN10 buck converter. This diode also prevents
VN10SW from going more than one diode drop negative with respect to VNN.
VN10 generator filter inductor. This inductor should be sized appropriately so that
LSW pass 0.5A of current without saturation, and VN10 does not overshoot with
respect to VNN during TK2350 turn on.
CSW
VN10 generator filter capacitors. The high frequency capacitor (0.1uF) must be
located close to the VN10 pins (pin 41 and 43 of the TP2350B) to maximize device
performance. The bulk capacitor (100uF) should be sized appropriately such that
the VN10 voltage does not overshoot with respect to VNN during TK2350 turn on.
VN10 generator feedback resistor. This resistor sets the nominal VN10 voltage.
With RSWFB equal to 1kΩ, the VN10 voltage generated will typically be 11V above
VNN.
RSWFB
CSWFB
VN10 generator feedback capacitor. This capacitor, in conjunction with RSWFB, filters
the VN10 feedback signal such that the loop is unconditionally stable.
13 of 36
TK2350, Rev 1.4/04.03
Tripath Technology, Inc. - Technical Information
Typical Performance
Efficiency versus Output Power
100
THD+N versus Output Power
10
5
R = 8Ω
L
f = 1kHz
90
BBM = 80nS
Vs =+28V
R = 4Ω
L
80
70
60
50
BW = 22Hz - 22kHz
2
1
R = 8Ω
R = 4Ω
L
L
0.5
0.2
0.1
40
f = 1kHz
30
20
10
0
BBM = 80nS
Vs =+28V
0.05
THD+N =<10%
0.02
0.01
2
5
10
20
50
100
200
0
20
40
60
80
100
120
Output Power (W)
Output Power (W)
Intermodulation Performance
Intermodulation Performance
RL = 4Ω
RL = 8Ω
+0
+0
-10
-10
19kHz, 20kHz, 1:1
Pout = 40W/Channel
0dBr = 12.65Vrms
Vs = +28V
19kHz, 20kHz, 1:1
Pout = 20W/Channel
0dBr = 12.65Vrms
Vs =+28V
-20
-30
-20
-30
BW = 22Hz - 22kHz
BW = 22Hz - 22kHz
-40
-40
-50
-50
-60
-60
-70
-70
-80
-80
-90
-90
-100
-110
-100
-110
-120
20
-120
50
100
200
500
1k
2k
5k
10k
20k
20
50
100
200
500
1k
2k
5k
10k
20k
Frequency (Hz)
Frequency (Hz)
Noise Floor
Channel Separation versus Frequency
-70
-75
-40
-45
V =+28V
S
BBM = 80nS
16K FFT
Pout = 40W/Channel @ 4Ω
Pout = 20W/Channel @ 8Ω
VS = +28V
-50
Fs = 48kHz
-80
-85
-90
-95
-55 BW = 22Hz - 22kHz
BW = 22Hz - 22kHz
-60
-65
-70
-75
-80
-85
-90
-95
-100
-105
-110
-115
RL = 4Ω
RL = 8Ω
-100
-105
-110
-115
-120
-120
20
50
100
200
500
1k
2k
5k
10k
20k
20
50
100
200
500
1k
2k
5k
10k
20k
Frequency (Hz)
Frequency (Hz)
14 of 36
TK2350, Rev 1.4/04.03
Tripath Technology, Inc. - Technical Information
Typical Performance
THD+N versus Frequency versus Break Before Make
THD+N versus Frequency versus Break Before Make
RL = 8Ω
RL = 4Ω
10
10
5
Pout = 20W/Channel
Vs =+28V
5
Pout = 40W/Channel
Vs =+28V
BW = 22Hz - 22kHz
BW = 22Hz - 22kHz
2
1
2
1
0.5
0.5
0.2
0.1
0.2
0.1
BBM = 120nS
BBM = 120nS
0.05
0.05
0.02
0.01
0.02
0.01
BBM = 80nS
BBM = 80nS
2k
20
50
100
200
500
1k
2k
5k
10k
20k
20
50
100
200
500
1k
5k
10k
20k
Frequency (Hz)
Frequency (Hz)
THD+N versus Frequency versus Bandwidth
THD+N versus Frequency versus Bandwidth
RL = 4Ω
RL = 8Ω
10
5
10
5
Pout = 40W/Channel
Vs =+28V
Pout = 20W/Channel
Vs =+28V
BBM = 80nS
BBM = 80nS
2
1
2
1
0.5
0.5
0.2
0.1
0.2
0.1
BW = 30kHz
BW = 22kHz
0.05
0.05
BW = 30kHz
0.02
0.01
0.02
0.01
BW = 22kHz
100 200
20
50
100
200
500
1k
2k
5k
10k
20k
20
50
500
1k
2k
5k
10k
20k
Frequency (Hz)
Frequency (Hz)
THD+N versus Output Power versus Supply Voltage
THD+N versus Output Power versus Supply Voltage
R = 4Ω
R = 8Ω
L
L
10
5
10
5
f = 1kHz
f = 1kHz
BBM = 80nS
BBM = 80nS
BW = 22Hz - 22kHz
BW = 22Hz - 22kHz
2
1
2
1
23V
28V
35V
23V
28V
35V
0.5
0.5
0.2
0.1
0.2
0.1
0.05
0.05
0.02
0.02
0.01
0.01
1
2
5
10
20
50
100
200
2
5
10
20
50
100
200
Output Power (W)
Output Power (W)
15 of 36
TK2350, Rev 1.4/04.03
Tripath Technology, Inc. - Technical Information
Typical Performance
Efficiency versus Output Power
THD+N versus Output Power
100
10
5
RL = 8Ω
f = 1kHz
BBM = 80nS
Vs =+45V
RL = 4Ω
BW = 22Hz - 22kHz
80
2
1
R = 8Ω
R = 4Ω
L
L
60
0.5
0.2
0.1
40
f = 1kHz
0.05
BBM = 80nS
20
Vs =+45V
THD+N<10%
0.02
0.01
0
0
50
100
150
200
250
300
2
5
10
20
50
100
200
500
Output Power (W)
Output Power (W)
Intermodulation Performance
Intermodulation Performance
L = 4Ω
RL = 8Ω
R
+0
+0
19kHz, 20kHz, 1:1
Pout = 30W/Channel
0dBr = 15.5Vrms
Vs =+45V
-10
-20
-30
-40
-50
-60
-70
-80
-90
-100
-110
19kHz, 20kHz, 1:1
Pout = 60W/Channel
0dBr = 15.5Vrms
Vs = +45V
-10
-20
-30
BW = 22Hz - 22kHz
BW = 22Hz - 22kHz
-40
-50
-60
-70
-80
-90
-100
-110
-120
-120
20
50
100
200
500
1k
2k
5k
10k
20k
20
50
100
200
500
1k
2k
5k
10k
20k
Frequency (Hz)
Frequency (Hz)
Noise Floor
Channel Separation versus Frequency
-70
-75
-40
-45
VS = +/-45V
BBM = 80nS
16kFFT
Pout = 60W/Channel @ 4Ω
Pout = 30W/Channel @ 8Ω
VS = +45V
-50
-55
FS = 48kHz
BW = 22Hz - 22kHz
-80 BW = 22Hz-22kHz
-60
-85
-90
-65
-70
-75
-95
-80
-85
RL = 4Ω
RL = 8Ω
-100
105
-90
-95
-100
-105
-110
-115
-110
-115
-120
-120
20
50
100
200
500
1k
2k
5k
10k
20k
10k
20
50
100
200
500
1k
2k
5k
20k
Frequency (Hz)
Frequency (Hz)
16 of 36
TK2350, Rev 1.4/04.03
Tripath Technology, Inc. - Technical Information
Typical Performance
THD+N versus Frequency versus Break Before Make
THD+N versus Frequency versus Break Before Make
RL = 4Ω
RL = 8Ω
10
5
10
Pout = 30W/Channel
Pout = 60W/Channel
Vs =+45V
Vs =+45V
5
BW = 20Hz - 22kHz
BW = 22Hz - 22kHz
2
1
2
1
0.5
0.5
0.2
0.1
0.2
BBM = 120nS
BBM = 120nS
0.1
0.05
0.05
BBM = 80nS
0.02
0.01
0.02
BBM = 80nS
5k
0.01
20
20
50
100
200
500
1k
2k
5k
10k
20k
50
100
200
500
1k
2k
10k
20k
Frequency (Hz)
Frequency (Hz)
THD+N versus Frequency versus Bandwidth
RL = 4Ω
THD+N versus Frequency versus Bandwidth
10
5
RL = 8Ω
10
Pout = 60W/Channel
Vs =+45V
Pout = 30W/Channel
Vs =+45V
BBM = 80nS
5
BBM = 80nS
2
1
2
1
0.5
0.5
0.2
0.1
0.2
0.1
BW = 30kHz
BW = 22kHz
BW = 30kHz
BW = 22kHz
0.05
0.05
0.02
0.01
0.02
0.01
20
50
100
200
500
1k
2k
5k
10k
20k
Frequency (Hz)
20
50
100
200
500
1k
2k
5k
10k
20k
Frequency (Hz)
THD+N versus Output Power versus Supply Voltage
THD+N versus Output Power versus Supply Voltage
R = 4Ω
R = 8Ω
L
L
10
5
10
5
f = 1kHz
f = 1kHz
BBM = 80nS
BW = 22Hz - 22kHz
BBM = 80nS
BW = 22Hz - 22kHz
2
1
2
1
39V
45V
54V
39V
54V
45V
0.5
0.5
0.2
0.1
0.2
0.1
0.05
0.05
0.02
0.02
0.01
0.01
1
2
5
10
20
50
100
200
500
2
5
10
20
50
100
200
500
Output Power (W)
Output Power (W)
17 of 36
TK2350, Rev 1.4/04.03
Tripath Technology, Inc. - Technical Information
Application Information
Figure 1 is a simplified diagram of one channel (Channel 1) of a TK2350 amplifier to assist in
understanding its operation.
TC2001
TP2350B
BBM0 26
51 OCS1HP
VPP
BBM1
25
22
OVER
DB
CS
CURRENT
DETECTION
RS
OAOUT1
VN10
RB
OCS1HN
50
57 VBOOT1
RI
RF
CI
V5
QO
RG
RG
INV1 23
ROFB
+
+
CB
0.1uF
-
Y1
9
17
16
48 HO1
47 HO1COM
CBAUX
+
CHBR
AGND
V5
VN10
OUTPUT
FILTER
QO
RS
Processing
&
Modulation
ROFB
10
Y1B
44 LO1
RL
ROFA
45
LO1COM
COF
2.5V
V5
54 OCS1LP
Offset Trim
Circuit
OVER
CURRENT
DETECTION
CA
5V
53 OCS1LN
BIASCAP
1
VNN
41,43
VN10
VN10
CS
CSW
VNN
VNN
42
6
MUTE 24
REF 15
OCR1
7
VNN
V5
5V
CS
5
AGND
RREF
RVNN1
16
OCR1
COCR
VNNSENSE 17
V5
VNN
VPP
OVER/
OVER
CURRENT
DETECTION
UNDER
VOLTAGE
DETECTION
ROCR
RVPP1
VPPSENSE
19
RFBA
RFBA
RFBC
RVNN2
RVPP1
6
7
FBKOUT1
FBKGND1
V5
V5
RFBC
CFB
V5
21
RFBB
RFBB
5V
CS
HMUTE
8
AGND 20
Analog Ground
Power Ground
F. BEAD
Figure 1: Simplified TK2350 Amplifier
TK2350 Basic Amplifier Operation
The audio input signal is fed to the processor internal to the TC2001, where a switching pattern is
generated. The average idle (no input) switching frequency is approximately 700kHz. With an
input signal, the pattern is spread spectrum and varies between approximately 200kHz and
1.5MHz depending on input signal level and frequency. Complementary copies of the switching
pattern is output through the Y1 and Y1B pins on the TC2001. These switching patterns are input
to the TP230 where they are level-shifted by the MOSFET drivers and then output to the gates
(HO1 and LO1) of external power MOSFETs that are connected as a half bridge. The output of
the half bridge is a power-amplified version of the switching pattern that switches between VPP
and VNN. This signal is then low-pass filtered to obtain an amplified reproduction of the audio
input signal.
The TC2001 processor is operated from a 5-volt supply. In the generation of the switching
patterns for the output MOSFETs, the processor inserts a “break-before-make” dead time between
the turn-off of one transistor and the turn-on of the other in order to minimize shoot-through
currents in the external MOSFETs. The dead time can be programmed by setting the break-
before-make control bits, BBM1 and BBM0. Feedback information from the output of the half-
bridge is supplied to the processor via FBKOUT1. Additional feedback information to account for
ground bounce is supplied via FBKGND1.
18 of 36
TK2350, Rev 1.4/04.03
Tripath Technology, Inc. - Technical Information
The MOSFET drivers in the TP2350B are operated from voltages obtained from VN10 and
LO1COM for the low-side driver, and VBOOT1 and HO1COM for the high-side driver. VN10 must
be a regulated 10V above VNN.
N-Channel MOSFETs are used for both the top and bottom of the half bridge. The gate resistors,
RG, are used to control MOSFET slew rate and thereby minimize voltage overshoots.
Circuit Board Layout
The TK2350 is a power (high current) amplifier that operates at relatively high switching
frequencies. The output of the amplifier switches between VPP and VNN at high speeds while
driving large currents. This high-frequency digital signal is passed through an LC low-pass filter to
recover the amplified audio signal. Since the amplifier must drive the inductive LC output filter and
speaker loads, the amplifier outputs can be pulled above the supply voltage and below ground by
the energy in the output inductance. To avoid subjecting the TK2350 to potentially damaging
voltage stress, it is critical to have a good printed circuit board layout. It is recommended that
Tripath’s layout and application circuit be used for all applications and only be deviated from after
careful analysis of the effects of any changes. Please refer to the TK2350 evaluation board
document, RB-TK2350, available on the Tripath website, at www.tripath.com.
The following components are important to place near either their associated TK2350 or output
MOSFET pins. The recommendations are ranked in order of layout importance, either for proper
device operation or performance considerations.
-
The capacitors, CHBR, provide high frequency bypassing of the amplifier power supplies
and will serve to reduce spikes across the supply rails. Please note that both mosfet
half-bridges must be decoupled separately. In addition, the voltage rating for CHBR
should be at least 150V as this capacitor is exposed to the full supply range, VPP-VNN.
-
CFB removes very high frequency components from the amplifier feedback signals and
lowers the output switching frequency by delaying the feedback signals. In addition, the
value of CFB is different for channel 1 and channel 2 to keep the average switching
frequency difference greater than 40kHz. This minimizes in-band audio noise. Locate
these capacitors as close to their respective TC2001 pin as possible.
-
-
To minimize noise pickup and minimize THD+N, RFBC should be located as close to the
TC2001 as possible. Make sure that the routing of the high voltage feedback lines is
kept far away from the input op amps or significant noise coupling may occur. It is best
to shield the high voltage feedback lines by using a ground plane around these traces
as well as the input section. The feedback and feedback ground traces should be
routed together in parallel.
CB, CSW provides high frequency bypassing for the VN10 and bootstrap supplies. Very
high currents are present on these supplies.
In general, to enable placement as close to the TK2350, and minimize PCB parasitics, the
capacitors CFB, CB and CSW should be surface mount types, located on the “solder” side of the
board.
19 of 36
TK2350, Rev 1.4/04.03
Tripath Technology, Inc. - Technical Information
Some components are not sensitive to location but are very sensitive to layout and trace routing.
-
To maximize the damping factor and reduce distortion and noise, the modulator
feedback connections should be routed directly to the pins of the output inductors. LO.
Please refer to the RB-TK2350 for more information.
-
-
The output filter capacitor, CO, and zobel capacitor, CZ, should be star connected with
the load return. The output ground feedback signal should be taken from this star point.
The modulator feedback resistors, RFBA and RFBB, should all be grounded and attached
to 5V together. These connections will serve to minimize common mode noise via the
differential feedback. Please refer to the RB-TK2350 evaluation board for more
information.
-
-
The feedback signals that come directly from the output inductors are high voltage and
high frequency in nature. If they are routed close to the input nodes, INV1 and INV2, the
high impedance inverting opamp pins will pick up noise. This coupling will result in
significant background noise, especially when the input is AC coupled to ground, or an
external source such as a CD player or signal generator is connected. Thus, care
should be taken such that the feedback lines are not routed near any of the input
section.
To minimize the possibility of any noise pickup, the trace lengths of INV1 and INV2
should be kept as short as possible. This is most easily accomplished by locating the
input resistors, RI and the input stage feedback resistors, RF as close to the TC2001 as
possible. In addition, the offset trim resistor, ROFB, which connects to either INV1, or
INV2, should be located close to the TC2001 input section.
TK2350 Grounding
Proper grounding techniques are required to maximize TK2350 functionality and performance.
Parametric parameters such as THD+N, Noise Floor and Crosstalk can be adversely affected if
proper grounding techniques are not implemented on the PCB layout. The following discussion
highlights some recommendations about grounding both with respect to the TK2350 as well as
general “audio system” design rules.
The TK2350 is divided into three sections: the input section, which is the TC2001, the MOSFET
driver section, which is the TP2350B, and the output (high voltage) section, which is the output
MOSFETs. On the TK2350 evaluation board, the ground is also divided into distinct sections, one
for the input and the MOSFET driver, and another one for the output. To minimize ground loops
and keep the audio noise floor as low as possible, the two grounds must be only connected at a
single point. Depending on the system design, the single point connection may be in the form of a
ferrite bead or a PCB trace.
The analog grounds, must be connected to pin 20 on the TC2001 and pin 5 on the TP2350B. The
ground for the V5 power supply should connect directly to pin 20 of the TC2001. Additionally, any
external input circuitry such as preamps, or active filters, should be referenced to pin 20 on the
TC2001.
For the power section, Tripath has traditionally used a “star” grounding scheme. Thus, the load
ground returns and the power supply decoupling traces are routed separately back to the power
supply. In addition, any type of shield or chassis connection would be connected directly to the
ground star located at the power supply. These precautions will both minimize audible noise and
enhance the crosstalk performance of the TK2350.
20 of 36
TK2350, Rev 1.4/04.03
Tripath Technology, Inc. - Technical Information
The TC2001 incorporates a differential feedback system to minimize the effects of ground bounce
and cancel out common mode ground noise. As such, the feedback from the output ground for
each channel needs to be properly sensed. This can be accomplished by connecting the output
ground “sensing” trace directly to the star formed by the output ground return, output capacitor,
CO, and the zobel capacitor, CZ. Refer to the Application / Test Circuit for a schematic description.
TK2350 Amplifier Gain
The gain of the TK2350 is the product of the input stage gain and the modulator gain for the
TC2001. Please refer to the sections, Input Stage Design, and Modulator Feedback Design, for a
complete explanation of how to determine the external component values.
A
A
VTK2350 = AVINPUTSTAG
E
* AV MODULATOR
R
F
R
FBC * (RFBA + RFBB )
VTK2350 ≈ −
+ 1
R
I
R
FBA * RFBB
For example, using a TC2001 with the following external components,
RI = 20kΩ
RF = 20kΩ
R
FBA = 1kΩ
RFBB = 1.07kΩ
RFBC = 13.3kΩ
20k Ω 13.3k Ω * (1.0k Ω + 1.07k Ω)
V
V
A
VTK2350 ≈ −
+ 1 = - 10.71
49.9k Ω
1.0k Ω * 1.07k Ω
Input Stage Design
The TC2001 input stage is configured as an inverting amplifier, allowing the system designer
flexibility in setting the input stage gain and frequency response. Figure 2 shows a typical
application where the input stage is a constant gain inverting amplifier. The input stage gain
should be set so that the maximum input signal level will drive the input stage output to 4Vpp.
The gain of the input stage, above the low frequency high pass filter point, is that of a simple
inverting amplifier:
R
F
A
VINPUTSTAG E = −
R
I
21 of 36
TK2350, Rev 1.4/04.03
Tripath Technology, Inc. - Technical Information
TC2001
OAOUT1
22
V5
CI
RI
RF
INV1 23
+
-
+
INPUT1
AGND
BIASCAP
V5
CI
RI
+
-
28
27
INV2
+
RF
INPUT2
AGND
OAOUT2
Figure 2: TC2001 Input Stage
Input Capacitor Selection
CIN can be calculated once a value for RIN has been determined. CIN and RIN determine the input
low-frequency pole. Typically this pole is set at 10Hz. CIN is calculated according to:
CIN = 1 / (2π x FP x RIN)
where: RIN = Input resistor value in ohms
FP = Input low frequency pole (typically 10Hz)
Modulator Feedback Design
The modulator converts the signal from the input stage to the high-voltage output signal. The
optimum gain of the modulator is determined from the maximum allowable feedback level for the
modulator and maximum supply voltages for the power stage. Depending on the maximum supply
voltage, the feedback ratio will need to be adjusted to maximize performance. The values of RFBA
,
RFBB and RFBC (see explanation below) define the gain of the modulator. Once these values are
chosen, based on the maximum supply voltage, the gain of the modulator will be fixed even with
as the supply voltage fluctuates due to current draw.
For the best signal-to-noise ratio and lowest distortion, the maximum modulator feedback voltage
should be approximately 4Vpp. This will keep the gain of the modulator as low as possible and
still allow headroom so that the feedback signal does not clip the modulator feedback stage.
Figure 3 shows how the feedback from the output of the amplifier is returned to the input of the
modulator. The input to the modulator (FBKOUT1/FBKGND1 for channel 1) can be viewed as
inputs to an inverting differential amplifier. RFBA and RFBB bias the feedback signal to approximately
2.5V and RFBC scales the large OUT1/OUT2 signal to down to 4Vpp.
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1/2 TC2001
V5
RFBA
RFBA
RFBC
FBKOUT1
Processing
&
Modulation
28
27
OUT1
FBKGND1
OUT1 GROUND
RFBC
RFBB
RFBB
AGND
Figure 3: Modulator Feedback
The modulator feedback resistors are:
R
R
FBA = User specified, typically 1KΩ
R
FBA * VPP
FBB
=
(VPP - 4)
R
FBA * VPP
R
A
FBC
=
4
R
FBC * (RFBA + RFBB )
V - MODULATOR
≈
+ 1
R
FBA * RFBB
The above equations assume that VPP=|VNN|.
For example, in a system with VPPMAX=52V and VNNMAX=-52V,
R
FBA = 1kΩ, 1%
RFBB = 1.08kΩ, use 1.07kΩ, 1%
RFBC = 13.0kΩ, use 13.3kΩ, 1%
The resultant modulator gain is:
13.3k Ω * (1.0k Ω + 1.07k Ω)
A
V - MODULATOR
≈
+ 1 = 26.73V/V
1.0k Ω * 1.07k Ω
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Tripath Technology, Inc. - Technical Information
Mute
When a logic high signal is supplied to MUTE, both amplifier channels are muted (both high- and
low-side transistors are turned off). When a logic level low is supplied to MUTE, both amplifiers
are fully operational. There is a delay of approximately 200 milliseconds between the de-assertion
of MUTE and the un-muting of the TK2350.
Turn-on & Turn-off Noise
If turn-on or turn-off noise is present in a TK2350 amplifier, the cause is frequently due to other
circuitry external to the TK2350. While the TK2350 has circuitry to suppress turn-on and turn-off
transients, the combination of the power supply and other audio circuitry with the TK2350 in a
particular application may exhibit audible transients. One solution that will completely eliminate
turn-on and turn-off pops and clicks is to use a relay to connect/disconnect the amplifier from the
speakers with the appropriate timing at power on/off. The relay can also be used to protect the
speakers from a component failure (e.g. shorted output MOSFET), which is a protection
mechanism that some amplifiers have. Circuitry external to the TK2350 would need to be
implemented to detect these failures.
DC Offset
While the DC offset voltages that appear at the speaker terminals of a TK2350 amplifier are
typically small, Tripath recommends that any offsets during operation be nulled out of the amplifier
with a circuit like the one shown connected to IN1 and IN2 in the Test/Application Circuit.
It should be noted that the DC voltage on the output of a TK2350 amplifier with no load in mute will
not be zero. This offset does not need to be nulled. The output impedance of the amplifier in
mute mode is approximately 10KΩ. This means that the DC voltage drops to essentially zero
when a typical load is connected.
HMUTE
The HMUTE pin on the TC2001 is a 5V logic output that indicates various fault conditions within
the device. These conditions include: over-current, overvoltage and undervoltage. The HMUTE
output is capable of directly driving an LED through a series 2kΩ resistor.
Over-current Protection
The TK2350 has over-current protection circuitry to protect itself and the output transistors from
short-circuit conditions. The TK2350 uses the voltage across a resistor RS (measured via
OCS1HP, OCS1HN, OCS1LP and OCS1LN of the TP2350B) that is in series with each output
MOSFET to detect an over-current condition. RS and ROCR are used to set the over-current
threshold. The OCS pins must be Kelvin connected for proper operation. See “Circuit Board
Layout” in Application Information for details.
When the voltage across ROCR becomes greater than VTOC (typically 0.97V) the TC2001 will shut
off the output stages of its amplifiers. The occurrence of an over-current condition is latched in the
TK2350 and can be cleared by toggling the MUTE input or cycling power.
Setting Over-current Threshold
RS and ROCR determine the value of the over-current threshold, ISC:
ISC = 3580 x (VTOC – IBIAS * ROCR)/(R OCR * RS)
R
OCR = (3580 x VTOC)/(ISC * RS+3580 * IBIAS
)
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where:
RS and ROCR are in Ω
VTOC = Over-current sense threshold voltage (See Electrical Characteristics Table)
= 0.97V typically
IBIAS = 20uA
For example, to set an ISC of 30A, ROCR = 9.63KΩ and RS will be 10mΩ.
As high-wattage resistors are usually only available in a few low-resistance values (10mΩ, 25mΩ
and 50mΩ), ROCR can be used to adjust for a particular over-current threshold using one of these
values for RS.
It should be noted that the addition of the bulk CHBR capacitor shown in the Application / Test
Diagram will increase the ISC level. Thus, it will be larger than the theoretical value shown above.
Once the designer has settled on a layout and specific CHBR value, the system ISC trip point can be
adjusted by increasing the ROCR value. The ROCR should be increased to a level that allows
expected range of loads to be driven well into clipping without current limiting while still protecting
the output MOSFETs in case of a short circuit condition.
Auto Recovery Circuit for Overcurrent Fault Condition
If an overcurrent fault condition occurs the HMUTE pin (pin 8 of the TC2001) will be latched high
and the amplifier will be muted. The amplifier will remain muted until the MUTE pin (pin 24 of the
TC2001) is toggled high and then low or the power supplies are turned off and then on again. The
circuit shown below in Figure 4 is a circuit that will detect if HMUTE is high and then toggle the
mute pin high and then low, thus resetting the amplifier. The LED, D1 will turn on when HMUTE is
high. The reset time has been set for approximately 2.5 seconds. The duration of the reset time
is controlled by the RC time constant set by R306 and C311. To increase the reset, time increase
the value of C311. To reduce the reset time, reduce the value of C311. Please note that this
circuit is optional and is not included on the RB-TK2350-X evaluation boards.
V5
R309
1kΩ , 5%
D1
LED
R311
R306
510kΩ , 5%
R307
10kΩ , 5%
R308
10kΩ , 5%
Q305
2N3906
1kΩ , 5%
R311
1kΩ , 5%
MUTE
Pin 24
C311
10uF, NP
Q302
2N3904
Q303
2N7002
R311
1kΩ , 5%
Jumper
HMUTE
Pin 8
Q304
2N3904
R310
1kΩ , 5%
remove jumper to
enable mute
AGND
Figure 4: Overcurrent Autorecovery Circuit
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Tripath Technology, Inc. - Technical Information
Over- and Under-Voltage Protection
The TC2001 senses the power rails through external resistor networks connected to VNNSENSE
and VPPSENSE. The over- and under-voltage limits are determined by the values of the resistors
in the networks, as described in the table “Test/Application Circuit Component Values”. If the
supply voltage falls outside the upper and lower limits determined by the resistor networks, the
TC2001 shuts off the output stages of the amplifiers. The removal of the over-voltage or under-
voltage condition returns the TK2350 to normal operation. Please note that trip points specified in
the Electrical Characteristics table are at 25°C and may change over temperature.
The TC2001 has built-in over and under voltage protection for both the VPP and VNN supply rails.
The nominal operating voltage will typically be chosen as the supply “center point.” This allows
the supply voltage to fluctuate, both above and below, the nominal supply voltage.
VPPSENSE (pin 19) performs the over and undervoltage sensing for the positive supply, VPP.
VNNSENSE (pin 17) performs the same function for the negative rail, VNN. When the current
through RVPPSENSE (or RVNNSENSE) goes below or above the values shown in the Electrical
Characteristics section (caused by changing the power supply voltage), the TK2350 will be muted.
VPPSENSE is internally biased at 2.5V and VNNSENSE is biased at 1.25V.
Once the supply comes back into the supply voltage operating range (as defined by the supply
sense resistors), the TK2350 will automatically be unmuted and will begin to amplify. There is a
hysteresis range on both the VPPSENSE and VNNSENSE pins. If the amplifier is powered up in
the hysteresis band the TK2350 will be muted. Thus, the usable supply range is the difference
between the over-voltage turn-off and under-voltage turn-off for both the VPP and VNN supplies. It
should be noted that there is a timer of approximately 200mS with respect to the over and under
voltage sensing circuit. Thus, the supply voltage must be outside of the user defined supply range
for greater than 200mS for the TK2350 to be muted.
Figure 5 shows the proper connection for the Over / Under voltage sense circuit for both the
VPPSENSE and VNNSENSE pins.
V5
VNN
TC2001
RVNN2
RVNN1
17
19
VNNSENSE
V5
VPP
RVPP2
RVPP1
VPPSENSE
Figure 5: Over / Under voltage sense circuit
The equation for calculating RVPP1 is as follows:
VPP
VPPSENSE
R
VPP1 =
I
Set RVPP2 = RVPP1 .
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The equation for calculating RVNNSENSE is as follows:
VNN
R
VNN1 =
I
VNNSENSE
Set
.
R
VNN2 = 3 × RVNN1
IVPPSENSE or IVNNSENSE can be any of the currents shown in the Electrical Characteristics
table for VPPSENSE and VNNSENSE, respectively.
The two resistors, RVPP2 and RVNN2 compensate for the internal bias points. Thus, RVPP1 and RVNN1
can be used for the direct calculation of the actual VPP and VNN trip voltages without considering
the effect of RVPP2 and RVNN2
.
Using the resistor values from above, the actual minimum over voltage turn off points will be:
VPP MIN_OV_TUR N_OFF = RVPP1 × IVPPSENSE (MIN_OV_TU RN_OFF)
VNN MIN_OV_TUR N_OFF = −(RVNN1 × IVNNSENSE (MIN_OV_TU RN_OFF)
)
The other three trip points can be calculated using the same formula but inserting the appropriate
IVPPSENSE (or IVNNSENSE) current value. As stated earlier, the usable supply range is the difference
between the minimum overvoltage turn off and maximum under voltage turn-off for both the VPP
and VNN supplies.
VPP RANGE = VPP MIN_OV_TUR N_OFF - VPP MAX_UV_TUR N_OFF
VNN RANGE = VNN MIN_OV_TUR N_OFF - VNN MAX_UV_TUR N_OFF
VN10 Supply and Switch Mode Power Supply Controller
VN10 is an additional supply voltage required by the TP2350B. VN10 must be 10 volts more
positive than the nominal VNN. VN10 must track VNN. Generating the VN10 supply requires
some care.
The proper way to generate the voltage for VN10 is to use a 10V-postive supply voltage
referenced to the VNN supply. The TP2350B has an internal switch mode power supply controller
which generates the necessary floating power supply for the MOSFET driver stage in the
TP2350B (nominally 11V with the external components shown in Application / Test Circuit). The
SMPSO pin (pin 60) provides a switching output waveform to drive the gate of a P channel
MOSFET. The source of the P channel MOSFET should be tied to power ground and the drain of
the MOSFET should be tied to the VN10 through a 100uH inductor. The performance curves
shown in this datasheet as well as the efficiency measurements were done using the internal
VN10 generator. Tripath recommends using the internal VN10 generator to power the TP2350B.
Figure 6 shows how the VN10 generator should be connected.
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Tripath Technology, Inc. - Technical Information
TP2350B
RSW FB 1kΩ
59 SW-FB
CSW FB
0.1uF
VN10
Switchmode
Power Supply
LSW
QP
VNN
RPG 10Ω
100uH
60
SMPSO
VN10
+
DSW
B1100DICT
CSW
0.1uF
CSW
100uF
VNN
Figure 6: VN10 Generator
In some cases, though, a designer may wish to use an external VN10 generator. The
specification for VN10 quiescent current (200mA typical, 250mA maximum) in the Electrical
Characteristics section states the amount of current needed when an external floating supply is
used. If the internal VN10 generator is not used, Tripath recommends shorting SMPSO(pin 60) to
VNN(pin 42) and SW-FB(pin 59) to VNN(pin 42).
One apparent method to generate the VN10 supply voltage is to use a negative IC regulator to
drop PGND down to 10V (relative to VNN). This method will not work since negative regulators
only sink current into the regulator output and will not be capable of sourcing the current required
by VN10. Furthermore, problems can arise since VN10 will not track movements in VNN. The
external VN10 supply must be able to source a maximum of 250mA into the VN10 pin. Thus, a
positive supply must be used and must be referenced to the VNN rail. If the external VN10 supply
does not track fluctuations in the VNN supply or is not able to source current into the VN10 pin, the
TP2350B will not work and can also become permanently damaged.
Figure 7 shows the correct way to power the TK2350:
VPP
V5
VPP
5V
AGND
VN10
PGND
VNN
10V
VNN
F. BEAD
Figure 7: Proper Power Supply Connection
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TK2350, Rev 1.4/04.03
Tripath Technology, Inc. - Technical Information
Output Transistor Selection
The key parameters to consider when selecting what MOSFET to use with the TK2350 are drain-
source breakdown voltage (BVdss), gate charge (Qg), and on-resistance (RDS(ON)).
The BVdss rating of the MOSFET needs to be selected to accommodate the voltage swing
between VSPOS and VSNEG as well as any voltage peaks caused by voltage ringing due to switching
transients. With a ‘good’ circuit board layout, a BVdss that is 50% higher than the VPP and VNN
voltage swing is a reasonable starting point. The BVdss rating should be verified by measuring
the actual voltages experienced by the MOSFET in the final circuit.
Ideally a low Qg (total gate charge) and low RDS(ON) are desired for the best amplifier performance.
Unfortunately, these are conflicting requirements since RDS(ON) is inversely proportional to Qg for a
typical MOSFET. The design trade-off is one of cost versus performance. A lower RDS(ON) means
lower I2RDS(ON) losses but the associated higher Qg translates into higher switching losses (losses
= Qg x 10 x 1.2MHz). A lower RDS(ON) also means a larger silicon die and higher cost. A higher
R
DS(ON) means lower cost and lower switching losses but higher I2RDSON losses.
The following table lists BVdss, Qg and RDS(ON) for MOSFETs that Tripath has used with the
TK2350:
Manufacturer
Manufacturer’s
Part Number
STW34NB20
STP19NB20
IRFB41N15D
IRFB31N20D
FQA34N20
BVdss
Qg
RDS(ON) (Max)
(Ohms)
0.075
(nanoCoulombs)
ST Microelectronics
ST Microelectronics
International Rectifier
International Rectifier
Fairchild
200
200
150
200
200
60
29
67
70
60
0.18
0.045
0.082
0.075
Gate Resistor Selection
The gate resistors, RG, are used to control MOSFET switching rise/fall times and thereby minimize
voltage overshoots. They also dissipate a portion of the power resulting from moving the gate
charge each time the MOSFET is switched. If RG is too small, excessive heat can be generated in
the driver. Large gate resistors lead to slower MOSFET switching, which requires a larger break-
before-make (BBM) delay.
Break-Before-Make (BBM) Timing Control
The half-bridge power MOSFETs require a deadtime between when one transistor is turned off
and the other is turned on (break-before-make) in order to minimize shoot through currents. The
TC2001 has BBM0 and BBM1 that are logic inputs (connected to logic high or pulled down to logic
low) that control the break-before-make timing of the output transistors according to the following
table.
BBM1
BBM0
Delay
120 ns
80 ns
40 ns
0 ns
0
0
1
1
0
1
0
1
Table 1: BBM Delay
The tradeoff involved in making this setting is that as the delay is reduced, distortion levels
improve but shoot-through and power dissipation increase. Both the 40nS and 0nS settings are
NOT recommended due the high level of shoot-thru current that will result. Thus, BBM1 should be
grounded in most applications. All typical curves and performance information was done with
29 of 36
TK2350, Rev 1.4/04.03
Tripath Technology, Inc. - Technical Information
using the 80ns or 120ns BBM setting. The actual amount of BBM required is dependent upon
other component values and circuit board layout, the value selected should be verified in the
actual application circuit/board. It should also be verified under maximum temperature and power
conditions since shoot-through in the output MOSFETs can increase under these conditions,
possibly requiring a higher BBM setting than at room temperature.
Output Filter Design
One advantage of Tripath amplifiers over PWM solutions is the ability to use higher-cutoff-
frequency filters. This means load-dependent peaking/droop in the 20kHz audio band potentially
caused by the filter can be made negligible. This is especially important for applications where the
user may select a 4-Ohm or 8-Ohm speaker. Furthermore, speakers are not purely resistive loads
and the impedance they present changes over frequency and from speaker model to speaker
model.
Tripath recommends designing the filter as a 2nd order, 100kHz LC filter. Tripath has obtained
good results with LF = 11uH and CF = 0.22uF.
The core material of the output filter inductor has an effect on the distortion levels produced by a
TK2350 amplifier. Tripath recommends low-mu type-2 iron powder cores because of their low
loss and high linearity (available from Micrometals, www.micrometals.com). The specific core
used on the RB-TK2350 was a T106-2 wound with 29 turns of 16AWG wire.
Tripath also recommends that an RC damper be used after the LC low-pass filter. No-load
operation of a TK2350 amplifier can create significant peaking in the LC filter, which produces
strong resonant currents that can overheat the output MOSFETs and/or other components. The
RC dampens the peaking and prevents problems. Tripath has obtained good results with RD =
20Ω and CD = 0.22uF.
Bridging the TK2350
The TK2350 can be bridged by returning the signal from VP1 to the input resistor at INV2. OUT1
will then be a gained version of VP1, and OUT2 will be a gained and inverted version of OAOUT1
(see Figure 8). When the two amplifier outputs are bridged, the apparent load impedance seen by
each output is halved, so the current capability of the output MOSFETs, as well their power
dissipation capability, must be accounted for in the design. In addition, the higher peak currents
caused by driving lower impedance loads will cause additional ringing on the outputs. Thus, the
layout and supply decoupling for low impedance (below 8 ohms) bridged applications must be
extremely good to minimize output ringing and to ensure proper amplifier performance.
30 of 36
TK2350, Rev 1.4/04.03
Tripath Technology, Inc. - Technical Information
TC2001
OAOUT1
22
V5
CI
RI
RF
INV1 23
+
-
+
INPUT1
AGND
BIASCAP
V5
20k
20k
+
-
INV2 28
AGND
OAOUT2
27
Figure 8: Input Stage Setup for Bridging
The switching outputs, OUT1 and OUT2, are not synchronized, so a common inductor may not be
used with a bridged TK2350. For this same reason, individual zobel networks must be applied to
each output to load each output and lower the Q of each common mode differential LC filter.
Low-frequency Power Supply Pumping
A potentially troublesome phenomenon in single-ended switching amplifiers is power supply
pumping. This phenomenon is caused by current from the output filter inductor flowing into the
power supply output filter capacitors in the opposite direction as a DC load would drain current
from them. Under certain conditions (usually low-frequency input signals), this current can cause
the supply voltage to “pump” (increase in magnitude) and eventually cause over-voltage/under-
voltage shut down. Moreover, since over/under-voltage are not “latched” shutdowns, the effect
would be an amplifier that oscillates between on and off states. If a DC offset on the order of 0.3V
is allowed to develop on the output of the amplifier (see “DC Offset Adjust”), the supplies can be
boosted to the point where the amplifier’s over-voltage protection triggers.
One solution to the pumping issue it to use large power supply capacitors to absorb the pumped
supply current without significant voltage boost. The low-frequency pole used at the input to the
amplifier determines the value of the capacitor required. This works for AC signals only.
A no-cost solution to the pumping problem uses the fact that music has low frequency information
that is correlated in both channels (it is in phase). This information can be used to eliminate boost
by putting the two channels of a TK2350 amplifier out of phase with each other. This works
because each channel is pumping out of phase with the other, and the net effect is a cancellation
of pumping currents in the power supply. The phase of the audio signals needs to be corrected by
connecting one of the speakers in the opposite polarity as the other channel.
Theoretical Efficiency Of A TK2350 Amplifier
The efficiency, η, of an amplifier is:
η = POUT/PIN
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The power dissipation of a TK2350 amplifier is primarily determined by the on resistance, RON, of
the output transistors used, and the switching losses of these transistors, PSW. For a TK2350
amplifier, PIN (per channel) is approximated by:
PIN = PDRIVER + PSW + POUT ((RS + RON + RCOIL + RL)/RL)2
where: PDRIVER = Power dissipated in the TA3020 = 1.6W/channel
PSW = 2 x (0.01) x Qg (Qg is the gate charge of M, in nano-coulombs)
RCOIL = Resistance of the output filter inductor (typically around 50mΩ)
For a 125W RMS per channel, 8Ω load amplifier using STW34NB20 MOSFETs, and an RS of
50mΩ,
PIN = PDRIVER + PSW + POUT ((RS + RON + RCOIL + RL)/RL)2
= 1.6 + 2 x (0.01) x (95) + 125 x ((0.025 + 0.11 + 0.05 + 8)/8)2 = 1.6 + 1.9 + 130.8
= 134.3W
In the above calculation the RDS (ON) of 0.065Ω was multiplied by a factor of 1.7 to obtain RON in
order to account for some temperature rise of the MOSFETs. (RDS (ON) typically increases by a
factor of 1.7 for a typical MOSFET as temperature increases from 25ºC to 170ºC.)
So,
η = POUT/PIN = 125/134.3 = 93%
Performance Measurements of a TK2350 Amplifier
Tripath amplifiers operate by modulating the input signal with a high-frequency switching pattern.
This signal is sent through a low-pass filter (external to the TK2350) that demodulates it to recover
an amplified version of the audio input. The frequency of the switching pattern is spread spectrum
and typically varies between 200kHz and 1.5MHz, which is well above the 20Hz – 22kHz audio
band. The pattern itself does not alter or distort the audio input signal but it does introduce some
inaudible noise components.
The measurements of certain performance parameters, particularly those that have anything to do
with noise, like THD+N, are significantly affected by the design of the low-pass filter used on the
output of the TK2350 and also the bandwidth setting of the measurement instrument used. Unless
the filter has a very sharp roll-off just past the audio band or the bandwidth of the measurement
instrument ends there, some of the inaudible noise components introduced by the Tripath amplifier
switching pattern will get integrated into the measurement, degrading it.
Tripath amplifiers do not require large multi-pole filters to achieve excellent performance in
listening tests, usually a more critical factor than performance measurements. Though using a
multi-pole filter may remove high-frequency noise and improve THD+N type measurements (when
they are made with wide-bandwidth measuring equipment), these same filters can increase
distortion due to inductor non-linearity. Multi-pole filters require relatively large inductors, and
inductor non-linearity increases with inductor value.
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TK2350, Rev 1.4/04.03
Tripath Technology, Inc. - Technical Information
TC2001 Package Information
28-pin SOIC
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TK2350, Rev 1.4/04.03
Tripath Technology, Inc. - Technical Information
TP2350B Package Information
64-pin LQFP
Please note that the heatslug on the bottom of the package is connected to VNN.
34 of 36
TK2350, Rev 1.4/04.03
Tripath Technology, Inc. - Technical Information
TP2350B Package Information
64-pin LQFP
35 of 36
TK2350, Rev 1.4/04.03
Tripath Technology, Inc. - Technical Information
PRELIMINARY – This product is still in development. Tripath Technology Inc. reserves the right to
make any changes without further notice to improve reliability, function or design.
This data sheet contains the design specifications for a product in development. Specifications may
change in any manner without notice. Tripath and Digital Power Processing are trademarks of Tripath
Technology Inc. Other trademarks referenced in this document are owned by their respective
companies.
Tripath Technology Inc. reserves the right to make changes without further notice to any products
herein to improve reliability, function or design. Tripath does not assume any liability arising out of the
application or use of any product or circuit described herein; neither does it convey any license under
its patent rights, nor the rights of others.
TRIPATH’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE
SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN CONSENT OF THE
PRESIDENT OF TRIPATH TECHNOLOGY INC.
As used herein:
1.
Life support devices or systems are devices or systems which, (a) are intended for surgical
implant into the body, or (b) support or sustain life, and whose failure to perform, when properly used
in accordance with instructions for use provided in this labeling, can be reasonably expected to result
in significant injury to the user.
2.
A critical component is any component of a life support device or system whose failure to
perform can be reasonably expected to cause the failure of the life support device or system, or to
affect its safety or effectiveness.
Contact Information
TRIPATH TECHNOLOGY, INC
2560 Orchard Parkway, San Jose, CA 95131
408.750.3000 - P
408.750.3001 - F
For more Sales Information, please visit us @ www.tripath.com/cont_s.htm
For more Technical Information, please visit us @ www.tripath.com/data.htm
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TP2401-TR
20MHz Bandwidth, Low Noise CMOS OpâampsWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
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3PEAK
TP24010000J0G
Barrier Strip Terminal BlockWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
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AMPHENOL
TP24013000J0G
Barrier Strip Terminal BlockWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
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AMPHENOL
TP24019000J0G
Barrier Strip Terminal BlockWarning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
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AMPHENOL
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